A NEW 246MHz ACTIVE LC BAND-PASS FILTER FOR IF SUB-SAMPLING GSM RECEIVERS

Mostafa A. I. Elmala', Mohamed A. I. Mostafa2, and Sherif H. K. Embabi2

'Electrical Eng. Dept., Texas A&M University. vexas Instruments, Inc., Wireless Communications Business, Unit.

ABSTRACT

A 246MHz bandpass filter is implemented using a new active inductor design. The filter is implemented for a GSM IF sub-sampling receiver, however it can be also used for dual mode GSM-WCDMA receivers. The active inductor is implemented using on-chip capacitance and two negative transconductance differential cells. The filter is implemented as a fourth-order prototype, using narrow band approximation. The BW of the filter is lOMHz, with automatic tuning for Q and center frequency. The design was fabricated in a 0 . 3 5 ~ CMOS process. Input 1dB compression point equals -19dBm. The power consumption is 33mW for 3V supply.

I. INTRODUCTION

Sub-sampling receiver is suitable for 2G and 3G standards, and allows for digital implementation of the IF to base-band mixer. In sub-sampling receivers, the sampling frequency f, is less than the IF frequency. This causes folding of the noise of the variable gain amplifier (VGA) stage to base-band, and subsequently, degrading the signal-to-noise ratio of the receiver [l]. A band-pass filter (BPF) is needed to attenuate the noise of the VGA before the sampling process takes place.

To achieve large attenuation for the noise, we need to use an LC BPF with 246MHz center frequency, which is the IF frequency for the GSM IF sub-sampling receiver shown in Fig. 1. At such frequency, the required values of the inductors will be large. Such values cannot be realized using on-chip spiral inductors because of the large die area and the low Q [2,3]. Active inductors can be used to achieve high Q with small die area [4]. The main disadvantage is the increase in noise figure (NF) proportionally to the inductor Q. The effect of the BPF noise on the total NF is not significant, because it is preceded by a high gain VGA stage.

In this paper, a new CMOS BPF for GSM IF sub- sampling system is described. System specifications require Q > 10 for an IF of 246MHz, and input 1dB compression point greater than -20dBm. The active inductor circuit used in the implementation of the BPF is introduced first. Then, the BPF and its automatic tuning

circuits are discussed, together with some measurement results.

IF BPF Digital

f, fJ4

Fig. 1. IF sub-sampling receiver.

11. ACTIVE INDUCTOR CIRCUIT

In the BPF circuit, we used a new active inductor design instead of on-chip inductors. The circuit implementation of the active inductor is shown in Fig. 2. The circuit consists of capacitor CO and a negative transconductance cell, MI and M2, which controls the quality factor of the active inductor together with the resistor 2R. The resistor 2R is implemented as an active resistance. Capacitance CO consists of on-chip capacitance and the gate-to-source parasitic capacitances of MI and Mz. To increase the equivalent parallel resistance of the active inductor, we used another negative transconductance cell, M3 and M4.

The differential input admittance, Yb, between nodes A and B in Fig. 2, can be derived using the small signal model of the circuit, shown in Fig. 3. This input admittance equals to:

g m ( G + g m o ) + sCo(G - g m o )

G - gm t. sCO Yin ( j ~ ) = 0.5

where g, is the transconductance of MI and Mz, gmo is the

transconductance Of M3 and M4, and G = R - . For the special case, g, = g,,, = G, the admittance is simplified to:

which is purely inductive with a value of:

aa 0-7803-6741 -3/901/$10.00 0 200 1 IEEE

The active inductor is clearly a gyrator implementation using the capacitor CO.

vdd I I I

'in

--LF 'in

7 +-- CO -

iqt

I I

Fig. 2. Active inductor circuit. ' v s s

,ii,

I I Fig. 3. Small signal model.

Cell M3 and M4 is used for several purposes. Firstly, it will increase the equivalent parallel resistance, and hence, increase the inductor Q. Secondly, it is used in tuning the center frequency of the BPF, for which this active inductor is used (inductance value can be varied, without affecting its Q value, by changing gmo). And finally, it improves the linearity of the BPF. The improvement in linearity is due to the fact that gm will be more constant if gate-to-source voltage of MI and Mz is large enough. Cell

M3 and & helps maintaining a large gate-to-source voltage for both MI and Mz.

In general, the equivalent inductance and resistance are frequency dependent and can be calculated from the following equation of the differential input impedance:

(G - g m ) (1 + afiw Zin ( j w ) = -

where a and p are defined as:

2 2 + j w ( a - B)) g m ( G + g m o ) (1 + f i2w2)

CO

G - g m

CO (C - gmo 1

g m (G + gmo )

a=-

B =

The parameters of the active inductor can be calculated from &,,(jo) to be:

g m ( G + g m o ) + ( G - g m ) ( G - gmo 1 QL ( W ) = d o 2 2

g m (G - g m )(G + gmo ) + CO ( G - gmo 1 where L,n(o), Rloss(o), and QL(o) are the inductance, its series resistive loss and its quality factor, respectively. For the special case, g,,= G, we have:

N O QL (U) = - G - g m

In this case, the condition of stability is to have Rl,,(o) > 0, which leads to the constraint g, c G. The quality factor equation clearly indicates that g, can control filter Q.

More accurate estimation for the parameters of the active inductor can be calculated by taking circuit parasitics into account. In this case, we have:

2(G+god-gm) (1+a$ + j4a-b) Zjntiw) =

gm(G+gmO+ god- gou)+godG+ god) (I+ $U2)

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where Cgd consists of the gate to drain parasitic capacitances of MI and M2, and the drain to source parasitic capacitance of the active resistor (2R), gd is the output conductance of the lower current source, and go, is the output conductance of the upper current source. The inductance and its associated resistance can be calculated from the real and imaginary parts of Zi,,(jm). It is important to notice that the active inductor will become unstable if R increases above a certain value, which corresponds to the solution of R when RloS = 0.

111. MPLEMENTATION OF BPF

The BPF is an active LC filter, which uses the active inductor circuit, proposed in the previous section. The prototype, shown in Fig. 4, is a fourth order ladder filter and consists of two LC resonators capacitively coupled by CZ2 in narrow-band approximation [5,6]. The resonance frequency of each section is given by:

- - - - Fig. 4. Fourth-order BPF prototype, using narrow band

approximation.

The values of the capacitors are C1 = lpF, C2 = l.lpF, C1, = 0.15pF, and Cz2 = 0.3pF. The equivalent inductance for both resonators is around 0.3p.H. A differential amplifier, with 8dE3 gain, is used after the filter to achieve a total gain of OdB.

The Q of the filter can be tuned by varying the Q of the inductor, which in turn, can be tuned by changing g,. The center frequency of the filter can be fine-tuned by changing g,. It is important to notice that g, variation has little effect on the Q of the filter. It is also possible to vary the filter Q and its center frequency simultaneously by changing the value of the active resistor (2R). This can be used in a feed-back system to automatically tune the filter response to compensate for process variations. The BPF response is sensitive to the variation in the value of Co. Therefore, an automatic tuning circuit is used to detect the variation in CO.

A PLL based automatic tuning scheme was implemented as shown in Fig. 5 . The VCO was

implemented as an active LC resonator put in a positive feedback loop with a limiter to realize an oscillator with a nominal oscillation frequency equal to the reference frequency. The active inductor used in this implementation has a scaled replica from capacitor CO. The reference frequency is equal to the sampling frequency in a sub-sampling receiver. The output of the VCO is compared to the reference clock using the mixer circuit shown in Fig. 6. The error signal is integrated using a Gm-C filter. The output signal, Vent, is used to control the active resistors in both the active filter and the vco.

4 Vent

U Fig. 5. PLL based automatk tuning system.

1 tv+

Fig. 6. Differential mixer used in the automatic tuning circuit.

IV. MEASUREMENTS RESULTS

The active BPF was fabricaled in a 0 . 3 5 ~ CMOS technology. Measurements were taken for a 3V supply. The measured frequency response is shown in Fig. 7, where the center frequency is 246MHz and fine frequency tuning is obvious when adjusting g,. The minimum measured BW is equal to 7MHz, which corresponds to a maximum Q of 35. Separate Q tuning control signals for the two resonators are needed to achieve larger Q than this value. 'The gain of the filter is OdB for a Q of 27. By adjusting g,, the filter Q can be tuned as shown in Fig. 8.

Group delay measurement was taken in the case of Q = 17 as shown in Fig. 9. The relative group delay in a

90

200KHz signal BW equals 6Ops, whereas it is lOns for a SMHz signal BW (corresponding to 3G wireless receivers).

Fig. 7. BPF frequency response and center frequency tuning measurement.

Fig. 8. BPF Q-tuning measurement.

Input 1dB compression point was measured and equals -19dBm. The measured IIP3 point is -8.5dBm. The corresponding third order inter-modulation measurement is shown in Fig. 10, where 2 tones separated by SoOKHz were used at 246.8MHz and 247.6MHz, respectively. The difference between the signal and the third inter-modulation component is 43.5dB for a corresponding input power of -36dBm.

Fig. 9. BPF group delay measurement.

R0U 30 kHz R F & I t 0 d0 > " I t a 1 i - 1 1 4 3 46 d3 VBU 3 kHz

-10 d0n -ern 0 l i C J : l J C kiiz SUT 300 ns U n l t dE.7

centcr 247.2 nnz 500 k H r i

T i t l e : M I X E R 1 osle: 311.0c~.2000 2 3 : 0 1 : 2 0

Fig. 10. BPF IM3 measurement.

The noise figure of the BPF was measured at the center frequency of 246MHz and is equal to 27dB. Although, the noise figure is relatively high, it is not critical because the BPF is preceded by a VGA with enough gain.

The effect of the BPF on the IF stage noise figure was tested. The total noise figure of the VGA, BPF, and ADC is only 12dB. This clearly shows the effect of the BPF on shaping the VGA noise, and subsequently, reducing the amount of folded noise. The power dissipation of the BPF is 20mW, and that of the tuning circuit is 13mW. The die photo of the filter and tuning circuits is shown in Fig. 11.

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REFERENCES

Fig. 11. Chip micrograph photo.

V. CONCLUSION

A 246MHz active LC BPF was designed and fabricated in a 0.35~ CMOS technology. The design is based on a new active inductor implementation to achieve high filter Q. Separate center frequency and Q-tunings were achieved. The filter is suitable for IF sub-sampling receivers, where it is preceded by a VGA stage, to reduce the folded noise due to sub-sampling. The input 1dB compression point is -19dBm. and the current consumption is l lmA @ 3V. The filter is suitable for dual mode GSM-WCDMA applications.

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[4] W. B. Kuhn, W. Stephenson,, and A. Elshabini-Rid, "A 200 MHz CMOS Q-enhanced LC bandpass filter," IEEE I. Solid-state Circuit:;, vol. 31, pp. 1112-1122, Aug. 1996.

[5 ] A. Zverev, Handbook of Filter Synthesis. New York Wiley, 1967.

[6] H. Khorramabadi and P. Gray, "High-frequency CMOS continuous-time filters," IEEE J. Solid-state Circuits, vol. SC-19, no.6, pp. 939-948, Dec. 1984.

2028-2034, Dec. 1998.

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