Application of FSS Structures to Selectively Control the ...€¦ · Figure 64: Planar scan of...

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ANTENNA SYSTEMS

Application of FSS Structures to Selectively Control the Propagation of signals into and out of buildings Annex 4: In-building propagation enhancement

M Philippakis, C Martel, D Kemp

S Massey

ERA Report 2004-0072 A4 ERA Project 51-CC-12033 FINAL Report Client : Ofcom Client Reference : AY4464 Report edited and checked by: Approved by:

Martin Shelley Project Manager

Robert Pearson Head of Antenna Systems

March 04Ref. Z:\AS_Projects\Custom Antennas and Consultancy_SW\12033_RA_in_and_out_building_FSS\Reporting\FINAL REPORTING\Annex 4 In-building enhancement.doc

ERA Report 2004-0072 Annex 4

2

Crown copyright 2004. Applications for reproduction should be made to HMSO.

This report has been prepared by ERA Technology Limited and its team for the Ofcom under Contract No. AY4464.

DOCUMENT CONTROL

The document may be distributed freely in whole, without alteration, subject to Copyright.

ERA Technology Ltd Cleeve Road Leatherhead Surrey KT22 7SA UK Tel : +44 (0) 1372 367000 Fax: +44 (0) 1372 367099 E-mail: [email protected]

Read more about ERA Technology on our Internet page at: http://www.era.co.uk/


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Contents

Page No.

1. Introduction 7

2. Improving transmission performance through concrete floors and walls using probes 7

2.1 Introduction 7

2.2 Enhancing transmission at 400MHz 7

2.3 Enhancing transmission at 900MHz 11

2.4 Use of sparse probe lattices 16

2.5 Manufacturing issues 19

2.6 Regulatory issues 20

2.7 Summary 24

3. Radio wave propagation under raised floors and in ceiling voids 24

3.1 Introduction 24

3.2 Basic analysis 24

3.3 Inclusion of realistic floor features 30

3.4 Benefits of the floor waveguide 31

3.5 Manufacturing and regulatory issues 34

3.6 Summary 34

4. Mode conversion frequency selective surfaces (MC-FSS) 35

4.1 Introduction 35

4.2 Basic principles of MC–FSS 35

4.3 Theoretical Design 39

4.4 Performance verification 62

4.5 Manufacturing technologies 77

4.6 Regulatory issues 77

4.7 Summary 78

5. References 79


4

Figure list

Page No.

Figure 1: Probe analysed in transmission mode 8 Figure 2: Reinforced concrete slab 9 Figure 3: E-field magnitude through a 30cm concrete slab with and without re-enforcing, 422 MHz,

60° incidence 9 Figure 4: -field magnitude through a 30cm concrete slab with and without re-enforcing, with metal

tray and probes, 422 MHz, 60° incidence 10 Figure 5: Transmission through 30 cm concrete slab under various conditions for 60° incidence 10 Figure 6: Transmission coefficient of concrete with periodic monopole probes 12 Figure 7: Probe with the choke, external and embedded 12 Figure 8: Comparison of transmission coefficient for various dipole configurations 13 Figure 9: Transmission using optimised embedded probed 14 Figure 10: Magnitude of the electric field in a concrete slab using various probes at 760 MHz 14 Figure 11: Transmission loss through 300 mm concrete slab 15 Figure 12: Transmission loss through 300 mm slab using embedded choke probes 15 Figure 13: Circuit model for passive repeater 16 Figure 14: Benefits of using sparse probe geometries 18 Figure 15: Wall cavity probes 19 Figure 16: Floor cavity probes 20 Figure 17: Modelling Geometry for a parallel plate ‘floor waveguide’ 25 Figure 18: Incidence at 0° for f=300 MHz, 400MHz, 500MHz, 800MHz and 900MHz 27 Figure 19: Incidence at 30° for f=300 MHz, 400MHz, 500MHz, 800MHz and 900MHz 28 Figure 20: Incidence at 45° for f=300 MHz, 400MHz, 500MHz, 800MHz and 900MHz 29 Figure 21: Realistic floor waveguide with a ‘tiled’ upper surface structure 30 Figure 22: Transmission loss in a realistic ‘under-floor’ waveguide’ 31 Figure 23: Operation of ‘under-floor’ waveguide in a more realistic building environment at 400

MHz 32 Figure 24: Field distributions along cut of Figure 23 with and without the coupling slots 33 Figure 25: Relative Field distributions along cut of Figure 23 33 Figure 26: Relative Field distribution along cut of Figure 23 34 Figure 27: Principle of operation of the mode conversion frequency selective surface (MC-FSS) 35 Figure 28: Generic geometry of a MC-FSS 36 Figure 29: Relationship between incident angles and reflected angles for various lattice sizes

expressed in wavelengths 37 Figure 30: specular term suppression levels for the Johansson MC-FSS at 9 GHz 38 Figure 31: Magnitude of scattered field in the absence (a) and in the presence (b) of the grid for an

incident angle of 35° (9 GHz) 38 Figure 32: Johansson FSS solved with full MoM and the MoM/PO solution 40 Figure 33: Cross dipole element size and lattice 41 Figure 34: Substrate topology 41


5

Figure 35: FSS performance compared with a metal sheet as reference 42 Figure 36: Full 3 dimensional plane wave response of a metal sheet and the MC-FSS to a plane wave

at 5° incidence in θ 42 Figure 37: Ray tracing model of MC-FSS in a right angle bend corridor 43 Figure 38: Ray tracing model of the right angle bend corridor without the MC-FSS 44 Figure 39: Ray tracing model of the right angle bend corridor using aperture source 44 Figure 40: Geometry of the unit cell. 45 Figure 41: Skew lattice geometry 46 Figure 42: Metallic plate test case 53 Figure 43: Reflected electric field on the computational surface, 30° illumination 54 Figure 44: E-field pattern in the direction cosine space, 30° illumination (metallic plate case) 54 Figure 45: Johansson test case 55 Figure 46: Reflected E-field on the computational surface, 20° illumination (Unmatched Johansson

case) 56 Figure 47: E-field pattern in the direction cosine space, 20° illumination (Unmatched Johansson case)

56 Figure 48: Reflected E-field on the computational surface, 20° illumination (Matched Johansson

case) 57 Figure 49: E-field pattern in the direction cosine space, 20° illumination (Matched Johansson case) 57 Figure 50: Ray tracing model of right angle bend corridor (top view) 58 Figure 51: Electric field inside corridor along vertical line (x=0.5;z=0.066). 58 Figure 52: Electric field inside corridor along horizontal line (Y=0;Z=0.0158m) 59 Figure 53: Field strength enhancement by FSS (crosses) inside the corridor along a longitudinal line

(Y=0, Z=0.0158 m) 59 Figure 54: Scattered field magnitude inside solid metal corridor after the right angle bend (29 GHz)

60 Figure 55: Scattered field magnitude inside the corridor after the right angle bend for MC-FSS case

(29 GHz) 61 Figure 56: Relative improvement of field strength (dB) inside the corridor (29 GHz) 62 Figure 57: Near field scanner measurement set up, (a) planar scanner, (b) FSS excitation 63 Figure 58: Comparison of the FSS grating lobe versus frequency 64 Figure 59: FSS and ground plane measurement at 27 GHz and 29GHz 65 Figure 60: Two samples from the flat panel FSS 65 Figure 61: Setup for VNA measurement (dimensions in mm) 66 Figure 62: Differential S21 of the FSS samples and the reference with the probe at 48 mm and 29mm

from the face of the corridor 67 Figure 63: Comparison between measurement and theory for the corridor measurement 68 Figure 64: Planar scan of corridor using the near field scanner 69 Figure 65: Differential amplitude plot in the x-z plane of the corridor at 27 GHz and 29GHz 70 Figure 66: Regions used to calculate mean field strengths 70 Figure 67: Mean differential amplitude levels inside the corridor for FSS1 mounted flush to the wall

71 Figure 68: Average improvement of field strength inside the corridor (29 GHz) 71


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Figure 69: MC-FSS panel dimensions 72 Figure 70: MC-FSS test panel picture 72 Figure 71: Predicted response of MC-FSS at 870 MHz 73 Figure 72: Measurement setup: (a) inclined metal plane, (b) FSS, (c) no FSS 74 Figure 73: Comparative performance of the MC-FSS 75 Figure 74: Comparative performance of the MC-FSS (at the peak grating lobe angle) 75 Figure 75: MC-FSS measurement setup 76 Figure 76: MC-FSS in the corridor 76 Figure 77: Comparative performance of the MC-FSS in a corridor 77

Table list

Page No.

Table 1: Transmission coefficient in dB of a 30 cm concrete slab at 900 MHz 11 Table 2: Impact of Building Regulations on wall mounted probes 21 Table 3: Impact of Building Regulations on floor mounted probes 23 Table 4: Impact of Building Regulations on MC-FSS 78


7

1. Introduction

This is Annex 4 to the Final Report provided under Ofcom Contract AY4464, Application of FSS Structures to Selectively Control the Propagation of signals into and out of buildings. It gives a detailed description of the work carried out concerned with propagation inside buildings.

Section 2 covers work that has been undertaken to assess the use of probes to improve propagation between floors and through concrete walls. Section 3 covers the work undertaken to assess to use of under-floor and ceiling cavities for propagation of RF signals deep into buildings. Section 4 covers the work which has been undertaken to develop FSS structures which redirect radio signals round bends and down stairwells. Finally, Section 5 provides a list of references.

2. Improving transmission performance through concrete floors and walls using probes

2.1 Introduction

In this task, the feasibility of using periodic probes to improve the transmission of RF energy through ceilings and floors is assessed. A basic floor model consisting of a 30cm layer of concrete is assumed. Reinforcing bars and/or a solid metal under tray are included, as appropriate. The probes assessed are monopoles passing through the floor/ceiling via holes in the concrete. Two modelling methods were used; Finite Element analysis using periodic boundary conditions and an equivalent circuit method.

2.2 Enhancing transmission at 400MHz

2.2.1 Probe geometries

A typical probe geometry is shown in Figure 1. The probe length (Ld) was adjusted for resonance at a frequency within the band of interest around 400 MHz. The coaxial section was air-filled. A 4mm diameter monopole was used, placed in an 8mm diameter hole. The structure in Figure 1 was analysed within a periodic array environment on a 375 x 375mm lattice (0.5λ at 400MHz). From this analysis, it has been found that setting the probe length Ld to 150 mm gives a resonance at 422 MHz. It is noted that this is not a balanced radiator and that currents are induced on the outer metal sheath, causing sub-optimal performance. This structure was used initially to assess the performance due to its simplicity.

Once the probe length had been optimised, the system was analysed to assess the level of coupling taking place between the two sides of a typical floor. The polarisation of the incident plane wave was assumed to be linear and TM. An acute angle of incidence was used (60° from normal) so that the incident plane wave impinged on the probe in a region where its radiation performance is good.


8

Figure 1: Probe analysed in transmission mode

Several different floor cases are investigated:

• Concrete, • Reinforced concrete, • Concrete with probes, • Reinforced concrete with probes, • Concrete on top of a solid metal under tray, with probes, • Reinforced Concrete on top of a solid metal under tray, with probes.

The calculation of the transmission coefficient is based on the ratio of the transmitted and incident power, ie:

( )( )( )( )

*

*

ˆRe( ) 10 log

ˆReout

in

t tS

i iS

E H n dsTrloss dB

E H n ds

× ⋅ = × ⋅

∫

∫

r r

r r

where inS and outS are the integration surfaces on the reflection and transmission planes respectively.

The subscripts t and i denote the total and incident fields respectively. E and H denote the electric and magnetic fields and n is the normal vector of the corresponding integration surface.

The geometry for the reinforcement bars, where used, is shown in Figure 2. The mesh was mounted centrally and had a lattice size of 187.5mm x 187.5mm. Where used, the probes passed through the centre of every other segment. The assumed electrical properties of the concrete were εr=7, tanδ=0.1.

Concrete Concrete

Port

dipole

metalmetal

airImpedance calculation

Ld

30 cm


9

Figure 2: Reinforced concrete slab

2.2.2 Results

Figure 3 shows the magnitude of the electric field across a concrete slab with and without reinforcement.

Incident plane wave

Concrete

Air

Air

Incident plane wave

Reinforcement

Air

Air

Concrete

Concrete

Figure 3: E-field magnitude through a 30cm concrete slab with and without re-enforcing, 422 MHz, 60° incidence

Figure 4 shows the equivalent plots fro concrete with a solid metal tray, using probes to couple through the metal tray. High field levels can be observed in the regions near to the probe.


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Incident plane wave

Metal plate

Probe

Air

Air

Concrete

Incident plane wave

Reinforcement

Air

Air

Concrete

Concrete

Probe

Metal plate

Figure 4: -field magnitude through a 30cm concrete slab with and without re-enforcing, with metal tray and probes, 422 MHz, 60° incidence

The use of probes in a concrete slab without the metal tray was also investigated, but the E-field plots are not shown. The transmission performance of all these cases is summarised in Figure 5. These results indicate that, by introducing probes, it is possible to achieve losses equivalent to those for concrete only. The introduction of re-enforcing bars degraded performance, by only to a minor extent.

-30.00

-25.00

-20.00

-15.00

-10.00

-5.00

0.00

240 260 280 300 320 340 360 380 400 420 440

Frequency (MHz)

Tran

smis

sion

loss

(dB

)

Concrete only

Concrete + Probes +metalConcrete + re-bars

Concrete + rebars +probes + metalConcrete + probes (nometal)Concrete + rebars +probes (no metal)

Figure 5: Transmission through 30 cm concrete slab under various conditions for 60° incidence


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2.3 Enhancing transmission at 900MHz

2.3.1 Preliminary results

Without re-optimising the probe geometry, or changing the lattice, further analysis was undertaken in the 700-900MHz band to assess the potential benefits of the technique at these higher frequencies.

Table 1 presents the values for transmission through reinforced and non-reinforced concrete with no probes and including probes and a metal under tray. The presence of the under tray clearly prevents all transmission. The use of comparatively sparse unmatched probes results degraded transmission of 7.8 dB and 6.4 dB for the non-reinforced and reinforced cases respectively.

Table 1: Transmission coefficient in dB of a 30 cm concrete slab at 900 MHz

Without reinforcement With reinforcement Concrete -7.2 -8.6 Metallised concrete with probes and undertray -15.0 -15.0

2.3.2 Re-optimised probes

The probe geometry was subsequently re-optimised to operate at 800 MHz and then analysed over the band running from 700 MHz to 900 MHz. The periodicity of the lattice was also changed to 187.5 mm (half-wavelength at 800 MHz), matching the pitch of the reinforcing bars. Figure 6 shows the transmission loss obtained for various probe lengths and hole diameters. The optimum probe was found to have a length of 100 mm and used a hole diameter of 16 mm. For this case, the transmission loss was around -4 dB from 760 MHz to 780 MHz and better than the concrete slab from 720 MHz to 810 MHz.

2.3.3 Periodic choked dipole probes

In order to improve further the transmission, the use of a balanced probe with a choke was investigated. Figure 7 shows the geometry used. A choke section about λ/4 at the operating wavelength was required to match the dipole and maximise the RF coupling. It was found that the dipole needed thicker cross-section above the choke for optimum performance; a diameter of 12 mm is used, compared to 4 mm for the unmatched probe. The length of the dipole, the length of the choke and the hole diameter were optimised for maximum transmission. The optimum configuration used a probe length of 150 mm, a choke length of 85 mm and a hole diameter of 16 mm. The transmission loss of a concrete slab with a lattice of choked dipole probes is shown in Figure 8. Higher levels of transmission are achieved with this configuration, peaking at –1.9 dB at 760 MHz. An analysis of the case where the choke section was embedded in the concrete was also undertaken, since this resulted in a significantly smaller protrusion above the concrete. This configuration is also shown in Figure 7. Results for this case are included in Figure 8, and it can be seen that there is very little difference


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between the performance with the probe fully exposed, or embedded. In general, the peak transmission is some 2dB higher than for the case using the unmatched probe.

-25

-20

-15

-10

-5

0

700 720 740 760 780 800 820 840 860 880 900

Frequency (MHz)

Tran

smis

sion

(dB

)

Concrete slab

Probe length = 75 mm, hole diameter = 8 mm

Probe length = 90 mm, hole diameter = 8 mm

Probe length =100 mm, hole diameter = 16 mm

Figure 6: Transmission coefficient of concrete with periodic monopole probes

Choke length

Dipolelength

Concrete Concrete

Choke

Holediameter

100 mm

12 mm

4 mm

4 mm

85

150

Concrete Concrete

Choke

16

12

4

4

air air

30

Figure 7: Probe with the choke, external and embedded


13

-25

-20

-15

-10

-5

0

700 720 740 760 780 800 820 840 860 880 900

Frequency (MHz)

Tran

smis

sion

(dB

)

concreteno chokenon embedded chokeembedded choke

Figure 8: Comparison of transmission coefficient for various dipole configurations

After further probe optimisation, resulting in an increased probe length of 190mm, a comparison was made between the use and absence of probes for un-strengthened concrete, reinforced concrete and probes in reinforced concrete with and without a metal under tray. These results are shown in Figure 9 below. It should be noted that the probes were optimised in the presence of the metal under tray. The poorer performance that is predicted when the metal tray is not present are a result of higher mis-match losses and of additional absorption in the concrete. It can be seen that the probes provide significant improvements in transmission in all cases.

The magnitude of the electric field across the concrete is shown in Figure 10 at 760 MHz for all four cases. From these field plots, it can be seen that the choke suppresses the reactive fields near the outer coax element in the concrete region, ensuring maximum propagation of energy through the coax section.

The effect of the angle of incidence was investigated for the optimum embedded choke case (dipole length of 190 mm). For comparison purposes, the transmission coefficient through a 300 mm concrete slab was calculated for various angles of incidence running from 0 to 80° without probes (Figure 11). Optimum transmission is seen for an incidence angle of 70° though it degrades rapidly after 70°. Figure 12 shows the transmission loss through the concrete with embedded choke dipoles for various angles of incidence. The transmission is very poor for low incidence angles, due to the inherent radiation characteristics of the dipole probe. However, the transmission for incident angles varying from 50 to 70° is greatly improved as compared to the concrete slab.


14

-25

-20

-15

-10

-5

0

700 720 740 760 780 800 820 840 860 880 900

Frequency (MHz)

Tran

smis

sion

(dB

)

concrete

Reinforced concrete

opt. Embed. Choke - With under-tray

opt. Embed. Choke - Without under-tray

Figure 9: Transmission using optimised embedded probed

Monopole probe Choked dipole probe Embedded choked dipole probes

Loaded, embedded choked dipole probes

Figure 10: Magnitude of the electric field in a concrete slab using various probes at 760 MHz


15

-10

-9

-8

-7

-6

-5

-4

-3

-2

-1

0

0.700 0.720 0.740 0.760 0.780 0.800 0.820 0.840 0.860 0.880 0.900

Frequency (GHz)

Tran

smis

sion

(dB

)

80 degrees70 degrees60 degrees50 degrees40 degrees30 degrees20 degrees10 degrees0 degree

Figure 11: Transmission loss through 300 mm concrete slab

-50

-45

-40

-35

-30

-25

-20

-15

-10

-5

0

700 720 740 760 780 800 820 840 860 880 900

Frequency (MHz)

Tran

smis

sion

(dB

)

80 degrees 70 degrees 60 degrees50 degrees 40 degrees 30 degrees20 degrees 10 degrees 0 degree

Figure 12: Transmission loss through 300 mm slab using embedded choke probes


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2.4 Use of sparse probe lattices

In many cases, the techniques described in the preceding sections will be very expensive to implement, since a large number of probes will be required (see Section 2.5 below). Hence, it is valuable to trade-off transmission performance against lattice geometry, considering sparse and non-periodic lattices. Simple analytical expressions, based on the first principles of wave reception and re-transmission, were used to undertake this trade-off.

The basic coupling mechanism is again based on the use of a double protruding probe (Figure 13). Assuming that a plane wave illumination exists in one side of the wall, the Norton equivalent circuit model shown in Figure 13 can be postulated. The basic assumption is that the Tx & Rx sides have similar resonating structures and hence a conjugate match is automatically guaranteed.

If a plane wave with amplitude Epw is assumed, the ‘open circuit’ voltage Voc at the terminal of the Rx part is given by the following equation.

( , )E

RX RX RX aoc pw

o

G ZVZ

q fl

p=

where,

GRX : is the power gain of the Rx part at the incident direction θRx, φRx, Za : the input impedance of Rx & Tx part, Zo : the impedance of free space (=377Ω), λ : is the wavelength.

Figure 13: Circuit model for passive repeater

Using the equivalent circuit of Figure 13, it is straightforward to calculate that the power available for retransmission in the lower part, PTx:

Rx

Tx

Za

Za

Voc

Plane wave E pw


17

2

4oc

T Xa

VPZ

=

It is assumed that this power will be retransmitted in the lower space in a Friis-type mode, ie on the basis that the contributions from each individual probe add on a power basis. This obviously neglects coherency effects such as constructive/destructive interference. However, in the arbitrary situation of radio wave propagation within a ‘cluttered’ radio environment, such as the inside of a building, this assumption is considered a fair representation of the average level. More sophisticated radio propagation models can be invoked; however, the associated complexity may obscure the basic conclusions from the assessment. Under this condition, the total radiation intensity J (W/m2) is simply given by:

2

( , )4

T X TX T X TXN P GJ

rq f

p=

where GTX is the power gain of the Tx part at the direction θTx, φTx. Here, a single characteristic distance, r, from the repeater array ‘centre of gravity’ to the observation point, has replaced the individual 1/r2 dependencies associated with each probe. In order to assess the benefit due to an array of N passive repeater probes, an “Advantage Factor” Q was defined which is the ratio of the field at the observation point with and within the probes present.

Assuming a transmission factor T characterising the insertion loss due to walls, the value of the direct illuminating field is simply defined as:

E Epw pw

in T=

Using straightforward manipulation of the above equations, the Advantage Factor Q can be defined as:

( , ) ( , )4E

pw

T XRX RX RX T X T X T Xin

EQ N G GrTl q f q f

p= =

If the probes on the upper surface are arranged in a regular lattice with an inter-element spacing δ then:

N = (S/δ)2

where, S is the dimension of a (square) room.


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While it is not necessary to use a regular probe lattice, if this is done, Q is a function of δ/λ . Where a non-regular spacing is used, δ can relate to an “average” inter-element spacing.

An observation point rA was defined as a special case, where h is the height of the room:

( )4 4 2AS S h= -r

This position in an average room may well represent a practical location of a wireless device. In a typical room the dimensions S, h may have values 6m and 2.5m respectively. If probes similar to those analysed above are used, a typical gain value ~1dBi can be assumed.

A family of curves was created, shown in Figure 14, which describe the benefits of using sparse probes for a number of wall losses and values for δ. Although a number of assumptions and specific numerical values are involved, this figure can be used as a simple 1st order design aid when considering the deployment of a sparce array of probes.

Passive Repeater Array

-40.00

-30.00

-20.00

-10.00

0.00

10.00

20.00

30.00

40.00

0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1

λ/d

Adv

anta

ge (

dB) T=-5 dB

T=-10 dBT=-15 dBT=-20 dBT=-25 dBT=-30 dBT=-35 dBT=-40 dB

Figure 14: Benefits of using sparse probe geometries

The most significant observations are summarised as follows:

• To get maximum benefit, very dense arrays must be used, • Sparse arrays are more beneficial in cases where large wall losses are present,


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• There is always a “break-even” probe density when transmission is equal to that of the basic wall. This depends on wall losses.

As an example, the line in Figure 14 represents the use of a single probe in the 6m x 6m room, operating at 400 MHz. It can be seen that this probe will provide an improvement in signal strength inside the room only if the wall attenuation is greater than about 28dB.

2.5 Manufacturing issues

2.5.1 Wall installation

The structure considered consists of a series of metal rods passing through a cavity wall and protruding about 100mm from the external and internal faces, see Figure 15.

Figure 15: Wall cavity probes

Within regulated spaces, the thermal and moisture bridging would need to be controlled by placing an additional insulation layer and containing structure on the inside. This would have to be in place to act as a protective element. At ground and lower floors the installation would require encasing on both sides to protect the public and deter people from climbing up the outside of the building. The probes would need to be installed at a slight angle to ensure any moisture which forms on the probes will travel towards the outer skin.

Due care must be taken to ensure that there is no loss of wall thickness due to drills punching through. Costs are likely to be of the order of £7/hole plus any fixed charges for access and power supply.

Inside Outside


20

The rods will be best installed after any settlement or differential movement between the skins has been finished. This will maintain the downward slope of the probes.

2.5.2 Floor / ceiling installation

The various configurations considered are shown in Figure 16 below.

Figure 16: Floor cavity probes

The probes could be installed after the concrete floor slabs have been cast, or holes for the probes could be cast into the slab. Both operations have considerable installation overheads in terms of time and cost. It is estimated that casting the tubes into the slabs would cost about £5/rod while drilling a pre-cast slab would cost £35/rod. In addition to this, the costs of manufacturing the tubes will be significant. Thus, the cost of using this system across any substantial floor area may be prohibitive

2.6 Regulatory issues

2.6.1 Wall installation

Table 2 below provides an initial assessment of the impact that wall-mounted probes will have on building regulations.


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Table 2: Impact of Building Regulations on wall mounted probes

Regulation Pass/Fail Comment

A Pass Best solution using straight through rods, viewed as a similar to a remedial wall tie replacement (see note 1)

B Special measures required to achieve compliance

Pass where precautions are taken to limit the conduction of heat and obstruction (see note 2)

C Special measures required to achieve compliance

Pass by ensuring drainage path E.g. use of inclined rods (see note 3)

D N/A

E Special measures required to achieve compliance

Pass by using suitable acoustic screening (see note 4)

F Special measures required to achieve compliance

Pass with suitable moisture traps on inside leaf. (see note 5)

G - K N/A

L Special measures required to achieve compliance

Pass by suitably limiting the heat loss through the fabric of the building (see note 6)

M Pass Depends how ‘a material alteration’ is defined with regard to restricting access widths.

N N/A

Workmanship and buildability

N/A Retro-fitting seems the most likely option.

1. Whilst the use of staggered probes to avoid bridging the cavity and creating moisture or thermal bridges is a reasonable approach it would probably create more problems than it solves. With new build there is little chance of the mortar joints on the inner and outer leaf lining up sufficiently to ensure that a regular grid is maintained and this is further compounded if some form of rigid insulation batt is used. In these instances the probe spacing would then be constrained to the ‘unit’ dimensions of the blockwork (250 or 450 mm), batts, and wall tie regulation of 2 ½ ties per metre, nominally 900 mm horizontal and 450 mm vertical spacing. Although the probes could substitute as wall ties, either as a square twisted rod or a flat profile with rippled strips, doing so would mean that the design has to be subject to EN testing procedures in order to comply as a wall tie. For remedial work where additional ties have to be inserted in the walls it is common practice to drill through both the inner and outer leaf and epoxy a stainless steel rod into the wall. Thus, extending the rod to protrude either side of the wall should not create any difficulties. This may create a thermal bridge and hazard on the internal face; these could simply be countered by encasing the protruding rods in a suitable false wall with additional insulating material (e.g. mineral fibre).


22

2. Provision would be needed to limit the spread of flame to any linings or rate of heat release so as not to propagate the fire. In the event of a fire there is the possibility that a substantial amount of heat could be transmitted via the rods. Such ‘hot spots’ would undoubtedly fail fire testing. This could be mitigated by ensuring the exposed ends are suitably encased and that the cavity fill material is not likely to ignite (i.e. something other than polystyrene). A further impact of this regulation would mitigate against using the rods as wall ties since excessive heating would cause loss of bond between the brick work and the mortar and disrupt the stability of the building. Normal wall ties have a reasonable degree of shielding from the effects of fire. In order to provide fire fighters with reasonable facilities there may be areas of the building where the use of the probes would be seen as an obstruction.

3. The walls, floors and roof of the building shall adequately resist the passage of moisture to the inside of the building. It may be necessary to incline the probes so that they slope down towards the outer leaf. Thus, any moisture will drain down the cavity. If larger rods of greater than 4mm diameter are required then it may be necessary to design a specific profile to promote drainage. Moist patches around the probe will promote a drop in the U value and there may be some condensation staining to finishes.

4. Screening is required for safety and fire; this will also mitigate against the spread of sound.

5. Screening required for safety and fire; this will also mitigate against thermal bridging into the room. Some provision would be required to deal with condensation on the cold metal work.

6. This can be achieved by suitably limiting the heat loss through the fabric of the building as per notes 2, 4, and 5

2.6.2 Floor / ceiling installation

Table 3 below provides an initial assessment of the impact that wall-mounted probes will have on building regulations.


23

Table 3: Impact of Building Regulations on floor mounted probes


A Special measures required to achieve compliance

Requires due control of the shear forces (see note 1)


Pass where precautions are taken to limit the conduction of heat and obstruction (see note 2)

C Special measures required to achieve compliance

Pass by ensuring adequate sealing and thermal insulation (see note 3)

D N/A

E Special measures required to achieve compliance

Pass by using suitable acoustic screening (see note 4)

F N/A

G N/A

H N/A

J N/A

K N/A

L Pass Requires insulation in certain cases (see note 5)

M N/A

N N/A


Special measures required to achieve compliance

High time and cost factors

1. Assuming that the steel reinforcement is not broken, the major issue is the generation of shear forces around critical components of the building. e.g. columns and the margins of the floor slabs. With the proviso that rods are placed only in appropriate areas then there should not be any further issues.

2. In the event of a fire there is the possibility that a substantial amount of heat could be transmitted via the rods. Such ‘hot spots’ would undoubtedly fail fire testing. This could be mitigated by ensuring the exposed ends are suitably encased. In order to provide fire fighters with reasonable facilities, there may be areas of the building where the use of the probes would be seen as an obstruction.

3. The rods and sleeves would need to be adequately sealed to stop the movement of moisture. Where the slab divides an unregulated space from a regulated space insulation will have to be provided on the regulated side.

4. Pass on the assumption that the suspended floors and ceiling provide sufficient sound insulation.


24

5. Where the slab divides an unregulated space from a regulated space insulation will have to be provided on the regulated side.

2.7 Summary

The insertion of probes into high loss wall structures such as re-enforced concrete can result in significant improvements in signal transmission. However, in order to offer major improvements, dense probe lattices must be used, and the cost of implementation may be high.

3. Radio wave propagation under raised floors and in ceiling voids

3.1 Introduction

Most modern buildings are constructed with a raised floor and/or a suspended ceiling. A regular lattice of metal posts supports the floor or ceiling at a distance of typically 200mm from the solid concrete slab. The floor itself takes usually takes the form of a series of 600 x 600mm tile blocks. These are often wood for the floor and plasterboard for the ceiling and may be metal backed for thermal control. The space between the tile blocks and the reinforced concrete slabs is used to run building services such as electric and IT cables, water pipes and ventilation systems.

In this section, the potential to use the cavities formed by these suspended floors and ceilings to propagate RF signals in the natural “parallel plate waveguide” structure created is assessed. The prime reason for investigating the efficiency of this propagation mechanism is to enhance radio wave energy transfer to rooms or spaces that cannot receive direct illumination from external wireless sources. A further benefit may be to reduce the shadowing effect of a building by allowing useful propagation through it.

3.2 Basic analysis

The propagation in a parallel plate environment can be simply described by well-known analytical modal expressions. However, the existence of posts complicates the situation. The current investigation is based on a theoretical evaluation using electromagnetic modelling tools including Finite Elements (Ansoft High Frequency Structural Simulator, HFSS) and Finite Difference Time Domain (Remcom XFDTD).

The initial study assessed the effects of an array of metal tile mounting posts in a metallic parallel plate waveguide, as shown in Figure 17. The ‘floor’ post array occupied a space of 6 x 6 m and the posts were placed on a 600 x 600 mm lattice. The diameter of the individual posts was assumed to be 40mm, while the parallel plate separation distance was 200mm. It was assumed that all materials were loss-less.

In the frequency band of interest in this instance (400-900 MHz), it can easily be seen that, for parallel polarisation incidence the waveguide is cut off for most frequencies in the band (<750 MHz). Even


25

for those above cut off (750-900 MHz), the waveguide does not propagate energy efficiently. However, vertically polarised energy (Figure 17) is always above cut off, hence its propagation in the parallel plate environment remains efficient. Hence, this guiding mechanism is polarisation selective. For this reason, results associated only with vertical polarisation incidence are presented here.

Figure 18 shows results of radio wave propagation for angle of incidence 0°, 30° and 45°. The quantity plotted is Relative Electromagnetic (EM) Energy distribution. This quality is the EM density inside the parallel plate waveguide referred to the same quantity when the same plane wave propagates in free space. If we assume a plane wave with Electric field magnitude of 1 V/m, the quantity plotted is:

2 212 2

oem

oW E Hm

e= +

ur ur

where E, H are the electric and magnetic fields considered. The values of εo and µo are the constituent parameters of the propagating medium, ie air.

Figure 17: Modelling Geometry for a parallel plate ‘floor waveguide’

It is evident that the propagation loss inside the structure is both frequency and angle of incidence dependent. It is worth noting the low transmission loss for frequencies in the TETRA and GSM

Perfect Electric

Conductor (PEC)

Plane Wave

E

0.04m dia Posts

Incidence Angle

PLAN VIEW


26

bands. For normal incidence on the parallel plate structure the loss is around 0.5dB/m. For other incidence angles, the losses increase and there are regions where the loss can increase to 1.5dB/m.

In common with every periodic scattering structure, the floor mounting post array has cut-off frequencies, and stop- and pass-bands. The characteristic values for this frequency selective propagation are angle dependent. This is because the periodicity of the ‘array elements’ along the given propagation line is different according to which angle of incidence is used.

In general, the floor suspension structure seems an efficient waveguide structure. However, coupling of external EM energy into this structure is not trivial and one of the other technologies or techniques described in this report, for instance, probe coupling, may be required to get sufficient energy into the structure in the first place. Even excluding the problem of energy loss due to the interaction of radio waves and the external features of the building (walls, windows etc), the energy available to the post guiding structure is only a fraction of the energy impinging on the building. Assuming a room with height of around 2.5m, the energy available to the floor guiding structure (height 0.2m) is, in principle, only around 10% (-10 dB) of the energy available to the room above (though, in principal, the capture area could be artificially increased using external antennas). However, when propagation inside the room is impaired significantly, or when coupling into rooms not directly illuminated from outside, the energy content in the floor guidance structure may be higher than the value of radio power available above. In these cases, extraction of energy from this guidance structure can improve radio coverage inside the building.

It is worth noting that, at higher frequencies, wall absorption is greater, while propagation in the cavities may actually improve. Hence, these concepts may become more valuable for cellular (GSM, UMTS) systems.


27

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3.3 Inclusion of realistic floor features

The analysis above assumed a post array scattering structure sandwiched between two uniform perfect metallic planes. Here, the upper part of the waveguide is modelled as an array of flat metallic tiles with a gap between each representing non-metallic tile supports, as shown in Figure 21.

3.6 m

3.0 m0.6 m

0.6 m

0.2 m

0.3 m

g

0.03 m

0.04 m

Figure 21: Realistic floor waveguide with a ‘tiled’ upper surface structure


31

Two gap sizes have been analysed, g=10 and 20 mm. The tiles themselves were considered to have a finite thickness of 0.03m.

The parallel plate region was analysed as a true waveguide structure with discrete input and output ports (Figure 21(a)). It was assumed that the ports were excited with a TEM mode. Figure 22 shows the predicted transmission loss for the 3 x 3m structure shown in Figure 21. The good propagating properties previously predicted for 400, 800 and 900 MHz are verified, though the performance between 800 and 900MHz is somewhat unstable. Significant stop bands around 550 and 650 MHz are predicted.

-20

-18

-16

-14

-12

-10

-8

-6

-4

-2

0

300 400 500 600 700 800 900Frequency (MHz)

Loss

(dB

)

No Gap10mm Gap20 mm gap

Figure 22: Transmission loss in a realistic ‘under-floor’ waveguide’

3.4 Benefits of the floor waveguide

The potential benefits of using the ‘under floor’ waveguide structure were assessed by creating a model for a ‘test’ building, as shown in Figure 23. As the model was physically large, and hence also electrically large, it was only practical to consider analysis at 400 MHz.

The case analysed was based on the following assumptions:

• A 3-room building • The test building had only walls on one side for reasons of computational resource savings • Neither the wall structure nor the post structure was matched to free space • The ‘under-floor’ guidance structure was open and hence the illumination had direct access to

the structure. • Illumination with a plane wave along a normal direction of incidence (Figure 23(a)). • Vertical polarisation incidence with electric field strength of 1V/m


32

• Walls made of concrete with dielectric properties εr=9 and tanδ=0.1 • Gap between the tiles with size g=0.1 m • Network of posts with cell size 0.6 x 0.6 m

Figure 23: Operation of ‘under-floor’ waveguide in a more realistic building environment at 400 MHz

The field distribution in the room furthest from the entry point of the direct radio wave illumination was assessed with and without the ‘under floor’ guide. In this way, it was possible to assess how leakage from the waveguide can improve coverage. Four inclined slots with a resonant length of around λ/2 at 400 MHz were used as the prime means for extracting energy from the ‘under floor’ waveguide. The coupling slot geometry is clearly seen in Figure 23.

Incident Wave

Field Cut

1.5 0.25 m

1.75 m

3.3 m

0.75 m

0.6 m

Field Cut

Wall Wall

Wall

Coupling Slot

6m


33

Figure 24 presents the individual field distributions along a vertical cut (Figure 23(b)) with and without the coupling slots. The high field intensity inside the guide system is evident. The leakage mechanism from the slot is also captured. Figure 25 combines both results to show the difference in the coverage made by the introduction of the coupling slots. It is clear from this comparison that significantly more RF energy can be made available to the space considered using under floor propagation. The same picture in the plan view cut (Figure 23(a)) can be seen in Figure 26. Again, a noticeable increase in the field strength is observed. This increase at points can be as high as 10 dB above the value of ordinary coverage.

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Relative Field distribution inside test building at f=400 MHz(Reference values the No coupling slot case)

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34

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Figure 26: Relative Field distribution along cut of Figure 23

3.5 Manufacturing and regulatory issues

There is unlikely to be any restriction in sending the signals through the voids with regard to the building regulations. Restrictions are more likely to be caused by practical issues arising from conflict with other services within the voids.

If large openings are required to couple energy into the voids, these will need to be incorporated into the design of the building. Inclusion of openings of this typical size through the skin of the building for other services is fairly routine and thus suitable designs are possible which do not compromise building regulations. Care would be required with regard to thermal bridging, moisture, and differential movement. Retrofitting openings on existing buildings could well compromise the integrity of the building and may not be possible.

Using dedicated waveguides to propagate energy through the void space would be another possibility and these could readily be designed to current standards for other similar services such as heating and ventilating duct work.

3.6 Summary

The under-floor (or above ceiling) area has the potential to transfer energy to parts of building which will normally be in a deep shadow situation. The initial theoretical investigation conducted here has demonstrated that the post guide can provide low loss RF power transfer. Although its performance has both frequency and angle of incidence dependence, characteristic dimensions found in modern buildings favour performance in the bands of interest.


35

4. Mode conversion frequency selective surfaces (MC-FSS)

4.1 Introduction

In this section, a novel structure which has the potential to enhance propagation down meandering corridors and stairwells is investigated. First, the principals of operation are described. Next, the implementation of a software model for the characterising the impact of the structure on an “ideal” right angle corridor is described. The results are then compared with practical measurements untaken on a test sample and in a full scale corridor demonstration.

4.2 Basic principles of MC–FSS

The MC-FSS is a flat structure which reflects EM energy in a direction other than its specular reflection angle, as shown in Figure 27. Incoming energy impinging on a flat metal sheet at an angle θ to the normal is reflected at an angle of -θ. Energy incident on the MC-FSS at an angle θ is reflected at a different angle. Non-specular reflections are achieved using a periodic structure which transfers the reflected energy from the lowest order mode to a higher order mode. The energy is then reflected in the direction inherent to the higher order mode. The behaviour is analogous to an antenna consisting of an array of radiating elements where the separation is greater than a wavelength; such an array has multiple lobes, including “grating lobes”. The function of the MC-FSS is to suppress energy in the main lobe and to radiate it in the direction of a “grating lobe”. This structure was first described by Johansson [Ref 1].

MC-FSS

Incoming wave

Reflected wave(conventional direction)

Reflected wave(unconventional direction)

Normal

Figure 27: Principle of operation of the mode conversion frequency selective surface (MC-FSS)

The generic structure capable of exhibiting such characteristics consists of one or more conductive surfaces containing periodic metal elements, separated from a ground plane by multiple dielectric layers. Figure 28 shows a generic MC-FSS structure, which is made up of 1 periodic surface and 3


36

dielectric layers. Ideally, the thickness of each dielectric layer can be optimised in order to achieve phase cancellation of the reflected field in the specular direction (the conventional direction). The grid is characterised by the shape and periodicity of the conductive elements used on it.

Dielectric layer 3

Dielectric layer 2

Dielectric layer 1 1 1,rε σ

2 2,rε σ

3 3,rε σ

Normal

Grid

Metal plate

Figure 28: Generic geometry of a MC-FSS

The reflection direction of higher order modes is a function of the following parameters:

• Separation between lattice elements, Sp • The wavelength of operation • The incident angle direction ( , )θ φ

Assume that an MC-FSS sheet is disposed in the xy plane. It can be shown that if the plane of incidence lies on the xz plane (i.e.: 0φ = ), the reflected angle of the first higher order mode will

occur in the same plane at an angle 1,0θ− satisfying the condition:

1,0sin sinpS

λθ θ− = −

Figure 29 shows this relationship for a series of pSλ

values. It can be seen that the first higher order

mode propagates for any angle of incidence when Sp is greater or equal to a wavelength. For cases where Sp is smaller than the wavelength, the first higher order mode propagates for only a range of incident angles. For instance, if the lattice size is 0.83λ then the first higher mode propagates when the incident angle is greater than 12°.


37

-90

-60

-30

0

30

60

90

0.0 2.5 5.0 7.5 10.0 12.5 15.0 17.5 20.0 22.5 25.0 27.5 30.0

Incident angle (degrees)

Ref

lect

ed a

ngle

(deg

rees

)

0.71 0.83 1.00 1.25 1.67 2.50 5.00

-0.17dB -0.15dB-0.32dB-0.55dB

-0.53dB

-1.28dB-5.87dB-4.48dB

-0.002dB -0.04dB

-0.03dB

-0.02dB

Figure 29: Relationship between incident angles and reflected angles for various lattice sizes expressed in wavelengths

Moreover, it can be seen that wide-angle reflections can be achieved from small incident angles. For instance, a reflected angle of 50° can be obtained from a 5° incident angle, based on a lattice size of 1.2λ.

The amount of energy reflected by the higher order mode depends primarily on the dimensions of the FSS elements. The reflection level of the Johansson MC-FSS, which is based on dipole elements, has been calculated for several incident angles. The reflection level expressed in dB is shown for a number of specific cases in Figure 29. Maximum reflectivity is observed (-0.002 dB) when the lattice size is about 1.2λ and the incident angle is 7.5°.

The reflectivity of the higher order modes can also be assessed by investigating the suppression of the specular reflection achieved by the MC-FSS in accordance with the conservation of energy principle. The suppression level is quantified as the field difference between the case where the grid is absent and the case where the grid is present. Figure 30 shows the level of suppression achieved by the Johansson MC-FSS. Maximum specular term suppression of –34 dB is seen for 35° incidence.

The scattered field magnitude caused by the Johansson MC-FSS is shown in Figure 31. The direction of reflection is clearly changed by the presence of the scattering grid.


38

-42

-36

-30

-24

-18

-12

-6

0

15 20 25 30 35 40 45

Theta incident (deg)

Supp

ress

ion

(dB

)

Figure 30: specular term suppression levels for the Johansson MC-FSS at 9 GHz

(a)

(b)

Figure 31: Magnitude of scattered field in the absence (a) and in the presence (b) of the grid for an incident angle of 35° (9 GHz)


39

4.3 Theoretical Design

Numerical modelling of FSS structures is normally done by analysing infinite periodic structures. In practice, an FSS will not be infinite and these conventional techniques do not incorporate truncation effects. The reason that finite structures are not normally modelled is because of the large computational time and memory requirements. Modelling of infinite FSS structures negates this problem by considering only one FSS element and its lattice. By applying the Floquet mode theorem to the single element, the response for an infinite number of elements is obtained.

A novel method has been developed to analyse finite FSS containing a ground plane. The method is based on the integration of two numerical techniques, the Methods of Moments (MoM) and Physical Optics (PO). The combination of the two techniques provides an efficient method in which to analyse finite structures.

The MoM technique has been widely used to analyse FSS and it is regarded as a highly accurate method. The only disadvantage of this technique is that its efficiency reduces as the problem space increases. This is because its solution matrix increases according to N2, where N is the number of unknowns, proportional to the size of the structure. This has a direct effect on the amount of memory required to solve the problem. The computational requirements are made even greater with the inclusion of a ground plane. Therefore, it is not feasible to use the MoM technique alone to predict the response of the FSS. The novel technique used to analyse finite FSS is an approximation to a full MoM solution. The basis of this technique is outlined in equation (1).

( ) ( )

( ) ( )

finite finite

finite infinite finite

MOM FSS GROUND

MOM FSS GROUND PO GROUND

+

≈ + + (1)

This equation states that the MoM solution for a finite FSS and ground can be approximated by a MoM solution for a finite FSS on an infinite ground and the effects of the truncation effects of the ground are accounted for by the PO solution.

4.3.1 FSS software

A general purpose software has been developed at ERA using the approach outlined in equation (1). The software combines two commercially available numerical tools; FEKO (a MoM package); and GRASP8, a package which can be used to calculate PO solutions of arbitrary metallic structures. In addition, proprietary software written by ERA, called GUIDE, is used to calculate the reflection and transmission response of any dielectric layers present in the model. A top-level program, using MATLAB, has been created to control the software operation. This code requests all of the required design parameters (ie, the element type, the number of layers between the ground and the FSS and their electrical properties, the number of elements, the polarisation and angle of incidence of the incoming plane wave). Many other parameters can also be defined and, in addition to this, parametric analysis of the aforementioned design parameters can be undertaken using the software, making it very flexible and powerful for finite FSS design.


40

To test the validity of this novel technique, a finite size FSS was analysed using the full MoM and the hybrid MoM / PO technique. The FSS under investigation is based on the design reported by Johansson, in which dipole elements are placed on top of four dielectric layers and a ground plane. The FSS is modelled using 10 x 10 elements to keep the computational time and the memory requirements within a realistic range for the full MoM solution. The results from the full MoM technique and the hybrid method are shown in Figure 32. The results show the response of the FSS at 9 GHz for an incidence angle 25°.

Figure 32: Johansson FSS solved with full MoM and the MoM/PO solution

A comparison of the results from the two methods show reasonable agreement in the prediction of the grazing lobe level and excellent prediction of the grating lobe angle. The real benefit of the hybrid method is in the computational time required to model the geometry. The full MoM solution took 2 days to run whereas the hybrid method took 10 minutes. The obvious advantage of the hybrid method is that it can be used to evaluate different FSS structures in a relatively short space of time with a high level of accuracy.

4.3.2 Design of a grating lobe FSS

There are two prime factors that determine the suppression and conversion of the reflected energy into a gating lobe; the FSS element size and period, and the electrical separation of the FSS and ground plane. For the initial test piece, the latter was fixed and was based on a standard thickness of FoamClad substrate manufactured by Arlon. FoamClad is a novel laminate constructed from a low dielectric constant microporous polymeric core on which impermeable copper clad polymeric membranes are bonded to both sides. The stated dielectric constant of the material is 1.2 for a 2.4 mm thick sample at 1MHz.

The FSS was designed using the element type and the periodicity to control the reflected and grating lobe for a fixed incidence angle of 5° in θ and 0° in φ, based on a standard Cartesian system. The element type chosen for the FSS was a cross dipole and the design frequency was 29GHz. An extensive parametric study in which the length and width of the elements and the lattice and ground plane spacing were varied, was undertaken. These properties were kept within realistic limits for


41

manufacturing purposes. The optimum element size and lattice spacing for a scaled design at 29GHz is shown in Figure 33, while substrate topology is shown in Figure 34.

X

Y

4.3

4.3

0.5

10.7

7

All dimensions in mm

Figure 33: Cross dipole element size and lattice

Microporous core

Polymer film

Polymer film

Ground Plane

FSS Elements

2.44 mm

Figure 34: Substrate topology

The optimum design was chosen to give the largest reduction in the specular level when compared to a level obtained from a metal reference sheet having the same dimensions as the FSS. Figure 35 shows the predicted performance of the optimised FSS design compared to a metal reference. The suppression of the specular lobe for this design is in the order of 25 dB. The response of the both the metal sheet and the FSS over an angular range of 180° in θ and 360° in φ is shown in Figure 36. The minimum amplitude level in these plots is -30 dB. They clearly show that no spurious energy is being transmitted into additional grating lobes and therefore all the energy is being transmitted in the plane of incidence.


42

Figure 35: FSS performance compared with a metal sheet as reference

Figure 36: Full 3 dimensional plane wave response of a metal sheet and the MC-FSS to a plane wave at 5° incidence in θ

4.3.3 Analysis of MC-FSS in a corridor

4.3.3.1 Analysis approach

Predicting the MC-FSS performance in a right angle bend corridor, as shown in Figure 37, is a difficult problem. No software tools are commercially available which can solve for the complete


43

problem without being too computationally extensive. Hence, a methodology has been developed as part of the work to enable the analysis of an MC-FSS in an enclosed metallic right angle bend corridor. The technique is based on a hybrid method used in conjunction to a plane wave spectra formulation.

First, the MC-FSS structure is characterised in isolation. The theory of Floquet modes is used to determine the amount of reflection achieved for each propagating mode. The scattered field strength is normalised and quantified to obtain the reflection coefficients of the MC-FSS in different look directions. Once the MC-FSS has been characterised for various directions of incidence, the rest of the environment is analysed without the MC-FSS structure, as shown in Figure 38. The field incident on the aperture (where the MC-FSS will be installed) is then expanded in terms of reflected Floquet coefficients in various directions. Combining each of these reflected Floquet coefficients with the normalised MC-FSS coefficients, it is possible to compute the scattered field caused by the MC-FSS at any point in space. By choosing to compute the scattered field on an aperture at the MC-FSS location, this field can be used as a source in a ray-tracing program to determine the field caused by the MC-FSS at the receiver point (see Figure 39).

It should be noted that, due to the limitations of the version of FEKO used in the analysis, only metallic “corridors” could be a analysed. Future versions of FEKO should allow the modelling to be carried out using lossy dielectric walls.

MC-FSS

Receiver

Transmitter

x

z

y

Figure 37: Ray tracing model of MC-FSS in a right angle bend corridor


44

Receiver

Transmitter

Aperturex

z

y

Figure 38: Ray tracing model of the right angle bend corridor without the MC-FSS

Receiver

Aperture sourcex

z

y

Figure 39: Ray tracing model of the right angle bend corridor using aperture source

4.3.3.2 The characterisation of the MC-FSS

The complete formulation of the scattered field at the receiver point is obtained by adding the direct contribution from the transmitter (in the absence of the MC-FSS) with the indirect contribution from the MC-FSS. This technique is described in detail below.

The periodic structure lies on the (x,y) plane as shown in Figure 40.


45

x

y

z

trr

zo

Incident unitplane wave

unit cell area

reflection plane

scatterer

Figure 40: Geometry of the unit cell.

A tangential vector position can be defined as:

ˆ ˆtr xx yy= +r (1)

The tangential incident propagation vector can be written as:

ˆ ˆt x yk k x k y= +r

(2)

The tangential modal propagation vector is:

( )1 2tpq tk k pk qk= − + +r r r r

(3)

where p and q are the Floquet mode indexes and 1kr

and 2kr

are expressed in terms of the skew lattice

components shown in Figure 41:

12 ˆˆ q

p

k z SA

π−= ×

r and (4)

22 ˆˆ q

p

k z SAπ

= ×r

(5)


46

pS is the periodicity along the x direction, qS is the periodicity along the y direction and pA is the

unit cell area.

ˆ ˆp p qA S S= × (6)

y

xz

Sq

Sp

α

Figure 41: Skew lattice geometry

The unit TM and TE modal vectors are defined as:

2ˆ ˆ tpqTMpq pq

tpq

ku u

k= =

r

r (7)

( )1ˆ ˆ ˆˆTEpq pq TMpqu u z u= = − × (8)

The TE and TM modal vectors:

( ) ( )1 11 ˆexp expTEpq pq pq tpq t pq

p

j z jk r uA

ψ ψ γ± ±= = ⋅ ± ⋅ − ⋅ ⋅rr r r

(9)

( ) ( )2 21 ˆexp expTMpq pq pq tpq t pq

p

j z jk r uA

ψ ψ γ± ±= = ⋅ ± ⋅ − ⋅ ⋅rr r r

(10)

where pqγ is the z component of the modal propagation vector:

22

pq tpqk kγ = −r

(11)


47

Note that TE incidence means that the electric field is directed along φ− and TM incidence means

that the electric field is directed along θ+ .

Developing the scalar vectors of equations (4) and (5), we obtain

ˆ ˆ ˆˆ sin cosq q qz S S x S yα α× = − + (12)

ˆˆ p pz S S y× =r

(13)

Therefore equations (4) and (5) can be written as

12 2ˆ ˆ

tanp p

k x yS Sπ π

α= −

r (14)

22 ˆsinq

k yS

πα

=r

(15)

The tangential modal propagation vector may be written as:

ˆ ˆtpq xpq ypqk k x k y= +r

where (16)

2

2 2tan sin

xpq xp

ypq yp q

pk kS

p qk kS S

π

π πα α

= − − = − + −

(17)

Rewriting the unit modal vector components, the following expression is obtained:

2

ˆ ˆˆ ˆ ˆxpq ypq

pq a btpq

k x k yu k x k y

k

+= = +r (18)

1 ˆ ˆpq b au k x k y= − (19)

where 2 2

xpqa

xpq ypq

kk

k k=

+ (20)

2 2

ypqb

xpq ypq

kk

k k=

+ (21)


48

Expanding the modal vector components, the following expressions are obtained:

1 1 1ˆ ˆpq xpq ypqx yψ ψ ψ± ± ±= +r

(22)

2 2 2ˆ ˆpq xpq ypqx yψ ψ ψ± ± ±= +r

(23)

where, ( ) ( )( )1 exp expbxpq pq xpq ypq

p

k j z j k x k yA

ψ γ± = ⋅ ± ⋅ − + (24)

( ) ( )( )1 exp expaypq pq xpq ypq

p

k j z j k x k yA

ψ γ± −= ⋅ ± ⋅ − + (25)

( ) ( )( )2 exp expaxpq pq xpq ypq

p

k j z j k x k yA

ψ γ± = ⋅ ± ⋅ − + (26)

( ) ( )( )2 exp expbypq pq xpq ypq

p

k j z j k x k yA

ψ γ± = ⋅ ± ⋅ − + (27)

The reflected tangential electric field can be expanded as a series of basis function (or modal vector functions) such as:

( ) ( )2

1t t kpq kpq t

k p q

E r a rψ+∞ +∞

= =−∞ =−∞

= ∑ ∑ ∑r rr r

(28)

Multiplying equation (28) by the complex conjugate of kpqψr and integrating over the unit cell area,

the following is obtained:

2

* *

1

( ) ( ) ( ) ( )p p

t t kpq t kpq kpq t kpq tk p qA A

E r r ds a r r dsψ ψ ψ+∞ +∞

= =−∞ =−∞

= ∑ ∑ ∑∫ ∫r r r r r

(29)

The modal reflection amplitude coefficient may be obtained by using the orthogonality property of the modal vectors.

*( ) ( )p

kpq t t kpq tA

a E r r dsψ= ⋅∫r rr r

(30)

Here, the modal reflection amplitude coefficients kpqa are expressed in terms of the tangential

scattered field on the reflection plane tEr

and the modal vector conjugate *kpqψr . The field tE

r can be

calculated using a full wave analysis tool such as the finite element software HFSS while the vectors *kpqψr can be found analytically. The integration of the right hand term of equation (30) may then be


49

carried out over the unit cell area pA with the appropriate sampling criteria (less than one twentieth of

the wavelength).

4.3.3.3 Modelling the right angle corridor without the MC-FSS

The hybrid method requires that the right angle bend in the corridor can be modelled without the MC-FSS as shown in Figure 38. A hole is now present where the MC-FSS structure is to be installed. The modelling of such a structure can easily be carried out using a ray tracing EM analysis program. The well-known FEKO software is used for this purpose. The objectives of this analysis are two fold. First, the direct contribution from the transmitter can be computed. Secondly, the incident field that would hit the MC-FSS structure if it were there can be quantified by simply computing the field on the aperture. The field incident on the aperture can then be expanded in terms of plane wave spectra; ie the incoming waves hitting the MC-FSS are represented as a series of plane waves coming from various directions in space and having different amplitudes.

Expressing the tangential incident electric field on the aperture as the sum of a series of plane wave, the following expression is obtained:

1 100 2 200( ) ( , ) ( , , ) ( , ) ( , , )it t t tE r b r b r d d

α β

α β ψ α β α β ψ α β α β+ += +∫ ∫r r rr r r

(31)

where α and β are the direction cosine of the plane waves

sin cosα θ φ=

sin sinβ θ φ=

Note that, for the incident wave, only the dominant Floquet mode is propagating, which is characterised by the indexes ( , ) (0,0)p q = . The amplitude coefficients 1b and 2b represent the field

strength incident on the MC-FSS from a given direction from the TE and TM polarisation respectively.

Separating the x and y components inside the double integration, equation (30) may also be written as:

ˆ ˆ( ) ( , , ) ( , , )i i it t tx t ty tE r r x r y d d

α β

α β α β α β= Φ + Φ∫ ∫r r r r

(32)

where 1 2( ) ( , ) ( ) ( , ) ( )itx t TExpq t TMxpq tr b r b rα β ψ α β ψ+ +Φ = +r r r

(33)

1 2( ) ( , ) ( ) ( , ) ( )ity t TEypq t TMypq tr b r b rα β ψ α β ψ+ +Φ = +r r r

(34)

Rewriting the incident tangential field in a more compact manner, the following expressions are obtained:


50

ˆ ˆ( ) ( ) ( )i i it t tx t ty tE r E r x E r y= +r r r r

(35)

( ) ( , , )i itx t tx tE r r d d

α β

α β α β= Φ∫ ∫r r

(36)

( ) ( , , )i ity t ty tE r r d d

α β

α β α β= Φ∫ ∫r r

(37)

Expanding the components of equations (33) and (34) leads to:

( ) ( )( )1 2 00 00 001( ) ( , ) ( , ) exp expi

tx t b a x yp

r b k b k j z j k x k yA

α β α β γΦ = + ⋅ ± ⋅ − +r

(38)

( ) ( )( )1 2 00 00 001( ) ( , ) ( , ) exp expi

ty t a b x yp

r b k b k j z j k x k yA

α β α β γΦ = − + ⋅ ± ⋅ − +r

(39)

It must be noted that in this case, pA represents the aperture shown in Figure 38 and should not be

mistaken with the unit cell aperture encountered earlier in this report to compute the kpqa coefficients

Assume now that we conveniently place the aperture at z=0 as shown in fig.mo2, we have:

( )00exp 1j zγ± = (40)

Moreover, the components of the tangential propagation vector can be expressed as:

00x xk k kα= − = − (41)

00y yk k kβ= − = − (42)

where 2k πλ

= (43)

Therefore equations (38) and (39) can be reduced to:

( )( )1 21( ) ( , ) ( , ) expi

tx t b ap

r b k b k jk x yA

α β α β α βΦ = + ⋅ + +r

(44)

( )( )1 21( ) ( , ) ( , ) expi

ty t a bp

r b k b k jk x yA

α β α β α βΦ = − + ⋅ + +r

(45)

Rewriting equations (36) and (37), we have:


51

( )( )( ) ( , ) expip tx t xA E r Q jk x y d d

α β

α β α β α β= + +∫ ∫r

(46)

( )( )( ) ( , )expip ty t yA E r Q jk x y d d

α β

α β α β α β= + +∫ ∫r

(47)

where 1 2( , ) ( , ) ( , )x b aQ b k b kα β α β α β= + (48)

1 2( , ) ( , ) ( , )y a bQ b k b kα β α β α β= − + (49)

Using the inverse fourier transform on equations (46) and (47), we obtain:

( )( )2( , ) ( ) expp ix tx t

x y

AQ E r jk x y dxdyα β α β

λ= ⋅ − +∫ ∫

r (50)

( )( )2( , ) ( ) expp iy ty t

x y

AQ E r jk x y dxdyα β α β

λ= ⋅ − +∫ ∫

r (51)

Solving the system of two equations (48) and (49) with two unknowns, we have:

1 2 2

( , ) ( , )( , ) b x a y

a b

k Q k Qb

k kα β α β

α β−

=+

(52)

2 2 2

( , ) ( , )( , ) a x b y

a b

k Q k Qb

k kα β α β

α β+

=+

(53)

The coefficients 1b and 2b have now been expressed in terms of ak , bk , xQ and yQ . The constant

ak and bk can be found analytically. xQ and yQ are calculated by integrating the field over the

aperture as shown in the equations (50) and (51) . This field is computed, in turn, by the ray-tracing program FEKO.

4.3.3.4 Predicting the MC-FSS contribution within the right angle bend corridor

Here, the formulation of the reflected field at the receiver point caused by the presence of the MC-FSS is described. The objective consists of combining the kpqa and the kb coefficients in a plane wave

spectrum formulation. To recap, the kpqa coefficient is the reflection coefficient of the MC-FSS for a

given mode. The kb coefficient is the field amplitude that is incident on the MC-FSS. Since the kpqa

coefficient has been obtained according to a specific direction of illumination, it may be combined


52

with the kb coefficient corresponding to the same direction of illumination. Hence, the kb coefficient

is a weighting factor to be applied to the normalised kpqa coefficient to obtain the true field value.

Having established the principle of the hybrid method, the electric field caused by an aperture at this point can be expressed as:

1 1 2 2ˆ ˆ( ) ( ) ( )E r E r u E r u= +r r r r

(54)

where 1u and 2u are respectively the TE and TM unit vectors and 1( )E rr and 2 ( )E rr are the TE and

TM field components.

1 1 1 1( ) ( , ) ( )pq pq

p qE r b a r d d

α β

α β ψ α β+∞ +∞

=−∞ =−∞

= ∑ ∑∫ ∫r r

(55)

22 2 22 2

( , )( ) ( )1

pq pqp q

bE r a r d dα β

α β ψ α βα β

+∞ +∞

=−∞ =−∞

=− −

∑ ∑∫ ∫r r

(56)

Additionally, it may be useful to express the field in terms of x and y components:

ˆ ˆ( ) ( ) ( )x yE r E r x E r y= +

r r r r (57)

The electric components can then be written as:

1 1 1

22 22 2

( , ) ( )

( )( , ) ( )

1

pq xpqp q

x

pq xpqp q

b a r

E r d db a rα β

α β ψ

α βα β ψα β

+∞ +∞

=−∞ =−∞

+∞ +∞

=−∞ =−∞

= + − −

∑ ∑∫ ∫

∑ ∑

r

r

r (58)

1 1 1

22 22 2

( , ) ( )

( )( , ) ( )

1

pq ypqp q

y

pq ypqp q

b a r

E r d db a rα β

α β ψ

α βα β ψα β

+∞ +∞

=−∞ =−∞

+∞ +∞

=−∞ =−∞

= + − −

∑ ∑∫ ∫

∑ ∑

r

r

r (59)

Equations (58) and (59) represent the scattered electric field components caused by the MC-FSS at the aperture. This field quantity can then be used as an aperture source to determine the scattered field caused by the MS-FSS inside the corridor at the receiver point (see Figure 39).


53

4.3.3.5 Predicting the field at the receiver point

Considering now the problem described in Figure 37, where energy is transmitted towards a right angle corridor with an MC-FSS, the field quantity at the receiver point is made up of two distinct contributions. The direct contribution is the energy travelling from the transmitter to the receiver end through bounces on the corridor walls without hitting the MC-FSS. The second contribution is from the MC-FSS, where the energy reflected by the MC-FSS is seen as a secondary source. Designating

the scattered field caused by the direct contribution as ( )dirsE rr r

and the scattered field caused by the

MC-FSS as ( )fsssE rr r

, the complete scattered field is:

( ) ( ) ( )dir fsss s sE r E r E r= +r r rr r r

(60)

where rr is the vector position at the receiver point.

4.3.3.6 Software verification

4.3.3.6.1 Metallic plate (9 GHz)

A FSS problem consisting of a metallic plate was used to test the accuracy of the program. A metallic plate 2 by 2λ wide at 9 GHz (i.e: 66.66 by 66.66 mm) is illuminated by an incident plane wave coming at an angle θ from the normal axis (see Figure 42).

Incident planewave

Reflected wave

Metal plate

Surface for fieldcalculation

θθ

Normal

z

x

Figure 42: Metallic plate test case

The electric field strength is computed on the computational surface using the plane wave spectra method in conjunction with Floquet theory. The angle of incident is 30°. Figure 43 shows the field


54

value obtained on the computational plane. It can be seen, as expected, the reflected field peak value is the same as the incident plane wave.

Figure 43: Reflected electric field on the computational surface, 30° illumination

For checking purposes, the reflected field of Figure 43 is Fourier transformed to obtain the field in terms of direction cosines (see Figure 44). Looking now at the reflected field from the angular point of view, we note that a beam is present in the direction corresponding to an angle θ of -30° ( 0.5α = − and 0β = ). This is a confirmation that the software is producing the results expected. Note that the graph only shows the relative strength of the field but not the absolute value.

Figure 44: E-field pattern in the direction cosine space, 30° illumination (metallic plate case)


55

4.3.3.6.2 Unmatched Johansson FSS (9 GHz)

The program is tested further by considering the FSS presented in [Ref 1] in an “unmatched” scenario., ie with only a single layer of air below the FSS grid. The FSS consists of a grid of dipoles, periodically spaced and oriented along the y-direction, as shown in Figure 45.

Incident planewave

Reflected wave(specular term)

Metal plate

Surface for fieldcalculation

Normal

z

x

Reflected wave(higher floquet mode)

0,1θ

0,0θ

FSS grid

iθ

Figure 45: Johansson test case

Analysing the FSS according to the same methodology described as for the metallic plate test case, the reflected field is calculated on the computational surface. The field can be seen in Figure 46. The angular pattern of Figure 47 shows that two reflected beams are present. One beam is the specular reflection at the angle 0,0θ and the other beam is the first higher order mode reflection at angle 0,1θ .

The peak value of the mode (0,1) is at 0.83u = and 0v = (or 0,1 56θ = ° and 0,1 0φ = ), which is in

agreement with the theoretical data.


56

Figure 46: Reflected E-field on the computational surface, 20° illumination (Unmatched Johansson case)

Figure 47: E-field pattern in the direction cosine space, 20° illumination (Unmatched Johansson case)

4.3.3.6.3 Matched Johansson FSS (9 GHz)

The third test case deals with the complete Johansson structure with the multilayer dielectrics (Figure 28). The reflected field is computed on the computational surface and is shown in Figure 48 for an incident angle of 20°. The angular pattern of this field is shown in Figure 49. It can be seen that the specular term is at the same location as in case test No2, but the specular reflection is much stronger.


57

This is attributed to the various dielectric layers, which have the effect of matching the structure for optimum specular suppression.

Figure 48: Reflected E-field on the computational surface, 20° illumination (Matched Johansson case)

Figure 49: E-field pattern in the direction cosine space, 20° illumination (Matched Johansson case)

4.3.3.6.4 Johansson MC-FSS within right angle bend corridor (9GHz)

The matched Johansson MC-FSS was characterised in terms of reflection coefficient and placed inside the right angle bend corridor shown in Figure 37. The field prediction at the other end of the corridor was carried out according to the modelling methodology presented earlier in this report.


58

Figure 51 shows the calculated scattered electric field along a vertical line going from floor to ceiling at the receiver location. The direct and the MC-FSS contributions are represented as well as the true field values, which correspond to the sum of these two contributions. In addition, the modelling of a right angle bend corridor without the presence of the MC-FSS has been carried out (see Figure 50). The achieved transmission is used for comparison purposes. From Figure 51, it can be seen that the Johansson MC-FSS improves the transmission around the right angle bend corridor significantly. On average, an improvement of about 6 dB is achieved.

Receiver

Transmitter

x

z

y

Figure 50: Ray tracing model of right angle bend corridor (top view)

0

2000

4000

6000

8000

10000

12000

14000

16000

18000

20000

-0.06 -0.04 -0.02 0.00 0.02 0.04 0.06Y (m)

Mag

nitu

de (V

/m)

(direct + fss) direct fss Normal right angle bend corridor

Figure 51: Electric field inside corridor along vertical line (x=0.5;z=0.066).

4.3.4 Theoretical design and performance prediction

The new MC-FSS design described in Section 4.3.2 was characterised in terms of reflection and transmission coefficients and analysed within the right angle bend corridor. The scale frequency was


59

used to facilitate the manufacture of a small test piece. Figure 52 shows the field strength inside the corridor after the right angle bend. Figure 53 shows the propagation enhancement realised by having the MC-FSS in place. The improvement is about 7 to 8 dB after 6 wavelengths from the MC-FSS panel.

0.E+00

2.E+05

4.E+05

6.E+05

8.E+05

1.E+06

1.E+06

0.000 0.020 0.040 0.060 0.080 0.100 0.120 0.140

X (m)

Efie

ld M

agni

tude

(V/m

)

With FSS (crosses) Without FSS

direct contribution FSS (crosses) contribution

Figure 52: Electric field inside corridor along the horizontal centre line (Y=0;Z=0.0158m)

-20

-15

-10

-5

0

5

10

15

20

0.000 0.020 0.040 0.060 0.080 0.100 0.120 0.140X (m)

dB

Figure 53: Field strength enhancement by FSS (crosses) inside the corridor along the horizontal centre line (Y=0, Z=0.0158 m)

The scattered field has also been calculated on a horizontal surface at the centre of the corridor after the right angle bend corridor. Figure 54 shows the strength of the scattered field for the metal right-angle bend corridor alone. Figure 55 shows the strength of the scattered field for the right-angle bend corridor containing the MC-FSS, showing individual contributions and the combined affects, as follows:


60

(a) The right-angle bend corridor with an aperture instead of the MC-FSS in such a way as to predict the so-called direct contribution (i.e.: the contribution which is independent of the MC-FSS characteristic.)

(b) The right-angle bend corridor with the excitation that the MC-FSS provides when it is illuminated by a point source. This case enables the calculation of the so-called MC-FSS contribution.

(c) The right-angle bend corridor in the presence of the MC-FSS. The field values are the addition of both the direct and the MC-FSS contributions.

It can be seen that the MC-FSS contribution is stronger than the direct contribution at places most distant from the bend (Figure 55 (a) and (b)). This suggests that the MC-FSS contributes significantly to sustain the propagation of energy after the right-angle bend.

The comparison of Figure 54 and Figure 55 clearly shows the advantage of the MC-FSS. The field strength inside the corridor after the right-angle bend is stronger in the presence of the MC-FSS.

The level of propagation enhancement is quantified by the calculation of the following term,

20logFSSsREFs

EE

∆ =

where FSSsE is the scattered electric field obtained when the MC-FSS is present and REF

sE is the

scattered electric field obtained when the MC-FSS is absent. Figure 56 presents the result of this calculation on the computational surface. Enhancement of field propagation ranging from 5 to 20 dB is observed in many areas in the corridor.

0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09 0.10 0.11X (m)

0

0.01

0.02

0.03

Z (m

)

0

5E+004

1E+005

2E+005

2E+005

3E+005

3E+005

4E+005

4E+005

5E+005

5E+005

6E+005

Figure 54: Scattered field magnitude inside solid metal corridor after the right angle bend (29 GHz)


61

0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09 0.10 0.11X (m)

0

0.01

0.02

0.03

Z (m

)

0

5E+004

1E+005

2E+005

2E+005

3E+005

3E+005

4E+005

4E+005

5E+005

5E+005

6E+005

(a)

0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09 0.10 0.11X (m)

0

0.01

0.02

0.03

Z (m

)

0

5E+004

1E+005

2E+005

2E+005

3E+005

3E+005

4E+005

4E+005

5E+005

5E+005

6E+005

(b)

0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09 0.10 0.11X (m)

0

0.01

0.02

0.03

Z (m

)

0

5E+004

1E+005

2E+005

2E+005

3E+005

3E+005

4E+005

4E+005

5E+005

5E+005

6E+005

(c)

Figure 55: Scattered field magnitude inside the corridor after the right angle bend for MC-FSS case (29 GHz)

MC-FSS contribution

Complete field

Direct contribution


62

0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09 0.10 0.11X (m)

0

0.01

0.02

0.03

Z (m

)

-30

-25

-20

-15

-10

-5

0

5

10

15

20

Figure 56: Relative improvement of field strength (dB) inside the corridor (29 GHz)

4.4 Performance verification

4.4.1 FSS characterisation at scaled frequency

A 610 by 610 mm flat panel FSS was manufactured to the optimum design and the panel was initially characterised using ERA’s near field scanning facility, where the grating lobe performance was compared to a metal sheet.

The measurement set-up is shown in Figure 57. The FSS was illuminated using an offset reflector system fed using a circular corrugated horn, which is shown in Figure 57(b). Using this approach, the FSS can be illuminated with a pseudo-plane wave for all frequencies of interest. The polarisation of the incident wave from the reflector is vertical, with the larger element spacing in the horizontal plane, exciting the grating lobe in the direction of the scanner. The reflector was parallel to the plane of scan of the measurement. The FSS was supported in a wooden frame and was offset from the reflector’s normal direction by an angle α. This offset angle determines the incidence angle of the field on the FSS.


63

Plan

e of

sca

n

Lathe carriage

Lathe bed

2.4

met

res

max

imum

sca

n ra

ngeProbe

FSS normal

Modedirection

Plan

e of

sca

n

α deg

Incident field direction

Incident fielddirectionFeed horn

Reflector

FSS

Reflector

(a)

(b)

Figure 57: Near field scanner measurement set up, (a) planar scanner, (b) FSS excitation

Using this configuration, the specular reflection propagated away from the scanner and therefore no direct field contribution was measured. This set-up therefore only determined the location and amplitude of the grating lobe relative to a ground plane and not the suppression of the reflected energy. However, assuming that there is only one dominant lobe and no significant losses, the energy in the lobe in the specular direction can be determined using the principal of conservation of energy.

Measurements using the set up outlined above were performed with an offset angle α of 5°, corresponding to the angle of incidence used to optimise the cross dipole FSS design. The FSS and


64

ground plane were measured for a frequency range of 25 GHz to 30 GHz so that any effects caused by manufacturing tolerances could be observed.

A comparison of the FSS grating lobe peak versus frequency for an incident angle of 5° is shown in Figure 58. Each plot is normalised to the maximum peak grating lobe level across the frequency range. The maximum peak was obtained at 27 GHz, shown as 0 dB in the figure. This indicates that the resonance of the FSS is around 27 GHz, which is below the predicted value of 28.8 GHz. One important factor that may be contributing to this effect is the permittivity of the FoamClad material. The manufacturer quotes a measured permittivity value of 1.2 for a frequency of 1 MHz and there is currently no data available for the electrical properties of the material at higher frequencies. Any difference in the permittivity of the material around 29 GHz will cause the electrical length between the FSS elements and the ground plane to change, altering the resonance of the structure.

Figure 58: Comparison of the FSS grating lobe versus frequency

The grating lobe response of the FSS, at 27 GHz and 29GHz, is compared with ground plane measurements in Figure 59. In both cases, the measurements have been normalised to the maximum peak of the grating lobe. The ability of the FSS to convert the reflection energy into grating energy can be easily observed from this figure. The difference between the peak grating lobe level and the ground plane response at 27GHz is over 40 dB. This indicates that the FSS is highly efficient in converting the reflected energy into grating lobe energy. At 29 GHz, a suppression of nearly 20dB is still demonstrated; however, it is clear that the performance of the FSS is not as good at this higher frequency. The structure appears to have a usable bandwidth of about 10%.


65

Figure 59: FSS and ground plane measurement at 27 GHz and 29GHz

4.4.2 Measurements in a right angle bend corridor at scaled frequency

The measurements of the flat panel described earlier provide an indication of the performance of the FSS under a fixed angle plane wave excitation. These results showed that the resonant frequency of the structure was about 27 GHz. However, the performance of the FSS under excitation from multiple plane waves incident onto the FSS from different angles and with various polarisations cannot be predicted from the measurements. To test the performance of the FSS under these conditions, a scaled right-angled corridor was manufactured out of aluminium. The real dimensions of a typical corridor have been scaled to reflect the frequency of operation of the FSS. At 29 GHz, the inner dimensions of the corridor are approximately 3λ. Therefore, the corridor’s equivalent size at the operating frequency of 400 MHz is 2.25 metres. It should be noted that the “corridor” was entirely metallic and only about 6λ long in each direction. These limitations were subsequently found to be significant since, in the metallic environment, the fields did not reach steady state by the end of the test piece.

Two samples of the flat panel FSS were cut from the large test panel to fit the inner dimensions of the corridor; these samples are shown in Figure 60 and are labelled FSS1 and FSS2.

FSS 1 FSS 2

Figure 60: Two samples from the flat panel FSS


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FSS1 has an element in the middle of the sample while, in FSS2, the elements are offset from FSS1 in the vertical direction by half a period. These two samples have been used to identify whether the interruption to the FSS periodicity has any effect on the measured performance.

4.4.2.1 VNA measurement

A vector network analyser allows a swept frequency measurement to be performed on the scaled corridor. The set up for the measurement is shown in Figure 61.

TX port

RX

port

Open endedwaveguide

FSS flush or short

30

E

LminLmax

FSS protruding

Lmin = 29Lmax = 48

Absorber Corridor face

Figure 61: Setup for VNA measurement (dimensions in mm)

Two open-ended waveguide probes were used as transmitter and receiver. The TX port excited Vertical polarisation (ie, out of the page as drawn) direction and its aperture was located at a fixed position of 30mm away from the face of the corridor. The probe was attached to a metal plate which was connected to the corridor face, which was backed with absorber to effectively terminate the corridor. The absorber used had a typical reflectivity for normal incidence of better than -20dB at the frequency of the measurement. A similar arrangement was used at the receive port. At this port, the location of the probe inside the corridor was adjusted to measure the forward reflection coefficient (S21) at different positions. The probe was moved between 29 and 48mm from the face of the corridor. Both these probes were equidistant from the vertical and horizontal walls of the corridor.

The FSS samples were mounted on a removable metal plug making it easy to place the samples within the corridor. When the metal plug was fully inserted into the corridor, it lay flush to the wall. This was used as the reference case. To observe the effects of placement of the samples within the corridor, the measurements were performed either with the FSS protruding inside the corridor or flush to the wall of the corridor. Figure 62 shows the two configurations.


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Figure 62: FSS locations

The forward reflection coefficient was measured over a frequency range of 27 GHz to 30GHz, and for five positions of the RX probe within the corridor. The measurement results for the probe located at 48 mm and 29mm from the face of the corridor are shown in Figure 63. In these measurements, the FSS samples were flush to the wall of the corridor. The figures show the differential transmission performance for a fixed location inside the corridor. These two plots have been chosen to highlight the effect on the results of the probe location within the corridor. It is evident from the plots that the differential S21 varies quite rapidly with position and frequency. For example, at 48mm, the differential S21 is always above 5 dB across the 27 to 28 GHz frequency band, which means that FSS is aiding the transmission round the corridors. However, at 29mm, the differential S21 across this same band is nearly always below -5dB, which implies that no benefit close to the bend is obtained using the FSS.

Figure 63: Differential S21 of the FSS samples and the reference with the probe at 48 mm and 29mm from the face of the corridor

FSS Flush FSS Protruding


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This variability is explained by the complex nature of the fields within the corridor. The corridor is essentially a waveguide in which many modes are propagating. At the RX port, the probe receives all of the propagating modes, which combine constructively or destructively depending on their relative phases. The amplitude and phase of the modes will vary with the spatial position of the probe inside the corridor and also with frequency. The inclusion of the FSS in the guide will alter these amplitudes and phases, and may introduce new modes. Therefore, the received signal with the FSS in situ will differ to that of the metal plate. The received signal will also vary with spatial position and frequency as can be seen in Figure 63. It is anticipated that, in a real (non-metallic) corridor, the benefits of using the MC-FSS may be greater.

The differential S21 at 29 GHz for the five probe locations is shown in Figure 64. The measurement is compared with the theoretical modelling outlined in section 4.3.4. In the theoretical work, the FSS is modelled flush to the wall of the corridor and the FSS is modelled with an element in the centre. This corresponds to the flush position of FSS1 in the measurements. In Figure 64, the measurements where the FSS is flush show good agreement with the theory. The levels are reasonably consistent with theory, but are slightly offset in position. This offset can be attributed to slight inaccuracies in the positioning of the RX probe within the guide. Furthermore, it is clear from the results that the overall improvements in transmission reduce when the FSS is protruding into the corridor.

Figure 64: Comparison between measurement and theory for the corridor measurement

4.4.2.2 Measurement using the near field scanner

The measurements on the VNA provide a means of assessing the swept frequency performance of the FSS within the corridor. These results show that the measurements are highly sensitive to the location of the probe and the frequency. Additional measurements have been performed to completely


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characterise the performance of the FSS within the corridor using ERA’s near field scanner. The near field data inside the corridor was obtained using a length of semi-rigid coaxial cable which was moved to various locations inside the corridor. A small loop at the end of the cable acts like an infinitesimal magnetic field probe, approximating a point source with an isotropic radiation pattern making direct comparisons with theoretical results possible.

The scanner was used to precisely control the measurement locations of the probe within the corridor and measurement data was collected on 2D planar surfaces. This data acquisition procedure is shown in Figure 65. The probe extended into the corridor (in the x direction) to the centre point of the FSS, the two other dimensions were limited by the dimensions of the corridor. A number of measurements were performed in the x-y plane for varying z locations. The multiple planar measurements were reconstituted into a volumetric scan of the corridor could be assessed.

2D planes

Measurement direction

Probe

TX port

RX port

Y

XZ

FSS

Figure 65: Planar scan of corridor using the near field scanner

Results for a cut the x-z plane in the centre of the corridor for 27 GHz and 29 GHz are shown in Figure 66. These plots show the differential amplitude of the field within the corridor which is plotted on a logarithmic scale, and is represented in the plots as a colour scale.

The results show the variability of the field strength within the corridor. In certain regions of the corridor the FSS generates higher field strengths while in other regions the levels are lower. The only way to provide an assessment of the performance of the FSS within the corridor, using this measurement data, is to use statistical methods.


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Figure 66: Differential amplitude plot in the x-z plane of the corridor at 27 GHz and 29GHz

The mean signal strength over a region is therefore used to evaluate the FSS performance. Figure 67 shows how the data is extracted from the measurements over planar regions containing the y and z axes for various x locations. The analysis of the results was restricted to FSS1.

Figure 67: Regions used to calculate mean field strengths

The mean levels at 27 GHz, 28 GHz and 29 GHz for FSS1 mounted flush to the corridor wall are shown Figure 68. The mean values were calculated from a set of data points on a series of corridor cross-sections referred to as “Data planes” shown in blue above. At 27 GHz, the inclusion of the FSS increases the signal strength, albeit at different levels. Similar improvements are also obtained at 28 GHz, while at 29 GHz the FSS does not enhance the field strengths. This result is consistent with the flat panel measurements, where the resonant frequency of the FSS was shown to be at 27 GHz.

TX port

RX port

X Z

FSS

Data planes

Y

Data lines


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Figure 68: Mean differential amplitude levels inside the corridor for FSS1 mounted flush to the wall

Figure 69 shows some predictions of the mean field strength improvement inside the corridor. In this case, the field strength was averaged along the “Data lines” shown in Figure 67, since full 3D data was not calculated. The average propagation improvement is 6 dB from 0.070m onwards.

Some comparison between predictions and measurements can be carried out by inspection of Figure 68 and Figure 69. The difference between the results is believed to be associated with the different scenarios used in the modelling and theoretical analysis. Although the results were obtained using a different averaging procedure, a general improvement is observed in both cases.

-2

0

2

4

6

8

0.030 0.040 0.050 0.060 0.070 0.080 0.090 0.100X (m)

Tran

smis

sion

impr

ovem

ent (

dB)

Figure 69: Predicted average improvement of field strength inside the corridor (29 GHz)


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4.4.3 Measurements in a right angle bend corridor at 1 GHz

Following the scale model measurements, the MC-FSS design was scaled to resonate at about 870 MHz. To represent a realistic building implementation, commonly used building materials were used to construct the FSS. A re-optimised and scaled design is shown in Figure 70. 9.5 mm plasterboard and 80 mm air layers replace the FoamClad substrate material used in the scaled design. The dimensions of the cross dipole elements increase to 123 mm in length and 17.22 mm in width, and have an inter-element period of 344.6 mm in the horizontal direction and 230.9 mm in the vertical.

17.22Plasterboard (9.5)

Air gap (80)

Ground plane

FSS elements

All dimensions in mm

344.6

123

123230.9

Copper FSSelements

Figure 70: MC-FSS panel dimensions

A 2.1m x 2.1m FSS panel was constructed from three interlocking panels, each with a width of 0.7 m and height of 2.1 m. Two columns were accommodated on each panel, each containing nine elements, giving a total of 54 elements. A ground plane with the same dimensions of the FSS was placed at right angles to improve the efficiency of the system, as shown in Figure 71.

Figure 71: MC-FSS test panel picture


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The predicted response of the (finite) FSS panel (without the additional ground plane) for a 5º-incidence plane wave is shown in Figure 72. The performance of the FSS is compared to a metal reference sheet having the same dimensions as the FSS. The FSS suppresses the specular component by 13 dB relative to the reference case. This energy is mode-converted into grating lobe energy, which radiates in a direction of 63º away from the normal direction. It should be noted that the suppression is significantly poorer than that predicted for the higher frequency panel. This is because the predicted suppression relates to a finite panel, and the lower frequency panel had significantly fewer elements than the 600 x 600mm 29GHz design.

Figure 72: Predicted response of MC-FSS at 870 MHz

4.4.3.1 Basic measurement setup

Extensive measurements were carried out on the panel to evaluate the propagation enhancement resulting from the presence of the FSS. Measurements were carried out in a free space environment, and in a right-angled corridor. The measurements used a vector network analyser (VNA) connected to two log-periodic antennas operating in transit and receive mode. The transmit antenna was positioned so that the main beam is incident on the FSS at approximately 5°. The receive antenna was located at 90º to the surface of the FSS, ie with its main beam orthogonal to the main beam of the transmit antenna.

This setup was used to perform a broadband frequency transmission measurement, which was transformed into the time domain in order to analyse the delay spread of the measurement environment. A time gating procedure was used to suppress the direct component of the transmitted signal, which allows only the mode-converted signals from the FSS panel to be included in the measurement. This time gated, time domain signal was subsequently re-transformed into the frequency domain. The gate was set by analysing the signal in the time domain when a 45º inclined metal plane was placed in the path between the transmit and receive points as shown in Figure 73 (a). In an open environment, this measurement shows two distinct peaks, one relating to the direct signal


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between the antennas, and one to the signal reflected off the angled metal plate; the former of these was gated out. This measurement was also used as a reference case, since this inclined plane was highly efficient in re-directing the transmitted signal power. The measurement was repeated with and without the FSS in-situ as shown in Figure 73 (b) and Figure 73 (c) respectively.

Inclined ground FSS No FSSMC-FSSMetallic groundTransmit / receive antennas

(a) (b) (c)

Figure 73: Measurement setup: (a) inclined metal plane, (b) FSS, (c) no FSS

4.4.3.2 MC-FSS in a free space environment

Initial measurements were performed within an open plan room with dimensions of approximately 20m x 5m. In this scenario, the direct transmitted signal level was large; the time gating procedure outlined above suppressed this component so that the only the signal contribution from the FSS was included. Measurements were taken with the receive antenna in different positions to determine the signal level improvements for various spatial locations mitigating the effects of multipath.

A total of five repeat measurements were taken at each spatial location, and the results were averaged to estimate the achievable improvements in signal quality due to the FSS. The performance of the inclined ground, the “no FSS” and the “with FSS” cases are shown respectively in Figure 74. At 870 MHz, the inclined ground plane produces a signal level of –33 dB, whereas the “no FSS” and “with FSS” cases produce levels of –54 dB and –44 dB respectively. For this scenario, the “with FSS” produces an improvement in the signal level of 10 dB relative to the “no FS” case.


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Figure 74: Comparative performance of the MC-FSS

The measurements were repeated with the receive antenna rotated to be in line with peak grating lobe direction of the FSS (ie, at approximately 60º). The averaged result from this location is shown in Figure 75 for the three measurement cases. Moving the receive antenna to the peak grating lobe angle had little effect on signal strength received using the inclined metal plane. At 870 MHz, the signal improvement produced by the FSS is 10 dB. An observation from this measurement is that even though, at resonance, the FSS improves the signal by the same amount, the operational bandwidth is much broader. When the receive antenna is located at 90º to the face of the FSS the bandwidth is 3%, whereas if it located toward maximum radiation the bandwidth is over 11%. When the measurement is performed close to the direction of the peak of the grating lobe, the frequency sensitivity is reduced since the grating lobe pattern roll-off is less in this region. Therefore, as measurement tends away from this region an equivalent frequency shift produces a greater change in pattern level and hence reduces the operational bandwidth of the FSS.

Figure 75: Comparative performance of the MC-FSS (at the peak grating lobe angle)


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4.4.3.3 MC-FSS in a right-angled corridor

The MC-FSS is primarily intended to improve signal levels in non-line-of-sight paths such as right angle corridors in buildings. Therefore, the measurements were repeated within a corridor located in a building adjacent to ERA. A plan view of the setup is shown in Figure 76 and a picture of the FSS in-situ is shown in Figure 77. The three measurement cases outlined in Figure 73 were repeated for various receive antenna locations within the corridor. Multiple measurements were performed for each spatial location and the results were again averaged to remove any multipath effects.

2.25

1.55

8.1Goods

lift

Doubledoors

FSS wallpaperGround plane

TX antenna

RX antenna

Doubledoors

Ceiling to floor = 2.35

2.1

1.55

VNA

All dimensions in metres

Figure 76: MC-FSS measurement setup

Figure 77: MC-FSS in the corridor


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A comparison of the received signal levels for the three measurement cases within the corridor is shown in Figure 78. At the design frequency of 870 MHz, the FSS generates an average signal level within the corridor of –43.5 dB, which is 13.5 dB above the case when no FSS is present. This measurement shows marginally better performance than for the free space measurement. This improvement over the free space case is caused by the corridor acting as a lossy waveguide which bounds the signal energy due to reflections towards the receiver, therefore increasing the signal level.

Figure 78: Comparative performance of the MC-FSS in a corridor

4.5 Manufacturing technologies

The manufacturing technologies that can be used to produce the MC-FSS panels are very similar to those described in Annex 3 for cavity wall FSS structures.

4.6 Regulatory issues

The MC-FSS structure is installed at the knee of corners and must be the full height of the corridor for best results. One wall may protrude from the wall (typically about 160 mm) and would be made from plasterboard (not foil backed) spaced from the wall with an air or foam cavity. The orthogonal section must be a suitable reflective material in the same plane as the corridor wall (e.g. foil backed plaster board). Table 4 below provides an initial assessment of the impact that the MC-FSS will have on building regulations.


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Table 4: Impact of Building Regulations on MC-FSS


A N/A


Pass where sufficient width is provided (see note 1)

C - L N/A

M Special measures required to achieve compliance

Pass where sufficient width is provided (see note 1)

N N/A


Pass

1. The protruding part of the system will locally reduce the corridor width, which is a dimension that may be controlled by Parts B and M of the Building Regulations. Part B1 of the regulations makes requirements for means of escape from buildings and Approved Document B gives recommendations for corridor widths when they are part of escape routes. However, the recommended minimum widths are dependent upon the number of people expected to use the route and conflicts with the regulations may occur in some cases. For example, Approved Document B recommends a minimum width of 900mm for a corridor intended as an escape route for up to 110 people and a width of 1100mm where the number of users is up to 220. Part M2 of the regulations requires reasonable provision to be made for disabled people to use the building. Approved Document M makes recommendations for the width of corridors to which wheelchair users have access, so that wheelchairs may be manoeuvred and other people may pass. In many circumstances, an unobstructed minimum width of 1200mm is recommended.

4.7 Summary

This work has shown that the MC-FSS technique can improve RF propagation in complex building environments considerably. Improvement of more than 10dB over quite wide operating bandwidths have been demonstrated. The MC-FSS structure appears to have significant potential to improve access to areas deep inside buildings, such as basements and internal stairwells more straightforward. The technique appears to offer better performance at higher frequencies where the structures used at electrically larger. Structures used for lower frequency applications cannot be designed on the basis of an “infinite array”, since truncation effects are significant. The technology could also be very useful for much higher frequency applications, for example future mm-wave Wireless LAN systems.

It should be noted that the structure was designed to operate with vertical polarisation and all the measurements were done using vertically polarised signals. Theoretical analysis showed that the screen acts like a metal sheet in response to horizontally polarised signals. One of the key aspects of


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any further work will be to assess the potential to design a screen that operates in all polarisations, probably based on using circularly symmetric elements such as rings with different lattice configurations.

It is considered that this technology is an ideal candidate for further work, which should look at techniques to make it usable for all RF signal polarisations, to extend its bandwidth capability and to address implementation issues relating to its use to enhance signals in stairwells and lift shafts as well as corridors.

5. References

[Ref 1] F. Stefan Johansson, Lars G. Josefsson, Torild Lorentzon, A novel frequency-scanned reflector antenna, IEEE Transactions on Antennas and Propagation, vol. 37, No. 8, August 1989.

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