Ultra-Wideband Power Amplifier Design - DiVA portal

Department of Science and Technology Institutionen för teknik och naturvetenskap Linköpings Universitet Linköpings Universitet SE-601 74 Norrköping, Sweden 601 74 Norrköping

ExamensarbeteLITH-ITN-ED-EX--06/010--SE

Ultra-Wideband PowerAmplifier Design

Magnus Ståhl

2006-03-03

LITH-ITN-ED-EX--06/010--SE

Ultra-Wideband PowerAmplifier Design

Examensarbete utfört i Elektronikdesignvid Linköpings Tekniska Högskola, Campus

Norrköping

Magnus Ståhl

Handledare Ian RayExaminator Adriana Serban Craciunescu

Norrköping 2006-03-03

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LITH-ITN-ED-EX--06/010--SE

Ultra-Wideband Power Amplifier Design

Magnus Ståhl

Power Amplifiers (PA) are large-signal amplifiers. This means that a large part of the load-line is usedduring signal operation. PAs are normally used as the last stage of communication electronics to providelarge enough signals to be transmitted. This thesis describes the design of an ultra-wideband power amplifier. As the first part of this thesis, thepower amplifier design is presented. In the second part of the thesis a printed circuit board (PCB) wasdesigned and together with the designed circuit tested. The amplifier was designed to have an output of5 W delivered at the frequency band from 500 MHz to 1 GHz with a gain of 15 dB.

Power Amplifier, wideband

Upphovsrätt

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© Magnus Ståhl

Design of an Ultra-Wideband Power Amplifier

Master Thesis

by

Magnus Ståhl

Supervisor: Ian Ray Examiner: Adriana Serban Craciunescu

Department of Science and Technology Linköping University SE-601 74 Norrköping, Sweden

Institutionen för teknik och naturvetenskap Linköpings Universitet

601 74 Norrköping


I

Abstract Power Amplifiers (PA) are large-signal amplifiers. This means that a large part of the load-line is used during signal operation. PAs are normally used as the last stage of communication electronics to provide large enough signals to be transmitted. The transistors used in PA applications needs to be able to handle large amounts of power and heat otherwise they may be destroyed when used in the application. Therefore there is a large demand for materials that can handle these amounts of power and heat. One of the most interesting materials for this is silicon-carbide (SiC), which has high breakdown-voltage and can handle more current than the more common material for transistors, silicon. One of the most difficult tasks when designing PA is to design them with a large bandwidth, make them wideband. To achieve the wideband one often has to sacrifice some of the gain. This thesis describes the design of an ultra-wideband power amplifier. As the first part of this thesis, the power amplifier design is presented. In the second part of the thesis a printed circuit board (PCB) was designed and together with the designed circuit tested. The amplifier was designed to have an output of 5 W delivered at the frequency band from 500 MHz to 1 GHz with a gain of 15 dB.


II

Sammanfattning Effektförstärkare har till uppgift att leverera en stor uteffekt. För att kunna göra det måste maximalt av arbetslinjen användas. Effektförstärkare används vanligtvis som sista steg i kommunikations elektronik där en stor uteffekt är efterfrågad. När stora effekter produceras medför det att komponenten blir väldigt het, vilket kan medföra att komponenten förstörs. Därför finns stor efterfrågan av halvledarmaterial som klarar av de höga temperaturerna. Ett av de mest intressanta materialen för det är kiselkarbid (SiC). SiC har en hög breakdown-voltage och klarar av att leverera mer ström än det mer vanliga halvledarmaterialet kisel. Ett av de största problemen vid konstruktion av effektförstärkare är att göra dem bredbandiga, ofta måste förstärkning offras för att få den önskade bandbredden. I detta examensarbete beskrivs konstruktionen av en ultra-wideband effektförstärkare. Som första del av arbetet presenteras konstruktionen av förstärkaren. I andra delen av arbetet gjordes tester på konstruktionen tillsammans med ett mönsterkort. Förstärkaren konstruerades för att leverera en uteffekt av 5 W i frekvensbandet 500 MHz till 1 GHz med en förstärkning av 15 dB.


III

Acknowledgments This report is the result of a master thesis done at the Department of Science and Technology (ITN) at the Technical University of Linköping in cooperation with Intrinsic Semiconductor in Kista. I want to take the chance to thank the people that have helped me complete this Master Thesis. First of all, I want to express my gratitude to Ian Ray for giving me the opportunity to perform my Master Thesis at Intrinsic Semiconductors and for being my supervisor. I also want to thank Jan Olsson, Chris Harris and Adriana Serban Craciunescu for their help during this Master Thesis. Last but not least I want to thank my family Maria, Julius and Indra for their love and support.


IV

Contents

1 INTRODUCTION.............................................................................................................. 1

1.1 INTRINSIC SEMICONDUCTOR AB ..................................................................................... 1 1.2 GOAL................................................................................................................................. 2

2 POWER AMPLIFIERS..................................................................................................... 3

2.1 MOSFET TRANSISTOR FUNDAMENTALS ........................................................................... 3 2.1.1 INTRODUCTION................................................................................................................ 3 2.1.2 BASIC STRUCTURE OF AN ENHANCEMENT N-CHANNEL MOSFET ................................... 3 2.1.3 PRINCIPLE OF OPERARATION........................................................................................... 4 2.1.4 DRAIN CURRENT IN THE SATURATION REGION................................................................ 5 2.1.5 DEVICE PARAMETERS ..................................................................................................... 6 2.2 LATERALLY DIFFUSED MOSFET.................................................................................... 7 2.3 AMPLIFIER CLASSES......................................................................................................... 8 2.3.1 CLASS A.......................................................................................................................... 8 2.3.2 CLASS B AND AB............................................................................................................ 9 2.3.3 CLASS C ........................................................................................................................ 10 2.3.4 BIASING A MOSFET FOR CLASS A ............................................................................... 10 2.4 AMPLIFIER CONFIGURATIONS ........................................................................................ 11 2.4.1 COMMON-SOURCE AMPLIFIERS...................................................................................... 11 2.4.2 COMMON-DRAIN AMPLIFIERS ........................................................................................ 12 2.4.3 COMMON-GATE AMPLIFIERS.......................................................................................... 13 2.5 POWER AMPLIFIERS ....................................................................................................... 14 2.6 IMPEDANCE MATCHING ................................................................................................. 15 2.6.1 WHY THE NEED FOR IMPEDANCE MATCHING ................................................................ 15 2.6.2 DIFFERENT APPROACHES .............................................................................................. 16 2.6.3 TWO-COMPONENT MATCHING NETWORK..................................................................... 17 2.7 DESIGNING BROADBAND AMPLIFIER ............................................................................. 18 2.8 TRANSISTORS FOR RF AMPLIFIER DESIGN .................................................................... 19 2.8.1 THE UNITY-GAIN FREQUENCY ...................................................................................... 20 2.8.2 POWER DISSIPATION ...................................................................................................... 20 2.8.3 DIMENSION AND DESIGN OF THE TRANSISTOR ............................................................... 20 2.8.4 DELIVERY STATUS......................................................................................................... 20 2.8.5 PRICE ............................................................................................................................ 21 2.9 DESIGNING A PRINTED CIRCUIT BOARD ......................................................................... 21

3 THE PD57006 TRANSISTOR ........................................................................................ 22

4 CLASS A POWER AMPLIFIER DESIGN................................................................... 23

4.1 THE CLASS A AMPLIFIER, BIAS OF THE TRANSISTOR .................................................... 23 4.1.1 I-V CURVE..................................................................................................................... 24 4.1.2 HEADROOM ................................................................................................................... 27 4.1.3 GATE CIRCUIT BIAS ....................................................................................................... 28


V

4.2 INPUT AND OUTPUT MATCHING CIRCUIT DESIGN........................................................... 28 4.2.1 INPUT MATCHING CIRCUIT ............................................................................................. 28 4.2.2 OUTPUT MATCHING CIRCUIT ......................................................................................... 31 4.3 THE FEEDBACK LOOP ..................................................................................................... 33 4.4 DC-BLOCKING CAPACITORS........................................................................................... 33 4.5 THE POWER AMPLIFIER IN CLASS A.............................................................................. 34

5 RESULTS AND DISCUSSION....................................................................................... 35

5.1 GAIN ................................................................................................................................ 35 5.2 THE POWER AMPLIFIER MEASUREMENTS ..................................................................... 35 5.3 BIAS ................................................................................................................................. 37 5.4 THE FINAL CIRCUIT ........................................................................................................ 38 5.5 OUTPUT POWER .............................................................................................................. 39

6 CONCLUSION................................................................................................................. 42

7 REFERENCES................................................................................................................. 43

APPENDIX A: GLOSSARY....................................................................................................I

APPENDIX B: PD57006 DATA SHEET .............................................................................. II


VI

List of Figures Figure 2-1 Basic structure of a n-channel MOSFET........................................................................................................................3 Figure 2-2 The drain currents relation to the gate voltage in a MOSFET........................................................................................4 Figure 2-3 I-V curve, showing the relationship between drain current and drain voltage in a MOSFET.........................................4 Figure 2-4 I-V curves for different gate voltages..............................................................................................................................5 Figure 2-5 The I-V curve of a MOSFET with its breakdown voltage ...............................................................................................5 Figure 2-6 Drain conductance .........................................................................................................................................................6 Figure 2-7 The transfer characteristics of a MOSFET.....................................................................................................................7 Figure 2-8 Structure of a LDMOSFET transistor .............................................................................................................................7 Figure 2-9 A class A Common-source amplifier ..............................................................................................................................8 Figure 2-10 Transistor driven out of the active region .....................................................................................................................9 Figure 2-11 Class B Common-source amplifier...............................................................................................................................9 Figure 2-12 Input voltage and output current waveforms ..............................................................................................................10 Figure 2-13 Schematic of a common-source configuration ...........................................................................................................11 Figure 2-14 Input and output signal of a common-source amplifier ..............................................................................................11 Figure 2-15 Common-drain amplifier .............................................................................................................................................12 Figure 2-16 Schematic of a common-gate amplifier ......................................................................................................................13 Figure 2-17 A class A power amplifier ...........................................................................................................................................14 Figure 2-18 Showing the input matching network between the source and transistor ..................................................................15 Figure 2-19 Impedance effect of series and shunt connections of L and C to a complex .............................................................16 load in the Smith Chart ...................................................................................................................................................................16 Figure 2-20 Showing the eight different two-component matching networks ................................................................................17 Figure 2-21 Smith Chart with the constant nodal quality factor lines.............................................................................................18 Figure 2-22 Feedback from the drain to the gate ..........................................................................................................................18 Figure 2-23 Frequency response with negative feedback.............................................................................................................19 Figure 2-24 Schematic of dc and ac feedback ..............................................................................................................................19 Figure 2-25 Showing how the gain decreases with increased frequency......................................................................................20 Figure 4-1 Characteristics of a class A amplifier ...........................................................................................................................23 Figure 4-2 I- curves with different gate-to-source voltages ...........................................................................................................24 Figure 4-3 I-V curves with load-line ...............................................................................................................................................24 Figure 4-4 Biaspoint taken from the I-V curves .............................................................................................................................25 Figure 4-5 Showing the components between the supply voltage and the drain ..........................................................................25 Figure 4-6 Showing the component values of the drain circuit ......................................................................................................26 Figure 4-7 Headroom for the chosen bias .....................................................................................................................................27 Figure 4-8 The complex conjugate to the input impedance is shown in the Smith Chart..............................................................28 Figure 4-9 Showing the movement in the constant reactance circle from

*inZ to CZ . This is equal to a capacitance in series

with the transistor shown to the right in the figure ..........................................................................................................................29 Figure 4-10 Moving form CY to LY in the constant-conductance circle which equals a inductor in shunt, shown to the right in

the figure.........................................................................................................................................................................................30 Figure 4-11 The complex conjugate to the output impedance inserted into the Smith Chart........................................................31 Figure 4-12 Movement in the constant-reactance circle from

*outZ to 2LZ which corresponds to a inductor in shunt shown to

the right...........................................................................................................................................................................................31 Figure 4-13 Movement in the constant-conductance circle from 2LY to 2CY which corresponds to a capacitance in shunt....32 Figure 4-14 Schematic of the class A power amplifier ..................................................................................................................34 Figure 5-1 Frequency sweep from 0 to 2GHz................................................................................................................................35 Figure 5-2 Ripple in the passband.................................................................................................................................................36 Figure 5-3 Frequency sweep from 0 to 6GHz to identify spurious IM products.............................................................................36 Figure 5-4 Picture of our final circuit ..............................................................................................................................................38 Figure 5-5 Closer picture of the final circuit ...................................................................................................................................38

List of tables Table 2-1 Amplifier classes_______________________________________________________________________________8 Table 5-1 I-V curve measurements________________________________________________________________________37 Table 5-2 Measurements done in Class A __________________________________________________________________39 Table 5-3 Measurements done in class C___________________________________________________________________41


1

1 Introduction The power amplifier (PA) is a main building block in communication systems. They are usually used as the last stage before the transmitting part, where their main task is to provide the transmitter with high power. PAs can be divided into different classes and configurations. The main classes are A, AB, B and C, there exists more but they are not as common as these four. The difference between the classes is how much of the signal is being amplified and is often given as the conduction angle, where o360 is the whole signal. Class A is the class that amplifies most of the signal o360 and is therefore the most linear but with the lowest efficiency. Low efficiency means that the amplifier will not deliver all power to the output, it will dissipate some power and this is usually done by heat. When using a class A amplifier which has a maximum efficiency of 50 %, and usually lower, at least half the power will dissipate in heat. If the amplifier is not able to handle that amount of power, it will in the worst case be destroyed. Due to the ever-increasing demands on electronics that can handle these high powers, high frequencies and high temperatures silicon (Si) has become almost obsolete. Silicon technology is already at its limits and does not have the capability to handle these demands. Silicon-carbide (SiC) on the other hand has the ability to handle the higher powers and does also work with higher frequencies and temperatures then Si. SiC also has a higher breakdown voltage, are more resistant to chemical attacks and radiation. This makes SiC very interesting for RF PAs. But with all this benefits there are some drawbacks. SiC wafers are more difficult to manufacture and costs more to fabricate. The major problem with manufacture has been micropipes. Micropipes comes from the growth of SiC and are holes in the crystal. However the technique of growing SiC is being improved, leading to that the micropipes and costs are decreasing. When designing PAs one often builds them with different blocks. For example as a first stage a class A amplifier can be used, this because there the need for high gain is desired and the efficiency can be neglected. As the second stage a class C amplifier can be used, here the designer can accept low gain with the benefit of high efficiency. Thus lower lost of power through dissipation. By building a PA like this high gain and high efficiency can be achieved.

1.1 Intrinsic Semiconductor AB Intrinsic Semiconductor AB are a privately owned manufacturer of wide bandgap semiconductor materials. They are located in Northern Virginia near Washington, DC. In October 2004 they acquired Advanced Microwave Device Solutions (AMDS) located in Kista, Sweden. AMDS was building their operation on a patented semiconductor device technology made for 3G Base Stations and many other advanced microwave applications. Intrinsic Semiconductor is today a company focusing on semiconductor materials and device technologies based on SiC and GaN. In Kista, one activity they are developing is a family of microwave power transistors. The latest of these transistors is a SiC Metal-Semiconductor Field Effect Transistor (MESFET), which will be used in many different applications from 500 MHz up to 3.5 GHz. In the best-case scenario, the MESFET will produce an output power of 50 W. In order to meet this specification the transistor needs to have up to 5 W of input power. For the sake of characterisation, a pre-amplifier is required to deliver the wanted input power in a linear and broadband fashion.


2

1.2 Goal The goal with this master thesis is to design a PA, which shall be used as a pre-amplifier to Intrinsics new MESFET. The pre-amplifier is to have the following specification:

• Work in the frequency rang from 500 MHz up to 1 GHz • Delivering at least 5 W of output power through the frequency range

• Have a gain of at least 15 dB

• Have an input and output impedance of 50 Ohm


3

2 Power Amplifiers In this chapter the fundamentals of the MOSFET, different classes and configurations of amplifiers together with some other relevant theory for the design of a power amplifier are presented

2.1 MOSFET Transistor fundamentals

2.1.1 Introduction The Metal-Oxide-Semiconductor Field Effect Transistor (MOSFET) is one of the FET family transistors. The FETs are voltage-controlled transistors in contrast to the Bipolar Junction Transistors (BJT) that are current-controlled. The term voltage-controlled means that in order to modulate the conductance of the device an electric field has to be applied to the gate. There are two different types of MOSFET devices, depletion mode (p-channel) and enhancement mode (n-channel). In this project the n-channel enhancement mode MOSFET was chosen.

2.1.2 Basic Structure of an enhancement n-channel MOSFET The basic structure of the enhancement n-channel MOSFET is shown in Figure 2-1. It is a p-silicon substrate into which two +n -regions have been diffused. The +n -regions are called drain and source. Between the source and the drain is a thin layer of silicon dioxide ( )2SiO . On top of the 2SiO layer there is a metal contact, called the gate.

Figure 2-1 Basic structure of a n-channel MOSFET As shown in the Figure 2-1 there exist no channel, and thus no current flow, between the source and drain when the gate voltage is zero. This makes this transistor a normally off device and it is off until the gate voltage exceeds the threshold voltage ( thV ) of the transistor.


4

2.1.3 Principle of Operaration The enhancement n-channel MOSFET are as mentioned a normally off device, which implies that when the gate voltage is below the thV it does not conduct any current. When the gate voltage is increases and exceeds the thV the device starts to conduct current, as can be seen in Figure 2-2.

Figure 2-2 The drain currents relation to the gate voltage in a MOSFET If a positiv gate voltage 1GV , which is greater then thV , is applied to the gate, a channel between the source and drain is induced. If then a small positive voltage DV is applied to the drain, the transistor will act as a constant resistor, seen in Figure 2-3. However if the drain voltage is increased further the resistance line will start to bend down from the initial resistance line this can also be seen in Figure 2-3. When the drain voltage reaches pinch-off the current changes from increasing according to Ohms law and becomes almost constant. The drain current has reached saturation.

Figure 2-3 I-V curve, showing the relationship between drain current and drain voltage in a MOSFET


5

If several different gate voltages are used, the curves will look like the curve in Figure 2-3, the only difference between them and the curve in Figure 2-3 is that they saturates at different drain currents. For example if a larger gate voltage than 1GV is applied the current will saturate at a higher value. In Figure 2-4 the I-V curves for different gate voltages are shown.

Figure 2-4 I-V curves for different gate voltages

2.1.4 Drain current in the Saturation Region When the drain current saturates it does not become constant, it increases slowly with the drain voltage, giving rise to a small drain conductance ( Dg ). If DV is increased even further the breakdown voltage is found. The breakdown voltage is the voltage at which an abrupt increase in the drain current occurs, as seen in Figure 2-5.

Figure 2-5 The I-V curve of a MOSFET with its breakdown voltage


6

2.1.5 Device Parameters

Two of the most important small-signal parameters are the drain conductance ( Dg ) and the transconductance ( mg ).

Drain conductance

The drain conductance is as explained in the Chapter 2.1.4 the small increas in drain current in the saturation region. It is the conductance on the output and this can be calculated using

DI∂ and DV∂ when GV is held constant.

constant=

∂∂

=GVD

DD V

Ig (2.1)1

Figure 2-6 Drain conductance

Transconductance

The transconductance is the change in drain current for a given change in gate-to-source voltage. To calculate the transconductance the drain voltage is to held constant while the drain current and gate voltage is to be varied as shown in equation 2.2. The transconductance is a very important parameter and is usually given in the data sheets. The data sheet for the transistor used in this project can be found in Appendix B.

constant=

∂∂

=DVG

Dm V

Ig (2.2)1

1 Man S. Tyagi (1991), Introduction to Semiconductor Materials and Devices


7

Figure 2-7 The transfer characteristics of a MOSFET

2.2 Laterally Diffused MOSFET The Laterally Diffused MOSFET (LDMOS) is a power MOSFET designed for low on-resistance and high blocking voltage. These features are obtained by creating a diffused p-type channel region in a low-doped n-type drain region. The low doping on the drain side results in a large depletion layer with high blocking voltage. The channel is short in the LDMOSFET and has therefore the capability to handle high currents. Because the LDMOS structure combines a short channel length with high breakdown voltage it is very useful for high power RF amplifiers. LDMOSFET is currently the device of choice for RF power amplifiers in base stations of wireless communications systems and also in radar systems. In Figure 2-8 the structure of a LDMOSFET can be seen.

Figure 2-8 Structure of a LDMOSFET transistor


8

2.3 Amplifier Classes Amplifiers are generally divided into different classes. The main classes are A, AB, B and C, there are more but they are not as common as these four. The difference between classes is how large the conduction angle is. The conduction angle is a measure of the time the transistor conducts current in a cycle. A conduction angle of 360 degrees equals that the transistor is conducting current during 100% of the input sine wave cycle. Large conduction angle produces a more linear representation of the input sine wave. Short conduction angles produce more efficiency of the amplifier, but less linearity. All classes can have the same topology but have different bias conditions. In Table 2-1 the conduction angle and the theoretical maximum efficiency for the four different classes are summarised.

Table 2-1 Amplifier classes Amplifier class Conduction angle

[degrees] Maximum theoretical

efficiency [%] Class A 360 50

Class AB 180 to 360 50 to 78.5 Class B 180 78.5 Class C 0 to 180 78.5 to 100

2.3.1 Class A The amplifier works in class A if it has a conduction angle of 360 degrees, conducts current for the whole input sine wave cycle. This means that the output signal is an amplified replica of the input signal, as shown in Figure 2-9.

Figure 2-9 A class A Common-source amplifier Class A amplifiers are usually biased at the centre of the load-line, as shown in Figure 2-9. By doing this, the transistor works in the active (saturation) region at as large time as possible and the signal is a linear function of the input signal. If the bias point Q is not centred the output signal runs the risk of being “clipped”. When a signal is being “clipped” it is outside the active region and will be lost, as shown in Figure 2-10.


9

Figure 2-10 Transistor driven out of the active region2 The class A amplifier has very good linearity but a poor efficiency. The advantage a class A amplifier has is that it amplifies the whole signal and therefore has little or no distortion. The big drawback is the efficiency, which is very low. The maximum efficiency of a class A amplifier is 50 %3 but in practice it is usually much less. The low efficiency is due to that the amplifier is biased at its maximum power consumption. This leads to that when there is no ac signal at the input the amplifier is consuming a very large power.

2.3.2 Class B and AB The amplifier operates in class B if it is biased at cut-off, with a conduction angle of 180 degrees. This means that only a half-cycle of the input signal is being amplified. At the other half-cycle the transistor is in the cut-off region and no current is flowing through the device.

Figure 2-11 Class B Common-source amplifier4

2 Thomas L. Floyd (2004), Electronic devices 7th edition 3 Thomas L. Floyd (2004), Electronic devices 7th edition 4 Thomas L. Floyd (2004), Electronic devices 7th edition


10

The advantage of a class B amplifier compared to the class A is that it is more efficient. In theory the maximum efficiency for a class B amplifier is 79 %5, but in reality is it lower. One disadvantage is that it only amplifies half the signal, so to amplify the whole signal two class B amplifiers has to be used. The problem is, when using two class B amplifiers, complementary push-pull, it is difficult to get linearity. Crossover distortion will occur when switching from positive to negative flank and vice versa. Crossover distortion is due to that when the dc gate voltage is zero, both transistors are off and before they will start to conduct the input signal must exceed the threshold voltage. Because of this, there will be a time-interval between positive and negative flank when neither transistor is conducting. A better way can be to use a class AB amplifier. It is a combination between class A and B, this means that the conduction angle is between 180 and 360 degrees. This makes it possible to overlap the positive and negative flank and reduce the distortion. The class AB has better efficiency than a class A and higher linearity than a class B.

2.3.3 Class C In the class C amplifier, amplification for less then 180 degrees is achieved. This means that the transistor does not conduct any current for more than a half-cycle. The result of this is a very high efficiency, up to 99%, but with a large distortion and a lower gain than for the other three classes. In Figure 2-12 an example of a class C amplifier is shown.

Figure 2-12 Input voltage and output current waveforms6

2.3.4 Biasing a MOSFET for class A

When biasing a MOSFET the load-line is inserted on top of the I-V curves. The load-line goes from ( )max0 == DD IV to ( )0== DD IbreakdownV . To get maximum output power the biaspoint needs to be in the middle of the graph between pinch-off and the breakdown voltage, so it can be allowed to swing the signal maximum up and down. 5 Thomas L. Floyd (2004), Electronic devices 7th edition 6 Thomas L. Floyd (2004), Electronic devices 7th edition


11

2.4 Amplifier configurations There are three different ways to connect the input and output signals to the transistor. Through common-source, common-drain or common-gate, which all have different behaviours and usefulness.

2.4.1 Common-source amplifiers In a common-source (CS) amplifier the input signal is applied at the gate and the output signal is taken out at the drain of the transistor, as shown in Figure 2-13.

Figure 2-13 Schematic of a common-source configuration7 In the common-source amplifier the input resistance is extremely high, ideally infinity. The high input resistance is caused by the insulated gate in the MOSFET. When looking at the output signal it can be seen that the phase is inverted, shown in Figure 2-14. This is due to that when the gate voltage is increased the drain current also increases leading to that more voltage is dropped over the drain resistance. Thus the drain-to-source voltage is decreased.

Figure 2-14 Input and output signal of a common-source amplifier

7 Thomas L. Floyd (2004), Electronic devices 7th edition


12

To find the voltage gain VA of a CS MOSFET the following equation can be used:

in

outv V

VA = (2.3)

md

ddv gI

RIA −= (2.4)

d

ddmV I

RIgA −= (2.5)

dmv RgA −= (2.6)

Where dR is the drain resistance in shunt with the load resistance and mg is the forward transconductance. The forward transconductance is as explained in Chapter 2.1.5.2 the change in drain current ( DI∆ ) for a given change in gate-to-source voltage ( GSV∆ ).

2.4.2 Common-drain amplifiers The input signal of a common-drain (CD) amplifier is as in the common-source applied to the gate but the output signal is taken out at the source. Therefore it is also called a source-follower. It has like the common-source a high input resistance but in contrast to the common-source is the phase of the output signal not inverted.

Figure 2-15 Common-drain amplifier8 The common-drain amplifier is usually used as the last stage in an amplifier circuit, as a buffer. This because it has a high input resistance and a gain of approximately one, which makes it very useful as a buffer. The voltage gain of a common-drain is:



13

in

outv V

VA = (2.7)

sgsmgs

sgsmv RVgV

RvgA

+= (2.8)

sm

smv Rg

RgA+

=1

(2.9)

where LSs RRR = . As seen in the equation 2.9 the voltage gain can’t be more than 1.

2.4.3 Common-gate amplifiers In a common-gate (CG) amplifier the input signal is not applied to the gate as in the common-source and common-drain. It is applied to the source and the output signal is taken at the drain like in the common-source. The common-gate does not invert the signal, as the common-source did. The input resistance in a common-gate amplifier is in contrast to the other two low. This is because the source is the input terminal. Due to the low input impedance this configuration is not as widely used as the other two, but it is used in certain high-frequency applications and also in high power applications.

Figure 2-16 Schematic of a common-gate amplifier9



14

The voltage gain for a common-gate is derived:

gs

d

in

outv V

VVVA == (2.10)

gs

ddv V

RIA = (2.11)

gs

dgsmV V

RVgA = (2.12)

dmv RgA = (2.13)

where LDd RRR = . This is as shown in Chapter 2.4.1 the same gain as for a common-source.

2.5 Power Amplifiers Power Amplifiers (PA) are large-signal amplifiers. This means that a large part of the load-line is used during signal operation. Power Amplifiers are normally used as the last stage of communication electronics to provide large enough signals to be transmitted. In Figure 2-17 the transfer function characteristics of a class A Power Amplifier is shown. As can be seen in the figure, the entire load-line is used.

Figure 2-17 A class A power amplifier


15

2.6 Impedance Matching As the frequency increases the wavelength of the electromagnetic waves become smaller and comparable with the dimensions of the discrete components. When this happens the components starts to deviate in their electric responses and the common electrical laws become obsolete. Then the voltage and current needs to be treated as waves and the circuit analyses has to be based on a distributed model.

2.6.1 Why the need for Impedance Matching To achieve maximum power transfer, the source needs to be matched to the load. This is easiest done using passive components to match the source and load to each other. Their function is not simply limited to matching for optimum power transfer they also helps to minimizing noise and reflection. The term impedance matching can be said to have the purpose to transform a given impedance to a more suited impedance.

Figure 2-18 Showing the input matching network between the source and transistor In Figure 2-18 the goal with the matching circuit is to transform the load impedance, here the transistor impedance ( )TZ , to the desired input impedance, here the source impedance ( )SZ . If maximum power transfer from source to load is required, the source impedance is to be equal to the complex conjugate of the load impedance.10 Thus the output impedance MZ of the matching network has to be equal to the complex conjugate of TZ , i.e. *

TM ZZ = .

10 Reinhold Ludwig and Pavel Bretchko (2000), RF Circuit Design


16

2.6.2 Different Approaches When designing a matching network there are different approaches:

• The analytically approach • The graphical approach (Smith Chart)

The analytical approach is a very precise but tedious way to design a matching network. It is suitable for small networks and for computer synthesis.11 The graphical approach is a faster, more intuitive and a simpler way to calculate the values of the passive components, but it is not as precise as the analytical approach. The graphical approach is done using a Smith Chart. The Smith Chart displays both the reflection coefficient and the impedance on the same graph. By moving from one impedance to another, for example from load to source impedance, the values of the component of the matching network can be calculated. The effect of adding a reactance in series or shunt results in a motion along either the constant-resistance circle or the constant-conductance circle, as can be seen in the Figure 2-19.

Figure 2-19 Impedance effect of series and shunt connections of L and C to a complex load in the Smith Chart12

11 Reinhold Ludwig and Pavel Bretchko (2000), RF Circuit Design 12

Reinhold Ludwig and Pavel Bretchko (2000), RF Circuit Design


17

2.6.3 Two-Component Matching Network The simplest way to impedance match a network is to use a two-component matching network, also known as a L-section. It consists of only two components as the name implies and is arranged in one of the eight configurations shown in Figure 2-20.

Figure 2-20 Showing the eight different two-component matching networks13 But it is not always best to have a matching circuit with as few components as possible. It is a simple solution but it can cause a big loss in bandwidth. A way to estimate the bandwidth is done using the nodal quality factor ( nQ ) of the matching network. When designing a matching network using Smith Chart one move from one node of the circuit to another. At each node the impedance can be expressed in terms of an equivalent series of impedance or admittance. At each node one can also find the nQ as the ratio of the absolute value of the reactance (X) to the corresponding resistance (R).

RX

Qn = (2.14)

or as the ratio of the absolute value of susceptance (B) to the conductance (G)

GB

Qn = (2.15)

When the nodal quality factor is known the loaded quality factor can be obtained by this formula:

2

nL

QQ = (2.14)14

and from it the bandwidth (BW) can be calculated

13 Reinhold Ludwig and Pavel Bretchko (2000), RF Circuit Design 14 Reinhold Ludwig and Pavel Bretchko (2000), RF Circuit Design


18

nL Qf

QfBW 00 2

== (2.15)15

where 0f is the resonance frequency. Thus the lower nodal quality factors the greater bandwidth. To simplify the matching network design process constant nQ -lines in the Smith Chart can be drawn. By keeping inside predefined lines the specified bandwidth can be achieved.

Figure 2-21 Smith Chart with the constant nodal quality factor lines

2.7 Designing broadband Amplifier When designing a power amplifier one of the most troublesome part is to make it wideband. There are some different ways to make an amplifier wideband. One of them is negative feedback, which was used in this project. When using negative feedback some of the output signal is coupled back to the input with the opposite phase to the input signal, thereby reducing it. This is shown in Figure 2-22.

Figure 2-22 Feedback from the drain to the gate

15 Reinhold Ludwig and Pavel Bretchko (2000), RF Circuit Design


19

By using negative feedback the amplifier produces a more flat gain response, is less sensitive to transistor-to-transistor parameter variations and a larger bandwidth is achieved. The disadvantage with negative feedback is that the power gain of the circuit is decreased. In Figure 2-23 it can be seen that the gain has decreased while the bandwidth has increased when negative feedback is used.

Figure 2-23 Frequency response with negative feedback The most general way to negative feedback a circuit is for dc to have a resistor in shunt with the transistor and in ac a resistor in series with an inductor in shunt with the transistor.

Figure 2-24 Schematic of dc and ac feedback

2.8 Transistors for RF amplifier design When looking for a transistor that has the appropriate parameters for the application one can be amazed over how difficult and time consuming it can be to find one. The first thing that needs to be established is what kind of application the amplifier will be used in. If it is to have large gain, is to be used as a digital component or if it is to be used in high power application. Depending on the application for the amplifier different kinds of transistors are to be preferred. For example if the gain is the most important parameter a BJT would be preferred and if it needs to handle large powers a LDMOSFET is a good choice. When the transistor model has been chosen the easiest way to find the transistor, that matches the requirements, is by searching the World Wide Web for large transistor manufacturers and their homepages.


20

When looking through the datasheets on the companies homepages some parameters a very obvious to look for, such as gain, output power etc. But some parameters are not that obvious but they need to be accounted for. Here are some very important parameters that have to be looked at.

2.8.1 The Unity-Gain frequency

The Unity-Gain frequency ( Tf ) is the frequency where the gain is equal to unity, as the name implies. When looking for a transistor one should count backward starting with Tf and the gain 1. Then by moving toward lower frequencies the gain increases with 6 dB, which is equal to a double in the voltage gain, for every time the frequency is halved. This is shown in the Figure 2-25.

Figure 2-25 Showing how the gain decreases with increased frequency

2.8.2 Power dissipation One of the most important parameters to look at is the power dissipation capability. The transistor must be able to handle the power dissipation or it will heat up and be destroyed. When choosing a transistor one should use a transistor with a power dissipation 2-3 times the specified maximum power.

2.8.3 Dimension and design of the transistor

An other important parameter to look for is the dimension and how the pins are placed on the transistor. This is important because the transistor has to fit in the application, example if the product is to be as small as possible the transistor should also be small. It can also be that the transistor needs to mount by hand and that some transistors don’t allow that, due to the design of it.

2.8.4 Delivery status

Also an important parameter to cheek, if the transistor is in storage or if it must be ordered from the manufacture. If it has to be ordered from the manufacturer it will delay the delivery time. Sometimes the transistors may not even be available at all, due to that it is old and has been taken out of production.


21

2.8.5 Price The price is a very important issue when it comes to choosing a transistor. This because everybody is trying to improve the costs of their products and therefore want the price of the components to be as low as possible. So when choosing between different transistors the less expensive is the one to be chosen.

2.9 Designing a printed circuit board The design of a printed circuit board (PCB) is usually done in a software program that is supplied by the manufacturer of the PCB. When the design is finished the file is sent to the manufacturer that then produces the PCB. When designing a PCB there are some issues that have to be taken into account.

• What is the allowed dimension of the PCB.

• What is the dimension of the components that will be mounted on the PCB. For standard components are these usually included, in the software programs, but for the other components the dimensions has to be found.

• Which components can be placed beside which and how close can they be mounted to

each other. If the wrong components are placed to close to each other they will interfere with each other.


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3 The PD57006 transistor The parameters of the PD57006 transistor that made it suite the projects power amplifier was:

• Frequency range: up to 960 MHz • Maximum output power: 6 W

• Maximum power dissipation: 20 W

• Gain: 15 dB @ 945 MHz • It is designed for high gain, broad band applications • It has a acceptable cost, approximately 160 SEK • The package of the transistor and its dimensions permit easy mounting on the PCB • The necessary technical information can be downloaded from STMicroelectronics

web site http://www.st.com/stonline/products/literature/ds/7236.pdf. The parameters are taken from the data sheet for the PD57006 transistor, which can be found in Appendix B.


23

4 Class A Power Amplifier design In this chapter the steps made to design the Power Amplifier are explained. How the bias circuit, impedance matching and the other solutions made to accomplish the specifications was made.

4.1 The class A amplifier, bias of the transistor The first thing when designing the amplifier were to bias the circuit. When biasing a MOSFET it is important to start with what value DSV can have.

Figure 4-1 Characteristics of a class A amplifier To ensure that the transistor does not go below the threshold voltage a voltage of 5 V are set as a lower voltage limit, as can be seen in Figure 4-1. Then the peak-to-peak voltage is added. The peak voltage can be calculated using the equation:

R

VP PEAK

2

2

= (4.1)

PRVPEAK 2= (4.2)

where the specified output power ( P) is 5 W and the output impedance (R) is approximately 6 Ohm, taken from the data sheet. When inserting these values in the equation the result is: 652 ××=PEAKV

VVPEAK 7.7≈ which gives a peak-to-peak voltage of 15.4 V. When adding this to the chosen lower voltage of 5 V a lowest value of the drain-to-source voltage is 20.4 V. Then by looking at the I-V


24

curves the bias point can be decided. An other criterion that has to be taken into account is

that DQI is not to be too large. The bias drain current is selected: ( )

2maxD

DQ

II = .

4.1.1 I-V curve

The output characteristics of the transistor were plotted using Matlab, shown in Figure 4-2, with the equations:

• ( )[ ]22 dsdsthgsD VVVVKI −−= , triode region • ( )2

thgsD VVKI −= , saturation region

Figure 4-2 I- curves with different gate-to-source voltages

On the I-V curve the load-line then was inserted, it goes from ( )AIVV DDS 1 0 == to ( )AIVV DDS 0 65 == .

Figure 4-3 I-V curves with load-line


25

In this case a drain current was to be approximately 0.5 A and the drain voltage to be approximately 25V. By looking at the I-V curve with the load-line on top of it an appropriate bias point can be chosen. In this case VVGSQ 6.4= , VVDSQ 5.25= and AIDQ 608.0= was proposed. This drain current is greater then specified but was proposed in order to keep DSV down.

Figure 4-4 Biaspoint taken from the I-V curves When the highest value of drain-to-source voltage and the peak-to-peak is known the lowest value the drain-to-source can have can be calculated, VVDS 1.104.155.25 =−= . By then setting VVD 8.17= when AIDQ 608.0= the desired swing can be allowed V 7.7± . To achieve this swing the drain impedance circuit ( DR in series with DL ) must be decided.

Figure 4-5 Showing the components between the supply voltage and the drain In this design the drain inductor ( )DL was chosen to 1 nH which makes sure no ac is taking this path. When this value is set the value of the drain resistance ( DR ) can be calculated. By knowing that the drop over DR in series with DL is 7.7 V, DR can be calculated using the equation with the frequency set to 950 MHz:


26

IVZ = (4.3)

A

VZ608.07.7

=

Ω= 66.12Z

LRZ ω+= (4.4)

( ) 9^1016^109502 −×××××+= πDRZ

97.5+= DRZ

97.5−= ZRD (4.5)

97.566.12 −=DR

Ω= 69.6DR By inserting these values in the circuit the desired voltage drop over DR and DL are achieved.

Figure 4-6 Showing the component values of the drain circuit


27

4.1.2 Headroom When calculating the values in the bias circuit it is useful to measure the headroom. Headroom is the difference between the lowest voltage allowed for the transistor and the lowest value the drain-to-source voltage has. The lowest value of the drain-to-source voltage is when the drain current is at its maximum. The lowest value the drain-to-source voltage has is: ( ) ( ) ( )DDDDSDS LRIVV ω+−= maxmin max

(4.6)

( ) ( )9^1016^10950269.615.25min −×××××+×−= πDSV

( )97.569.615.25(min) +×−=DSV

VVDS 84.12(min) =

(max)DI is taken from the data sheet, which can be found in Appendix B. In this design it is decided to not go below 5 V and it is therefore needed to subtract this from the calculated value. The headroom/difference between lowest allowed voltage and lowest value of the drain-to-source voltage is: ( ) allowedowestDS VVHeadroom _lmin −= (4.7)

VVHeadroom 584.12 −=

VHeadroom 84.7=

This is rather high headroom, which is good. If the headroom had been smaller there is always a possibility that the transistor will drive out of its operating region. But with this high headroom there is no risk of doing that.

Figure 4-7 Headroom for the chosen bias


28

4.1.3 Gate circuit bias The gate circuit’s main task is to provide an appropriate bias at the gate. When the bias was applied some extra components had to be inserted. This was done to prevent the ac to be able to take this path to the voltage source, which the ac would see as a ground. To prevent the ac to take this path an inductor was inserted into the circuit, which blocks the ac signal. Into the circuit a resistor was also inserted, to achieve the required bias.

4.2 Input and output matching circuit design The circuit was to be impedance matched to 50 Ohm on both the input and output.

4.2.1 Input matching circuit In the data sheet for the PD57006 the input impedance is given for the frequency range 925-960 MHz, the data sheet can be found in Appendix B. In this frequency range the impedance didn’t change much so the input impedance to be used for matching the source impedance to is selected to 326.2886.5 jZin −= . To impedance match the complex conjugate is needed, which is 326.2886.5* jZin += . Then it is inserted into the Smith Chart as shown in the Figure 4-8.

Figure 4-8 The complex conjugate to the input impedance is shown in the Smith Chart The goal is to move this point to the centre of the Smith Chart, which is the desired 50 Ohm. The movement along the constant-resistance circle in a counter clockwise direction responds to a capacitor in series. This is shown in Figure 4-9 where the movement from

326.2886.5* jZin += to 33.16886.5 jZC −= is shown.

*inZ


29

Figure 4-9 Showing the movement in the constant reactance circle from *inZ to CZ . This is equal to a capacitance in series

with the transistor shown to the right in the figure The value of the capacitor is derived: *

inCC ZZjX −= (4.8)

( ) ( )326.2886.533.16886.5 jjjX C +−−=

656.18jjX C −=

Cj

jX C ω1

= (4.9)

CX

Cω−

=1 (4.10)

656.18109502

16 ××××

=π

C

pFC 9≈ From point CZ the movement along the constant-conductance circle, counter clockwise, is achieved by a shunt inductor. By converting the impedance to admittance the calculation is made easier. The movement form 054.002.0 jYC += to 002.0 jYL −= is shown in Figure 4-10.


30

Figure 4-10 Moving form CY to LY in the constant-conductance circle which equals a inductor in shunt,

shown to the right in the figure The value of the inductor is calculated: CLL YYjB −= (4.11)

( ) ( )054.002.0002.0 jjjBL +−+=

054.0jjBL −=

Lj

jBL ω1

= (4.12)

LB

Lω−

=1 (4.13)

054.0109502

16 ××××

=π

L

nHL 3≈

The calculated values are not always the best ones. By inserting the values in a simulator it is possible to trim the impedance matching by changing the values of the components. By doing this, the inductor in this circuit was changed to 2.5nH, because it gave a better result.


31

4.2.2 Output matching circuit The output impedance matching is done in the same way as the input impedance matching, only now the output impedance is used.

Figure 4-11 The complex conjugate to the output impedance inserted into the Smith Chart The movement along the constant-reactance circle clockwise in the chart, is achieved by a series inductor.

Figure 4-12 Movement in the constant-reactance circle from *outZ to 2LZ which corresponds to

a inductor in shunt shown to the right


32

The value of the inductor is: *

22 outLL ZZjX −= (4.14)

( ) ( )999.5578.603.17578.62 jjjX L −−+=

029.232 jjX L =

22 LjjX L ω= (4.15)

ω

22

LXL = (4.16)

62 109502029.23

×××=

πL

nHL 86.32 ≈

From point 22 LL YZ = the movement to 2CY is achieved by the insertion of a shunt capacitor.

Figure 4-13 Movement in the constant-conductance circle from 2LY to 2CY which

corresponds to a capacitance in shunt


33

The value of the capacitor is: 222 LCC YYjB −= (4.17)

( ) ( )051.002.0002.02 jjjBL −−−=

051.02 jjBC =

22 CjjBC ω= (4.18)

ω

22

CBC = (4.19)

62 109502051.0

×××=

πC

pFC 5.82 =

After trimming the circuit with the help of the simulator the inductor was changed to 2.5nH and the capacitor to 9pF, which gave a better impedance matching.

4.3 The feedback loop To achieve a flat gain and a wider bandwidth a negative feedback loop was inserted. The feedback loop that was chosen is a resistor in series with an inductor, in shunt with the transistor. The goal with the negative feedback loop is to minimize the ripple in the passband and achieve a broader bandwidth with as little as possible decrease of the power gain. To achieve this only a small part of the signal is to be coupled back to the input, thus a large resistor and inductor is to be used. The simplest and best way to calculate the values of the components is to use a simulation program to do this. In this master thesis a simulation program that Ian Ray has written was used. By simulating the feedback loop with different values for the components, these values was chosen Ω= 100feedbackR and nHLfeedback 20= .

4.4 DC-blocking capacitors At the port where the signal is inserted and also where it is taken out two large capacitors was inserted. This was done to make sure that no dc from outside the circuit would affect the bias in the circuit and also to keep the dc inside the circuit. By using rather large capacitors the dc was kept out and only the ac was able to affect the circuit. Things that had to be aware of were that too large capacitor couldn’t be used because then they may attenuate the signal. Also an important factor to be aware of is that the signals not passing the capacitors will not be reflected back in the system, which is done by the use of filters. Typical values of the dc-blocking capacitors for microwave amplifiers are 1nF.


34

4.5 The Power Amplifier in class A When all the different parts of the PA were designed and the values of the components were calculated the final circuit could be put together, this is shown in Figure 4-14. The circuit consists of, from the left: A resistor, which symbolises the 50 Ohm that the input should be impedance matched to. Then moving to the right there is a blocking capacitor that prevents dc, from outside the circuit, to interfere with the bias. After that is the input matching circuit, which is a capacitor in shunt with an inductor. The values of the components are calculated in Chapter 4.2.1. The gate bias in the circuit has no values of the components, this is because it depends on what value GGV has. The PD57006 is the transistor used in the design and the parameters of it can be seen in Appendix: B. There is a negative feedback loop in shunt with the transistor, its task is to achieve a flat gain and broader bandwidth. To achieve the desired bias, when VVDD 5,25= , a drain impedance circuit is used, the values of the components are calculated in Chapter 4.1.1. Then there is the output matching circuit, which is an inductor in shunt with a capacitor, values of the components are calculated in Chapter 4.2.2. On the output there is a dc blocking capacitor as at the input, the output is also matched to 50 Ohm, which the 50 Ohm resistor symbolises.

Figure 4-14 Schematic of the class A power amplifier


35

5 Results and discussion In this chapter the amplifier test measurements are presented with a concluding discussion about the results.

5.1 Gain One of the most important parameters of an amplifier is the gain. How much gain does it have, over what frequency region does this gain stay constant, how much gain ripple is there in the passband and are there any unwanted spurious intermodulation products (IM) that have to be eliminated. When measuring the gain of the amplifier a Vector Network Analyser (VNA) was used. The VNA measures both the forward and reflected s-parameters, from both sides of the amplifier. These are termed s11, s21, s12 and s22.

5.2 The Power Amplifier measurements First the device performance without the feedback loop included, in order to measure the stand-alone performance of the transistor, was measured. Next the feedback loop was inserted to measure the improved constant gain bandwidth. The measurements revealed that the impedance matching could be improved. There was also poor coupling at higher microwave frequencies due to component placement and orientation which could be corrected so one further revision of CAD was required. Because this project was time limited and a new layout was required there was no chance to optimize the feedback loop, but it would be a simple matter to perform later when the layout is correct. In the Figure 5-1 the frequency response (s21) up to 2 GHz is shown to prove that approximately 15 dB of gain was obtainable from 700 MHz to 1100 MHz. The passband ripple could be improved to less than 1 dB by inserting the feedback loop, which also broaden the passband. The goal of the project was to have the amplifier to work in the 500 MHz to 1 GHz region, which is not accomplished in this measurement. But with some improvement/trimming of the matching networks and the feedback loop the specification is to be accomplished.

Figure 5-1 Frequency sweep from 0 to 2GHz


36

Figure 5-2 Ripple in the passband

To ensure that not too high IM products was produced the measurement region was increased. This is done in Figure 5-3 where the frequency range of 0-6GHz is shown.

Figure 5-3 Frequency sweep from 0 to 6GHz to identify spurious IM products Here some spurious products at 2.4, 3.0 and 3.6 GHz can be seen, some of which are only 15 dB below the carrier, which is not good. These were due to reflections from certain components. With further circuit improvements, such as steepening the filter, extra decoupling on the dc bias lines, improved impedance matching these spurious resonances can be minimised or even eliminated. This also reduces the passband ripple as some of the intermodulation products may fall within the passband. Consequently there may be an increase in the passband gain.


37

5.3 Bias Because the transistor data sheets did not show the I-V curves, they had to be measured to decide the load-line. This was done using following method. The Gate voltage was increased in 1.0 V steps from 0 V to 4 V, the Drain voltage was increased from 0 V to 24 V in steps and the Drain current was measured. This produced the I-V curve for each Gate bias voltages. From this the Gate bias voltage to be use could be determinated. The I-V measurements are shown in Table 5-1.

Table 5-1 I-V curve measurements

[ ]VVGS [ ]VVDD [ ]mAI D 0 20 0

1.0 20 0 2.0 20 0 3.0 0.2 10 3.0 0.6 20 3.0 1.0 20 3.0 2.6 30 3.0 20 30 4.0 0.2 10 4.0 0.5 20 4.0 0.9 30 4.0 1.3 40 4.0 1.8 50 4.0 2.2 60 4.0 2.6 70 4.0 3.0 80 4.0 3.4 90 4.0 3.9 100 4.0 4.3 110 4.0 4.7 120 4.0 5.1 130 4.0 5.6 140 4.0 6.0 150 4.0 6.5 160 4.0 6.8 170 4.0 7.3 180 4.0 7.7 190 4.0 8.1 200 4.0 10.5 250 4.0 13.2 300 4.0 24.1 350


38

5.4 The final circuit When doing the power measurements some improvements had been done to reduce the resonance in the circuit, which was not done fore the measurements presented in Chapters 5.2 and 5.3. The changes are not described in the report, this because it was done with the help from my supervisor who has several years of experience in the area which is needed to do this. And to explain why we choose to do it like we did would be very difficult. These changes will not change the results from the previous presented measurements that much, so it was decided not to do the measurements for them again to save time. In the Figure 5-4 and 5-5 the finished circuit can be seen.

Figure 5-4 Picture of our final circuit

Figure 5-5 Closer picture of the final circuit


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5.5 Output power To see how much output power the amplifier could produce measurements on the amplifier performance was done in different classes of operation (Class A and C). In all tests DDV was set to 28V. First the measurements were done in class A with the following parameters.

Table 5-2 Measurements done in Class A

Freq GSV DDV DQI DI inP outP Gain MHz V V mA mA dBm dBm dB 1000 3.9 28.0 340 340 0 +15 +15 1000 3.9 28.0 340 340 +10 +25 +15 1000 3.9 28.0 340 380 +20 +35 +15 1000 3.9 28.0 340 480 +25 +37 +12 1000 3.9 28.0 340 540 +28 +38 +10 1000 3.9 28.0 340 570 +30 +38.5 +8.5

900 3.9 28.0 340 340 0 +20 +20 900 3.9 28.0 340 350 +10 +30 +20 900 3.9 28.0 340 490 +20 +37.5 +17.5 900 3.9 28.0 340 590 +25 +39 +14 900 3.9 28.0 340 630 +30 +39.5 +9.5

800 3.9 28.0 340 340 0 +22 +22 800 3.9 28.0 340 370 +10 +30 +20 800 3.9 28.0 340 460 +15 +35 +20 800 3.9 28.0 340 630 +20 +37 +17 800 3.9 28.0 340 760 +25 +39 +14

In the Table 5-2 the measurements from a class A configuration with three different frequencies is shown. The measurements are done for three different frequencies because the amplifier is to be broad banded and therefore one needs to do the measurements for more than one frequency. In the table the most interestingly parameters are outP and the Gain which both are outgoing parameters. By increasing the input power the maximum output power

dBmPout 5.39(max) = ( ) Watts9≈ can be found. This is a very good result, the output power is even larger than the manufacture say their transistor can produce. The results prove that the requirements from the specifications (5 W) are by far achieved. In table 5-2 it can be seen that the amplifier has a rather flat gain up to an output power of

dBmPout 35+= . It can also be seen that the gain decreases when the amplifier is working in higher frequencies, which is expected for a class A amplifier. The gain of the amplifier is good, the requirement of 15 dB is reached with all the frequencies. The drawback with a class A amplifier is the poor efficiency, which can be calculated using the equation.


40

S

L

PPLoad

==PowerSupply

Power η (5.1)

where the load power ( )LP is:

L

outL R

VP

2

21

×= (5.2)

here

outL

out PR

V=

2

(5.3)

The Supply Power is: DDDS IVP ××= 2 (5.4) and if the values from the table 5-2 are inserted into the equation the efficiency of 6-12% is calculated.


41

Table 5-3 Measurements done in class C

Freq GSV DDV DQI DI inP outP Gain MHz V V mA mA dBm dBm dB 1100 2.4 28.0 0 10 +10 +3 -7 1100 2.4 28.0 0 30 +15 +14 -1 1100 2.4 28.0 0 100 +20 +23 +3 1100 2.4 28.0 0 240 +25 +30 +5 1100 2.4 28.0 0 480 +30 +36 +6 1100 2.4 28.0 0 600 +32 +38 +6 1100 2.4 28.0 0 640 +33 +38.5 +5.5 1100 2.4 28.0 0 740 +36 +39 +3

800 2.4 28.0 0 10 +0 +6 +6 800 2.4 28.0 0 30 +5 +16 +11 800 2.4 28.0 0 80 +10 +24 +14 800 2.4 28.0 0 190 +15 +30 +15 800 2.4 28.0 0 330 +20 +35 +15 800 2.4 28.0 0 460 +25 +37 +8 800 2.4 28.0 0 490 +30 +38 +8

Here the measurements is done for a class C configuration with two different frequencies, the lower 800 MHz and the higher 1100 MHz. By only looking at the results one can see that they have the classical characteristics of a class C amplifier. The classical characteristics of a class C amplifier are low gain at low output powers and improving when the output power is increased. The gain of the amplifier continues to improve until it levels of at a certain value, after that it begins to roll off. This can be seen in Figure 5-3 where the amplifier reaches a maximum gain (+6 dB) at an output power of +36 dBm for 1100 MHz and (+15 dB) at +30 dBm for 800 MHz and then the gain begins to roll off. In the table it can be seen that the gain is lower for a class C configuration than for class A, which it also should be. The benefit of using the amplifier in class C is that the efficiency is much higher, approximately over 75%, than it is for class A. Thus it doesn’t lose as much power in form of heat as it would when using it in a class A configuration. The class C amplifier is very useful to have as a second stage in a Power Amplifier configuration. It doesn’t deliver much gain, that is the first amplifier stage task, but it delivers high output power to a good efficiency. The need of sufficient high input power is achieved from the first stage in the Power Amplifier, the class A amplifier.


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6 Conclusion From the results it can be conclude that when using the PD57006 MOSFET transistor in a class A amplifier a very good gain for the specified frequency region can be achieved. Also a high output power (9 W) can be produced. The drawback when using the PD57006 in a class A configuration is that it has a poor efficiency (6-12%), thus lots of power is lost through heat in the amplifier. When losing large amounts of power one always runs the risk of overheating the transistor, which could destroy it. When using the PD57006 in a class C configuration good efficiency can be achieved, which implies a low loss of power in form of heat. Also a high output power (8 W) can be produced. The drawback with a class C configuration is the lower gain compared to a class A. In the upper frequency region 1100 MHz only 6 dB is achieved, this is not acceptable when the requirement is a minimum of 15 dB. The conclusion from this is that when using the PD57006 in either of these two configurations a Power Amplifier that satisfies the requirements can not be achieved. In class A it has too large power dissipation and in class C it has too low gain. But if they are used together, the class A amplifier as a first stage, to get the wanted gain, and the class C amplifier as a second stage to improve the efficiency a power amplifier that satisfies the requirements are achieved. By using the class C amplifier as the second stage the power dissipation is dramatically decreased compared to when using it in class A. By using a class A amplifier as the first stage the required gain is achieved. To conclude: When building a two stage Power Amplifier with the PD57006 transistor where a class A amplifier is the first stage and a class C amplifier is the second a power amplifier that satisfies the requirements is achieved and can be used as a pre-amplifier to Intrinsics new MESFET, which it was designed for.


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7 References Floyd, Thomas L. (2004) Electronic devices 7th edition Upper Sadle River, NJ: Prentice Hall ISBN 0-13-127827-4 Floyd, Thomas L. and Buchla, David (2002) Fundamentals of Analog Circuits 2nd edition Upper Saddle River, N.J.: Prentice Hall ISBN 0-13-060619-7 Ludwig, Reinhold and Bretchko, Pavel (2000) RF Circuit Design Upper Saddle River: Prentice Hall ISBN 0-13-095323-7 Edwards, T.C. and Steer, M.B. (2000) Foundations of interconnect and microstrip design 3 edition Chichester: Wiley ISBN 0-471-60701-0 Molin, Bengt (1993) Förstärkarteknik 3 edition Stockholm: Kompendieutgivningen ISBN 91-7582-142-7 Tyagi, Man S. (1991) Introduction to Semiconductor Materials and Devices New York: Wiley ISBN 0-471-60560-3 http://en.wikipedia.org/wiki/Semiconductor 2005 http://www.st.com/stonline/ 2005-04-12


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Appendix A: Glossary A – Ampere

VA – Voltage Gain BJT – Bipolar Junction Transistor CD – common-drain CG – common-gate CS – common-source dB – decibel F – Farad

Tf – Unity-Gain frequency FET – Field Effect transistor H – Henry LDMOS – Laterally Diffused Metal-Oxide-Semiconductor MOSFET – Metal-Oxide-Semiconductor Field Effect Transistor MESFET – Metal-Semiconductor Field Effect transistor P – Power

OUTP – Output power

DISSP – Power dissipation 21S – The through parameter in the Network Analyser measurements. The signal

is applied at port 1 and taken out at port 2. V – Volt W – Watt p – pico ( )1210− n – nano ( )910− µ – micro ( )610− m – milli ( )310− k – kilo ( )310 M – Mega ( )610 G – Giga ( )910


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Appendix B: PD57006 Data Sheet


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Ultra-Wideband Power Amplifier Design - DiVA portal

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