Research Article Simple BiCMOS CCCTA Design and Resistorless Analog Function...

Research ArticleSimple BiCMOS CCCTA Design and Resistorless AnalogFunction Realization

Worapong Tangsrirat

Faculty of Engineering, King Mongkut’s Institute of Technology Ladkrabang (KMITL), Ladkrabang, Bangkok 10520, Thailand

Correspondence should be addressed to Worapong Tangsrirat; [email protected]

Received 9 March 2014; Revised 2 June 2014; Accepted 18 June 2014; Published 16 July 2014

Academic Editor: Leopoldo Angrisani

Copyright © 2014 Worapong Tangsrirat. This is an open access article distributed under the Creative Commons AttributionLicense, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properlycited.

The simple realization of the current-controlled conveyor transconductance amplifier (CCCTA) in BiCMOS technology isintroduced.TheproposedBiCMOSCCCTArealization is based on the use of differential pair and basic currentmirror, which resultsin simple structure. Its characteristics, that is, parasitic resistance (𝑅

𝑥) and current transfer (𝑖

𝑜/𝑖𝑧), are also tunable electronically

by external bias currents. The realized circuit is suitable for fabrication using standard 0.35𝜇m BiCMOS technology. Some simpleand compact resistorless applications employing the proposed CCCTA as active elements are also suggested, which show that theircircuit characteristics with electronic controllability are obtained. PSPICE simulation results demonstrating the circuit behaviorsand confirming the theoretical analysis are performed.

1. Introduction

Since an introduction of the newly defined active buildingblock, namely, the current conveyor transconductance ampli-fier (CCTA), in 2005 [1], this device has been gaining anincreasing attention that led to a great number of analog func-tion circuits. Basically, the CCTA device can be realized bycascading the second-generation current conveyor with themultioutput transconductance amplifier in monolithic form.By combining the advantages of both circuit technologies, theCCTA possesses low power consumption, wide bandwidth,high dynamic range, and high-slew rate. Considering thesereasons, the CCTA is suitable for a class of analog signal pro-cessing which can process both current and voltage signals.Hence, a great number of numerous analog adjustable func-tions are available in open literature [1–5].

In 2008, the current-controlled conveyor transconduc-tance amplifier (CCCTA), which is a modified version ofthe CCTA, was introduced in bipolar technology [6]. Since itsintroduction, the CCCTA has been widely used in appli-cations for continuous-time signal processing [6–9]. Theparasitic resistance looking into the 𝑥-terminal (𝑅

𝑥) of the

circuit is used to advantage in current-controlled circuitparameter, because it is easily adjusted by an external biasing

current.This advantage allows the implementation of numer-ous electronically tunable circuits without requiring externalpassive resistors, which is especially important for integratedcircuit implementation.

In recent integrated circuit technology, there are two basictechnologies that are known as bipolar and CMOS technolo-gies. For the bipolar transistor technology, they have highertransconductance gain (𝑔

𝑚), low-noise performance, and

better high-frequency performance than their CMOS coun-terparts [10, 11]. On the other hand, the advantage of theCMOS technology includes high-input impedance level, lowpower dissipation, and small chip area.The circuit realized inBiCMOS technology will therefore provide the advantages ofboth technologies.

The aim of this work is to realize a current-controlledconveyor transconductance amplifier (CCCTA) structuresuitable for integration in BiCMOS technology.The proposedCCCTA has relatively simple structure, since it is composedof solely differential pairs and simple current mirrors. Theimportant circuit parameters, that is, parasitic resistance 𝑅

𝑥

and current transfer characteristic 𝑖𝑜/𝑖𝑧, can be adjusted elec-

tronically and linearly through the external bias currents.The characteristics of the proposedCCCTAare demonstratedby PSPICE simulation results using 0.35𝜇m BiCMOS real

Hindawi Publishing Corporatione Scientific World JournalVolume 2014, Article ID 423979, 7 pageshttp://dx.doi.org/10.1155/2014/423979

2 The Scientific World Journal

z

y

CCCTA

x

IA IB

iy

ix

�y

�x

±io

iz

�o±o±

�z

(a)

y 1

x z

iy

iz

ix

ix Rx

o±

±io±gm�z

(b)

Figure 1: The CCCTA: (a) circuit symbol and (b) its equivalent circuit.

y

x z

M10 M13

M4M3B

M3AM2M1 M5 M6M7

M12M11

M8

M9

o−o+

+V

−V

Q1 Q2Q3 Q4 Q5 Q6

io io

izix

IAIB

Figure 2: BiCMOS realization of the proposed CCCTA.

process parameters. The results show good agreement withthe expected values. Some application examples in realiz-ing resistorless analog function circuits using the proposedCCCTA as active building blocks are also given. The circuitsolutions with fewer components are realized in order todemonstrate the easy applicability of the proposed circuit andto obtain simple and compact circuit designs.

2. Basic Concept of the CCCTA

The CCCTA is conceptually a combination of second-gener-ation current-controlled conveyor (CCCII) and transconduc-tance amplifier. Its electrical symbol and equivalent circuitcan be shown in Figure 1. As shown, the CCCTA device con-sists of two input terminals (𝑦 and 𝑥) and two output termi-nals (𝑧 and 𝑜). The 𝑥-terminal has a parasitic serial resistance(𝑅𝑥), where its value usually depends on an external supplied

current. The 𝑦-terminal is the high-input impedance termi-nal, while the 𝑧- and 𝑜-terminals are two types of high-outputimpedance terminals. The property of the CCCTA can bedescribed by the following matrix:

[[[

[

𝑖𝑦

V𝑥

𝑖𝑧

𝑖𝑜±

]]]

]

=[[[

[

0 0 0 0

𝑅𝑥1 0 0

1 0 0 0

0 0 ±𝑔𝑚0

]]]

]

⋅[[[

[

𝑖𝑥

V𝑦

V𝑧

V𝑜±

]]]

]

, (1)

where 𝑅𝑥and 𝑔

𝑚are the finite parasitic resistance looking

into the 𝑥-terminal and the transconductance gain of the

CCCTA, respectively. Here, 𝑅𝑥and 𝑔

𝑚depend on the exter-

nal DC bias currents 𝐼𝐴and 𝐼𝐵, respectively.

3. Proposed BiCMOS CCCTA Realization

The schematic BiCMOS realization of the proposedCCCTA is shown in Figure 2. The circuit mainly consists ofsecond-generation current-controlled conveyor (CCCII) andtransconductance amplifier. It is designed by combiningbipolar and CMOS technologies in order to utilize the mainadvantages of each technology, that is, higher transcon-ductance, higher frequency, low power consumption, andsmall silicon area. The groups of transistors 𝑄

1-𝑄2, 𝑄3-𝑄4,

and 𝑄5-𝑄6, which are assumed to be well matched, act as

transconductance amplifiers to convert the voltage signal tothe current signal. The current mirroring has been achievedby simple current mirror circuits (𝑀

1–𝑀3), (𝑀

4-𝑀5),

(𝑀6-𝑀7), (𝑀

8–𝑀10), and (𝑀

11–𝑀13). Mirroring actions

between 𝑀1and 𝑀

2, 𝑀4and 𝑀

5, and 𝑀

6and 𝑀

7force

equal bias current in 𝑄1and 𝑄

2, 𝑄3and 𝑄

4, and 𝑄

5and

𝑄6, respectively. By applying the translinear principle to the

base-emitter voltages (VBE) of 𝑄1and 𝑄

2, the differential

input voltage (V𝑦− V𝑥) can be derived as

V𝑦− V𝑥= VBE1 − VBE2 = 𝑉𝑇 (ln

𝑖𝑐1

𝐼𝑆

− ln𝑖𝑐2

𝐼𝑆

) , (2)

where 𝑉𝑇≅ 26mV at 27∘C is the thermal voltage, 𝐼

𝑆is the

reverse saturation current, and 𝑖𝑐1and 𝑖𝑐2are, respectively, the

collector currents of transistors 𝑄1and 𝑄

2.

The Scientific World Journal 3

As shown in Figure 2, the relationship of the currentflowing through the 𝑥-terminal (𝑖

𝑥) is equal to

𝑖𝑥= 𝑖𝑐2− 𝑖𝑐1, (3)

where

𝑖𝑐1=

𝐼𝐴

1 + 𝑒−(V𝑦−V𝑥)/𝑉𝑇,

𝑖𝑐2=

𝐼𝐴

1 + 𝑒(V𝑦−V𝑥)/𝑉𝑇.

(4)

Substituting (4) into (3), the current 𝑖𝑥can be rewritten as

𝑖𝑥= 𝐼𝐴tanh(

V𝑥− V𝑦

2𝑉𝑇

) . (5)

For simplification, if we assume that (V𝑥−V𝑦) ≪ 2𝑉

𝑇, then

the term tanh(V𝑥− V𝑦/2𝑉𝑇) can be approximately reduced to

(V𝑥− V𝑦/2𝑉𝑇). Hence, (5) can also be given by

𝑖𝑥≅ 𝐼𝐴(V𝑥− V𝑦

2𝑉𝑇

) . (6)

From the above expression, the parasitic resistance lookinginto the 𝑥-terminal (𝑅

𝑥) of the CCCTA when the 𝑦-terminal

is connected to ground has been derived as

𝑅𝑥≅2𝑉𝑇

𝐼𝐴

. (7)

It should be noted from (7) that the resistance 𝑅𝑥is control-

lable electronically by adjusting the bias current 𝐼𝐴.

Similarly, the small-signal transconductance gain (𝑔𝑚) of

the CCCTA derived from transconductors 𝑄3-𝑄4(𝑄5-𝑄6)

can be expressed as [12]

𝑔𝑚=𝑖𝑜

V𝑧

=𝐼𝐵

2𝑉𝑇

. (8)

Also note that the 𝑔𝑚-value can be controlled electronically

and linearly by changing the 𝐼𝐵-value.

4. Simulation Results

To confirm the theoretical study, the proposed CCCTA struc-ture in Figure 2 has been simulated with PSPICE using stan-dard 0.35𝜇m BiCMOS process parameters. The circuit wasbiased with ±1 V supply voltages. The transistor aspect ratios(𝑊/𝐿 in 𝜇m/𝜇m) were chosen as 7/0.7 and 8.5/0.7 for allthe PMOS and NMOS transistors, respectively.

In Figure 3, the theoretical and simulated values of theparasitic resistance 𝑅

𝑥of the proposed CCCTA against the

biasing current 𝐼𝐴are plotted. In the plots, the biasing current

𝐼𝐴was adjusted from5𝜇A to 300𝜇A.As shown, the simulated

results agree well with the theory. Next, the DC current trans-fer behavior has been investigated. A DC sweep simulationhas been performed, to demonstrate the range where thecurrent through 𝑧-terminal is equal to the one applied to 𝑥-terminal. The resulting plots when 𝐼

𝐴= 50 𝜇A are shown

200 6040 80 120100 140 160 180 200 220 240 260 280 3000.1

1

10

TheorySimulated

Rx

(kΩ

)

IA (𝜇A)

Figure 3: Variation of 𝑅𝑥as a function of 𝐼

𝐴.

0 50 100

0

50

100

ixiz

−50

−50−100

−100

Curr

ent (𝜇

A)

ix (𝜇A)

Figure 4: Simulated DC current transfer characteristic between 𝑖𝑥

and 𝑖𝑧.

in Figure 4, which can observe that the output offset currentat the 𝑧-terminal was found to be approximately 300 nA.Figure 5 shows the current transfer characteristic from 𝑥-terminal to 𝑧-terminal as a function of frequency. One canmeasure from the simulation results that the −3 dB band-width of the current transfer 𝑖

𝑧/𝑖𝑥has a value of about

47MHz.Figure 6 shows the plots of both output currents 𝑖

𝑜+and

𝑖𝑜−

against the input voltage V𝑧. It can be measured from

the plots that the voltage V𝑧linearly converts into output

signal currents 𝑖𝑜+

and 𝑖𝑜−

with nonlinearity of less than 1%for the input voltage range of −50mV to 50mV. The simu-lated frequency responses of the transconductance gain 𝑔

𝑚

characteristic for three different values of 𝐼𝐵, that is, 𝐼

𝐵=

50 𝜇A, 100 𝜇A, and 150 𝜇A, are depicted in Figure 7. Theresults indicate that the bandwidth in order of megahertz


Table 1: Main performances of the CCCTA reported in [6] and the proposed one of Figure 2, for 𝐼𝐴= 50 𝜇A and 𝐼

𝐵= 100𝜇A.

Parameters CCCTA of [6] Proposed CCCTA of Figure 2Technology ALA400 bipolar 0.35 𝜇m BiCMOSSupply voltages ±1.5 V ±1 VPower dissipation 1.48mW 0.13mW−3 dB bandwidth for V

𝑥/V𝑦

105MHz 82MHz−3 dB bandwidth for 𝑖

𝑧/𝑖𝑥

34MHz 47MHz−3 dB bandwidth for 𝑖

𝑜±/V𝑧

30MHz 40MHz𝑅𝑥

260Ω (@𝐼𝐵1

= 50 𝜇A) 1.04 kΩ (@𝐼𝐴= 50 𝜇A)

𝑔𝑚

0.95mA/V–2.78mA/V(@𝐼𝐵2

= 50 𝜇A–150 𝜇A)0.925mA/V–2.64mA/V(@𝐼𝐵= 50 𝜇A–150 𝜇A)

Parasitic resistance at port 𝑦 (𝑅𝑦) 7.24MΩ 262 kΩ

Parasitic resistance at port 𝑧 (𝑅𝑧) 123.26 kΩ 80 kΩ

Parasitic resistance at port 𝑜± (𝑅𝑜±) 207.87 kΩ 740 kΩ

0

20

i z/i x

(dB)

10k 100 k 1M1k 10M 10G100M 1GFrequency (Hz)

−60

−40

−20

−80

Figure 5: Simulated frequency response of the current transfer 𝑖𝑧/𝑖𝑥

characteristic.

40

20

0

0 10 20 30 40 50−50 −40−40

−30 −20

−20

−10

�z (mV)

Curr

ent (𝜇

A)

io− io+

Figure 6: Simulated DC voltage-to-current transfer characteristicbetween V

𝑧and 𝑖𝑜±.

is achieved. In addition, the total power dissipation of thiscircuit was found to be <0.13mW. Table 1 compares themajorperformances between the proposed BiCMOS CCCTA ofFigure 2 and the previous bipolar CCCTA reported in [6].They were obtained with the bias currents 𝐼

𝐴= 50 𝜇A and

𝐼𝐵= 100 𝜇A.

3

2

1

010k 100 k 1M1k 10M 10G100M 1G

Frequency (Hz)

IB = 150𝜇A

IB = 100𝜇A

IB = 50𝜇Agm

(mA

/V)

Figure 7: Simulated frequency responses of the transconductancegain 𝑔

𝑚characteristic.

5. Resistorless Analog Function Realization

In order to underline the potential of the proposed CCCTA,two illustrative analog function circuits have been imple-mented and discussed in the following subsections. Thecircuits were realized based on the employment of the mini-mum number components, thereby reducing the total powerconsumptions. Also, they do not need any additional passiveresistor, which result in the integrable circuit design, as wellas simple and compact structures.

5.1. Current-Mode Universal Filter. As the first applicationexample, the proposed CCCTA was used to realize the resis-torless current-mode universal filter. A simple three-inputsingle-output (TISO) filter using a single CCCTA and onlytwo grounded capacitors is described. The configuration isshown in Figure 8. By straightforward analysis, the single-output current function realized by this configuration isfound to be

𝐼out (𝑠) =𝐷 (𝑠) 𝐼in3 − (𝑠𝐶1𝑅𝑥 + 1) 𝑔𝑚𝐼in2 + 𝑔𝑚𝐼in1

𝐷 (𝑠), (9)


z

x

CCCTA

y

iin1

C1

IA IBiin3

iout

iin2

o−

o+

C2

Figure 8: CCCTA-based TISO current-mode universal filter.

where

𝐷 (𝑠) = 𝑠2

𝑅𝑥𝐶1𝐶2+ 𝑠𝐶2+ 𝑔𝑚. (10)

From the above expressions, the following can be sum-marized.

(1) The LP response is obtained with 𝐼in1 = 𝐼in (an inputcurrent signal) and 𝐼in2 = 𝐼in3 = 0.

(2) The BP response is obtained with 𝐼in1 = 𝐼in2 = 𝐼in and𝐼in3 = 0.

(3) The HP response is obtained with 𝐼in2 = 𝐼in3 = 𝐼in,𝐼in1 = 0, and 𝐶2 = 𝑔𝑚𝑅𝑥𝐶1.

(4) The BS (bandstop) response is obtained with 𝐼in1 =𝐼in2 = 𝐼in3 = 𝐼in and 𝐶2 = 𝑔𝑚𝑅𝑥𝐶1.

(5) The AP (all pass) response is obtained with 𝐼in1 =𝐼in2 = 𝐼in3 = 𝐼in and 𝐶2 = 2𝑔𝑚𝑅𝑥𝐶1.

Clearly, the configuration of Figure 8 can be used as a three-input single-input current-mode universal filter that canrealize all the five standard types of the biquad filter functions.Also from (9) and (10), the natural angular frequency (𝜔

𝑜) and

bandwidth (BW) of the filter in all cases are given, respec-tively, by

𝜔𝑜= √

𝑔𝑚

𝑅𝑥𝐶1𝐶2

,

BW =1

𝑅𝑥𝐶1

.

(11)

It can be observed that we can tune the values of the filterparameters 𝜔

𝑜and BW by controlling 𝑔

𝑚and/or 𝑅

𝑥.

The TISO resistorless current-mode universal filter ofFigure 8 was also simulated. The active and passive compo-nent values have been chosen as 𝐼

𝐴= 𝐼𝐵= 50 𝜇A and 𝐶

1=

𝐶2= 100 pF, to obtain the filter responses with a natural ang-

ular frequency of 𝑓𝑜= 𝜔𝑜/2𝜋 ≅ 1.53MHz. The theory and

simulated frequency characteristics for LP, BP, HP, and BS areshown in Figures 9 and 10, respectively. Figure 11 also showsthe AP frequency characteristics of the filter in Figure 8 when𝐶1= 50 pF and 𝐶

2= 100 pF.

20

0BP

HP LP

Curr

ent g

ain

(dB)

−60

−40

−20

Frequency (Hz)

SimulatedTheory

10k 100 k 1M 10M 100M−80

Figure 9: Theory and simulated LP, BP, and HP current responsesfor the universal filter of Figure 8.

20

Curr

ent g

ain

(dB)

0

Frequency (Hz)

SimulatedTheory

10k 100 k 1M 10M 100M−60

−40

−20

(a)

Frequency (Hz)

100

Phas

e (de

g)

50

0

−50

−100100 k 1M 10M 100M

SimulatedTheory

10k

(b)

Figure 10: Theory and simulated BS frequency characteristics forthe filter of Figure 8: (a) gain responses and (b) phase responses.


100

0

5

10 0

Gai

n (d

B)

Phas

e (de

g)

10k 100 k 1M1k 10M 100M 1G−360

−270

−180

−90

Frequency (Hz)

−5

−10

SimulatedTheory

Figure 11: Theory and simulated AP frequency characteristics forthe filter of Figure 8.

z

yCCCTA

x 1

zxCCCTA

y 2

o−

o+

Zin

i1

�1

i2

�2C2

C1

IA1 IB1

IA2 IB2

Figure 12: CCCTA-based floating inductance simulator circuit.

5.2. Floating Inductance Simulator. Figure 12 represents thelossless floating inductance simulator circuit consisting oftwo CCCTAs and two grounded capacitors. Taking 𝑔

𝑚=

𝑔𝑚1

= 𝑔𝑚2

and 𝐶 = 𝐶1= 𝐶2, the input impedance of the

simulator is obtained as

𝑍in = 𝑠𝐿eq ≅𝑠 (𝑅𝑥1+ 𝑅𝑥2) 𝐶

𝑔𝑚

, (12)

where 𝑅𝑥𝑖

is the parasitic resistance 𝑅𝑥of the 𝑖th CCCTA

(𝑖 = 1, 2). It is obvious that the realized equivalent inductancevalue is found to be 𝐿eq = (𝑅

𝑥1+ 𝑅𝑥2)𝐶/𝑔𝑚, which is

electronically controllable by adjusting 𝑅𝑥𝑖and/or 𝑔

𝑚. Addi-

tionally, if V2= 0, a grounded inductance simulator can also

be realized from the configuration of Figure 12.To verify the performance of the derived inductance

simulator of Figure 12, the circuit was simulated and com-pared with the ideal inductor. For this purpose, the followingcomponent values were taken as 𝑅

𝑥1= 𝑅𝑥2= 2.6 kΩ (𝐼

𝐴1=

𝐼𝐴2≅ 20 𝜇A), 𝑔

𝑚1= 𝑔𝑚2= 0.96mA/V (𝐼

𝐵1= 𝐼𝐵2≅ 50 𝜇A),

and 𝐶 = 𝐶1= 𝐶2= 50 pF, which results in 𝐿eq ≅ 0.27mH.

The simulated voltage and current waveforms of the floatinginductance simulator circuit of Figure 12 when a 1MHz

0 1 2 3 4

0

50

100

Volta

ge (m

V)

−100

−50

0

50

100

−100

−50 Curr

ent (𝜇

A)

Time (𝜇s)

iL�L

Figure 13: Simulated voltage and current waveforms for the induc-tance simulator of Figure 12.

Phas

e (de

g)

Frequency (Hz)

300

100

200

0

100

1Phase

Magnitude10k

1M

Mag

nitu

de (Ω

)

10k 100 k 1M1k 10M 100M

SimulatedTheory

Figure 14: Theory and simulated frequency responses of theinductance simulator in Figure 12.

sinusoidal signal is applied are shown in Figure 13. From theresults, the phase shift between the current and voltageis about 93∘, which is in close correspondence with theexpected value equal to 90∘. Further, the frequency-depend-ent impedance of the simulator is shown in Figure 14. It maybe noted that the simulator operates correctly along the fre-quency range 10 kHz to 4MHz. Figure 15 also shows the fre-quency characteristics of the inductance simulator for threedifferent values of 𝑅

𝑥, where 𝑅

𝑥= 𝑅𝑥1= 𝑅𝑥2(𝐼𝐴= 𝐼𝐴1

=

𝐼𝐴2).The simulations were performed by varying𝑅

𝑥= 5.2 kΩ

(𝐼𝐴≅ 10 𝜇A), 𝑅

𝑥= 2.6 kΩ (𝐼

𝐴≅ 20 𝜇A), and 𝑅

𝑥= 1.04 kΩ

(𝐼𝐴≅ 50 𝜇A), to obtain 𝐿eq ≅ 0.54mH, 0.27mH, and

0.108mH, respectively.

6. Closing Remarks

In this paper, a simplified structure of the current-controlledconveyor transconductance amplifier (CCCTA) in BiCMOStechnology has been introduced and characterized. Thecircuit is capable of operating at ±1 V supply voltages andcan operate to a frequency of about 40MHz. The proposedCCCTA is implemented with standard 0.35𝜇m BiCMOSreal process parameters. Some resistorless circuit implemen-tations with minimum component count and the addedadvantage of electronic tuning property realizing from theproposed CCCTA are also given. The simulation results have


100

110k 100 k 1M1k 10M 100M

Frequency (Hz)

10k

1M

Mag

nitu

de (Ω

)

IA = 10𝜇AIA = 20𝜇AIA = 50𝜇A

Figure 15: Simulated magnitude responses of the inductance simu-lator in Figure 12 by varying 𝐼

𝐴.

been performed for the designed CCCTA and its applicationsto verify the theoretical analysis.

Conflict of Interests

The author declares that there is no conflict of interestsregarding the publication of this paper.

Acknowledgment

This work was supported in part by the Faculty of Engi-neering, KingMongkut’s Institute of Technology Ladkrabang(KMITL).

References

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Research Article Simple BiCMOS CCCTA Design and Resistorless Analog Function...

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