LECTURE NOTESvemu.org/uploads/lecture_notes/03_01_2020_65460003.pdfDepartment of Electrical and...

Subject code:15A02503 Power Electronics

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LECTURE NOTES

ON

POWER ELECTRONICS

III B. Tech I Semester (JNTUA-R15)

Dr.D.CHANDRA SEKHAR, M.TECH,P.hD,MISTE.

PROFESSOR, HOD,EEE.

Department of Electrical and Electronics Engineering VEMU INSTITUTE OF TECHNOLOGY::P.KOTHAKOTA

NEAR PAKALA, CHITTOOR-517112

(Approved by AICTE, New Delhi & Affiliated to JNTUA, Anantapuramu)


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SYLLABUS

UNIT I: POWER SEMI CONDUCTOR DEVICES

Semiconductor Power Diodes, Thyristors – Silicon Controlled Rectifiers (SCR’s) – TRIACs, GTOs -

Characteristics and Principles of Operation and other Thyristors – Classification of Switching Devices

Based on Frequency and Power Handling Capacity-BJT – Power Transistor - Power MOSFET – Power

IGBT -– Basic Theory of Operation of SCR – Static Characteristics – Turn On and Turn Off Methods-

Dynamic Characteristics of SCR - Two Transistor Analogy – Triggering Circuits––– Series and Parallel

Connections of SCR’s – Snubber Circuits – Specifications and Ratings of SCR’s, BJT, IGBT.

UNIT II: PHASE CONTROLLED CONVERTERS

Phase Control Technique – Single Phase Line Commutated Converters – Mid Point and Bridge

Connections – Half Controlled Converters, Fully Controlled Converters with Resistive, RL Loads and

RLE Load– Derivation of Average Load Voltage and Current – Line Commutated Inverters -Active and

Reactive Power Inputs to the Converters without and with Free Wheeling Diode, Effect of Source

Inductance – Numerical Problems. Three Phase Line Commutated Converters – Three Pulse and Six

Pulse Converters – Mid Point and Bridge Connections - Average Load Voltage with R and RL Loads –

Effect of Source Inductance–Dual Converters (Both Single Phase and Three Phase) - Waveforms –

Numerical Problems.

UNIT III: CHOPPERS AND REGULATORS

Commutation Circuits – Time Ratio Control and Current Limit Control Strategies – Step Down and Step

up Choppers Derivation of Load Voltage and Currents with R, RL and RLE Loads- Step Up Chopper –

Load Voltage Expression– Problems. Study of Buck, Boost and Buck-Boost regulators, buck regulator

TPS54160, hysteretic buck regulator e.g.LM3475, Switching Regulator and characteristics of standard

regulator ICs – TPS40200, TPS40210, TPS 7A4901, TPS7A8300

UNIT IV: INVERTERS

Inverters – Single Phase Inverter – Basic Series Inverter – Basic Parallel Capacitor Inverter Bridge

Inverter – Waveforms – Simple Forced Commutation Circuits for Bridge Inverters – Single Phase Half

and Full Bridge Inverters-Pulse Width Modulation Control-Harmonic Reduction Techniques-Voltage

Control Techniques for Inverters – Numerical Problems, Three Phase VSI in 1200 And 1800 Modes of

Conduction.

UNIT V: AC VOLTAGE CONTROLLERS & CYCLO CONVERTERS

AC Voltage Controllers – Single Phase Two SCR’s in Anti Parallel – With R and RL Loads – Modes of

Operation of TRIAC – TRIAC with R and RL Loads – Derivation of RMS Load Voltage, Current and

Power Factor Wave Forms – Firing Circuits -Numerical Problems - Thyristor Controlled Reactors;

Switched Capacitor Networks.

Cyclo Converters – Single Phase Mid Point Cycloconverters with Resistive and Inductive Load

(Principle of Operation only) – Bridge Configuration of Single Phase Cycloconverter (Principle of

Operation only) – Waveforms

.

TEXT BOOKS:

1. Power Electronics, M. D. Singh and K. B. Khanchandani, Mc Graw Hill Education (India) Pvt.

Ltd., 2nd Edition, 2007, 23rd Reprint 2015.

2. Power Electronics: Circuits, Devices and Applications, Muhammad H. Rashid, Pearson, 3rd

Edition, 2014, 2nd Impression 2015.

REFERENCE BOOKS:


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1. Power Electronics, K. R. Varmah, Chikku Abraham, CENGAGE Learning, 1st Edition, 2016.

2. Power Electronics, P. S. Bimbhra, Khanna Publishers, 2012.

3. Power Electronics: Devices, Circuits, and Industrial Applications, V. R. Moorthi, OXFORD

University Press, 1st Edition, 2005, 12th Impression 2012.


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UNIT-I

POWER SEMICONDUCTOR DEVICES


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Main power source

Power

Electronic

circuit

Load

POWER ELECTRONICS

The control of electric motor drives requires control of electric power. Power electronics have

eased the concept of power control. Power electronics signifies the word power electronics

and control or we can say the electronic that deal with power equipment for power control.

Ref signal

Power electronics based on the switching of power semiconductor devices. With the

development of power semiconductor technology, the power handling capabilities and

switching speed of power devices have been improved tremendously.

Power Semiconductor Devices

The first SCR was developed in late 1957. Power semiconductor devices are broadly

categorized into 3 types:

1. Power diodes (600V,4500A)

2. Transistors

3. Thyristors (10KV,300A,30MW)

Thyristor is a four layer three junction pnpn semiconductor switching device. It has 3

terminals these are anode, cathode and gate. SCRs are solid state device, so they are compact,

possess high reliability and have low loss.

Control

Circuit

Feedback Signal

Digital

Circuit


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SCR is made up of silicon, it act as a rectifier; it has very low resistance in the forward

direction and high resistance in the reverse direction. It is a unidirectional device.

Static V-I characteristics of a Thyristor

The circuit diagram for obtaining static V-I characteristics is as shown

Anode and cathode are connected to main source voltage through the load. The gate and

cathode are fed from source 𝐸𝑆.

A typical SCR V-I characteristic is as shown below:


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𝑉𝐵𝑂=Forward breakover voltage

𝑉𝐵𝑅=Reverse breakover voltage

𝐼𝑔=Gate current

𝑉𝑎=Anode voltage across the thyristor terminal A,K.

𝐼𝑎=Anode current

It can be inferred from the static V-I characteristic of SCR. SCR have 3 modes of

operation:

1. Reverse blocking mode

2. Forward blocking mode ( off state)

3. Forward conduction mode (on state)

1. Reverse Blocking Mode

When cathode of the thyristor is made positive with respect to anode with switch open

thyristor is reverse biased. Junctions 𝐽1 and 𝐽2 are reverse biased where junction 𝐽2 is

forward biased. The device behaves as if two diodes are connected in series with reverse

voltage applied across them.

• A small leakage current of the order of few mA only flows. As the thyristor is

reverse biased and in blocking mode. It is called as acting in reverse blocking

mode of operation.

• Now if the reverse voltage is increased, at a critical breakdown level called

reverse breakdown voltage 𝑉𝐵𝑅,an avalanche occurs at 𝐽1 and 𝐽3 and the reverse


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current increases rapidly. As a large current associated with 𝑉𝐵𝑅 and hence more

losses to the SCR.

This results in Thyristor damage as junction temperature may exceed its maximum

temperature rise.

2. Forward Blocking Mode

When anode is positive with respect to cathode, with gate circuit open, thyristor is said to

be forward biased.

Thus junction 𝐽1 and 𝐽3 are forward biased and 𝐽2 is reverse biased. As the forward voltage

is increases junction 𝐽2 will have an avalanche breakdown at a voltage called forward

breakover voltage𝑉𝐵𝑂. When forward voltage is less then 𝑉𝐵𝑂thyristor offers high

impedance. Thus a thyristor acts as an open switch in forward blocking mode.

3. Forward Conduction Mode

Here thyristor conducts current from anode to cathode with a very small voltage drop

across it. So a thyristor can be brought from forward blocking mode to forward

conducting mode:

1. By exceeding the forward breakover voltage.

2. By applying a gate pulse between gate and cathode.

During forward conduction mode of operation thyristor is in on state and behave like a

close switch. Voltage drop is of the order of 1 to 2mV. This small voltage drop is due to

ohmic drop across the four layers of the device.

Different turn ON methods for SCR

1. Forward voltage triggering

2. Gate triggering

3. 𝑑𝑣

triggering 𝑑𝑡

4. Light triggering

5. Temperature triggering

1. Forward voltage triggering


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A forward voltage is applied between anode and cathode with gate circuit open.

• Junction 𝐽1 and 𝐽3 is forward biased.

• Juntion 𝐽2 is reverse biased.

• As the anode to cathode voltage is increased breakdown of the reverse biased junction

𝐽2 occurs. This is known as avalanche breakdown and the voltage at which this

phenomena occurs is called forward breakover voltage.

• The conduction of current continues even if the anode cathode voltage reduces below

𝑉𝐵𝑂till 𝐼𝑎 will not go below𝐼ℎ. Where 𝐼ℎ is the holding current for the thyristor.

2. Gate triggering

This is the simplest, reliable and efficient method of firing the forward biased SCRs. First

SCR is forward biased. Then a positive gate voltage is applied between gate and cathode. In

practice the transition from OFF state to ON state by exceeding 𝑉𝐵𝑂 is never employed as it

may destroy the device. The magnitude of 𝑉𝐵𝑂, so forward breakover voltage is taken as final

voltage rating of the device during the design of SCR application.

First step is to choose a thyristor with forward breakover voltage (say 800V) higher than the

normal working voltage. The benefit is that the thyristor will be in blocking state with normal

working voltage applied across the anode and cathode with gate open. When we require the

turning ON of a SCR a positive gate voltage between gate and cathode is applied. The point

to be noted that cathode n- layer is heavily doped as compared to gate p-layer. So when gate

supply is given between gate and cathode gate p-layer is flooded with electron from cathode

n-layer. Now the thyristor is forward biased, so some of these electron reach junction 𝐽2 .As a

result width of 𝐽2 breaks down or conduction at 𝐽2 occur at a voltage less than 𝑉𝐵𝑂.As 𝐼𝑔

increases 𝑉𝐵𝑂 reduces which decreases then turn ON time. Another important point is

duration for which the gate current is applied should be more then turn ON time. This means


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that if the gate current is reduced to zero before the anode current reaches a minimum value

known as holding current, SCR can’t turn ON.

In this process power loss is less and also low applied voltage is required for triggering.

3. dv/dt triggering

This is a turning ON method but it may lead to destruction of SCR and so it must be avoided.

When SCR is forward biased, junction 𝐽1 and 𝐽3 are forward biased and junction 𝐽2 is reversed

biased so it behaves as if an insulator is place between two conducting plate. Here 𝐽1 and 𝐽3

acts as a conducting plate and 𝐽2 acts as an insulator. 𝐽2 is known as junction capacitor. So if we

increase the rate of change of forward voltage instead of increasing the magnitude of voltage.

Junction 𝐽2 breaks and starts conducting. A high value of changing current may

damage the SCR. So SCR may be protected from high𝑑𝑣. 𝑑𝑡

𝑞 = 𝑐𝑣

𝐼 = 𝑐 𝑑𝑣

4. Temperature triggering

𝑎

𝐼𝑎

𝑑𝑡

𝛼 𝑑𝑣

𝑑𝑡

During forward biased, 𝐽2 is reverse biased so a leakage forward current always associated

with SCR. Now as we know the leakage current is temperature dependant, so if we increase

the temperature the leakage current will also increase and heat dissipitation of junction

𝐽2occurs. When this heat reaches a sufficient value 𝐽2 will break and conduction starts.

Disadvantages


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This type of triggering causes local hot spot and may cause thermal run away of the device.

This triggering cannot be controlled easily.

It is very costly as protection is costly.

5. Light triggering

First a new recess niche is made in the inner p-layer. When this recess is irradiated, then free

charge carriers (electron and hole) are generated. Now if the intensity is increased above a

certain value then it leads to turn ON of SCR. Such SCR are known as Light activated SCR

(LASCR).

Some definitions:

Latching current

The latching current may be defined as the minimum value of anode current which at must

attain during turn ON process to maintain conduction even if gate signal is removed.

Holding current

It is the minimum value of anode current below which if it falls, the SCR will turn OFF.

Switching characteristics of thyristors

The time variation of voltage across the thyristor and current through it during turn on

and turn off process gives the dynamic or switching characteristic of SCR.

Switching characteristic during turn on

Turn on time

It is the time during which it changes from forward blocking state to ON state. Total turn

on time is divided into 3 intervals:

1. Delay time

2. Rise time

3. Spread time

Delay time

If 𝐼𝑔and 𝐼𝑎 represent the final value of gate current and anode current. Then the delay time

can be explained as time during which the gate current attains 0.9 𝐼𝑔 to the instant anode

current reaches 0.1 𝐼𝑔 or the anode current rises from forward leakage current to 0.1 𝐼𝑎.

1. Gate current 0.9 𝐼𝑔 to 0.1 𝐼𝑎.

2. Anode voltage falls from 𝑉𝑎 to 0.9𝑉𝑎.

3. Anode current rises from forward leakage current to 0.1 𝐼𝑎.


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Rise time (𝒕𝒓)

Time during which

1. Anode current rises from 0.1 𝐼𝑎 to 0.9 𝐼𝑎

2. Forward blocking voltage falls from 0.9𝑉𝑎 to 0.1𝑉𝑎. 𝑉𝑎 is the initial forward blocking

voltage.

Spread time (𝒕𝒑)

1. Time taken by the anode current to rise from 0.9𝐼𝑎 to𝐼𝑎.

2. Time for the forward voltage to fall from 0.1𝑉𝑜 to on state voltage drop of 1 to 1.5V.

During turn on, SCR is considered to be a charge controlled device. A certain amount

of charge is injected in the gate region to begin conduction. So higher the magnitude

of gate current it requires less time to inject the charges. Thus turn on time is reduced

by using large magnitude of gate current.

How the distribution of charge occurs?

As the gate current begins to flow from gate to cathode with the application of gate

signal. Gate current has a non uniform distribution of current density over the cathode

surface. Distribution of current density is much higher near the gate. The density decrease

as the distance from the gate increases. So anode current flows in a narrow region near

gate where gate current densities are highest. From the beginning of rise time the anode

current starts spreading itself. The anode current spread at a rate of 0.1mm/sec. The

spreading anode current requires some time if the rise time is not sufficient then the anode

current cannot spread over the entire region of cathode. Now a large anode current is

applied and also a large anode current flowing through the SCR. As a result turn on losses

is high. As these losses occur over a small conducting region so local hot spots may form

and it may damage the device.

Switching Characteristics During Turn Off

Thyristor turn off means it changed from ON to OFF state. Once thyristor is oON there is

no role of gate. As we know thyristor can be made turn OFF by reducing the anode

current below the latching current. Here we assume the latching current to be zero

ampere. If a forward voltage is applied across the SCR at the moment it reaches zero then

SCR will not be able to block this forward voltage. Because the charges trapped in the 4-

layer are still favourable for conduction and it may turn on the device. So to avoid such a

case, SCR is reverse biased for some time even if the anode current has reached to zero.

So now the turn off time can be different as the instant anode current becomes zero to the

instant when SCR regains its forward blocking capability.

𝑡𝑞 =𝑡𝑟𝑟+𝑡𝑞𝑟 Where,


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𝑡𝑞 is the turn off time,𝑡𝑟𝑟 is the reverse recovery time, 𝑡𝑞𝑟 is the gate recovery time

At 𝑡1 anode current is zero. Now anode current builds up in reverse direction with same 𝑑𝑣

slope. This is due to the presence of charge carriers in the four layers. The reverse 𝑑𝑡

recovery current removes the excess carriers from 𝐽1 and 𝐽3 between the instants 𝑡1 and𝑡3. At

instant 𝑡3 the end junction 𝐽1 and 𝐽3 is recovered. But 𝐽2 still has trapped charges which decay

due to recombination only so the reverse voltage has to be maintained for some more

time. The time taken for the recombination of charges between 𝑡3 and 𝑡4 is called gate

recovery time 𝑡𝑞𝑟. Junction 𝐽2 recovered and now a forward voltage can be applied across

SCR.

The turn off time is affected by:

1. Junction temperature

2. Magnitude of forward current 𝑑𝑖 during commutation. 𝑑𝑡

Turn off time decreases with the increase of magnitude of reverse applied voltage.

GTO(Gate turn off thyristor)


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A gate turn off thyristor is a pnpn device. In which it can be turned ON like an ordinary SCR

by a positive gate current. However it can be easily turned off by a negative gate pulse of

appropriate magnitude.

Conventional SCR are turned on by a positive gate signal but once the SCR is turned on gate

loses control over it. So to turn it off we require external commutation circuit. These

commutation circuits are bulky and costly. So due to these drawbacks GTO comes into

existence.

The salient features of GTO are:

1. GTO turned on like conventional SCR and is turned off by a negative gate signal of

sufficient magnitude.

2. It is a non latching device.

3. GTO reduces acoustic and electromagnetic noise.

It has high switching frequency and efficiency.

A gate turn off thyristor can turn on like an ordinary thyristor but it is turn off by negative

gate pulse of appropriate magnitude.

Disadvantage

The negative gate current required to turn off a GTO is quite large that is 20% to 30 % of

anode current

Advantage

It is compact and cost less

Switching performance


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1. For turning ON a GTO first TR1is turned on.

2. This in turn switches on TR2 so that a positive gate current pulse is applied to turn on the

GTO.

3. Thyristor 𝑇1 is used to apply a high peak negative gate current pulse.

Gate turn-on characteristics

1. The gate turn on characteristics is similar to a thyristor. Total turn on time consists of

delay time, rise time, spread time.

2. The turn on time can be reduced by increasing its forward gate current.


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GATE TURN OFF

Turn off time is different for SCR.Turn off characteristics is divied into 3 pd

1. Storage time

2. Fall time

3. Tail time

Tq=ts+tf+tt

At normal operating condition gto carries a steady state current.The turn off process starts as

soon as negative current is applied after t=0.

STORAGE TIME

During the storagepd the anode voltage and current remains constant.The gate current rises depending upon

the gate circuit impedance and gate applied voltage.The beginning of pd is as soon as negative gate current is

applied.The end of storage pd is marked by fall in anode current and rise in voltage,what we have to do is

remove the excess carriers.the excess carriers are removed by negative carriers.


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FALL TIME

After ts, anode current begins to fall rapidly and anode voltage starts rising.After falling to a certain value,then

anode current changes its rate to fall.this time is called fall time.

SPIKE IN VOLTAGE

During the time of storage and fall timethere is achange in voltage due to abrupt current change. TAIL TIME

During this time ,the anode current and voltage continues towards the turn off values.The transient overshoot is

due to the snubber parameter and voltage stabilizes to steady state value.


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THE TRIAC

As SCR is a unidirectional device,the conduction is from anode to cathode and not from

cathode to anode. It conducts in both direction.It is a bidirectional SCR with three terminal.

TRIAC=TRIODE+AC

Here it is considered to be two SCRS connected in anti parallel.As it conducts in both

direction so it is named as MT1,MT2 and gate G.

SALIENT FEATURES

1.Bi directional triode thyristor

2.TRIAC means triode that works on ac

3.It conduct in both direction

4. It is a controlled device


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5. Its operation is similar to two devices connected in anti parallel with common gate

connection.

6. It has 3 terminals MT1,MT2 and gate G

Its use is control of power in ac.

POWER BJT

Power BJT means a large voltage blocking in the OFF state and high current carrying capability in the ON state. In

most power application, base is the input terminal. Emitter is the common terminal.

Collector is the output terminal.

SIGNAL LEVEL OF BJT

n+ doped emitter layer ,doping of base is more than collector.Depletion layer exists more towards the collector

than emitter


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POWER BJT CONSTRUCTION

The maxium collector emitter voltage that can be sustained across the junction, when it is

carrying substantial collector current.

Vceo=maxium collectorand emitter voltage that can be sustain by the device.

Vcbo=collector base breakdown voltage with emitter open

PRIMARY BREAKDOWN

It is due to convention avalanche breakdown of the C-B junction and its associated large

flow of current.The thickness of the depletion region determines the breakdown voltage of

the transistor.The base thickness is made as small as possible,in order to have good

amplification capability. If the thickness is too small, the breakdown voltage is

compromised.So a compromise has to be made between the two.


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THE DOPING LEVELS-

1.The doping of the emitter layer is quite large.

2.The base doping is moderate.

3.n- region is lightly doped.

4.n+ region doping level is similar to emitter.

1.THICKNESS OF DRIFT REGION-

It determines the breakdown length of the transistor.

2.THE BASE THICKNES –

Small base thickness- good amplification capability

Too small base thickness- the breakdown voltage of the transistor has ti be compromised.

For a relatively thick base,the current gain will be relatively small.so it is increase the

gain.Monolithicesigns for darlington connected BJT pair have been deveploed.

SECONDARY BREAKDOWN

Secondary breakdown is due to large power disspation at localized site within the semi

conductor.

PHYSICS OF BJT OPERATION-

The transistor is assumed to operate in active region. There is no doped collector drift

region. It has importance only in switching operation, in active region of operation.

junction is forward biased and C-B junction is reverse biased. Electrons are injected

into base from the emitter. Holes are injected from base into the emitter.

QUASI SATURATION-

Intially we assume that, the transistor is in active region. Base current is allowed to increase

then lets see what happens.first collector rises in response to base current.So there is a

increase voltage drop across the collector load.So C-E voltage drops.

Because of increase in collector current, there is a increase in voltage in drift region. This

eventually reduces the reverse biased across the C-B junction.so n-p junction get

smaller, at some point the junction become forward bised. So now injection of holes from

base into collector drift region occurs. Charge neutrality requires the electron to be injected

in the drift region of the holes. From where these electron came. Since a large no of

electron is supplied to the C-B junction via injection from emitter and subsequent

diffusion across the base. As excess carrier build up in the drift region begins to occur

quasi saturation region is entered. As the injected carrires increase in the drift region is


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gradually shotred out and the voltage across the drift region drops. In quasi saturation the

drift region is not completely shorted out by high level injection.Hard saturation obtained

when excess carrier density reaches the n+ side.

During quasi saturation, the rate of the collector fall.Hard saturation occurs when excess

carriers have completely swept across the drift region .

THYRISTOR PROTECTION

OVER VOLTAGE PROTECTION

Over voltage occurring during the switching operation causes the failure of SCR.

INTERNAL OVERVOLTAGE

It is due to the operating condition of SCR.


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During the commutation of SCR ,when the anode current decays to zero anode current

reverses due to stored changes. First the reverse current rises to peak value, then reverse

current reduces abruptly with large 𝑑𝑖⁄𝑑𝑡. During series inductance of SCR large

transient large voltage i.e 𝐿 𝑑𝑖⁄𝑑𝑡. is generated.

EXTERNAL OVER VOLTAGE

This is due to external supply and load condition. This is because of

1. The interruption of current flow in an inductive circuit.

2. Lightening strokes on the lines feeding the thyristor systems.

Suppose a SCR converter is fed from a transformer, voltage transient occur when

transformer primary will energise or de-energised.

This overvoltages cause random turn ON of a SCR.

The effect of overvoltage is minimized using

1. RC circuits

2. Non linear resistor called voltage clamping device.

Voltage clamping device is a non linear resistor.It is connected between

cathode and anode of SCR. The resistance of voltage clamping device

decreases with increasing voltages. During normal working condition Voltage

clamping (V.C) device has high resistance, drawing only leakage current.

When voltage surge appears voltage clamping device offers a low resistance and it

create a virtual short circuit across the SCR. Hence voltage across SCR is

clamped to a safe value.


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When surge condition over voltage clamping device returns to high resistance

state.

e.g. of voltage clamping device

1.Seleniumthyrector diodes

2.Metal Oxide varistors

3.Avalanche diode supressors

OVER CURRENT PROTECTION

Long duration operation of SCR, during over current causes the

1.junction temp. of SCR to rise above the rated value,causing permanent

damage to device.

SCR is protected from overcurrent by using

1.Circuit breakers

2.Fast acting fuses

Proper co-ordination is essential because

1..fault current has to be interrupted before SCR gets damaged.

2.only faulty branches of the network has to be replaced.

In stiff supply network,source has negligible impedance.So in such system

the magnitude and rate of rise of current is not limited.Fault current hence

junction temp rises in a few miliseconds.

POINTS TO BE NOTED-

1. Proper coordination between fast acting fuse and thyristor is essential.

2. The fuse is always rated to carry marginal overload current over

definite period.

3. The peak let through current through SCR must be less than sub cycle

rating of the SCR.

4. The voltage across the fuse during arcing time is called arcing or

recovery voltage and is equal to sum of the source voltage and emf

induced in the circuit inductance during arcing time.

5. On abrupt interruption of fuse current, induce emf would be high, which

results in high arcing voltage.

Circuit Breaker (C.B)

has long tripping time. So it is used for protecting the device against continuous

overload current or against the surge current for long duration. In order that fuse

protects the thyristor realiably the 𝐼2𝑡 rating of fuse current must be less than that of

SCR.

ELECTRONIC CROWBAR PROTECTION


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For overcurrent protection of power converter using SCR, electronic crowbar are

used. It provide rapid isolation of power converter before any damage occurs.

HEAT PROTECTION- To protect the SCR

1. From the local spots

2. Temp rise

SCRs are mounted over heat sinks.

GATE PROTECTION-

Gate circuit should also be protected from

1. Overvoltages

2. Overcurrents Overvoltage across the gate circuit causes the false triggering of SCR


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Overcurrent raise the junction temperature. Overvoltage protection is by zener diode across

the gate circuit.

INSULATED GATE BIPOLAR TRANSISTOR(IGBT)-

BASIC CONSTRUCTION-

The n+ layer substrate at the drain in the power MOSFET is substituted by p+ layer substrate and called

as collector. When gate to emitter voltage is positive,n- channel is formed in the p- region.This n-

channel short circuit the n- and n+ layer and an electron movement in n channel cause hole injection

from p+subtrate layer to n- layer.


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POWER MOSFET

A power MOSFET has three terminal device. Arrow indicates the direction of current

flow. MOSFET is a voltage controlled device. The operation of MOSFET depends on flow

of majority carriers only.

(Circuit diagram) (circuit symbol)

Switching Characteristics:-

The switching characteristic is influenced by

1. Internal capacitance of the device.

2. Internal impedance of the gate drive circuit.

Total turn on time is divided into

1. Turn on delay time

2.Rise time

Turn on time is affected by impedance of gate drive source. During turn on delay time gate

to source voltage attends its threshold value 𝑉𝐺𝑆𝑇.

After 𝑡𝑑𝑛 and during rise time gate to source voltage rise to 𝑉𝐺𝑠𝑝, a voltage which is

sufficient to drive the MOSFET to ON state.

The turn off process is initiated by removing the gate to source voltage. Turn off time is

composed of turn off delay time to fall time.

Turn off delay time


DEPT.OF EEE VEMUIT

To turn off the MOSFET the input capacitance has to be discharged . During 𝑡𝑑𝑓 the input

capacitance discharge from 𝑉1to 𝑉𝐺𝑠𝑝. During 𝑡𝑓 , fall time ,the input capacitance discharges

from 𝑉𝐺𝑠𝑝 to 𝑉𝐺𝑆𝑇. During 𝑡𝑓 drain current falls from 𝐼𝐷 to zero.

So when 𝑉𝐺𝑠 ≤ 𝑉𝐺𝑆𝑇 , MOFSET turn off is complete.

Fig. Switching waveform of power MOSFET

Insulated Gate Bipolar Transistor (IGBT)

IGBT has high input impedance like MOFFSET and low on state power lose as in BJT.

IGBT Characteristics

Here the controlling parameter is gate emitter voltage As IGBT is a voltage controlled device.

When 𝑉𝐺𝐸 is less than 𝑉𝐺𝐸𝑇 that is gate emitter threshold voltage IGBT is in off state.


DEPT.OF EEE VEMUIT

Fig. a Fig. b. Fig. c

Fig. a (Circuit diagram for obtaining V-I characteristics) Fig. b (Static V-I

characteristics)

Fig. c (Transfer characteristic)

Switching characteristics: Figure below shows the turn ON and turn OFF characteristics of

IGBT

Turn on time


DEPT.OF EEE VEMUIT

Time between the instants forward blocking state to forward on -state .

Turn on time = Delay time + Rise time

Delay time = Time for collector emitter voltage fall from 𝑉𝐶𝐸 to 0.9𝑉𝐶𝐸

𝑉𝐶𝐸=Initial collector emitter voltage

𝑡𝑑𝑛=collector current to rise from initial leakage current to 0.1Ic

Ic= Final value of collector current

Rise time

Collector emitter voltage to fall from 0.9𝑉𝐶𝐸 to 0.1𝑉𝐶𝐸.

Ic to Ic

After 𝑡𝑜𝑛 the device is on state the device carries a steady current of Ic and the collector

emitter voltage falls to a small value called conduction drop 𝑉𝐶𝐸𝑆.

Turn off time

1) Delay time 𝑡𝑑𝑓

2) Initial fall time 𝑡𝑓1

3) Final fall time 𝑡𝑓2

𝑡𝑂𝑓𝑓 =𝑡𝑑𝑓

+ 𝑡𝑓1+ 𝑡𝑓2

𝑡𝑑𝑓 = Time during which the gate emitter voltage falls to the threshold value 𝑉𝐺𝐸𝑇.

Collector current falls from Ic to 0.9Ic at the end of the 𝑡𝑑𝑓 collector emitter voltage begins to

rise.

Turn off time = Collector current falls from 90% to 20% of its initial value Ic OR The time

during which collector emitter voltage rise from 𝑉𝐶𝐸 to 0.1𝑉𝐶𝐸.

𝑡𝑓2 = collector current falls from20% to 10% of Ic.

During this collector emitter voltage rise 0.1𝑉𝐶𝐸 to final value of 𝑉𝐶𝐸.

Series and parallel operation of SCR

SCR are connected in series for h.v demand and in parallel for fulfilling high current

demand. Sting efficiency can be defined as measure of the degree of utilization on SCRs in a

string.

String efficiency < 1.

Derating factor (DRF)


DEPT.OF EEE VEMUIT

1 – string efficiency.

If DRF more then

no. of SCRs will more, so string is more reliable.

Let the rated blocking voltage of the string of a series connected SCR is 2𝑉1 as shown in the

figure below, But in the string two SCRs are supplied a maximum voltage of 𝑉1+𝑉2.

Significance of string efficiency .

𝜂 = 𝑉1 + 𝑉2

2𝑉1

Two SCRs are have same forward blocking voltage ,When system voltage is more then the

voltage rating of a single SCR. SCRs are connected in series in a string.

There is a inherent variation in characteristics. So voltage shared by each SCR may not be

equal. Suppose, SCR1 leakage resistance > SCR2 leakage resistance. For same leakage

current 𝐼0 in the series connected SCRs. For same leakage current SCR1 supports a voltage

𝑉1 , SCR2 supports a voltage 𝑉2,

So string 𝜂 for two SCRs = 𝑉1+𝑉2 = 1(1 + 𝑉

2) < 1 .

2𝑉2 2 𝑉1

So, 𝑉1 > 𝑉2,

The above operation is when SCRs are not turned ON. But in steady state of operation , A

uniform voltage distribution in the state can be achieved by connect a suitable resistance

across each SCRs , so that parallel combination have same resistance.

But this is a cumbersome work. During steady state operation we connect same value of

shunt resistance across each SCRs. This shunt resistance is called state equalizing circuit.

Suppose,


DEPT.OF EEE VEMUIT

Let SCR1 has lower leakage current 𝐼𝑏𝑚𝑛 , It will block a voltage comparatively larger than

other SCRs.

Voltage across SCR1 is 𝑉𝑏𝑚 =𝐼1𝑅.

Voltage across (n-1)SCR is (n-1) 𝐼2R, so the voltage equation for the series circuit is

Vs = 𝐼1𝑅+ (𝑛 − 1)𝐼2𝑅 = 𝑉𝑏𝑚+ (n-1)R (𝐼 − 𝐼𝑏𝑚𝑥 )

As 𝐼1 = 𝐼 − 𝐼𝑏𝑚𝑛

𝐼2= 𝐼 − 𝐼𝑏𝑚𝑥

So, Vs = 𝑉𝑏𝑚+ (n-1)R [𝐼1 − (𝐼𝑏𝑚𝑥 − 𝐼𝑏𝑚𝑛)] If

Δ𝐼𝑏 = 𝐼𝑏𝑚𝑥 − 𝐼𝑏𝑚𝑛

Then Vs = 𝑉𝑏𝑚+(n-1)R (𝐼1 − Δ𝐼𝑏)

Vs = 𝑉𝑏𝑚+ (n-1)R 𝐼1-(n-1)R Δ𝐼𝑏

R𝐼1= 𝑉𝑏𝑚

So, Vs = 𝑉𝑏𝑚+ (n-1) 𝑉𝑏𝑚 -(n-1)R Δ𝐼𝑏

= n 𝑉𝑏𝑚- (n-1)R Δ𝐼𝑏

⇒R = 𝑛𝑉𝑏𝑚−𝑉𝑆

(n−1)Δ𝐼𝑏

SCR data sheet usually contain only maximum blocking current , 𝐼𝑏𝑚𝑥

so we assume 𝐼𝑏𝑚𝑛 = 0

So Δ𝐼𝑏=𝐼𝑏𝑚𝑥

So the value of R calculated is low than actually required.

SCRs having unequal dynamic characteristics:


DEPT.OF EEE VEMUIT

It may occur that SCRS may have unequal dynamic characteristics so the voltage

distribution across the SCR may be unequal during the transient condition.

SCR 1 and SCR 2 have different dynamic characteristics. Turn ON time of SCR 2 is more

than SCR 1 by time Δ𝑡𝑑 .

As string voltage is 𝑉𝑆 so voltage shared by each SCRs be 𝑉𝑆/2. Now both are gated at same

time so SCR 1 will turn ON at 𝑡1 its voltage fall nearly to zero so the voltage shared by SCR

2 will be the string voltage if the break over voltage of SCR 2 is less than 𝑉𝑆 then SCR 2 will

turn ON .

* In case 𝑉𝑆 is less than the breakoverer voltage, SCR 2 will turn ON at instant 2. SCR 1

assumed to have less turn off 𝑡𝑞1 time then SCR 2, so 𝑡𝑞1

< 𝑡𝑞2 . At 𝑡2 SCR 1 has recovered

while SCR 2 is developing recovery voltage at 𝑡1 both are developing different reverse

recovery voltage.

At 𝑡2 SCR 1 has recovered while SCR2 is developing reverse recovery voltage .

Conclusion :

* Series connected SCR develop different voltages during turn ON and turn OFF

process. Till now we connect a simple resistor across the diode for static voltage equalizing

circuit .

* During turn ON and turn OFF capacitance of reverse biased junction determine the

voltage distribution across SCRs in a series connected string . As reverse biased junction

have different capacitance called self capacitance , the voltage distribution during turn ON

and turn Off process would be different.


DEPT.OF EEE VEMUIT

* Under transient condition equal voltage distribution can be achieved by employing

shunt capacitance as this shunt capacitance has the effect of that the resultant of shunt and

self capacitance tend to be equal. The capacitor is used to limits the dv/dt across the SCR

during forward blocking state. When this SCR turned ON capacitor discharges heavy current

through the SCR . The discharge current spike is limited by damping resistor 𝑅𝑐 . 𝑅𝑐 also

damps out high frequency oscilation that may arise due to series combination of 𝑅𝑐 ,C and

series inductor . 𝑅𝑐 & C are called dynamic equalizing circuit

Diode D is used during forward biased condition for more effective charging of the capacitor.

During capacitor discharge 𝑅𝑐 comes into action for limiting current spike and rate of change

of current di/dt .

The R, 𝑅𝑐 & C component also provide path to flow reverse recovery current. When one SCR

regain its voltage blocking capability. The flow of reverse recovery current is necessary as it

facilitates the turning OFF process of series connected SCR string. So C is necessary for both

during turn ON and turn OFF process. But the voltage unbalance during turn OFF time is

more predominant then turn ON time. So choice of C is based on reverse recovery

characteristic of SCR .


DEPT.OF EEE VEMUIT

2

SCR 1 has short recovery time as compared to SCR 2. Δ𝑄 is the difference in reverse

recovery charges of two SCR 1 and SCR 2. Now we assume the SCR 1 recovers fast . i.e it

goes into blocking state so charge Δ𝑄 can pass through C . The voltage induced by 𝑐1 is

𝛥𝑄/C , where is no voltage induced across 𝐶2 .

The difference in voltage to which the two shunt capacitor are charged is 𝛥𝑄/C .

Now thyristor with least recovery time will share the highest transient

voltage say 𝑉𝑏𝑚,

So, 𝑉𝑏𝑚 - 𝑉2 = Δ𝑄/C

So, 𝑉2= 𝑉𝑏𝑚 - Δ𝑄/C

As 𝑉1 = 𝑉𝑏𝑚

𝑉𝑆 = 𝑉1+𝑉2

= 𝑉𝑏𝑚+(𝑉𝑏𝑚 - Δ𝑄/C)

𝑉𝑆 = 2𝑉𝑏𝑚-Δ𝑄/C

⇒ 1 (𝑉 + Δ

𝑄) = 𝑉

2 𝑠 𝐶 𝑏𝑚

⇒ 𝑉2= 𝑉𝑏𝑚 - Δ𝑄/C

1[𝑉𝑠 - Δ𝑄/C]

Now suppose that there are n series SCRs in a string.


DEPT.OF EEE VEMUIT

𝑛

Let us assume that if top SCR has similar to characteristic SCR 1. Then SCR 1 would

support a voltage 𝑉𝑏𝑚

* If the remaining (n-1) SCR has characteristic that of SCR 2 .Then SCR 1

would recover first and support a voltage 𝑉𝑏𝑚 . The charge (n-1) Δ𝑄 from the remaining (n -

1) SCR would pass through C.

𝑉1 = 𝑉𝑏𝑚

𝑉2 = 𝑉𝑏𝑚 - Δ𝑄/C

Voltage across (n-1) slow thyristors

𝑉 = (n-1) (𝑉𝑏𝑚 - Δ𝑄/C)

So, 𝑉𝑆 = V1+(n-1) 𝑉2

= 𝑉𝑏𝑚 + (n-1) (𝑉𝑏𝑚 - Δ𝑄/C)

By simplifing we get ,

Parallel operation:

𝑉𝑏𝑚 = 1 [𝑉𝑠+(n-1) Δ𝑄/C ]

C =[ (n-1) Δ𝑄/( n𝑉𝑏𝑚 -𝑉𝑆)

𝑉2 = (𝑉𝑆 - Δ𝑄/C )/ n .

When current required by the load is more than the rated current of single thyristor ,

SCRs are connected in parallel in a string .


DEPT.OF EEE VEMUIT

For equal sharing of current, SCRs must have same 𝑉 − 𝐼characteristics during forward

conduction. 𝑉𝑇 across them must be same. For same 𝑉𝑇 , SCR 1 share 𝐼1and SCR 2 share I2 .

If 𝐼1 is the rated current

𝐼2 < 𝐼1

The total current 𝐼1+𝐼2 and not rated current 2𝐼1.Type equation here.

Thus string efficiency ,

𝐼1+𝐼𝐼2 =

1 (1 +

𝐼𝐼2)

2𝐼1 2 𝐼1

Middle conductor will have more inductance as compared to other two nearby conductor. As

a result less current flow through the middle conductor. Another method is by magnetic

coupling.

Thyristor gate characteristics:-

𝑉𝑔 = +ve gate to cathode voltage.

𝐼𝑔= +ve gate to cathode current.

As the gate cathode characteristic of a thyrister is a p-n junction, gate characteristic of the

device is similar to diode.

Curve 1 the lowest voltage value s that must be applied to turn on the

SCR.


DEPT.OF EEE VEMUIT

Curve 2 highest possible voltage values that can be safely applied to get circuit.

𝑉gm= Maximum limit for gate voltage .

𝐼gm= Maximum imilt for gate current.

𝑃gav = Rated gate power dissipation for each SCR.

These limits should not be crossed in order to avoid the permanent damage of the

device junction 𝐽3.

OY = Minimum limit of gate voltage to turn ON .

OX = minimum limit of gate current to turn ON.

If 𝑉gm, 𝐼gm, 𝑃gav are exceeded the thyristor will damage so the preferred gate drive area of

SCR is bcdefghb.

oa = The non triggering gate voltage , If firing circuit generates +ve gate

signal prior to the desired instant of triggering the SCR.It should be ensured that this un

wanted signal should be less than the non –triggering voltage oa.

𝐸𝑆 = 𝑉𝑔 + 𝐼𝑔𝑅𝑆

𝐸𝑆 =Gate source voltage

𝑉𝑔 = Gate cathode voltage

𝐼𝑔 = Gate current

𝑅𝑆 = Gate source resistance

𝑅𝑆 = The internal resistance of the trigger source

𝑅1 is connected across the gate cathode terminal, which provides an easy path to the flow of

leakage current between SCR terminal. If 𝐼𝑔𝑚𝑛 , 𝑉gmn are the minimum gate current and gate

voltage to turn ON the SCR.

𝐸𝑆 = (𝐼𝑔𝑚𝑛+ 𝑉gmn/ 𝑅1) 𝑅𝑆 + 𝑉𝑔𝑚𝑛


DEPT.OF EEE VEMUIT

UNIT-II

PHASE CONTROLLED CONVERTERS


DEPT.OF EEE VEMUIT

RECTIFIER

Rectifier are used to convert A.C to D.C supply.

Rectifiers can be classified as single phase rectifier and three phase rectifier. Single phase

rectifier are classified as 1-Փ half wave and 1-Փ full wave rectifier. Three phase rectifier are

classified as 3-Փ half wave rectifier and 3-Փ full wave rectifier. 1-Փ Full wave rectifier are

classified as1-Փ mid point type and 1-Փ bridge type rectifier. 1-Փ bridge type rectifier are

classified as 1-Փ half controlled and 1-Փ full controlled rectifier. 3-Փ full wave rectifier are

again classified as 3-Փ mid point type and 3-Փ bridge type rectifier. 3-Փ bridge type rectifier

are again divided as 3-Փ half controlled rectifier and 3-Փ full controlled rectifier.

Single phase half wave circuit with R-L load

Output current 𝑖𝑜 rises gradually. After some time 𝑖𝑜 reaches a maximum value and then

begins to decrease.

At π, 𝑣𝑜=0 but 𝑖𝑜 is not zero because of the load inductance L. After π interval SCR is reverse

biased but load current is not less then the holding current.

At β>π, 𝑖𝑜 reduces to zero and SCR is turned off.

At 2π+β SCR triggers again

α is the firing angle.


DEPT.OF EEE VEMUIT

R2 + X

2

Vm

β is the extinction angle.

v = − = conduction angle

Analysis for 𝑉𝑇.

At 𝜔𝑡 = 𝐼,𝑉𝑇 = 𝑉𝑚𝑠𝑖𝑛𝐼

During = 𝐼 𝑡𝑜 𝐼 , 𝑉𝑇 = 0;

When = 𝐼 , 𝑉𝑇 = 𝑉𝑚𝑠𝑖𝑛𝐼;

Vm sin t = Ri0 L di0

dt

is = sin( t−)

Where,

= tan−1 X

R

X = L

Where 𝐼 is the angle by which 𝐼𝑠 lags 𝑉𝑠.

The transient component can be obtained as

Rit + L

di0 = 0 dt

So 𝑖𝑡 = 𝐴𝑒−(𝑅𝑡⁄𝐿)

𝑖0 = 𝑖𝑠 + 𝑖𝑡

𝑉𝑚 sin( 𝜔𝑡 − 𝐼) + 𝐴𝑒−(𝑅𝑡⁄𝐿)

𝑧

Where 𝑧 = √𝑅2 + 𝑋2

At 𝛼 = 𝜔𝑡, 𝑖𝑜 = 0;

0 = 𝑉𝑚

sin( 𝛼 − 𝐼) + 𝐴𝑒−(𝑅𝛼⁄𝐿𝜔); 𝑧

𝐴 = −𝑉𝑚

sin( 𝛼 − 𝐼)𝑒(𝑅𝛼⁄𝐿𝜔)

𝑧

𝑖 = 𝑉𝑚

sin( 𝜔𝑡 − 𝐼) − 𝑉𝑚

sin( 𝛼 − 𝐼)𝑒−𝑅(𝜔𝑡−𝛼)⁄𝐿𝜔

𝑜 𝑧 𝑧


DEPT.OF EEE VEMUIT

Therefore,

𝜔𝑡 = 𝛽, 𝑖0 = 0;

So sin(𝛽 − 𝛼) = sin(𝛼 − 𝐼)𝑒−(𝛽−𝛼)/(𝜔𝐿)

β can obtained from the above equation.

The average load voltage can be given by

1

V0 = 2

Vm sin td (t)

𝑉𝑚 (cos(𝛼) − cos(𝛽)) 2𝜋

Average load current

I0 = Vm

2 R (cos − cos )

Single phase full converter

1 +

V0 =

Vm sin(t)d (t)

= 2Vm cos

T₁,T₂ triggered at α and π radian latter T₃, T₄ are triggered.


DEPT.OF EEE VEMUIT

Single phase half wave circuit with RLE load

The minimum value of firing angle is


DEPT.OF EEE VEMUIT

Vm sin(t) = E

So,

1

= sin

−1 E

Vm

Maximum value of firing angle

2 = −2

The voltage differential equation is

Vm sin(t) = Ri0 +

L di0 + E dt

is = is1 + is 2

Due to source volt

is1

= Vm sin(t −) Z

Due to DC counter emf


DEPT.OF EEE VEMUIT

t

is 2 = −(E / R)

i = Ae−( R/ L)t

Thus the total current is given by

is1 + is 2 + it

= Vm sin(t −) −

E + Ae

−(R/L) t

Z R

is0

= Vm sin(t −) −

E + Ae

−(R/L) t

Z R

At t = i0 = 0

A = [ E

− V

m sin( −)]e− R L

R Z

So

V { − R

(t − )} E =

{

− R (t − )

i m [sin(t −) − sin( −)e L − [1− e L ] 0

Z R

Average voltage across the inductance is zero. Average value of load current is

I = 1

(V sin t − E) d(t)

0 2 R m

= 1

[V

(cos − cos ) − ( − )]

2 R m

Conduction angle = −

= + v

I0 = 1

2 R [Vm (cos − cos( + v)) − (v)]

cos A − cosB = 2sin A + B

sin A − B

2 2

So

I0 =

1

2 R

[2Vm

sin( + v

) sin v

− E. ] 2 2


DEPT.OF EEE VEMUIT

) sin ]

= E + I0 R

= E+ 1

2 [2Vm sin( +

v ) sin

v − E.

] 2 2

= E(1− v

) + [Vm sin( +

v v

2 2 2

If load inductance L is zero then

= 2

And

v = − = 2 −

But 2 = −1

So = 2 = −1

And v = −1 −

So average current will be

I0 = 1

2 R [Vm (cos − cos( −1 )) − E( −1 − )]

So V0=E+I0R

= Vm (cos + cos ) +

E (1+

1 + )

2 1

2

For no inductance rms value of load current

I = [ 1 −

2 1/2

0 2 R2

(Vm sin( t) − E) d (t )]

Power delivered to load

P = I

2 R + I E

or 0

Supply power factor

I

2 R + I E

Pf = or 0

Vs Ior

Single phase full wave converter:


DEPT.OF EEE VEMUIT


DEPT.OF EEE VEMUIT

V = 1 +

V

sin(t)d (t)

0 m

= 2Vm sin

Single phase semi converter:

V = 1

V sin(t)d (t)

0 m

= Vm cos

full converter:


DEPT.OF EEE VEMUIT

2

m

steady state analysis

V = Ri + L di0 + E

s o dt

V0 = RI0 + E

V = 2Vm cos

0

So in case of DC motor load

V0 = ra Ia +mm

2Vm cos − r I

a a

m =

So m

T = m Ia

Ia = Te

Put

Ia =

m

Te

m

(

2Vm ) cos

= − raTe

So m m


DEPT.OF EEE VEMUIT

UNIT-III

CHOPPERS AND REGULATORS


DEPT.OF EEE VEMUIT

0

1

CHOPPER

A chopper is a static device that converts fixed DC input voltage to variable output voltage directly.

Chopper are mostly used in electric vehicle, mini haulers.

Chopper are used for speed control and braking. The systems employing chopper offer smooth

control, high efficiency and have fast response.

The average output voltage is

1 t1

1

Va = T

V0dt = T

Vs (t1 ) = ft1Vs = Vs

The average load current

I = Va

=

Vs

a R R

Where, T=chopping period Duty

cycle of chopper =

= t1

T

f=chopping frequency The rms value of output voltage is

1 0 T 0 s

V = ( V 2dt)2 = V

0

If we consider the converter to be loss less then the input power is equal to the output power and is given by


DEPT.OF EEE VEMUIT

o

I a

1 T

1 T

V 2

Pi = T

V0idt = T R

dt 0 0

1 V 2 V 2

= s (T ) = s

T R R

The effective input resistance seen by the P source is

P = Vs =

Vs = R

i Vs

R

The duty cycle α can be varied by varying t₁ , T of frequency.

Constant frequency operation:

1) The chopping period T is kept constant and on time is varied. The

pulse width modulation ,the width of the pulse is varied.

2) Variable frequency operation, the chopping frequency f is varied.

Frequency modulation, either on time or off time is kept constant.

This type of control generate harmonics at unpredictable frequency and filter design is often difficult.

TYPES OF CHOPPER:

FIRST QUADRANT OR TYPE A CHOPPER:

When switch ON

V0 = Vs .


DEPT.OF EEE VEMUIT

Current i₀ flows in the same direction when switch off. V₀=0,

i₀=0

So, average value of both the load and the current are positive.

SECOND QUADRANT OR TYPE B CHOPPER:

When switch are closed the load voltage E drives current through L and switch. During Ton

L stores energy.

When switch off

V0 exceeds source voltage

Vs .

V = E + L di

0

dt

Diode D₂ is forward biased.power is fed back to supply. As V₀is more than sourse voltage. So such chopper is

called step up chopper.

V = E + L di

0

dt

So current is always negative and V₀ is always positive.

TWO QUADRANT TYPE A CHOPPER OR, TYPE C CHOPPER:


DEPT.OF EEE VEMUIT

Both the switches never switch ON simultaneously as it lead direct short circuit of the supply.

Now when sw2 is closed or FD is on the output voltage V₀ is zero. When sw1 is

ON or diode D conducts output voltage is V₀ is +Vs’ CURRENT ANANLYSIS:

When CH1 is ON current flows along i0. When CH1 is off current continues to flow along i0 as FD is

forward biased. So i0 is positive.

Now when CH2 is ON current direction will be opposite to i0. When sw2 is off D2 turns ON.

Load current is –i0. So average load voltage is always positive. Average load current may be positive or

negative.

TWO QUADRANT TYPE B CHOPPER, OR TYPE D CHOPPER:


DEPT.OF EEE VEMUIT

When CH1 and CH2 both are on then V0=Vs.

When CH1 and CH2 are off and D1 and D2 are on V 0=-Vs.

The direction of current is always positive because chopper and diode can only conduct in the direction

of arrow shown in fig.

Average voltage is positive when Ton>Toff

FOUR QUADRANT CHOPPER, OR TYPE E CHOPPER


DEPT.OF EEE VEMUIT

FIRST QUADRANT:

CH4 is kept ON

CH3 is off

CH1 is operarted

V0=Vs

i0 = positive

when CH1 is off positive current free wheels through CH4,D2 so V0

and I2 is in first quadrant.

SECOND QUADRANT:

CH1,CH3,CH4 are off.

CH2 is operated.

Reverse current flows and I is negative through L CH2 D4 and E. When CH2

off D1 and D4 is ON and current id fed back to source. So

E + L di

dt is more than source voltage Vs.

As i0 is negative and V0 is positive, so second quadrant operation. THIRD

QUADRANT:

CH1 OFF, CH2 ON

CH3 operated. So both V0 and i0 is negative.

When CH3 turned off negative current freewheels through CH2 and D4.

FOURTH QUADRANT:


DEPT.OF EEE VEMUIT

2

CH4 is operated other are off.

Positive current flows through CH4 E L D2.

Inductance L stores energy when current fed to source through D3 and D2.V0 is negative.

STEADY STATE ANALYSIS OF PRACTICAL BUCK CHOPPER:

The voltage across the inductor L is e i=Ldi/dt.

V −V = L d (i2 − i1 ) = L

i

t1 t1

t1 = iL

V −V s a .

The inductor current falls linearly from I2 to I1 in time t2 as Vs =0.

So

−Va

=

L(i1 − i2 ) t

2

If I2-I1=ΔI then

−Va = − LI t

2

t = LI

Va .

ΔI=I2-I1= peak to peak ripple current.

I = (Vs −Va ) t1 =

Vat2

L L

Now t1=αT, t2=(1-α)T

V = V t1 = V

a s T

s

Α<1 so it is a step down or buck converter. If the

circuit is lossless then VsIs=Va Ia=αVsIa Is =α Ia.

Now switching period T can be expressed as

s a


DEPT.OF EEE VEMUIT

s

T=1/f= t1+ t2=ΔIL/(Vs-Va) + ΔIL/(Va)

=ΔILVs/Va(Vs-Va)

So peak to peak ripple current

I = Va (Vs −Va )

fLVs

I = Va (1− )

fL

The peak to peak voltage of the capacitor is

Vc =

I

8 fc

So from above equation

V = Va (Vs −Va ) .

Vs (1− )

c 8Lcf

2V 8Lcf

2

Condition for continuous inductor current and capacitor voltage :

If IL is the average inductor current

IL = 2IL ….as

Va = Vs

Vs (1− ) =

fL

I2 − I1 = I

As 2 L

So I = 2IL

Vs (1− )

fL …..eq (2)

Vs (1− ) = 2I

fL L = 2Ia =

2Vs

R

…..eq(4)


DEPT.OF EEE VEMUIT

1

1

As Va = Vs

I = Vs

so a

R

2Ia = 2Vs

R

So equation 4 gives

L = (1− )R

c 2 f

Which is the critical value of inductor

Vc = 2Va

2V = Vs (1− )

= 2V

a 8Lcf

2 s

c = 1−

16Lf 2

Peak to peak ripple voltage of capacitor:

Vc = Vc −Vc (t = 0)

= 1 t1

I dt = 1

t1

I

= Iat1

c 0 c

c 0 a c

t = Va −Vs

So Vaf

t = Va −Vs

Vaf

1− = Vs

Va

1− t1 =

Vs

T Va

t1 = Va −Vs

Va f

Vc

So

= Ia (

Va −Vs )

c Vaf


DEPT.OF EEE VEMUIT

Vc = Ia

fc

Condition for continuous imductor current and capacitor voltage:

If IL= average inductor current then

I = I

L 2

I = Vs

= 2I fL

L = 2Ia = 2Vs

(1− )R

Va = As

Vs

1−

2Ia = 2Vs

(1− ) R

I = 2I = 2I = 2Vs = Vs

So

Lc

L L a

= (1− a)R

2 f

(1− )R fL

Vc = 2Va

Ia = 2V = 2I R

cf a a

c =

2 fR


DEPT.OF EEE VEMUIT

UNIT-IV

INVERTERS


DEPT.OF EEE VEMUIT

⁄

4

01 𝑆

INVERTERS

The device that converts dc power into ac power at desired output voltage and frequency is called an

inverter.

Single phase voltage source inverters

1 𝑇0⁄

𝑉2

𝑉𝑆

𝑉𝑜(𝑟𝑚𝑠) = 𝑇0 ∫0

2

2 𝑆 dt =

∞

𝑉0=

𝑎0 + ∑(𝑎𝑛 cos(𝑛𝜔𝑡) + 𝑏𝑛 sin(𝑛𝜔𝑡)) 2

𝑛=1

Due to symmetry along x-axis

𝑎0 = 0 , 𝑎𝑛 = 0

𝑏𝑛 = 4𝑉𝑆

𝑛𝜋

The instantaneous output voltage

∞ 2𝑉𝑆

sin(𝑛𝑤𝑡) 𝑣0 = ∑𝑛=1,3,5… 𝑛𝜋

=0, n=2,4.....

The rms value of the fundamental output voltage

𝑉 = 2𝑉𝑆 =0.45𝑉

√2𝜋

2


DEPT.OF EEE VEMUIT

01 01

1 𝑆

𝑅

So if 𝑉0 = ∑∞ 2𝑉𝑆 sin(𝑛𝑤𝑡)

𝑛=1,3,5… 𝑛𝜋

=∑∞ 2𝑉𝑆 sin(𝑛𝜔𝑡 − 𝜃𝑛)

𝑛=1,3,5…. 𝑛𝜋√𝑅2+(𝑛𝜔𝐿)2

𝑃 = (𝐼 )2𝑅 = [ 2𝑉𝑆

√2𝜋√𝑅2+(𝜔𝐿)2 ]2R

DC Supply Current

Assuming a lossless inverter, the ac power absorbed by the load must be equal to the average

power supplied by the dc source.

∫𝑇 𝑖𝑠(𝑡)𝑑𝑡 = 1 ∫

𝑇 √2𝑉01 sin(𝜔𝑡) √2𝐼0 sin(𝜔𝑡 − 𝜃1) 𝑑𝑡=𝐼𝑆

0 𝑉𝑠 0

𝑉01 =Fundamental rms output output voltage

𝐼0=rms load current

𝜃1=the load angle at the fundamental frequency Single

phase full bridge inverter

For n=1, 𝑉 = 4𝑉𝑆 =0.9𝑉 (The rms of fundamental) √2𝑉𝑆

Instantaneous load current 𝑖0 for an RL load

𝑖0 = ∑∞ 4𝑉𝑆 sin(n𝜔𝑡 − 𝜃𝑛) 𝑛=1,3,5… 𝑛𝜋√𝑅2+(𝑛𝜔𝐿)2

𝜃𝑛=tan−1(𝑛𝜔𝐿)

The rms output voltage is 𝑉 =(

2 ∫𝑇⁄2 𝑉2)

1⁄2 =𝑉

0 𝑇0 0 𝑆 𝑆


DEPT.OF EEE VEMUIT

The instantaneous output voltage in a fourier series

𝑣0

∞

= ∑ 4𝑉𝑆

sin(𝑛𝜔𝑡) 𝑛𝜋

𝑛=1,3,5…

Single phase bridge inverter

INVERTER

Inverters are of the two types

1) VSI

2) CSI

Pulse width model

The VSI can be further divided into general 3 categories:

1.Pulse width modulated inverters

2. Square wave inverters

3. Single phase inverter with voltage cancellation


DEPT.OF EEE VEMUIT

Pulse width modulated inverters

The input dc voltage is of constant magnitude . The diode rectifier is used to rectify the line voltage.The

inverter control the magnitude and frequency of the ac output voltage.

This is achieved by PWM technique of inverter switches and this is called PWM inverters.

The sinusoidal PWM technique is one of the PWM technique to shape the output voltage to as close as

sinusoidal output.

Basic concepts of switch mode inverter


DEPT.OF EEE VEMUIT

During interval 1 𝑣0and 𝑖0 both are positive During

interval 3 𝑣0 and 𝑖0 both are negative

Therefore during 1 and 3 the instantaneous power flow is from dc side to corresponding to inverter

mode of operation.

In contrast during interval 2 and 4 𝑣0 and 𝑖0 are of opposite sign i.e. power flows from ac side to dc

side corresponding to rectifier mode of operation.

Pulse width modulated switching scheme

We require the inverter output to be sinusoidal with magnitude and frequency controllable.

In order to produce sinusoidal output voltage at desired frequency a sinusoidal control signal at

desired frequency is compared with a triangular waveform as show.

The frequency of the triangular waveform established the inverter switching frequency.

The triangular waveform is called carrier waveform. The triangular waveform establishes switching

frequency 𝑓𝑠 , which establishes with which the inverter switches are applied.

The control signal has frequency 𝑓𝑠 and is used to modulate the switch duty ratio.

𝑓1 is the desired fundamental frequency of the output voltage.

The amplitude modulation ratio 𝑚𝑎 is defined as

m 𝑉𝑐𝑜𝑛𝑡𝑟𝑜𝑙

a= 𝑉𝑡𝑟𝑖


DEPT.OF EEE VEMUIT

f

𝑉𝑐𝑜𝑛𝑡𝑟𝑜𝑙 is the peak amplitude of control signal.

𝑉𝑡𝑟𝑖 peak amplitude of triangular signal. The

frequency modulation ratio 𝑚𝑓

m = 𝑓𝑠 𝑓1


DEPT.OF EEE VEMUIT

1

2

When 𝑉𝑐𝑜𝑛𝑡𝑟𝑜𝑙 > 𝑉𝑡𝑟𝑖 𝑇+ is ON 𝑉𝐴𝑂 = 1 𝑉𝑑

𝐴 2

𝑉𝑐𝑜𝑛𝑡𝑟𝑜𝑙 < 𝑉𝑡𝑟𝑖 𝑇− is ON 𝑉𝐴𝑂 = 1 𝑉𝑑

𝐴 2

So the following inferences can be drawn

The peak amplitude of fundamental frequency is matimes

2 𝑉𝑑

𝑉𝐴𝑂=𝑚𝑎𝑉𝑑

𝑉𝐴𝑂=𝑉𝑐𝑜𝑛𝑡𝑟𝑜𝑙 ∗

𝑉𝑑 𝑉𝑐𝑜𝑛𝑡𝑟𝑜𝑙 ≤ �̂�𝑡𝑟𝑖 �̂�𝑡𝑟𝑖 2

The foregoing arguments shown why Vcontrolis chosen to be sinusoidal to provide sinusoidal output

voltage with fewer harmonics

Let the Vcontrol vary sinusoidal with frequency f1,which is the desired frequency of the inverter

output voltage.

Let Vcontrol=V̂controlsin𝜔1t

V̂control ≤ V̂tri


DEPT.OF EEE VEMUIT

2

1

�̂�𝑡𝑟𝑖=�̂�𝑡𝑟𝑖

𝑡1 𝑇𝑠⁄4

At t=𝑡1 , 𝑣𝑡𝑟𝑖 =𝑣𝑐𝑜𝑛𝑡𝑟𝑜𝑙

So 𝑣𝑐𝑜𝑛𝑡𝑟𝑜𝑙=

�̂�𝑡𝑟𝑖

𝑡1 𝑇𝑠⁄4

𝑡1=�̂�𝑐𝑜𝑛𝑡𝑟𝑜𝑙*

𝑇𝑆

�̂�𝑡𝑟𝑖 4

𝑇𝑜𝑛=2𝑡1+𝑇𝑆

𝑇 2𝑡1+𝑇𝑆

𝐷1= 𝑜𝑛= 2

𝑇𝑠

=2

+

2

2𝑡1

𝑇𝑠

𝐷 1

1 �̂�𝑐𝑜𝑛𝑡𝑟𝑜𝑙

1=2+

2(

�̂�𝑡𝑟𝑖 )

Three phase inverter

When three single-phase inverters are connected in parallel a three phase inverter is formed.

The gating signal has to be displaced by 1200 with respect to each other so as achieve three phase

balanced voltages.

A 3-phase output can be achieved from a configuration of six transistors and six diodes.


DEPT.OF EEE VEMUIT

Two type of control signal can be applied to transistors, they are such as 1800 or 1200 conduction.

180- degree conduction

When 𝑄1 is switched on, terminal a is connected to the positive terminal of dc input voltage. When 𝑄4 is

switched on terminal a is brought to negative terminal of the dc source.

There are 6 modes of operation is a cycle and the duration of each mode is 600.

The conduction sequence of transistors is 123,234,345,456,561,612. The gating signals are shifted

from each other by 600 to get 3-𝜑 balanced voltages.

Switching states for the three phase voltage inverters


DEPT.OF EEE VEMUIT


DEPT.OF EEE VEMUIT

4,5,6 5,6,1

3,4,5 6,1,2

2,3,4 1,2,3

𝑉𝑅𝑁 𝑉𝑌𝑁 𝑉𝐵𝑁 𝑉𝑅𝑌 𝑉𝑌𝐵 𝑉𝐵𝑅 𝑉1

𝑉

3 −2𝑉

3

𝑉

3

𝑉𝑎𝑐 −𝑉𝑑𝑐 0 2 (3300)

√3

2𝑉

3

−𝑉

3

−𝑉

3

𝑉𝑑𝑐 0 −𝑉𝑑𝑐 2

(300) √3

𝑉3

𝑉

3

−2𝑉

3

0 V -V 2 (900)

√3

−𝑉

3

2𝑉

3

−𝑉

3

-V V 0 2 (1500)

√3

−2𝑉

3

𝑉

3

𝑉

3

-V 0 0 2 (2100)

√3

−𝑉

3

−𝑉

3

2𝑉

3

0 -V 0 2 (2700)

√3

Fourier analysis

If we go for harmonic analysis 𝑉𝑅𝑌=∑∞

4𝑉𝑠𝑖𝑛

𝑛𝜋 sin 𝑛(𝜔𝑡+𝜋⁄ )

𝑛=1,3,5…𝑛𝜋 3 6


DEPT.OF EEE VEMUIT

𝑉 ∑∞

4𝑉𝑆𝑠𝑖𝑛 𝑛𝜋

sin 𝑛(𝜔𝑡−𝜋⁄ ) 𝑌𝐵= 𝑛=1,3,5… 𝑛𝜋 3 2

𝑉 ∑∞

4𝑉𝑆𝑠𝑖𝑛 𝑛𝜋

sin 𝑛(𝜔𝑡−𝜋⁄ ) 𝐵𝑅= 𝑛=1,3,5… 𝑛𝜋 3 6

All even harmonics are zero all triple n harmonics are zero. The

rms nth component of the line voltage is

4𝑉 𝑛𝜋 4𝑉

=√2𝑛𝜋

sin 3

=√2𝜋

sin(60)

For n=1

=0.7797𝑉𝑆

Three phase 1200mode VSI

The circuit diagram is same as that for 1800 mode of conduction.

Here each thyristor conducts for 1200.There are 6 steps each of 600 duration, for completing

one cycle of ac output voltage.

5,6 6,1

4,5 1,2

3,4 2,3


DEPT.OF EEE VEMUIT

Step 1: 6,1 conducting

𝑉𝑎𝑛 = 𝑉𝑆 , 𝑉𝑦𝑛 =

−𝑉𝑆, 𝑉𝑐𝑛=0 2 2


DEPT.OF EEE VEMUIT


𝑉𝑎𝑛 = 𝑉𝑆 , 𝑉𝑏𝑛 = 0, 𝑉𝑐𝑛=

−𝑉𝑆

2 2


𝑉𝑎𝑛 =0, 𝑉𝑏𝑛=𝑉𝑆, 𝑉𝑐𝑛=

−𝑉𝑆

2 2


𝑉𝑎𝑛 = −𝑉𝑆, 𝑉𝑏𝑛=

𝑉𝑆, 𝑉𝑐𝑛=0 2 2


𝑉𝑎𝑛== − 𝑉𝑆, 𝑉𝑦𝑛 = 0, 𝑉𝑏𝑛 =

𝑉𝑆

2 2


𝑉𝑎𝑛=0, 𝑉𝑏𝑛 = − 𝑉𝑆, 𝑉𝑐𝑛=

𝑉𝑆

2 2

1200 conduction mode

Step

Thyristor

conducting

𝑉𝑅𝑛

𝑉𝑌𝑛

𝑉𝐵𝑛

→ V

1 6,1 Vs⁄2 −𝑉𝑠⁄

2 0 √3VS

(−300) 2

2 1,2 𝑉𝑠⁄2 0 −𝑉𝑠⁄

2 √3VS

(300) 2

3 2,3 0 𝑉𝑠⁄2 −𝑉𝑠⁄

2 √3VS

(900) 2

4 3,4 −𝑉𝑠⁄2 𝑉𝑠⁄

2 0 √3VS

(1500) 2

5 4,5 −𝑉𝑠⁄2 0 𝑉𝑠⁄

2 √3V

(2100) 2

6 5,6 0 −𝑉𝑠⁄2 𝑉𝑠⁄

2 √3V

(−300) 2


DEPT.OF EEE VEMUIT


DEPT.OF EEE VEMUIT

UNIT-V

AC VOLTAGE CONTROLLERS

&

CYCLO CONVERTERS


DEPT.OF EEE VEMUIT

This lesson provides the reader the following:

(i) AC-AC power conversion topologies at fixed frequency

(ii) Power converter options available for the conversion

(iii) Ability to formulate equations describing the current waveform for the PAC

(iv) Ability sketch the current waveform by observation of the circuit

(v) Ability to assess the performance of the converter of the topologies

Introduction

AC to AC voltage converters operates on the AC mains essentially to regulate the output

voltage. Portions of the supply sinusoid appear at the load while the semiconductor switches

block the remaining portions. Several topologies have emerged along with voltage regulation

methods, most of which are linked to the development of the semiconductor devices.

Fig 26.1 Some single phase AC-AC voltage regulator topologies. (a) Back-to-back

SCR; (b) One SCR in (a) replaced by a four-diode full wave diode bridge; (c) A

bi-directionally conducting TRIAC; (d) The SCR in (b) replaced by a transistor.

The regulators in Fig 26.1 (a), (b) and (c) perform quite similarly. They are called Phase

Angle Controlled (PAC) AC-AC converters or AC-AC choppers. The TRIAC based converter

may be considered as the basic topology. Being bi-directionally conducting devices, they act on

both polarities of the applied voltage. However, dv dtre−applied their ratings being poor, they tend

to turn-on in the opposite direction just subsequent to their turn-off with an inductive load. The

'Alternistor' was developed with improved features but was not popular. The TRIAC is common

only at the low power ranges. The (a) and (b) options are improvements on (c) mostly regarding

current handling and turn-off-able current rating.

A transistorised AC-AC regulator is a PWM regulator similar to the DC-DC converters. It

also requires a freewheeling path across the inductive load, which has also got to be bi-

directional. Consequently, only controlled freewheeling devices can be used.


DEPT.OF EEE VEMUIT

Operation with resistive loads

Fig. 26.2 illustrates the operation of the PAC converter with a resistive load. The device(s) is

triggered at a phase-angle '' in each cycle. The current follows the voltage wave shape in each

half and extinguishes itself at the zero crossings of the supply voltage. In the two-SCR topology,

one SCR is positively biased in each half of the supply voltage. There is no scope for conduction

overlap of the devices. A single pulse is sufficient to trigger the controlled devices with a

resistive load. In the diode-SCR topology, two diodes are forward biased in each half. The SCR

always receives a DC voltage and does not distinguish the polarity of the supply. It is thus

always forward biased. The bi-directional TRIAC is also forward biased for both polarities of the

supply voltage.

Fig. 26.2 Operation of a Phase Angle Controlled AC-AC converter with a resistive load

The rms voltage Vrms decides the power supplied to the load. It can be computed as

Vrms =

= V

1

2V sin 2 2

t dt

1 −

+ sin2

2


DEPT.OF EEE VEMUIT

Fig. 26.3 The rms output voltage and the most important harmonics versus triggering

angle

As is evident from the current waveforms, the PAC introduces significant harmonics both into

the load and the supply. This is one of the main reasons why such controllers are today not

acceptable. The ideal waveform as shown in Fig 26.2 is half wave symmetric. However it is to be

achieved by the trigger circuits. The controller in Fig. 26.4 ensures this for the TRIAC based

circuit. While the TRIAC has a differing characteristic for the two polarities of biasing with the

32V DIAC - a two terminal device- triggering is effected when the capacitor voltage reaches 32 V. This ensures elimination of DC and even components in the output voltage.

Fig. 26.4 DIAC based trigger circuit for a TRIAC to ensure symmetrical triggering

in the two halves of the supply.

For the SCR based controllers, identical comparators for the two halves of the AC supply, which

generates pulses for the two SCRs ensures DC and even harmonic free operation.

The PAC operates with a resistive load for all values of ranging from 0o

The fundamental current, if can be represented as

i =

sin 2 1 cos 2 2V

R ( −

2 +

2

)sin t − ( 2

− 2 )cost

26.1

In machine drives it is only the fundamental component, which is useful. However, in

resistance heating type of application all harmonics are of no consequence. The corrupted supply

f


DEPT.OF EEE VEMUIT

current nevertheless is undesirable.


DEPT.OF EEE VEMUIT

Power Factor

The power factor of a nonlinear deserves a special discussion. Fig. 26.2 shows the supply

voltage and the non-sinusoidal load current. The fundamental load/supply current lags the

supply voltage by the , 'Fundamental Power Factor' angle. Cos is also called the

'Displacement Factor'. However this does not account for the total reactive power drawn by the

system. This power factor is inspite of the actual load being resistive! The reactive power is

drawn also y the trigger-angle dependent harmonics. Now

power factor = average power =

P 26.2

apparent voltamperes VIL

= VIL1 cos1

VIL

distortion factor = IL1

IL

The Average Power, P drawn by the resistive load is 1 2 1 2V

2 2

26.3

P = 2 0

viL dt =

= 2V

2

sin 2

R

− 2

+ 2

sin R

t dwt

The portion within square brackets in Eq. 26.5 is identical to the first part of the expression within brackets in Eq. 26.1, which is called the Fourier coefficient 'BB1'. The rms load

voltage can also be similarly obtained by integrating between and and the result can be

combined with Eq. 26.5 to give

power factor = per − unit

=

rms load − current

per − unit load power


DEPT.OF EEE VEMUIT

Fig. 26.5 Variation of various performance parameters with triggering angle

= B 1

p.u..


DEPT.OF EEE VEMUIT

Operation with inductive loads

With inductive loads the operation of the PAC is illustrated in Fig 26.5. The current builds up

from zero in each cycle. It quenches not at the zero crossing of the applied voltage as with the

resistive load but after that instant. The supply voltage thus continues to be impressed on the load

till the load current returns to zero. A single-pulse trigger for the TRIAC 26.1 (c) or the anti-

parallel SCR (b) has no effect on the devices if it (or the anti-parallel device) is already in

conduction in the reverse direction. The devices would fail to conduct when they are intended to,

as they do not have the supply voltage forward biasing them when the trigger pulse arrives. A

single pulse trigger will work till the trigger angle > where is the power factor angle of the

inductive load. A train of pulses is required here. The output voltage is controllable only between

triggering angles and 180o.

The load current waveform is further explained in Fig. 26.6. The current is composed of two

components. The first is the steady state component of the load current, iss and the second, itr is

the transient component.

Fig. 26.6 Operation of a single phase PAC with an inductive load

Fig 26.7 Load current for a single phase AC-AC converter with a R_L load. Vs -

supply voltage, iss -steady state current component , itr - transient current

component and iload - load current (= iss + itr).


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)

With an inductance in the load the distinguishing feature of the load current is that it must

always start from zero. However, if the switch could have permanently kept the load connected

to the supply the current would have become a sinusoidal one phase shifted from the voltage by

the phase angle of the load, This current restricted to the half periods of conduction is called

the 'steady-state component' of load current iss. The 'transient component' of load current itr, again

in each half cycle, must add up to zero with this iss to start from zero. This condition sets the

initial value of the transient component to that of the steady state at the instant that the

SCR/TRIAC is triggered. Fig. 26.6 illustrates these relations.

When a device is in conduction, the load current is governed by the equation

L di dt

+ Ri =vs

iload

2V =

Z

( ) ( − R (

−t )

sin t − + sin − e L

Since at t = 0, iload = 0 and supply voltage vs = Vsint the solution is of the form

The instant when the load current extinguishes is called the extinction angle . It can be

inferred that there would be no transients in the load current if the devices are triggered at the

power factor angle of the load. The load current I that case is perfectly sinusoidal.

AC-AC Chopper

Fig. 26.8 A complete Transitorised AC-AC chopper topology of the version shown in

Fig. 26.1 and the corresponding load voltage and current waveforms for an inductive

load. The output voltage is shown to be about 50% for a 0.5 Duty Ratio chopping.

The AC-AC converter shown in Fig 26.1 has to be augmented with two additional controlled

devices clamping the load as indicated in Fig. 26.7. A large capacitor across the supply terminals

is also to be inserted. These devices which are mostly transistors of the same variety as used for

the chopper are necessary to clamp the voltages generated by the switching-off of the current

carrying inductors in the load while the input capacitor takes care of the line inductances. The

harmonics in the line current and load voltage waveforms are significantly different from those

with the PACs. Mostly switching frequency harmonics are present in both the waveforms.


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PAC as a static switch

Both single phase and three phase PACs are often used as static switches for applications like

switching on of highly inductive loads without transients or for regulating output AC voltages by

switching in tapings of a transformer. Such sequence control PACs while controlling the output

voltage also permit improvement of the power factor as seen by the source. Sequence control can

be two or multiple phase depending upon the application. Typical load voltage and current

waveforms are shown in Fig. 26.8. The outer TRIACs connected to thwe higher voltage leads of

the input transformer are triffered at the desired angle to realize the required load voltage.

Obviously this voltage is greater than that available at the low voltage terminal of the

transformer. This device continues conduction into the next half of the supply voltage till the

load current falls to zero. The inner TR2 starts conduction subsequently, requiring a wide pulse

or a train of pulses. TR1 can be however triggered by a single pulse.

Fig. 26.9 Load voltage and current control with a two-stage sequence control

Practice Questions and Problems with Answers

Q1 A single-phase transformer, assumed to have a negligible resistance compared to its

inductance is switched on via a PAC. At what trigger angle will the operation be free from

transients?

A1 For the transformer load L

Therefore for transient free operation =

Q2 For the load described in Q1, the PAC is triggered by a single pulse at = Sketch the

load current waveform.


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A2 Since L, the load current should have been continuous. However, the current in the

SCR first triggered extinguishes at a the total load current, iL = iss + itr = 0. For this load

which can be considered to be highly inductive , say . Thus the first SCR

conducts till that angle. The anti-parallel SCR is triggered at = corresponding to a

+ = when it is still reverse biased. It fails to conduct. The load thus sees

only a unipolar current. The load current and voltage waveforms are illustrated in Fig 26.

A2. Note that both the load voltage and current waveforms contain DC components.

Fig. 26. A2 The load current waveform and its steady-state and transient

components when a highly inductive load is switched using single narrow trigger

pulses.


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CYCLO CONVERTERS


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Instructional Objectives

Study of the following:

• The cyclo-converter circuits − basic principle of operation

• The circuit for the single-phase to single-phase cyclo-converter using thyristors

• The operation of the above cyclo-converter circuit, along with the voltage waveforms

Introduction

Earlier in the last three (4.1-4.3) lessons (first half) of this module, the circuit and operation

of ac to ac voltage controllers − both single-phase and three-phase, were described in detail. The

devices used are either triac, or thyristors connected back to back. In this lesson (4.4) − first one

in the second half of this module, the cyclo-converter is introduced as a type of power controller,

where an alternating voltage at supply frequency is converted directly to an alternating voltage at

load frequency (normally lower), without any intermediate dc stage. As will be shown in the last

(fifth) module, an alternating voltage at any frequency (output) is obtained using an inverter as a

power controller from a dc voltage fed at its input. This input, i.e. dc voltage, is again obtained

using a rectifier (converter) with ac voltage (normally at supply frequency) fed at its input. This

type has been described in module 2. Note that this is a two-stage process with an intermediate

dc stage. Now-a-days, the power switching devices used in the inverter circuit belong to

transistor family (termed as self-commutated ones), starting with power transistors, whereas

thyristors are still being used in the converter (rectifier) circuits. These devices are called force-

commutated ones, when used in dc chopper circuits (described in module 3), but in this case, i.e.

converter circuits, line commutation takes place. As stated earlier, the output frequency of the

cyclo-converter is limited to about one-third of supply (line) frequency of 50 Hz.

Initially, the basic principle of operation used in a cyclo-converter is discussed. Then, the

circuit of a single-phase to single-phase cyclo-converter using thyristors is presented. This is

followed by describing the operation of the above cyclo-converter circuit, along with voltage

waveforms. The readers at this stage, have gone through the following lessons − single-phase

fully controlled converter using thyristors, for obtaining dc output voltage from ac supply (#2.2),

and ac to ac voltage controllers − both single-phase and three-phase, using either triac, or

thyristors connected back to back (#4.1-4.3). In the above cases, the output voltage obtained is,

in the form of phase-controlled one, as can be observed from the waveforms shown in the above

lessons. In the present case, the output voltage of the cyclo-converter circuit (single-phase) using

thyristors, is synthesized from the above phase-controlled voltage waveforms, so as to obtain an

ac waveform (output) of low frequency, with the input being an ac voltage of higher frequency,

say line. The angle, at which the thyristors are triggered, is controlled to obtain the desired

waveform.

Keywords: Single-phase to single-phase cyclo-converter using thyristors, Voltage waveforms.

Cyclo-converter

Basic Principle of Operation


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The basic principle of operation of a cyclo-converter is explained with reference to an equivalent circuit shown in Fig. 29.1. Each two-quadrant converter (phase-controlled) is


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eo

iO +

ac load

represented as an alternating voltage source, which corresponds to the fundamental voltage

component obtained at its output terminals. The diodes connected in series with each voltage

source, show the unidirectional conduction of each converter, whose output voltage can be either

positive or negative, being a two-quadrant one, but the direction of current is in the direction as

shown in the circuit, as only thyristors − unidirectional switching devices, are used in the two

converters. Normally, the ripple content in the output voltage is neglected.

eP = Em sin ot

iP iN

eN = Em sin ot

Positive (P)

converter Negative (N)

converter

er = Er sin ot

Fig. 29.1: Equivalent circuit of cyclovonverter

The control principle used in an ideal cyclo-converter is to continuously modulate the firing

angles of the individual converters, so that each produces the same sinusoidal (ac) voltage at its

output terminals. Thus, the voltages of the two generators (Fig. 29.1) have the same amplitude,

frequency and phase, and the voltage of the cyclo-converter is equal to the voltage of either of

these generators. It is possible for the mean power to flow either ‘to’ or ‘from’ the output

terminals, and the cyclo-converter is inherently capable of operation with loads of any phase

angle − inductive or capacitive. Because of the uni-directional current carrying property of the

individual converters, it is inherent that the positive half-cycle of load current must always be

carried by the positive converter, and the negative half-cycle by the negative converter,

regardless of the phase of the current with respect to the voltage. This means that each two-

quadrant converter operates both in its rectifying (converting) and in its inverting region during

the period of its associated half-cycle of current.

The output voltage and current waveforms, illustrating the operation of an ideal cyclo-

converter circuit with loads of various displacement angles, are shown in Fig. 29.2. The

displacement angle of the load (current) is 0 (Fig. 29.2a). In this case, each converter carries

the load current only, when it operates in its rectifying region, and it remains idle throughout the whole period in which its terminal voltage is in the inverting region of operation. In Fig. 29.2b,

the displacement angle of the load is 60 lagging. During the first 120 period of each half-

cycle of load current, the associated converter operates in its rectifying region, and delivers

Control Circuit

-


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power to the load. During the latter 60 period in the half-cycle, the associated converter


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operates in its inverting region, and under this condition, the load is regenerating power back into

the cyclo-converter output terminals, and hence, into the ac system at the input side. These two

are illustrative cases only. Any other case, say capacitive load, with the displacement angle as

leading, the operation changes with inverting region in the first period of the half-cycle as per

displacement angle, and the latter period operating in rectifying region. This is not shown in Fig.

29.2, which can be studied from a standard text book.


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Single-phase to Single-phase Cyclo-converter

The circuit of a single-phase to single-phase cyclo-converter is shown in Fig. 29.3. Two full-

wave fully controlled bridge converter circuits, using four thyristors for each bridge, are

connected in opposite direction (back to back), with both bridges being fed from ac supply (50

Hz). Bridge 1 (P – positive) supplies load current in the positive half of the output cycle, while

bridge 2 (N – negative) supplies load current in the negative half. The two bridges should not

conduct together as this will produce short-circuit at the input. In this case, two thyristors come

in series with each voltage source. When the load current is positive, the firing pulses to the

thyristors of bridge 2 are inhibited, while the thyristors of bridge 1 are triggered by giving pulses

at their gates at that time. Similarly, when the load current is negative, the thyristors of bridge 2

are triggered by giving pulses at their gates, while the firing pulses to the thyristors of bridge 1

are inhibited at that time. This is the circulating-current free mode of operation. Thus, the firing

angle control scheme must be such that only one converter conduct at a time, and the change

over of firing pulses from one converter to the other, should be periodic according to the output

frequency. However, the firing angles the thyristors in both converters should be the same to

produce a symmetrical output.

1-phase

ac

supply

1-phase

ac

supply

Bridge 1 (Positive) Bridge 2 (Negative)

Fig. 29.3: Single-phase to single-phase cycloconverter (using thyristor bridges)

When a cyclo-converter operates in the non-circulating current mode, the control scheme is

complicated, if the load current is discontinuous. The control is somewhat simplified, if some

amount of circulating current is allowed to flow between them. In this case, a circulating current

limiting reactor is connected between the positive and negative converters, as is the case with

dual converter, i.e. two fully controlled bridge converters connected back to back, in circulating-

current mode. The readers are requested to refer to any standard text book. This circulating

current by itself keeps both converters in virtually continuous conduction over the whole control

range. This type of operation is termed as the circulating-current mode of operation. The

operation of the cyclo-converter circuit with both purely resistive (R), and inductive (R-L) loads

is explained.

iP iN

P1 P + iO

2 N1 N2

eO

-

P4 P3 N4 N3

l

o

a

d


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Resistive (R) Load: For this load, the load current (instantaneous) goes to zero, as the input

voltage at the end of each half cycle (both positive and negative) reaches zero (0). Thus, the


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0 5

2 3 4 6

3

3

conducting thyristor pair in one of the bridges turns off at that time, i.e. the thyristors undergo

natural commutation. So, operation with discontinuous current (Fig. 29.4) takes place, as current

flows in the load, only when the next thyristor pair in that bridge is triggered, or pulses are fed at

respective gates. Taking first bridge 1 (positive), and assuming the top point of the ac supply as

positive with the bottom point as negative in the positive half cycle of ac input, the odd-

numbered thyristor pair, P1 & P3 is triggered after phase delay (1 ), such that current starts

flowing through the load in this half cycle. In the next (negative) half cycle, the other thyristor

pair (even-numbered), P2 & P4 in that bridge conducts, by triggering them after suitable phase

delay from the start of zero-crossing. The current flows through the load in the same direction,

with the output voltage also remaining positive. This process continues for one more half cycle

(making a total of three) of input voltage ( f1 = 50 Hz). From three waveforms, one combined

positive half cycle of output voltage is produced across the load resistance, with its frequency

being one-third of input frequency ( f2 = f1 / 3 = 16 2 Hz). The following points may be noted.

The firing angle () of the converter is first decreased, in this case for second cycle only, and

then again increased in the next (third) cycle, as shown in Fig. 29.4b. This is, because only three

cycles for each half cycle is used. If the output frequency needed is lower, the number of cycles

is to be increased, with the firing angle decreasing for some cycles, and then again increasing in

the subsequent cycles, as described earlier.

es

Mean output

e0 voltage

(a)

0 α2 α3 4 α 5 6

α1 2 3

(b)

5

α4 α6

Fig. 29.4: Input (a) and output (b) voltage waveforms of a cyclo-

converter with an output frequency of 16 2 Hz for resistive

(R) load

To obtain negative output voltage, in the next three half cycles of input voltage, bridge 2 is

used. Following same logic, if the bottom point of the ac supply is taken as positive with the top

point as negative in the negative half of ac input, the odd-numbered thyristor pair, N1 & N3

conducts, by triggering them after suitable phase delay from the zero-crossing. Similarly, the

even-numbered thyristor pair, N2 & N4 conducts in the next half cycle. Both the output voltage

and current are now negative. As in the previous case, the above process also continues for three

consecutive half cycles of input voltage. From three waveforms, one combined negative half

cycle of output voltage is produced, having same frequency as given earlier. The pattern of firing


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angle − first decreasing and the increasing, is also followed in the negative half cycle. One

positive half cycle, along with one negative half cycle, constitute one complete cycle of output


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2 3 4 5 6 7 8

3 (load) voltage waveform, its frequency being 16 2 Hz as stated earlier. The ripple frequency of

the output voltage/ current for single–phase full-wave converter is 100 Hz, i.e., double of the

input frequency. It may be noted that the load (output) current is discontinuous (Fig. 29.4c), as

also load (output) voltage (Fig. 29.4b). The supply (input) voltage is shown in Fig. 29.4a. Only

one of two thyristor bridges (positive or negative) conducts at a time, giving non-circulating

current mode of operation in this circuit.

Inductive (R-L) Load: For this load, the load current may be continuous or discontinuous

depending on the firing angle and load power factor. The load voltage and current waveforms are

shown for continuous and discontinuous load current in Fig. 29.5 and 29.6 respectively.

es

0

Mean o/p output 0

(a)

Mean o/p output voltage

i0

(b)

0 α1 α2 α

(c)

Fig. 29.5: Input (a) and output (b) voltage, and current

(c) waveforms for a cyclo-converter with discontinuous

voltage

5 6 7 0 2 3 4 8

3

e


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es

0

e0

(b)

i0

0

i0

0

Fig. 29.6: Input (a) and output (b) voltage, and current (c, d) waveforms

for a cyclo-converter with continuous load current.

(a) Discontinuous load current

The load current in this case is discontinuous, as the inductance, L in series with the

resistance, R, is low. This is somewhat similar to the previous case, but difference also exists as

described. Here, also non-circulating mode of operation takes place, with only one of the bridges

− #1 (positive), or #2 (negative), conducting at a time, but two bridges do not conduct at the

same time, as this will result in a short circuit. In this case, the output frequency is assumed as

( f 2 = 12.5 Hz), the input frequency being same as ( f1 = 50 Hz), i.e., f1 = 4 f 2 , or f 2 = f1 / 4 .

So, four positive half cycles, or two full cycles of the input to the full-wave bridge converter

(#1), are required to produce one positive half cycle of the output waveform, as the output

frequency is one-fourth of the input frequency as given earlier. As in the previous case with

resistive load, taking bridge 1, and assuming the top point of the ac supply as positive, in the

positive half cycle of ac input, the odd-numbered thyristor pair, P1 & P3, is triggered after phase

(c)

Mean o/p output voltage

(a)

0 5 6 7 8

α1

2 3 4

α2 α3 α4 α5 α6 α7 α8

(d)

2 3 4 5 6 7 8


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delay ( = t = 1 ), such that current starts flowing the inductive load in this half cycle. But


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here, the current flows even after the input voltage has reversed (after = ), till it reaches zero

at ( = 1 ) with ( + 2) 1 , due to inductance being present in series with resistance, its

value being low. It may be noted that the thyristor pair is, thus, naturally commutated. In the next

(negative) half cycle, the other thyristor pair (even-numbered), P2 & P4, is triggered at ( + 2) .

The current flows through the load in the same direction, with the output voltage also remaining

positive. The current goes to zero at ( + 2) , with ( + 3) 2 . This procedure continues

for the next two half cycles, making a total of four positive half cycles. From these four

waveforms, one combined positive half cycle of output voltage is produced across the inductive

load. The firing angle () of the converter is first decreased, in this case for second half cycle

only, kept nearly same in the third one, and finally increased in the last (fourth) one, as shown in

Fig. 29.5b.

To obtain negative output voltage, in the next four half cycles of output voltage, bridge 2 is

used. Following same logic, if the bottom point of the ac supply is taken as positive in the

negative half of ac input, the odd-numbered thyristor pair, N1 & N3 conducts, by triggering them

after phase delay ( = 4 + 1 ). The current flows now in the opposite (negative) direction

through the inductive load, with the output voltage being also negative. The current goes to zero

at ( 4 + 1 ), due to load being inductive as given earlier. Similarly, the even-numbered

thyristor pair, N2 & N4 conducts in the next half cycle, after they are triggered at ( 5 + 2 ). The

current goes to zero at ( 5 + 2 ). Both the output voltage and current are now negative. As in

the previous case, the above process also continues for two more half cycles of input voltage,

making a total of four. From these four waveforms, one combined negative half cycle of output

voltage is produced with same output frequency. The pattern of firing angle − first decreasing and then increasing, is also followed in the negative half cycle. It may be noted that the load

(output) current is discontinuous (Fig. 29.5c), as also load (output) voltage (Fig. 29.5b). The

supply (input) voltage is shown in Fig. 29.5a. One positive half cycle, along with one negative

half cycle, constitute one complete cycle of output (load) voltage waveform, its frequency being

12.5 Hz as stated earlier. The ripple frequency remains also same at 100 Hz, with the ripple in

load current being filtered by the inductance present in the load.

(b) Continuous load current

As given above, the load current is discontinuous, as the inductance of the load is low. If the

inductance is increased, the current will be continuous. Most of the points given earlier are

applicable to this case, as described. To repeat, non-circulating mode of operation is used, i.e.,

only one of the bridges − #1 (positive), or #2 (negative), conducts at a time, but two bridges do

not conduct at the same time, as this will result in a short circuit. Also, the ripple frequency in the voltage and current waveforms remains same at 100 Hz. The output frequency is one-fourth of

input frequency (50 Hz), i.e., 12.5 Hz. So, for each half-cycle of output voltage waveform, four

half cycles of input supply are required. Taking bridge 1, and assuming the top point of the ac

supply as positive, in the positive half cycle of ac input, the odd-numbered thyristor pair, P1 &

P3, is triggered after phase delay ( = t = 1 ), such that current starts flowing the inductive

load in this half cycle. But here, the current flows for about one complete half cycle, i.e., up to

the angle, ( + 1 ) or ( + 2 ), whichever is higher, even after the input voltage has reversed,

due to the high value of load inductance. In the next (negative) half cycle, the other thyristor pair

(even-numbered), P2 & P4, is triggered at ( + 2) . At that time, reverse voltage is applied across


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each of the conducting thyristors, P1/P3, and the thyristors turn off. The current flows through the

load in the same direction, with the output voltage also remaining positive. Also, the current


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flows for about one complete half cycle, i.e., up to the angle, ( + 2 ) or ( + 3 ), whichever is

higher. This procedure continues for the next two half cycles, making a total of four positive half

cycles. From these four waveforms, one combined positive half cycle of output voltage is

produced across the inductive load. The firing angle () of the converter is first decreased, in this

case for second half cycle only, kept nearly same in the third one, and finally increased in the last

(fourth) one, as shown in Fig. 29.6b.

To obtain negative output voltage, in the next four half cycles of output voltage, bridge 2 is

used. Following same logic, if the bottom point of the ac supply is taken as positive in the negative half of ac input, the odd-numbered thyristor pair, N1 & N3 conducts, by triggering them

after phase delay ( = 4 + 1 ). The current flows now in the opposite (negative) direction

through the inductive load, with the output voltage being also negative. The current flows for

about one complete half cycle, i.e., up to the angle, ( 5 . + 1 ) or ( 5 + 2 ), whichever is

higher, as the load is inductive. Similarly, the even-numbered thyristor pair, N2 & N4 conducts in

the next half cycle, after they are triggered at ( 5 + 2 ). As described earlier, both the

conducting thyristors turn off, as reverse voltage is applied across each of them. Both the output

voltage and current are now negative. Also, the current flows for about one complete half cycle,

i.e. up to the angle, ( 5 + 2 ) or ( 5 + 3 ), whichever is higher. As in the previous case, the

above process also continues for two more half cycles of input voltage, making a total of four.

From these four waveforms, one combined negative half cycle of output voltage is produced

with same output frequency of 12.5 Hz. The pattern of firing angle − first decreasing and then

increasing, is also followed in the negative half cycle. It may be observed that the load (output)

current is continuous (Fig. 29.6c), as also load (output) voltage (Fig. 29.6b). The load (output)

current is redrawn in Fig. 29.6d, under steady state condition, while the supply (input) voltage is

shown in Fig. 29.6a. One positive half cycle, along with one negative half cycle, constitute one

complete cycle of output (load) voltage waveform.

Advantages and Disadvantages of Cyclo-converter

Advantages

1. In a cyclo-converter, ac power at one frequency is converted directly to a lower frequency in

a single conversion stage.

2. Cyclo-converter functions by means of phase commutation, without auxiliary forced commutation circuits. The power circuit is more compact, eliminating circuit losses

associated with forced commutation.

3. Cyclo-converter is inherently capable of power transfer in either direction between source

and load. It can supply power to loads at any power factor, and is also capable of

regeneration over the complete speed range, down to standstill. This feature makes it

preferable for large reversing drives requiring rapid acceleration and deceleration, thus suited

for metal rolling application.

4. Commutation failure causes a short circuit of ac supply. But, if an individual fuse blows off,

a complete shutdown is not necessary, and cyclo-converter continues to function with

somewhat distorted waveforms. A balanced load is presented to the ac supply with

unbalanced output conditions.


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5. Cyclo-converter delivers a high quality sinusoidal waveform at low output fre-quencies,

since it is fabricated from a large number of segments of the supply waveform. This is often

preferable for very low speed applications.

6. Cyclo-converter is extremely attractive for large power, low speed drives.

Disadvantages

1. Large number of thyristors is required in a cyclo-converter, and its control circuitry becomes

more complex. It is not justified to use it for small installations, but is economical for units

above 20 kVA.

2. For reasonable power output and efficiency, the output frequency is limited to one-third of

the input frequency.

3. The power factor is low particularly at reduced output voltages, as phase control is used with

high firing delay angle.

3-phase

ac supply

Converter +

-

Vdc

Inverter

3-phase

output

Fig. 29.7: DC link converter

The cyclo-converter is normally compared with dc link converter (Fig. 29.7), where two

power controllers, first one for converting from ac input at line frequency to dc output, and the

second one as inverter to obtain ac output at any frequency from the above dc input fed to it. The

thyristors, or switching devices of transistor family, which are termed as self-commutated ones,

usually the former, which in this case is naturally commutated, are used in controlled converters

(rectifiers). The diodes, whose cost is low, are used in uncontrolled ones. But now-a-days,

switching devices of transistor family are used in inverters, though thyristors using force

commutation are also used. A diode, connected back to back with the switching device, may be a

power transistor (BJT), is needed for each device. The number of switching devices in dc link

converter depends upon the number of phases used at both input and output. The number of

devices, such as thyristors, used in cyclo-converters depends on the types of connection, and also

the number of phases at both input and output. It may be noted that all features of a cyclo-

converter may not be available in a dc link converter. Similarly, certain features, like Pulse

Width Modulation (PWM) techniques as used in inverters and also converters, to reduce the

harmonics in voltage waveforms, are not applied in cyclo-converters. The various circuits used

and their operational aspects are discussed in detail in the next (last) module (#5) on DC to AC

Converters termed as Inverters.

Advantages and Disadvantages of DC Link Converter

Advantages

1. The output frequency can be varied from zero to rated value, with the upper frequency limit,

load


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being decided by the turn-off time of the switching devices, which is quite low due to the use

of transistors in recent time.


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2. .The control circuit here is simpler, as compared to that used in cyclo-converter.

3. It has high input power factor, if diode rectifier is used in the first stage. If phase-controlled

thyristor converter is used, power factor depends upon phase angle delay.

4. It is suitable for higher frequencies, as given earlier.

Disadvantages

1. The conversion is in two stages, using two power controllers − one as converter and other as

inverter, as stated earlier.

2. Forced commutation is required for the inverter, if thyristors are used, even though phase

control is used in converter, where natural commutation takes place.

3. The feature of regeneration is somewhat difficult, and also is involved to incorporate in a dc

link converter.

4. The output waveform of the inverter is normally a stepped one, which may cause non-

uniform rotation of an ac motor at very low frequencies (< 10 Hz). The distorted waveform

also causes system instability at low frequencies. This can be reduced by using PWM

technique as given earlier.

In this lesson, the first one in the second half of this module (#4), the cyclo-converter is first

introduced, along with the basic principle of operation. The circuit and the operation of single-

phase to single-phase cyclo-converter, with both resistive and inductive loads, are described in

detail, with voltage and current waveforms. The current is discontinuous, with resistive and

inductive (with low value of inductance) loads, but can be continuous, if the inductance is higher.

In the next lesson, the circuit and operation of three-phase to single-phase cyclo-converter,

followed by three-phase to three-phase one, will be described in detail.

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