Design and Construction of an EV Driveline Prototype With an Integrated Flywheel

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UPTEC ES10 013 Examensarbete 30 hp April 2010 Design and Construction of an EV Driveline Prototype with an Integrated Flywheel Nils Finnstedt

Transcript of Design and Construction of an EV Driveline Prototype With an Integrated Flywheel

Page 1: Design and Construction of an EV Driveline Prototype With an Integrated Flywheel

UPTEC ES10 013

Examensarbete 30 hpApril 2010

Design and Construction of an EV Driveline Prototype with an Integrated Flywheel

Nils Finnstedt

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Teknisk- naturvetenskaplig fakultet UTH-enheten Besöksadress: Ångströmlaboratoriet Lägerhyddsvägen 1 Hus 4, Plan 0 Postadress: Box 536 751 21 Uppsala Telefon: 018 – 471 30 03 Telefax: 018 – 471 30 00 Hemsida: http://www.teknat.uu.se/student

Abstract

Design and Construction of an EV Driveline Prototypewith an Integrated Flywheel

Nils Finnstedt

Research shows that flywheels have a significant potential for improving theperformance of EV (Electric Vehicle) drivelines. Flywheels can be used as powerbuffers that even out the energy flow between the primary energy storage device andthe EV traction motor. This improves the potential energy density and extends thelifetime of the primary energy storage device of the EV.

In this degree project a prototype of a flywheel-buffered driveline was constructed.The flywheel chosen was an electric motor/generator constructed at the Division ofElectricity at Uppsala University. Lead acid batteries were used as the primary energystorage device in the driveline and the traction motor was a DC-motor.

Two DC/DC buck converters were designed for the driveline. The first limited thecurrent from the batteries to the flywheel and the second controlled the power fromthe flywheel to the traction motor. Both converters were controlled bymicrocontrollers. The current limiter was controlled by a hysteresis controller andthe DC-motor power was regulated manually, under the constraint of a maximumcurrent PI-controller. The buck circuits were simulated in MATLAB Simulink prior totheir construction.

The performance of the driveline was satisfactory, despite the poor efficiency of theDC-motor. The results showed that the efficiency of the flywheel and the powerconverters was relatively high and that the flywheel had excellent power-bufferingproperties.

ISSN: 1650-8300, UPTEC ES10 013Examinator: Kjell PernestålÄmnesgranskare: Hans BernhoffHandledare: Janaína Gonçalves och Johan Lundin

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Sammanfattning Forskning har visat att svänghjul har potential att förbättra prestanda hos drivlinor för elfordon. Svänghjul kan användas för att jämna ut effektflödet mellan drivlinans primära energilager och dess drivmotor. Det primära energilagret kan vara optimerat för hög energitäthet istället för hög effekttäthet om effekten från det är utjämnat och maxeffekten reducerad. För batterier, som är det vanligaste primära energilagret för elfordon, ökar också livslängden och förlusterna sjunker om de kan leverera en konstant effekt istället för den varierande effekten med de höga maxströmmarna som drivmotorn kräver. Svänghjul är lämpliga att använda som energibuffertar i drivlinor för elfordon eftersom de både har goda effekthanterings- och energilagringsegenskaper. En prototyp av en elektrisk drivlina med ett integrerat svänghjul har designats och konstruerats i detta examensarbete. Detta har varit en del av ett större forskningsprojekt på Avdelningen för Elektricitetslära på Uppsala Universitet. Det svänghjul som användes i drivlinan var en specialdesignad elektrisk motor/generator som tidigare konstruerats på avdelningen. Drivlinan bestod förutom svänghjulet av blybatterier, en DC-motor som drivmotor, en last som mekaniskt belastade DC-motorn, samt ett antal elektriska kretsar för att kontrollera effektflödet mellan drivlinans olika delar. De elektriska kretsarna som designades i arbetet var två DC/DC-konverterare. Den ena hade syftet att begränsa strömmen fån batterierna till svänghjulets drivsystem och den andra att kontrollera effekten från svänghjulet till DC-motorn. Kretsarna designades något olika för att jämförelser av olika systemlösningar skulle kunna göras. Båda kretsarna kontrollerades av digitala mikrokontrollers. Kretsarna datorsimulerades innan de konstruerades. Bortsett från DC-motorns verkningsgrad, visade mätningar att drivlinans prestanda var god. Svänghjulet och dess drivkretsar visade sig ha relativt hög verkningsgrad och dess förmåga att jämna ut effektflödet från batterierna till drivmotorn var mycket god.

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Table of Contents 1 Introduction...................................................................................................................................... 1 2 Background ...................................................................................................................................... 3 3 Aims and objectives......................................................................................................................... 4 4 Method ............................................................................................................................................. 5 5 Theory .............................................................................................................................................. 6

5.1 Electric machines ...................................................................................................................... 6 5.1.1 DC-Machines ..................................................................................................................... 6 5.1.2 PM Synchronous Machines ............................................................................................... 7

5.2 Batteries .................................................................................................................................... 8 5.3 Power Converters...................................................................................................................... 8

5.3.1 DC/AC ............................................................................................................................... 8 5.3.2 AC/DC ............................................................................................................................... 8 5.3.3 DC/DC ............................................................................................................................... 9

5.4 Electronic components............................................................................................................ 10 5.4.1 Switches ........................................................................................................................... 10 5.4.2 Diodes .............................................................................................................................. 11 5.4.3 Microcontrollers............................................................................................................... 11 5.4.4 Gate Drivers ..................................................................................................................... 12

5.5 Snubber Circuits...................................................................................................................... 12 5.6 Control Methods ..................................................................................................................... 12

5.6.1 Hysteresis Control............................................................................................................ 12 5.6.2 PID Controlled PWM ...................................................................................................... 13

5.7 Current Limitation .................................................................................................................. 14 5.8 Power Quality ......................................................................................................................... 14 5.9 PCB Design............................................................................................................................. 14

6 Basic Choices of Design ................................................................................................................ 16 7 Simulations .................................................................................................................................... 18

7.1 Current limiter simulations ..................................................................................................... 18 7.2 Traction motor drive simulations............................................................................................ 20

8 Final Choices of Design................................................................................................................. 23 8.1 Current limiter design ............................................................................................................. 23

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8.2 Traction Motor Controller Design .......................................................................................... 25 9 Results............................................................................................................................................ 28

9.1 Power Converter Performance................................................................................................ 28 9.1.1 Current Limiter Performance........................................................................................... 28 9.1.2 Traction Motor Controller Performance .......................................................................... 30

9.2 Driveline Performance ............................................................................................................ 34 9.2.1 Steady State...................................................................................................................... 34 9.2.2 Drive Cycle ...................................................................................................................... 35

9.3 PCB......................................................................................................................................... 37 10 Discussion .................................................................................................................................... 39 11 Future Work ................................................................................................................................. 41 12 Conclusions.................................................................................................................................. 42 Acknowledgements........................................................................................................................... 42 References......................................................................................................................................... 43 Appendix........................................................................................................................................... 44

A. DC-motor dynamics................................................................................................................. 44 B. Power MOSFETs and IGBTs................................................................................................... 46 C. Snubber Design ........................................................................................................................ 48 D. Zieger-Nichols Rules ............................................................................................................... 51 E. Microcontroller codes............................................................................................................... 52

Current Limiter Code................................................................................................................ 52 Traction Motor Controller Code ............................................................................................... 53

F. Switch Losses ........................................................................................................................... 57

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Nomenclature C Capacitor Cb Snubber capacitor in parallel with batteryCd Snubber capacitor in parallel with diode Cs Snubber capacitor in parallel with switch Cw Drag Coefficient Kp Proportional PID-coefficient I Current Ia IC IL IR IS

Armature current Capacitor Current Inductor Current Resistive Load Current Switch Current

L Inductance M Torque R Resistance Ra Armature resistance Rs Snubber-resistor resistance Td Differential PID-coefficient Ti Integral PID-coefficient U VL

Voltage Inductor Voltage

VRVSfn

Resistive Load Current Switch Voltage Natural ringing frequency (Hertz)

fo Cut-off frequency n Motor speed

δΦ Magnetic flux in air gap

nω Natural ringing frequency (Radians) ζ Damping coefficient

Abbreviations AC Alternating Current CAD Computer-Aided Design DC Direct Current EMF Electro Motive Force ESR Effective Series Resistance EV Electric Vehicle FTP Federal Test Procedure IGBT Insulated Gate Bipolar Transistors MOSFET Metal-Oxide-Semiconductor Field-Effect TransistorsPCB Printed Circuit Board PI Proportional, Integral PID Proportional, Integral, Differential

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PM Permanent Magnet PWM Pulse Width Modulation RC Resistor, Capacitor RCD Resistor, Capacitor, Diode VRLA Valve Regulated Lead Acid

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1 Introduction One of the greatest challenges facing mankind today is to reduce the impact of human beings on the environment. Limiting climate change is essential for the future of human beings and other forms of life on this planet. One way of achieving this is to limit the use of fossil fuels. To do this as efficiently as possible, we need to both increase the percentage of alternative energy sources used, and to decrease our total energy consumption. One efficient method of decreasing our total energy consumption is to make new technology more energy efficient. The transport sector consumes large amounts of fossil fuels with low efficiency. It is, with the exception of trains, almost exclusively driven by combustion engines powered by fossil fuels. An alternative to the fossil fuel-powered combustion engine needs to be found to reduce the environmental impact from traffic. Electric motors are an alternative to combustion engines in vehicles. Electric motors can with high efficiency be powered from renewable energy sources and do not cause emissions where they are being used. Electric vehicles (EVs) have been manufactured as long as other types of automobiles [1]. The most common primary energy storage system used in EVs is batteries. The low energy density of batteries has, however, until recently prevented the EV from competing with combustion engine-powered vehicles. In recent years tremendous improvements have been made in terms of the energy density of batteries, but the range of EVs and battery lifetimes are still low compared to the range and performance of vehicles driven by conventional combustion engines. Research has shown that one way of increasing the lifetime and energy density of EV primary energy storages is to integrate power buffers in the drivelines, in the form of flywheels [2] [3]. A flywheel is a rotating mass that stores energy in the form of kinetic energy. The principle is that the average power of the EV’s traction motor is transferred to the flywheel from the primary energy storage, while the varying traction motor power can be provided by the flywheel if the vehicle accelerates. Power can also be transferred from the traction motor to the flywheel when the EV brakes. This principle prolongs the lifetime of batteries, as they can work at a smooth and optimal discharge rate and because the number of charge/discharge cycles can be reduced [4]. The energy density of the primary energy source can be increased by the flywheel, as it enables the primary energy source to be optimised for high energy density instead of for high maximum powers. Thus by integrating a flywheel in the driveline, the range of the EV can be extended. An electric machine is a suitable device for charging and discharging a flywheel in an EV driveline.

Simulations have shown that the average power needed to propel a car differs greatly from the instantaneous power needed [5]. Figure 1 shows the simulation results of the average and the instantaneous power (excluding internal losses) of a car with a mass of 1500kg, a dimensionless drag coefficient, Cw, of 1.35, and a frontal area of 1.73m2 when driving according to a standard FTP 75 (Federal Test Procedure) urban drive cycle. The fact that the average power is so much less than the maximum power indicates the advantages of incorporating a flywheel into the driveline.

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0 200 400 600 800 1000 1200 1400-30

-20

-10

0

10

20

30

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Time [s]

Pow

er [k

W]

Power - US urban drive cycle, FTP 75

Min = -25kW

Max = 33kW

Average = 2.2kW

Figure 1: Calculated power for US urban drive cycle

The amount of storable energy is an important design parameter for a flywheel. If the flywheel is only to be used as a power buffer, it does not have to be very large. Figure 2 shows simulations of the energy stored in a flywheel if it is charged/discharged with the powers illustrated in Figure 1 [5]. The fact that the difference between the maximum and minimum energy in the flywheel is only 0.34kWh means that the flywheel only needs to be able to store that amount of energy. A flywheel that can store more energy can, on the other hand, have the advantage that it can be used for buffering energy when the EV is being charged from the grid.

0 200 400 600 800 1000 1200 14000

0.5

1

1.5

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Time [s]

Ene

rgy

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rage

[kW

h]

0 200 400 600 800 1000 1200 14000

20

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]

Energy stored in flywheel and battery

Vehicle SpeedEnergy in BatteryEnergy in Flywheel

0.34 kWh

Figure 2: Energy in flywheel and battery

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It is that the flywheel has both high specific energy and specific power that makes it suitable as an EV power buffer. In comparison with batteries, flywheels can handle a lot more power, and in comparison with capacitors, they can store a lot more energy. Figure 3 shows that flywheels fill the gap between batteries and capacitors [4]. Low losses and a very long lifetime are two other advantages of flywheels.

Figure 3: Specific energy and power for electric energy storage devices

Purely mechanical flywheels have been used for a long time in many different applications. The technology of electrically powered flywheels for EV applications is, however, not especially mature technology. So far, only EV prototypes with integrated flywheels have been constructed and tested.

2 Background Research has since 2005 been carried out on energy storage in flywheels for EV applications at the Division of Electricity at Uppsala University. The novelty of the concept developed in Uppsala is that the driveline is divided into two voltage levels. The windings in the flywheel machine are arranged so that they divide the electric system into one high voltage level and one low voltage level, similar to an electric transformer. The advantage of this is that batteries and fuel cells, which can be used as primary energy storages for EVs, work intrinsically at low voltage levels while traction motors work more efficiently at a higher voltage level [5]. The two-voltage-level flywheel allows the batteries to be connected to the low voltage side of the flywheel and the traction motor to the high voltage side. The aim of the research project is to design the complete EV driveline prototype that is illustrated in Figure 4. So far two flywheel prototypes have been designed and constructed at Uppsala University. Until this degree project was carried out, the flywheels had, however, never been tested

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in a driveline. The tests that were performed previously used a grid-connected power supply as a primary power source for the flywheel and a variable resistor as a load.

AC/DC/ACEnergy storage(battery, fuel cellgas turbine etc.)

AC/DC

High voltageHigh power

Low voltageLow power

Motor/Generator

Flywheel

Figure 4: Driveline design

3 Aims and objectives The aim of this degree project was to design and construct an EV driveline prototype in which one of the flywheels designed at the Division of Electricity could be integrated. The driveline needed to include the eight parts presented in the block diagram in Figure 5 below.

Figure 5: Drive line components

The flywheel controller, the flywheel and the rectifier are the parts of the driveline that had been previously constructed by the research group. These parts were not changed, except that the flywheel controller circuit was replaced by a Printed Circuit Board (PCB) which was designed and mounted as a part of the degree project. Apart from this, a battery system and a traction motor with a braking load were selected and obtained, and a current limiting circuit and a motor controller circuit were designed and constructed. To limit the scope of the project, it was decided to make the driveline unidirectional, which means that power can only be transferred in one direction: from the battery towards the load. The driveline was designed with a limited budget and should be seen as a first prototype that can help the process of making a more advanced system in the future. The objective of the degree project was to construct a test driveline, which would enable interesting measurements to be made to give information about how a flywheel affects the dynamics of a driveline system. Earlier, only tests of the flywheel itself, and not an entire driveline system were made. A driveline is a complex system with many parts connected in series, and to understand its dynamics fully practical tests need to be made. Apart from the dynamics of the total system, the project can provide information about how power converter circuits should be designed. To demonstrate the performance of the system constructed, it was decided that a number of measurements should be made, once the construction was complete.

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4 Method This degree project was carried out according to the method presented below:

• Literature study within the relevant area

• Choice of a model to simulate

• Simulation of the model

• Evaluation of the simulations

• Design of a system to construct

• Construction of the system

• Measurements of the system performance

• Evaluation of the system

• Conclusions and observations This method was chosen as it was considered to be the most efficient way to achieve the aims of the project. A literature study provides the basic knowledge about how the different parts in a system work. Simulations give knowledge about the dynamics of the specific system to be constructed, so that a suitable design can be chosen for a prototype. Finally the construction of a prototype gives knowledge about how the system works in reality and how the parts interact with each other. A literature study and simulations save a lot of time and expense when constructing a complex technical system. The following report describes the way in which the method was carried out in practice, explains in further detail the motivation for the choices made and presents the results.

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5 TheoryThe degree project started with a literature study within the relevant area. This provided the background theory needed to choose a suitable design and to interpret the results. The relevant theory is summarised in the following section.

5.1 Electric machines There are several different types of electric motors and generators. Three types of machines commonly found as traction motors in EVs are synchronous motors, asynchronous motors and DC-motors [6]. As the work in this thesis has been focused on a DC-motor, and the flywheel in the driveline is a three phase synchronous AC-motor, the theory will focus on these two motor types.

5.1.1 DC-Machines As implied by their name, DC-motors can be fed directly from a DC power source. Apart from this, DC-motors are suitable as EV motors because they have a high starting torque and they are easy to speed control [7]. Like all rotating electric machines, the DC-motor has a rotor and a stator. A commutator that is integrated in the motor transmits the relative rotation between the magnetic field in the stator and the rotor in the DC-motor. In contrast to other types of motors, the DC-motor’s main magnetic flux is constant with respect to the stator, and rotating with respect to the rotor. The stator of a DC-motor contains field windings or field magnets if it is a permanent magnet (PM) machine. The rotor contains armature windings that are isolated from each other and placed in axial slots in the laminated rotor and connected to the commutator. The commutator consists of copper segments that are isolated from each other and are in contact with connectors on the motor via brushes. When the rotor rotates, the brushes slides across the different segments on the commutator so that different armature windings carry the armature current, which means that the magnetic field made by the armature windings shifts so that there is always a magnetic torque applied on the rotor. Figure 6 shows the different parts of a DC-motor.

Field windings

Brushes

Stator

Rotor Commutator

Figure 6: DC-motor DC-motors can be magnetised either by permanent magnets or by electro-magnets. The field windings in electrically magnetized DC-machines can be fed in different ways. They can be fed by

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a separate DC-source or by the same source as the armature, in parallel with the armature, in series or in a combination of both. The dynamics of the motor will depend on how the armature winding is fed. The traction motor used in the driveline constructed in this degree project is a compound motor. This means that there are two field windings; one is in series with the armature, called the series winding, and one in parallel with the armature and the series winding, called the shunt winding. More about how these windings affect the dynamics of DC-motors, and the dynamics of DC-motors in general, can be found in Appendix A. There are many ways of controlling the output of a DC-motor. The currents through the different windings in the motor, that decide the torque and speed, can be regulated both by connecting resistors in series with them or by regulating the supply voltage to the motor. The outputs can be controlled via the inputs, manually or automatically or in a combination of the two.

5.1.2 PM Synchronous Machines Synchronous electric machines are AC-powered machines in which the rotor rotates at the same speed as the magnetic field from the stator. All synchronous machines have magnets in their rotors. PM machines are magnetised by permanent magnets. Traditional PM synchronous machines, in contrast to traditional DC-machines, have the magnets in their rotor inside their stators. There are, however, other types of synchronous machines. In the driveline built in this degree project, the electric machine that works as a flywheel is an axial flux machine [8]. This means that the stator is placed axially between the two rotor discs that carry the permanent magnets. The stator is made of bakelite and carries two sets of windings; one high voltage winding and one low voltage winding. The machine can be seen in Figure 7.

Stator

Rotor

Figure 7: Flywheel machine

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5.2 Batteries Batteries are the most common energy storage devices used in EVs. There are several types of battery technologies that are suitable for different applications. Essential features of EV traction batteries are the following [9]: - High energy density - High charging and discharging power density - Long lifetime with maintenance-free and high safety mechanisms - Wide acceptance as a recyclable battery from an environmental standpoint - Price Four types of batteries commonly found as traction batteries in EVs are lead acid, nickel cadmium, nickel metal hydride and lithium-ion batteries. However, in this degree project only lead acid batteries were used. Lead acid batteries have a relatively low energy density (30-50Wh/kg), but are the most economical solution for larger power applications where weight is of little concern. In addition to being inexpensive, lead acid batteries are robust and the technology is well-established and used in many applications. There are two types of lead acid batteries: flooded lead acid batteries that require maintenance by periodic replenishment of distilled water, and valve regulated lead acid (VRLA) batteries that are maintenance free [10].

5.3 Power Converters Power converters are electronic systems that convert power flows from one current and voltage level to another. The following section will provide a theoretical discussion of the three different types of power converters that are involved in this degree project. The three converters used are based on the same components. They are composed of switches, diodes, drivers, capacitors, inductors, resistors and control systems. Capacitors and inductors are passive components that store energy in electric fields and magnetic fields, and are used to stabilise voltage and current.

5.3.1 DC/AC DC/AC converters, or inverters, convert direct power to alternating power. The DC/AC converter used in the driveline constructed in this degree project, powers the variable speed flywheel machine on its low voltage side. This means that it needs to vary both the output frequency and the voltage. The DC/AC converter used is controlled by a microcontroller and consists of an IGBT bridge and an output filter [11].

5.3.2 AC/DC AC/DC converters, or rectifiers, convert alternating power to direct power. The rectifier used in the driveline constructed in this degree project is used for extracting DC power from the three-phase flywheel machine on its high voltage side.

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The AC/DC converter used in the driveline is a full-wave passive three-phase rectifier. It is constructed with six power diodes that are connected in a bridge as shown in Figure 8 [12].

Diode bridge3-phase AC source

Load

Figure 8: Passive rectifier

When the rectifier is fed by a three-phase AC current the diodes that conduct are the two connected to the phases with the momentarily highest and lowest phase voltages. The output DC voltage from the rectifier can be smoothened by capacitors in parallel with the load.

5.3.3 DC/DC As implied by the name, DC/DC converters are power converters that convert DC power at one level of current and voltage to another. Some converters increase the voltage and decrease the current, while others do the opposite. There are also DC/DC converters that both can increase and decrease voltage depending on how they are controlled. The types of DC/DC converters designed in this degree project are step-down converters.

Buck Converters Buck converters, or step-down converters, are converters from which one can obtain an output signal (current or voltage) which is lower than the input. How they work can be understood by studying Figure 9 that shows a circuit diagram of a buck converter.

DC source

Switch

LoadDiode

Inductor

Capacitor

Figure 9: Buck converter

When the switch of the converter is conducting, the inductor current will increase and the inductor will produce a negative voltage as it builds up energy in a magnetic field. When the switch opens, the energy in the inductor will produce a positive voltage that will power the load so that the current flows through the diode instead of the power source. The capacitor filters the output voltage. The output voltage of the buck circuit increases when the switch is closed and decreases when it is open. If the inductor current never goes to zero, the average output voltage will be a linear function of the average ratio of the switch position [13]. For example, if the switch is closed 50 % of the

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total switching period, the average voltage across the load will be 50 % of the power source voltage. Figure 10 shows typical buck converter waveforms. It shows a sequence where the converter switch switches from its on-position to its off-position two times. VL is the voltage across the inductor, IL is the current through the inductor, VR is the voltage across the resistive load, IR is the current through the resistive load, VS is the voltage across the switch, IS is the current through the switch and IC is the current through the capacitor.

Figure 10: Buck converter waveforms

5.4 Electronic components Electric systems are built of many different connected parts. In this section the electronic devices used in the power converters built in this degree project will be examined.

5.4.1 Switches Discrete power switches are essential components of power converters. There are many types of switches, all of which have different advantages and disadvantages. All switches used in switched mode power converters are semiconductors. The reason for this is that they have much shorter switch times than any mechanical switch, which is of great importance in reducing switching

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losses. Values that are important when selecting a switch can be summarised in the following list [14]: - Maximum current carrying capability - Maximum voltage blocking capability - Forward voltage drop during ON and its temperature dependency - Leakage current during OFF - Thermal capacity - Switching transition times during both turn-on and turn-off - Capability to stand dV/dt when switch is OFF or during turn-off - Capability to stand dI/dt when switch is ON or during turn-ON - Controllable dI/dt or dV/dt capability during switching transition - Ability to withstand both high current and voltage simultaneously - Switching losses - Control power requirement and control circuit complexity As mentioned, switching losses occur when switches have long switching times. Ideally, a discrete switch has no losses. During the switching process, however, for a short period of time some current will pass through the switch and there will be a voltage drop across it simultaneously. The power that is burnt off during the switching processes defines the switching losses. The longer time the switching process takes, and the more often it happens, the larger the losses will be. All types of switches are controlled by control signals, some based on current and some on voltage. Both types of switches used in this degree project, MOSFETs and IGBTs, are controlled by voltage. When a certain positive voltage is applied to their control input, known as gate, relative to their cathode, they will conduct. If the gate is at the same or lower potential than the cathode, the switch will block the current from running from the anode to the cathode. A comparison between IGBTs and MOSFETs can be found in Appendix B. The switching time of a voltage-controlled switch is determined by the time it takes for the gate voltage to rise and fall. Even though both MOSFETs and IGBTs have electrically isolated gates, it takes some time to change their potential, as they have built-in parasitic capacitors, both between the gate and the anode and between the gate and the cathode. The switching time is the time it takes to charge or discharge these capacitors. To achieve a short switching time, a switch with a low gate charge, which is the charge that is required to charge both the gate capacitors, should be chosen.

5.4.2 Diodes Diodes are passive semiconductors that conduct current in one direction and block current in the opposite direction. Power diodes are useful in applications such as rectifying circuits and to provide current paths for inductive loads [15]. Diodes conduct when the voltage of the anode, in relation to the cathode exceeds a certain voltage named knee voltage, often around 0.7V [15]. If the voltage is increased above this level, the conducting current increases dramatically.

5.4.3 Microcontrollers The switches in a power converter system need a system that controls their operations. If the power flow in the converter is to be automated, an automatic control system needs to be implemented. A

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control system needs components that can interpret the system outputs and transform them into an input signal. A suitable component for this is the microcontroller. A microcontroller is a small computer on a single integrated circuit. It contains a central processing unit, a clock, I/O ports, and a memory. The microcontroller can be programmed, which gives great freedom to the user. Suitable microcontrollers for power switch control also have analogue-to-digital converters that make it possible for them to interpret analogue measurement signals.

5.4.4 Gate Drivers Gate drivers are the interface between control systems and high power electronic systems. As switching losses depend on the time taken to charge and discharge the gate capacitors of power switches, they should not be driven directly by the logic outputs from devices such as microcontrollers. A gate-driving circuit for a voltage-controlled switch is a circuit that is made for injecting or removing the gate charge fast. The larger currents the driver can handle, the faster the gate charge can be injected or removed and the more efficient the power circuit will be. It is therefore of great importance for the efficiency of the power system to have a well-designed gate driver circuit [16].

5.5 Snubber Circuits Snubber circuits, or snubbers, are small circuits that are added in many power converter circuits to protect sensitive semiconductor devices from being damaged. Snubbers also fulfill other tasks. The improvements snubber circuits can make to a power converter system are summarised in the following list [17]. - Reduce or eliminate voltage or current spikes - Limit dI/dt or dV/dt - Transfer power dissipation from the switch to a resistor or a useful load - Reduce total losses due to switching - Reduce electro magnetic interference by damping voltage and current ringing There are many different types of snubbers that are specialised for the different functions listed above. The snubber that has been constructed in this degree project has been optimised to fulfill two functions: to limit voltage spikes and to reduce ringing. A description of what causes these problems and how snubber circuits should be designed to solve them, can be found in Appendix C.

5.6 Control Methods There are several ways to control power systems. Different control strategies suit different power systems and different control parameters. In all switched mode circuits, the positions of the switches can be controlled. The two control strategies used in this degree project are hysteresis control and PI-controlled PWM.

5.6.1 Hysteresis Control Hysteresis control is a controlling strategy that is suitable for controlling current and voltage in buck converters. It works through measuring the output that is to be controlled and comparing it with a reference value. If the value of the output is larger than the reference value, the switch is turned off, and if it is smaller, the switch is turned on. To prevent the switch from chattering when

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it tries to keep the output at the reference value, a hysteresis dead band is created around the reference value [18]. This means that the switch is turned off when the output is slightly above the reference value and turns on when it is slightly below it. The larger the dead band is, the slower the switching frequency will be, at the expense of causing a larger ripple in the output. Hysteresis control is easy to implement and very robust. It gives an optimal input-output response time, and the overshoot is eliminated [18]. A drawback of the hysteresis control strategy is that it puts high demands on the output measurements. The measuring frequency needs to be a magnitude higher than the desired switching frequency and the measuring noise a magnitude smaller than the hysteresis band.

5.6.2 PID Controlled PWM Pulse width modulation (PWM) is a commonly used method for modulating switching devices. The basic idea is that a switching frequency is defined, and the duty ratio, which is defined as the time that the switch is on in relation to the PWM period, is varied to change a given system output. If the duty ratio is to be regulated automatically by some kind of feedback system from the output, a controlling function must be defined that relates the duty ratio to the error of the output. The simplest regulator is a P-regulator (proportional) that adjusts the duty ratio proportionally to the output error. More sophisticated regulators can also integrate the error and form a PI-regulator (proportional, integral), or can also take the derivative of the error in what is known as a PID (proportional, integral, differential) regulator. A lot can be said about the PID regulator. The basics can, however, be understood fairly easily. The P term of the regulator works by forming a control signal by multiplying the output error with a constant Kp. A large Kp constant gives a fast regulator, but can make the output oscillate. The effect of the P term of the regulator is large when the error is large, but small when the error declines. In many systems, the error does not converge to zero if only a P regulator is used. This is the reason why PI controllers are used. The I term of the regulator forms the control signal by multiplying the integral of the output error with a constant Ti. If the output error does not converge, the accumulated error grows and the PI controller reacts. The D term of the regulator is used to increase its speed without making the output oscillate. It works by multiplying the derivative of the output error with a constant Td. [19]. In order to get a good response from a P, PI or PID regulator it is essential to tune it. If a mathematical model of the regulated system can be derived, there are several analytical methods to find suitable values for the proportional, integrating and derivative constants. If a mathematical model cannot be found, e.g. if the system is very complex, there are several experimental methods that can be used to tune the regulator. One method that can be used is the Zieger-Nichols rules for tuning PID controllers [20]. This method can be found in Appendix D. One of the advantages of PWM is that the duty ratio does not have to be regulated as fast as the switching frequency. This means that the output can be sampled at lower frequency than that required for hysteresis control. It also means that there will be time to filter the measurements and consequently eliminate their noise.

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One of the drawbacks of PWM regulation is that it is not as fast as hysteresis control. If the speed of the PID regulator is increased by using a too large proportional constant, instability problems with large overshoots will occur [18].

5.7 Current Limitation When an electric motor is started, the current can exceed the maximum level that its power system can handle. This is due to that the back EMF, that normally limits the motor current, is low when the motor is staring as it is proportional to the speed of the motor. Because of this, many motor drives have some kind of current-limiting system that works when the motor shall be started. The easiest way to limit the current to a motor is to connect resistors in series with it. Another way is to make a buck circuit that decreases the voltage applied to the motor when the current level is too high. Using a buck circuit to limit the current limits the losses in comparison to using resistors.

5.8 Power Quality Ideally the power in electrical systems is delivered by single frequency AC or DC voltages and currents. In reality this is not the case. Voltages and the currents in all power systems contain both noise and harmonics. The content of power distortions in a system defines the power quality. The power quality issues that have been taken in account in this degree project are the current ripple created by the DC/DC-converters and the voltage spikes created when switches turn off. There are many reasons why high power quality is desirable. Current and voltage ripple leads to decreased efficiency and shortens the lifetime of power systems [21]. Many power electronic devices are sensitive to voltage distortion and can malfunction or shut down if the voltage quality is too bad. Moreover, voltage distortion leads to current distortion in all types of loads. Bad current quality leads, amongst other things, to increased losses. As resistive losses are given by I2R, the total resistive energy loss in a system will increase the more uneven the current is. In addition to this, due to the skin effect, the resistance in a conductor increases with the frequency. This means that the higher the frequency is, the more the current is concentrated to the surface of the conductor and less of its cross section area is utilized [21]. Another reason for reduced efficiency in systems with bad current quality is the fact that the magnetic flux will vary around the conductors, if the current in them varies. Varying magnetic flux leads to hysteresis losses and eddy losses in the iron in electric machines. The hysteresis losses are proportional to the frequency of magnetic flux variations and the eddy losses to the frequency squared [21]. There are many ways to improve the power quality in a power system. There are various types of filters, both passive and active, that can be implemented. Filters can, however, create losses themselves. In the case of switched DC-DC converters, increasing the switching frequency or the size of filter inductors can decrease the current ripple. The switching frequency and the size of the filter should be optimised to minimise the total losses in the system.

5.9 PCB Design A printed circuit board (PCB) is a board specially designed for a certain circuit, on which components can be soldered. The board consists of a substrate and copper traces that are printed on it. The substrate is an isolator that provides a structure that physically holds the circuit components and the printed wires in place [22]. The wires can be placed on both sides of the substrate, and also

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inside the substrate in multi-layer PCBs. The conductors in the different layers can be connected to each other through holes called vias. PCBs are manufactured industrially by photoplotters and CNC machines. A digital description of the board design is needed in order to manufacture it. This description is made by the PCB designer with a computer-aided design (CAD) program [22].

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6 Basic Choices of Design The first choices of system design were made based on the literature study, earlier experience, design criteria and practical limitations. The following section gives a description of and motivation for the basic choices that were made before any simulations of the system, or practical experiments were carried out. The parts of the driveline that these choices affected are presented in the list below:

• Battery • Current limiter • Traction motor controller • Traction motor • Load

The traction motor used was a compound DC-motor rated 10kW at 60V and 205A. This was chosen for practical reasons. The motor was originally bought for other purposes but fitted well in the system. No modifications were made to the motor, except for a mounting frame and a shaft coupling between the motor and its load. The mechanical load used to brake the motor was a DC-generator rated 1.9kW. The electrical load used to brake the generator was a variable resistor rated 0-630Ω. These loads were chosen mostly for practical reasons. Both the generator and the variable resistor were found in the lab where the driveline was built. Beside the convenience of using these loads, they were suitable in the driveline. Both the braking torque and the output power could easily be adjusted by regulating the field current in the generator and the resistance of the resistor. Lead acid batteries were chosen for powering the driveline. Four 12V batteries connected in series gave an output voltage of 48V. This was enough to bring the flywheel to a speed which gave a high voltage output of 80V. As the DC-motor was rated at 60V, this low voltage level was suitable. The batteries chosen were 45Ah gel batteries. Lead-acid batteries were chosen as they are relatively cheap, safe and easy to charge. 45Ah was a suitable value for the energy content of the batteries, as it allows the flywheel to be run for the time it takes to do the tests likely to be done during one day (~4 hours at 500W average input power). Gel batteries which are a type of VRLA batteries were chosen as they are maintenance free. Two buck circuits were chosen for the current limier and the DC-motor controller. Buck circuits were chosen as they could perform what was required of the two power converter systems and as they are efficient. The control systems were based on microcontrollers, as they are cheap, fast and robust. The current and the voltage quality are important design parameters of a power system. Finding the optimal level of the power quality needs a lot of measurements. These measurements could not be made before the driveline had been constructed. However the power quality had to be set to some level for the power converters to be designed for. A current ripple of 0.5A was chosen as a suitable value to aim for. This value can easily be changed in the future when an investigation of the losses has been made.

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Table 1 summarises the basic choices of design that were made in the beginning of the project. Table 1: Basic choices of design Battery VRLA batteries 48V Power converters Buck converters Power quality 0.5A maximum DC current ripple Traction motor Compound DC-motor Driveline load DC generator with variable load resistor

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7 Simulations After a literature study on the types of power circuits required in the driveline, and a decision about their topography, the circuits were simulated on a computer. The aim of the simulations was to understand the dynamics of the system and to discover what results to expect when certain values of the components in the system were changed. The simulations also provided an indication of which components should be ordered. This saved both time and money. The simulations were made in MATLAB Simulink, which is a toolbox for MATLAB made for modelling and simulating dynamic systems. The programming interface is graphical and consists of block programming tools. The method used to make the simulations of the buck circuits was to control the switches by feedback loops from current-measuring device in the system. The current measurements gave an output signal that was converted to a logic input for the switch via a hysteresis control block. This had the advantage that for a certain current quality, given by the hysteresis band width, and a certain current, given by the hysteresis band level, the switching frequency could be found. If the frequency was too high, larger inductors could be added to the system, as they make the current level more stable.

7.1 Current limiter simulations To reduce the complexity of the simulations some simplifications of the system were made. The load for the current limiter was in reality the flywheel, driven by the DC/AC-converter. This was approximated as a constant resistive load, with the same resistance as the windings of the flywheel. In reality, the load had inductance and a back EMF, which were ignored here. The inductance of the flywheel is, however, small (0.19mH), as it is coreless and the back EMF is small in the start-up phase, when the current limiter will be used. As the inductance of the windings and the back EMF make the current level more stable, the approximation can be seen as a worst case scenario. Designing the current limiter so that it can limit the current without being dependent on inductance from the load increased the reliability of the system. Other simplifications made include the imperfections in the used components. For example the reverse recovery time of diodes and parasitic inductances in conductors were ignored. These are very important factors for the system, but as the simulations were not used for studying transients, these simplifications were possible. Many different simulations were made and the model used is shown in Figure 11.

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Figure 11: Simulink current limiter model

Figure 12 and Figure 13 show two different plots from a simulation of the current limiter. Figure 12 shows the control signal to the switch and the inductor current measured by the system during the beginning of a simulation. Figure 13 shows the load current and load voltage during the same time period and simulation as in Figure 12. The simulation is made for a hysteresis band between 9.75A and 10.25A, a source voltage of 48V, an inductor of 3.75mH and a capacitor of 4.7mF.

0.5 1 1.5 2 2.5

x 10-3

9

9.5

10

10.5Hysteresis controlled current

Mea

sure

d cu

rrent

(A)

Time (sec)0.5 1 1.5 2 2.5

x 10-3

0

1

Sw

itch

cont

rol s

igna

l

Figure12: Inductor current and control signal

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0.5 1 1.5 2 2.5

x 10-3

8.8

9

9.2

9.4

9.6

9.8

10

10.2Load current and voltage

Load

cur

rent

(A)

Time (sec)0.5 1 1.5 2 2.5

x 10-3

11

11.5

12

12.5

13

Load

vol

tage

(V)

Figure 13: Load current and voltage

The simulations showed that the current ripple through the resistive load was smallest if the current measuring device was placed between the diode and the capacitor. If the device was between the capacitor and the load, the current ripple was the same as the hysteresis band. If it was placed before the capacitor, the capacitor evened out the current and decreased the ripple. The simulations also showed that the greater the filter capacitance was, the smaller the voltage ripple across the load was, up to the level when the capacitor current increased and decreased relatively linearly with respect to the time. Increasing its size of the capacitor above this level did not improve the power quality significantly. The smaller the hysteresis band used, the smaller the capacitor needed to be. Figure 12 and Figure 13 show that the switching frequency of the current limiter needs to be around 4.9 kHz to keep the inductor current ripple within 0.5A.

7.2 Traction motor drive simulations Simplifications were also made in the simulations of the DC-motor controller. The motor was simplified by two parallel current paths with the impedance of the DC-motor’s shunt winding and its series and armature winding. As in the current limitation circuit, the back EMF and the imperfections in the components were ignored. The inductance of the motor was, however, not ignored as it is much larger than in the flywheel. The simplifications were motivated in the same way as regards the current limiter. As the motor was simplified to its equivalent impedance, no control system based on the output of the motor could be simulated. This was the reason why the simulations were made by controlling the input to the motor. The current was controlled in the same way as in the simulations of the current limiting circuit. However, this was not a problem, as the objective of the simulations was to give information about the power converter system, not the motor output dynamics. Simulations were performed both with and without an extra inductor in the buck circuit. As the motor is an

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inductive load, an extra inductor was not strictly required. Another difference between the current controller buck circuit and the motor control buck circuit was the filter capacitor. The model used for the motor controller simulations is shown in Figure 14.

Figure 14: Simulink DC-motor controller model

Figure 15 and Figure 16 show two different plots of a simulation of the motor control power system. Fig15 shows the control signal to the switch and the current measured by the control system during the beginning of a simulation. Figure 16 shows the armature current and voltage during the same time period as in Figure 15. The simulations were made for a hysteresis band between 19.75A and 20.25A, a source voltage of 60V without an extra inductor in the buck circuit.

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0.6 0.8 1 1.2 1.4 1.6 1.8 2 2.2 2.4

x 10-4

19

19.5

20

20.5Hysteresis controlled curent

Mea

sure

d cu

rrent

(A)

Time (sec)0.6 0.8 1 1.2 1.4 1.6 1.8 2 2.2 2.4

x 10-4

0

1

Sw

itch

cont

rol s

igna

l

Figure 15: Total motor current and control signal

0.6 0.8 1 1.2 1.4 1.6 1.8 2 2.2 2.4

x 10-4

19

19.5

20

20.5Armature current and voltage

Arm

atur

e cu

rrent

(A)

Time (sec)0.6 0.8 1 1.2 1.4 1.6 1.8 2 2.2 2.4

x 10-4

-10

0

10

20

30

40

50

Arm

atur

e vo

ltage

(V)

Figure 16: Armature current and voltage

The simulations showed that it was not suitable to use filter capacitors anywhere in the motor control buck circuit. As all parts of the load are inductive, it was not suitable to add a filter capacitor in parallel with any part of the load. Connecting a capacitor in parallel with an inductor causes a resonance phenomenon that is associated with increased losses. Figure 15 and Figure 16 show that the switching frequency of the DC-motor controller needs to be around 34 kHz to keep the current ripple within 0.5A without any filter inductor or capacitor in the buck circuit.

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8 Final Choices of Design After the computer-based simulations, practical experiments were made. Before the experimental process started further designing parameters were chosen. These decisions were necessary prior to starting the experiments, as a lot of components needed to be ordered. Some of the decisions were motivated by the simulations and some by practical reasons. As the designing of the driveline was part of a research project, some choices were motivated by the fact that interesting conclusions could be drawn by choosing and evaluating certain system solutions. As two buck-converters were constructed, different system solutions could be made for them both and their function could be compared. The following section describes the choices that were made after the simulations were completed and during the experimental process. Table 2 summarises the system solutions for the two buck-converters. Table 2: Final choices of design Part of the Power Converter Current Limiter Traction Motor Controller Switch IGBT MOSFET Buck output filter Diode, inductor, capacitor Diode Snubber circuit No Yes Switch driver IR2110 IR2110 Microcontroller dsPIC30F 2010 dsPIC30F 2010 Current sensor Hall Effect sensor Hall Effect sensor Control strategy Hysteresis control PWM, PI maximum-current control

8.1 Current limiter design The current limiter was the first of the two DC/DC converters to be designed. This fact affected its design as efforts were made not to make it too complex. If the first converter was simple, more time could be spent on the second converter, when experience from the first one had been gained. The switch that was used in the current limiting circuit was an IGBT. This was partly decided as the research group that the degree project was made for, had experience of the actual IGBT. Another reason for choosing an IGBT was that the current limiting switch only switches when the flywheel is in its start-up phase. The rest of the time the switch is continuously conducting. This means that the most important performance of the switch is its static conducting performance, which means that the IGBT is a more suitable switch than a MOSFET. IGBTs are also in general rated for higher voltages than MOSFETs. This makes the requirement for voltage-spike-reducing snubber-circuits smaller when using IGBTs, which was a reason for using an IGBT in the first converter to be designed. The actual IGBT that was used was a module containing two transistors, which meant that the body diode of one of the switches could be used as a protection diode in the buck circuit. The IGBT was a SKM600GB066D made by Semikron, rated at 600V and 690A (continuously). By studying the simulations it can be seen that if the current smoothening inductor in the current limiting circuit is 3.75mH, and the current ripple is set to a maximum of 0.5A, a switching frequency of 4.9kHz is required. This is a suitable switching frequency for the IGBT as it can be made both higher or lower if the specifications of the power quality are changed. On the basis of this conclusion, an inductance value of 3.75mH was used in the experimental setup. The filtering

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capacitor was set to 4.7mF, as the simulations showed that this was adequate. The inductors that were used were chosen as they were readily available and had suitable inductance values. The conductors of the inductors were, however, under dimensioned for the application. The IGBT driver and microcontroller were also chosen because of earlier experience. The microcontroller chosen was a dsPIC30F 2010 made by Microchip. It is a 16-bit digital signal controller with a 10-bit 1 Msps Analog-to-Digital Converter. The driver was an IR2110 made by International Retifier. It can deliver an output current of 2.5A and has a high side and a low side that can be used for driving two switches. However, only the low side was used in the setup. The driver voltage was set to 15V. Hysteresis control was used for the current controller. This was mainly motivated by the fact that it is fairly straightforward to implement. The code made for the hysteresis controller can be found in Appendix E. A Hall Effect current sensor was used. This was motivated by the fact that they have good accuracy and do not create excessive losses in the power circuit ass when measuring current with a shunt resistor. The current limiter was designed so that the level at which the current was limited could be adjusted with a potentiometer connected to the microcontroller. The current sensor was a HAL 50-s made by LEM. Figure 17 shows a circuit diagram and Figure 18 shows a photo of the current limiter circuit.

Input

Hysteresis Control System

ControllerFlywheel3.75 mH

L

IG

BT

4.7 mFC48V

BATTERY

Current Sensor

Figure 17: Current limiter circuit diagram

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Capacitor

Hysteresis Control System

Inductor

Current Sensor

IGBT

Figure 18: Photo of current limiter circuit

8.2 Traction Motor Controller Design Many of the design parameters chosen for the DC-motor controller differed from those chosen for the current limiter. This decision was made because interesting conclusions could be made by comparing different system solutions. One difference was that a MOSFET was chosen for the switch in the motor controller buck circuit. The choice was motivated by the fact that the switch in the motor driving circuit has to switch under all conditions, except for when maximum current is required, and therefore the dynamic performance of the switch is very important. Also, as a MOSFET was chosen, it was possible to have a sufficiently high switching frequency so that no extra inductor was required in the power converter. The chosen MOSFET was an IXFN140N20P made by IXYS. Its rating is 200V and 140A (continuously). As the MOSFET has a faster switching time than the IGBT and has a lower voltage rating, measurements showed that a snubber circuit was required in combination with the MOSFET. The first snubber to be designed was intended to reduce voltage peaks. Four 22μF polypropylene capacitors were connected in parallel with the power source. This reduced the voltage spikes essentially, but to make them even smaller a 10nF capacitor was connected in parallel with the MOSFET. As a MOSFET is an effectively resistive conductor, the discharge of this capacitor should not be a problem. This will have to be proved by testing the lifetime of the switch. The 10nF

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capacitor was chosen because it reduced the current spike essentially. If this reduction had been made by increasing the size of the capacitors in parallel with the power source, their size would have had to be increased substantially. This would have been a more costly and space-consuming solution than connecting the capacitor in parallel with the MOSFET. To reduce the ringing, an extra capacitor and resistor were connected in parallel with the MOSFET according to the damping theory presented in Appendix C. The ringing frequency was measured to 7MHz. This ringing frequency in combination with the 10nF voltage snubber capacitor in parallel with the switch, means, according to Equation C3 (found in Appendix C), that the damping resistor should be 2.27Ω. A 10Ω thick film power resistor was used as a damping resistor, as it was readily available. Equation C6 motivated that a 15nF capacitor was used as high pass filter capacitor. An attempt to build a RCD-snubber was made. The result, however, was not satisfactory. The same control system hardware was chosen as for the current limiter. The same microcontroller, current sensor and gate driver were used in both systems. The software was, however, designed differently. As a comparison between hysteresis control and PWM control is of interest, it was decided to use a PWM control for the motor controller. As the time for the degree project was limited, no closed loop control system for the motor speed, torque or power were made. However, a closed loop controller for limiting the maximum current to the motor was constructed, for protection reasons. The motor controller was designed so that the motor torque and speed could be regulated by varying the duty ratio of the switch via a potentiometer connected to the microcontroller, on the condition that the current did not exceed its maximal value. If the potentiometer was adjusted to a duty ratio that would lead to a current larger than the maximum current, the duty ratio was automatically adjusted by a PI-controller. If the current exceeded its maximum value, which was set in the microcontroller code, the PI controller regulated the duty ratio so that the current remained on the maximum level until the duty ratio was decreased manually with the potentiometer or the back EMF of the motor reduced the current below the maximum level. The PI regulator was tuned by using Zieger-Nichols rules, which can be found in Appendix D. The critical value of the proportional regulating constant Kcr that first gave sustained oscillations of the output was 0.2. If the proportional constant was set to 0.2, the period of the oscillations Pcr was measured to 1.2ms. According to Table D1 in Appendix D the values for the PI constants should be set to Kp=0.45*Kcr and Ti=Pcr /1.2. The constants were consequently set to Kp=0.09 and Ti=0.001. The duty ratio of the PWM signal was updated every other PWM cycle. The code made for the PWM controller can be found in Appendix E. Figure 19 shows a circuit diagram and Figure 20 shows a photo of the traction motor controller.

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M

MOSFET

..

10 μFC15 μF C

Trac

tion

Mot

or

Rec

tifie

r Out

put

2.2 μF

C

..2.2 μF

C

..2.2 μF

C

..2.2 μF

C

..

PI - PWM Control system

Current Sensor

10 Ω R

..

..

Figure 19: Traction motor controller circuit diagram

Current Sensor

Figure 20: Photo of traction motor controller circuit

MOSFET and Diode

Snubber Capacitors

Circuit Braker PI – PWM Control System

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9 Results As the main objective of this degree project was to design and construct a driveline, the main result is the actual driveline prototype. Figure 21 shows a photo of the final driveline prototype. To demonstrate the functioning of the prototype some measurements of the system performance have been carried out. Many more measurements can be made in the future. The measurements that have been made so far are described below.

Figure 21: Photo of driveline

Traction Motor

9.1 Power Converter Performance The power converters are important parts of the driveline prototype. They affect both the efficiency and the dynamics of the system. Some measurements of the performance of the current limiter and the traction motor controller are given below. As the other two power converters were not designed as part of this degree project, no measurements of their performance have been made.

9.1.1 Current Limiter Performance As mentioned, when the DC-current to the flywheel reached a certain level it was restricted within the limits of a hysteresis band with a buck converter. Figure 22 shows the inductor current in the buck converter, which is the current to the flywheel controller, and the control signal to the IGBT, when the current was limited to 10A.

Flywheel

Batteries

Generator

Current limiter

Flywheel Controller

RectifierMotor

Controller

Load

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Figure 22: Current limiter performance

Figure 22 shows that the switching frequency is around 14.2 kHz when the current is limited between approximately 9.7A and 10.3A. The switching frequency controlled by the hysteresis controller varies, partly because of varying back EMF from the flywheel, and partly because of noise on the current sensor signal. The current sensor used for measuring the current illustrated in Figure 22 was a Hall Effect sensor, as was the one used by the control system. The current measurements made by the control system were made with a sample rate of 170 kHz. Figure 23 shows the emitter-collector voltage and the gate-collector voltage during a turn-on sequence for the IGBT.

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0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1

x 10-5

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60IBGT Turn-On

Em

titte

r-col

lect

or v

olta

ge (V

)

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x 10-5

0

5

10

15

Gat

e-co

llect

or v

olta

ge (V

)

Figure 23: IGBT turn-on sequence

Figure 23 shows that the turn-on time for the IGBT is around 1.4 μs, if it is measured as the time when there is a current through and a voltage drop across the device simultaneously. The current starts flowing at approximately the same time as the positive gate voltage is applied, and the emitter-collector voltage drops to zero, as can be seen in Figure 23, when the gate voltage exceeds a certain threshold voltage. The total losses in the IGBT were calculated by measuring the drain current and the emitter-collector voltage. The reliability of the results can however be questioned, because of the limited bandwidth of the current sensor used. The results can be found in Appendix F.

9.1.2 Traction Motor Controller Performance The same measurements that were made on the current limiter, were also made on the traction motor controller. Figure 24 shows the current to the motor and the control signal to the MOSFET when the current was limited to approximately 20A by the PI-controller at a PWM frequency of 57.6 kHz.

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0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1

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30PWM controlled current

Mot

or c

urre

nt (A

)

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er v

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)

Figure 24: DC-motor controller performance

Figure 24 shows that the current ripple is between approximately 20.9A and 20.3A when the PWM frequency is 57.6 kHz if the spikes are neglected. In addition to the PWM frequency, the current ripple depends on the voltage level produced by the flywheel and the back EMF produced by the traction motor. The higher the flywheel voltage is and the nearer to 50% the duty ratio is, the larger the current ripple is. The current measurement in Figure 24 was made with a Hall Effect sensor, but not the same one as the control system used. The rectified output voltage from the flywheel used in the test shown in Figure 24 was 60V. Figure 25 shows the emitter-collector voltage and the gate-collector voltage during a turn-on sequence for the MOSFET in the DC-motor controller.

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0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2

x 10-6

-10

0

10

20

30

40

50

60MOSFET Turn-On

Dra

in-s

ourc

e vo

ltage

(V)

Time (sec)0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2

x 10-6

-10

0

10

20

30

40

50

60MOSFET Turn-On

Dra

in-s

ourc

e vo

ltage

(V)

Time (sec)0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2

x 10-6

0

5

10

15

Gat

e-so

urce

vol

tage

(V)

Figure 25: MOSFET turn-on sequence

By measuring the turn-on time in the same way as in Figure 23, it can be seen that the turn-on time for the MOSFET is approximately 175 ns. Measurements of the MOSFET switching losses were carried out, as for the IGBT. These results are more unreliable than those from the IGBT as the bandwidth of the current sensor was limited and the switching process was faster for the MOSFET than for the IGBT. The results of the measurements can be found in Appendix F.

Snubber performance A snubber circuit was required for the motor controller switch. The three figures below illustrate the effect of the snubber. Figure 26 shows the drain to source voltage across the MOSFET before the snubber was implemented.

3.768 3.77 3.772 3.774 3.776 3.778 3.78

x 10-3

-20

0

20

40

60

80

100

120

140

160Drain-source voltage, no snubber

Dra

in-s

ourc

e vo

ltage

(V)

Time (sec) Figure 26: Turn-off sequence with no snubber

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Figure 26 shows that the voltage spikes and ringing are major problems when the MOSFET is turned off. When the duty ratio was increased above the level shown in Figure 26, the ringing reached such a high level that it affected the gate charge so that the switch started to conduct. Figure 27 shows the drain to source voltage across the MOSFET when the voltage-spike-reducing snubber had been implemented.

3.772 3.7725 3.773 3.7735 3.774 3.7745 3.775 3.7755 3.776 3.7765 3.777

x 10-3

-20

0

20

40

60

80

100

120Drain-source voltage, voltage snubber

Dra

in-s

ourc

e vo

ltage

(V)

Time (sec) Figure 27: Turn-off sequence with voltage-spike-reducing snubber

Figure 27 shows that the voltage spike was reduced by the snubber from 140V to100V. The ringing was also reduced by the capacitors. Note that the duty ratio was increased in Figure 27 in comparison to Figure 26. Figure 28 shows the drain to source voltage across the MOSFET when the final snubber had been implemented.

3.772 3.7725 3.773 3.7735 3.774 3.7745 3.775 3.7755 3.776 3.7765 3.777

x 10-3

-20

0

20

40

60

80

100Drain-source voltage, final snubber

Dra

in-s

ourc

e vo

ltage

(V)

Time (sec) Figure 28: Turn-off sequence with final snubber

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Figure 28 shows that the final snubber reduces the ringing and the voltage spike to 80V. The duty ratio is the same in Figure 27 and Figure 28.

9.2 Driveline Performance To analyse the performance of the total driveline, simultaneous measurements were made at several points in the driveline. By feeding the flywheel, the traction motor, and varying the generator load in different ways, interesting observations about the system performance and dynamics could be made. Two different driveline performance tests were carried out. Both tests measured the same parameters. In addition to the traction motor speed and the flywheel speed, the currents and voltages were measured in the three measuring points shown in the Figure 29.

Figure 29: Driveline measuring points

The current sensors used by the current limiter and the motor controller were used for the driveline performance tests. As the sensors had fixed positions in the driveline, the current measurements were made at these points. To be able to calculate the power on both sides of the flywheel, voltage measurements were performed at the same points. As the load of the generator was a resistor, it was enough to measure voltage across it to be able to calculate the current and power passing through it.

9.2.1 Steady State The first test was carried out when the driveline was running at a steady state. This was achieved by letting both the current limiter and the traction motor controller limit the current to the flywheel and the motor, and tuning the generator load so that the speed of the flywheel and the traction motor were as constant as possible. Figure 30 shows the driveline voltages, the flywheel speed and the motor speed from the steady state test. The current to the flywheel was kept at 11A and the traction motor current at 26A. The load for the generator was 630Ω. The measurements have been smoothened so that their values can more easily be read in the plot. The measuring points in the driveline were the ones shown in Figure 29.

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0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 50

10

20

30

40

50Driveline voltages and speeds at steady state

Vol

tage

(V)

Time (sec)

0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 50

500

1000

1500

2000

2500

3000

Spe

ed (r

pm)

0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 50

500

1000

1500

2000

2500

3000Limited Battery VoltageResistor VoltageFlywheel SpeedMotor SpeedMotor Voltage

Figure 30: Steady state measurements

The power to the flywheel and the power to the traction motor could be calculated from the measurements presented in Figure 30. The average power to the flywheel was 467.9 W and to the traction motor 362.1W. This means that the combined efficiency of the flywheel controller, the flywheel and the traction motor controller was 73.6%. When calculating the efficiency of the system, the speed of the flywheel was approximated as constant. The electric power extracted from the generator was very low. Almost all mechanical power that was developed by in the traction motor was dissipated by friction in the motor and the generator. The average power in the measurements presented in Figure 30 that was dissipated from the generator by the resistive load was 2.0W.

9.2.2 Drive Cycle The second test that was carried out was a simple drive cycle. The drive cycle was achieved by programming the traction motor controller to vary the current to the motor, and letting the current to the flywheel be kept constant. At the same time, the magnetisation of the load generator was varied so that the traction motor was kept within reasonable speed limits. If the speed and the back EMF of the traction motor became too high, when the current to the motor was held at a certain level, the power extracted from the flywheel to the motor could increase so much that the flywheel could stop spinning. The resistance of the generator load was kept constant at 630Ω. Figure 31 shows the current to the flywheel and to the traction motor during the drive cycle performed. Figure 32 shows the speeds of the flywheel and the traction motor measured in the same test. Figure 33 shows the driveline voltages measured in the test. The measuring points in the driveline used in the tests were the ones shown in Figure 29. In all the figures the measurements have been smoothened.

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0 1 2 3 4 5 6 7 8 9 100

10

20

30

40Driveline currents during drive cycle

Time (sec)

Cur

rent

(A)

Motor CurrentLimited Battery Current

Figure 31: Drive cycle currents

0 1 2 3 4 5 6 7 8 9 100

500

1000

1500

2000Driveline speeds during drive cycle

Time (sec)

Spe

ed (r

pm)

Motor SpeedFlywheel Speed

Figure 32: Drive cycle speeds

0 1 2 3 4 5 6 7 8 9 10

0

50

100

150

200

250Driveline voltages during drive cycle

Time (sec)

Vol

tage

(V)

Motor VoltageLimited Battery VoltageResistor Voltage

Figure 33: Drive cycle voltages

By comparing Figure 31 and Figure 32 it can be seen that there is a delay between the current fed to the traction motor and the motor speed. This is due to the moment of inertia of the traction motor

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and generator. It takes time to change the speed of the rotating mass. Figure 32 also shows that more power is extracted from the flywheel than is fed to it, as the flywheel speed decreases throughout the test. Figure 32 and Figure 33 show that the voltages that the motor and the flywheel are fed with are more correlated with the speeds of the machines than with the currents that they are fed with. This is an effect of that the winding resistances are small and it can be understood by studying Equation A2 in Appendix A. The power to the flywheel controller, to the traction motor and to the generator load can be calculated from the measurements presented in Figure 31 and Figure 33. Figure 34 shows a plot of these calculations. The motor power is the power fed from the traction motor controller to the traction motor, the limited battery power is the power fed from the current limiter to the flywheel controller and the resistor power is the power fed from the generator to the resistor.

0 1 2 3 4 5 6 7 8 9 100

200

400

600

800

Driveline powers during drive cycle

Time (sec)

Pow

er (W

)

Motor PowerLimited Battery PowerResistor Power

Figure 34: Drive cycle powers

Figure 34 shows the positive effects of the flywheel. The power to the traction motor varies between approximately 150W and 700W, while the power feeding the flywheel from the batteries is almost constant at approximately 375W. The figure also shows that the power extracted from the generator to the resistive load is small.

9.3 PCB In this degree project a PCB was designed and constructed for the flywheel controller. Figure 35 shows a drawing of the board made in the program Eagle. Figure 36 shows a photo of the board when the components were soldered on to it. The PCB was used in the tests presented previously.

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Figure 35: PCB layout

Figure 36: Photo of mounted PCB

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10 Discussion The results of the degree project were generally positive. The functionality of the driveline was satisfactory, but the results also demonstrated that some of the design details were suboptimal. Here follows a discussion about the project results, how they have been achieved and how they could have been improved. The method used for designing and constructing the driveline worked well. The simulations implied that the switching frequencies would need to be considerably lower than the practical measurements showed that they needed to be to keep the current ripple within 0.5A. Nevertheless, the simulations gave an indication of the magnitude of the frequencies. The reason that the frequencies were lower in the simulations than in reality was presumably that the measurements of the values of the components (inductors, resistances) that were used for simulating the system were not very precise. Multimetres were used for making the measurements. The functionality of the two constructed power converters was proven. As the current-measuring equipment available was suboptimal, only limited conclusions could be drawn from the results. Had faster current sensors been available, measurements could have been made that would indicate whether the efficiency of the chosen converter design was optimal. However, the driveline constructed can be used to make more precise measurements with more suitable equipment in the future. The measurements of the turn-on time of the switches could be used to compare the efficiency of the two power converters. The IGBT had a turn-on time approximately eight times as large as the turn-on time of the MOSFET. The MOSFET, on the other hand, had four times as high switching frequency. According to this, the IGBT should have double as high switching losses as the MOSFET. Thus, the switching losses can be reduced by using MOSFETs and a high switching frequency. In addition to reduced switching losses, resistive losses in the inductor can be reduced by using higher switching frequencies as this allows smaller inductors to be used. On the other hand, the conducting losses are lower in the IGBTs than in the MOSFETs. The snubber circuit that the MOSFET requires also creates losses. Higher switching frequencies also give rise to increased skin effect in conductors and increased eddie and hysteresis losses in the electric machines. Designing an optimal power converter is a complex issue. Hopefully the prototype constructed can be used for making experiments that can lead to an optimal design. Economy is one of the many other factors, besides efficiency, that determine how good a driveline design is. Economy has not been an issue when designing the prototype. But if it is taken into consideration, the power converter solution with small inductors, high switching frequencies and MOSFETs is better than the solution with IGBTs and lager inductors. MOSFETs are in general cheaper than IGBTs and high power inductors cost more the higher inductance they have. A good solution for the current limiter, from an economic and efficiency point of view, would be to use the existing switches in the flywheel controller and modify its control system so that it measures the flywheel current and limits the current itself. In addition to the switches and the inductors, the two constructed power converter systems differ in that they are controlled by different control systems. Both systems work efficiently. The main difference is that the hysteresis-controlled current limiter gives a constant current ripple and the

39

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PI/PWM-controlled traction motor controller gives a constant switching frequency. It can not be predicted which of the two systems is favourable for the applications. However as the switching frequency is only an important property in the sense that it may not reach a too high level, the best solution may be to use a hysteresis controller with a maximum switching frequency limit. The results from the tests of the driveline performance were very positive apart from the low efficiency of the traction motor and the generator. The combined efficiency of the flywheel controller, the flywheel, the rectifier and the DC-motor controller of 73.6% is high in view of the fact that the driveline has not been especially optimised. It would be interesting to measure all the parts of the driveline individually, to discover which parts need to be improved. The part of the driveline that clearly had the largest losses was the traction motor and the generator. Probably this was due to friction losses in the motor. As the motor was rated for a much higher current than what was used, it had unnecessarily large brushes that gave arise to large friction losses. The friction losses would have been less significant if the motor had been driven at the power it was rated for (10 kW). The friction losses became dominating at the low power of the driveline. The coupling between the motor and generator was also not optimal. The motor and the generator were not totally aligned, which resulted in energy being dissipated by making the set-up vibrate. Another reason for the low efficiency of the traction motor and the generator was that the motor was rated for a high current and low voltage. For a motor to be as efficient as possible, it should be rated at a low current and a high voltage. A motor powered by a higher voltage, closer to the output voltage from the flywheel, at the relevant power would be more suitable than the used motor. In the constructed driveline, the flywheel was on its low voltage side powered by a voltage higher than the traction motor voltage that was fed by the flywheel’s high voltage side. It would have been better to use the DC-machine that was used as a generator, as a traction motor. It was rated at 1.5kW at 220V, as compared to10kW at 60V. This machine could therefore have been driven at a much higher voltage, at the relevant power. Much can be said about the performance of the DC-motor and the generator. However, the combined efficiency of the motor and the generator is less interesting from the research point of view that the driveline was designed for. The aim of the research is to see the effects of the flywheel. The low efficiency of the traction motor is not especially interesting, as it does not have anything to do with the flywheel. That the flywheel had a relatively high efficiency and that it had a very good capability of buffering power is a much more interesting result than that the traction motor had low efficiency.

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11 Future Work Many measurements need to be made to evaluate the driveline constructed in the project. Apart from pure measurements, a lot of modifications can be made to the system, to find a more optimal driveline design. Here follow some suggestions of future work that can be carried out:

• The power can be measured at all points of the driveline so that the losses can be identified • The switching frequencies of the power converters can be varied to see how that affects the

driveline performance • The traction motor can be changed to one rated at a lower power and a higher voltage, e.g.

the one used as a generator • Measurements can be made for a certain drive cycle, with and without the flywheel to see

how the batteries are affected In addition to these measurements and small modifications, much work still needs to be done on the driveline before it would function well in an EV. Four major tasks are listed below:

• The system power needs to be scaled up • The speed of the flywheel needs to be taken into account in the control system, so that the

level of energy extracted from the flywheel does not become so large that it stops the flywheel from spinning during operation

• The power converters need to be made bidirectional so that energy can be transferred from the traction motor to the flywheel when the EV is braked, and from the flywheel to the batteries when the flywheel is to be stopped

• The losses in the flywheel need to be reduced, e.g. by designing magnetic bearings and a vacuum enclosure

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12 Conclusions The main achievement of this degree project was the design and construction of the driveline. The driveline proved to have good performance and can be used for future research. The results showed that the flywheel controller, the flywheel and the traction motor controller had a combined efficiency of 73.6%. They also showed that the flywheel had excellent power buffering properties. The traction motor and the generator, which acted as a mechanical load for the motor, had a low efficiency. The motor, which was over-dimensioned and had large friction losses, can easily be replaced by a more suitable one. The two DC/DC-converters that were designed proved to have good functionality. Their designs demonstrated the differences between two different types of semiconductor switches and two different control strategies. The results proved that the MOSFET had a shorter turn-on time than the IGBT. However the IGBT did not require a snubber circuit. The RC-snubber that was designed for the MOSFET reduced the voltage spikes across it and the ringing remarkably. Both hysteresis control and PI-controlled PWM proved to be good control strategies for current limitation. The driveline is well-suited for experimentally determining which of these switches and control methods are best for the two applications.

Acknowledgements Janaina Goncalves de Oliviera, Magnus Hedlund, Johan Lundin, Juan de Santiago, Johan Abrahamsson, Venugopalan Kurupath, Ulf Ring, Mårten Einarsson, Hans Bernhoff, Mimi Finnstedt, Finn Finnstedt and Stella Andermo Thank you - you have been the most important components in my driveline!

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References [1] D. Naunin: Elehrische StraBenfahrzeuge. Expert Verlag, 1994. [2] P. P. Acarnley, B. C. Mecrow, J. S. Burdess, J. N. Fawcett, J. G. Kelly, P. G. Dickinson.

Design Principles for a Flywheel Energy Store for Road Vehicles, IEEE Trans on Ind. Appl., vol. 32, no. 6, 1996.

[3] M. M. Flynn, J. J. Zierer, R. C. Thompson. Performance Testing of a Vehicular Flywhwwl Energy System. SAE 2005 World Congress & Exhibition, 2005.

[4] O. Briat, J.M. Vinassa, W. Lajnef, S. Azzopardi, E. Woirgard. Principle, design and experimental validation of a flywheel-battery hybrid source for heavy-duty electric vehicles. IET Electr. Power Appl., 2007.

[5] J. Santiago, J. G. Oliveira, J. Lundin, J. Abrahamsson, A. Larsson, H. Bernhoff. Design and parameters calculation of a novel driveline for electric vehicles. World ElectricVehicle Journal Vol. 3, 2009. [6] I. M. Gottlieb. Electric motors and control techniques s.257. McGraw-Hill/Tab Electronics

1994, Edition 2. [7] H. Mogensen. Elmaskiner. Liber 1999, Edition 2. [8] J. Santiago, AFPM Motor/Generator Flywheel for Electric Power Stabilization, Uppsala Universitet, 2009. [9] S.Dhameja. Electric Vehicle Battery Systems - chap. 1. Elsevier Newnes 2001.[10] I. Buchmann (2006), What's the best battery? [Online] Available from: http://www.batteryuniversity.com/partone-3.htm [16 Mars 2010]. [11] J. G. Oliviera, Power Control Systems for PM Synchronous Flywheel Alternators, Uppsala Universitet, 2009. [12] M. M. Swamy. The Power Electronics Handbook – chapter 4.2. CRC Press 2001. [13] J. Mahdavi, A. Agah, A. Emadi. The Power Electronics Handbook – chapter 2.2. CRC Press 2001. [14] K. Rajashekara. The Power Electronics Handbook – chapter 1.1. CRC Press 2001. [15] S. Anwar. The Power Electronics Handbook – chapter 1.2. CRC Press 2001. [16] M. H. Rashid. Power Electronics Handbook: Devices, Circuits, and Applications - chapter 20. Academic Press 2007. Edition 2. [17] R. Severns. DESIGN OF SNUBBERS FOR POWER CIRCUITS, [Online] Available from: http://www.cde.com/tech/design.pdf [16 Mars 2010] [18] H. Salehfar. The Power Electronics Handbook – chapter 7.7. CRC Press 2001. [19] J. Zambada (2005). Sinusoidal Control of PMSM Motors with DSPIC30F DSC, [Online] Available from: http://ww1.microchip.com/downloads/en/AppNotes/01017A.pdf [16 Mars 2010]. [20] K. Ogata. Modern Control Engineering – Chapter 10. Pretice Hall 2002, Edition 4. [21] T. L. Skvarenina. The Power Electronics Handbook – chapter 17.2. CRC Press 2001. [22] K. Mitzner. Complete PCB Design Using OrCAD Capture and PCB Editor. Newnes 2009. [23] R. Staus. The Power Electronics Handbook – chapter 9.1. CRC Press 2001. [24] A. Q. Huang. The Power Electronics Handbook – chapter 1.9. CRC Press 2001. [25] V. Barkhordarian. The Power Electronics Handbook – chapter 1.6. CRC Press 2001. [26] J. Hagerman (1995), Calculating Optimum Snubbers. [Online] Available from: http://www.hagtech.com/pdf/snubber.pdf [16 Mars 2010]. [27] C. Nordling, J Österman. Physics Handbook for Science and Engineering. Studentlitteratur, 2006.

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Appendix

A. DC-motor dynamics When the rotor in a DC-machine rotates the armature windings are exposed to a rotating magnetic field from the poles in the stator. This results in a varying magnetic flux through the armature windings which, according to Faraday’s law and Lenz’s law, gives an electro motive force (EMF) that is opposed to the voltage from the power source, which is called a back EMF. Equation shows that the back EMF is proportional to the speed of the motor (n) and of the magnetic flux (Φδ) in the air gap [23].

Equation A1 δΦ∝ *nEMF

The voltage across the armature (U) will, according to Ohm’s law and Kirchhoff’s law, if the derivative of the armature current is small, be given by Equation A2 where Ra is the armature resistance and Ia is the armature current [23]:

aa IREMFU *+= Equation A2 This formula is valid even if the back EMF is larger than the supply voltage (U) across the armature. This means that the current (Ia) must be negative. In this mode, the motor will be working as a generator that delivers energy from the shaft of the motor to the “power source”. This process can be taken advantage of to charge the batteries when braking an EV with regenerative brakes. As mentioned, the traction motor used in the driveline constructed in this degree project is a compound motor. Figure A1 shows an equivalent circuit of a compound motor.

Series Winding

ArmatureShunt Winding

Figure A1: Equivalent circuit of compound motor

The advantage of the compound motor in this application is its relation between torque and speed. This can be understood by studying Equation A3 that gives the torque (M) [23] and Equation A4 that gives the speed (n) [7] of a DC-motor.

δφ*aIM ∝ Equation A3

δφaa IRU

n*−

∝ Equation A4

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As the magnetic flux in the air gap is a function of the current in the field windings, it will be large when the motor is to be started, as the armature current is large when the back EMF is small, and the armature current passes the series winding. This means, according to Equation A3, that the starting torque in a compound motor is large. When the speed of the motor is high, however, and the back EMF almost reduces the armature current to zero, the magnetic flux will not be reduced to zero, as the current will pass through the shunt winding. This means, according to Equation A4 that the speed of the compound motor will not increase to infinity when the armature current diminishes to zero. The resistance in the series winding of a compound motor is less than in the shunt winding, as the maximum current passing through it needs to be a lot larger. The inductance, on the other hand, is a lot larger in the shunt winding, so that a small current can provide a high enough magnetic flux.

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B. Power MOSFETs and IGBTs

Power MOSFETs Power MOSFETs (metal-oxide-semiconductor field-effect transistors) are the most commonly used power switches in applications below 600 V [24]. The circuit symbol for a MOSFET is shown in Figure B1. The anode of the MOSFET is called drain and the cathode source.

Drain

Source

Gate

Figure B1: MOSFET circuit symbol

The greatest advantage of the power MOSFET is its dynamic performance. MOSFETs allow high switching frequencies and high efficiency in combination. The main disadvantage of the power MOSFET is its static performance. The losses in the transistor when it is in its ON state can be critical, if it is conducting large currents. The MOSFET is effectively a resistive load, which means that the losses will be proportional to the collector current squared [24]. This is not the case for all semiconductor switches, i.e. the voltage drop across the current controlled power bipolar junction transistors (BJTs) does not depend on the current passed from its anode to cathode, but on the current flow through its gate. Current can pass from the cathode to the anode in a power MOSFET, even in its OFF state. This can be seen upon like an integrated body diode

IGBTs IGBTs (Insulated Gate Bipolar Transistors) are the most commonly used power switches in applications between 600V and 3000V [24]. The development of the IGBT was a way to overcome the problem with the poor current-handling capability of the MOSFET and the high gate currents of the BJT. Some BJTs need as high gate currents as a fifth of the collector current to stay in the ON state [25]. The construction of the IGBT is based on a combination between a MOS isolated gate transistor and a BJT. Figure B2 shows the circuit symbol of an IGBT. The anode of the IGBT is called emitter and the cathode collector.

Collector

Emitter

Gate

Figure B2: IGBT circuit symbol

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The IGBT does not have as good dynamic performance as the MOSFET. To achieve the same switching time for an IGBT as for a MOSFET of the same ratings, a more advanced gate driving circuit is required. The advantage of the IGBT, on the other hand, is its very low conducting losses. The conducting losses are proportional to the current, and not to the current squared, as in the case of the MOSFET. IGBTs can unlike MOSFETs be manufactured without an integrated body diode [16].

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C. Snubber Design Here follows the theory about the snubber circuits that have been designed in this degree project. The circuit that was built had the purpose to reduce the voltage spikes and the voltage ringing across a switch. Voltage spikes over semiconductors occur if they shift from conducting mode to blocking mode in a short time. This is a phenomena created by inductance. If an inductor current is decreased, the stored energy in the magnetic field of the inductor will work like a voltage source that aims to keep the current constant. If a switch is powering an inductive load, there will often be a freewheeling diode anti-parallel with the load, as in the buck converter shown in Figure 9. This will prevent the load inductance from producing voltage spikes across the switch. There is, however, always parasitic inductance in all conductors. The parasitic inductance is often small, but can cause severe voltage spikes across switches and diodes during their turning off processes, especially if the switching times are short and the conducting currents large. A way to reduce voltage spikes is to use capacitors. By charging electrical fields, capacitors can absorb the energy stored in the magnetic fields of inductors, and thereby reduce voltage spikes. Another way of explaining this mechanism is that if the current flowing through a switch or a diode can be led into a capacitor when the device blocks the current, the derivative of the current will be smaller and the voltage spike created by the parasitic inductance will decrease. Parasitic inductance in power circuits does not only create problems with voltage spikes, but also a problem called ringing. When either snubber capacitors or parasitic capacitors in devices are charged up after a turn-off process, they will start discharging. This will keep on until the capacitors have become negatively charged and start to get charged again. A resonance phenomenon will consequently occur every time the device blocks current. This current and voltage resonance is called ringing. Ringing causes problems related to electromagnetic interference. Ringing can be solved by adding a resistor to the snubber circuit. The resistor burns off the energy stored in the parasitic inductor and thereby solves the problem. If a snubber consists of capacitors an resistors it is called a RC-snubber. Snubber capacitors used for reducing voltage spikes, are often connected in parallel with a switch or a diode, as the capacitors Cs and Cd shown in Figure C1. In a buck converter, capacitors can also be connected in a parallel with the battery as the capacitor Cb in Figure C1.

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Load

Switch

bDiode

dC

Inductor

C

..C

DC source

s

Figure C1: Snubber capacitors

Placing a capacitor in parallel with a switch can cause problems. The capacitor will be charged when the switch is open, and will discharge when the switch closes. The discharging current can become so large that the device gets damaged, depending on the turn-on time and the type of switch. When capacitors are placed in parallel with the batteries in a buck circuit this problem does not arise. When choosing a capacitor for voltage spike reduction, it is important to choose the right type of capacitor. The capacitor should have as low effective series resistance (ESR) as possible. If the ESR value is large, the capacitor will have a large time constant, which will result in it not being able absorb current fast enough, and the voltage spike will remain. The value of the capacitance is also important for the time constant. The capacitance should be set so that the voltage spikes reach the required levels. Capacitors placed in parallel with switches increase the switching losses in the system in addition to being a risk of damaging the switches. To reduce ringing resistors can be connected in parallel with a switch. In reference [26] a derivation of the voltage across a RC-snubbed device is made. It has the form of a damped oscillation where the natural frequency ( nω ) of the ringing, in radians, and the damping coefficient (ζ ) is given by Equation C1 and Equation C2, where L is the parasitic inductance, C is the capacitance in parallel with the switch and is the snubber resistor. sR

LCn1

=ω Equation C1

CRsnωζ

21

= Equation C2

When designing a snubber, the snubber resistor that gives an optimal damping to the system should be found. A too small damping coefficient allows oscillations to continue and too large damping leads to unnecessary losses. A suitable value for the damping coefficient is 0.5 [26]. This results in a snubber resistor value given by Equation C3.

CL

CCR

nns ===

ωζω1

21 Equation C3

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Page 57: Design and Construction of an EV Driveline Prototype With an Integrated Flywheel

Reducing ringing across semiconductor devices by connecting resistors in parallel with them is not a good idea, as current will flow through the resistor as long as the device is in its OFF mode. This will lead to substantial system losses. A solution to this problem is to connect the resistor to a high-pass filter, that easily passes the ringing frequency, but blocks all lower frequencies and DC. The snubber circuit will then have an equivalent circuit, as shown in Figure C2.

Figure C2: RC-snubber

The cut-off frequency of a high-pass filter is given by Equation C4, where is the filter capacitor [27].

of sC

sSo CR

fπ2

1= Equation C4

If the cut-off frequency of the high-pass filter is set to the ringing frequency, the damping coefficient of the snubber will be affected. Instead a somewhat lower cut-off frequency should be chosen. A suitable value for the cut-off frequency is 2π times lower than the ringing frequency fn [26]. This gives Equation C5 and Equation C6.

⇔====LC

fCR

f nn

sSo 22 4

1422

1ππ

ωππ

Equation C5

Snss R

LCfR

C π21== Equation C6

A way of combining a voltage spike reduction snubber with a ringing reduction snubber, without having a capacitor connected directly across a switch as in Figure C1, which may risk damaging the switch, is to make a RCD-snubber. RCD stands for resistor, capacitor and diode. The diode in a RCD-snubber is connected in parallel with a resistor, which in its turn is connected in series with a capacitor, as shown in Figure C3.

sC

Figure C3: RCD-snubber

With a RCD-snubber the current blocked by the switch will be able to flow into the capacitor so that the voltage spike is reduced. The time constant of the snubber will be low, as the diode will bypass the resistor. When, on the other hand, the capacitor discharges, the diode will be biased in reverse and the current will have to pass the resistor. This can both solve the problem of ringing and the problem of large discharging currents through the switch.

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Page 58: Design and Construction of an EV Driveline Prototype With an Integrated Flywheel

D. Zieger-Nichols Rules Zieger-Nichols rules for tuning PID controllers can be used for finding values for the PID constants that give a good input/output response [20]. The constants can be chosen by studying the output of the system when it is only proportionally regulated and Kp is set to the critical value Kcr that first gives sustained oscillations of the output. The PID constants can be set according to the Table D1, where Kcr is the critical proportional constant and Pcr is the period of the oscillations. Table D1: Zieger-Nichols rules Type of Controller Kp Ti TdF 0.5 Kcr ∞ 0 PI 0.45 Kcr Pcr /1.2 0 PID 0.6 Kcr 0.5Pcr 0.125 Pcr

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Page 59: Design and Construction of an EV Driveline Prototype With an Integrated Flywheel

E. Microcontroller codes

Current Limiter Code /*********************************************/ /* Connect RB0 ---> Current sensor */ /* Connect RB1 ---> Driver input */ /* Connect RB2 ---> Potentiometer */ /*********************************************/ #include "p30f2010a2.h" /* Setup Configuration For ET-dsPIC30F2010 */ _FOSC(XT_PLL16); // Enable Clock Switching,Enable Fail-Salf Clock // Closk Source = Primary XT + (PLL x 16) _FWDT(WDT_OFF); // Disable Watchdog _FBORPOR(PBOR_ON & BORV_45 & PWRT_64 & MCLR_EN); // Enable Brown-Out = 4.5V,Power ON = 64mS,Enable MCLR _FGS(CODE_PROT_OFF); // Code Protect OFF /* End Configuration For ET-dsPIC30F2010 */ /* Definition of global variables*/ int channel0Result = 0; // The read value from the current sensor int channel2Result = 0; // The read value from the potentiometer /* ADC interrupt */ void __attribute__((__interrupt__)) _ADCInterrupt(void); /* Start main program here */ int main(void) /** Digital output that can be used for seeing how the process is runnuing **/ TRISBbits.TRISB3 = 0; LATBbits.LATB3 = 1; TRISBbits.TRISB4 = 0; LATBbits.LATB4 = 1; /* Intialize the ADC */ ADPCFG = 0xFFFA; /** AN0 and AN2 are analog inputs**/ TRISEbits.TRISE0 = 0; /** Config RE0 = Output **/ ADCON2bits.VCFG = 0; /** VCFG=000 means: Vdd Vss are voltage references**/ ADCON2bits.CSCNA = 1; /** 1 = Scan inputs **/ ADCON3bits.ADCS = 3; /** Tad = internal 2*Tcy.**/ ADCON2bits.CHPS=0; /** CHPS = 00 Converts CH0**/ ADCON1bits.SSRC=7; /** SSRC bit = 111 implies internal**/ /** counter ends sampling and starts converting **/ ADCSSL = 0x5; /** Let AN0 and AN2 be scanned**/ ADCHS = 0; /** 0000 = Channel 0 positive input is AN0**/ ADCON3bits.SAMC =0; /** AUTO-Sample time = O**/ ADCON1bits.ASAM = 1; /** auto start sampling**/ ADCON2bits.SMPI = 1; /** Interrupts at the completion of conversion for**/ /** each 2nd sample/convert sequence.**/

ADCON2bits.BUFM = 0; /** Buffer configured as one 16-word buffer**/ /** ADCBUF(15...0)**/

ADCON2bits.ALTS = 0; /** Always use MUX A input multiplexer settings**/ /* Set up the Interrupts */ IFS0bits.ADIF = 0; /** Clear AD Interrupt Flag **/ IPC2bits.ADIP = 4; /** Set ADC Interrupt Priority **/ IEC0bits.ADIE = 1; /** Enable the ADC Interrupt **/ ADCON1bits.ADON = 1; /** turn ADC ON **/ /** Digital output B1 gives the control signal to the driver **/ TRISBbits.TRISB1 = 0; LATBbits.LATB1 = 1; while (1) /** Loop continue**/ // Compare the value from the current sensor and the potentiometer if (channel0Result>channel2Result-10)

// -10 sets the level of the upper hysteresis limit

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Page 60: Design and Construction of an EV Driveline Prototype With an Integrated Flywheel

LATBbits.LATB1 = 0; // turn the switch of if the current is to large if (channel0Result<channel2Result - 35)

// -35 sets the level of the lower hysteresis limit. // Change this to adjust the hystersis band!

LATBbits.LATB1 = 1; // Turn the switch on if the current is low enough

//This can be uncommented to se how fast the main loop is operating //LATBbits.LATB3 = !LATBbits.LATB3; /** ADC interrupt **/ void __attribute__((interrupt, no_auto_psv)) _ADCInterrupt(void) channel0Result = ADCBUF0; /* Get the conversion result current limiter*/ channel2Result = ADCBUF1; /* Get the conversion result potentiometer*/ // This can be uncommented to show how fast the ADC interrupt is operating // LATBbits.LATB4 = !LATBbits.LATB4; IFS0bits.ADIF = 0; /* Clear ADC Interrupt Flag */

Traction Motor Controller Code /****************************************************/ /* Connect RB0 ---> Potentiometer */ /* Connect RB1 ---> Current sensor */ /* Connect RE8(FLTA) ---> Error Motor Stop */ /* Connect RE1(PWM1H) ---> Driver input */ /****************************************************/ #include "p30f2010a2.h" // For dsPIC30F2010 MPU Register #include "pwm.h" // Used MCPWM Library Function #include "adc10.h" // Used 10 Bit ADC Library Function /* Setup Configuration For ET-dsPIC30F2010 */ _FOSC(XT_PLL16); // Enable Clock Switching, Enable Fail-Salf Clock // Clock Source = Primary XT + (PLL x 16) _FWDT(WDT_OFF); // Disable Watchdog _FBORPOR(PBOR_ON & BORV_45 & PWRT_64 & MCLR_EN); // Enable Brown-Out = 4.5V, Power ON=64mS, Enable MCLR _FGS(CODE_PROT_OFF); // Code Protect OFF /* End Configuration For ET-dsPIC30F2010 */ /* pototype section */ void init_mcpwm(void); // Initial MCPWM Function void init_adc(void); // Initial RB0 and RB1 = 10 Bit ADC void delay(unsigned long int); // Delay Time Function void __attribute__((__interrupt__)) _ADCInterrupt(void); // ADC interrupt with PI maximum current limiter /* Definition of global variables */ unsigned int gas = 0; // The value read from the potentiometer unsigned int current = 0; // The value read from the current sensor unsigned int max = 350; // The maximum current, 350 gives ~33A unsigned int Duty = 0; // The duty ratio calculated by the PI regulator int error; // The difference between the actual current and the // maximum current int integral = 0; // The integral of the error float Kp = 0.09; // The proportional constant in the PI regulator float Ki = 0.001; // The integral constant in the PI regulator /* Start Main Program Here */ int main(void)

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Page 61: Design and Construction of an EV Driveline Prototype With an Integrated Flywheel

init_mcpwm(); // Initial MCPWM = 57.6 KHz init_adc(); // Initial the ADC //Digital outputs that can be used for seeing how the process is running TRISBbits.TRISB2 = 0; LATBbits.LATB2 = 0; TRISBbits.TRISB3 = 0; LATBbits.LATB3 = 0; //Limits for the current if the drive cycle programmed in the continuous loop shall be run //(uncomment the three following lines) // int tempmax=300; //Max current in the drive cycle // int tempmin=150; //Min current in the drive cycle // max = tempmax; //Set the max current to tempmax while(1) // Loop Continue //A drive cycle that increases/decrease the current linearly between tempmax and tempmin, if //the following two loops are uncommented //while(max > tempmin) // // max = max - 1; // delay(20000); //This sets the time for the decreasing of the current // //while (max < tempmax) // // max = max + 1; // delay(20000); //This sets the time for the increasing of the current // //LATBbits.LATB2 = !LATBbits.LATB2; //This can be uncommented to show how fast the main //loop is operating /********************************/ /* Initial PWM for dsPIC30F2010 */ /* -> PWM Frequency = 57.6 KHz */ /********************************/ void init_mcpwm() CloseMCPWM(); // Disable MCPWM Before New Config // Config MCPWM Interrupt Control ConfigIntMCPWM(PWM_INT_DIS & // Disable PWM Interrupt PWM_INT_PR6 & // PWM Interrupt Priority = 6 PWM_FLTA_DIS_INT & // Disable Fault-A Interrupt PWM_FLTA_INT_PR7); // Fault-A Interrupt Priority = 7 SetMCPWMFaultA(PWM_OVA1H_INACTIVE & // Enable Fault-A Control PWM1H = OFF PWM_FLTA_MODE_LATCH & // Fault-A Mode = Latch PWM_FLTA1_EN); // Enable Fault-A CH1 //*************************************************** // XTAL = 7.3728MHz // Fosc = 7.3728 MHz x 16 = 117.9648 MHz // Fcy = Fosc / 4 // = 117.9648 / 4 = 29.4912 MHz // Tcy = 1 / 29.4912 MHz // = 33.90842 nS //*************************************************** // PWM Clock = Prescale = 1 // 1 Cycle PWM = 1 / 29.4912 MHz // Desire MAX Duty Cycle = 1024 // PWM Period = 1024/2 // = 512

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Page 62: Design and Construction of an EV Driveline Prototype With an Integrated Flywheel

// Frequency = 29.4912 MHz/ 512 // = 57.6 KHz // Duty Cycle = 1...1024 //****************************************************

OpenMCPWM(512, // PTPER = Period = 57.6 KHz 400, // SEVTCMP = Special Time, the AD conversion is trigged

//when the PTMR value is 400 // PTCON PWM_EN & // Enable PWM Function PWM_IDLE_STOP & // Disable PWM in IDLE Mode PWM_OP_SCALE1 & // PWM Post Scale = 1 PWM_IPCLK_SCALE1 & // PWM Input Clock Prescale = 1 PWM_MOD_FREE , // PWM = Free Running // PWMCON1 PWM_MOD1_IND & // PWM1 = Free Mode PWM_PEN1H, // PWM1 High // PWMCON2 PWM_SEVOPS1 & // Special Even Post Scaler = 1:1 PWM_OSYNC_PWM & // Override Sync. With PWM Clock PWM_UEN); // Enable PWM Update /********************************/ /* Initial ADC for dsPIC30F2010 */ /* Used RB0 and RB1 = 10 Bit ADC */ /********************************/ void init_adc()

ADCON1bits.ADON = 0; /** Turn-OFF ADC Before Change Config**/ /* Intialize the ADC */ ADPCFG = 0xFFFC; /** AN0 and AN1 are analog inputs**/ ADCON2bits.VCFG = 0; /** VCFG=000 means: Vdd Vss are voltage references**/ ADCON2bits.CSCNA = 1; /** 1 = Scan inputs **/ ADCON3bits.ADCS = 4; /** Tad = internal Tcy*5/2.**/ ADCON2bits.CHPS=0; /** CHPS = 00 Converts CH0**/

ADCON1bits.SSRC=3; /** SSRC bit = 011 implies Motor Control PWM **/ /**interval ends sampling and starts conversion**/

ADCSSL = 0x3; /** let AN0 and AN1 be scanned**/ ADCHS = 0; /** 0000 = Channel 0 positive input is AN0**/ ADCON1bits.ASAM = 1; /** auto start sampling**/ ADCON2bits.SMPI = 1; /** Interrupts at the completion of conversion for**/ /** each 2nd sample/convert sequence.**/

ADCON2bits.BUFM = 0; /** Buffer configured as one 16-word buffer **/ /**ADCBUF(15...0)**/

ADCON2bits.ALTS = 0; /** Always use MUX A input multiplexer settings**/ /* Set up the Interrupts */ IFS0bits.ADIF = 0; /** Clear AD Interrupt Flag **/ IPC2bits.ADIP = 4; /** Set ADC Interrupt Priority **/ IEC0bits.ADIE = 1; /** Enable the ADC Interrupt **/ ADCON1bits.ADON = 1; /** turn ADC ON **/ /***********************/ /* Delay Time Function */ /***********************/ void delay(unsigned long int count1) while(count1 > 0) count1--; // Loop Decrease Counter /************************************************/ /* ADC interupt with PI maximum current limiter */ /************************************************/ void __attribute__((interrupt, no_auto_psv)) _ADCInterrupt(void)

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Page 63: Design and Construction of an EV Driveline Prototype With an Integrated Flywheel

//LATBbits.LATB3 = !LATBbits.LATB3; //This can be uncommented to show how fast the main //loop is operating

gas = ADCBUF0; /* Get the conversion result from the potentiometer*/ current = ADCBUF1; /* Get the conversion result from the current sensor*/ // PI-Regulator error = max - current; integral = integral + error; Duty = Duty + Kp * error + Ki * integral; //Avoid negative duty ratios if (Duty < 0) Duty = 0; //Compare the value of the potentiometer and the calculated maximum duty ratio if (Duty < gas) SetDCMCPWM(1,Duty,0); //(PDC1 Register,Duty Cycle,PWMCON2.UDIS=0(Enable

//Update PWM1) else SetDCMCPWM(1,gas,0); //(PDC1 Register,Duty Cycle,PWMCON2.UDIS=0(Enable

//Update PWM1) Duty = gas; //Make the memory of the regulator accurate when the

//current limiter does not set the duty ratio integral = 0; //Set the integrated error to 0 when the current

//limiter does not set the duty ratio IFS0bits.ADIF = 0; /* Clear ADC Interrupt Flag */

56

Page 64: Design and Construction of an EV Driveline Prototype With an Integrated Flywheel

F. Switch Losses An attempt was made to measure the switch losses for the IGBT and the MOSFET in the power converters designed in this degree project. The current through and the voltage across the switches were measured. By multiplying these, the losses in the switches could be found. The device used to measure the current was a Hall Effect sensor that was rated for a maximum AC-current of 50 kHz. The current measured was a DC-current, but it was switched at high speed. It could not be proved that the speed of the current sensors were too slow, but the results of the calculations of the MOSFET losses indicate that it was so. The temperature of the heat sink of the MOSFET was not sufficiently high for the measurements to be correct. As the switching speed of the IGBT was much lower than that of the MOSFET, the losses measured for the IGBT are more reliable than the ones for the MOSFET. Figure F1 and Figure F2 show the measurements of the switch voltages and switch currents that were made to allow calculations of the switch losses.

1.8 1.9 2 2.1 2.2

x 10-3

0

50

100

IBGT voltage and current

Em

titte

r-col

lect

or v

olta

ge (V

)

Time (sec)1.8 1.9 2 2.1 2.2

x 10-3

0

5

10

15

Em

itter

cur

rent

(A)

Figure F1: IGBT losses

According to the measurements, the IGBT losses were 15.8W.

57

Page 65: Design and Construction of an EV Driveline Prototype With an Integrated Flywheel

3.76 3.765 3.77 3.775 3.78 3.785 3.79

x 10-3

0

50

100

MOSFET voltage and current

Dra

in-s

ourc

e vo

ltage

(V)

Time (sec)3.76 3.765 3.77 3.775 3.78 3.785 3.79

x 10-3

0

5

10

15

Dra

in c

urre

nt (A

)

Figure F2: MOSFET losses

According to the measurements, the MOSFET losses were 57.2W.

58