978-1-4673-8617-3/16/$31.00 (c) 2016 IEEE

A Performance Comparison of SiC and Si devices in a Bi-Directional Converter for Distributed Energy

Storage Systems

Thilini Daranagama, Florin Udrea The Department of Engineering The University of Cambridge Cambridge, United Kingdom.

e-mail: td325@cam.ac.uk, fu10000@cam.ac.uk

Tom Logan Paramita Electronics

Cambridge, United Kingdom. e-mail: tgl@paramita-electronics.com

Richard McMahon WMG, The University of Warwick

Coventry, United Kingdom. e-mail:

R.McMahon.1@warwick.ac.uk

Abstract—Due to its superior electrical characteristics resultant from material properties such as high critical electric field, larger bandgap and higher thermal conductivity in comparison to Silicon (Si), Silicon Carbide (SiC) devices are becoming increasingly popular in the power electronics industry. Many SiC device types have been commercialised during the past decade. In this paper, a performance comparison between SiC MOSFET, SiC BJT and SiC SJT based bi-directional converters, targeting energy storage applications, has been made along with a Si IGBT version. Conduction losses and switching losses have been calculated and compared with the aid of measured forward I-V curves and double-pulse testing in a clamped-inductive test configuration, respectively. In order to validate the continuous operation, the device performances when operating the bi-directional converter in DC/DC conversion mode have been compared. The efficiency of the converters have been assessed at 20 and 40 kHz switching frequencies and up to 6.6 kW output powers and 27 A input current levels per half-bridge. From the results discussed, it is clear that the SiC devices surpass Si IGBTs in switching performance. During 250 V to 500 V boost conversion at 20 kHz, a loss reduction of 50.8% has been observed at a 4.9 kW power output in SiC MOSFET converter, in comparison to the Si IGBTs. At 40 kHz, peak efficiencies of 98.8% and 98.6% have been demonstrated by SiC MOSFET and SiC BJT converters, respectively. The aim of this comparison is to identify and set a guideline as to the most appropriate circumstances for using each device technology.

Keywords—Silicon Carbide; SiC BJT;SiC MOSFET; SiC SJT; Si IGBT; Bi-directional Converter; Energy Storage; DC-DC Conversion

I. INTRODUCTION

Solar PV (photovoltaic) capacity has grown dramatically in recent years – the UK, for example, reached 5.4 GW installed capacity in the 3rd quarter of 2015, up from 2.9 GW in 2013 [1]. The number of heat pumps and electric vehicle charging stations are likely to show similar growth rates in coming years, and must if governments are to meet their targets. Houses with these technologies installed show much wider power demand fluctuations than typical consumers of the past,

with the potential to be substantial net exporters on sunny days and to be substantial loads as vehicles are charged overnight. The increase in households following this pattern will place additional strain on the grid, reducing the overall capacity factor of the generating plant at the national level, and making it more difficult to keep supplies within voltage limits at the local level.

One potential method of reducing these power fluctuations is to install energy storage plants across the network and a variety of technologies have been proposed and trialled. In a domestic or commercial setting battery energy storage would appear to be the most practical technology in terms of installability and ease of operation. Battery systems also have excellent dynamic performance and can be scaled to domestic capacities while still showing good efficiencies and reasonable cost.

The key part of the battery energy storage devices is a bi-directional AC/DC converter to transfer power between the battery and the mains. One class of battery converter uses high voltage batteries, where the battery voltage is greater than the rectified mains voltage. The simplest three phase case is shown in Fig. 1. In this configuration the battery is connected across the DC link of the converter, with a half bridge switched to control the current through an inductor into each mains phase to obtain the necessary sinusoidal current draw/injection. For 400 V mains, the battery voltage will typically vary within the

Fig. 1. Simplest form of a 3-phase bi-directional AC/DC converter for grid connected batteries.

600-900 V range and typically 1.2 kV switching devices would be used.

The relatively slow switching speed of IGBTs means that large inductance values and a multi-stage filter are required to keep the ripple current low enough to meet EMC requirements on the power lines. Moving to a higher switching frequency allows smaller ripple currents to be obtained using a lower cost and/or higher efficiency filter. This cannot be achieved with silicon IGBTs because the resulting switching losses require a bulky and unacceptably expensive heat sink while causing a poor overall efficiency. This problem can be averted by replacing Si IGBTs with SiC devices.

SiC power devices have attracted increasing interest in the power electronics industry because of their ability to switch faster while maintaining lower conduction losses. SiC technology has evolved considerably during the past decade and many device types have been introduced commercially. Previous work found in literature has compared individual static and switching performances of the SiC devices with Si devices [2],[3],[4],[5]. However, a performance comparison that investigates and validates the continuous performance of the devices in a real application is important for the end users. Currently, the SiC MOSFET is fairly established and the benefits are well known within the power electronics industry as a number of publications have assessed and compared their performance in applications. In [6], authors have made a detailed performance analysis of a SiC MOSFET in a DC-DC boost converter and have compared the results with Si IGBT and Si BiMOSFET converters up to 10 kW power and 10 kHz switching frequency. In more recent years, SiC bipolar devices have been commercialised. SiC BJTs are normally-OFF devices free of gate-oxide problems and have low on-state voltages, faster switching performance and are easy to parallel owing to the positive temperature coefficient of the on-state resistance [7]. Therefore, it is important to compare their performance against the SiC MOSFET as well.

Therefore, in this paper, a comparison between different SiC device types have been considered in a bi-directional converter designed for a domestics battery energy storage system. We have assessed the performances of 1.2 kV rated SiC MOSFETs, SiC BJTs and normally-OFF SiC Junction Transistors (SJTs) and compared against a Si IGBT. In the first section of the paper, individual device performances have been compared by calculating conduction losses with the aid of forward I-V curves as well as switching losses obtained by double-pulse testing in a clamped-inductive test configuration. Finally, in order to validate continuous operation, the next section compares the device performance when operating the bi-directional converter in DC/DC conversion mode.

The electrical ratings of the 1.2 kV devices chosen for this comparison along with the specification of the respective driver circuit have been listed in Table I. For a fair comparison between each device type, the devices with similar electrical ratings have been chosen. Even though the SiC JFET was the first SiC transistor to be commercialised, we have not considered JFETs in this work as a commercial sample was not available during the time of this work.

II. CHARACTERISATION AND COMPARISON OF ON-STATE AND

HARD SWITCHING PERFORMANCE

A. Forward Characterisation

The forward characterisation of all the devices has been performed using two Keithley 2651A high power system source meters for obtaining high output currents, and a Keithley 2602B system source meter at the input for biasing the gate/base of the devices. The Kelvin sense measurement approach was employed to eliminate the influence of contact and wire resistances on measurements. The forward characteristics of the devices for the corresponding biasing requirements used in the driver circuits discussed in the following sections, are compared in Fig. 2. All measurements were obtained at 25oC. It is clear that the SiC bipolar devices exhibit the lowest forward voltage drop at all current levels, particularly the SiC BJT. The reduced on-state voltage (VCESAT) results from the cancellation of the junction voltages of the base-emitter and base-collector junctions in the saturation region of operation [8],[9]. At current levels below 43 A, the SiC MOSFET demonstrates lower forward voltages. Since the critical electric field of SiC is about 10 times higher than Si, for the same breakdown voltage, the drift region of SiC devices are much shorter than Si devices. Therefore, despite being a unipolar device, the SiC MOSFET exhibits a lower on-state drop than the Si IGBT. Due to the conductivity modulation of the drift region, the Si IGBT has a lower on-state voltage drop at higher current levels.

Based on the forward characteristics, the conduction energy losses (Econd) for the devices have been calculated using

Fig. 2. Comparison of forward characteristics at 25oC for the Si-IGBT (VGE = 15 V), SiC-MOSFET (VDS = 20 V), SiC-BJT (IB = 1.3 A), and SiC-SJT (IG = 1.3 A).

TABLE I. LIST OF TESTED 1200V DEVICES AND RATINGS.

Device Manufacturer Max. IC

or ID (A)

Max. TJ (oC)

Driver Output

Si-IGBT IRG7PH42UD

International Rectifier

45 A @ 100 oC

150 oC -5V/15V, RG = 10 Ω

SiC-MOSFET C2M0040120D

CREE 40 A @ 100 oC

150 oC -5V/20V, RG = 3 Ω

SiC-BJT FSICBH017A120

Fairchild Semiconductor

50 A @ 100 oC

175 oC 1.3 A

SiC-SJT GA50JT12-247

GeneSiC Semiconductor

50 A @ 145 oC

175 oC 1.3 A

Econd = (Vfwd * Icond * D) / fsw (1)

where Vfwd, Icond, D and fsw are forward voltage drop of the device, average conduction current, duty ratio and switching frequency, respectively. In these calculations, ripple current is assumed to be small and insignificant. Conduction energy losses calculated for 50 % duty ratio and 20 kHz switching frequency have been plotted against Icond in Fig. 3. Due to the low forward voltage, the SiC BJT exhibits the lowest conduction losses.

B. Experimental Set-up For Hard-Switching Characterisation

To characterise the switching performance of the devices under hard-switched conditions, the double-pulse (DP) test method has been used. The equivalent circuit is illustrated in Fig. 4(a). In this schematic, the DUT (device under test) represents one of the four devices listed in Table I. A 277 μH inductor (LDP) designed for a 60 A saturation current has been used in the experiment and a 1200V CREE C4D40120D SiC Schottky diode (DDP) have been used as the free-wheeling diode. A Tektronix DPO 5104 digital oscilloscope, which has a bandwidth of 1 GHz, has been used for measuring the switching waveforms. The details of the measurement probes

have been listed in Table II. All current and voltage probes have been deskewed with reference to the TPP1000 voltage probe to eliminate delays between signals so that the power losses can be accurately estimated using the measured current and voltage waveforms.

The pulse sequence for the DP test is as seen in Fig. 4(b). At the end of the first pulse, the inductor current (ILDP) ramps up to the desired value, set by the first pulse width (TON1). Then the device is switched off and the turn-off characteristics are recorded. During the off period (TOFF) this current flows in the LDP and DDP. It is assumed the current stays constant throughout the free-wheeling period as the inductor is designed with a time constant much greater than TOFF. After this, the device is turned-on again to measure the hard-switched turn-on characteristics. In the DP method, it can be assumed the device junction temperature remains constant due to the short pulse widths. Since the battery voltage of the application is up to ± 450 V, all devices have been tested at a 900 V DC bus voltage. Driver circuits for each device type has been designed separately with the outputs listed in Table I. The Si IGBT and SiC MOSFET are voltage driven devices. The IGBT has been switched with a 10 Ω gate resistance (RG) and a gate voltage of +15 V during turn on and -5 V during turn-off. An RG of 3 Ω has been used for the SiC MOSFET with a +20 V ON-voltage and a -5 V OFF-voltage. The SiC BJT and SiC SJT are current driven bipolar devices and the dual source base drive circuit discussed in [10] has been adapted to provide a 1.3 A steady-state current output. The block diagram for the drive circuit is illustrated in Fig. 5. The driver IC 1 provides the continuous steady-state base current of 1.3 A through RB1 to ensure the transistor is operating in the saturation region during on-state. The driver IC 2 provides the transient current required for faster switching of the device, i.e. for a faster discharging and charging of the miller capacitance CBC during turn-on and turn-off, respectively. The minimum steady-state base current required to keep the SiC BJT in saturation during on-state (IB(min)), for the maximum collector current expected in the application (IC(max)), and at the maximum junction temperature (TJ(max)), is calculated as

TABLE II. DETAILS OF THE MEASUREMENT PROBES

Measurement Probe Bandwidth Deskew

Base/Gate Voltage

Tektronix TPP1000 1 GHz 0 ns

Collector/Drain Voltage

Tektronix TPP0850 800 MHz 350 ps

Collector/Drain Current

PEM CWT Ultra Mini Rogowski Coil

30 MHz 30.4 ns

(b) (a)

Fig. 4. (a) Schematic of the DP test set-up used for hard-switched characterisation and (b) Double-pulse switching waveform.

(2)

Fig. 3. Comparison of conduction energy losses at different current levels, at 20 kHz switching frequency and 50 % duty ratio.

where βT(max) is the current gain at the maximum operating temperature of the system. For a IC(max) of 50 A and TJ(max)) of 125oC, the current gain (β) of the SiC SJT is typically 57 [11]. Hence, the driver circuits for both the SiC bipolar devices have been designed to provide a steady-state base current of 1.3 A. A safety margin of 50% have been used in the steady-state base current calculations, as recommended in application notes provided by Fairchild Semiconductor [12].

C. Hard-Switching Characterisation and Loss Analysis

The turn-on and turn-off switching waveforms of the Si IGBT, SiC MOSFET, SiC BJT and SiC SJT at 25 oC can be seen in Fig. 6, 7, 8 and 9, respectively. PCB layouts have been carefully designed to minimise the effects of stray inductances on switching waveforms.

Based on the switching characterisations, switching energy losses have been calculated by integrating the product of device voltage and device conduction current over the switching time period. The calculated turn-on energy losses (EON), turn-off energy losses (EOFF) and total switching losses (ET(SW)) have been plotted as a function of switching current in Fig. 10. As expected, the SiC devices have a clear advantage over the Si IGBT in switching performance, especially at turn-off. As a bipolar device, Si IGBT relies on conductivity modulation during on-state to achieve a lower forward voltage drop. However, this has an adverse effect on switching performance, as the stored charge in the collector drift layer must be removed during turn-off, leading to a long current tail in the turn off waveform and hence increased turn-off losses.

Even though both the SiC BJT and SiC SJT are bipolar devices, they exhibit low switching losses. This is more significant at higher current levels. Due to the high critical electric field strength of SiC, BJTs with thin and highly doped collector drift regions can be designed [13]. As a result of the high collector-drift doping, combined with the lower bulk carrier lifetimes of SiC, the conductivity modulation phenomenon is relatively low in SiC bipolar devices [14]. This results in fast switching performance in SiC bipolar devices. The lower switching losses of the SiC MOSFET are due to the absence of stored-charge issues arising from the unipolar nature of the device, combined with the lower capacitances due to the SiC material properties.

(b) (a)

VCE

VCE

VBE

VBE

IC IC

Fig. 8. Switching characteristics of the SiC BJT at 25 oC. DC voltage is 900 V and IC is 40 A. Scale: IC → 10 A/div, VCE → 200 V/div, VBE → 5 V/div, Time → 50 ns/div. (a) Turn-on. (b) Turn-off.

VDS

VGS VGS

ID

(b) (a)

Fig. 7. Switching characteristics of the SiC MOSFET at 25 oC. DC voltage is 900 V, ID is 50 A and RG is 3 Ω. Scale: ID → 10 A/div, VDS → 200 V/div, VGS→ 5 V/div, Time → 50 ns/div. (a) Turn-on. (b) Turn-off.

VDS

Fig. 5. Schematic of the dual source base driver used for the SiC BJT and SiC SJT.

(b) (a)

VCE

VCE

VGE

VGE

IC IC

Fig. 6. Switching characteristics of the Si IGBT at 25 oC. DC voltage is 900 V, IC is 50 A and RG is 10 Ω. Scale: IC → 10 A/div, VCE → 200 V/div, VGE → 5 V/div, Time → 100 ns/div. (a) Turn-on. (b) Turn-off.

ID

(b) (a)

VDS

VDS

VGS

VGS

ID ID

Fig. 9 Switching characteristics of the SiC SJT at 25 oC. DC voltage is 900 V and ID is 50 A. Scale: ID → 10 A/div, VDS → 200 V/div, VGS → 5 V/div, Time → 50 ns/div. (a) Turn-on. (b) Turn-off.

Total energy losses, taking in to account the turn-on and turn-off switching losses as well as the conduction losses calculated for 25 A and 50 A switching currents and 20 kHz switching frequency at 50 % duty ratio, are summarised in Fig. 11. At 25 A, there are no significant differences in the energy losses between the SiC bipolar devices and the SiC MOSFET. However at the higher current level, SiC bipolar devices have a clear advantage over the SiC MOSFET, predominantly due to the reduction in conduction losses.

III. CONVERTER OPERATION AND EFFICIENCY COMPARISON FOR

DC-DC CONVERSION

Forward and DP switching characterisation results have indicated the clear advantages of SiC power devices, particularly the SiC BJTs. However, continuous operation of the devices in a real hard-switched application is necessary to demonstrate and validate the advantages of SiC devices. In this section, the performance of the devices listed in Table I have been compared in four bi-directional converters designed for a grid connected energy storage application. A single phase leg of the three phase converter of Fig. 1 was constructed and tested as a DC/DC converter using the setup shown in Fig. 12. Although in the application, the circuit will be operated in AC/DC mode, operating in DC/DC mode allows a more straightforward comparison between the performances of the devices and is equivalent to a single point within the AC cycle. Appropriate gate drive circuits have been designed for every converter to harness the maximum performance form each device type, as discussed in Section II. The PCB layouts have been carefully designed to minimise the stray inductances in the current commutation path. The devices have been mounted beneath the PCBs on to the heatsinks of type LA 6/150 from Fischer Electronik, containing 12 V fans for forced air cooling and thermal resistance of 0.175 °C/W. In a real application, it is more convenient to use one heatsink to mount all power devices; therefore, Sil-Pad K-6 thermally conductive insulator pads have been used for electrical isolation between the device and the heatsink. The temperature of the heatsinks is adjusted by heating four 2.2 Ω resistors attached to either sides of the heatsinks and was kept at 30 °C by closed-loop control. A temperature sensor was mounted on the heatsink close to the power device. By this method, the case temperature of the device can be kept constant at 30 °C during the tests. 1200V CREE C4D40120D SiC Schottky diodes have been used as the anti-parallel diode in all switches. Ferrite cored inductors have been designed for each switching frequency to hold the inductor ripple current constant in each case. The Details of the passive components and the measurement equipment have been listed in Table III and Table IV.

IC

VCE

VGE

IC

(c)

(b)

(a)

Fig. 10. Comparison of switching losses at different switching currents and 900 V DC bus voltage. 25 oC. (a) Turn-on energy losses (EON). (b) Turn-off energy losses (EOFF) (c) Total switching losses (ET(SW)).

As shown in Fig. 12(a), in this configuration, two converters are connected back-to-back. A is always the converter being tested and is connected to a 250 V power supply and operated at a constant duty cycle of 50%. Converter A is switched to step-up the 250 V input (Vin ) to 500 V (Vout) and the input current flow (Iin) is adjusted by changing the duty cycle of converter B, which is operated in voltage step-down

mode from 500 V. An example of the switching waveforms when operating the converters in this configuration (low-side switches of the two converters and inductor ripple currents) can be seen in Fig. 13. The performance of the converters have been measured up to 6.6 kW output power and 27 A input current levels. The efficiency of the converters have been evaluated from the measured input and output powers as,

The energy losses in the drive circuits are not significant at

the power levels tested and hence, have been ignored in the efficiency calculations. The results of the 250 V to 500 V voltage boost operation of the four converters at 20 kHz switching frequency (fSW) have been plotted in Fig. 14. A clear improvement in efficiency can be seen in the SiC device curves in comparison to the Si IGBTs; especially at low power levels where the switching and conduction losses of the power devices are the dominant loss mechanisms of the converters. At the power levels tested, the SiC MOSFET converter demonstrated the highest efficiency, with a peak efficiency of 98.9% recorded at 3.2 kW output power. However, the difference in efficiencies between the three SiC converters are not significant at these power levels. This is consistent with the double-pulse test results of Fig. 10, where total switching losses are similar amongst the SiC devices at current levels below 25 A.

In order to investigate the switching performance of the SiC devices at a higher switching frequency, the SiC converters were tested at 40 kHz using a 235 μH inductor. The efficiency

Fig. 12. (a) Schematic of the test configuration. (b) Experimenatal set-up.

B A (b)

(a)

TABLE IV. DETAILS OF THE MEASUREMENT EQUIPMENT

Measurement Equipment Details

VCE / VDS Pico technology TA042 (100 MHz) differential probes

Inductor Ripple Currents

Agilent N2783B 100 MHz current probes

Oscilloscope Waveforms

Keysight 200 MHz InfiniiVision DSO-X 2024A Digital Oscilloscope

Input and output powers, voltages and currents

Newtons4th Ltd PPA5500 KinetiQ Precision Power Analyzer (0.03% basic accuracy)

(3)

(a)

(b)

Fig. 11. Comparison of turn-on (EON) and turn-off (EOFF) switching losses (900 V) with conduction losses (ECOND) at 20 kHz switching frequency at 50 % duty ratio while operating at 25 oC. (a) 25 A switching current (b) 50 A switching current.

TABLE III. DETAILS OF PASSIVE COMPONENTS

Component Value Details

RLOAD 4Ω (TE1500B1R0J)

Four 1 Ω, 1.5 kW chassis mount resistors connected in series.

L (20 kHz) 470 μH Ferrite core inductor

L (40 kHz) 235 μH Ferrite core inductor

Capacitors (CAL, CAH, CBL, CBH )

1000 μF

2 x Vishay MAL209527102E3 450V Electrolytic Capacitors connected in series

measurements have been plotted against the output power in Fig. 15. The graph also shows the operation of the same circuits at 20 kHz with a similar ripple current to the 40 kHz experiments. Even at 40 kHz, the highest recorded efficiency is demonstrated by the SiC MOSFET converter, i.e., 98.8% at 2.7 kW output power. SiC bipolar converters closely follows with a peak efficiency of 98.6% demonstrated by the SiC BJTs at 2.6 kW output power. At power levels below 1 kW, an improvement of efficiency can be seen for 40 kHz in all the converter. Around 1 kW, a drop in the efficiencies can be observed in the 40 kHz graphs. At this point the mean inductor current first exceeds half the ripple current, i.e. the inductor current goes from having positive and negative components to being continuously positive. This causes a change in the switching mechanism in that the current is carried by different diodes during the deadtime below and above this critical current level and this appears to change the losses in the circuit. Above 2.5 kW, the efficiency of the SiC MOSFET converter has decreased during 40 kHz operation. SiC BJT and SJT converters demonstrate similar efficiencies for 20 kHz and 40 kHz operation, between 2-5 kW power outputs. However after about 6 kW, the graph for the 40 kHz operation of the SiC SJT converter shows a trend of increasing efficiency.

IV. CONCLUSIONS

In recent years, a number of SiC devices have become commercially available. In this paper, we have assessed and compared the performance of 1.2 kV rated SiC MOSFETs, BJTs and SJTs with Si IGBTs. Forward characterisation and hard-switch characterisation with a double-pulse test set-up have been performed. Furthermore, continuous operation of the devices has been assessed and validated in a bi-directional converter designed for energy storage systems. The converters have been operated in the DC/DC voltage boost conversion mode, in order to allow a more straightforward comparison between the performances of the devices at a particular operating point. From the results presented, it is clear that the SiC devices outperform the Si IGBT in terms of switching losses and efficiency. Even at 1.2 kV, the conduction loss analysis reveals a clear advantage of SiC bipolar devices over

SiC MOSFETs, especially at higher current levels. The reduced on-state losses of SiC BJTs will be of greater benefit for applications requiring high powers and high voltage rated devices, such as HVDC voltage-source converters (VSCs). At the high voltages required in such applications (>10 kV), SiC MOSFETs will have significant conduction losses due to the large drift resistance arising from the unipolar nature. Despite being bipolar devices, SiC BJTs and SJTs show superior switching performance which is comparable to the SiC MOSFETs at lower current levels and exceeds at current levels above 40 A. SiC has a large critical electric field and this is beneficial for the design of bipolar devices in SiC with very low VCE(SAT) values, while avoiding operation in deep saturation. Consequently, achieving a faster switching performance. To date, the power electronics industry has been reluctant to adopt SiC bipolar technology as they are current driven devices. However, the results presented in this paper have been achieved with a simple and a compact driver design,

Fig. 15. Comparison of efficiencies of the SiC converters at 20 kHz and 40 kHz switching frequencies during 250V to 500 V boost conversion.

Fig. 14. Converter efficiencies measured for 250V to 500 V boost conversion.. FSW = 20 kHz.

Fig. 13. Converter operation for 5.7 kW input power. Converter A → SiC MOSFETs, fSW = 20 kHz, Converter B → SiC SJTs, fSW = 40 kHz. Switching voltages are for the low-side switches. Scale: IL(A) → 10 A/div, IL(B) → 10 A/div, VDS(A) → 200 V/div, VDS(B) → 200 V/div, Time → 10 μs/div.

VDS (B)

IL (B)

VDS (A)

IL (A)

with similar PCB dimensions to the voltage driven devices.

From the DC/DC conversion results for the tested power levels, SiC MOSFET converter demonstrated the highest efficiency at 20 kHz as well as at 40 kHz. However, SiC bipolar devices follow closely. From the double-pulse test results, it is expected that SiC bipolar converters will enforce a greater challenge to the SiC MOSFETs at higher output power levels. However, this could not be experimentally validated in the converters due to the limitations of the available test equipment. In these experiments, a lower value inductor has been used at 40 kHz compared to 20 kHz, keeping the ripple current constant and suffering only a small efficiency reduction. This demonstrates the ability to reduce the weight and size of passive components by operating SiC devices at higher switching frequencies while still achieving good efficiencies.

Therefore, it can be concluded that for 1-10 kW power applications such as the battery energy storage converters assessed in this paper, SiC MOSFETs have a slight advantage over SiC bipolar devices. Future work will include assessing the performance of the converters at higher voltage and power levels.

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