A Multi Channel, Wireless Telemetric Micro System

8/8/2019 A Multi Channel, Wireless Telemetric Micro System

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IEEE SENSORS JOURNAL, VOL. 6, NO. 1, FEBRUARY 2006 187

A Multichannel, Wireless Telemetric Microsystemfor Small Animal Studies

Chung-Chiun Liu, Edward OConnor, and Kingman P. Strohl

AbstractConventional means of collecting biophysiological pa-rameters in small animals often involve cumbersome direct wiringand/or restraint of the animal. At present, there is no system forvery small animals that can provide multichannel monitoring ofbiopotentials without restraining the animal or small enough insize or light enough in weight for studies with smaller animals. Forlarger animals, such as monkeys or larger rodents, systems havebeen proposed where the transmitter of the system has dimensionssuch as 2 5 2 5 1 3 cm3 and the weight is 9 g; this is fartoo high for smaller animals. Also, the battery life of that systemis relatively short ( 10 h). In this study, a multichannel wirelesstelemetric microsystem for biopotential monitoring in small ani-

mals, such as mice or rats, has been designed, fabricated, and eval-uated. This microsystem has four input channels with one calibra-tion channel. There are 8 channels on the chip, of which five, thefour electroencephalogram (EEG)/electromyogram (EMG) chan-nels, and the calibration channel, are now in use. The system canalso be expanded to more than eight input channels, if desired. Inthat case, a larger ASIC chip and larger circuit substrate might berequired, depending on the type of biopotentials being measured.The amount of ASIC and circuit substrate space consumption islarger for biopotentials such as EEG or EMG than for others suchas temperature or pressure. However, the same clocking-demod-ulation system could be retained up to 128 channels. The mul-tichannel telemetric chip for the present embodiment is approxi-mately 2 2 mm, and the overall size of the microsystem is ap-proximately 1 0 1 0 5 mm, including the enclosure package

and battery, with a total weight of 1 g. The power consumed bythis four-channel version, where two channels are EEG and twoare EMG, is 0.41 mW, and the fabrication process is AMI_ABN.There is a magnetic on/off provision. The microsystem has beenused to monitor EEG, Theta activity, and nuchal EMG in mice withexcellent results. This wireless telemetric microsystem can be ef-fectively used to record multiple biopotentials from freely movingsmall animals. This platform microsystem can be extended to in-clude other physiological parameters, such as temperature, pres-sure, and biological parameters.

Index TermsBiopotentials, electroencephalogram (EEG),electromyogram (EMG), multichannel, small animals, wirelesstelemetry.

I. INTRODUCTION

PHYSIOLOGICAL investigation of various biological is-

sues usually requires the collection of biophysical and

biochemical parameters in animal studies. It is often that small

Manuscript received June 29, 2004; revised August 3, 2005. The associateeditor coordinating the review of this paper and approving it for publicationwas Prof. Ralph Etienne-Cummings.

C.-C. Liuand E. OConnor arewith theDepartment of ChemicalEngineeringand the Electronics Design Center, CaseWestern Reserve University, Cleveland,Ohio 44106 USA (e-mail: [email protected]; [email protected]).

K. P. Strohl is with the School of Medicine, Case Western Reserve Univer-sity and the Louis Stokes VA Hospital, Cleveland, OH 44106 USA (e-mail: [email protected]).

Digital Object Identifier 10.1109/JSEN.2005.860358

animals, such as a mouse or rat, are used. Regardless of the

sensing elements employed for the biophysical and biochem-

ical parameters, conventional techniques to transmit the sensor

outputs to the external environment involve wire connections

and restraint tethering which limit the animal movement and

the recording conditions. It will be desirable if the transmis-

sion interface can be accomplished using a wireless telemetric

approach. This allows the monitoring of various biological

functions of an unrestrained small animal.

One multichannel telemetry system used a sequential conver-

sion of the input signals to a current to control a current-con-trolled oscillator based upon a monolithic chip [2], [3]. This

chip (3 3 mm) was complementary bipolar (BJT), and con-

tained a single set of amplifiers, reference circuits and a current

controlled oscillator (CCO). It lacked clocking provisions and

required the addition of commercial CMOS chips. Other limi-

tations included an inability for providing proper preamplifica-

tion, filtering, or input impedance for weak biological signals,

such as EEG or EMG. The overall package size (not including

the battery) was large, cm, precluding its use in

small animals.

Fryer et al. [4] described a multichannel telemetry system

using a time-sharing sequential multiplexing. However, the size

of the system, cm not including the battery, was largeand designed for signals such as strain gauges or electrocar-

diogram (EKG) rather than far more difficult to detect signals

such as EEG and EMG. Input impedance was 150 K for EKG.

Input-referred noise was 20 uV p-p at 50-Hz bandwidth. The

current drain was 2.5 mA using two 1.35-V mercury cells.

The frequency deviation of the FM transmitter was required to

be trimmed to match the discriminator of a system FM receiver.

No in vivo EKG recordings were shown.

Ruedin et al. [5] described a miniaturized EEG transmitter

with two asymmetric channels that was anchored to the skull of

a small animal with screws. The size was large,

cm. The input impedance was stated to be 2 10e6 ; forEEG and EMG recordings, this was internally shunted to 6.8

10e4 (68 K) . Other versions of EEG transmitters [6], [7] have

four single-ended rather than differential channels and had the

same construction and size disadvantages as that by Ruedin et

al. [5]. A system reported by Borbely et al. [8] had only a single

channel.

An intraperitoneal telemetry device that transmits EEG is

commercially available. However, this unit has only one EEG

channel and is large, cm in size. The single EEG

channel unit employs a pair of built-in silicone-insulated double

helix stainless steel EEG leads [9], and users cannot connect

their own electrodes.

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188 IEEE SENSORS JOURNAL, VOL. 6, NO. 1, FEBRUARY 2006

TABLE ICOMPARISON OF WIRELESS MICROSYSTEM PERFORMANCE

Commercial implantable transmitters to monitor preterm

labor by measuring pressure changes are available [10]. A

biotelemetry system for EMG and tendon force measurement

in rats has been proposed [11], as are implantable biotelemetry

transmitters for mice, sensing temperature and pressure, and

incorporating ion-selective microelectrodes and biosensors

[12]. However, the existing units in this type of devices are

fairly large (20 8 mm diameter).

A 128-channel EEG monitoring system was described using

time multiplexing with a clock rate of, e.g., 25.6 KHz [13]. This

system utilized large commercial integrated circuit packages

and discrete parts, and operated via a cable rather than telemetry.

However, in such applications, a larger chip using the system

described herein could accommodate more channels and enable

such EEG monitoring to be accomplished by telemetry rather

than cable.

Irazoqui-Pastor et al. [14] described a miniaturized neural or

EEG device, operating at an r.f. carrier frequency of 3.2 GHzusing analog FM. It was inductively powered and a 100-W Ham

radio transmitter driving a large external coil with a passive

impedance matching circuit was required. It was approximately

cm and required a 2.5-cm monopole transmitting

antenna. The power consumption was approximately 5.8 mW.

It appeared to be single channel. The input-referred noise of the

OTA portion of the design was 8 uV, but the crest factor ([15,

p. 299]) was not stated.

Mohseni et et al. [16] described a wireless neural mi-

crosystem with three signal channels and a marker channel

operating with FM in the 88108 FM band. The marker

channel was used for sequencing the channels in the systemsdemultiplexer. The size of the microsystem was 1.8 1.3 cm

without the two 1.5-V batteries. The power consumption was

2.44 mW. The input-referred noise was 7.5-uV RMS but

the crest factor was not stated. The device had not been tested

in vivo. The dc offset of the input signals was over a range of

0.25 V. The ASIC used the AMI ABN process but required

laser trimming, both to set clocking rate and to control the

amount of frequency modulation.

Takeuchi et al. [17] described a hybrid neural device which

used commercial parts. It was a single channel and operated

with an 8090 MHz FM r.f. carrier. It was tested with a 500-uV

test signal. The dimensions were relatively large, 1.5 0.8 cm,

not including any powering system. The power consumptionof this device was 10 mW. It operated for only 30 min with a

silver-oxide battery. It was mounted with an adhesive material

directly on the back of an insect.

DeMichele et al. [18] described a 16-channel inductively-

powered system for neural or EEG signals using an ASIC

of .46 .46 cm, fabricated in the AMI ABN process. The

overall package size with an enclosure or encapsulation was not

specified. It drew 3.8 mA at 4.75 V with a power consumption

of 18 mW. It required gain adjustment. The range of the

transmitter was about 3 ft at 385 MHz, with a 1 antenna

connected to the transmitter. The r.f. bandwidth was specified

as 15 MHz and the device was tested with a signal consisting of

a 100-uV 6-Hz square wave, but it was not tested in vivo. The

stated amplifier input-referred noise was high, e.g., 121-uV

RMS. Switching noise injected by the scanning process was

found to be a significant problem. The amplifiers demonstrated

an operating point shift in the presence of r.f. interference

including the VCO.

Harrison et al. [19] described a neural amplifier built in a1.5- m CMOS process (AMI_ABN), with six amplifiers on a

2.2 2.2 mm chip. The amplifier was designed with MOSFETs

and on-chip capacitors. The supply voltage was 5-V split-supply

and it had a frequency range for use with neural electrodes

from millihertz to 7 KHz. The measured input-referred noise

was 14-uV p-p. The RMS was 2.2 uV; this would lead to a

crest factor of 3.8 [15, p. 299]. The neural amplifier was re-

designed for low-frequency biosignal applications such as EEG

or brain-surface electrodes, to exhibit a bandwidth of below 1 to

30 Hz. It was stated that the input-referred noise voltage for the

EEG version was 1.6-uV RMS. There was a neural waveform

recording but there was no in vivo EEG recording presented inthis study.

Table I summarizes the results of our telemetry system and

those offive others referenced in this paper. The input-referred

noise, power dissipation, detectable signal, and transmission

range, as well as the parameters of telemetry link frequency,

number of data channels, power supply, system clock frequency,

communication scheme, number of external components, total

weight, and package dimensions are presented. Our system

shows the lowest input-referred noise, and our system has the

lowest power dissipation of a complete telemetry system. By

comparison, our system has a small footprint. The telemetry link

frequency of our system is higher than those of [ 16] and [17],

but lower than those of [14] and [18]. Our system has more datachannels with the exception of [18]; in that system, a much larger



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LIU et al.: MULTICHANNEL, WIRELESS TELEMETRIC MICROSYSTEM 189

ASIC chip is used. Table I also provides information on the total

weight (where provided by the investigators) and the power

requirements of the systems cited, and our system compared

favorably in these aspects. The communication scheme of our

system is amplitude-modulation-based (onoff keying) while

those of the other complete telemetry systems are FM-based;

this use of an FM-based system is apparently associated withhigher power dissipation. The power supply voltage of our

system also compares well with other systems.

The advance in microfabrication techniques as well as micro-

electronics in recent years provides an opportunity to develop

microsize, wireless telemetric interface suitable for small an-

imal monitoring. Sensors for physiological parameters include

those for common parameters such as EEG, EMG, EKG, pres-

sure, temperature, and also for various chemical and biochem-

ical parameters. Thus, a wireless telemetric interface needs to

be capable of transmitting these types of sensor outputs.

EEG refers to recording graphically the electric activity of

the brain, particularly the cerebral cortex, by means of exter-

nally placed electrodes. The frequency range is 0.5100 Hz.

Theta EEG refers to a recording from the temporal region,

having a frequency range of 47 Hz. Problems associated

with recording EEG are low signal levels and high 5060 Hz

power line interference. EMG refers to recording the electrical

impulses that pass through a muscle as it contracts and relaxes.

EKG refers to recording electrical impulses as they vary during

the cardiac (heart) cycle.

In this study, we had developed a multichannel (four channels

with a calibration channel) wireless telemetric microsystem for

EEG and EMG monitoring for small animals. Specifically, this

microsystem was used for the study of sleep disorders using

mice as the animal model. From the physiological viewpoint,sleep apnea is initiated and sustained by instability in the

respiratory control system [20]. Short-term potentiation of

ventilation (STP), also called ventilatory after-discharge,

is evoked by brief hypoxia, promotes ventilatory stability,

and protects against dysrhythmic breathing or posthypoxic

frequency and ventilatory decline [21]. An absence of STP

promotes the appearance of repetitive apneas, as supported by

studies on patients with obstructive sleep hypo-apnea syndrome

(OSAHS) [22] or congestive heart failure (CHF) patients

with Cheyne-Stokes respiration (CSR) [23]. Hence, ventilatory

instability and periodicity are common to CSR, OSAHS, and

the appearance of periodic breathing at altitude [24]. Centraland obstructive apneas may occur in the same patient over a

night [25]. These studies indicated that posthypoxic behavior

and periodicity are fundamental features in the pathophysiology

of sleep apnea syndromes.

The complexity of understanding the pathophysiology of

sleep disorder would require the investigation of certain organs

such as the brain involved in both central chemosensory and

coordination of chemical and nonchemical reflexes. Such inves-

tigation would need to monitor the EEG and EMG developed

from a small animal model. It would be important that the

animal is not restrained during this study. Thus, it is mean-

ingful to have a multichannel wireless microsystem capable

of monitoring EEG and EMG that can be used in studies withunrestrained small animals.

Fig. 1. Transmitter package (ca.1 2 1 2 0 : 5

cm).

II. DESIGN AND FABRICATION OF THE

INTERFACE MICROSYSTEMIn our laboratory, a low-power 2 2 mm integrated circuit

signal processor chip was developed and applied to a transmitter

package as shown in Fig. 1, that could be anchored to the head

of a small animal such as a rat or mouse. The dimensions of

the transmitter including the enclosure package and battery are

approximately cm. The vertical spacefor the circuitry

in the circuit compartment of the package is about 1.1 mm.

The input signals of the bipolar (differential) EEG or EMG

channels modulated the period of a sub-carrier oscillator and

the time-multiplexed sub-carrier oscillator output was converted

to pulses, which gated a wireless transmitter on and off. The

wireless transmitter was an r.f. oscillator with a tank coil as thetransmitting antenna. The pulsed RF output from the tank coil

or other antenna structure was then picked up by a radio re-

ceiver, which drove a demodulator to reconstruct the individual

input signals and output them to a PC with waveform acquisi-

tion software.

The digital clocking on the chip provided for up to eight mul-

tiplexed channels; four signal channels plus a fifth on-chip cal-

ibration channel. The purpose of the calibration channel was to

produce a reference output the amplitude of which corresponded

to an EEG or EMG input level of 50-uV p-p. It also served as

an error detector in that it indicated by its frequency, waveshape

and output channel that the signal was properly received by the

radio receiver and processed by the demodulator.For animal study chronically implanted brain or muscle elec-

trodes were connected to the transmitter and used to record

spontaneous or evoked brain potentials (EEG) and neck muscle

activity (EMG). The package itself could be anchored to the

skull with cranioplastic cement or fitted with pins for insertion

into a small socket mounted on the animals head.

The details of this multichannel wireless telemetric mi-

crosystem are given in the following sections.

A. Technology

A 1.5- 2-metal 2-poly CMOS process with an NPN op-

tion, which can operate at 3 V, was used to implement an in-tegrated circuit chip providing the main part of the circuitry for



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Fig. 2. Block diagram of the ASIC.

the telemetry transmitter package. This fabrication process al-

lows p-channel and n-channel enhancement mode MOSFETs,

bipolar NPNs and other electronic structures such as resistors,

capacitors and diodes to be integrated. This process, AMI ABN

[16], [18], [19], and [26], is applicable to low-noise analog de-

signs or mixed-signal designs.

B. Process Characteristics

Resistors can be fabricated from n-wells in this process. Poly1and poly2 can be used to form on-chip capacitors. Diodes may

be laid out for the bonding pads for ESD protection. Guard

rings may be laid out for signal isolation between circuit blocks

and for latch up prevention. An NPN option can be used for

BiCMOS digital speed-up purposes and was used for analog

purposes in this application.

C. Circuit Considerations

This CMOS process having an NPN option [26] was used so

that along with the digital clocking circuitry, the bipolar devices

could be used to provide a higher transconductance, a better

matching, lower offsets and lower flicker noise for the analogEEG and EMG amplifiers that were needed for the application.

A block diagram of the ASIC chip is shown in Fig. 2. The cir-

cuit was designed with 4 input differential preamplifiers, four

selectable second-stage amplifiers that could activate sequen-

tially from pulse inputs from the clocking circuitry, and a cur-

rent-controlled subcarrier oscillator. The gain of each selectable

second-stage amplifier was a function of the pulse current of the

select pulse for that channel. The gain of each input differential

preamplifier stage was fixed. Between each input differential

preamplifier stage and the associated selectable second-stage

amplifier was a pair of capacitors (indicated between dashed

lines), of which one coupled the signal and the other provided

roll-off to the band of signal frequencies. As previously indi-cated there was a calibration circuit for producing a reference

signal. There was also CMOS clocking circuitry, a block of

monostable multivibrators with outputs combined by gating to

provide output pulses and synchronization pulses, and a power

toggle-on/toggle-off switch circuit.

The power switch was toggled by an external magnetic sensor

such as a Hall-effect sensor or a reed switch turning the trans-

mitter on and off with a magnet. The Hall-effect sensor was

an SMT having a footprint of mm and was incorpo-

rated onto the circuit substrate of the hybrid package along withthe ASIC and the bare-die BJT r.f. oscillator/transmitter chip.

A reed switch selected from currently-available types would

have required a slightly larger overall package. The Hall-effect

sensor, a pole-independent device with a latched digital output,

worked by producing an output which went high to low as a

small magnet was brought near and went low to high again when

it was withdrawn, thus toggling the chip supply voltage via the

on-ASIC power switch circuit mentioned above. The onoff cir-

cuitry of the ASIC does not have significant static power con-

sumption; the Hall-effect sensor, an Allegro A3212 ELHLT, has

a static power consumption of 15 uW. The manufacturers data

sheet is numbered as Allegro Microsystems 27 622.61G.

The set of four selectable amplifier stages converted the input

EEG or EMG signals into linearly proportional output currents,

which, along with a reference current from the calibration cir-

cuit, were fed into the current controlled subcarrier oscillator.

These currents were fed in sequence so that the channels were

time-division multiplexed. A diagram describing the encoding

scheme is shown in Fig. 3. Each channel was turned on for two

complete cycles of the subcarrier oscillator; a half-cycle on each

end of this period served to provide setup time, and one sub-

carrier oscillator cycle was used for the data measurement of

an input channel or for the calibration signal. Each channels

value was encoded in the duration between RF pulses, i.e., the

subcarrier oscillators period was modulated by the channelsvalue. An increase in input current to the subcarrier oscillator



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Fig. 3. Encoding scheme diagram.

Fig. 4. Timing and reference/calibration signals diagram.

resulted in a shortening of the period. If the subcarrier oscil-

lator is operated at 1015 kHz it will produce a minimum sam-

pling frequency per channel of 625 Hz and a Nyquist frequency

of 312 Hz, which exceeds the 100-Hz bandwidth which is re-

quired for the EEG and EMG signals in this application. The

CMOS timing circuit was a chain of toggle-connected master-

slave D-type flip-flops, some of which drove a 1-of-8 logic de-

coder block. Three more input channels could have been used

with the 1-of-8 decoder arrangement, but the physical space was

not available on the 2 2 mm chip to accommodate the input

amplifiers for the three extra channels. Additional toggle-con-

nected D flip-flops divided down the clock signal from the sub-

carrier oscillator to about 6 Hz, which was used to drive thecalibration circuit and inject a square wave as a reference signal

onto one of the multiplexed channels. A timing and reference

signals diagram is shown in Fig. 4.

Simulated waveforms at circuit nodes are shown in Figs. 57.

Fig. 5 simulates the output of an input channel amplifier-pream-

plifier pair, having an input signal of 70 Hz at 5 000-uV p-p am-

plitude. The second amplifier is gated on and off by the select

pulse input for that channel. In this simulation, the select pulse

is actually a SPICE pulse rather than an SCO clock pulse; the

period of the SCO clock would actually be varying. The output

current is 70 uA p-p. Fig. 6 simulates the select pulse (with pe-

riod held constant for simplification purposes) for the channel.

The amplitude is 3 V, the on time is 162 uS and the frame time is

1296 uS. Fig. 7 simulates the frequency response of the channelfor a differential signal input level of 50 uV p-p.



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Fig. 5. Simulation of output of signal channel stage.

Fig. 6. Simulation of channel amplifier select pulse.

Fig. 7. Simulation of channel amplifier frequency response.

Separate parts of the block diagram are shown in Figs. 811.

They are the input preamplifier, Fig. 8, the subcarrier oscillator

(SCO), Fig. 9, the divide-by-128 section (DFFs 511), Fig. 10,and the monostable multivibrators block, Fig. 11. In the SCO,

the magnitudes of the current sources are controlled by an ex-

ternal current, e.g., the current from a signal channel amplifier

or from the calibration section. A latching circuit is formed fromtwo 2-input CMOS NAND gates. The positive feedback around



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Fig. 8. Signal channel input preamplifier stage diagram.

Fig. 9. SCO block diagram.

Fig. 10. DFFs 511 (divide-by-128) block diagram.

the loop is used to ensure that only one of the MOS transis-

tors is on at a time. The switching points of the comparators

combined with the current source determine the oscillator fre-

quency of the SCO. Two 22-pF timing capacitors were required

for the SCO and they were implemented by very small 0402 or

0201 external chip capacitors, which were mm or

smaller. The divide-by-128 chain of toggle-connected CMOS

DFFs 511 provides a low frequency for the calibration signal.

In the Monostable Multivibrators block, each one-shot consists

of a 2-input CMOS NOR gate, a CMOS inverter, an on-chip ca-

pacitor, and an on-chip resistor. The trigger input of a one-shot

can be longer than the output pulse width.

The power for the chip and transmitter unit was suppliedfrom a CR1025 3-V watch cell which was located in the bat-

tery compartment of the enclosure package, above the circuit

compartment. The battery was located inside of the wireless

transmitter coil, which was wound around the outside of the

enclosure package of the transmitter unit.

The battery was an Li/MnO2 watch cell with a nominal

voltage of 3 V and an average capacity of 30 mAh to 2.0 V.

The volume of the battery is 0.2 cm . It is speci fied by the

manufacturer that if the load is such that the current drain is

64 uA and the operation is 24/7 the time to cutoff voltage (2.0

V) is 467 h. The wireless transmitter drew 40 uA and the Hall

sensor drew 12 uA; the ASIC drew 93 uA. The current drain

of the telemetry hybrid transmitter including ASIC, wireless

transmitter and Hall-effect sensor was 145 uA, and, thus, thissystem could operate for 168 h (7 days, 24/7) or longer. It was



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Fig. 11. Monostable multivibrators block.

found by measurement that the CR1025 dropped below 3 to

2.9 V within a few hours; it then gradually reached 2.83 V

and remained at this level during most of its lifetime. When

it reached the end of its lifetime it then dropped within a few

hours, to 2.0 V. The effective supply voltage of the ASIC andhybrid transmitter was, thus, approximately 2.83 V.

The power consumption of the entire hybrid transmitter was

0.41 mW. This was considerably less than that of other multi-

plexed or wireless designs referred to above.

This device, like most battery-operated devices, was subject

to the effects of battery drop and parameters were expected to

change somewhat during operational life of the battery as a re-

sult. Such changes, as,e.g., changes in channel gains, were small

due to the use of on-chip voltage regulators and the use of cur-

rent mirrors tending to act as current regulators. However, it was

found that the change in signal amplitude was tracked by the

change in calibration waveform amplitude over the range of 3.2to 2.7 V. The amplitude change for both signal output wave-

form and calibration channel output square-wave amplitude was

2.0% per 0.1-V change in battery voltage over the range of 3.0

to 2.7 V. Thus, the amplitude change throughout the operational

life of the battery was small but would be correctible by refer-

encing the demodulators signal-channel gain to the calibration

square-wave amplitude.

The conductive substrate in epi wafers can have currents in-

duced into it by the magnetic field of a (spiral) on-chip inductor,

and this inductor-induced noise through the substrate can affect

other circuits on the same chip [27]. Although our device did not

utilize an on-chip inductor, the integrated circuit chip used was

within the field of an inductor. No degradation of performancewas observed in the miniaturized unit in comparison to larger

prototypes with the coil remote from the chip.

D. Circuit Implementation

The ASIC constituted the processor section of the complete

telemetry link. The only requirements to complete the imple-

mentation of the telemetry function of the transmitter were a

single external bare-die BJT chip, a resistor and a capacitor on

the hybrid substrate, plus a tank circuit. The resistor could have

been included on the ASIC but it was put instead on the cir-

cuit substrate so that it would be physically closer to the wire-

less oscillator section and, thus, tend to better decouple RF fromthe ASIC. The transistor and capacitor were not laid out on the

Fig. 12. Microphoto of ASIC chip (ca. 2.22

2.2 mm).

ASIC because of considerations of space and flexibility in re-

gard to the wireless link, as well as to avoid any problems which

might arise from having an RF generating circuit directly on

the ASIC in close proximity to analog circuitry which was pro-

cessing low-level signals such as EEG and EMG. A micropho-

tograph of the ASIC processor chip of the telemetry system is

shown in Fig. 12.

The block diagram of the ASIC chip was shown previously in

Fig. 2. The input stages utilized dc blocking capacitors of very

small physical size (0402 and 0603) on the hybrid substrate insuch a way that capacitive dc blocking, a standard precaution

with EEG amplifiers, was implemented.

The input impedance of the input preamplifier stages was re-

lated to circuit parameters and represented a tradeoff between

stage current drain, electronic flicker noise and chip size. The

measured input impedance was 670 K and was found to be

adequate for the EEG and EMG signals of the application; how-

ever, in future versions, this impedance may be increased to

3.8 M as indicated below. The in vivo electrode-tissue inter-

face impedances of the animal electrodes used in the in vivo

system testing were relatively low. Information about the an-

imal electrodes used is given later in this paper.

In a test, the result of loading (reducing the input impedance)of standard Grass EEG amplifiers was examined for in vivo



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mouse EMG and EEG at 300 K resistive shunt impedance.

In vivo EMG signals were not influenced by the 300-K shunt

across the input terminals of the Grass amplifier. The in vivo

EEG signals were not significantly attenuated (less than 3 dB

below 5 Hz) by the 300-K shunt. The higher frequency

components were not noticeably attenuated. The equivalent

electrode impedance appeared to increase as the frequencydecreases; however, the signals appeared entirely usable even

at 300 K.

It is of interest that earlier, transistorized commercial and re-

search EEG equipment utilized lower input impedances than the

value mentioned above; in [5], cited above, the input impedance

was stated to be approximately 2 10e6 ; however, for EEG

and EMG recordings, this was stated by the authors to be inter-

nally shunted to only 6.8 10e4 (68 K) .

Noise is considered to be a dominant factor in EEG equip-

ment, particularly in CMOS input amplifiers [14], [16], [18],

and [19] where the electronic (circuit-generated) noise, such as

flicker noise, is often too high. In EEGequipment, both the RMS

and the crest factor [15, p. 299] should be stated in equipmentspecifications but frequently are not.

In this development, BJTs were used for input stages of the

EEG/EMG telemeter because the flicker noise of MOSFETs is

known to be as much as 10 to 1000 times larger [15, p. 123]

unless space-and-power-consuming techniques are used.

In a BJT, the total equivalent input noise is

Also

where ;

; ;

;

' ;

;

;

: :

. The above equation for Eni^2 [15,pp. 142143] is valid for a 1-Hz bandwidth at the frequency f.

Increasing the transistor quiescent collector current increases

the 1/f noise [15, p. 126]. As the collector current drops, elec-

tronic 1/f noise of the stage may be expected to decrease, while

small-signal input impedance rises. The CMRR of long-tailed

pair Q1, Q2 in Fig. 8 increases as emitter resistor R7 is in-

creased. Thus, because collector current drops but CMRR in-

creases as R7 increases, R7 was made as large as space on the

chip permitted. Because R1 and R2 shunt the input impedance

of the stage, they were also both made as large as space per-

mitted. The resulting input impedance was 670 K but simula-

tion showed that the unshunted input impedance was 4 M .

R1 and R2 may, thus, be replaced in future versions by currentsources to increase the channel input impedance. R7 may also

be replaced by a current source to further increase the CMRR

in future versions. The output node of the input preamplifier

stage EXT1, drove filter capacitors C1 and C2, connected to

nodes EXT2 and EXT3 of the following stage. The purpose of

C1 and C2 was to set the high and low frequency roll-offs as

narrow as possible, to pass only the spectrum required, since

the greater the amplifier bandwidth the greater the output noiseand input-referred noise [15, p. 125]. Essentially, C1 of Fig. 8

may be regarded as coupling the signal to the associated se-

lectable second-stage amplifier (not shown in Fig. 8 but indi-

cated in Fig. 2) while C2 may be regarded as providing roll-off

to the band of signal frequencies. These capacitors, indicated

in Fig. 8 as being separated from the amplifier circuitry by a

dashed line, are also indicated (between a pair of dashed lines)

in Fig. 2, and separated from the input differential preamplifier

stage by the first of the dashed lines.

The input-referred noise of the EEG channels, from trans-

mitter channel input to system demodulator output, was found

by measurements and calculation [15, p. 275] to be 0.69 uV

RMS with a crest factor [15, p. 299] of 3.84 at 100-Hz BW,and 0.36-uV RMS with a crest factor of 3.69 at 30-Hz BW.

The input-referred noise of the EMG channels was found to be

0.63-uV RMS with a crest factor of 3.84 at 100-Hz BW.

Thus, the EEG signal processing in this system exhibited less

noise than the systems using CMOS circuitry for signal pream-

plification [14], [16], [18], and [19].

E. Wireless Transmitter

The train of output pulses from the ASIC, which processed

the input EEG and EMG signals and converted the signals to

a pulse format, was applied to the input resistor R1 of an r.f.

oscillator used as a wireless transmitter, as shown in Fig. 13.It was an Armstrong oscillator in which the collector winding

coil L1 and the tank circuit winding coil L2 of the transformer

were combined into an autotransformer single tapped tank coil

L, serving as the transmitting antenna of the telemetry system

and having a tuning capacitor C1 across it. Coupling capacitor

C2 provided feedback to the base of Q1. Capacitor C3 placed the

tap point at r.f. ground. Resistor R1 provided base current and

set the forward bias on Q1 when the output pulses of the ASIC

were present, thereby causing a burst of r.f. oscillation during

each ASIC output pulse. The circuit operated class C during

the intervals when it was gated on by the output pulses of the

ASIC. Because of the flow of r.f. current in L, an r.f. electromag-

netic field was generated by the coil; this r.f. field then served to

propagate the wireless signal. The use of crystal control was not

found necessary. Because the transistor and coupling capacitor

were not incorporated on the ASIC, any r.f generating circuit

which can be onoff keyed, using any carrier frequency or any

antenna type, of sufficiently small footprint, might be substi-

tuted in this system.

F. Demodulator

The train of transmitter pulses from the output of the radio

receiver was applied to the input of the demodulator of the

system, a block diagram of which is shown in Fig. 14. An

LED was turned on by a pulse-sensing circuit to indicate signalreception from the transmitter. These pulses were applied to



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Fig. 13. Wireless transmitter circuit diagram.

a pulse separator circuit which separated the active interval,

or signal channel pulses, from the synchronization, or marker

pulses which were also sent by the transmitter, so that the

demodulator was able to sequence the channels. When the

marker and signal interval pulses were separated, the signal

interval pulses were applied to the clock input of a flip-flop;

a pulse at the beginning of a signal channel interval set the

flip-flop output high and the pulse at the end of the channel

interval reset the flip-flop output low. Thus, a series of pulses

were generated at the output of the flip-flop, whose high levels

corresponded in duration to the signal channel intervals. These

high levels were then used to gate a clock input of a counter

in such a way that when the flip-flop output pulses were high

the counter up-counted, and when the flip-flop pulses went

low, the up-counter stopped counting. At the end of each signal

channel interval of the transmitter, a binary (digital) output was,

thus, generated. This digital output was applied to a D-to-Aconverter, which converted the digital to an analog voltage,

corresponding to the EEG or EMG signal sample, and held this

sample until the next frame of signal channel intervals. The

up-counter was then reset and at the start of the next signal

channel interval the process started again. There were eight

D-to-As, corresponding to eight channels, of which four were

used for signals, one was used for the calibration signal and

three of which were reserved for future ASIC versions. At the

end of each frame of eight channels, the flip-flop mentioned

above was reset by the marker pulses from the pulse-separation

circuit mentioned previously. The output of each D/A was

then passed through a dc amplifier/filter to eliminate the D/Aincrement-noise and calibrate the signal level so that 1 V p-p

output from the demodulator corresponded to 50-uV p-p input

to the transmitter. The output of each amplifier/filter was then

passed through an additional low-pass filter to limit channel

bandwidth to 100 Hz, for the purpose of reducing electronic

flicker noise [15, pp. 19-20]. The amplified and filtered signal

was then passed through a dc block and a buffer to eliminate

dc offset. In addition, because the subcarrier oscillator of the

transmitter was current-to-frequency-modulated, a 1/f converter

was added between the low-pass filter and the dc blocking filter

in the four signal channels. Although eight channels were

sent from the transmitter, only four signal channels and one

calibration channel were used, so that there were five outputamplifier/filter/buffer circuits in the demodulator. Any channel

mismatch on the chip was trimmed out in the demodulator as

the system was calibrated channel-by-channel with a 50-uV

p-p signal being applied to the channel input of the hybrid

transmitter containing the particular ASIC and the channel was

trimmed for a 1.0-V p-p output signal from the corresponding

output of the demodulator unit being calibrated for that specific

transmitter. In the future, the use of a different BiCMOS typeof process in which p-n-ps as well as n-p-ns are available

may allow closer on-chip matching between channels, by

using BJTs to replace less-well-matched MOSFETs in analog

circuitry, thus allowing for less need for trimming.

G. Radio Receiver

The output of the systems oscillator/transmitter was sensed

using a small antenna such as a half-dipole or loop, which fed a

single-conversion receiver, a block diagram of which is shown

in Fig. 15. The receiver incorporated a front end, an i.f. section

and a second detector. A commercial front end having a tuning

range of 50810 MHz in four bands was used in this receiver.

An internal switch was used to select the correct band for re-

ceiving the transmitter. The i.f. bandwidth of this receiver was

adjusted to correspond to the spectral content of the transmitter

pulses while rejecting external interference; the i.f. circuitry was

broadly-tuned and the bandwidth was in the range of 2 MHz.

The receiver gain was limited by the adjustment of the front-end

and IF gains, to suppress interference of extraneous signals and

noise with the transmitter pulses. The second detector circuit in

this receiver produced a pulse output, which corresponded to

the pulse output of the transmitters ASIC chip, which onoff

keyed the transmitter units wireless transmitter circuit. The re-

ceiver second detectors pulse output was fed to the demodu-

lator unit, to be processed in order to recover the transmitterchannel interval and marker signals, and subsequently the EEG

or EMG input signal information. The systems receiver also in-

corporated an audio section, which produced an audible signal

to facilitate tuning the receiver to the transmitter signal. The RF

receiver is not a commercial one and it, therefore, has no manu-

facturer information or data sheet reference; however, the front

end is a Zenith 175-00 014 CATV tuner made by Zenith Elec-

tronics Corporation, Glenview, IL.

H. Miniature Enclosure for Telemetry Transmitter

The miniature enclosure for the telemetry transmitter was

a micromachined box that could be made of alumina ceramicor macor ceramic. A microphotograph of the box is shown in

Fig. 1. A cover slid in low-friction micromachined grooves, and

served to clamp the battery against the internal battery contacts

as well as to hold the battery. The battery shelf also acted as

the top of the circuitry compartment, which was only 1.1 mm

in height. The enclosure box was set on the circuit substrate,

which carried the ASIC chip, a SMT Hall-effect sensor, an R.F.

oscillator chip BJT, and the various chip capacitors and a chip

resistor; the .22- and .001-uF chip capacitors and the chip re-

sistor were 0402 or 0201 size and some of the remaining chip

capacitors were 0603 size. The ASIC chip and the chip BJT were

connected to the circuit substrate with wire bonds. Flip-chip at-

tachment could also have been used. A photograph of the com-plete multichannel wireless telemetry circuitry on the alumina



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Fig. 14. System demodulator block diagram.

Fig. 15. System radio receiver block diagram.

Fig. 16. Microphoto of transmitter hybrid substrate (ca. 12

1 cm).

substrate is shown in Fig. 16. The size of this circuit substrate

was cm and was smaller than the footprint of a U.S.

dime. It may be assembled manually or by automated methods.A transmitting coil of a few turns of wire of approximately #40

TABLE IITABLE OF OFF-CHIP COMPONENTS OF THE TRANSMITTER

gauge was wound around the top of the enclosure, leaving clear-

ance for the sliding cover. The assembly was held together by

a cement. Leads from the electrodes implanted in the animal,

or from a connector on which the box might be mounted, were

brought to the side or bottom terminals. The replacement of

the battery was relatively simple by sliding the top cover of the

miniature enclosure to access the battery.

A list of the off-chip components of the hybrid substrateand enclosure package is shown in Table II. These off-chip



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Fig. 17. B6 Mouse recording (active) made with ASIC system.

Fig. 18. B6 Mouse recording (slow wave) made with ASIC system.

components are annotated as to the components shown in the

schematics included in this paper.

III. RESULTS AND DISCUSSIONS

After the ASIC chip was fabricated and the hybrid trans-

mitter was built, the in vivo performance of this system wasassessed by recording the EEG and EMG waveforms of small

mice. Two C57BL/6J and two A/J mice (Jackson Laboratory,

Bar Harbor, ME) were implanted with stainless steel electrodes

for the recording of the cortical and theta EEGs and of the EMGs

of the nuchal (neck) muscles. A midline incision was made to

expose the skull and neck muscles posterior to the skull. Two

pairs of stainless steel wires 0.21 mm in diameter and stripped

for 0.5 mm at the ends were surgically placed to contact the durafor bipolar theta and cortical EEG recordings. Two other pairs of



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Fig. 19. AJ Mouse recording (REM) made with ASIC system.

Fig. 20. B6 Mouse recording (slow arousal) made with ASIC system.

stainless steel electrodes made by knotting stainless steel wires

and stripping the knotted portion were sutured into the surface

of the neck muscle for bipolar EMG recordings.

The electrodes were connected to the telemetry transmitter

and the animal was placed in a Lucite chamber (10 cm in di-

ameter and 6 cm high) with bedding and food and water. Eachanimal was studied for three days.

A data analysis program was used to view the demodulated

EEG and EMG data via an analog conditioning filter/amplifier

(CWE, Inc.) and a 12-bit Data Acquisition System (National In-

struments PCI-MIO-16E) or 16-bit PCI-6033E DAQ. The Data

Acquisition System (DAS) was used in a LabView environment

on a P-III desktop computer. The resultant data was sampledat 512 Hz and stored on a hard disk. Records were scored for



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TABLE IIITABLE OF SPECIFICATIONS OF THIS SYSTEM

sleep stage, i.e., assignment was made of the kind of waveform

as, e.g., Waking, Slow-Wave, or REM.

For each mouse, several segments of data, each about 5 h

long, were recorded. The data was then analyzed. Example

waveforms telemetered by the telemetry system are shown.

Each recording has visible at the left the labels EEG, Theta,

EMG1, EMG2, CAL. Each recording also has vertical bars

in the center that are labeled to show the signal level corre-

sponding to the height of the bar.

Fig. 17 shows cortical EEG, Theta activity and nuchal EMG

from a B6 mouse in active wakefulness. Fig. 18 shows the EEG,

Theta and EMG from a B6 mouse in slow wave sleep. Fig. 19

shows a set of waveforms from a B6 mouse in slow wave sleep

with arousal from sleep. Fig. 20 shows signals from an AJ

mouse in REM sleep. The calibration signal, a square wave of

56 Hz, is shown on the bottom channel of each recording.

The bandwidth used for the cortical EEG and the EMG was

100 Hz. The bandwidth used for the theta EEG was reduced

by the recording instrumentation to improve the display of the

characteristics of this waveform parameter.

By referring to the waveform recordings, which represented

the entire system as well as the ASIC and hybrid transmitter, itcould be seen that system CMRR was sufficient so that the effect

of 5060 Hz power line interference, often a problem with EEG

systems, was not observed. The system channel crosstalk effects

were visually absent, since clearly evidence of the calibration

square wave was not seen in the signal channels, or vice versa,

and evidence of the EEG waveforms was not seen in the EMG

or calibration channels, or vice versa.

The calibration signal served an important purpose in therecordings in that it indicated at all times during the recordings,

by continuously providing a waveform of known wave-shape,

frequency, and amplitude, that the system was working and that

r.f. transmission and reception were being properly achieved. It

indicated also that demodulation was properly accomplished,

and all channels including signal channels were being displayed

in their proper positions on the recording. In other systems

there has often been no indicator to assure that transmission

was being adequately accomplished or that channels were

being properly demodulated or output in proper sequence,

so that, for example, what might have seemed to be EEG

might have been not a true EEG waveform but possibly only

a system artifact. The calibration channel, thus, served thepurpose of an error detector in the recordings as well as an

amplitude reference.

In the recorded waveforms, the noise levels represented

the resultant of all noise sources, including digital, 5060 Hz

power line interference, and ambient r.f. interference as well as

electronic system noise such as flicker, shot and thermal. There

was good signal integrity as compared with, e.g., recordings

made directly with Grass amplifiers, despite these noise sources

and despite changes in light and temperature as well during

the in vivo recording experiment. The measured EEG channel

electronic noise parameters at 30-Hz BW have been previously

stated in this paper to be 0.36-uV RMS with a crest factorof 3.69, and at 100-Hz BW to be 0.69 uV with a crest factor

of 3.84. This was the overall system noise from the channel

input of the hybrid transmitter to the channel output of the

system demodulator. It is evident that this noise is considerably

less than that of the amplifier alone at 30-Hz BW in [18]

above, which is an all-CMOS design in the same process [26].

In the future, however, the use of another BiCMOS type of

chip process where both polarities of BJTs are available, may

allow still further reduction in noise to be achieved, by using

BJTs to replace noisier MOSFETs in the analog circuitry.

Specifications of the system are indicated in Table III. A com-

parison of this wireless microsystem with other wireless mi-

crosystems is shown in Table I, above. In Table III, the parameter

RF BW is measured on the bench by tuning a calibrated com-

mercial telemetry receiver between the points where the signal

reception from the telemetry transmitter drops off. The commer-

cial receiver has switchable bandpass filters with bandwidths

suitable for television testing purposes, e.g., several megahertz.

IV. CONCLUSION

We have designed and tested a multichannel wireless

telemetry system for up to four biopotential signals plus a

calibration channel for use in the monitoring of small animals

(mouse or rat). It is expandable to more channels. The systemhas a package size of approximately cm (including



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the enclosure package and the battery power source) appropriate

for monitoring small animals (mouse or rat). The footprint of

the complete package including ASIC, BJT transmitter chip,

Hall-effect SMT, and all other parts including enclosure, is

smaller than that of a small coin, such as a U.S. dime. To

achieve specifications of such small size, a monolithic I.C.

chip, i.e., ASIC, was fabricated. The I.C. chip is a low-powersignal processor chip, only 2 2 mm in size. This signal

processor chip amplifies, filters and time division multiplexes

the signals that are in turn transmitted via an RF link, i.e.,

BJT chip and associated parts as described above, contained

within the package, to an external radio receiver. The receiver

drives a demodulator to reconstruct the individual signals for

display or analysis by waveform acquisition software. Finally,

the system incorporates a Hall-effect sensor, i.e., an SMT also

in the package, as mentioned above, providing magnetic onoff

capability, initially for conservation of power, but which also

could be used for interactive procedures.

This development demonstrates the feasibility of recording

of multiple biopotentials using the miniature telemetric systemfor a freely moving small animal. The ASIC chip design used

in the telemetry system is flexible and can accommodate more

channels and both unipolar and bipolar signals, as well as other

physical and biochemical sensor outputs. This forms the tech-

nical foundation for future research in this wireless telemetric

microsystem for small animal study.

ACKNOWLEDGMENT

This study was approved by the IACUC of Louis Stokes, VA,

Research Center, Cleveland, OH, and complied with the Na-tional Institutes of Health Guide for the Care and Use of Lab-

oratory Animals.

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Chung-Chiun Liu is the Wallace R. Persons Pro-fessor of Sensor Technology and Control and aProfessor of chemical engineering at Case WesternReserve University, Cleveland, OH, where he isalso the Director of the Center for Micro and NanoProcessing. His research areas include chemical andbiological sensors and sensor arrays, applicationsof microfabrication to the development of chemicaland biological microsystems, wireless telemetric

interface technology, and microelectrochemical en-ergy sources, including microfuel cells and printablebatteries. He has authored 190 journal publications and holds 12 U.S. patents.



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Edward OConnor is a Technical Specialist in elec-tronics at the Center for Micro and Nano Processing,Case Western Reserve University, Cleveland, OH, apost which he has held for the past 31 years. He hasbeen working in the areas of biomedical electronicsand radio frequency telemetry. He has attendedthe Case Institute of Technology where he studiedelectrical engineering and has published his work in

biotelemetry. He holds four U.S. patents.

Kingman P. Strohl received the B.S. in anthro-pology from Yale University, New Haven, CT, in1970, the M.D. degree from Northwestern Uni-versity, Evanston, IL, in 1974, and he completedhis training in internal medicine at the Universityof Kentucky, Lexington, in 1977, and a researchfellowship in respiratory physiology and pulmonarymedicine at Peter Bent Brigham Hospital, Harvard

School of Health, Cambridge, MA, in 1980.Since 1980, he has been with Case Western Re-serve University, Cleveland, OH, and is now a Pro-

fessor of medicine, anatomy, and oncology at the School of Medicine. Thethemes advanced in funded research over this period of time include the me-chanical properties of the upper airway (1981 to 1988), biomarkers of hypoxia(1988 to 1997), and generic features of breathing and sleep (1997 to present),all relevant to clinical disorders of sleep apnea.

A Multi Channel, Wireless Telemetric Micro System

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