542 IEEE ANTENNAS AND WIRELESS PROPAGATION LETTERS, …

542 IEEE ANTENNAS AND WIRELESS PROPAGATION LETTERS, VOL. 9, 2010

Circularly Polarized Stacked Annular-RingMicrostrip Antenna With Integrated Feeding

Network for UHF RFID ReadersXi Chen, Guang Fu, Shu-Xi Gong, Ya-Li Yan, and Wei Zhao

Abstract—A novel circularly polarized (CP) annular-ring mi-crostrip antenna (ARMSA) is proposed for American ultrahighfrequency (UHF) radio frequency identification (RFID) readers.The annular ring is fed by an equal-split power divider with 90phase difference. For the compactness of the entire antenna, thedivider with the shape of Archimedes gradual-change lines isdesigned at the inside of the annular ring. A circular parasiticpatch is suspended above the ring to improve the impedancematching and bandwidth. It also enhances the performances ofaxial ratio (AR). The impedance bandwidth for dBand 3-dB AR bandwidth are 870–967 MHz (10.6% at 915 MHz)and 893–948 MHz (6%), respectively. The CP gain is 8.9 dBiat 915 MHz with the front-to-back ratio (F/B) of 16.5 dB. Theproposed antenna is low profile as the total height is (

is the free-space wavelength).

Index Terms—Annular-ring, axial ratio (AR), circularly polar-ized (CP), feeding network, microstrip antenna.

I. INTRODUCTION

C IRCULARLY polarized (CP) microstrip antenna hasbeen widely used in satellite communication, global

positioning system (GPS), and radio frequency identification(RFID). Rectangular and circular patch microstrip antennasare traditionally applied to implement circular polarization [1],[2]. In recent years, annular-ring microstrip antenna (ARMSA)is becoming another available candidate. It can be achievedby cutting a concentric circular slot on a circular radiatingpatch. At a fixed operating frequency, an annular-ring patch canmaintain similar radiation characters with a smaller dimensioncompared to that of a circular patch; it is mainly because thebehavior of slotting on the patch extends the flowing path of thesurface current [3]. The CP operation of the annular ring alsoinvolves introducing perturbation segments [4], [5] or addingCP feeding network [6], [7]. However, the very large inputimpedance of ARMSA is an obstacle when the annular ringworks in and modes [8].

Some techniques have been reported to solve the problem.In [4], the authors adopt microstrip line as an impedancetransformer to realize impedance matching with 50 , and thetransmission loss caused by the very narrow microstrip line hasto be considered carefully. In [5] and [9], the coupling feedingis applied to improve the impedance matching. The approaches

Manuscript received April 07, 2010; revised May 03, 2010; accepted May 25,2010. Date of publication June 07, 2010; date of current version June 14, 2010.

The authors are with the National Key Laboratory of Antenna and Mi-crowave Technology, Xidian University, Xi’an 710071, China (e-mail:[email protected]; [email protected]; [email protected])

Digital Object Identifier 10.1109/LAWP.2010.2051791

above hardly provide wide impedance and axial ratio (AR)bandwidths because of the single resonant mode. Stacking aparasitic patch above the annular ring can reduce the value ofinput impedance greatly, and the bandwidth can be broadenedobviously due to the excitation of two adjacent resonant modes[7], [10].

In this letter, we propose a novel CP ARMSA. The circularpolarization is produced by an equal-split power divider, whichis reshaped and arranged at the inside of the annular ring forcompactness. A stacked parasitic circular patch is used to im-prove the performances of impedance and AR. The effect ofground plane size is also studied for the consideration of smallexternal dimensions and good antenna performances.

II. ANTENNA DESIGN

The configuration of the proposed antenna is shown inFig. 1(a). It only contains three components: a printed circuitbroad (PCB) with an annular ring and a feeding network etchedon it, a circular parasitic patch, and a square metallic ground.The PCB is a circular substrate with radius mm( at 915 MHz, is the free-space wavelength), thick-ness mm ( ), and relative permittivity .The annular-ring patch is etched on it. The inner radius andouter radius of the ring are determined by the operatingfrequency. When the ring is designed to work in mode,the resonant frequency can be expressed as [5]

(1)

where represents the speed of light in free space, andrepresent respectively the inner and outer radius consideringthe fringing effect, and represents the effective permittivityaround the ring. ( ) can be seen as the mean circumfer-ential length of the annular ring. From (1), there is

(2)

so the mean circumferential length of the ring is approximatelythe guided wavelength in effective permittivity. Some literaturehas reported the uses for the inner space of the ring. [4] ar-ranges the meandered impedance transformer lines at the insideof the ring. [7] integrates a 3-dB branch-line coupler in an an-nular ring of mode, but the mean circumferential lengthof the ring is approximately twice the guided wavelength in ef-fective permittivity. In our application, it is unsuitable to arrangea traditional divider in the annular ring of mode, so a re-shaped equal-split power divider is designed and etched at theinside of the ring. To obtain the reduction of the divider size and

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CHEN et al.: CP STACKED ANNULAR-RING MICROSTRIP ANTENNA WITH INTEGRATED FEEDING NETWORK FOR UHF RFID READERS 543

Fig. 1. Configurations of the proposed antenna. (a) Full view. (b) Local view. mm, mm, mm ( ), mm( ), mm, mm, mm, mm,

mm, mm, , and mm ( ).

the smoothness of the branch lines, Archimedes gradual-changelines are used to construct the branch lines of the divider.

Fig. 1(b) shows the local view of the feeding network andthe annular ring. The principle of the network is identical withthat of the traditional divider except the shapes. The length ofBranch 1 is approximately and is controlled by and ,where is the guided wavelength; the length of Branch 2 is ap-proximately and is controlled by and . The segmentadjusting 90 -phase shift is controlled by its corresponding cen-tral angle . The widths and of the microstrip lines deter-mine their characteristic impedances of 70 and 50 , respec-tively. The final designing sizes are listed below Fig. 1. As iswell known, introducing CP feeding network can provide widerbandwidths of impedance and AR, but a large insertion loss willlead to low radiation efficiency [6]. In this application, the iso-lation resistor is omitted, so the insertion loss of the network isvery small and the radiation characters are well maintained.

To improve the impedance matching and bandwidth of theannular ring, we suspended a 0.8-mm-thickn circular parasiticpatch above the PCB at the height . When the diameter

Fig. 2. and AR of the proposed antenna with three groups of and .(a) . (b) AR.

of the patch is about a half-wavelength in free space, a goodradiation feature can be obtained. Fig. 2 shows the and ARcurves of the proposed antenna with the different groups ofand . It indicates the antenna has the feature of double resonantmodes, and as decreases and increases, the bandwidths ofimpedance and AR are both broadened. When mm and

mm, the AR values are lowest and its bandwidth isnarrowest; when mm and mm, the ARvalues are partially over 3 dB. A good compromise betweenlower AR values and wider AR bandwidth can be obtained by

mm and mm. The obvious effect comes fromthe strong electromagnetic (EM) mutual coupling between theARMSA and the parasitic patch. According to the theoreticalanalysis in [7], the mutual coupling can be express as

(3)

(4)

(5)

where is a coupling coefficient of the two resonant modes,and and are self-inductances of the ARMSA and the par-asitic patch, respectively. It can be explained that as the para-


Fig. 3. Simulated CP gain and F/B versus ground side length .

sitic patch approaches the annular ring, the enhancement of mu-tual coupling will broaden the bandwidth . However, thetarget values of the concerned electrical performances also haveto be considered.

The square metallic ground is attached to the bottom sur-face of the substrate. The ground size is studied for a suitablechoice. Fig. 3 shows the curves of the simulated CP gain and thefront-to-back ratio (F/B) versus the ground side length from180 ( ) to 320 mm ( ) by 20-mm steps. The increaseof is beneficial to both the gain and the F/B. The gain risesflatly and uniformly, while the F/B rises rapidly before 240 mmand tends to be slow after that. Therefore, for better radiationperformances and smaller total antenna dimensions, it is suit-able to decide in the range of 200–240 mm. Fig. 4 showsthe simulated and AR curves of the proposed antenna withdifferent . The impedance bandwidth for dB isalmost unchangeable, and the 3-dB AR bandwidth is also main-tained well except at 180 mm. In practice, we choose 220 mm( ) as the designing size for calculations and experiments.Of course, a smaller ground size can be chosen for miniatureones if the adjustment of other parameters is provided.

III. EXPERIMENTAL RESULTS AND DISCUSSION

To confirm the theoretical results, we fabricated prototypeswith mm, mm, and mm.The total external dimensions are 14 220 220 mm( ), so a low profile is still main-tained compared to some RFID reader antennas [11]. Fig. 5shows the measured compared to simulated values.The measured impedance bandwidth for dB is870–967 MHz (10.6% at 915 MHz), and it agrees well with thesimulated one. The difference between the measured values andthe simulated ones is mainly due to the manufacture tolerances,e.g., the state of the soldering can cause a few changes to theinput impedance.

Fig. 6 shows the measured CP gain and AR of the pro-posed antenna compared to the simulated results at broadsidedirection. The measured values of AR, CP gain, and the radi-ation patterns (RP) are all obtained through phase-amplitudemethod. From the figure, the measured 3-dB AR bandwidthis 893–948 MHz (6% at 915 MHz), and the values slightly

Fig. 4. Simulated and AR of the proposed antenna with different groundsizes. (a) . (b) AR.

Fig. 5. Measured and simulated of the proposed antenna.

shift toward higher frequency compared to the simulated one.The CP gain in this band also agrees with the simulated valuesdespite some disturbances, which come from the measuringerrors. The gain of 8.9 dBi is obtained at 915 MHz, and thegain is more than 8.5 dBi from 893 to 948 MHz. Therefore,

CHEN et al.: CP STACKED ANNULAR-RING MICROSTRIP ANTENNA WITH INTEGRATED FEEDING NETWORK FOR UHF RFID READERS 545

Fig. 6. Measured and simulated AR and CP gain.Simulated AR, mea-sured AR, Simulated CP gain, measured CP gain.

Fig. 7. Measured RP at 915 MHz. (a) At plane. (b) At plane.LHCP pattern; RHCP pattern.

the proposed antenna has better radiation performances andsmaller ground electrical sizes than those of recent applications[6], [9], and [11].

The measured left-hand circular polarization (LHCP) andright-hand circular polarization (RHCP) RP at 915 MHz areshown in Fig. 7. The half-power beamwidth (HPBW) is 65and 67 , respectively, in plane and plane. Obviously, a

good symmetrical radiation behavior is acquired; it is mainlybecause the feeding network is integrated in the ring and doesnot destroy the symmetry of the entire antenna. Due to the suit-able ground size, the F/B reaches 16.5 dB. It is predicted thatthe F/B will be enhanced further if the ground size increases.

IV. CONCLUSION

A novel CP stacked annular-ring microstrip antenna is pro-posed in this letter for UHF RFID application. The antenna con-tains an equal-split power divider, which is reshaped and ar-ranged at the inside of the ring for compactness of the entire an-tenna. The parasitic patch above the annular ring can improvethe bandwidths of impedance and AR obviously because twoadjacent resonant modes are caused simultaneously. Further-more, through experimental and theoretical analysis, we findthe parameters of the parasitic patch have a sensitive effect onthe values and bandwidth of AR. The experimental results in-dicate the impedance bandwidth for and 3 dB ARbandwidth are 870–967 (10.6%) and 893–948 MHz (6%), re-spectively. The overlapped bandwidth can well cover AmericanUHF RFID band and others. High CP gain of 8.9 dBi and goodF/B of 16.5 dB is acquired respectively at 915 MHz due to theoptimized ground size.

ACKNOWLEDGMENT

The authors would like to thank Mr. J.-L. Zhang for antennafabrication, and the reviewers for helpful suggestions.

REFERENCES

[1] Nasimuddin, K. P. Esselle, and A. K. Verma, “Wideband circularly po-larized stacked microstrip antennas,” IEEE Antennas Wireless Propag.Lett., vol. 6, pp. 21–24, 2007.

[2] F. Ferrero, C. Luxey, G. Jacquemod, and R. Staraj, “Dual-band circu-larly polarized microstrip antenna for satellite applications,” IEEE An-tennas Wireless Propag. Lett., vol. 4, pp. 13–15, 2005.

[3] Y. S. Wu and F. J. Rosenbaum, “Mode chart for microstrip ring res-onators,” IEEE Trans. Microw. Theory Tech., vol. MTT-21, no. 7, pp.487–489, Jul. 1973.

[4] H. M. Chen and K. L. Wong, “On the circular polarization operationof annular-ring microstrip antennas,” IEEE Trans. Antennas Propag.,vol. 47, no. 8, pp. 1289–1292, Aug. 1999.

[5] Y. F. Lin, H. M. Chen, and S. C. Lin, “A new coupling mechanism forcircularly polarized annular-ring patch antenna,” IEEE Trans. AntennasPropag., vol. 56, no. 1, pp. 11–16, Jan. 2008.

[6] Y. X. Guo, L. Bian, and X. Q. Shi, “Broadband circularly polarizedannular-ring microstrip antenna,” IEEE Trans. Antennas Propag., vol.57, no. 8, pp. 2474–2477, Aug. 2009.

[7] H. Ohmine, Y. Sunahara, and M. Matsunaga, “An annular-ring mi-crostrip antenna fed by a co-planar feed circuit for mobile satellitecommunication use,” IEEE Trans. Antennas Propag., vol. 45, no. 6,pp. 1001–1008, Jun. 1997.

[8] W. C. Chew, “A broad-band annular-ring microstrip antenna,” IEEETrans. Antennas Propag., vol. AP-30, no. 5, pp. 918–922, May 1982.

[9] K. F. Tong and J. J. Huang, “New proximity coupled feeding methodfor reconfigurable circularly polarized microstrip ring antennas,” IEEETrans. Antennas Propag., vol. 56, no. 7, pp. 1860–1866, Jul. 2008.

[10] S. H. Al-Charchafchi, W. K. Wan Ali, and S. Sinkeree, “A stackedannular-ring microstrip patch antenna,” in IEEE APS Int. Symp. Dig.,1997, vol. 2, pp. 948–951.

[11] Z. N. Chen, X. M. Qing, and H. L. Chung, “A universal UHF RFIDreader antenna,” IEEE Trans. Microw. Theory Tech., vol. 57, no. 5, pp.1275–1282, May 2009.

IEEE ANTENNAS AND WIRELESS PROPAGATION LETTERS, VOL. 10, 2011 49

Dual-Band Circularly Polarized StackedAnnular-Ring Patch Antenna for GPS Application

Xiaoye Sun, Zhijun Zhang, Senior Member, IEEE, and Zhenghe Feng, Senior Member, IEEE

Abstract—A dual-band circularly polarized stacked an-nular-ring patch antenna is presented in this letter. This antennaoperates at both the GPS L1 frequency of 1575 MHz and L2frequency of 1227 MHz, whose frequency ratio is about 1.28.The proposed antenna is formed by two concentric annular-ringpatches that are placed on opposite sides of a substrate. Wideaxial-ratio bandwidths (larger than 2%), determined by 3-dBaxial ratio, are achieved at both bands. The measured gains at1227 and 1575 MHz are about 6 and 7 dBi, respectively, with theloss of substrate taken into consideration. Both simulated andmeasured results are presented. The method of varying frequencyratio is also discussed.

Index Terms—Annular-ring patch, circular polarization, dual-band, microstrip antennas.

I. INTRODUCTION

W ITH THE rapid development of the satellite com-munication and positioning system, more and more

attention has been paid to circularly polarized (CP) antennas.The Global Positioning Satellite (GPS) system operates at 1575and/or 1227 MHz with RHCP signals. To meet additionallyimportant and demanding applications, antennas operating atboth frequencies are needed.

Various dual-band circularly polarized patch antennas, whichadopt the configuration of single-layer and single-feed, havebeen proposed in [1]–[5]. In [1], two pairs of arc-shaped slots,with different subtending angles, are placed at the edge of thecircular patch. Perturbation, caused by the asymmetric slots,achieves the performance of circular polarization. Two orthog-onal and modes are excited at different bands. Amethod similar to [1] is used in [2], which proposes a squarepatch, with T-shaped and Y-shaped slots, operating at the reso-nant frequencies of and modes. A low-frequency-ratio (about 1.21) CP antenna with a single layer and singlefeed is reported in [3]. Having an unequal lateral cross-slot as

Manuscript received December 06, 2010; revised December 30, 2010;accepted January 18, 2011. Date of publication January 31, 2011; date ofcurrent version March 14, 2011. This work was supported in part by theNational Basic Research Program of China under Contract 2009CB320205,the National High Technology Research and Development Program of China(863 Program) under Contract 2007AA01Z284, the National Natural ScienceFoundation of China under Contract 60771009, and the National Science andTechnology Major Project of the Ministry of Science and Technology of China2010ZX03007-001-01.

The authors are with the State Key Laboratory of Microwave and Communi-cations, Tsinghua National Laboratory for Information Science and Technology,Tsinghua University, Beijing 10084, China (e-mail: [email protected];[email protected]; [email protected]).

Color versions of one or more of the figures in this letter are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/LAWP.2011.2109365

a perturbation structure on the ground plane, it has a bidirec-tional radiation. A slot-coupled square patch antenna with anS-shaped slot on the patch [4] operated at the two GPS frequen-cies, whose circular polarization bandwidth is about 0.6% at thehigh frequency, is presented. In all these designs, a perturbationmethod is used to achieve circular polarization, and the relativeaxial-ratio (AR) bandwidth is normally around 1% or less.

In [6]–[11], stacked patch antennas with dual-resonant unitsare presented and measured. With three layers of substrates anddual feed [6], the stacked antenna operating at both the L1 andL2 frequencies of GPS are excited by a cross slot, which isfed by a two-stage Wilkinson power divider network. In [7],with a dual-layer structure, two stacked corner-truncated squarepatches are excited by a single probe. In the design of [7] and [8],the reduction of substrate is achieved by a single feed, whosecost is the decrease of axial-ratio bandwidths and gains. In thesethree kinds of stacked structures[6]–[8], strong coupling existsbetween the two patches since the bottom patch acts as a groundplane when the top patch resonates.

In [12], the introduction of an annular-ring patch can reducethe dimension of the antenna compared to a circular patch withthe same outer radius. Similarly, perturbation on an annular ringalso makes the relative axial-ratio bandwidth less than 1%.

In this letter, a dual-band, dual-layer, dual-feed stacked an-nular-ring patch antenna excited by two orthogonal H-shapedslots, fed by a 3-dB hybrid, with axial-ratio bandwidths greaterthan 2% at L1 and L2 frequencies of GPS and slight couplingbetween the patches is proposed and measured. With the lossof substrate taken into consideration, the measured gains at L1and L2 are about 7 and 6 dBi, respectively. In addition, the fre-quency ratio of the two bands can be varied from 1.28 to 1.83by adjusting the radius of the patch.

II. ANTENNA DESIGN AND STRUCTURE

Fig. 1 shows the geometry and dimensions of the proposedstacked annular-ring patch antenna, which is fabricated with twolayers of square FR4 mmsubstrates, and the substrates are separated by an air layer withheight mm. This antenna is assembled with four nyloncolumns and eight nylon screws. Two concentric annular-ringpatches, one smaller and the other one larger, are on the top andbottom of the upper substrate, respectively. The inside radiusand outside radius of the top ring are mm and

mm, respectively. The inside radius and outside radius ofthe bottom ring are mm and mm, respec-tively. The dimension of the patches is only about 0.87 times ofthat in [1].

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Fig. 1. Geometry of the proposed antenna.

With the loss of the substrate taken into consideration, a thinsubstrate should be adopted. Using an annular-ring can achievean antenna within relatively small dimensions and at the sametime decrease the coupling between the two patches. To furthercontrol the coupling, a gap of 2 mm between the two rings isrequired, namely the should be larger than 2 mm. In addition,since the frequencies of L1 and L2 are quite close, the dimen-sions of the rings have to be nearly the same, which might in-crease the coupling between the two patches. Thus, the largerring for low frequency has to be printed on the bottom of theupper substrate in order to enlarge its dimensions and at thesame time shrink the radius of the top ring.

The top layer of the lower substrate is the ground. Dual feed isadopted to achieve wide axial-ratio bandwidths. Two orthogonalH-shaped apertures were etched on the ground. A 3-dB hybrid,which functions as the feeding network, is on the bottom of thelower substrate. The positions and dimensions of the aperturesand 3-dB hybrid are listed in Table I, respectively. The feedingport is shown in Fig. 1. As the 3-dB hybrid has two ports, the

TABLE IDIMENSIONS AND POSITION OF APERTURE AND 3-dB HYBRID

ON THE LOWER SUBSTRATE

Fig. 2. Simulation results of frequencies of smaller and larger patches with thechanging of .

antenna can support both LHCP and RHCP. When measuringthe antenna in the RHCP mode, the antenna is fed from one port,and a 50- load is attached to the other one.

Because the two frequencies of the dual-band antenna are rel-atively close, a single-band 3-dB hybrid is used. The center fre-quency of the 3-dB hybrid is 1.4 GHz. By adjusting the param-eter of the 3-dB hybrid, it can operate at both the L1 and L2frequencies with an equal power at the output ports, with phaseshift of about 90 between them and lower than 13 dB.

III. PARAMETRIC STUDY

With mm, mm, and mmfixed, Fig. 2 shows the simulated results (using HFSS) of highresonant frequency and low resonant frequency of the twopatches by varying from 0.4 to 17.6 mm, without the 3-dBhybrid (two ports assigned at the two feeding lines with equalamplitude and 90 phase shift). When is larger than 2 mm,with increased, the inner and outer radii of the large patchincrease, which makes the low frequency decrease. When isless than 2 mm, strong coupling exists between the two patches,which makes the high frequency lower and low frequencydecrease even more rapidly. A probable explanation for this isthat the adjacent patches might act as a parallel capacitor toeach other, which enlarges their equivalent electrical length.This explication is also suitable for the slight variation of highfrequency when is larger than 4 mm since the increasinggap between patches gradually decreases the coupling betweenthem.

Coupling between the two patches affects the adjustment ofthe frequency ratio. Fig. 3 shows the simulated results of thefrequency ratio of the two resonant frequencies under the samecondition as that in Fig. 2. When is larger than 2 mm, thefrequency ratio monotonically increases with the increasing of

. In this proposed structure, with larger than 2 mm, variationof the frequency ratio from 1.28 to 1.83 can be easily achieved

SUN et al.: DUAL-BAND CIRCULARLY POLARIZED STACKED ANNULAR-RING PATCH ANTENNA FOR GPS APPLICATION 51

Fig. 3. Simulation results of frequency ratio with the changing of .

Fig. 4. Photograph of the fabricated antenna.

Fig. 5. Simulated and measured .

by only adjusting the gap between patches. When is less than2 mm, strong coupling exists between the two patches, whichmakes the frequency ratio increase and difficult to control.

IV. SIMULATED AND MEASURED RESULTS

The proposed antenna is fabricated and measured. Fig. 4 isa photograph of the assembled antenna. Fig. 5 shows the sim-ulated and measured with good consistency between them.Although good matching is observed from 1227 to 1575 MHz,not all of them are due to good antenna matching. Because a3-dB hybrid is used, the 50- load can absorb all reflected powereven when the antenna works at mismatched frequency points.The AR of right-hand circularly polarized (RHCP) radiation at1227 and 1575 MHz is 0.7 and 1.9 dB, respectively. Simulationand measurement of AR and gain at these two bands are pre-sented in Fig. 6, in which relative axial-ratio bandwidths deter-mined by 3-dB axial ratio are larger than 2%. About 6- and 7-dBi

Fig. 6. Simulated and measured axial ratio and gain.

Fig. 7. Measured radiation pattern. (a) 1227 MHz at plane. (b) 1227 MHzat plane. (c) 1575 MHz at plane. (d) 1575 MHz at plane.

gains at the broadside of the antenna are illustrated at 1227 and1575 MHz, respectively. Normalized radiation pattern at 1227and 1575 MHz at plane and planes are plotted in Fig. 7,respectively.

V. CONCLUSION

A novel dual-band antenna structure of stacked annular-ringpatches operated at GPS L1 and L2 frequencies, whose fre-quency ratio is 1.28, is presented in this letter. This antenna isexcited by two orthogonal H-shaped apertures fed by a 3-dBhybrid. Relatively small dimensions are achieved by the appli-cation of the annular ring. It has relative axial-ratio bandwidthslarger than 2% at both bands. Measured gains at the broadsidedirection at L1 and L2 are about 7 and 6 dBi, respectively. Themethod of varying the frequency ratio is also discussed in thisletter.

REFERENCES

[1] G.-B. Hsieh, M.-H. Chen, and K.-L. Wong, “Single feed dual-bandcircularly polarized microstrip antenna,” Electron. Lett., vol. 34, pp.1170–1171, 1998.


[2] K. P. Yang and K. L. Wong, “Dual-band circularly-polarized squaremicrostrip antenna,” IEEE Trans. Antennas Propag., vol. 49, no. 3, pp.377–382, Mar. 2001.

[3] X. L. Bao and M. J. Ammann, “Dual-frequency circularly-polarizedpatch antenna with compact size and small frequency ratio,” IEEETrans. Antennas Propag., vol. 55, no. 7, pp. 2104–2107, Jul. 2007.

[4] Nasimuddin, Z. N. Chen, and X. M. Qing, “Dual-band circularly polar-ized S-shaped slotted patch antenna with a small frequency-ratio,” IEEETrans. Antennas Propag., vol. 58, no. 6, pp. 2112–2115, Jun. 2010.

[5] F. Yang and Y. Rahmat-Samii, “Switchable dual-band circularly po-larised patch antenna with single feed,” Electron. Lett., vol. 37, pp.1002–1003, 2001.

[6] D. M. Pozar and S. M. Duffy, “A dual-band circularly polarized aper-ture-coupled stacked microstrip antenna for global positioning satel-lite,” IEEE Trans. Antennas Propag., vol. 45, no. 11, pp. 1618–1625,Nov. 1997.

[7] C.-M. Su and K.-L. Wong, “A dual-band GPS microstrip antenna,” Mi-crow. Opt. Technol. Lett., vol. 33, pp. 238–240, 2002.

[8] J.-Y. Jan and K.-L. Wong, “A dual-band circularly polarized stackedelliptic microstrip antenna,” Microw. Opt. Technol. Lett., vol. 24, pp.354–357, 2000.

[9] G. Z. Rafi, M. Mohajer, A. Malarky, P. Mousavi, and S. Safavi-Naeini, “Low-profile integrated microstrip antenna for GPS-DSRCapplication,” IEEE Antennas Wireless Propag. Lett., vol. 8, pp.44–48, 2009.

[10] J.-H. Oh, Y.-P. Hong, and J.-G. Yook, “Dual circularly-polarizedstacked patch antenna for GPS/SDMB,” in Proc. Int. Conf. Microw.Millim. Wave Technol., 2010, pp. 252–255.

[11] K. Geary, J. H. Schaffner, H. P. Hsu, H. J. Song, J. S. Colburn, andE. Yasan, “Single-feed dual-band stacked patch antenna for orthogonalcircularly polarized GPS and SDARS applications,” in Proc. IEEE Veh.Technol. Conf., 2008, pp. 1–5.

[12] H.-M. Chen and K.-L. Wong, “On the circular polarization operationof annular-ring microstrip antennas,” IEEE Trans. Antennas Propag.,vol. 47, no. 8, pp. 1289–1292, Aug. 1999.

Progress In Electromagnetics Research C, Vol. 12, 65–77, 2010

PARAMETRIC STUDIES ON THE CIRCULARLY PO-LARIZED STACKED ANNULAR-RING MICROSTRIPANTENNA

X. Chen, G. Fu, S. X. Gong, Y. L. Yan, and J. Chen

National Key Lab. of Antenna and Microwave TechnologyXidian UniversityXi’an 710071, China

Abstract—In this paper, we propose a circularly polarized (CP)stacked annular-ring microstrip antenna (SARMSA) with an integratedfeeding network in the UHF RFID band. A circular parasitic patch issuspended above the annular ring to improve the impedance matchingand bandwidth. Through the parametric studies on SARMSA, the CPcharacters of the entire antenna are well understood, and an optimizedCP character is obtained. Prototypes are fabricated to confirm thetheoretical results. The experimental results indicate the impedancebandwidth for S11 < −10 dB is 870–967 MHz (10.6% at 915MHz),and the 3 dB AR bandwidth is 893–948 MHz (6%). Meanwhile, themeasured CP gain reaches 8.9 dBic at 915 MHz.

1. INTRODUCTION

Rectangular and circular patch microstrip antennas are traditionallyused to implement circular polarization [1, 2]. In recent years, annular-ring microstrip antenna (ARMSA) is another available structure torealize circular polarization [3–11]. It can be achieved by cutting aconcentric circular slot on the circular radiating patch. At a fixedoperating frequency, an annular-ring patch can realize more miniaturedimensions than that of a circular patch, which is because the behaviorof slotting on the patch extends the path of current flowing [12]. TheCP operation of the annular ring also involves introducing perturbationsegments [3, 4] or adding feeding network [10, 11]. However, the verylarge input impedance of ARMSA is an obstacle when an annularring works in TM11 and TM21 modes [13]. In the literatures, some

Corresponding author: X. Chen ([email protected]).

66 Chen et al.

techniques have been proposed to solve this problem. In [3], the authorsuse microstrip line as impedance transformer to realize matching withthe characteristic impedance of 50 Ω. In [4–8], the technique of couplingfeeding is used to improve the impedance matching. Loading gaps,slits, or stubs on the ring is another method to be involved [9], but thepatch mainly works at linear polarization, and polarization rotationcan be caused by the loading. However, the above approaches hardlyprovide wide impedance bandwidth because of the single resonantfrequency. Stacking parasitic patch above the annular ring can greatlyreduce the value of input impedance [10, 11], and the bandwidth can bebroadened obviously due to exciting two adjacent resonant frequencies.But in [10], the authors did not provide the relative parametric studieson the antenna performance, especially, the CP performance when theantenna works in circular polarization.

In this paper, we design a CP stacked annular-ring microstripantenna (SARMSA) working in TM11 mode. The CP character isimplemented by an integrated power divider. The parameters of theARMSA are studied in detail since they have a significant effect on theperformance of the proposed antenna. The experimental results areprovided to verify the correctness of the analysis.

2. ANTENNA CONFIGURATION

The configuration of the proposed antenna is shown in Fig. 1(a). Themain components involve a circular parasitic patch, a printed circuitbroad (PCB) with an annular ring and a feeding network on it, anda square metallic ground. The annular ring with inner radius a andouter radius b is etched on a circular substrate with radius Rs = 55 mm(0.168λ at 915 MHz) and thickness ts = 1 mm (0.003λ). For theconsideration of low cost, PTFE with relative permittivity εr = 2.65is selected as the substrate material. At the inner of the ring, areshaped equal-split power divider is also etched on the top surfaceof the substrate. For the size reduction of the feeding network and thesmoothness of the microstrip lines, Archimedes gradual-change linesare used to construct branch lines of the divider. Fig. 1(b) is the localview of the feeding network. The length of the branch 1 is about λg/4and controlled by r1 and r2, where λg is the guided wavelength, whilethe length of the branch 2 is about λg/2 and controlled by r3 and r4.The segment adjusting 90 phase shift is controlled by its correspondingcentral angle θ. The widths w1 and w2 of the microstrip lines determinetheir characteristic impedances of 70 Ω and 50Ω respectively. In thisapplication, the isolation resistor is omitted, so the insert loss of thenetwork is small. To improve the impedance matching and bandwidth

Progress In Electromagnetics Research C, Vol. 12, 2010 67

of the annular ring, we suspend a 0.8 mm-thickness circular parasiticpatch above the PCB at the height h. When the diameter of the patchis about a half wavelength in free space, a good radiation feature canbe obtained. The square metallic ground is attached to the bottomsurface of the substrate, and its size has an obvious effect on the gainand the ratio of front to back (F/B) but less effect on the impedanceand AR bandwidths. After adjustment and optimization based onthe initial sizes, the designed parameters are fixed as a = 25.8mm,b = 42.6mm, dr = 167.9 mm (0.51λ), h = 10 mm (0.03λ), r1 = 17 mm,r2 = 22mm, r3 = 17 mm, r4 = 25mm, w1 = 1.3mm, w2 = 2.4mm,θ = 120, and Lg = 220 (0.67λ).

Figure 2 shows the power distribution ratio and phase differenceof the designed feeding network when the output ports connect to50Ω. It is observed that a good CP character is provided. Since thecharacter of power divider is widely known, its parametric studies arenot performed here. In the following sections, the parameters of theSARMSA are studied in detail for understanding the performance ofthe entire antenna. The commercial software HFSS is used to guidethe studies.

Lg

Annular ring Feeding

network

Feed point

Parasitic patch Substrate (1 mm)Metallic

ground

Top view

Side viewFeed point

dr

h

z

x y

z

x

y

a

b

w1

w2 r 2r1

r 3

r 4θ

Feeding

networkAnnular ring

Branch 1

Branch 2

Substrate

(a) (b)

Figure 1. Configurations of the proposed antenna. (a) Full view. (b)Local view.

68 Chen et al.

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Figure 2. Power distribution ratio and phase difference of theproposed feeding network at output ports.

3. PARAMETRIC STUDIES

3.1. ARMSA

ARMSA is the foundation of the proposed antenna. The characterof ARMSA has been studied in theory and practice [13–16]. Inpractical applications, its resonant frequency, input impedance, andimpedance bandwidth are the keys to be concerned. In general, whenthe annular-ring patch works in TM11 mode, the resonant frequencycan be expressed as [4]

f11 =c

π(a + b)√εeff(1)

where c is the speed of light in free space, π(a + b) the meancircumferential length of the annular-ring patch, and εeff the effectivepermittivity. Obviously, when the material and sizes of the substrateiare fixed, the resonant frequency is determined by a and b. Figs. 3(a)and (b) show the simulated resonant frequency and input resistanceversus the ratio of a to b (a/b) with two different substrate thicknesses,where the outer radius b is fixed at 42.6 mm. It illustrates, at a fixedfrequency, the thinner substrate has lower resonant frequency andsmaller input impedance than that of the thicker one. In addition,Fig. 3(b) also indicates that as the width of the annular ring increases,i.e., a/b decreases, the input impedance drops dramatically. Accordingto this rule, we can change a and b jointly to adjust the value of inputimpedance greatly at a fixed frequency. The precise sizes of a and bare decided by the value of input impedance and its bandwidth afterstacking the parasitic patch. We select a = 25.8 mm as designingparameter, i.e., a/b = 0.6. The resonant frequency and the inputimpedance of the ARMSA are recorded in Table 1.


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Input

resi

stan

ce (

ohm

)

Figure 3. Simulated characters on ARMSA versus a/b with differentsubstrate thickness. —¥— 1 mm, —¤— 2 mm. (a) Resonantfrequency. (b) Input impedance.

Table 1. Comparison of resonant characters between ARMSA andSARMSA.

````````````TypesCharacters

f/MHz Rin/Ω

ARMSA 910 1114SARMSA 892 948 68 60

3.2. SARMSA

Stacking a parasitic patch above the annular ring has been used toimprove the impedance matching [10, 11]. However, we have notseen the relative studies on the relationships between the parametersof the parasitic patch and the characters of SARMSA, especially,the impedance matching, impedance bandwidth, AR value, and ARbandwidth when the antenna works in circular polarization by afeeding network. Therefore, to illustrate the relationships between theSARMSA and entire antenna, we present the parametric studies of theparasitic patch in detail below. To accurately understand the effect ofthese critical parameters on the input impedance and AR performance,the sizes of the annular ring and the feeding network are fixed as thosementioned in Section 2.

(1) The diameter of the parasitic patch — dr.Figure 4 shows the impedance loci of the SARMSA with the

parasitic patch of different diameters dr, as well as no parasitic patch.First, it indicates that after stacking parasitic patch, the impedance

70 Chen et al.

0.2 0.5 1.0 2.0 5.0

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938MHz

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0.2j

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0.5j

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1.0j

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2.0j

-5.0j

5.0j

Figure 4. Impedance loci of ARMSA with various dr. —¤—dr = 165 mm, — dr = 167.9mm, —¥— dr = 171 mm, —×— noparasitic patch.

850 860 870 880 890 900 910 920 930 940 950 960 970 980 990

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(d

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11

Frequency (MHz) Frequency (GHz)

dr=165mm

dr=167.9mm

dr=171mm

dr=165 mm

dr=167 .9mm

dr=171 mm

Figure 5. S11 and AR on the entire antenna with various dr. (a) S11.(b) AR.

matching is improved remarkably in a wide band, i.e., a loop isformed on impedance loci. When dr = 167.9mm, two adjacentresonant frequencies are excited simultaneously around 915 MHz. Thefrequencies and their input resistances are also recorded in Table 1for comparing. In fact, the middle frequency of the two resonantfrequencies decides the single resonant frequency of the ARMSA,and the values of the input resistance decide that of the ARMSA.Accordingly, the sizes of a and b are determined in this way. Secondly,


as the diameter decreasing, the loop is transformed from capacitivecharacter to inductive one, while the dimension of the loop changeslittle. It implies a stable impedance bandwidth. Meanwhile, themiddle point of the two resonant frequencies is shifted toward lowerfrequency. Figs. 5(a) and (b) show the variations of S11 and AR of theentire antenna respectively for different dr. It is observed that bothof the performances express frequency shift versus dr. The impedancebandwidth for S11 < −10 dB changes slightly as that of the SARMSA,and the AR bandwidth is also maintained. However, the values of S11

and AR degenerate totally or partially. This point is easy to be ignoredin the study of linear polarization.

(2) The height between parasitic patch and annular ring — h.Figure 6 shows the impedance loci of SARMSA for various heights

h between the parasitic patch and annular ring. It is observed thatexcept for a little change on the character of reactance, the impedancebandwidth increases as the height decreases, i.e., the impedance loopbecomes a large one from a point. For the entire antenna, theimpedance and AR bandwidth, shown in Fig. 7, are also enhanced byreducing h, e.g., the impedance bandwidth of h = 8 mm is 15.2% widerthan that of h = 10 mm. It is beneficial for linear polarization. But itsAR values exceed 3 dB partially. In contrary, the case of h = 11 mmhas a good AR value in a smaller bandwidth. It is indicated that agood impedance matching of the SARMSA can cause a lower AR valuebut a narrower AR bandwidth.

0.2 0.5 1.0 2.0 5.0

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0.5j

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1.0j

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2.0j

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5.0j

Figure 6. Impedance loci of ARMSA with various h. —×—h = 8 mm, —¤— h = 9 mm, — h = 10 mm, —¥— h = 11 mm.

72 Chen et al.

0.86 0.88 0.90 0.92 0.94 0.96 0.98 1.00

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11

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l ra

tio (

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)

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8mm

9mm

10 mm

11 mm

8mm

9mm

10 mm

11 mm

Figure 7. S11 and AR on the entire antenna with various h. (a) S11.(b) AR.

0.2 0.5 1.0 2.0 5.0

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0.2j

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0.5j

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1.0j

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2.0j

-5.0j

5.0j

Figure 8. Impedance loci of ARMSA with three groups of h and dr.—¤— h = 9.3mm, dr = 168.6mm, — h = 10 mm, dr = 167.9mm,—¥— h = 11 mm, dr = 166.1mm.

According to the results above, we have a consideration of dr andh in common to obtain different characteristics of the entire antenna.Fig. 8 illustrates three impedance loci of the ARMSA according tothree groups of dr and h. Several equal-VSWR circles are drawn onthe Smith chart for reference. And the corresponding S11 and ARof the entire antenna are shown in Fig. 9. It is observed that whenh = 11 mm and dr = 166.1mm, the impedance loci form a pointin the circle of VSWR< 1.2, but the bandwidth of the ARMSA isnarrow. In this case, the entire antenna has good AR values, whichare less than 1.68 dB from 896 MHz to 938MHz. When h = 9.3mm and


0.86 0.88 0.90 0.92 0.94 0.96 0.98 1.00

-30-28-26-24-22-20-18-16-14-12-10

-8-6-4-20

(a)

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0

3

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(b)

S

(d

B)

11

Axia

l ra

tio (

dB

)

Frequency (GHz) Frequency (MHz)

h=11 mm, dr=166.1mm

h=10 mm, dr=167.9mm

h=9.3 mm ,dr=168.6mm

h=11mm, dr=166. 1mm

h=10 mm, dr=167. 9mm

h=9.3mm, dr=168. 6mm

Figure 9. S11 and AR on the entire antenna with three groups of hand dr. (a) S11. (b) AR.

dr = 168.6mm, the entire antenna forms two 3 dB AR bandwidth apartfrom 45 MHz. Here, the impedance loop of the ARMSA is in the circleof VSWR< 1.6. A good compromise on the AR value and bandwidthcan be obtained when h = 10 mm and dr = 167.9mm, and itsimpedance loop of ARMSA is in the circle of VSWR< 1.4. It is notedfrom the Fig. 9(a) that the impedance bandwidth for S11 < −10 dB hasa small change with the three groups’ sizes. The experimental resultsare mainly attributed to the condition of impedance matching betweenthe ARMSA and feeding network. Due to no isolation resistor in thefeeding network, the equal-power distribution is satisfied well in thecase of good impedance matching between the ARMSA and feedingnetwork. But for SARMSA, as shown in Fig. 8, the values of inputimpedance and its bandwidth are contradictive. Therefore, lower ARvalues and wider AR bandwidth cannot be obtained together, but wecan obtain the ARMSA with high CP purity or dual-frequency CPcharacter respectively in terms of demands.

4. EXPERIMENTAL RESULTS AND DISCUSSIONS

According to the conclusions of the parametric studies, we fabricateprototypes with h = 10mm and dr = 167.9mm for a balancedconsideration. Their photos are shown in Fig. 10. The measuredS11, compared with simulated values, is shown in Fig. 11. Themeasurements are implemented in an Aglient network analyzerE8363B. As shown in the figure, the measured impedance bandwidthfor S11 < −10 dB is 870–967MHz (10.6% at 915 MHz), and it agreeswell with the simulated one despite of a little higher S11 values. Itis mainly due to the manufacture tolerances, e.g., the state of thesoldering causes the change of the input impedance.

74 Chen et al.

Figure 10. Photos of the proposed antenna.

0.86 0.88 0.90 0.92 0.94 0.96 0.98 1.00

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S

(d

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Frequency (GHz)

simulated

Figure 11. Measured andsimulated S11.

870 880 890 900 910 920 930 940 950 960 970

0

3

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4

5

6

7

8

9

10A

xia

l ra

tio (

dB

)

CP

gai

n (

dB

ic)

Frequency (MHz)

Figure 12. Measured andsimulated AR and CP gain. —simulated AR, ¥ measured AR,− − − simulated CP gain, Nmeasured CP gain.

Figure 12 shows the measured CP gain and AR of the proposedantenna compared with the simulated results at broadside direction.The measured values of AR and CP gain are both derived from the twoorthogonal linearly polarized vectors measured by a standard gain hornantenna [17]. It is observed that the measured 3 dB AR bandwidthis 893–948 MHz (6% at 915 MHz), and the values shift toward highfrequency slightly compared with the simulated one. The CP gainin this band also agrees with the simulated values in spite of somedisturbance. It is also observed that the gain of 8.9 dBic is obtainedat 915 dBic, and the gain is more than 8.5dBic from 893 MHz to948MHz. It is noted that the proposed antenna has better radiationfeatures and smaller ground sizes than those in recent literatures [6, 7].Furthermore, if we could sacrifice some radiation capability, i.e., gainand F/B, smaller antenna dimensions could be achieved.


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Figure 13. Simulated CP radiation patterns in x-z plane. —¥—LHCP, —¤— RHCP. (a) At 900MHz, F/B = −17.6 dB, HPBW =65. (b) At 915 MHz, F/B = −17.4 dB, HPBW = 64. (c) At 940 MHz,F/B = −16.9 dB, HPBW = 64.

Due to the limitation of measurements, the measured circularlypolarized radiation patterns (RP) are not acquired. But from theagreement of CP gain between measurement and simulation, it isbelieved that the simulated data have the value of reference. Thesimulated CP RP in x-z plane at 900 MHz, 915 MHz, and 940 MHz areprovided separately in Fig. 13. The F/Bs and the half power beamwidths (HPBW) are listed below the figures.

5. CONCLUSION

A CP stacked annular-ring microstrip antenna with an integratedfeeding network is proposed in this paper. Parametric studies areperformed on the SARMSA to illustrate the CP characters of the

76 Chen et al.

proposed antenna. For the proposed antenna, different CP characterscan be obtained in terms of the condition of the parasitic patch.Prototypes are fabricated to confirm the theoretical studies. Theexperimental results agree well with the simulated ones, and it isindicated that the impedance bandwidth for S11 < −10 dB is 870–967MHz, 10.6% at 915MHz, and the 3 dB AR bandwidth is 893–948MHz, 6% at 915MHz. The overlapped bandwidth can well coverAmerican UHF RFID band, i.e., 902–928MHz. Meanwhile, themeasured CP gain can reach 8.9 dBic at 915 MHz.

REFERENCES

1. Kasabegoudar, V. G. and K. J. Vinoy, “A broadband suspendedmicrostrip antenna for circular polarization,” Progress InElectromagnetics Research, PIER 90, 353–368, 2009.

2. Yang, S. L. S., K. F. Lee, and A. A. Kishk, “Design and studyof wideband single feed circularly polarized microstrip antennas,”Progress In Electromagnetics Research, PIER 80, 45–61, 2008.

3. Chen, H. M. and K. L. Wong, “On the circular polarizationoperation of annular-ring microstrip antennas,” IEEE Trans.Antennas Propag., Vol. 47, No. 8, 1289–1292, 1999.

4. Lin, Y. F., H. M. Chen, and S. C. Lin, “A new coupling mechanismfor circularly polarized annular-ring patch antenna,” IEEE Trans.Antennas Propag., Vol. 56, No. 1, 11–16, 2008.

5. Row, J. S., “Design of aperture-coupled annular-ring microstripantennas for circular polarization,” IEEE Trans. AntennasPropag., Vol. 53, No. 5, 1779–1784, 2005.

6. Tong, K. F. and J. J. Huang, “New proximity coupled feedingmethod for reconfigurable circularly polarized microstrip ringantennas,” IEEE Trans. Antennas Propag., Vol. 56, No. 7, 1860–1866, 2008.

7. Guo, Y. X., L. Bian, and X. Q. Shi, “Broadband circularly po-larized annular-ring microstrip antenna,” IEEE Trans. AntennasPropag., Vol. 57, No. 8, 2474–2477, 2009.

8. Ridgers, G. M., J. W. Odendaal, and J. Joubert, “New feedingmechanism for annular-ring microstrip antenna,” Electron. Lett.,Vol. 36, No. 7, 605–606, 2000.

9. Latif, S. I. and L. Shafai, “Polarization characteristic of multibandloaded microstrip annular ring antennas,” IEEE Trans. AntennasPropag., Vol. 57, No. 7, 2788–2793, 2009.

10. Al-Charchafchi, S. H., W. K. Wan Ali, and S. Sinkeree, “A stacked


annular-ring microstrip patch antenna,” IEEE APS InternationalSymposium Digest, Vol. 2, 948–951, 1997.

11. Ohmine, H., Y. Sunahara, and M. Matsunaga, “An annular-ringmicrostrip antenna fed by a co-planar feed circuit for mobilesatellite communication use,” IEEE Trans. Antennas Propag.,Vol. 45, No. 6, 1001–1008, 1997.

12. Wu, Y. S. and F. J. Rosenbaum, “Mode chart for microstrip ringresonators,” IEEE Trans. Microwave Theory Tech., Vol. 21, 487–489, 1973.

13. Liu, H. and X. F. Hu, “Input impedance analysis of a microstripannular-ring antenna with a thick substrate,” Progress InElectromagnetics Research, PIER 12, 177–204, 1996.

14. Chew, W. C., “A broad-band annular-ring microstrip antenna,”IEEE Trans. Antennas Propag., Vol. 30, No. 5, 918–922, 1982.

15. Lee, K. F. and J. S. Dahele, “Theory and experiment on theannular-ring microstrip antenna,” Annals of Telecommunications,Vol. 40, 508–515, 1985.

16. Dahele, J. S., K. F. Lee, and D. P. Wong, “Dual-frequency stackedannular-ring microstrip antenna,” IEEE Trans. Antennas Propag.,Vol. 35, No. 11, 1281–1285, 1987.

17. Lo, Y. T. and S. W. Lee, Antenna Handbook, Chapman and Hall,New York, NY, 1993.

Progress In Electromagnetics Research Letters, Vol. 23, 99–107, 2011

CIRCULARLY-POLARIZED STACKED ANNULAR-RINGMICROSTRIP ANTENNA

M. Ramırez and J. Parron

Q-Building, Campus of UABUniversitat Autonoma de Barcelona (UAB)Bellaterra (Cerdanyola del Valles), Barcelona 08193, Spain

Abstract—In this paper, we propose the aperture feeding as atechnique for bandwidth enhancement of multi-band microstrip ringantennas. In particular, we present a dual-band stacked annular-ring microstrip antenna fed by four bow-tie apertures with circularpolarization. Furthermore, we show that the size and shape of thesubstrate that supports the radiating elements of the antenna plays animportant role in the quality of the axial ratio.

1. INTRODUCTION

Circularly-polarized microstrip patch antennas are widely used innavigation systems and satellite communications due to their low cost,light weight and ease of fabrication. Circular polarization is usuallyobtained either using a single feeding point and perturbing the regularstructure of the microstrip antenna [1, 2], or using two different feedingpoints with equal amplitudes and 90 phase shift (without perturbingthe regular structure of the antenna) [3–5].

The main drawback of microstrip antennas is the size. Forthis reason, new designs for achieving a reduction in size have beenpresented in recent years [6, 7]. In particular, microstrip ring antennashave become very attractive because for a given frequency, a ringantenna has a smaller patch size as a microstrip square antenna [8–10]. However, ring microstrip antennas suffer limitations in impedanceand axial-ratio bandwidths. The input impedance of ring microstripantennas depends on the width of the rings. As width is reduced, inputimpedance increases and therefore it is more difficult to obtain goodimpedance match. Various techniques have been employed in literaturein order to enhance impedance matching and axial-ratio bandwidths.

Received 22 March 2011, Accepted 21 April 2011, Scheduled 28 April 2011Corresponding author: Monica Ramırez ([email protected]).

100 Ramırez and Parron

In [11, 12] capacitive feeding was used in a multi-band ring stackedconfiguration obtaining narrow bandwidths. In [13] a circular parasiticpatch was suspended above an annular-ring to enhance the impedancematching and bandwidth. The annular-ring was fed by an equal-splitpower divider placed inside the ring. 3-dB axial ratio bandwidth of10.6% was obtained. In [14] circular polarization of a square-ring patchwas achieved by a simple microstrip feeding line through the couplingof a square patch in the same plane of the antenna. A 10-dB returnloss bandwidth of 1.1% and 3-dB axial-ratio bandwidth about 0.03%was achieved. In [15] a proximity coupling technique was used to feed areconfigurable circularly polarized microstrip ring antenna, obtaininga 10-dB return loss bandwidth of 5% and a 3-dB axial ratio bandwidthof 1.32%.

Although ring microstrip antennas with good impedance andaxial-ratio bandwidths can be found in literature, it must be notedthat most of them are single-band.

The aim of this paper is to use aperture feeding as an alternativetechnique to improve the performance (impedance and axial ratiobandwidth) of multi-band ring microstrip antennas. In particular, adual-band circularly-polarized antenna based on stacked annular-ringmicrostrip patches fed by four bow-tie apertures is designed, fabricatedand measured. A broadband feeding network has been used to obtainthe required phase shift thus allowing to improve the axial ratio in thedirection of maximum radiation. This feeding network is based on thebroadband phase shifter presented in [16]. We will also show that thesize and shape of the substrate that supports the radiating elementsof the antenna, i.e., the annular-rings, plays an important part in thequality of the axial ratio for low elevation angles.

2. ANTENNA CONFIGURATION

Our aim is to design a dual-band antenna with right-hand circularpolarization. The lower band must cover Galileo (E5a, E5b), GPS(L5) and GLONASS (L3) bands (bandwidth 50 MHz, 4.2%, at thecenter frequency of 1.189 GHz), whereas the upper band must coverGalileo (E1, E2, L1) and GPS (L1) bands (bandwidth 30MHz, 2%, atthe center frequency of 1.575GHz).

2.1. Antenna Design

Figure 1 shows a detailed drawing of the proposed antenna. Theantenna is composed of three substrate layers separated by air gaps.The parasite annular-ring is in the top layer, the active annular-ring is

Progress In Electromagnetics Research Letters, Vol. 23, 2011 101

Y

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DD

1in 1out

2out2in

Figure 1. Geometry of the stacked annular-ring microstrip antenna.

in the middle layer whereas the feeding network and the four aperturesare placed in the bottom layer. The parasite annular-ring has an innerdiameter D2in and outer diameter D2out, whereas the active annular-ring has an inner diameter D1in and outer diameter D1out.

In the first step of the design the antenna is excited through fourapertures using four microstrip lines, obtaining a dual-band antennawith two orthogonal linearly-polarized modes. The resonant frequencyof the lower band is mainly determined by the parasite annular-ringwhile the resonant frequency of the higher band is mainly determinedby the active annular-ring. The fine tuning of the antenna is carried outadjusting, the lengths of the tuning stubs, the air gaps between layersand the length and width of the apertures so as to get the desiredbandwidths.

In general, the shape, length (la) and width (wa) of the apertureare designed to optimize the coupling between stubs and rings. In ourcase, the apertures bow-tie shaped, which offer, in the lower band,a relative bandwidth 9.5% wider than using conventional rectangularapertures of the same area. On the other hand, in the upper band therelative bandwidth is the same.

The apertures are positioned in the bottom layer, very close to


the outer edge of the annular-ring (x0). The input impedance of theantenna is about 50 Ω close to the outer edge of the annular-ring anddecreases when the apertures are moved towards the center of the rings.

When the length of stubs (ls) is decreased the bandwidth of thelower band is slightly reduced and shifted downwards in frequencywhereas the upper band is slightly shifted upwards in frequency.Finally, when the gaps between layers are increased (h1, h2) the lowerand upper bands are shifted upwards in frequency.

Once the dual-band antenna with two orthogonal linearly-polarized modes is designed, the four stubs must be connected to therespective output ports of the feeding network to generate right-handcircular polarization. Figure 1 shows the feeding network and the stubstogether. The feeding network is composed of three Wilkinson powercombiners and four broadband 90 phase shifters [16]. It is importantto remark that, with this feeding network, we obtain a balanced powerdistribution in the slots, with maximum error in phase lower than 2from 1.1 to 1.6 GHz.

Finally, the last step in the design procedure is to determine theshape and size of the substrate that supports the rings (Figure 1). Theaim is to improve the axial ratio for low elevation angles. Figure 2shows the simulated 4-dB axial ratio beamwidth as a function offrequency for the antenna shown in Figure 1 for different substrateshapes and sizes. In the lower band, the beamwidth of each antenna

1.1 1.12 1.14 1.16 1.18 1.2 1.22140

160

180

1.55 1.56 1.57 1.58 1.59 1.6120

140

160

180

4 d

B A

xia

l R

atio B

eam

wid

th (

)

o

lower band

upper band

Frequency (GHz)

Figure 2. Simulated 4 dB axial ratio beamwidth as a functionof the frequency for the dual-band antenna shown in Figure 1, forsquare substrate with D = 180mm (solid line), circular substratewith D = 180mm (solid line with circles) and circular substrate withD = 150 mm (dashed line), ϕ = 0 cut.


is very similar, meanwhile, in the upper band, circular substrates havewider beamwidth than the square substrate. Besides, the reduction inthe diameter of circular substrates reduces the beamwidth. However,it is important to point out that the size of the circular substrate canbe reduced significantly with a low degradation in the beamwidth.

2.2. Antenna Prototype

The proposed antenna is fabricated on Taconic TRF-45 (εr = 4.5,tan δ = 0.0035). The thicknesses of the substrates are h3 = 3.26mmand h4 = 1.63mm respectively. The feeding network is designed at thecenter frequency of both bands (1.4GHz). Initially, each annular ringis designed to resonate approximately at the center frequency of eachband. Next, the size and place of the aperture along with the length ofthe stubs and the air gaps between layers are adjusted, following thetendencies explained in the previous section, to obtain the specifiedbandwidth and center frequency of each band. The final dimensionsof our antenna are (Figure 1): h1 = h2 = 4 mm, D1in = 44 mm,D1out = 80 mm, D2in = 15.6mm, D2out = 71.2mm, wa = 12 mm,la = 50mm, x0 = 35mm and ls = 10 mm.

The substrate that supports the rings has circular shape with adiameter D = 100 mm. Figure 3 shows the fabricated antenna. Itshould be noted that 3 nylon screws and 9 nylon nuts are used toachieve the air gaps between layers.

Figure 3. Fabricated antenna.

3. EXPERIMENTAL RESULTS

Figure 4 shows the measured reflection coefficient for the designedantenna. Note that the response is matched from 1.1 to 1.7 GHz, dueto the feeding network behavior.


1.1 1.2 1.3 1.4 1.5 1.6 1.7-30

-25

-20

-15

-10

-5

0

Frequency (GHz)

S

(d

B)

11

Figure 4. Measured input reflection coecient for the fabricated dual-band circularly-polarized antenna.

-2

0

2

4

6

8

10

1.1 1.2 1.3 1.4 1.5 1.6 1.7

Frequency (GHz)

Gain

Axial Ration

Ga

in &

Axia

l R

atio

n (

dB

)

Figure 5. Measured gain (continuous line) and axial ratio (dashedline) as a function of frequency in the direction of maximum radiation(z-axis).

Figure 5 shows the measured axial ratio and gain as a functionof frequency. At the lower band (1.2 GHz) the gain is 6 dB and theaxial ratio 1.4 dB whereas at the upper band (1.575GHz) the gain is6.5 dB and the axial ratio is 0.4 dB. Figure 6 shows the simulated axialratio as a function of the elevation angle for the antenna prototype.Both bands have a wide angular coverage. The 4 dB axial ratiobeamwidth is higher than 156 for all frequencies. It is important


0

2

4

6

-90 -60 -30 0 30 60 90

f = 1.165 GHz

f = 1.175 GHz

f = 1.185 GHz

f = 1.195 GHz

f = 1.205 GHz

f = 1.215 GHz

f = 1.565 GHz

f = 1.575 GHz

f = 1.585 GHz

f = 1.595 GHz

-90 -60 -30 0 30 60 90

0

2

4

6

Theta ( )o

Axia

l R

atio (

dB

)

Figure 6. Simulated axial ratio as a function of angle θ, for differentfrequency values, ϕ = 0 cut.

to point out that the simulated and measured (not shown) axial ratioare in a good agreement except for low elevation angles, where themeasured axial ratio is degraded due to the effects of the metalliccage that supports the antenna during the measurement process. Thisdegradation reduces the measured 4 dB axial ratio beamwidth to 110for the center frequencies of both bands.

Comparing the obtained results with previous dual-band mi-crostrip antennas with circular polarization referenced in the intro-duction part, it can be verified that our measured reflection coefficientpresents a wider bandwidth than [1], and is similar to [3, 4] where feed-ing networks were also used. On the other hand, the measured gainand axial ratio as a function of frequency is very similar to the gainand axial ratio presented in [3, 4]. However, the proposed antenna hasa more compact size thanks to the use of the annular-rings and thereduction of the substrate size. Finally, it must be emphasized thatthe dual-band antenna shown in [6] presents a compact size because itis composed of only one layer. However, it only has a good axial ratioat boresight in the interesting bands, worse 4 dB axial ratio beamwidthand lower values of gain than our.

4. CONCLUSION

In this paper, a compact dual-band annular ring microstrip antennawith circular polarization has been presented. The proposed antennacovers the bands of Galileo (E5a, E5b, E1, E2, L1), GPS (L1, L5) and


GLONASS (L3). Aperture feeding has been used as an alternativetechnique to enhance the bandwidth and improve the input impedanceof dual-band annular-ring microstrip antennas. A broadband feedingnetwork has been used to improve the axial ratio at boresight. It hasalso been discussed as the shape and size of the substrate plays animportant role in the quality of the axial ratio for low elevation angles.Measurements show as the proposed antenna improves the performanceof previous designs that can be found in literature.

ACKNOWLEDGMENT

This work was supported by the Spanish Ministry of Scienceand Innovation through grant TEC2009-1387-C03-02 and theCONSOLIDER-INGENIO 2010 program, Ref-CSD2008-00068.

Measurements were carried out at the anechoic chamber of theAntenna Lab group of the Universitat Politecnica de Catalunya (UPC).The authors would like to thank Dr. J. M. Gonzalez Arbesu for hisvaluable comments and help during the prototype measurements.

REFERENCES

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