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CMOS UNIVERSAL REAL-TIME LABEL-FREE DNAANALYSIS SYSTEM-ON-CHIP

by

Hamed Mazhab Jafari

A thesis submitted in conformity with the requirementsfor the degree of Doctorate of Philosophy

Graduate Department of Electrical and Computer EngineeringUniversity of Toronto

Copyright c⃝ 2013 by Hamed Mazhab Jafari

II

Abstract

CMOS UNIVERSAL REAL-TIME LABEL-FREE DNA ANALYSIS

SYSTEM-ON-CHIP

Hamed Mazhab Jafari

Doctorate of Philosophy

Graduate Department of Electrical and Computer Engineering

University of Toronto

2013

Amperometric electrochemical DNA sensors have emerged as a low-cost, high-

throughput and real-time alternative to the conventional optical sensory methods. This

thesis presents the design, implementation, and validation of a fully integrated, mixed-

signal CMOS thermally controlled amperometric electrochemical DNA-sensing system-

on-chip (SoC).

The microsystem includes 54 current-to-digital channels, 600 on-chip nanostruc-

tured DNA sensors and 54 on-chip pH sensors. It reuses key circuits to perform three

key functions: 1) cyclic voltammetry and pH sensing, 2) impedance spectroscopy, and

3) temperature regulation.

Cyclic voltammetry DNA analysis and pH sensing is implemented by utilizing

the current-to-digital channel and an on-chip programmable waveform generator. The

current-to-digital channel is multiplexed between a bank of DNA sensors and a pH sen-

sor. The on-chip programmable waveform generator provides a wide range of user-

controlled rate, shape, and amplitude parameters of the sensor interrogation wave-

form, with a maximum scan range of 1.2V and a scan rate ranging from 0.1mV/sec

to 300V/sec.

Impedance spectroscopy DNA analysis is implemented by utilizing frequency re-

sponse analysis (FRA) to extract the impedance components of the biosensor. The most

III

computationally intensive operations, the multiplication and integration, required by

the FRA algorithm are performed by the in-channel dual-slope multiplying ADC in the

mixed-signal domain resulting in small integration area and power consumption.

The on-chip heating and temperature-sensing elements are implemented without

any post-CMOS processing. Temperature is regulated to within 0.5C using PID feed-

back control. This enables precise thermal control of on-chip DNA hybridization. The

two computationally intensive operations, multiplication and subtraction, required by

the PID algorithm are also efficiently performed by the same in-channel dual-slope mul-

tiplying ADC in the mixed-signal domain. A digital ultra-wideband transmitter based

on a delay line architecture provides wireless data communication capabilities.

The 3mm×3mm prototype fabricated in a 0.13µm standard CMOS technology has

been experimentally validated in the context of prostate cancer synthesized DNA marker

detection. Each recording channel occupies an area of only 0.06mm2 and consumes

42µW of power from a 1.2V supply. The digital ultra-wideband transmitter provides

wireless data communication capabilities with data rates of up to 50 Mb/s while con-

suming 400µW.

IV

Acknowledgements

First and foremost I would like to express my sincere thanks and deep appreciation

to my supervisor, Professor Roman Genov, for his thoughtful and constant guidance

throughout the course of this thesis, and for giving me the opportunities for the academic

advancements I have made. His regular encouragement, support and technical insights

were invaluable to me throughout the course of this work. Further, it enabled me to

perform to the best of my abilities and he provided me with opportunities and exposure

that I would otherwise never have had. I am thankful to work with such an outstanding

researcher and mentor.

I would like to thank my defense committee: Professor Glen Gulak, Professor David

Johns, Professor Ng and my external examiner, Professor Mason for their feedback that

helped improve this thesis. I would also like to thank our collaborators at the McMaster

University, Professor Leyla Solyemani for assisting us in DNA testing. I would also like

to thank CMC for fabrication access to to IBM 130nm CMOS, Jaro for CAD support

and NSERC for providing funding for my project.

I also thank the following members of Professor Genovs Microelectronics Research

Laboratory Alireza Nilchi, Ritu Raj Singh, Farzaneh Shahrokhi, Ruslana Shulyzki,

Derek Ho, Arezu Bagheri, Hossein Kassiri, Nima Soltani, and Arshya Feyzi for their

support, comments and assistance during the course of this research. I especially thank

Karim Abdulhalim, who was always helpful to me during the Ph.D program and spe-

cially during the tape-out.

Finally, I would like to express my deepest acknowledgements and appreciation to

my dear friend Morvarid Akhbari, for being a constant and unconditional source of

support during the writing of this thesis. Thank you for all you have done for me during

the past year and thank you for all your caring and support specially during the hard

time. This thesis is dedicated to you.

V

Contents

List of Tables xi

List of Figures xiii

List of Acronyms xxiii

1 Introduction 1

1.1 Basics of DNA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

1.1.1 DNA Structure and Chemical Properties . . . . . . . . . . . . . 3

1.1.2 DNA Concentration . . . . . . . . . . . . . . . . . . . . . . . 4

1.1.3 DNA Amplification by the Polymerase Chain Reaction (PCR) . 5

1.2 Conventional Bench-top DNA Analysis Methods . . . . . . . . . . . . 8

1.2.1 Optical Fluorescent DNA Sensing . . . . . . . . . . . . . . . . 8

1.2.2 Commercial Optical DNA-Sensing Instruments . . . . . . . . . 10

1.2.3 pH-based DNA Sensing . . . . . . . . . . . . . . . . . . . . . 11

1.2.4 Commercial pH-based DNA Sensing Instruments . . . . . . . . 12

1.3 Existing Integrated Circuits for DNA Sensing . . . . . . . . . . . . . . 14

1.3.1 CMOS Fluorescent DNA Contact Imaging . . . . . . . . . . . 15

1.3.2 CMOS Magnetic Field-Based DNA Sensing . . . . . . . . . . . 15

1.3.3 CMOS Capacitance-Based DNA Sensing . . . . . . . . . . . . 17

1.4 Amperometric Electrochemical Sensing Principles . . . . . . . . . . . 17

1.4.1 Three-electrode Sensing Configuration . . . . . . . . . . . . . 19

VI

1.4.2 Electrode-Electrolyte Interface . . . . . . . . . . . . . . . . . . 21

1.4.3 Reduction-Oxidation Current . . . . . . . . . . . . . . . . . . . 22

1.4.4 Labeled vs Label-free DNA Sensing . . . . . . . . . . . . . . . 22

1.4.5 Electrode Material . . . . . . . . . . . . . . . . . . . . . . . . 23

1.5 Amperometric Electrochemical Sensing Methods . . . . . . . . . . . . 24

1.5.1 Fast-Scan Cyclic Voltammetry . . . . . . . . . . . . . . . . . . 26

1.5.2 Amperometric Impedance Spectroscopy . . . . . . . . . . . . . 26

1.5.3 Temperature Regulation for DNA Analysis . . . . . . . . . . . 27

1.5.4 Commercial Amperometric Instruments . . . . . . . . . . . . . 28

1.6 Integrated Circuits for Amperometric DNA Sensing . . . . . . . . . . 28

1.7 Main Specifications for Amperometric Electrochemical DNA Sensing

Integrated Circuits Design . . . . . . . . . . . . . . . . . . . . . . . . 30

1.7.1 Accuracy . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

1.7.2 Dynamic Range . . . . . . . . . . . . . . . . . . . . . . . . . . 32

1.7.3 Frequency Range . . . . . . . . . . . . . . . . . . . . . . . . . 32

1.7.4 Non-electrical Design Specification . . . . . . . . . . . . . . . 33

1.8 Additional Considerations for DNA-sensing Amperometric Integrated

Circuits Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

1.8.1 Low-Level Current Acquisition for Amperometry . . . . . . . . 34

1.8.2 Wireless Data Transmission of DNA Results . . . . . . . . . . 35

1.8.3 DNA Temperature Regulation . . . . . . . . . . . . . . . . . . 37

1.8.4 On-chip Computation for DNA Analysis: Analog vs. Digital

vs. Mixed-Signal VLSI Multiplication . . . . . . . . . . . . . . 38

1.9 Thesis Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42

1.9.1 Complete DNA Analysis SoC . . . . . . . . . . . . . . . . . . 47

1.9.2 Cyclic Voltammetry and Sample pH Level Sensing . . . . . . . 48

1.9.3 Impedance Spectroscopy . . . . . . . . . . . . . . . . . . . . . 48

1.9.4 Temperature Regulation . . . . . . . . . . . . . . . . . . . . . 49

1.10 Thesis Outline . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49

VII

2 Current Acquisition Circuits for Electrochemical Amperometric Biosen-

sors 51

2.1 Transimpedance Amplifier (TIA) . . . . . . . . . . . . . . . . . . . . . 52

2.1.1 Circuit Implementation . . . . . . . . . . . . . . . . . . . . . . 53

2.1.2 Experimental Results . . . . . . . . . . . . . . . . . . . . . . . 55

2.2 Current Conveyer (CC) . . . . . . . . . . . . . . . . . . . . . . . . . . 56

2.2.1 Circuit Implementation . . . . . . . . . . . . . . . . . . . . . . 57

2.2.2 Channel Noise Analysis . . . . . . . . . . . . . . . . . . . . . 61

2.2.3 Experimental Results . . . . . . . . . . . . . . . . . . . . . . . 66

2.3 Comparative Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . 67

2.4 Chapter Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72

3 Cyclic Voltammetry and pH sensing 73

3.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74

3.2 DNA Detection Principle . . . . . . . . . . . . . . . . . . . . . . . . . 75

3.3 Integrated Sensors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76

3.3.1 DNA Sensing Microelectrodes . . . . . . . . . . . . . . . . . . 76

3.3.2 pH Sensors . . . . . . . . . . . . . . . . . . . . . . . . . . . . 77

3.4 VLSI Architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . 79

3.4.1 Top-Level VLSI Architecture . . . . . . . . . . . . . . . . . . 79

3.4.2 Channel VLSI Architecture . . . . . . . . . . . . . . . . . . . 81

3.5 Circuit Implementation . . . . . . . . . . . . . . . . . . . . . . . . . . 84

3.5.1 Current Conveyer . . . . . . . . . . . . . . . . . . . . . . . . . 84

3.5.2 Dual-slope ADC Comparator . . . . . . . . . . . . . . . . . . 88

3.5.3 Ultra-wideband Transmitter . . . . . . . . . . . . . . . . . . . 88

3.6 Electrical Experimental Results . . . . . . . . . . . . . . . . . . . . . . 89

3.7 Experimental Electrochemical Results . . . . . . . . . . . . . . . . . . 95

3.8 Synthetic Prostate Cancer DNA Detection . . . . . . . . . . . . . . . . 102

3.9 Chapter Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 110

VIII

4 Impedance Spectroscopy DNA 111

4.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111

4.2 DNA Detection Principle . . . . . . . . . . . . . . . . . . . . . . . . . 112

4.3 Impedance Spectroscopy VLSI Architecture . . . . . . . . . . . . . . . 113


4.4.1 Multi-channel System-Level Architecture . . . . . . . . . . . . 116

4.4.2 Waveform Generator and Pattern Generator . . . . . . . . . . . 118

4.4.3 Dual-slope Multiplying ADC Channel . . . . . . . . . . . . . . 120

4.4.4 Three-electrode Regulation Loop . . . . . . . . . . . . . . . . 125

4.5 Electrical Experimental Results . . . . . . . . . . . . . . . . . . . . . . 126

4.6 Analog vs Digital vs Mixed-Signal Multiplication . . . . . . . . . . . . 131

4.7 Electrochemical Experimental Results . . . . . . . . . . . . . . . . . . 135

4.8 Chapter Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 138

5 Temperature Regulation 140

5.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 141

5.2 Thermal Control Principles . . . . . . . . . . . . . . . . . . . . . . . . 142

5.2.1 Temperature regulation . . . . . . . . . . . . . . . . . . . . . . 142

5.2.2 Temperature sensing . . . . . . . . . . . . . . . . . . . . . . . 144

5.2.3 Temperature Regulation . . . . . . . . . . . . . . . . . . . . . 148

5.3 VLSI Architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . 149


5.4.1 Current-to-Digital Channel . . . . . . . . . . . . . . . . . . . . 150

5.4.2 CTAT Current Source . . . . . . . . . . . . . . . . . . . . . . . 153

5.4.3 PTAT Current Source . . . . . . . . . . . . . . . . . . . . . . . 155

5.4.4 Digital Pulse Width Modulator . . . . . . . . . . . . . . . . . . 156

5.5 Experimental Results . . . . . . . . . . . . . . . . . . . . . . . . . . . 157

5.6 Chapter Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 165

IX

6 Conclusions and Future Work 167

6.1 Contributions and Related Publications . . . . . . . . . . . . . . . . . . 167

6.2 Future Work . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 171

6.2.1 Power Harvesting and Wireless Communication Chip . . . . . . 171

6.2.2 Improvements to the SoC . . . . . . . . . . . . . . . . . . . . . 171

References 173

Appendix 187

A Supplementary Hardware and Software Documentation 187

A.1 Board Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 187

A.2 DNA Sensing protocol . . . . . . . . . . . . . . . . . . . . . . . . . . 190

A.3 Data Acquisition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193

A.4 FPGA Verilog Code . . . . . . . . . . . . . . . . . . . . . . . . . . . . 197

A.4.1 Top Level Module . . . . . . . . . . . . . . . . . . . . . . . . 197

A.4.2 USB Controller . . . . . . . . . . . . . . . . . . . . . . . . . . 206

A.4.3 Master Clock Divider . . . . . . . . . . . . . . . . . . . . . . . 208

A.4.4 USB Clock Divider . . . . . . . . . . . . . . . . . . . . . . . . 209

A.4.5 One Channel FIFO Controller . . . . . . . . . . . . . . . . . . 210

A.4.6 FIFO Controller . . . . . . . . . . . . . . . . . . . . . . . . . 214

A.4.7 16-bits Parallel to Serial Converter . . . . . . . . . . . . . . . . 217

A.4.8 Timing PLL . . . . . . . . . . . . . . . . . . . . . . . . . . . . 218

A.4.9 Waveform Generator . . . . . . . . . . . . . . . . . . . . . . . 224

A.4.10 Master Reset . . . . . . . . . . . . . . . . . . . . . . . . . . . 226

A.4.11 Seven Bit Counter . . . . . . . . . . . . . . . . . . . . . . . . 227

A.4.12 Chip Timing Coefficients . . . . . . . . . . . . . . . . . . . . . 228

A.4.13 SRAM Timing Coefficients . . . . . . . . . . . . . . . . . . . 229

A.4.14 Channel Range Controller Coefficients . . . . . . . . . . . . . 233

A.4.15 Impedance Spectroscopy SRAM Coefficients . . . . . . . . . . 235

X

A.4.16 Temperature Regulation SRAM Coefficients . . . . . . . . . . 236

A.4.17 Test Channel Controller . . . . . . . . . . . . . . . . . . . . . 237

A.5 Example CV Recording Verilog Code . . . . . . . . . . . . . . . . . . 238

A.6 Example IS Recording Verilog Code . . . . . . . . . . . . . . . . . . . 247

A.7 Example Temperature Regulation Verilog Code . . . . . . . . . . . . . 258

XI

List of Tables

1.1 Comparative Analysis of Amperometric Biomolecular Sensory Microsys-

tems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44

2.1 TIA OTA Transistors Sizing . . . . . . . . . . . . . . . . . . . . . . . 58

2.2 Current Conveyer Transistor Sizing . . . . . . . . . . . . . . . . . . . . 66

2.3 Experimentally Measured Transimpedance Amplifier (TIA) and Cur-

rent Conveyer (CC) Characteristics . . . . . . . . . . . . . . . . . . . . 71

3.1 ADC Comparator Transistor Sizing . . . . . . . . . . . . . . . . . . . 85

3.2 Experimentally Measured Electrical Characteristics . . . . . . . . . . . 100

3.3 Comparative Analysis of Amperometric Sensory Microsystems . . . . . 109

4.1 OTA1 and Comparator Transistor Sizing . . . . . . . . . . . . . . . . . 123

4.2 Analog Integrator Noise Summary . . . . . . . . . . . . . . . . . . . . 124

4.3 OTA2 and OTA3 Transistors Sizing . . . . . . . . . . . . . . . . . . . 126

4.4 Experimentally Measured Characteristics . . . . . . . . . . . . . . . . 130

4.5 Serial Digital Multiplier Area Breakdown . . . . . . . . . . . . . . . . 134

4.6 Comparison of Mixed-Signal vs Analog vs Digital Multiplication . . . . 135

4.7 Comparative Analysis of Reported Amprometric Sensory Microsystem 136

5.1 OTA Transistor Sizing . . . . . . . . . . . . . . . . . . . . . . . . . . 152

5.2 Experimentally Measured Characteristics . . . . . . . . . . . . . . . . 161

XII

5.3 Comparative Analysis of Reported Thermally Regulated Biosensory

Microsystems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 163

XIII

List of Figures

1.1 The structure of the DNA double helix [2]. . . . . . . . . . . . . . . . . 4

1.2 Schematic of polymerase chain reaction cycles. . . . . . . . . . . . . . 6

1.3 A fluorescence-based DNA microarray [14]. . . . . . . . . . . . . . . . 9

1.4 A typical scanner for fluorescence-based DNA detection [14]. . . . . . 10

1.5 Commercially available fluorescence-based DNA microarray scanners:

(a) Affymetrix GeneChip Scanner 3000 7G [35], (b) Agilent SureScan

Microarray Scanner [36], and (c) Spartan RX microarray scanner [37]. . 11

1.6 pH-based DNA sensing: (a) system cross-section, (b) complementary

DNA sequence and (c) and non-complementary DNA sequence [41]. . . 13

1.7 Commercially available pH-based DNA-sensing instruments: (a) Ion

Torrent [41], (b) DNA Electronics [44], and (b) GeneOnyx [42]. . . . . 14

1.8 Conceptual view of an electrochemical amperometric sensing microsys-

tem. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

1.9 Charge profile of the electrode-electrolyte interface and its equivalent

circuit model. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

1.10 Working electrode material examples: (a) diamond [71], (b) polymer [20],

and (c) flat gold [70]. . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

1.11 Fast-scan cyclic voltammetry principle of operation (a) cyclic redox po-

tential applied between the working and reference electrode, and (b)

cyclic voltammogram in the absence and presence of the target chemical. 25

XIV

1.12 Impedance spectroscopy principle of operation (a) potential difference

between working and reference electrode, and (b) change in redox cur-

rent frequency response due to the change in the chemical concentration. 27

1.13 Commercially available amperometric instruments: (a) Early Warning

biohazard water analyzer [38], (b) Palmsens 3 [39], and (c) PSTAT

mini [40]. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29

1.14 Biosensor SNR vs target analyte concentration [23]. . . . . . . . . . . . 32

1.15 Conceptual view of current acquisition circuits for electrochemical am-

perometric sensory system. . . . . . . . . . . . . . . . . . . . . . . . . 34

1.16 Conventional transimpedance amplifier (TIA) interfaced to a biosensor

model. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35

1.17 Conventional current conveyer (CC) interfaced to a biosensor model. . . 35

1.18 Trojan UV3000PTP wastewater solution [31]. . . . . . . . . . . . . . . 36

1.19 Analog multiplier block diagram [103]. . . . . . . . . . . . . . . . . . 39

1.20 Analog multiplier circuit schematic. . . . . . . . . . . . . . . . . . . . 39

1.21 The block diagram of the 16×16-bit serial multiplier [128]. . . . . . . . 40

1.22 (a) The critical path of the 16×16-bit parallel multiplier and (b) the

block diagram of the 16×16-bit two-cycle parallel multiplier [128]. . . 41

1.23 Reuse of the SoC architecture resources (blocks outlined in bold are

reused in all modes) (a) complete DNA analysis SOC, (b) cyclic voltam-

metry configuration, (c) impedance spectroscopy configuration and (d)

temperature regulation configuration. . . . . . . . . . . . . . . . . . . . 46

2.1 Chopper-stabilized bidirectional-input transimpedance amplifier (TIA). 53

2.2 Circuit schematic diagram of the OTA within the TIA. . . . . . . . . . 54

2.3 Simulated input-referred noise spectrum of the TIA from 0.01Hz to 1kHz. 55

2.4 Transimpedance amplifier noise summary: (a) flicker noise contribu-

tions, and (b) thermal noise contributions. . . . . . . . . . . . . . . . . 56

XV

2.5 Experimentally measured relative error of the output of the TIA for the

input current of (a) 10pA to 350nA, and (b) -350nA to -10pA. The

results are measured from one typical channel on one chip. . . . . . . . 57

2.6 Experimentally measured TIA output current of 16 channels (from 16

chips, one channel each) for the input current of (a) 100pA, and (b)

100nA. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58

2.7 Conventional pseudo-bidirectional current conveyer with a DC offset

current. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59

2.8 Chopper-stabilized bidirectional current conveyer VLSI architecture. . . 59

2.9 Chopper-stabilized bidirectional current conveyer circuit schematic di-

agram. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 60

2.10 Simulated current conveyer AC response . . . . . . . . . . . . . . . . . 61

2.11 Simplified noise model of the chopper-stabilized current conveyer. . . . 62

2.12 Simulated input-referred noise spectrum of the current conveyer from

0.01Hz to 1kHz. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66

2.13 Current conveyer noise summary: (a) flicker noise contributions, and

(b) thermal noise contributions. . . . . . . . . . . . . . . . . . . . . . . 67

2.14 Experimentally measured relative error of the output of the current con-

veyer for the input current of (a) 10pA to 350nA, and (b) -350nA to

-10pA. The results are measured from one typical channel on one chip. . 68

2.15 Experimentally measured current conveyer output current of 16 chan-

nels (from 16 chips, one channel each) for the input current of (a)

100pA and (b) 100nA. . . . . . . . . . . . . . . . . . . . . . . . . . . 69

2.16 Average current injected into the working electrode for (a) TIA and (b)

CC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70

2.17 Cyclic voltammogram of 2µM potassium ferricyanide in 1M potas-

sium phosphate buffer solution experimentally recorded with the tran-

simpedance amplifier (TIA) and the current conveyer (CC) using a 50µm×50µm

gold electrode. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70

XVI

2.18 Cyclic voltammogram of 2µM potassium ferricyanide in 1M potas-

sium phosphate buffer solution experimentally recorded with the tran-

simpedance amplifier (TIA) and the current conveyer (CC) using a 2µm×2µm

gold electrode. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72

3.1 Label-free electrochemical DNA detection principle. (a) Bare elec-

trode: maximum charge transfer between working and reference elec-

trode in the absence of negatively charged probe and target DNA; (b)

non-complementary target DNA: reduction in the charge transfer rate

due to the presence of negatively charged probe DNA, and (c) comple-

mentary DNA: further reduction in the charge transfer rate due to the

presence of negatively charged target and probe DNA. . . . . . . . . . 75

3.2 (a) Passivation opening in standard CMOS and added metal layers of a

flat (2D) microelectrode after electroless nickel-palladium-gold plating,

(b) and (c) SEM photographs of such 55µm×55µm working electrodes. 78

3.3 Nanostructured DNA sensing working electrodes (NMEs): (a) Cross-

sectional view of a 2µm×2µm passivation opening in standard CMOS,

(b) SEM photograph of a 2µm×2µm working electrode passivation

opening over an aluminum base, (c) nanostructured 2µm×2µm work-

ing electrode grown on the passivation opening over an aluminum base

in standard CMOS, (c) (d) and (e) SEM photographs of nanostructured

microelectrodes grown at different electrodeposition conditions on the

passivation opening in (b). . . . . . . . . . . . . . . . . . . . . . . . . 79

3.4 Wireless DNA analysis microsystem functional block diagram. . . . . . 80

3.5 Simplified top-level VLSI architecture of one chopper-stabilized inte-

grated current-to-digital channel. . . . . . . . . . . . . . . . . . . . . . 81

3.6 Detailed implementation of the current conveyer OTA with internal

chopping and dynamically-matched low-current regulation. . . . . . . . 81

3.7 High-speed latched comparator circuit schematic diagram. . . . . . . . 84

XVII

3.8 (a) Ultra-wideband transmitter circuit schematic diagram, and (b) schematic

of one current-starved inverter. . . . . . . . . . . . . . . . . . . . . . . 86

3.9 Timing diagram of the ultra-wideband transmitter. . . . . . . . . . . . . 86

3.10 Die micrograph of the 3mm×3mm 54-channel wireless DNA analysis

SoC. The SoC was fabricated in a 0.13µm standard CMOS technology. 87

3.11 Experimentally measured spectrum of the electrochemical recording

channel output for a 15Hz sinusoidal full-scale (350nA) input. . . . . . 89

3.12 Experimentally measured transfer characteristics of the current-to-digital

channel for three sampling frequencies. . . . . . . . . . . . . . . . . . 90

3.13 Experimentally measured output ENOB of 32 channels (from 16 chips,

two channels each) for a 15Hz 350nA sinusoidal input (a) without cali-

bration (b) with in-channel calibration. . . . . . . . . . . . . . . . . . . 91

3.14 Experimentally measured ENOB vs. input frequency for the in-channel

ADC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 92

3.15 Experimentally measured relative error of the output digital code of the

current conveyer connected with the dual-slope ADC for (a) 10pA to

350nA and (b) -350nA to -10pA input current. . . . . . . . . . . . . . . 93

3.16 Experimentally measured output current of 32 channels (from 16 chips,

two channels each) for the input current of (a) 100pA without DEM

[32] and (b) 100pA with DEM (this work). . . . . . . . . . . . . . . . . 94

3.17 Experimentally measured output current of 32 channels (from 16 chips,

two channels each) for input current of (a) 100nA without DEM [32]

and (b) 100nA with DEM (this work). . . . . . . . . . . . . . . . . . . 95

3.18 Experimentally measured (a) Manchester-encoded input data to the UWB

transmitter and (b) the output pulses. . . . . . . . . . . . . . . . . . . . 96

3.19 Wirelessly measured UWB pulse at the distance of 5cm using a custom-

built UWB receiver. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 97

XVIII

3.20 Experimentally measured UWB transmitter output spectrum (direct out-

put of the transmitter driving a 50 ohm load). The output spectrum is

compliant with the 0-1GHz FCC UWB band output power criteria . . . 97

3.21 (a) Manchester-encoded input data to the UWB transmitter and (b) the

corresponding data received wirelessly at a 5cm distance. . . . . . . . . 98

3.22 Experimentally recorded cyclic voltammograms of 1M potassium phos-

phate buffer and 20µM potassium ferricyanide solution using the 55µm×55µm

working electrode in Fig. 3.2(b) . . . . . . . . . . . . . . . . . . . . . 98

3.23 Experimentally recorded cyclic voltammograms of 10µM, 20µM, 30µM

and 40µM potassium ferricyanide solution using the 55µm×55µm work-

ing electrode in Fig. 3.2(b) . . . . . . . . . . . . . . . . . . . . . . . . 99

3.24 Calibration curve for the peak reduction current of potassium ferri-

cyanide solution for the 55µm×55µm Au . . . . . . . . . . . . . . . . 99

3.25 Experimentally recorded peak reduction current of the 40µM potassium

ferricyanide solution recorded using the 55µm×55µm working elec-

trode shown in Fig. 3.2(b) by 48-channel on the CMOS DNA analysis

SoC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100

3.26 Experimentally measured on-chip pH sensor calibration curve relative

to pH of 7. A total of 60 measurements from 3 chips, 20 measurements

each, have been performed. The corresponding 3σ error bars are shown. 101

3.27 Experimentally measured cyclic voltammetry results of 5µM prostate

cancer synthetic DNA detection from the 55µm×55µm flat gold work-

ing electrode in Fig. 3.2(b) . . . . . . . . . . . . . . . . . . . . . . . . 103

3.28 Experimentally measured 5µM prostate cancer synthetic DNA cyclic

voltammetry recording 3σ error bars from 3 chips 60 measurements

each from 55µm×55µm flat gold working electrodes in Fig. 3.2(b). . . 104

3.29 Experimentally measured cyclic voltammetry results of 100aM prostate

cancer synthetic DNA detection, from 2µm×2µm nanostructured work-

ing electrodes in Fig. 3.3(e) . . . . . . . . . . . . . . . . . . . . . . . . 105

XIX

3.30 Experimentally measured 100aM prostate cancer synthetic DNA cyclic

voltammetry recording 3σ error bars from 3 chips, 60 measurements

each, from 2µm×2µm nanostructured working electrodes in Fig. 3.3(e) 105

3.31 Experimentally measured microelectrode characteristics, detection lim-

its and dynamic ranges in prostate cancer synthetic DNA detection us-

ing the three electrodes types shown in Fig. 3.3(d) and (e) and Fig. 3.2(b).

Error bars (3 sigma) are from 3 chips, 100 measurements each. . . . . . 106

4.1 Block diagram of a frequency-response analyzer (FRA) system for biosen-

sor impedance spectroscopy. . . . . . . . . . . . . . . . . . . . . . . . 113

4.2 Impedance spectroscopy microsystem functional block diagram. . . . . 114

4.3 Waveform generator functional block diagram. . . . . . . . . . . . . . 115

4.4 Principle of the sine wave generation. . . . . . . . . . . . . . . . . . . 115

4.5 Absolute value of the relative error of the biosensor R-C model impedance

as a function of frequency due to stepwise approximation of the inter-

rogation signal for the : (a) real, and (b) imaginary components. . . . . 116

4.6 (a) Dual-slope multiplying ADC VLSI architecture, and (b) timing dia-

gram of all relevant signals. . . . . . . . . . . . . . . . . . . . . . . . . 117

4.7 Timing diagram illustrating the ADC multiplication function. . . . . . . 118

4.8 Chopper-stabilized folded-cascode OTA1 in the analog integrator. . . . 122

4.9 Comparator circuit diagram. (a) One of the three identical gain stages,

and (b) the high-speed latch. . . . . . . . . . . . . . . . . . . . . . . . 122

4.10 Simulated input-referred noise spectrum of the analog integrator with

the chopper on and off (clocked at 100kHz) from 0.01Hz to 10kHz. . . 123

4.11 (a) Circuit architecture of the three-electrode regulation loop, and (b)

OTA2 and OTA3 circuit schematic. . . . . . . . . . . . . . . . . . . . . 124

4.12 Die micrograph of the 54-channel integrated impedance spectrum ana-

lyzer fabricated in a 0.13µm standard CMOS technology. . . . . . . . . 126

XX

4.13 Experimentally measured transfer characteristics of the impedance spec-

trum analyzer channel for three sampling frequencies. . . . . . . . . . . 127

4.14 Experimentally measured output spectrum of the ADC for a 10Hz sinu-

soidal input. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 127

4.15 Experimentally measured MADC SNDR vs. the multiplication factor. . 128

4.16 Waveform generator DNL. . . . . . . . . . . . . . . . . . . . . . . . . 128

4.17 Waveform generator INL. . . . . . . . . . . . . . . . . . . . . . . . . . 129

4.18 Total harmonic distortion of waveform generator DAC. . . . . . . . . . 129

4.19 Off-chip biosensor model impedance as a function of frequency exper-

imentally measured by the impedance spectroscopy microsystem: (a)

real, and (b) imaginary components. . . . . . . . . . . . . . . . . . . . 131

4.20 Absolute value of the relative error of the off-chip biosensor model

impedance as a function of frequency: (a) real, and (b) imaginary com-

ponents. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 132

4.21 Potassium ferricyanide solution impedance as a function of frequency

experimentally measured by the impedance spectroscopy microsystem:

(a) real, and (b) imaginary components. . . . . . . . . . . . . . . . . . 133

4.22 (a) Impedance spectrum of 5µM prostate cancer synthetic DNA probe,

complementary, non-complementary targets and (b) the corresponding

3-sigma error bars (from 3 chips, 20 measurements each). . . . . . . . . 138

5.1 Block diagram of PID implementation of the temperature regulation loop.142

5.2 (a) A pair of BJTs for generating ∆VBE and VBE , (b) Temperature de-

pendence of the base-emitter voltage, (c) generation of PTAT and CTAT

current, (d) generation of temperature dependent current and (e) utiliza-

tion of a current-to-digital channel for temperature measurements. . . . 143

5.3 Top-level VLSI architecture of temperature regulation loop (shaded boxes

are in-channel). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 147

5.4 Timing diagram of the temperature regulation loop. . . . . . . . . . . . 148

XXI

5.5 Top-level VLSI architecture of one temperature sensing channel. . . . . 149

5.6 In-channel heater schematic. . . . . . . . . . . . . . . . . . . . . . . . 150

5.7 Timing diagram of the dual-slope multiplying ADC. . . . . . . . . . . . 151

5.8 Circuit diagram of the CTAT sensor and CTAT current source . . . . . . 153

5.9 Chopper-stabilized folded-cascode OTA in the CTAT (OTA1) and PTAT

(OTA2) current sources. . . . . . . . . . . . . . . . . . . . . . . . . . . 153

5.10 Simulated input-referred noise spectrum of the OTA with chopper en-

abled and disabled from 0.01Hz to 1kHz. . . . . . . . . . . . . . . . . 154

5.11 Simulated OTA noise summary: (a) thermal noise contribution, and (b)

flicker noise contribution. . . . . . . . . . . . . . . . . . . . . . . . . . 155

5.12 Circuit diagram of the PTAT sensor and PTAT current source. . . . . . . 156

5.13 Block diagram of the digital pulse width modulator. . . . . . . . . . . . 157

5.14 Die micrograph of the 54-cell wireless temperature regulated DNA anal-

ysis SoC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 157

5.15 Experimentally measured output spectrum of the ADC for a 15Hz sinu-

soidal input. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 158

5.16 Experimentally measured transfer function of the pulse-width-modulator. 158

5.17 Experimentally measured digital output of the dual-slope ADC vs tem-

perature (in-air). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 159

5.18 TP04390A ThermoStream [169] unit used for temperature sensor char-

acterization. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 160

5.19 Temperature sensor error experimentally measured on seven dies from

one wafer (in-air). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 161

5.20 Experimentally measured temperature from 54 channels in one chip (in-

air). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 162

5.21 Temperature regulation cycle with steps at 35, 45, 55 and 65 degree

Celsius (in-liquid). . . . . . . . . . . . . . . . . . . . . . . . . . . . . 164

5.22 Measured absolute value of the relative error of PID regulation loop

(in-air). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 165

XXII

6.1 Die micrograph of the 2mm×3mm power harvesting and wireless com-

munication chip. The chip was fabricated in a 0.13µm standard CMOS

technology. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 172

A.1 PCB fabricated for characterizing the DNA analysis SoC. . . . . . . . . 188

XXIII

List of Acronyms

ADC analog-to-digital converter

DAC digital-to-analog converter

SNR signal-to-noise ratio

VLSI very-large-scale integration

OTA operational transconductance amplifier

TIA transimpedance amplifier

FOM figure-of-merit

LOC lab-on-chip

RMS root mean square

SC switched capacitor

DSP digital signal processing

LFSR linear feedback shift register

OPAMP operational amplifier

IC integrated circuit

1

Chapter 1

Introduction

The genetic blue print of every living organism is defined by its genome which is com-

prised of a sequence of nucleotide bases that contains deoxyribo-nucleic acid (DNA).

The microbiological culture techniques currently employed to identify disease-causing

pathogens are time consuming, labor-intensive, often have a high false positive rate,

do not provide the necessary information about the potential pathogenicity or virulence

of the identified organisms, and are not well suited for in-field managing of a large

number of samples. These limitations have stimulated efforts to develop DNA-based

pathogen detection methods. The advantage of DNA-based methods is the opportu-

nity to pinpoint precisely the unique DNA and ribonucleic acid (RNA) signature that

each pathogen carries, differentiating it from other organisms having vastly different

virulences.

This thesis focuses on the design, implementation, and experimental validation

of a fully integrated mixed-signal CMOS thermally controlled amperometric electro-

chemical DNA-sensing system-on-chip (SoC). The microsystem includes 54 current-

to-digital channels, 600 on-chip nanostructured DNA sensors, and 54 on-chip pH sen-

sors. The microsystem is universal since it reuses key circuits to perform three key

CHAPTER 1. INTRODUCTION 2

functions: (1) cyclic voltammetry and pH sensing, (2) impedance spectroscopy, and

(3) temperature regulation. Chapter 1 of this thesis provides a brief overview of the

DNA structure and DNA sensing techniques, which is followed by a literature review

and an overview of the work that is to be presented. Chapter 2 presents two different

analog current-recording front-end circuits for electrochemical sensing applications and

compares their advantages and disadvantages. Chapter 3 focuses on the cyclic voltam-

metry DNA-sensing capability of the SoC and presents a fully integrated 54-channel

wireless label-free cyclic voltammetry DNA analyzer. Chapter 4, which focuses on the

impedance spectroscopy DNA sensing capability of the SoC, shows the SoC configured

as an impedance analyzer that utilizes frequency response analysis (FRA) to extract the

real and imaginary impedance components of the biosensor. Chapter 5 discusses the

temperature regulation capabilities of the SoC and introduces a 54-site mixed-signal

CMOS thermal regulator with a compact mixed-signal PID controller. The conclusions

of the thesis and the possible directions of future work are discussed in Chapter 6.

In the introductory section of Chapter 1, basics of DNA analysis are presented in

Section 1.1. An overview of conventional bench-top DNA analysis methods and a sum-

mary of commercially available DNA sensing and analysis instruments are provided

in Section 1.2. Prior research works on non-amperometric CMOS integrated DNA

analysis SoCs are presented in Section 1.3. Background information and an overview

of amperometric electrochemical DNA-sensing principles and different amperometric

electrochemical sensing methods are provided in Sections 1.4 and 1.5. Prior research

works on CMOS amperometric integrated DNA analysis SoCs are presented in Sec-

tion 1.6. The specifications and requirements of an electrochemical DNA analysis SoC

are described in Section 1.7. Summary of additional design considerations for DNA-

sensing amperometric integrated circuits are provided in Section 1.8. Overview of the

work presented in the thesis is provided in Section 1.9. Thesis outline is presented in


Section 1.10.

1.1 Basics of DNA

1.1.1 DNA Structure and Chemical Properties

Deoxyribo-nucleic acid (DNA) is a nucleic acid that contains the genetic blueprint re-

quired for development and functioning of the living organisms. DNA is composed of

two polymer strands made of units called nucleotides. The backbone of DNA is made

of sugars and phosphate groups joined by ester bonds. Each nucleotide consists of three

units: a phosphate group, a five carbon deoxyribose group, and a nitrogen base group.

The phosphate group loses a proton in a neutral solution giving rise to the overall neg-

ative charge of the DNA. The nucleotides units in DNA include adenine (A), cytosine

(C), guanine (G), and thymine (T) as shown in Fig. 1.1. The four bases are bonded to

the sugar/phosphate DNA backbone to form the complete nucleotide [1].

DNA does not exist as a single molecule in any living organism, but exists as a

pair of molecules that are bonded together. Two long strands of DNA are twisted to-

gether like vines, in the shape of a double helix. The double helix DNA is stabilized by

hydrogen bonds between nucleotides [2].

The double helix DNA molecule can be separated into two single-stranded DNA

(ssDNA) molecules in a process called de-naturing. De-naturing is normally achieved

by heating the double helix DNA. In reverse, two complementary ssDNA molecules

can form a double-stranded DNA molecule in a process known as hybridization or re-

naturation [2].

Polymerase chain reaction (PCR) amplification technique is commonly used to am-

plify specific regions of a DNA strand for more accurate sensing (i.e, to make many

copies of these DNA regions in order to increase and then measure their concentration).


Figure 1.1: The structure of the DNA double helix [2].

1.1.2 DNA Concentration

Concentration of a constituent (e.g., DNA molecules) in a chemical mixture is defined

as the quantity of a constituent divided by the total volume of a mixture. There are

several mathematical description that can quantify concentration such as: mass con-

centration, molar concentration, number concentration, and volume concentration. The

term concentration can be applied to any kind of chemical mixture, but most frequently

it refers to solutes and solvents in solutions.

The most commonly used term for quantifying concentration is molar concentration.


The molar concentration is defined as the amount of a constituent in units of mol divided

by the volume of the mixture. The unit of molar concentration is mol/L.

Given the molar concentration of a sample, number of molecules in the sample can

be calculated as follows:

1. Calculate the absolute amount of the sample in units of mol:

Concentration(mol/L)×volume(L)= Amount(mol)

2. Next use the Avogadro constant to get the number of the molecules in the sample

as follows:

Amount(mol)×6.02214129×1023 mol−1 = number of the molecules in the volume

For example, 1mL solution containing 10aM concentration of a target DNA would

containing 10aM×1mL×6.02214129×1023 target DNA molecules. This corresponds

to 602 target DNA molecules.

1.1.3 DNA Amplification by the Polymerase Chain Reaction (PCR)

The polymerase chain reaction (PCR) is a procedure in biochemistry to amplify a single

or a few copies of a DNA sequence, generating up to millions of copies of the same

DNA sequence [62–65]. This method is utilized to increase the concentration of the

target DNA in a sample, thus increasing the detection accuracy. PCR relies on thermal

cycling which consists of cycles of repeated heating and cooling for DNA melting and

enzymatic replication of the DNA. Short DNA fragments (less than 100 base pairs [65]

containing sequences complementary to the regions of the target DNA sequence are

key components to enable selective and repeated amplification. As PCR progresses, the

DNA generated is itself used as a template for replication, setting a chain reaction in

which the DNA template is exponentially amplified.

An example of PCR is shown in Fig. 1.2. A brief descriptions of the PCR steps is

as follows:


Figure 1.2: Schematic of polymerase chain reaction cycles.


1. Initialization step: Heating the DNA solution to a temperature of 94 to 96 oC and

holding the solution at that temperature for 1 to 9 minutes [62].

2. Denaturation step: Heating the DNA solution to 94 to 98 oC for 20 to 30 seconds.

This results in melting of the DNA template by breaking the hydrogen bonds between

complementary bases, resulting in single-stranded DNA molecules.

3. Annealing step: Reducing the DNA solution temperature to 50 to 65 oC for 20

to 40 seconds resulting in annealing of the DNA oligonucleotides to the single-stranded

DNA template. DNA-DNA hydrogen bonds are formed when the DNA oligonucleotides

sequence closely matches the template sequence. The polymerase binds to the oligonucleotides-

template hybrid and begins DNA formation.

4. Extension step: The temperature is increased to 75 to 80 oC [62, 63]. At this

step the DNA polymerase synthesizes a new DNA strand complementary to the DNA

template strand by adding deoxynucleotide triphosphates that are complementary to

the template in 5’ to 3’ direction, condensing the 5’-phosphate group of the deoxynu-

cleotide triphosphates with the 3’-hydroxyl group at the end of the nascent DNA strand.

The amount of DNA target is doubled, resulting in exponential increase of the specific

DNA fragment over time.

5. Final elongation: This step performed at a temperature of 70 to 74 oC for 5 to 15

minutes after the last PCR cycle ensuring that any remaining single-stranded DNA is

fully extended.

6. Final hold: Temperature is reduced to 4 to 15 oC for an indefinite time for short-

term storage of the DNA samples.


1.2 Conventional Bench-top DNA Analysis Methods

Analysis of specific biomolecules, such as proteins and nucleic acids, finds applica-

tions that range from disease diagnostics to forensics. Detection, identification, and

quantification of nucleic acid sequences such as DNA, messenger RNA (mRNA), and

micro-RNA have recently been exploited for the diagnosis, prognosis, and treatment

choices for heredity and infectious diseases [3,4], the monitoring of the quality of food

and water [5], and the detection of bio-warfare agents [6].

1.2.1 Optical Fluorescent DNA Sensing

Fluorescence-based DNA sensing is currently the gold standard technique for DNA

analysis. It finds applications in gene expression [29], pharmacogenomics [32] and

drug development analysis [33].

As shown in Fig. 1.3, in this method groups of ssDNA probe molecules are immo-

bilized in an array format on a passive substrate. The DNA probes can be constructed

using solid-phase chemical synthesis techniques and they are immobilized on the mi-

croarray though spotting or non-contact ink-jet printing. ssDNA target molecules are

tagged with chemical labels. After absorbing a photon at a specific wavelength, these

labels emit photons at a higher wavelength. Common labels are fluorescent dyes and

fluorescent which emit lights with wavelength in the range of 500 to 700nm.

When the target ssDNA molecules are introduced onto the microarray, the probe

molecules and the complementary target molecules hybridize. The array is usually

exposed to the target molecules for several hours so that the hybridization reaches an

equilibrium. Next the array is washed so that the un-bonded targets are removed. The

washing step is a major drawback of optical DNA analysis method as it eliminates the

possibility of real-time monitoring of the hybridization. This is due to the fact that the


Modify substrate to

accept DNA probes

Immobilize and

pattern probesReact with DNA

target analyte

Scan and process

microarray

DNA target DNA probe

Fluorescence labeled

Sequence 3

Sequence 1

Sequence 4

Sequence 2

Figure 1.3: A fluorescence-based DNA microarray [14].

analyte must be removed from the array before a meaningful measurement can be done.

Next, as shown in Fig. 1.4, the microarray is illuminated with a laser light source

and the fluorescence light emitted from the hybridized DNA strands is recorded using

an image sensor (charge coupled device (CCD) or photomultiplier tube (PMT)) with

the help of necessary optical lenses and filters. The intensity of the emitted light is a

measure of the number of target DNA strands that are hybridized on the surface of the

microarray.

Optical DNA sensing instruments are in general bulky and expensive due to the need

for a laser source, optical lenses and filters. Another drawback of optical DNA sensing

is the labeling requirement of the target DNA, which is a costly and time consuming

process. Optical DNA sensing methods are in general slow. This is due to sequen-

tial scanning of the DNA microarrays by the image sensor, which requires mechanical

movement of the microarray or the image sensor as shown in Fig. 1.4. On the other


LASER

PMT

Excitation

lter

Emission

lter

Pin holeBeam

splitter

Mirror

Mirror

Sample

Figure 1.4: A typical scanner for fluorescence-based DNA detection [14].

hand the detection limit of DNA sensing instruments is very low and they can detect

signals down to 10 million hybridized target molecules per centimeter squared [34].

1.2.2 Commercial Optical DNA-Sensing Instruments

The Affymetrix GeneChip Scanner 3000 7G (Fig. 1.5 (a)) is an example of commer-

cially available optical fluorescent-based DNA microarray. Its dimensions are 33 × 46

× 56 cm3, and its weight is 32kg [35].

Agilent offers a variety of fluorescent-based DNA microarray scanners for DNA

sequencing. SureScan Microarray Scanner (Fig. 1.5 (b)) is one such example. Its di-

mensions are 42 × 43 × 67 cm3, and its weight is 56.8kg [36].

Spartan Bioscience produces a 4 kg portable fluorescent PCR-based DNA analyzer

(Fig. 1.5 (c)) that processes 12 samples, typically medical ones, in one hour [37].

There are also hand held lateral flow immunoassay pathogen testers such as the one

from Idetekt [43], but they are limited in their specificity and sensitivity.


(a) (b) (c)

Figure 1.5: Commercially available fluorescence-based DNA microarray scanners: (a)Affymetrix GeneChip Scanner 3000 7G [35], (b) Agilent SureScan Microarray Scanner [36],and (c) Spartan RX microarray scanner [37].

1.2.3 pH-based DNA Sensing

pH-based DNA sensing is a new alternative method to the optical DNA sensing that is

commonly used for DNA sequencing. DNA sequencing is the process of determining

the order of nucleotides (adenine, guanine, cytosine, and thymine) within a DNA strand.

As shown in Fig. 1.6 (a) [41], in this method an array of pH sensors are implemented

on a chip. Next an array of micromachined wells is placed on top of the pH sensors and

a single ssDNA strand probe is placed in each well.

The principle of label-free electrochemical DNA detection using pH sensors is de-

scribed in detail in [5]. In general the ssDNA strand is activated with an enzyme at

one of its end. The enzyme facilitates the bonding of the complementary base pair to

the ssDNA strand. Once the bonding of one base pair is done, the enzyme moves up

the DNA strand by one base pair thus enabling the next location to bond with the next

complimentary base pair molecule.

In a pH-based DNA sequencing microsystem, DNA base pairs (A, T, C or G) are

sequentially washed over the ssDNA. When a base pair bonds to the ssDNA strand, a

H+ ion is released changing the pH of the reagents inside the sensing well in which

the match took place. pH sensors are sensitive to the pH level (concentration of H+


ions) of reagents inside the sensing well and they convert this change in the pH into

an electrical signal as shown in Fig. 1.6(b). No signal is generated if there is no match

between the DNA base pairs and the probe DNA as shown in Fig. 1.6(c). This change

in the electrical signal can be used to sequence the target DNA.

pH sensors are in general implemented as an ion sensitive field effect transistor

(ISFETs) that can be readily fabricated on a CMOS chip with no post processing re-

quirements. This allows for low production cost and scalability of the DNA sensor. Ion

Torrent has reported a pH based DNA sensing CMOS chip with 1024 sensing sites on

a single chip [41]. The drawback of the pH based DNA sensing is large variation in

the electrical sensitivity of the on-chip pH sensors and also the drift in their sensing

properties over time. Extensive digital calibration circuitry is implemented to cancel

out the sensor-to-sensor variations and calibrate a DNA sequencing system over time

for the drift in the pH sensors electrical properties.

1.2.4 Commercial pH-based DNA Sensing Instruments

Ion Torrent offers a 60 kg rapid, genome-scale benchtop sequencing system (Fig. 1.7

(a)) based on pH sensors that produces one Gb of sequence data in a one run of two

hours. The system is capable of processing both short-base and long-base pair DNA

sequences with ranges from 35 to 400 base pair [41].

DNA Electronics is developing a hand-held, USB-connected, DNA sensing microar-

ray based on pH sensors as shown in Fig. 1.7 (b). The DNA microarray contains 40

sensing sites and enables nucleic acid tests (NATs) from sample to detection in under

30 minutes. The system is currently under development.

GeneOnyx offers a tissue box-size DNA analyzer based on pH sensors [42], as

shown in Fig. 1.7 (c). GeneOnyx is primarily used in skin care and is tailored for

recommending skin care solution and cosmetic products based on individual genetic


(a)

(b)

(c)

Figure 1.6: pH-based DNA sensing: (a) system cross-section, (b) complementary DNA se-quence and (c) and non-complementary DNA sequence [41].


(a) (b) (c)

Figure 1.7: Commercially available pH-based DNA-sensing instruments: (a) Ion Torrent [41],(b) DNA Electronics [44], and (b) GeneOnyx [42].

variation.

1.3 Existing Integrated Circuits for DNA Sensing

The size, weight, and price of today’s commercial DNA-sensing and sequencing mi-

crosystems make them impractical for use in point-of-care diagnostics applications. If

nucleic acid detection is to be widely used globally and at the point-of-care, the de-

tection process has to offer high throughput and automated, portable analysis at a low

cost.

A number of lab-on-a-chip systems based on complementary-metal-oxide-semiconductor

(CMOS) integrated circuit technology have been developed that translate nucleic hy-

bridization events to optical [15], pH [97], magnetic [16], and capacitive [17] signals.

These systems typically feature an array of single-stranded nucleic acid probe DNA

sequences that are immobilized on a solid surface. At the time of the capture of com-

plementary target sequences through Watson Crick base pairing, the sensor transduces

a measurable signal. A brief description of the most popular methods is presented next.


1.3.1 CMOS Fluorescent DNA Contact Imaging

Fluorescent DNA on-CMOS contact imaging relies on the same principle as bench-

top optical fluorescence-based DNA sensing as described in Section 1.2. Unlike the

standard optical fluorescence-based DNA sensing, in contact imaging the DNA probes

and targets are placed close to the CMOS image sensor focal plane [55–57]. This elimi-

nates the need for bulky and expensive lenses and mirrors, thus enabling miniaturization

of the DNA detection microsystem. Various on-chip color filters and color-selective

CMOS imagers have also been developed [58–61]. The latter eliminates the need for

optical filters thus further reducing the cost and the size of the DNA sensing microsys-

tem. The only part of the system that can not be integrated into the CMOS chip is the ex-

citation light source [60], which can be bulky and expensive. This method also requires

DNA labeling which is a major drawback compared to electrochemical amperometric

DNA sensing CMOS microsystems described next which can be label-free [58].

1.3.2 CMOS Magnetic Field-Based DNA Sensing

In recent years magnetically labeled biosensors have shown promising results in provid-

ing an alternative solution to the optical instruments used by conventional fluorescence-

based sensors [45–49]. Magnetic biosensors have the potential to achieve high sensitiv-

ity at low cost and low power consumption levels for a variety of applications such as

in-field medical diagnostics, epidemic disease control, biohazard detection and analy-

sis.

In general, magnetic field-based on-CMOS DNA sensing requires tagging of DNA

strands with a magnetic microbead, and also needs a giant magnetoresistance (GMR)

[45] and a bead counter [45, 46]. One major drawback of the traditional magnetic

biosensor is the requirement of externally generated magnetic biasing fields (to imple-

ment the GMR) [46, 47] in conjunction with exotic post-fabrication processes [48, 49].


This significantly increases the system form-factor, increases total power consumption

and cost.

A typical magnetic based DNA sensor operates in the following steps [45]:

1. ssDNA probes are immobilized on a gold electrode surface using gold-thiol chem-

istry.

2. Target DNA strands are extracted from the sample, denatured, and labeled with

the magnetic microbeads.

3. Target DNA is allowed to hybridize with the probes and the sensor spots are then

washed to remove any un-hybridized strands.

4. The sensor spots are subjected to an applied magnetic field (using the GMR), and

the CMOS chip picks up the resulting bead signal [45].

Recently an RF-based on-CMOS magnetic DNA sensors has been proposed to elim-

inate the need for the bulky GMR and the bead counter [25]. The system was imple-

mented in bulk CMOS and provides single bead detection sensitivity without any ex-

ternal magnets. In this method an RF electrical current is passed through the on-chip

inductor generating a magnetic field which polarizes the magnetic particles present in its

vicinity. Probe DNA are deposited on the surface of the on-chip inductor. Hybridization

of the the target DNA (tagged with the magnetic bead) with the probe DNA results in a

change in the inductor magnetic field, thus changing the inductance of the on-chip in-

ductor. A voltage-controlled oscillator (VCO) combined with a frequency-shift sensing

scheme is utilized to measure the change in the inductance value thus quantifying the

number of target DNA that have hybridized. Such a system [25] did not achieve results

comparable to the traditional magnetic field-based DNA sensors due to the complexity

of the RF receiver and sensitivity of the RF receiver to the environmental noise.


1.3.3 CMOS Capacitance-Based DNA Sensing

Another common method for on-CMOS DNA sensing is capacitance-based DNA sens-

ing [17, 21, 50, 51]. This method is a sub-set of the impedance spectroscopy method

[144] described in Section 1.5.2 as it senses the change in the electrode capacitance

CDB due to the DNA hybridization. In general in this method two gold electrodes are

placed in close proximity (1-10µm) [17] resulting in a simple two-plate capacitor with

the electrolyte solution acting as the dielectric. Next ssDNA probes are immobilized

on the surface of one, working, electrode using gold-thiol chemistry. A capacitance

measurement is conducted at this stage to measure the electrode capacitance before

hybridization. Next, target DNA is allowed to hybridize with the probe DNA and the

electrodes are washed after hybridization to remove any un-hybridized DNA strands. It

has been observed that the capacitance CDB decreases when the target DNA molecules

bond with the probe DNA [17]. The change in the capacitance can be mapped to the

target DNA concentration.

There are various methods to measure the change in the electrode capacitance in

CMOS. The most commonly used method is capacitance-to-frequency conversion as

described in [21, 50, 51].

1.4 Amperometric Electrochemical Sensing Principles

As described in Section 1.2, many conventional DNA-based pathogen detection sys-

tems rely on the gold-standard optical signal readout. Such systems are costly, require

bulky optics, typically require PCR for high sensitivity, and are often slow due to the

serial nature of the optical signal readout. DNA analyzers with non-optical parallel elec-

tronic readout, such as those described in Section 1.3, are a promising cost-effective,

highly sensitive and fast alternative to optical methods, often with their own limitations.


Another such method is known as amperometric electrochemical DNA sensing.

Amperometric DNA sensing methods have gained interest in recent years but are not

yet as developed as the optical or pH-based DNA sensing techniques. They have the

potential to reduce the instrumentation cost and provide portable diagnostic platform

for DNA analysis and detection. The main advantages of amperometric sensing versus

gold-standard optical are the elimination of light as the source of excitation and thus of

bulky optical equipment, and the real-time nature of DNA sensing. This provides an

opportunity to design and prototype a portable real-time DNA sensing microsystem for

point-of-care applications.

The polymerase chain reaction (PCR) is commonly utilized to increase the con-

centration of the target DNA in a sample, thus increasing the detection accuracy. As

described in Section 1.1.3, PCR is a procedure in biochemistry to amplify a single or a

few copies of a DNA sequence, generating up to millions of copies of the same DNA

sequence [62, 63]. This method relies on thermal cycling which consists of cycles of

repeated heating and cooling for DNA melting and enzymatic replication of the DNA.

This is generally costly and time consuming.

Signal transduction is typically performed by coupling the target sequence with a

reduction-oxidation (redox) or fluorescent label with a well-defined and easily detected

electrochemical or optical signature, respectively. This is known as label-based sensing

which requires sample labeling and the corresponding cost and time.

Label-free detection of DNA hybridization is also possible, by monitoring the elec-

trical signals, such as current, voltage, impedance, and conductance, at the sensor’s

solid-liquid interface. There are several label-free electrical nucleic acid detection plat-

forms, but most of these techniques rely on off-chip, expensive, and bulky instrumenta-

tion for signal readout and processing, characteristics that make the techniques unsuit-

able for many point-of-need and in-field applications.


Amperometric electrochemical DNA sensors [20, 21, 23–28, 99] have emerged as

a low-cost, high-throughput, and real-time alternative to conventional optical and pH-

based sensory methods. Electrochemical amperometric DNA analysis techniques have

the potential to provide real-time, label-free, PCR-free sensing in portable detection

platforms.

1.4.1 Three-electrode Sensing Configuration

Electrochemical sensing involves a chemical reaction that occurs on the surface of an

electrode in the presence of a potential difference. The chemical reaction results in a

charge transfer which results in a current or a voltage output. The current-mode output

has a wide dynamic range and is commonly used. Sensing methods in which a current

is measured are known as amperometric electrochemical sensing methods [66, 67].

A block diagram of a three-electrode electrochemical amperometric sensing sys-

tem is depicted in Fig. 1.8 [68]. It consists of a working electrode (WE), a reference

electrode (RE), a counter electrode (CE), a waveform generator and a current-to-digital

channel. In this configuration the waveform generator drives the reference electrode.

The working electrode is held at a known potential, VWE by the current-to-digital chan-

nel. In general the voltage difference between the working and reference electrode is set

to VREDOX . This is the voltage at which the chemical under test is reduced or oxidized.

The IWE current generated due to the voltage difference between the working and ref-

erence electrodes is recorded by the current-to-digital channel. If the voltage difference

is set to VREDOX , the resulted current is called IREDOX , which is the current generated

due to the reduction and oxidization of the chemical species.

In applications where the redox current level is high (greater than 1µA), there could

be a significant voltage drop between the working and reference electrode (due to the

finite impedance of the electrolyte solution) if the redox current is to flow from the


DIGITAL

OUTPUTCURRENT-TO

DIGITAL

CHANNEL

RE

RE CE

PROBE

DNA

TARGET

DNA

CE

WE

WAVEFORM

GENERATOR

REV

CEV

WEV

REDOXI

WEV

Figure 1.8: Conceptual view of an electrochemical amperometric sensing microsystem.

reference electrode to the working electrode. As a results of this voltage drop, the ac-

tual voltage in the vicinity of the working electrode will not be the same as the voltage

that was set by the reference electrode. This can significantly affect the redox poten-

tial required by the electrochemical experiment. To solve this issue a third (counter)

electrode is added to the electrochemical cell. In this configuration no current flows

from the reference electrode to the working electrode. The counter electrode provides

the current required to keep the voltage difference between the working and reference

electrodes exactly at the redox potential. In general the counter electrode is not required

for low-current applications (less than 1µA).

In amperometric electrochemical DNA sensing applications the working electrode

is coated with a single-stranded DNA (ssDNA) probe. Binding of the probe ssDNA

with the target ssDNA results in a variation of the working electrode surface properties

such as impedance or surface charge. The variation of the surface properties results in a

change in the recorded value and waveform features of the redox current, thus indicating


CWE

RS

RCT

CDB

+

+

-

-

-

-

-

+

+

+

WERE

RE WE

ELECTROLYTE

SOLUTION

DIFFUSION

LAYER

DOUBLE

LAYER

+

-PROBE

-TARGET

-

-

-

-

-

++

+ +

-REDUCTION

-OXIDATION+

+

-H O2

Figure 1.9: Charge profile of the electrode-electrolyte interface and its equivalent circuit model.

the thermodynamics and kinetics of chemical reactions at the sensory interface. In most

biochemical sensing applications the recorded redox current is in the range of pA to

nA [69–71].

Electrochemical amperometry provides a wide range of concentration measurement

because of its linear behavior. Cyclic voltammetry and impedance spectroscopy are

two common amperometric sensing methods. In general, the reference electrode poten-

tial, VRE , is set to a bidirectional ramp voltage for cyclic voltammetry or to a small-

amplitude sinusoid for impedance spectroscopy.

1.4.2 Electrode-Electrolyte Interface

When an electrode is in the presence of an electrolyte solution containing solvated and

unsolvated anions and cations and water molecules, a layer of charge forms at the

electrode-electrolyte interface in order to maintain the electrical neutrality, as shown

in Fig. 1.9. This layer can be electrically modeled by a set of resistors and capacitors.

A generic R-C biosensor impedance model is also depicted in Fig. 1.9. The counter

electrode is omitted for simplicity. In this model RS represents the electrolyte resistance

between the working and reference electrodes, CWE represents the diffusion layer ca-


pacitance, CDB models the interfacial double-layer capacitance at the WE-electrolyte

interface and RCT models the charge transfer resistance at the WE-electrolyte inter-

face [84]. The double layer consists of solvated positive ions physically separated from

the working electrode surface by a monolayer of water molecules and ions attached to

the working electrode surface. The diffusion layer is the layer beyond the double layer

where the capturing probe molecules are located. The ssDNA probes are negatively

charge, thus attracting the positively charged ions from the electrolyte solution.

1.4.3 Reduction-Oxidation Current

An example of the mechanism of generation of the redox current is also shown in

Fig. 1.9. In the presence of a potential difference between the working electrode and

the reference electrode the electroactive chemical species of the electrolyte solution

are either reduced (gain an electron) or oxidized (lose an electron). The amount of

charge transferred between the electrolyte solution and the working electrode depends

on the surface charge. The surface charge in term depends on the density of the target

molecules that are bonded to the probes. Measuring the current corresponding to the

charge transfer yields a measure of the concentration of the target molecules.

1.4.4 Labeled vs Label-free DNA Sensing

Amperometric electrochemical DNA sensors can be divided into two main categories:

label-based and label-free. In the label-based techniques an electroactive chemical label

is attached to the target DNA to provide a positive electrochemical signal. This signal

can also be derived from a redox indicator that associates with a ssDNA differently than

with a DNA duplex. On the other hand, a label-free DNA sensor does not require any

supporting redox species, indicator molecule or enzymes to provide the electrochemical

signal. This method relies on the oxidation of individual DNA bases or change in the


electrode-electrolyte interface properties such as capacitance and resistance changes

due to DNA hybridization.

1.4.5 Electrode Material

In general an electrochemical cell consists of a working electrode, a counter electrode

and a reference electrode.

A wide range of materials are used to fabricate working electrodes. These include

carbon, polymers and metals as shown in Fig. 1.10. The working electrode material is

chosen such that it provides a polarizable interface when it is in contact with the elec-

trolyte solution [26]. In this work gold (Au) has been chosen as the working electrode

material. Gold has the advantage of being electrochemically inactive and it does not

react with the electrolyte solution under normal operating conditions. Also, gold work-

ing electrodes can easily be fabricated on a CMOS chip, forming on-chip integrated

working electrodes. Another advantage of gold working electrodes is that it enables

straight-forward DNA attachment chemistry [26].

The reference electrode must meet the following criteria [78]:

• Must have a high exchange current density and be non-polarizable.

• The electrode potential must be reproducible.

• The electrode potential drift must be small during the life time of the electrode.

Depending on the application and solution chemistry, different reference electrodes

can be used. Due to its simple construction and the above-mentioned properties, silver

(Ag) coated by a layer of silver chloride (Ag/AgCl) is the most commonly used material

for a reference electrode. A conventional commercially available reference electrode

consists of a silver wire coated with silver chloride that is immersed in a Cl ion solution


such as KCl. The silver wire and the KCl solution are enclosed in a glass tube and

separated from the solution that is under test by an ion exchange membrane and a salt

bridge [79].

Another common type of the reference electrode is the saturated calomel electrode

(SCE), which consists of mercury covered by mercury chloride pates that are in contact

with saturated KCl solution and enclosed in a glass tubing. The advantage of this elec-

trode is its simple construction and stable potential. However, the toxicity of mercury

makes its use hazardous [80].

Another common reference electrode in electro-chemical sensing is the standard

hydrogen electrode. The electrode is constructed with a platinum surface that is covered

with black platinum. The black platinum helps with the reduction of protons and is a

catalyst for the electrode reaction. The platinum is submerged in an acidic solution.

The reference electrode cell is confined in a glass enclosure and is pressure- controlled

to keep the hydrogen pressure fixed and to maintain the electrode’s potential [81, 82].

Due to its non-reactive nature, platinum (Pt) is the most commonly used material

choice for counter electrode. Other nobel metals such as gold can be used as well [26].

In this work Au electrodes are used for the working and counter electrodes and off-

chip commercially available Ag/AgCl electrodes are used as the reference electrode.

In future work we are planning to implement the Ag/AgCl electrode on-chip using

the post-processing method described in [83] applied to the implemented on-chip Al

reference electrode bases.

1.5 Amperometric Electrochemical Sensing Methods

Different amperometric electrochemical sensing methods are distinguished by the wave-

form of the voltage difference between the working and reference electrode. The two


WE

(c)(b)

60µm

(a)

Figure 1.10: Working electrode material examples: (a) diamond [71], (b) polymer [20], and (c)flat gold [70].

OXIDATION

VOLTAGE

REDUCTION

VOLTAGE

VW

E-V

RE

REVERSE

SCAN

FORWARD

SCAN

TIME

Epc

Epa

VWE

I RE

DO

X

-VRE

OXIDATION

PEAK

REDUCTION

PEAK

BACKGROUNDTARGET CHEMICAL

REVERSE

SCAN

FORWARD

SCAN

Epa Epc

(a) (b)

Figure 1.11: Fast-scan cyclic voltammetry principle of operation (a) cyclic redox potential ap-plied between the working and reference electrode, and (b) cyclic voltammogram in the absenceand presence of the target chemical.

most commonly used electrochemical sensing methods are: cyclic voltammetry (CV)

and impedance spectroscopy (IS). Both have to be thermally regulated as the redox cur-

rent is highly dependent on the temperature [19]. In this work all these three methods

are implemented on-chip to perform thermally-regulated label-free DNA detection and

cross-validation.


1.5.1 Fast-Scan Cyclic Voltammetry

In the fast-scan cyclic voltammetry (CV) method, a cyclic ramp potential is intermit-

tently applied between the working and reference electrodes, as shown for one period

in Fig. 1.11(a). The time between scans, which can vary, determines the temporal reso-

lution of the technique. The halt time prevents successive scans from influencing each

other. The cyclic voltammogram shown in Fig. 1.11(b), represents the redox current

versus the applied redox potential and provides chemical information about the sub-

stance under measurement. For example, the location of the reduction and oxidation

peaks acts as a unique chemical identifier for different chemicals. The reduction and

oxidation peak heights change for different chemical concentrations. A background

current is generated due to the transient changes of the applied voltage (Fig. 1.11(b)).

This current occurs mainly because of the charging and discharging of the double layer

capacitance associated with the electrode-electrolyte interface. The background current

is proportional to the scan rate and also to the double layer capacitance.

1.5.2 Amperometric Impedance Spectroscopy

Impedance spectroscopy (IS) is a popular method of quantitative and qualitative moni-

toring of chemical reactions in many biosensors. A wide range of biosensors have been

developed which rely on impedance spectroscopy, including sensors for detection of

enzymes, antibodies and DNA [85, 86].

In the IS method, a small-amplitude perturbing sinusoidal voltage, as shown in

Fig. 1.12(a), is applied between working and reference electrode and the subsequent

current response is measured. The impedance is calculated as the ratio of the applied

voltage to the resulting current. A frequency response plot of a biosensor, as shown

in Fig. 1.12(b), is computed by applying a variable frequency excitation signal and

computing the biosensor impedance at each frequency point. The biosensor frequency


VWE-VRE

TIME FREQUENCY

CHANGE IN CHEMICAL

CONCENTRATION

I RE

DO

X

(a) (b)

Figure 1.12: Impedance spectroscopy principle of operation (a) potential difference betweenworking and reference electrode, and (b) change in redox current frequency response due to thechange in the chemical concentration.

response is directly related to the concentration of the electroactive substance.

1.5.3 Temperature Regulation for DNA Analysis

Temperature reliance is a key constraint in DNA sensing. DNA hybridization is highly

dependent on the temperature [4–6,19], and as a result, temperature plays an important

role in the sensitivity of DNA sensors. It is shown in [7] that the redox current is highly

dependent on the temperature. Generally the redox current drops by 10 percent, given

a 10 degree increase in the temperature above the ambient temperature. A temperature

regulation system for DNA sensing that has on chip-heating, temperature sensing and

regulation circuits, would be of great benefit for both the characterization and sensing

of DNA. Such a system would increase the accuracy of the measurements, improve the

speed and efficiency of the analysis, and also reduce reagent usage and improve the

sensitivity limit of the measurements.


1.5.4 Commercial Amperometric Instruments

Early Warning Inc. offers a beta-version of its amperometric biohazard water analyzer

that employs carbon nanotube DNA sensors to detect E.coli O157:H7, as shown in

Fig. 1.13(a). The analyzer weighs 200 kg and promises to detect up to 25 pathogens

in 10L of water within three hours, with the sensitivity of one bacterium cell per

100mL [38].

Palm Sens Inc offers Palmsens 3, Fig. 1.13(b), which is a portable wireless cur-

rent recording frontend (potentiostat) for electrochemical recording applications [39].

The unit consist of one current-to-digital recording channel and a wireless bluetooth

transmitter. The unit weighs 450g and achieves an 8 hour wireless battery powered

operation. It has an input dynamic range of 100pA to 10mA.

Metrohm Inc offers PSTAT mini [40], Fig. 1.13(c), which is a single channel portable

potentiostat weighting 0.985kg. The unit achieves an input dynamic range of 2nA to

200A and can be directly interfaced to DNA sensing microelectrodes.

The bulky and expensive commercially available instruments described in Sections

1.2.2, 1.2.4 and 1.5.4 generally attain very high sensitivity (0.1nM). Routine quantita-

tive pathogen detection with low-cost, fast (under one hour), highly specific (multiple

strands), and medium to high sensitivity (10µM down to 0.1nM) biosensors in an inte-

grated and portable lab-on-chip platform is a sought-after goal that is not yet commer-

cially available.

1.6 Integrated Circuits for Amperometric DNA Sensing

Several electrochemical DNA detection microsystems utilizing cyclic voltammetry (CV)

have been reported [20, 21, 26, 96, 97, 99]. The design in [99] is a 50-channel pro-

grammable electrochemical biosensor array implemented in a 0.13µm standard CMOS


(a) (b) (c)

Figure 1.13: Commercially available amperometric instruments: (a) Early Warning biohazardwater analyzer [38], (b) Palmsens 3 [39], and (c) PSTAT mini [40].

technology. The microsystem includes flat gold electrodes and analog recording chan-

nels, and utilizes impedance spectroscopy for DNA detection. The implementation

in [96] consists of one recording channel and 24×16 recording electrodes implemented

in a 0.5µm CMOS technology. The design, which consists of a three-electrode regula-

tion loop and an analog recording channel, utilizes CV for DNA detection and analysis.

A 128-channel DNA analysis microsystem implemented in a 0.5µm CMOS technology

is presented in [21]. The microsystem consists of on-chip gold electrodes, a three-

electrode regulation loop, and an in-pixel ADC. The design presented in [20] is imple-

mented in a 0.5µm CMOS technology and consists of 24 recording channels with an

in-channel ADC, 24×24 polymer-functionalized sensing electrodes, and a temperature

sensor. The microsystem utilizes CV for DNA detection and analysis. The design

in [26] presents the first fully-integrated CMOS DNA analysis microsystem, which

consists of 16 recording channels, a three-electrode regulation loop, a flat gold DNA

sensing microelectrode, and an in-channel ADC.

Several DNA analysis SoCs based on impedance spectroscopy have been reported

[84, 130, 131]. The design in [130] is a single-channel impedance extractor based on a

lock-in amplifier that extracts the sensor impedance from 1Hz to 10kHz. The implemen-

tation in [84] is a 100-channel impedance-to-digital converter based on a delta-sigma

modulator capable of extracting sensor impedance from 1mHz to 10kHz at the expense


of a long conversion cycle. A direct conversion receiver without an on-chip ADC [131]

extracts the electrode impedance from 10Hz to 50MHz at the cost of consuming 104mW

of power.

1.7 Main Specifications for Amperometric Electrochem-

ical DNA Sensing Integrated Circuits Design

1.7.1 Accuracy

Amperometric electrochemical DNA sensors use selective binding and interaction be-

tween probe molecules and target molecules to generate a detectable electrical signal

that correlates with the presence or absence of target molecules. The main components

of any electrochemical DNA sensing system are a molecular recognition layer, a sig-

nal generator, and a readout unit. To generate an electrical target-specific signal, first

the target molecule needs to come into proximity with the probe molecule. The target

molecule’s motion is typically dominated by diffusive spreading, which is a probabilis-

tic mass-transfer. In addition, the nature of the chemical bond forming the interaction

between the probe molecules and the analyte solution is also probabilistic, and the result

is more uncertainty (noise) in the biosensor [9–12, 23].

In addition to these processes, the interrogation signal generation circuits and the

readout circuits also add noise to the biosensor. On top of the noise generated by the

binding process and the electronic circuit, the binding of other species to the probe

molecules (non-specific binding) also adds noise to the system. Non-specific binding

is generally less frequent than specific binding if the concentration of both the specific

and non-specific analyte is in the same order of magnitude. If the concentration of non-

specific molecules becomes much larger than the concentration of the target molecules,


then the non-specific binding (noise) can overcome the detectable signal and thus in-

creases the minimum detectable level (MDL) [13]. Based on the biosensor noise, we

can define the MDL, the highest detection level (HDL), and the detection dynamic range

(DR).

In [23], a noise model was developed to capture the biosensor MDL, HDL, DR, and

SNR (signal-to-noise ratio). The model was applied to a simple biosensor that consisted

of a signal generator, an electrode-eletrolyte interface, and a generalized recording cir-

cuit. Fig. 1.14 plots the SNR of the generalized biosensor model vs. the analyte concen-

tration (DNA molecule concentration) in different background solution concentrations.

In an ideal system, increasing the concentration of analyte generates more noise, but, at

the same time, generates larger signals, a process that results in a linear increase of SNR

vs. analyte concentration. In a practical biosensor, this outcome is not possible, since

the probe saturation occurs, and, as a result, SNR drops as the analyte concentration

increases. As shown in Fig. 1.14, the HDL is limited by saturation and the finite value

of the probe density. The maximum SNR of the generalized biosensor model in [21]

is about 55dB, and the limiting factor is probe saturation and biochemical noise. The

overall dynamic range of the system for an SNR of 30dB varies from 84dB to 104dB,

depending on background solution concentration.

Based on Fig. 1.14 results, the conclusion may be drawn that the minimum SNR of

the interrogation signal generation and recording circuitry should be larger than 55dB,

which corresponds to approximately effective number of bits (ENOB) of 9-bits.

The model developed in [23] is only valid for flat gold electrode. The concentration

range that is presented in this plot is similar to the DNA concentration range that was

successfully detected by the CMOS DNA detection microsystem as presented in Section

3.8. The model developed in [23] is not valid for the nanostructured electrodes. Similar

model for the nanostructured electrodes have not been developed in the literature.


Figure 1.14: Biosensor SNR vs target analyte concentration [23].

1.7.2 Dynamic Range

In [26–28, 69, 70, 99], the output redox current level from a typical on-chip working

electrode (10µm to 100µm diameter) in electrochemical DNA-sensing applications is

shown to be in the range of low pA to mid nA. As a result, in this work, the current

range specification of the system is set to cover a range from one pA to 500 nA.

1.7.3 Frequency Range

In electrochemical sensing methods such as cyclic voltammetry and impedance spec-

troscopy, a high-frequency interrogation waveform generates a large background cur-

rent due to the charging and discharging of the double-layer electrode-electrolyte ca-

pacitor. The large background current puts stringent requirements on the accuracy and

dynamic range of the electrochemical recording front end. To relax the accuracy and


dynamic range requirements in general, the interrogation waveform frequency range of

electrochemical sensing systems is limited to a maximum 10 kHz [26].

1.7.4 Non-electrical Design Specification

The ultimate goal in this thesis is design and development of a low-cost semiconductor-

based technology for rapid and sensitive in-field DNA analysis of various DNA strands.

As a result, small form-factor, low-cost, high-throughput, as well as label-free and PCR-

free operation are the most important non-electrical design requirements of the DNA

sensing microsystem.

The size and the cost of the DNA sensing microsystem needs to be minimized to

enable widespread in-field utilization. This would require integration of both DNA

sensing electrodes and circuitry on the same chip. CMOS integrated circuits provide

versatile signal acquisition and processing functionalities at a low cost. The large scale

of integration allows for hundreds of sensors to be placed directly on a chip for simulta-

neous high-throughput sensing. On-chip microsensors are directly suitable for CMOS

surface functionalization yielding high sensitivity and selectivity thus eliminating the

need for the PCR amplification step which is both time consuming and expensive. Di-

rect deposition of sensors on a chip eliminates costly excessive wiring and minimizes

the interference noise. Label-free operation of the microsystem also eliminates the

costly and time-consuming labeling process further reducing the overall cost of the

DNA sensing microsystem.


WERE

ADC

AM

PE

RO

ME

TR

IC

RE

AD

OU

T

WE

CC

V

EXCITATION

WAVEFORM

IREDOX

TIA

CURRENT

ACQUISITION

CIRCUIT OUTV

OUTI

BIOSENSOR-

FUNCTIONALIZED

SURFACE

Figure 1.15: Conceptual view of current acquisition circuits for electrochemical amperometricsensory system.

1.8 Additional Considerations for DNA-sensing Amper-

ometric Integrated Circuits Design

1.8.1 Low-Level Current Acquisition for Amperometry

The use of a transimpedance amplifier (TIA) or a current-conveyer (CC) are the most

common approaches for redox current acquisition [26], as shown in Fig. 1.15.

In the TIA approach, the transimpedance amplifier sets a virtual potential at the

working electrode and at the same time generates an output voltage that is propor-

tional to the redox current. This can be implemented as a resistive feedback configu-

ration [89] or a switched-capacitor circuit [90]. The size of the resistor in the resistive

feedback configuration makes it impractical for massively-parallel array sensing imple-

mentations. Additionally in a resistive feedback configuration the thermal noise can be

injected back into the biosensor. The switched-capacitor TIA configuration is a com-

mon choice for arrayed implementations [99]. A circuit diagram of a conventional

transimpedance amplifier with a captive feedback is shown in Fig. 1.16.

A current conveyer (CC), as shown in Fig. 1.17, is another common compact circuit

for measuring the redox current [91]. The WE is held at a virtual potential. Instead


−

+

IIN

VWE

VOUTCWE

RS

RCT

RE

CDB

VRE

WE

Figure 1.16: Conventional transimpedance amplifier (TIA) interfaced to a biosensor model.

IIN

VWEIOUT

−

+

CWE

RS

RCT

WERECDB

VRE

Figure 1.17: Conventional current conveyer (CC) interfaced to a biosensor model.

of converting the redox current directly to voltage, it is conveyed from WE to a high-

impedance output node and can then be converted to a voltage. A number of current

conveyer designs for amperometric sensing applications have been reported [91–95]. In

general, these designs do not support bidirectional current recording or offer pseudo-

bidirectional current acquisition capability by means of adding a large bias current to

the input so the current becomes unidirectional [69].

1.8.2 Wireless Data Transmission of DNA Results

Wireless communications capabilities is necessary in applications such as at-home health

care, food safety and water quality monitoring where in-field DNA sensing and analysis

are required.

One such example is placement of the DNA analysis microsystem in a water treat-

ment plant to quickly and accurately monitors the water quality and adjust the treatment

methods to disinfect water for municipal, industrial and residential customers. For ex-

ample in Trojan UV 3000PTP waste water solution [31], shown in Fig. 1.18, the waste


Figure 1.18: Trojan UV3000PTP wastewater solution [31].

water is passed thought a set of UV lights that are used for disinfecting the waste water.

The intensity of the UV light in this system is adjusted based on the contamination level

in the water. Currently water samples are taken on daily basis and the samples are sent

to a lab for analysis to determine the effectiveness of the water treatment. Based on the

lab results the UV light level is adjusted to maximize the effectiveness of the disinfec-

tion. The wireless DNA analysis microsystem integrated with a microfluidic chamber

(capable of taking and conditioning the water samples) can be placed inside the waste

water flow. The microsystem can take samples periodically, analysis the contamination

level (by looking for bacterial DNA) and send the results wirelessly to a nearby central

unit. The central unit can transmitter the results over cellular network to the water treat-

ment control center which can adjust the UV light intensity based on the DNA analysis

results.

High number of recording channels and wireless operation of the DNA sensing mi-

crosystem put stringent requirement on the design and choice of the wireless transmitter.

In DNA sensing applications, high number of recording electrodes is required to pro-

vide a sufficient number of probe strands and enough redundancy for accurate results.


For example, if a DNA analysis microsystem consists of 500 recording channels each

outputting data at 10kbps, this results in a 5MB/s data rate requirement for the wireless

transmitter. Also, power consumption of the wireless transmitter should be small to

increase the battery life of the wireless DNA analysis microsystem.

A single channel wireless DNA system-on-chip (SoC) based on a monolithic polysil-

icon nanowire CMOS process is reported in [98]. The design utilizes polysilicon nanowires

as DNA sensors and consists of a single recording channel, an on-chip temperature sen-

sor and a wireless OOK (on-off-keying) transmitter.

In this work an ultra-wideband (UWB) transmitter was chosen for wireless data

transmission. Compared to other wireless transmitter architectures such as phase-locked

loop (PLL) based transmitters [72,73] and open loop voltage controlled oscillator (VCO)

transmitters [74–77], the ultra-wideband transmitter achieves lower power consumption

and integration area for the same data rate as the ultra-wideband protocol allows for a

lower-power all-digital implementation of the transmitter as described in Chapter 3.

1.8.3 DNA Temperature Regulation

A number of CMOS temperature sensing and regulation integrated circuits for chemi-

cal sensing applications have been reported in the literature. These microsystems use

a resistive heating and sensing element to form what is known as a micro-hotplate.

The work in [132] presents a temperature sensing and regulation microsystem for bi-

ological applications that achieves a single fixed temperature of 37 degrees Celsius on

the back side of a CMOS chip, with an off-chip digital proportional-integral-derivative

(PID) controller. The work in [135] presents a 3×3 micro-hotplate array for a protein

sensing application. In this design, the temperature of the chip is regulated up to 45

degrees Celsius, with an off-chip digital PID controller. The work presented in [25]

is a 16-channel, frequency-shift CMOS magnetic biosensor array for a protein sens-


ing application. The design consists of an array of in-channel heaters, with a single

temperature sensor and an off-chip analog PID controller. The design in [136] is a

digital CMOS micro-hotplate array architecture for a gas sensing application. The ar-

ray consists of 3×3 gas sensing sites, and each site consists of a heater, a temperature

sensors, and a delta-sigma modulated PID controller. The micro-hotplate achieves an

operating temperature of up to 350 degrees Celsius. A 40-channel DNA analysis SoC

based on ISFET sensors for rapid point-of-care DNA detection is reported in [97]. The

design includes an in-channel delta-sigma modulated ADC, in-channel temperature-

sensing elements and heaters, and a dedicated ADC for temperature sensing. A fully

digital on-chip PID controller is utilized for temperature regulation. In summary, exist-

ing closed-loop temperature regulation microsystems generally use either off-chip PID

controllers or on-chip digital PID controllers. This requires large integration area and

high power, typically prohibitive for generating accurate 2D temperature profiles on a

CMOS die as needed for arrayed biosensing applications.

1.8.4 On-chip Computation for DNA Analysis: Analog vs. Digital

vs. Mixed-Signal VLSI Multiplication

Multiplication is the most computationally intensive operation required by the impedance

extraction algorithm (Chapter 4) and the temperature regulation algorithm (Chapter 5).

If the multiplication required by these algorithms can be performed within the recording

channel re-using available mixed-signal VLSI circuits, then significant area and power

can be saved. Conventional analog and digital VLSI multiplication techniques are re-

viewed next and our mixed-signal VLSI approach is introduced last in this section.

A number of analog multipliers have been reported in literature [103–112]. Ana-

log multipliers find applications in a variety of analog signal processing circuits such

as adaptive filters and frequency modulators. The basic idea of an analog multiplier is


V (t)1

V (t)2

Nonlinear

Device

V =aV +bV

+cVi i

i

2

3

Nonlinearity

cancellation

scheme

V (t)1

V (t)2

V =O

O

V i

Figure 1.19: Analog multiplier block diagram [103].

Figure 1.20: Analog multiplier circuit schematic.

shown in Fig. 1.19. In an analog multiplier two signals V1(t) and V2(t) are applied to a

nonlinear device. The non-linear device can be characterized by a high-order polyno-

mial function as shown in Fig. 1.19 [103]. This polynomial function generates a number

of terms besides the desirable input product. Next a nonlinearity cancelation circuit is

used to remove the undesired terms and produce the desired V1(t)V2(t) signal.

An example of a fully differential low-power state-of-the-art analog multiplier is

shown in Fig. 1.20 [113]. The input signals are applied to transistors (P1 to P4) and

(N1 to N4) and the products of the input signals are taken at VO1 and VO2. The analog

multiplier achieves a linearity of 4.5 percent for a 200mV input signal while consuming

45µW with a bandwidth of 1GHz. All the transistors in this implementation have the

same size of W/L = 0.8µm/0.35µm. This results in a total area of 40µm2 (including

the routing requirements). Another implementation of a low-power analog multiplier is

presented in [114]. In this work the analog multiplier achieves a linearity of 4.2 percent

for an input range of 120mV while consuming 6.7µW with a bandwidth of 200kHz.

The area and power requirements, the limited input range and linearity degradation


Figure 1.21: The block diagram of the 16×16-bit serial multiplier [128].

of analog multipliers makes them unsuitable for high-throughput sensory applications

with stringent power, area and accuracy requirements.

There are several techniques for digital multiplication implementation [115–117,

126,127]. These vary from basic serial multipliers [125] to advanced parallel multipliers

[120–124]. The serial multiplier implements the multiplication in N cycles for an N ×

N-bit multiplier with minimum hardware requirements. Serial multipliers are based on

the principle of shift and add algorithm [126–128]. An efficient adder design provides

the required energy and delay requirements for serial multipliers. A state-of-the-art

implementation of a serial multiplier is shown in Fig. 1.21. The multiplier fabricated

in 45nm CMOS generates an output every 16 clock cycles and consists of 436 logic

cell. It consumes 550pJ/MMPS (multiplications per second) while operating at 0.5V

supply [128].

Parallel multipliers are commonly used in high-speed applications and require more

hardware compared to serial multipliers in order to achieve the speed requirements. The

most commonly used architectures for parallel multiplication are single cycle parallel

multiplier [128] and two-cycle parallel multiplier [129]. The schematic diagrams for

both these implementations are shown in Figs. 1.22 (a) and (b). The single cycle par-


(a)

(b)

Figure 1.22: (a) The critical path of the 16×16-bit parallel multiplier and (b) the block diagramof the 16×16-bit two-cycle parallel multiplier [128].


allel multiplier fabricated in 45nm CMOS technology generates an output every clock

cycle and requires 1794 logic cells. It consumes 2.5pJ/kMPS while operating at 0.5V

supply [128]. The two-cycle parallel multiplier fabricated in 45nm CMOS technol-

ogy generates an output every clock cycle and requires 1901 logic cells. It consumes

2.6pJ/MMPS while operating at 0.5V supply [128]. Although the digital multipliers

can achieve very low power operation, the area requirements of these multipliers make

them unsuitable for sensory array applications.

In this work the multiplications required by the impedance extraction and the tem-

perature regulation algorithms are performed by an in-channel dual-slope multiplying

ADC (MADC) in the mixed-signal VLSI domain resulting in small area and low power

consumption.

1.9 Thesis Overview

Table 1.1 provides a comparative analysis of existing amperometric biochemical sen-

sory microsystems.

Based on this survey of the current state-of-the-art electrochemical DNA analysis

SoCs, there is currently no wireless DNA analysis SoC capable of performing all key

electrochemical analysis techniques while regulating the test environment temperature.

Integrating a large number of DNA sensing electrodes with on-chip circuits required to

perform various electrochemical analysis techniques and temperature regulation poses

many challenges. The challenges include excessive power consumption, large silicon

area and digital circuits interference with sensitive analog circuits. The work previously

reported in the literature in general lacks sensitive on-CMOS DNA sensing electrodes

capable of performing PCR-free and label-free DNA detection. Also these systems

perform only one electrochemical sensing protocol such as CV or IS and are not capable


of regulating the chip temperature while performing DNA sensing.


Tabl

e1.

1:C

ompa

rativ

eA

naly

sis

ofA

mpe

rom

etri

cB

iom

olec

ular

Sens

ory

Mic

rosy

stem

s

Syst

emIS

SCC

08JS

SC08

JSSC

09IS

SCC

10B

IOC

AS

09T

BIO

CA

S07

1IS

SCC

101

ISSC

C10

[20]

[26]

[84]

[131

][1

35]

[132

][2

5][9

7]Te

chno

logy

(CM

OS)

0.18

µm

0.25

µm

0.5µ

m0.

35µ

m0.

5µm

0.5µ

m0.

13µ

m0.

6µm

Pow

er25

mW

160m

W0.

6mW

84.5

mW

30m

W—

165m

WN

/ASu

pply

Volta

ge5.

0V2.

5V3.

0V3.

3V5V

5.0V

1.2V

3.3V

Chi

pA

rea

11.2

mm

215

mm

22.

25m

m2

4mm

23m

m2

6mm

27.

5mm

225

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m2

Ele

ctro

deC

ount

576

5410

010

09

116

40C

hann

elSe

nsin

gPr

otoc

olC

VC

VIS

ISC

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Mag

netic

CA

Cha

nnel

Cou

nt24

1610

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116

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peof

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ctro

des

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2D2D

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roho

tpla

teM

icro

hotp

late

Mag

netic

2DPo

lym

erG

old

Gol

dG

old

pHPo

wer

N/A

10m

W6µ

W0.

84m

WN

/AN

/AN

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ynam

icR

ange

N/A

60dB

58dB

N/A

Off

-Chi

pO

ff-C

hip

55dB

50dB

Con

vers

ion

Rat

e10

Hz

10kH

z10

kHz

N/A

——

—1H

zSe

nsiti

vity

97pA

240p

A10

kHz

330p

A—

—0.

3Hz

25µ

VE

NO

B11

bits

9bi

ts8

bits

No

AD

C—

—D

irec

tCon

vers

ion

12bi

tsW

avef

orm

Gen

erat

orN

oN

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esY

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pe—

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uare

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uare

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e—

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neW

ave

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eque

ncy

(Per

iod)

——

10kH

z50

MH

z—

—1G

Hz

—Po

wer

——

——

——

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smitt

erN

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oPr

otoc

ol—

——

——

——

—D

ata

Rat

e—

——

——

——

—Po

wer

——

——

——

——

Tem

pera

ture

Sens

ing

Yes

——

—Y

esY

esY

esY

esSe

nsor

Cou

nt1

——

—9

11

40C

ontr

olN

o—

——

PID

PID

PID

PID

——

——

(Ana

log)

(Ana

log)

(Ana

log)

(Dig

ital)

——

——

Off

-Chi

pO

ff-C

hip

Off

-Chi

pO

n-C

hip

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urac

y—

——

—0.

7o

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5o

CSe

nsor

sTy

peD

NA

DN

APr

otei

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NA

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ein

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NA

DN

AO

n-di

eY

esY

esY

esY

esN

o—

Yes

Yes

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el-f

ree

No

Yes

No

No

No

—N

oN

oPC

R-f

ree

No

No

No

No

No

—N

oY

esB

iom

olec

ule

Type

30B

ase

18B

ase

Bila

yerL

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m—

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Poly

mor

phis

ms

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cent

ratio

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N/A

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CV

:cyc

licvo

ltam

met

ry,I

S:im

peda

nce

spec

tros

copy

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Non

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ric

bios

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dfo

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plet

enes

s.


This thesis presents a CMOS microsystem for amperometric electrochemical DNA

sensing and analysis that overcomes these challenges by employing advanced mixed-

signal VLSI computing circuits to efficiently implement computationally intensive al-

gorithms required for the DNA analysis and by reusing on-chip circuits for multiple

functions in order to save both area and power. The presented system is capable of

performing DNA analysis using both cyclic voltammetry and impedance spectroscopy

while regulating the chip temperature using an on-chip temperature regulator. This

make it universally suitable for a wide range of DNA analysis applications. Addition-

ally, in this work, on-chip nano-textured DNA sensing electrodes were fabricated to

enable label-free DNA analysis utilizing potassium-ferrocyanide redox reporter.


D

NA

p

H

CT

AT

PT

AT

×5

4

PID

SR

AM

WE

WE

P

ID

SR

AM

MA

DC

I RE

F

I IN

WAVEFORM

GENERATOR CE

RE

CC

/

TIA

CL

OC

K

G

EN

BIA

S ∑

WAVEFORM

GENERATOR SRAM

HE

AT

ER

PW

M

MU

LT

IPL

ICA

TIO

N

C

OE

FF

ICIE

NT

S

SR

AM

GL

OB

AL

CL

OC

K

GL

OB

AL

B

IAS UWB TX

I RE

F

CH

AN

NE

L

D

NA

×5

4

SR

AM

WE

MA

DC

CE

RE

TIA

CL

OC

K

G

EN

BIA

S ∑

HE

AT

ER

PW

M G

LO

BA

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CL

OC

K

GL

OB

AL

B

IAS UWB TX

I RE

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CH

AN

NE

L

CT

AT

PT

AT

×5

4

PID

SR

AM

P

ID

SR

AM

MA

DC

CC

CL

OC

K

G

EN

BIA

S ∑

HE

AT

ER

PW

M

MU

LT

IPL

ICA

TIO

N

C

OE

FF

ICIE

NT

S

SR

AM

GL

OB

AL

CL

OC

K

GL

OB

AL

B

IAS UWB TX

CH

AN

NE

L

(a)

(c)

(d)

I RE

F

I IN

I RE

F

I IN

(b)

MU

LT

IPL

ICA

TIO

N

CO

EF

FIC

IEN

TS

SR

AM

WAVEFORM

GENERATOR WAVEFORM

GENERATOR SRAM

CL

K1

.2 V

I BIA

SV B

IAS

CL

K1

.2 V

I BIA

SV B

IAS

CL

K1

.2 V

I BIA

SV B

IAS

D

NA

p

H

×5

4

SR

AM

WE

WE

MA

DC

I RE

F

I IN

CE

RE

CC

CL

OC

K

G

EN

BIA

S ∑

HE

AT

ER

PW

M G

LO

BA

L

CL

OC

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GL

OB

AL

B

IAS UWB TX

I RE

F

CH

AN

NE

L

WAVEFORM

GENERATOR WAVEFORM

GENERATOR SRAM

CL

K1

.2 V

I BIA

SV B

IAS

CE

LL

CE

LL

CE

LL

CE

LL

MU

LT

IPL

ICA

TIO

N

C

OE

FF

ICIE

NT

S

SR

AM

Figu

re1.

23:

Reu

seof

the

SoC

arch

itect

ure

reso

urce

s(b

lock

sou

tline

din

bold

are

reus

edin

allm

odes

)(a

)co

mpl

ete

DN

Aan

alys

isSO

C,(

b)cy

clic

volta

mm

etry

confi

gura

tion,

(c)i

mpe

danc

esp

ectr

osco

pyco

nfigu

ratio

nan

d(d

)tem

pera

ture

regu

latio

nco

nfigu

ratio

n.


1.9.1 Complete DNA Analysis SoC

The functional block diagram of the wireless electrochemical DNA analysis SoC is

shown in Fig. 1.23(a) [87]. The microsystem includes a 9×6 array of low-power

low-noise sensory circuit cells, a programmable waveform generator, on-chip SRAM

memories, a shared proportional-integral-derivative (PID) controller for regulating the

chip temperature, a digital pulse-width-modulator (PWM), an all-digital ultra-wideband

transmitter, and on-chip biasing and clock generation circuits. Each sensory cell in-

cludes a current-to-digital channel, an array of DNA sensors, a pH sensor (only for

sample pH level sensing not for indirect DNA sensing), several BJTs to generate the

CTAT and PTAT currents required for temperature sensing and a heater required for

temperature regulation. To enable independent cell programmability, each cell includes

a bias voltage generator, clock generation circuitry, and SRAM memory.

Each current-to-digital channel consists of either a transimpedance amplifier (TIA)

or a current conveyer (CC) and a multiplying ADC (MADC). The selectable TIA or

CC acquires the sensory current at a low impedance. The dual-slope multiplying ADC

multiplies the sensor response with a set of digital coefficients and outputs the corre-

sponding digital word. The digital output of each channel is serialized on the chip and is

wirelessly transmitted, using the all-digital ultra-wide band transmitter [137]. The SoC

can be configured to perform either cyclic voltammetry (Fig. 1.23(b)), or impedance

spectroscopy (Fig. 1.23(c)), or temperature regulation (Fig. 1.23(d)). The SoC reuses

a number of blocks (denoted by bold outlines in Fig. 1.23) in different configurations

resulting in significant saving in the integration area. For completeness the three key

distinct functionalities of the SoC: cyclic voltammetry, impedance spectroscopy, and

temperature regulation, are briefly summarized next in sections 1.9.2-1.9.4 respectively.

The cyclic voltammetry (and concurrently pH sensing) integrated circuit configuration

is then further detailed in Chapter 3 [137]. The impedance spectroscopy integrated cir-


cuits configuration is detailed in Chapter 4 [144]. The temperature regulation integrated

circuits configuration is detailed in Chapter 5.

1.9.2 Cyclic Voltammetry and Sample pH Level Sensing

The functional block diagram of the wireless electrochemical DNA analysis SoC con-

figured for DNA cyclic voltammetry analysis and pH sensing is shown in Fig. 1.23(b).

In this configuration of the SoC, each channel is multiplexed among a bank of DNA sen-

sors and a pH sensor. The sensors are interrogated by the on-chip programmable wave-

form generator that is shared among all channels [137]. The current conveyer front-end

is utilized in this configuration. The digital data representing the stimulation waveform

properties are stored in the on-chip waveform generator SRAM. The current-to-digital

channel quantizes the input current and outputs the corresponding digital word. No

computation is required in this case and the MADC digital multiplication coefficients

are set to one. The chip temperature is held at a constant value by the pre-configured

on-chip PWM controlling the in-channel heating element. The pH sensors are only used

to measure the pH level of the electrolyte solution under test.

1.9.3 Impedance Spectroscopy


figured for impedance spectroscopy is shown in Fig. 1.23(c). In this mode, the SoC uti-

lizes the 54 current-to-digital recording channels to extract the DNA sensor impedance.

The microsystem utilizes the frequency response analysis (FRA) algorithm to extract

real and imaginary components of the biosensor [144]. Two computationally intensive

operations, the multiplication and integration required by the FRA algorithm, are per-

formed by the in-channel MADC in the mixed-signal domain resulting in small area

and power consumption. The waveform generator produces a variable frequency sinu-


soidal interrogation waveform and drives the reference electrode with it. The front-end

transimpedance amplifier acquires DNA sensor response. The MADC multiplies the

biosensor response with a set of digital FRA algorithm coefficients (stored in the multi-

plication coefficient SRAM). Next the MADC accumulates the results over one period

of the interrogation signal (integration), thus extracting the biosensor impedance. In

this mode the channel temperature is also held constant using the pre-configured PWM

and the in-channel heater.

1.9.4 Temperature Regulation


figured for temperature regulation is shown in Fig. 1.23(d). In this mode, the SoC uti-

lizes the 54 current-to-digital recording channels to measure temperature. The on-chip

PID controller is used to regulate the 2D chip temperature profile, on-chip SRAMs are

utilized to store PID and multiplication coefficients. The channel measures temperature

by taking the ratio of a CTAT to PTAT currents. The CTAT current is used as the input

and the PTAT current is used as the reference to the current-to-digital recording chan-

nel. The CTAT and PTAT currents are generated using the in-cell BJTs. The measured

temperature is fed to the on-chip PID controller. The PID controller regulates the 2D

chip temperature profile by modulating the in-cell heaters. Two computationally inten-

sive operations, the multiplication and subtraction required by the PID algorithm, are

performed by the in-channel MADC in the mixed-signal domain, yielding small area

and power consumption.

1.10 Thesis Outline

The rest of this thesis is organized as follows:


• Chapter 2 focuses on the current acquisition front end of the DNA analysis SoC

that is shared among all mode of operations, as shown in Fig. 1.23(a). This

chapter presents design, comparative analysis and experimental results of two

low-noise bidirectional current acquisition circuits for interfacing with electro-

chemical amperometric biosensor arrays: a transimpedance amplifier (TIA) and

a current conveyer (CC).

• Chapter 3 focuses on the cyclic voltammetry DNA sensing capability of the SoC,

as shown in Fig. 1.23(b). This chapter presents a fully integrated 54-channel

wireless label-free cyclic voltammetry DNA analyzer. The microsystem includes

600 nanostructured DNA sensors and 54 pH sensors, and reuses key circuits for

cyclic voltammetry and for pH sensing.

• Chapter 4 focuses on the impedance spectroscopy DNA sensing capability of the

SoC, as shown in Fig. 1.23(c). This chapter presents a 54-channel mixed-signal

CMOS DNA impedance analyzer that utilizes frequency response analysis (FRA)

to extract the real and imaginary impedance components of the biosensor.

• Chapter 5 focuses on the temperature regulation capabilities of the DNA analysis

SoC, as shown in Fig. 1.23(d). This chapter presents a 54-sites mixed-signal

CMOS thermal regulator with a compact mixed-signal PID controller.

• Chapter 6 contains a summary of the original contributions in this thesis and

provides directions for future work.

51

Chapter 2

Current Acquisition Circuits for

Electrochemical Amperometric

Biosensors

We present a study of the utility of the switched-capacitor TIA and CC circuits in ac-

quisition of bidirectional currents in electrochemical amperometric biosensing applica-

tions. Advantages and disadvantages of the TIA and CC circuits in such applications

are described and experimentally confirmed. This work builds on the theoretical noise

analysis presented in [133, 134]. Specifically, we show that the main drawbacks of the

switched-capacitor TIA is the injection of the the switching noise into the sensor work-

ing electrode. The switching noise injected into the working electrode alters the charge

at the working electrode surface, thus degrading the electrochemical recording results

accuracy.

We introduce a solution to this problem, a low-noise bidirectional current conveyer

for small currents. The advantage of this method, compared to a transimpedance ampli-

fier, is that the working electrode is isolated from the clocked circuitry (e.g, switches,

CHAPTER 2. CURRENT ACQUISITION CIRCUITS FOR ELECTROCHEMICAL AMPEROMETRICBIOSENSORS 52

ADC, comparator) thus reducing the effect of charge injection into the working elec-

trode due to high-frequency switching. Generally current conveyers suffer from the

amplifier flicker noise as it is not integrated as in the case of the TIA. Internal OTA

chopper stabilization is utilized to reduce the effect of the flicker noise, the dominant

noise of the amplifier. Full chopping at the input and output of the OTA was not pos-

sible due to the single ended nature of the design. The CC achieves a dynamic range

of 8.6pA to 350nA. Finally, we present a comparative analysis of a bidirectional-input

TIA and a bidirectional CC fabricated in 0.13µM CMOS technology.

This chapter extends on an earlier report of the current conveyer design presented

in [166], and offers a more detailed analysis of the design and additional electrochemi-

cal experimental results characterizing the circuit implementation and comparing the

current conveyer implementation to a transimpedance amplifier implementation for

electrochemical recording. The rest of the chapter is organized as follows. Section II

presents the TIA design and implementation and includes experimentally measured re-

sults. Section III details the circuit implementation of the CC and also includes exper-

imentally measured results. Section IV presents a comparative analysis of the TIA and

CC designs and their utility in electrochemical sensing applications.

2.1 Transimpedance Amplifier (TIA)

As described in Chapter I, most electrochemical biosensing applications require sourc-

ing and sinking the redox current. One way to measure a bidirectional redox current

is to directly convert it to a voltage. The conversion can be done by a resistive or a

capacitive feedback across an OTA, known as a transimpedance amplifier (TIA). To

convert a low-level current in the nano-ampere or pico-ampere range, a very large re-

sistor is required. A capacitive feedback allows to keep the area small. In addition, a


−

+

S

IIN

VWE

CF

VOUTCWE

RS

RCT

RE

CDB

VRE

WE

IINJECTION

Figure 2.1: Chopper-stabilized bidirectional-input transimpedance amplifier (TIA).

capacitive TIA has an averaging behavior and acts as a low-pass filter to remove the

high-frequency noise [90].

2.1.1 Circuit Implementation

The circuit diagram of the presented TIA is shown in Fig. 2.1. The TIA consist of a

chopper-stabilized OTA with a capacitive feedback. The TIA operates in two phases.

In phase one switch S is open. In this phase the OTA feedback ensures that the negative

terminal of the OTA is at the working electrode potential. The input current is integrated

over the feedback capacitor. This is the integrating phase of the TIA. In the next phase,

switch S is closed to discharge the feedback capacitor. This removes any residual charge

on the feedback capacitor at the end of one full cycle.

The OTA has a folded cascode topology as shown in Fig. 2.2. To reduce the effect

of flicker noise, the amplifier utilizes input PMOS devices with a high aspect ratio and

internal chopping. One set of the chopper switches is placed at the input of the OTA.

Another set is placed at the folding nodes as depicted in Fig. 2.2. This significantly

reduces the flicker noise and DC offsets [134] caused by the input pair transistors, M1

and M2, and the NMOS current mirror transistors, M3 and M4. The size of the feedback

capacitor is 10pF. The chopper clock frequency can be set up to 20kHz. This is lim-


VWE

VbiasP

VbiasN

VcascP

cascNV

IIN

IOUT

M1M

2

M3 M

4

M5

M6

M7

M8

M9 M

10M

11

Figure 2.2: Circuit schematic diagram of the OTA within the TIA.

ited by the bandwidth of the OTA. The chopper frequency should be set high enough

such that the low frequency noise that is up converted is placed beyond the OTA 3dB

frequency (to be filtered out). Also the chopping frequency should be below the OTA

3dB bandwidth so that the OTA provides adequate gain (speed) required for the noise

up conversion at the chopping frequency [52,53]. The chopping frequency is in general

optimized by circuit level simulations.

Correlated double sampling is another circuit technique to reduce the OTA input

offset and flicker noise [54]. This method was not utilized in this work as the leakage

from the extra required switches would have reduced the overall system accuracy.

The simulated input-referred noise of the integrator for the cases where the chopper

is disabled and enabled is shown in Fig. 2.3. The integrated input-referred noise from

0.01Hz to 1kHz is 0.12pA when the chopper is disabled and 0.07pA when the chopper

is enabled. In this design the internal chopping yield a noise improvement of -5dB

at 10Hz as shown in Fig. 2.3. The integrator transistor sizes are shown in Table I.

Contribution of each transistor to the total input-referred noise is shown in Fig. 2.4.

When the chopper is disabled the main contributions are from the OTA current mirror


10−2

10−1

100

101

102

103

−130

−120

−110

−100

−90

−80

−70

FREQUENCY (Hz)

CHOPPER ON

CHOPPER OFF

INPUT NOISE ( A / Hz ) (dB)

rms

1/2

Figure 2.3: Simulated input-referred noise spectrum of the TIA from 0.01Hz to 1kHz.

transistors M3 and M4 and the input pair transistors M1 and M2. When the chopper

is enabled, the current mirror transistors M9 and M10 are the main contributors to the

input-referred noise.

2.1.2 Experimental Results

The transimpedance amplifier was fabricated in a 0.13µm CMOS process with a 1.2V

supply and occupies an area of 80µm×60µm.

The experimentally measured relative errors of the digital output for the input cur-

rent swept between ±10pA and ±350nA are shown in Fig. 2.5. The relative error stays

below 5 percent over the whole operating range. The TIA achieves a dynamic range of

10pA to 350nA. The lower limit is defined by the LSB of the on-chip ADC. The higher

limit is defined by the input current that saturates the TIA.

Figs. 2.6(a),(b) show the experimentally recorded output current distribution for

the input currents of 100pA and 100nA, respectively, measured from 16 channels on

16 chips (one channel per chip). The mean output current and the corresponding stan-

dard deviation are 80.89pA and 6.9pA, respectively, for the input current of 100pA.


M 3,4

M 1,2

M 9,10

M 5,6

M 7,8

TO

TA

L N

OIS

E (

%)

0

10

20

30

TRANSISTOR

CHOPPER ON

CHOPPER OFF

(a)

TO

TA

L N

OIS

E (

%)

5

0

10

15

20

25

3,4 1,2 9,10 5,6M M M M M

7,8

TRANSISTOR

CHOPPER ON

CHOPPER OFF

(b)

Figure 2.4: Transimpedance amplifier noise summary: (a) flicker noise contributions, and (b)thermal noise contributions.

They are 100.10nA and 17.8pA for input current of 100nA. Table III summarizes the

experimentally measured characteristics of the TIA.

2.2 Current Conveyer (CC)

Another common method to acquire a bidirectional current is to use a unidirectional

current conveyer and to add a DC offset current to its input as shown in Fig. 2.7 [7-

12]. This requires high resolution in the subsequent ADC and adds noise to the redox

current, given that the current mirror generating the DC currents directly contributes

thermal and flicker noise to the output noise of the current conveyer. Also, depending

on the WE impedance, a portion of the DC offset current, IERROR, can leak into the

electrochemical cell and disturb the charge balance on the WE-electrolyte interface.


-4

-2

0

2

4

6

10p 100p 1n 100n 1µINPUT CURRENT (A)

10nRE

LA

TIV

E E

RR

OR

(%

)

(a)

-1µ -100n -10n -1n -100p -10pINPUT CURRENT (nA)

-2

0

2

4

6

-4RE

LA

TIV

E E

RR

OR

(%

)

(b)

Figure 2.5: Experimentally measured relative error of the output of the TIA for the input currentof (a) 10pA to 350nA, and (b) -350nA to -10pA. The results are measured from one typicalchannel on one chip.

2.2.1 Circuit Implementation

The top-level VLSI architecture of the presented bidirectional current conveyer is shown

in Fig. 2.8. It is comprised of a PMOS and an NMOS transistors Mn and Mp connected

in the feedback of the OTA. The negative feedback ensures a known potential at the

working electrode is set by the voltage at the non-inverting input of the OTA. It also

enables the current conveyer to source and sink input current without the need for a

DC offset current. The currents through Mn and Mp are mirrored to the output of the

current conveyer.

Internal OTA chopping has been utilized to reduce the effect of its flicker noise. The

current mirrors are implemented in a low-current regulated cascode topology to enable

accurate current copying down to the pA level. To facilitate a comparative analysis with


CURRENT (pA)0 20 40 60 80 100 120 140

0

1

2

3

4

5

6MEAN= 80.89pA

SD(3σ) = 6.9pA

N= 16

NU

MB

ER

OF

OC

CU

RA

NC

ES

(a)

CURRENT (nA)100.00 100.05 100.10 100.15 100.20 100.25100.30012

3

45

67

MEAN= 100.10nA

SD(3σ) = 17.8pA

N= 16

NU

MB

ER

OF

OC

CU

RA

NC

ES

(b)

Figure 2.6: Experimentally measured TIA output current of 16 channels (from 16 chips, onechannel each) for the input current of (a) 100pA, and (b) 100nA.

the TIA presented in section IV, the output current is integrated on the load capacitor

CF of 10pF and sampled.

The schematic diagram of the current conveyer is shown in Fig. 2.9. The OTA is

implemented as a folded-cascode amplifier and the current mirrors are implemented as

a low-current regulated cascode current mirror comprised of the transistors M12 to M20

and M21 to M29 [11]. The regulated cascode current mirrors ensure high precision and

Table 2.1: TIA OTA Transistors Sizing

Transistor W/L (µm)M1,2 8× 1/0.4M3,4 1× 0.5/5M5,6 4× 0.5/4M7,8 8× 0.5/4M9,10 2× 0.5/0.5M11 4× 1/4MS 4× 1/0.13


IIN

VWE

Mp

IOUT

IDC

CWE

RS

RCT

WERECDB

VRE

IERROR

−

+

Figure 2.7: Conventional pseudo-bidirectional current conveyer with a DC offset current.

IN

I OUTVWE

Mn

Mp

CcI

CWE

RS

RCT

WERECDB

VREV OUT

S

CF

−

+

Figure 2.8: Chopper-stabilized bidirectional current conveyer VLSI architecture.

high output impedance of the current conveyer. The regulated current mirrors monitor

IIN through M13 and M22. The input current is used to bias the regulating amplifiers

consisting of transistors M14, M16, M17, M19, M20 and M23, M25, M26, M28, M29. The

regulating amplifies sense the drain voltage of M15 and M24 through transistors M14

and M26 and adjust the gate voltage of M18 and M27 such that the drain-source voltage

of M12-M15 pair and M21-M24 pair are equal pairwise thus ensuring accurate current

copying down to pA level.

In more detail, for the NMOS section of the regulated cascode current mirror, tran-

sistors M14, M17 and M20 form a common-source amplifier. The common-source am-

plifier forms a negative feedback loop with the help of source follower transistor M18.

This ensures a high impedance at the output of the current mirror. Vds of M15 should

be equal to Vds of M12 to ensure accurate current copying. This is achieved as follows:

the common-source amplifier is biased with the current IIN using a (1:1) current mirror

consisting of transistors M19 and M20. The sizes of transistors M14 and M13 are equal

to those of the transistors M12 and M15 respectively. Transistors M16 and M17 are sized


VWE

VbiasP

VbiasN

VcascP

cascNVIIN

IOUTCc

M1 M2

M3 M4

M5 M6

M7 M8

M9 M10M11

M12M13 M14 M15

M16M17

M18

M19 M20

M21 M22 M23 M24

M25

M26

M27

M28 M29

M

Mn

p

Figure 2.9: Chopper-stabilized bidirectional current conveyer circuit schematic diagram.

such that they provide enough voltage drop to ensure that transistors M12, M13, M14,

M15 have all the same Vds. The value of the capacitor Cc is set to 0.5pF to ensure OTA

stability over the operating current range.

Similarly to the TIA, internal OTA chopping has been implemented to reduce the

effect of both flicker noise and the input offset voltage. As shown in Fig. 2.9, chopper

switches are placed at the input of the OTA. Another set is placed after the NMOS tail

current source. This significantly reduces the flicker noise and offsets due to the input

pair transistors and the NMOS tail current source transistors. Minimum size switches

are utilized to reduce the effect of charge injection into the working electrode. The

current conveyer transistor sizes are shown in Table IV.

In typical electrochemical sensing applications, the current conveyer operates in the

frequency range of 0.01Hz to 1kHz. To achieve efficient flicker noise reduction, the

chopper frequency needs to be higher than the input signal frequency. The chopper

clock frequency was set to 10kHz. As a result the current conveyer bandwidth should

be higher compared to the case where no chopper stabilization is utilized. Simulated

bandwidth of the current conveyer is shown in Fig. 2.10. The current conveyer achieves


102

103

104

105

−5

−4

−3

−2

−1

0

13dB = 35.7 kHz

CURRENT GAIN (dB)

FREQUENCY (Hz)

Figure 2.10: Simulated current conveyer AC response

a 3dB bandwidth of 35.7kHz.

2.2.2 Channel Noise Analysis

An important consideration in the design of the current conveyer is its intrinsic noise

as it limits the sensitivity of the recording channel. The output noise originates from

the OTA, the feedback transistors Mp, Mn and the regulated cascode output current

mirror as shown in Fig. 3.6. The sub-circuit in Fig. 2.11 (active during the positive

current recording) includes these elements and is used for noise analysis to simplify the

derivation.

The impedance between the reference and working electrodes is modeled with a

generic R-C biosensor impedance model shown in Fig. 2.11, left. In this model RS

(typical value of 1MΩ) represents the electrolyte resistance between the working and

reference electrodes, CWE (typical value of 500pF) represents the diffusion layer ca-

pacitance, and CDB (typical value of 300pF) models the interfacial double-layer capac-

itance at the WE-electrolyte interface and RCT (typical value of 1GΩ) models the charge

transfer resistance at the WE-electrolyte interface [84]. The input-referred noise of the

OTA is due to its thermal and flicker noise. Both noise sources can be referred to the

positive input of the OTA and are modeled as the voltage source Vn in Fig. 2.11 [100].

According to the simplified model given in Fig. 2.11, the output noise power of the


OUT

M12M13

M14

M15

M16M17

M18

M19M20

Mp

IN

VWE

I

WE

Inp

In12

In16

In19

In13 In14

In17

In20In18

In15

I

CWE

RS

RCT

RECDB

VRE Vn

−

+

Figure 2.11: Simplified noise model of the chopper-stabilized current conveyer.

current conveyer is given by

I2n,OUT = β1

(∣∣∣∣ gmpA

1 + gmpAZ

∣∣∣∣2 V 2n +

∣∣∣∣ 1

1 + gmpAZ

∣∣∣∣2 I2np)

+ I2n,18 + I2n,15 + g2m18

(β2(

∣∣∣∣ gmpA

1 + gmpAZ

∣∣∣∣2 V 2n

+

∣∣∣∣ 1

1 + gmpAZ

∣∣∣∣2 I2n) + I2n,13,14,16,17,18,19,20

), (2.1)

where V 2n is the OTA input-referred noise, I2np is the noise due to the feedback transistor

Mp, gmp is the transconductance of the PMOS feedback transistor Mp, A is the open-

loop gain of the OTA and Z is the equivalent impedance of the electrode. β1 and β2 are

the current mirror ratios between the transistors M12, M13 and M12, M15, respectively.

In this design both ratios are set to one.

For practical values of gmp, A and Z, the contribution of I2np in equation (1) is

negligible, resulting in the total output current noise


I2n,OUT ≈ β1

(∣∣∣∣ 1Z∣∣∣∣2 V 2

n

)+ I2n,18 + I2n,15

+ g2m18

(β2(

∣∣∣∣ 1Z∣∣∣∣2 V 2

n )

+ I2n,13,14,16,17,18,19,20

). (2.2)

Since the electrolyte resistance, RS , is typically small, and the charge transfer resis-

tance, RCT , is very large, the electrode equivalent impedance is approximately capaci-

tive

Z ≈ 1

jω(CWE + CDB), (2.3)

where CWE represents the diffusion layer capacitance and CDB models the interfacial

double-layer capacitance at the WE-electrolyte interface as shown in Fig. 2.11. Substi-

tuting equation (3) into (2) results in

I2n,OUT ≈ β1

(|2πf(CWE + CDB)|2 V 2

n

)+ I2n,18 + I2n,15

+ g2m18

(β2(|2πf(CWE + CDB)|2 V 2

n )

+ I2n,13,14,16,17,18,19,20

). (2.4)

According to equation (4), the output noise level is proportional to the input-referred

noise of the OTA as shown in Fig. 2.11, the current noise contributions from the reg-

ulated cascode current mirror transistors and the electrode capacitance. The electrode

capacitance is determined by the electrode surface area. In general the larger the area

of the electrode the more biomolecules there are to undergo a redox reaction resulting


in a larger input signal. As a result, the electrode size and thus capacitance do not affect

the SNR of the electrochemical recording system significantly.

The input-referred noise of the OTA consists of two components, the thermal noise

power and the 1/f noise. The thermal noise component can be expressed as [100]

V 2n,thermal

∆f=

4KT

gm1

(4

3

)(1 +

gm3

gm1

+gm9

gm1

), (2.5)

where K is Boltzmann constant and T is the absolute temperature. The noise contribu-

tion of M3,4,9,10 is minimized by biasing M3,4,9,10 such that gm3,4,9,10<<gm1,2. The key

to minimize the gm ratios is to bias the input differential pair transistors in the weak

inversion region, where the transconductance efficiency, gm/ID, is high, and to bias

M3,4,9,10 in the strong inversion region to lower their gm.

The input-referred 1/f noise power can be expressed as [100]

V 2n,1/f

∆f=

2KP

COXW1L1f+

2KN

COXW3L3f

(gm3

gm1

)2

+

2KP

COXW9L9f

(gm9

gm1

)2

, (2.6)

where COX is gate oxide capacitance per unit area. In order to minimize the 1/f noise,

large PMOS input-pair transistors, long-channel current mirrors and internal OTA chop-

ping are employed.

The noise contribution of the output regulated cascode current mirror is mostly gov-

erned by the transistors transconductance. The channel noise of a transistor is given

by [100]

I2n = 4KT

(2

3

)gm +

K

WLCOXfgm. (2.7)


Thus, the gm of the current mirror transistors should be reduced and the length of the

transistor should be increased in order to reduce the current mirror noise contribution

to the output of the current conveyer.

The simulated input-referred noise of the current conveyer for the cases where the

chopper is disabled and enabled is shown in Fig. 2.12. The electrode model shown in

Fig. 2.11 was included in the noise simulation to model the effect of the voltage ripple

and the leakage current [155]. In this simulation RS was set to 1MΩ, CWE was set to

500pF, CDB was set to 300pF and RCT was set to 1GΩ. This takes into account the

effect of the high sensor output impedance on the noise performance of the chopper.

The integrated input-referred noise from 0.01Hz to 1kHz is 0.27pA for the case when

the chopper is disabled and is 0.13pA when the chopper is enabled.

The chopper implementation yields an 8dB improvement in the noise floor with

the realistic high-impedance electrode model. Off-chip access to the analog output of

the current conveyer is not available, and as a result, the noise can not be measured

directly before the quantization noise is added. The dual-slope ADC integrates the

output current of the current conveyer in every ADC conversion cycle. This integration

significantly reduces the effect of the ripples caused by chopper switches on the output

of the current recording channel. Also, given that the chopper is implemented inside

the OTA (internal OTA chopping) the limited bandwidth of the OTA combined with the

2pF integrating capacitor, act as a low-pass filter thus further reducing the ripple levels.

The contribution of each transistor to the total input-referred noise is shown in

Fig. 2.13. When the chopper is disabled the main contributions are from the OTA

current mirror transistors M3,4 and the input pair transistors M1,2. When the chopper

is enabled, the current mirror transistors M12,21,24,15 are the main contributors to the

input-referred noise.


10−2

10−1

100

101

102

103

−140

−120

−100

−80

FREQUENCY (Hz)

CHOPPER ONCHOPPER OFF

INPUT NOISE ( V / Hz ) (dB)

rms

1/2

Figure 2.12: Simulated input-referred noise spectrum of the current conveyer from 0.01Hz to1kHz.

Table 2.2: Current Conveyer Transistor Sizing

Transistor W/L (µm) Transistor W/L (µm)M1,2 8× 3/0.4 M16,17,18 1× 0.5/4M3,4 1× 0.5/5 M19,20 8× 0.3/5M5,6 4× 0.5/4 M21,22,23,24 2× 0.4/8M7,8 8× 0.5/4 M25,26,27 4× 0.7/3M9,10 2× 0.5/0.5 M28,29 4× 0.3/4M11 4× 1/4 Mp 1× 1/1M12,13,14,15 1× 0.4/8 Mn 1× 0.4/4

2.2.3 Experimental Results

The current conveyer was fabricated in a 0.13µm CMOS process with a 1.2V supply

and occupies an area of 100µm×100µm.

The experimentally measured relative errors (absolute error divided by the magni-

tude of the exact value) of the digital output for the input current swept between ±10pA

and ±350nA are shown in Fig. 2.14. The relative error stays below 8 percent over the

whole operating range. The current conveyer achieves a dynamic range of 8.6pA to

350nA. The lower limit is defined by the ADC LSB and the higher limit is defined by

the input current that saturates the current conveyer. Fig. 2.15 shows the experimen-

tally recorded output current distribution for the input currents of 100pA and 100nA

measured from 16 channels on 16 chips (one channel per chip). The mean output cur-

rent and the corresponding standard deviation are 81.19pA and 20.31pA, respectively,


CHOPPER ON

CHOPPER OFF

M 3,4

M 1,2

M 12

M 15

M 21

M 24

0

0.005

0.010

0.015

0.020

0.025

INT

EG

RA

TE

D I

NP

UT

NO

ISE

( p

A )

0.035

(a)

CHOPPER ON

CHOPPER OFF

M 3,4

M 1,2

M 21

M 12

0

0.005

0.010

0.015

0.020

0.025

0.030

INT

EG

RA

TE

D I

NP

UT

NO

ISE

( p

A )

(b)

Figure 2.13: Current conveyer noise summary: (a) flicker noise contributions, and (b) thermalnoise contributions.

for the input current of 100pA. They are 100.21nA and 29.0pA for the input current of

100nA. Table III summarizes the experimentally measured characteristics of the current

conveyer.

2.3 Comparative Analysis

The electrical characteristics of TIA and CC are compared first. When the chopper

is enabled, the TIA achieves and input-referred noise of 0.07pArms and CC achieves

an 0.13pArms. This is due to the fact that the TIA integrates noise over one sampling


-8

-4

0

4

6

8

10p 100p 1n 100n 1µINPUT CURRENT (A)

10n

RE

LA

TIV

E E

RR

OR

(%

)

(a)

-1µ -100n -10n -1n -100p -10pINPUT CURRENT (nA)

0

2

4

6

8

−2RE

LA

TIV

E E

RR

OR

(%

)

(b)

Figure 2.14: Experimentally measured relative error of the output of the current conveyer forthe input current of (a) 10pA to 350nA, and (b) -350nA to -10pA. The results are measured fromone typical channel on one chip.

period.

The chip-to-chip output current variation of TIA, in Fig. 2.6, is lower compared

to that of the CC design shown in Fig. 2.12. The variation in CC is mainly due to

the mismatch in the output current mirrors. The maximum relative error of the output

over the operating current range is 5 percent for the TIA and 8 percent for the CC.

Mismatch in the regulated cascode current sources limits the linearity of the current

conveyer. The mismatch in the regulated cascode current mirrors is due to the mismatch

in the transistor pairs M12-M15 and M21-M24, as shown in Fig. 2.9. Dynamic element

matching (DEM) can be employed to reduce the effect of the mismatch in the current

mirrors [166].

To study the effect of charge injection into the working electrode, two sets of simu-


CURRENT (pA)

NU

MB

ER

OF

OC

CU

RA

NC

ES

0 20 40 60 80 100 120 14001

2

3

4

5

6MEAN= 81.19pA

SD(3σ) = 20.31pA

N= 16

(a)

CURRENT (nA)100.15 100.2 100.25 100.3 100.35 100.401

234

5

67

MEAN= 100.21nA

SD(3σ) = 29pA

N=16

NU

MB

ER

OF

OC

CU

RA

NC

ES

100.5

(b)

Figure 2.15: Experimentally measured current conveyer output current of 16 channels (from 16chips, one channel each) for the input current of (a) 100pA and (b) 100nA.

lations were performed with the working electrode model in both Fig . 2.1 and Fig. 2.8

connected to the voltage VRE . The average current integration into the working elec-

trode, due to current integration and sampling, is calculated for the cases where the

sampling frequency is varied from 1kHz to 12kHz and the chopping frequency is set to

20kHz. As it can be seen from Figs. 2.16(a) and (b), the average current injected into

the working electrode at 6kHz for the case of the TIA is significantly higher compared

to the average current injected by the CC.

To compare the performance of the TIA and CC in electrochemical sensing appli-

cations, two sets of CV scans of a DNA reporter, potassium ferricyanide, have been

performed. Potassium ferricyanide K3[Fe(CN)6] is commonly used in electrochemi-

cal sensing systems as a redox reporter [87]. Cyclic voltammetry recordings of 2µM

potassium ferricyanide in a 1M potassium phosphate buffer (pH 7.3) have been carried

out. A 100mV/sec 0.7V peak-to-peak CV waveform with 50ms resting period was ap-


0 2 4 6 8 10 12

INP

UT

CU

RR

EN

T (

fA)

130

180

230

280

330

SAMPLING FREQUENCY (kHz)

(a)

0 2 4 6 8 10 12

INP

UT

CU

RR

EN

T (

fA)

20

40

60

80

100

SAMPLING FREQUENCY (kHz)

(b)

Figure 2.16: Average current injected into the working electrode for (a) TIA and (b) CC.

−700 −600 −500 −400 −300 −200 −100 0−12

−10

−8

−6

−4

−2

0

2

4

6

8

V -V (mV)

CU

RR

EN

T (

nA

)

OXIDATION

VOLTAGE

RE WE

REDUCTION

VOLTAGE

TIA

CC

Figure 2.17: Cyclic voltammogram of 2µM potassium ferricyanide in 1M potassium phosphatebuffer solution experimentally recorded with the transimpedance amplifier (TIA) and the currentconveyer (CC) using a 50µm×50µm gold electrode.


Table 2.3: Experimentally Measured Transimpedance Amplifier (TIA) and Current Conveyer(CC) Characteristics

TIA CCTechnology 0.13µm CMOS 0.13µm CMOSSupply Voltage 1.2V 1.2VArea 80µm×60µm 100µm×100µmSensitivity 135fA 8.6pAMax Relative Error 5.1 8.0(10pA to 350nA)Max Relative Error 4.8 7.8(10pA to 350nA)Input Referred Noise 0.07pA 0.13pA(0.01Hz to 1kHz)Charge Injection at 6kHz 221fA 53fA(Simulated)Power Consumption

Core Circuit 1µW 2µWBiasing 1µW 1µWDigital 1µW 1µWTotal 3µW 4µW

plied between a 55µm×55µm on-chip gold working electrode and an off-chip Ag-AgCl

reference electrode. The resulting CV curves recorded by the chopper-stabilized TIA

and the chopper-stabilized CC are shown in Fig. 2.17. The CV curves for both TIA

and CC show two distinct peaks at the reduction and oxidation voltages of potassium

ferricyanide at -250mV and -450mV respectively. The measurements match as well.

Next the same set of CV recordings have been conducted using a 2µm×2µm on-chip

gold working electrode. The resulting CV curves recorded by the chopper-stabilized

TIA and the chopper-stabilized CC are shown in Fig. 2.18. The CV curve for the CC

shows two distinct peaks at the reduction and oxidation voltages of potassium ferri-

cyanide, as expected. The CV curve for the TIA shows no reduction or oxidation peaks.

In the TIA case the switching charge from switch S is injected into the working elec-

trode. This disturbs the charge balance at the electrode-electrolyte interface thus affect-

ing the electrochemical reaction required for reduction and oxidation of the potassium

ferricyanide. As expected, this effect is more pronounced for smaller electrode size,

given that the amount of the charged injected into the working electrode is comparable

to the double layer capacitance charge.


−700 −600 −500 −400 −300 −200 −100 0

−0.6

−0.4

−0.2

0.0

0.4

0.6

0.8

1.0

V -V (mV)

CU

RR

EN

T (

nA

)

OXIDATION

VOLTAGE

RE WE

REDUCTION

VOLTAGE

TIA

CC

0.2

−0.8

Figure 2.18: Cyclic voltammogram of 2µM potassium ferricyanide in 1M potassium phosphatebuffer solution experimentally recorded with the transimpedance amplifier (TIA) and the currentconveyer (CC) using a 2µm×2µm gold electrode.

Both circuits inject small amount of charge generated by the chopper switches into

the biosensor. This injected noise is negligibly small compared to the switching noise

due to the feedback switch in the TIA and is due to the chopper switches mismatch.

2.4 Chapter Summary

Designs of two low-noise chopper-stabilized bidirectional current acquisition circuits

for electrochemical sensing applications have been presented. The first design has a

switched-capacitor transimpedance amplifier topology. The second one has a current

conveyer topology. Both designs are implemented in a 0.13µm CMOS technology.

Electrical and electrochemical performance of both design has been characterized. The

TIA and CC consume 3µW and 4µW from a 1.2V supply, respectively. It is shown

that the TIA marginally outperforms the CC for high-amplitude input currents. For

small input currents corresponding to low concentration of biochemicals the CC is the

preferred choice as it better isolates the working electrode from current injection.

73

Chapter 3

Cyclic Voltammetry and pH sensing

A fully integrated 54-channel wireless fast-scan cyclic voltammetry DNA analysis SoC

is presented. The microsystem includes 546 3D nanostructured and 54 2D gold DNA

sensing microelectrodes as well as 54 pH sensors. Each channel consists of a chopper-

stabilized current conveyer with dynamic element matching. It is utilized as the am-

perometric readout circuit with a linear resolution from 8.6pA to 350nA. The on-chip

programmable waveform generator provides a wide range of user-controlled rate and

amplitude parameters with a maximum scan range of 1.2V, and scan rate ranging be-

tween 0.1mV/sec to 300V/sec. A digital ultra-wideband transmitter based on a de-

lay line architecture provides wireless data communication with data rates of up to 50

Mb/sec while consuming 400µW. The 3mm×3mm prototype fabricated in a 0.13µm

standard CMOS technology has been validated in prostate cancer synthetic DNA detec-

tion with 10aM label-free PCR-free detection limit. Each channel occupies an area of

only 0.06mm2 and consumes 42µW of power from a 1.2V supply.

CHAPTER 3. CYCLIC VOLTAMMETRY AND PH SENSING 74

3.1 Introduction

We have reported in Nature Nanotechnology amperometric electrochemical sensors

fabricated on passive silicon, not on CMOS, that do not require cumbersome tagging of

DNA with chemical or optical labels [138]. These gold microelectrodes have fine-tuned

nanostructured patterns on their surface that yield an over 140dB input dynamic range

and 10aM detection limit sufficient for PCR-free DNA detection.

In this paper, we present a 0.13µm CMOS DNA analysis SoC with 600 such nanos-

tructured microelectrodes (NMA) grown directly on the die. This paper extends on

an earlier report of the principle and demonstration in [87], and offers a more detailed

analysis of the design and additional experimental results characterizing the circuit im-

plementation and the DNA detection performance. This SoC performs label-free PCR-

free DNA analysis using fast-scan cyclic voltammetry with a 10aM detection limit and

pH sensing for cancer detection. The microsystem consists of a fully programmable

arbitrary waveform generator with an on-chip memory and 54 chopper-stabilized cur-

rent recording channels. The chopper-stabilized current conveyer frontend, with an

input-referred noise of 0.13pArms over one kHz bandwidth, is utilized as the ampero-

metric readout circuit in each channel. The current conveyer achieves linear resolution

from 10pA to 400nA. A chopper-stabilized dual-slope ADC is utilized to digitize the

recorded current. The waveform generator provides stimulation waveforms with a max-

imum scan range of 1.1V and a scan rate ranging from 0.1mV/sec to 300V/sec. A fully

digital 10Mb/s ultra-wideband (UWB) transmitter performs wireless communication.

The rest of this paper is organized as follows. Section II provides background on

DNA detection principles. Section III describes the process of fabrication of the nanos-

tructured DNA sensing microelectrodes. Section IV presents the DNA analysis SoC

VLSI architecture. Section V details the circuit implementation of the VLSI archi-

tecture. Section VI demonstrates the electrical experimental results obtained from the


PR

OB

E D

NA

TAR

GE

T D

NA

K [Fe(CN) ] 6

K [Fe(CN) ] 6

e e

e

e e

e

e

e

e

e

e e

e

e

e

e

e

e

e

e

e

BA

RE

GO

LD

I RE

DO

X1

I RE

DO

X1

I RE

DO

X2

<

4 3

(a) (b)

Au WE

e e

Au WE

e

Au WE

(c)

VRE

I RE

DO

X1

-VWE

VRE

I RE

DO

X2

-VWE V

RE

I RE

DO

X3

-VWE

Ag/AgCl RE Ag/AgCl RE Ag/AgCl RE

I RE

DO

X2

I RE

DO

X3

<

REDUCTION OXIDATION

Figure 3.1: Label-free electrochemical DNA detection principle. (a) Bare electrode: maximumcharge transfer between working and reference electrode in the absence of negatively chargedprobe and target DNA; (b) non-complementary target DNA: reduction in the charge transfer ratedue to the presence of negatively charged probe DNA, and (c) complementary DNA: furtherreduction in the charge transfer rate due to the presence of negatively charged target and probeDNA.

0.13µm CMOS prototype. In Section VII, the results of on-chip electrochemical record-

ing of calibration chemicals are presented. In Section VIII, the results of on-chip CV

recording of a synthetic DNA marker in prostate cancer screening are presented.

3.2 DNA Detection Principle

The principle of the label-free DNA detection method based on potassium ferricyanide

reporter is shown in Fig. 3.1. Potassium ferricyanide K4[Fe(CN)6] is a negatively

charged redox complex with a well-defined electrochemical signature exhibiting oxi-

dation and reduction currents at VRE-VWE voltage of -450mV and -250mV, respec-

tively. Maximum electron transfer between the bare gold electrode and potassium fer-

ricyanide is achieved in the absence of both the DNA target and probe, as denoted by


IREDOX1, in Fig. 3.1(a). Electron transfer is decreased when a negatively charged self-

assembled monolayer of probe DNA (SSDNA) is deposited on the electrode, as shown

in Fig. 3.1(b). This corresponds to smaller redox current IREDOX2, which results in

relatively smaller reduction or oxidization peaks. Upon bonding of the probe DNA and

target DNA (if present) the resulting DSDNA is more negatively charged and causes

potassium ferricyanide to be repelled farther from the electrode surface reducing the

generated faradaic current, as shown in Fig. 3.1(c). The redox current IREDOX3 is sig-

nificantly smaller compared to the first two cases and lacks the reduction and oxidation

peaks. In other words, the presence of negatively charged DNA on the biosensor sur-

face translates to a decrease in the potassium ferricyanide oxidation/reduction current

creating a detectible signal change [140–142].

3.3 Integrated Sensors

3.3.1 DNA Sensing Microelectrodes

To improve the sensitivity and dynamic range of the DNA sensor, nanostructured micro-

electrodes (NMEs) [138] are grown on the CMOS aluminum working electrode base,

using a combination of electroless plating and electroplating techniques.

It is shown in [138] that nanostructuring the working electrode allows for fabri-

cation of DNA sensors on passive silicon that have a broad range of sensitivities and

dynamic ranges. Highly branched electrodes with fine nanostructuring are capable of

achieving a 10aM detection limit [138]. It is postulated that the DNA probes which are

functionalized on nanostructured electrodes are more accessible and, as a result, bond

much easier and faster with target molecules. Microelectrodes with different degrees

of nanostructuring result in different sensitivities and dynamic ranges. By placing an

array of different electrodes on the same CMOS chip the sensor system can achieve a


sensitivity of two to six orders of magnitude [138].

In this design DNA sensing working electrodes are created by forming 2µm×2µm

passivation openings on the top metal layer (aluminum) of the CMOS chip (as it is

commonly done for bond pads) as shown in Figs. 3.2(a) and (b). An electroless metal

plating technique is employed to sequentially deposit nickel (Ni), palladium (Pd) and

then flat gold (Au) base on the exposed Al surface to form an electrode foundation as

shown in Fig. 3.3(c). Next NMEs are grown electrostatically in a solution containing

69µL of gold solution (544385-10G Aldrich) diluted in 2.5mL of deionized (DI) water

2.5mL of 5µM HCl [138] as also shown in Fig. 3.3(c). The shape and the size (defin-

ing the sensitivity and dynamic range) of the NMEs depend on the potential difference

between the working electrode and the reference electrode and the duration of the elec-

troplating. Examples of NMEs grown on a CMOS chip for 60sec at 100mV, 0 and

-100mV voltage difference between an on-chip Au working electrode and an off-chip

(Ag/AgCl) reference electrode are shown in Figs. 3.3(d), (e) and (f), respectively. Two

examples of arrays of NMEs grown on a CMOS chip are also shown in Fig. 3.3(f),

middle and right. For comparison purposes, large flat (2D) working electrodes have

also been fabricated on-CMOS. These flat gold electrodes are fabricated using the same

electroless plating technique as that used for the NME foundation fabrication, as shown

in Fig. 3.2(a). For example, the SEM photographs of such a gold-plated 55µm×55µm

on-CMOS flat working electrode are shown in Figs. 3.2(b) and (c).

3.3.2 pH Sensors

The in-channel ion-sensitive-field-effect-transistor (ISFET) based pH sensor is imple-

mented by a floating gate PMOS with the size of 0.5µM×0.35µM. The poly-gate

of the PMOS is connected to the top metal layer to form a floating gate electrode,

and the CMOS passivation layer (Si3N4 and SiO2) is used as the pH-sensitive mem-


(a)

(b) (c)

(a)

(c)

FLAT Au WE

25µm 1µm

FLAT Au WE

1.0μm0.2μmPd

NiAl

Au

SiO2

Si N43

POLYAMIDE

Figure 3.2: (a) Passivation opening in standard CMOS and added metal layers of a flat (2D)microelectrode after electroless nickel-palladium-gold plating, (b) and (c) SEM photographs ofsuch 55µm×55µm working electrodes.

brane [101]. It is shown in [143] that the passivation layer (exposed section where

there is no polyamide) gives a linear pH response with a sensitivity of approximately

56mV/pH [101], depending on the stoichiometry of the passivation layer. The 54 pH

sensors are directly interfaced to the 54 current-recording channels. The source of the

PMOS is connected to the VDD (1.2V), and the drain is connected to the input of a

current conveyer. The pH sensor gate voltage is set by the on-chip reference electrode.

In this configuration, both the Vgs and the Vds of the pH sensor PMOS transistor are

fixed. Any change in the pH level effectively changes the PMOS threshold voltage.

This change results in a corresponding change in the drain current, which is digitized

by the recording channel.


5µm

5µm

30µm

VRE

-V = -100 mVWE

RE-V = 0 mV

WE

VRE

-V = -100 mVWE

555555µµµµµmmmmm3µm

6µm1µm

Al BASE

VRE

-V = 100 mVWE

VR

mV RE

-V

= 100 mV

WE

VRE

-V = 100 mVWE

(b)

V

(d)

(e)

(f)

VRE -V = -100 m

V

WE

VRE

-V = 0mVWE

5µm7µm

VRE

-V = 100 mVWE

GROWN @

GROWN @

GROWN @

1µm

Au BASE

PdNiAl

1.0μm0.2μmSiO2

Si N43

POLYAMIDE

~5.0μmAu WE

(c)

Al

SiO2

Si N43

POLYAMIDE

~2.0μm

(a)

VRE

-V = -100 mVWE

40µm

(d)

(e)

VRE

-V = 0 mVWE

5µm

RE-V = 0 mV

WEV

5µm

Figure 3.3: Nanostructured DNA sensing working electrodes (NMEs): (a) Cross-sectional viewof a 2µm×2µm passivation opening in standard CMOS, (b) SEM photograph of a 2µm×2µmworking electrode passivation opening over an aluminum base, (c) nanostructured 2µm×2µmworking electrode grown on the passivation opening over an aluminum base in standard CMOS,(c) (d) and (e) SEM photographs of nanostructured microelectrodes grown at different elec-trodeposition conditions on the passivation opening in (b).

3.4 VLSI Architecture

3.4.1 Top-Level VLSI Architecture

The top-level VLSI architecture of the wireless DNA analysis SoC is shown in Fig. 3.4.

The SoC consists of 54 current-to-digital recording channels. Each channel is multi-

plexed between a bank of DNA sensors and a pH sensor.

The sensors are interrogated by the on-chip arbitrary waveform generator that is

shared among all channels. The arbitrary waveform generator consists of a 8-bit R-2R

DAC, an 8-bit up-down counter and a 3-electrode-configuration RE voltage regulation

circuit [144]. The waveform generator provides stimulation waveforms with a maxi-

mum scan range of 1.1V and the scan rate ranging from 0.1mV/sec to 300V/sec. It


DNA

PH

×54

SRAM

WE

WE IREF

IIN

WA

VE

FO

RM

GE

NE

RA

TO

R

CE

RE

CLK 1.2 VIBIAS VBIAS

CCII

CLOCK

BIAS

GLOBAL

CLOCK

GLOBAL

BIAS

U

WB

TX

IREF

CHANNEL

DUAL-

SLOPE

ADC

TIMING

COEFFICIENTS

SRAM

WAVEFORM

GENERATOR SRAM

Figure 3.4: Wireless DNA analysis microsystem functional block diagram.

consumes 900µA from a 1.2V supply when driving a 5nF load at the maximum scan

rate of 300V/sec. This maximum rate is not required for the DNA sensing applica-

tion as the scan rate is limited to low 100s of mV/sec. Other amperometric biochemical

sensing applications (such as, for example, neurotransmitter sensing [70]) require much

higher scan rates of up to 300V/sec. The microsystem presented here is designed so that

it can also be used in applications other than DNA sensing. As a result, the waveform

generator is designed such that it meets requirements for a general purpose biochemical

sensing microsystem but with the power scaling with the frequency. The digital data

representing the stimulation waveform properties are stored in the on-chip waveform

generator SRAM (Fig. 3.4).

A current conveyer is placed at the frontend of each channel to acquire the result-

ing sensory current at a low impedance. A dual-slope ADC quantizes the input redox

current and outputs a corresponding digital word. The digital output of each channel is

serialized on the chip and is wirelessly transmitted at a data rate of up to 10Mbps, using

an all-digital ultra-wide band transmitter. To enable independent channel programma-

bility, each channel also includes a bias voltage generation circuit, a clock generation

circuit, and an in-channel SRAM for setting the channel dynamic range and sensitivity.


−

+

S1

S2

S2

VWE

CONTROL

LOGIC

IREF+

IREF-

CINT

VOUT2

(S1,S2)

−

+

VWE

CHOPPER

STABILIZATION

VLATCH

S2

S1

DIGITAL

OUTPUT

−

+

CURRENT CONVEYER DUAL-SLOPE ADC

WEs

VWE

Mn

Mp

Cc

DYNAMIC ELEMENT

MATCHING

−

+ IOUT

IIN

9-BIT

COUNTER

CHOPPER CLK

GENERATOR

−

+

Figure 3.5: Simplified top-level VLSI architecture of one chopper-stabilized integrated current-to-digital channel.

VWE

VbiasP

VbiasN

VcascP

cascNVIIN

IOUTCc

M1 M2

M3 M4

M5 M6

M7 M8

M9 M10M11

M12M13 M14 M15

M16M17

M18

M19 M20

M21 M22 M23 M24

M25

M26

M27

M28 M29

M

Mn

p

I-to-V

I-to-V

Figure 3.6: Detailed implementation of the current conveyer OTA with internal chopping anddynamically-matched low-current regulation.

3.4.2 Channel VLSI Architecture

The top-level VLSI architecture of one current-to-digital channel of the integrated elec-

trochemical sensory microsystem is shown in Fig. 3.5. Each channel consists of a

chopper-stabilized bidirectional current conveyer (Fig. 3.5, left) and a 9-bit dual-slope

ADC (Fig. 3.5, right).

The current conveyer buffers the input current and maintains the working electrode

at a fixed potential, VWE , as needed to induce a redox reaction. DNA analysis applica-

tions require both sourcing and sinking the redox current. A number of current conveyer

designs for electrochemical sensing applications have been reported [91–94]. In gen-


eral, existing designs do not support bidirectional current recording and suffer from the

amplifier flicker noise and the mismatch within current mirrors. Fig. 3.5(left) depicts a

low-noise and accurate current conveyer VLSI architecture that overcomes these lim-

itations. Internal OTA chopper stabilization is utilized to reduce the effect of flicker

noise. The current conveyer utilizes low-current regulated-cascode current mirrors to

record small (i.e., as small as 10pA) bidirectional currents. Dynamic element matching

is utilized to improve the accuracy by averaging the mismatch in the current mirrors.

The current conveyer is comprised of a PMOS and an NMOS transistors Mn and

Mp connected in the feedback of the chopper-stabilized OTA. The negative feedback

ensures a known potential, VWE , at the working electrode is set by the voltage at the

negative terminal of the OTA. It also enables the current conveyer to source and sink

input current without the need for a DC offset current [92], which can disturb the DNA

charge balance. The currents through Mn and Mp are mirrored by dynamically-matched

current mirrors to the output of the current conveyer and are added.

Based on previously published results of DNA hybridization experiments on NME

working electrodes [138], it is determined that the on-chip ADCs must be able to dig-

itize bidirectional current in the 10pA to 100nA range or greater, and to cover a fre-

quency range of 0.01Hz to at least 1kHz. The dual-slope ADC architecture is selected

for this purpose because its dynamic range, sampling frequency, and nominal resolution

suit these requirements and can all be easily adjusted.

The dual-slope ADC shown in Fig. 3.5(right) consists of an integrating on-chip

variable capacitor CINT (adjustable from 1pF to 10pF, all the measurements here are

done using a 2pF capacitor value), regulated-cascode current sources IREF+ and IREF−,

a four-stage track-and-latch comparator, a 9-bit digital counter and control logic. All

switches are implemented as low-leakage switches as shown in an inset in Fig. 3.5.

The reference current sources are implemented as regulated-cascode current mirrors


to ensure accurate current sourcing over the operating dynamic range. The IREF+ is

implemented with PMOS devices and the IREF− is implemented with NMOS devices.

This can result in some mismatch between the IREF+ and IREF−. The effect of the

mismatch between the positive and negative current source does affect the linearity of

the ADC. These effects are within the specification and are reflected in the measured

spectrum of the ADC output and its ENOB presented in Section VI. The first stage of

the comparator is chopper-stabilized to reduce the effect of its offset and low-frequency

noise.

The dual-slope ADC operates in two phases. In phase one, the integrating capac-

itor CINT is charged by the input current IIN for a predetermined period of time T1.

Next, during the second phase of the operation, the capacitor is discharged to zero

by a DC reference current IREF (IREF+ or IREF−). By counting the duration of the

second phase, the time T2, a digital representation of IIN can thus be obtained as -

sign(IREF )×(T2/T1)×|IREF |. In this design the value of the IREF is programmable

(using an off-chip variable resistor) between 100pA to 50nA.

The in-channel SRAM can also be used to adjust the duration of the charging and

discharging cycles of the dual-slope ADC for the purpose of channel gain calibration.

For example if IREF is higher in the first channel compared to the second channel, then

the duration of the charging time T2 can be reduced for the first channel to compensate

for larger IREF and thus generating the same output digital code for both channels for a

given input current. This effectively calibrates each channel independently and reduces

the channel-to-channel gain mismatch.


VLATCH

V

OUT+V

OUT-

VLATCHVLATCH

VBIAS

M1

M2

M3

M4 M

5M

6 M7

M9 M10

M8

M11 M12

M13M14 M15 M16

VIN+

VIN-

−

+

STAGE2 STAGE3

−

+

Figure 3.7: High-speed latched comparator circuit schematic diagram.

3.5 Circuit Implementation

3.5.1 Current Conveyer

As shown in Fig. 3.6 the OTA has been implemented as a folded-cascode amplifier

to provide a wide input dynamic range and a high gain. In this design, internal OTA

chopping has been implemented to reduce the effect of both flicker noise and the input

offset voltage. A set of chopper switches are placed at the input of the OTA. Another

set is placed after the NMOS tail current source. This significantly reduces the flicker

noise and offsets due to the input pair transistors and the NMOS tail current source

transistors. Minimum size switches are utilized to reduce the effect of charge injec-

tion into the working electrode. The output current mirrors are implemented using a

low-current regulated cascode topology. The regulated current mirrors M12, M15, M18

and M21, M24, M27 replicate IIN (fed through the NMOS and PMOS transistors) at the

output node with a high output impedance. The I-to-V blocks consisting of transistors

M13,14,16,17,19,20 and M22,23,25,26,28,29 adjust the gate voltage of the NMOS and PMOS

output cascode transistors M18 and M27 such that the drain-source voltages of the cur-

rent mirror transistor pairs M12, M15 and M21, M24 are pairwise equal thus ensuring

accurate current copying down to the pA level.

Mismatch in these regulated cascode current sources can significantly reduce the

linearity of the current conveyer. Dynamic element matching (DEM) [145] is employed


Table 3.1: ADC Comparator Transistor Sizing

Transistor W/L (µm) Transistor W/L (µm)

M1,2 24/1 M9,10 3/0.4M3 2/4 M11,12 36/0.5M4,5 4/1 M13,14 6/4M6,7 8/1 M15,16 36/0.5M8 4/4 — —

to reduce the effect of the mismatch in the current mirrors. The main source of mis-

match in the regulated cascode current mirrors is due to the mismatch in the transistor

pairs M12, M15 and M21, M24. To reduce the effect of the mismatch between these tran-

sistors, the DEM technique is applied by means of the chopper switches at the drains

of the current source transistors, so that the critical transistor pairs are dynamically

matched. In this method, the locations of the transistors M12, M15 and M21, M24 are

swapped periodically, at 500Hz, effectively averaging the current mirrors mismatch.

Ideally the error due to the mismatch in the current mirrors is reduced with a higher

DEM switching frequency which results in better averaging over one ADC conversion

cycle. Due to the non-ideality of the switches, an increase in the switching frequency

results in high-frequency switching noise and an increase in the charge injected into the

current path. This in turn causes an error at the output of the current conveyer. Based

on these considerations the 500Hz DEM frequency was chosen.

To achieve efficient flicker noise reduction, the chopper frequency needs to be higher

(at least twice) than the input signal maximum frequency (1kHz). The chopper clock

frequency was set to 10kHz to place the switching noise well outside the operating

frequency range. As a result the current conveyer bandwidth should be higher compared

to the case where no chopper stabilization is utilized so that the output settles in each

switching period. The current conveyer 3dB bandwidth is 35.7kHz.


V6

V5

V4

V3

V2

V1

OUTV

D

C

C

C

CBP

M

ON-OFF KEYING (OOK)

MANCHESTER MODULATION

D IN

OUTVINV

PV

NV

(a) (b)

Figure 3.8: (a) Ultra-wideband transmitter circuit schematic diagram, and (b) schematic of onecurrent-starved inverter.

V6

V5

V4

V3

V2

V1

OUTV

DM

Figure 3.9: Timing diagram of the ultra-wideband transmitter.


1Х6 BANK OF CROSS-VALIDATION CHANNELS

UWB TX

WAVEFORM

GENERATOR & SRAMCONTROL

LOGIC

9×6 ARRAY OF

CHANNELSRECE

WE

2X2μm2

55X55μm

ISFET

5X5μm2X2μm

2

2

2ONE CHANNEL

8Х8 ARRAY OF

2Х2μm WE2

ARRAY OF

WEs

Figure 3.10: Die micrograph of the 3mm×3mm 54-channel wireless DNA analysis SoC. TheSoC was fabricated in a 0.13µm standard CMOS technology.


3.5.2 Dual-slope ADC Comparator

The ADC comparator is implemented with three stages of pre-amplifiers, with a total

gain of 60dB and the last stage with a high-speed latch as shown in Fig. 3.7. The first

stage of the comparator is implemented as a cross-coupled diode-connected gain stage.

This topology provides a moderate gain and a high frequency bandwidth. Chopper-

stabilization suppresses the input offset and ensures 9-bit accuracy. The second and

third stages are identical to the first one but with no chopping. The high-speed latch is

implemented with an NMOS input pair gain stage and a NMOS-PMOS cross-coupled

load. This topology provides high accuracy, low offset and a high frequency bandwidth.

The comparator transistor sizes are listed in Table 4.1.

3.5.3 Ultra-wideband Transmitter

The circuit diagram of the all-digital pulsed UWB transmitter is shown in Fig. 3.8(a).

The input data are modulated using on-off keying (OOK) Manchester modulation.

UWB pulses are generated on the rising edge of the modulated data (DM ). A delay line

bank is employed together with a capacitively coupled output combiner [150] as shown

in Fig. 3.8(a). The modulated data are passed through a delay line, and a delayed ver-

sion of the data are passed through three pulse generators. The pulse generators shape

a first-order Gaussian pulse at the rising edge of the input data. The presented digital

UWB transmitter achieves both power efficiency and spectral compliance in a much

smaller chip area compared to earlier designs [151, 152].

As illustrated in Fig. 3.9, each pulse generator forms pulses that are delayed, and

have opposite signs. By capacitively combining the three paths, the opposite signs

are canceled, and the zero-DC double-differentiated Gaussian pulse propagates to the

single-ended antenna [151, 152]. The width of the output pulse depends on the delays

in the delay line. The delay cells in all the paths are implemented as current-starved


0 5 10 15 20 25 30 35 40 45 50−140

−120

−100

−80

−60

−40

−20

0

2nd HARMONIC

3rd HARMONIC

SFDR = 59 dB

NORMALIZED POWER (dB)

SNR = 57 dB

FREQUENCY (Hz)

I = 350nAin

ELECTROCHEMICAL RECORDING

CHANNEL OUTPUT SPECTRUM

Figure 3.11: Experimentally measured spectrum of the electrochemical recording channel out-put for a 15Hz sinusoidal full-scale (350nA) input.

inverters, shown in Fig. 3.8(b), to allow for tuning of the UWB pulse width.

3.6 Electrical Experimental Results

The fabricated prototype die micrograph is depicted in Fig. 3.10. The 54 channels are

arranged in a 9×6 array on a 3mm×3mm 0.13µm CMOS die. Two channel types with

two different WE aluminum base configurations are implemented. A set of 48 channels

of the first type scan 4 WEs each, in order to perform initial detection of DNA. They

have three different WE aluminum base sizes of 2µm×2µm (twice), 5µm×5µm and

55µm×55µm each as needed to cover a wide combined dynamic range. An additional

set of 6 channels of the second type (at the bottom of the array in Fig. 3.10) additionally

scan a sub-array of 8×8 2µm×2µm WEs each. These redundant-electrode sub-arrays

are utilized for DNA detection results cross-validation and for titer DNA concentration

measurements.


100

101

102

103

104

SNDR=48dB LSB=8.6pA

f = 23 kHzs

SNDR=53dBLSB=132pA

SNDR=59dB LSB=1.05nA

f = 2 kHzs

10p 100p 1n 10n 100n 1μ

DIG

ITA

L O

UT

PU

T C

OD

E

f = 11kHzs

INPUT CURRENT (A)

CHANNEL

TRANSFER

CHARACTERISTIC

Figure 3.12: Experimentally measured transfer characteristics of the current-to-digital channelfor three sampling frequencies.

Dynamic performance of the entire channel was measured by applying a 15Hz full-

scale (350nA) sinusoidal input current sampled at 23kHz. Fig. 3.11 shows the 65536-

point FFT of the measured ADC output. The strong second harmonic is due to the

single-ended nature of the architecture of the ADC. The resulting effective number of

bits (ENOB) is 9.1.

For static performance characterization the input DC current of one typical channel

was swept between 10pA and 350nA as shown in Fig. 3.12. The input dynamic range

is 93dB cumulatively for the three sampling frequency settings, or 48dB at one fixed

sampling frequency of 2kHz. The dynamic range for each setting is computed by taking

the ratio of the maximum signal that saturates the ADC to the LSB for a given sampling

frequency setting.

Two sets of ENOB measurements were conducted to study the effectiveness of the

in-channel gain calibration using the in-channel SRAM to adjust the ADC timing. In

the first measurement no calibration has been performed and the timing parameters


ENOB (BITS)

NU

MB

ER

OF

OC

CU

RA

NC

ES

8.2 8.4 8.6 8.8 9.0 9.2 9.4 9.60

2

4

6

8

10

12

MEAN= 9.01 BITS

SD(3σ) = 0.307 BITS

N= 32

(a)

ENOB (BITS)

NU

MB

ER

OF

OC

CU

RA

NC

ES

8.2 8.4 8.6 8.8 9.0 9.2 9.4 9.60

2

4

6

8

10

12

MEAN= 9.15 BITS

SD(3σ) = 0.252 BITS

N= 32

(b)

Figure 3.13: Experimentally measured output ENOB of 32 channels (from 16 chips, two chan-nels each) for a 15Hz 350nA sinusoidal input (a) without calibration (b) with in-channel cali-bration.

of all channels are set to a constant value (all the ADCs have the same charging and

discharging phases duration). Fig. 3.13(a) shows the experimentally recorded ENOB

for a 15Hz full-scale (350nA) sinusoidal input current from 32 channels on 16 chips

(two channels per chip), with the ADC clocked at 12MHz. The mean ENOB and the

corresponding standard deviation are 9.01 and 0.307 respectively. Next, the same set

of experiments were repeated with the calibrated channels, as described at the end of

Section IV.B. Fig. 3.13(b) shows the experimentally recorded ENOB for the same input

tone as the pervious case. The mean ENOB and the corresponding standard deviation

are 9.15 and 0.252 respectively. The calibration improves the ENOB standard deviation

by 17 percent.

Fig. 3.14 shows the ADC ENOB versus the frequency for a full-scale (350nA) si-


101

102

103

104

5

6

7

8

9

10

EN

OB

(B

ITS

)

FREQUENCY (Hz)

I = 350nAin

Figure 3.14: Experimentally measured ENOB vs. input frequency for the in-channel ADC.

nusoidal input current. The ADC maintains an ENOB of greater than 8.5 bits at up to

3.4kHz. The drop in the ENOB is due to the limited bandwidth of the frontend current

conveyer and high-frequency switching interference noise.

The experimentally measured relative errors of the digital output for the input cur-

rent swept between ±10pA and ±350nA are shown in Fig. 3.15. The relative error stays

below 6 percent over the whole operating range. This is an improvement of 33 percent

compared to the design without the DEM [155]. Fig. 3.15 illustrates an improvement

in the output relative error of approximately 25 percent due to the use of DEM in this

design as compared to a previously reported design without DEM [155]. The current

conveyer achieves a dynamic range of 8.6pA to 350nA or 93dB. The lower limit is de-

fined by the ADC LSB and the higher limit is defined by the input current that saturates

the current conveyer.

Fig. 3.16 shows the experimentally recorded output current distribution for the input

current of 100pA measured from 32 channels on 16 chips (two channels per chip) with-

out dynamic element matching [155] and with dynamic element matching implemented

in this design. The mean output current and the corresponding standard deviation with-

out dynamic element matching [155] are 81.26pA and 20.2pA, respectively. In this de-


−4

-2

0

2

4

6

10p 100p 1n 100n

INPUT CURRENT (A)

10n

RE

LA

TIV

E E

RR

OR

(%

)

(a)

-100n -10n -1n -100p -10p

INPUT CURRENT (nA)

−2

0

2

4

6

−4

RE

LA

TIV

E E

RR

OR

(%

)

(b)

Figure 3.15: Experimentally measured relative error of the output digital code of the currentconveyer connected with the dual-slope ADC for (a) 10pA to 350nA and (b) -350nA to -10pAinput current.

sign, with dynamic element matching added, they are 92.12pA and 9.2pA, respectively.

Adding DEM results in a 54 percent improvement in channel-to-channel accuracy.

As shown in Fig. 3.17 the same experiment is repeated with the input current level

of 100nA. The mean output current and the corresponding standard deviation without

dynamic element matching [155] are 100.26nA and 34pA, respectively. In this design,

with dynamic element matching added to the design, they are 100.18nA and 22pA,

respectively. Adding DEM results in a 35 percent improvement in channel-to-channel

accuracy.

The input Manchester-encoded data to the UWB transmitter and its measured output


CURRENT (pA)

NU

MB

ER

OF

OC

CU

RA

NC

ES

0 20 40 60 80 100 120 1400

2

4

6

8

10

12

MEAN= 81.26pA

SD(3σ) = 20.2pA

N= 32

(a)

CURRENT (pA)

NU

MB

ER

OF

OC

CU

RA

NC

ES

0 20 40 60 80 100 120 140

0

2

4

6

8

10

12

MEAN= 92.12pA

SD(3σ) = 9.2pA

N= 32

(b)

Figure 3.16: Experimentally measured output current of 32 channels (from 16 chips, two chan-nels each) for the input current of (a) 100pA without DEM [32] and (b) 100pA with DEM (thiswork).

UWB pulses are shown in Fig. 3.18. The UWB pulses are measured using custom-built

UWB antennas (5cm spacing between the transmitter and receiver) and an custom-built

receiver. A zoomed-in version one such the measured UWB pulse overlayed on a simu-

lated UWB pulse is shown in Fig. 3.19. As it can be seen the measured pulse resembles

the expected UWB pulse but includes minor ringing due to the package bondwire in-

ductance. The measured output power spectrum of the UWB transmitter is plotted in

Fig. 3.20. The power spectrum complies with the FCC-defined 0-1 GHz UWB spectrum

(mask) also shown. An example of the input data to the UWB transmitter Manchester-

encoded at the rate of 10Mb/s and the data received at the distance of 5cm using a

custom-built UWB receiver is shown in Fig. 3.21.

Table II provides a summary of experimentally measured characteristics of the


CURRENT (nA)100.10 100.15 100.20 100.25 100.30 100.35 100.400

1

2

3

4

5

6

7MEAN= 100.26nA

SD(3σ) = 34pA

N= 32

NU

MB

ER

OF

OC

CU

RA

NC

ES

(a)

CURRENT (nA)100.10 100.15 100.20 100.25 100.30 100.35 100.400

1

2

3

4

5

6

7

MEAN= 100.18nA

SD(3σ) = 22pA

N= 32

NU

MB

ER

OF

OC

CU

RA

NC

ES

(b)

Figure 3.17: Experimentally measured output current of 32 channels (from 16 chips, two chan-nels each) for input current of (a) 100nA without DEM [32] and (b) 100nA with DEM (thiswork).

integrated CMOS DNA analyzer SoC.

3.7 Experimental Electrochemical Results

To validate the performance of the channel in electrochemical sensing applications,

CV, first, scans of a DNA reporter potassium ferricyanide and a buffer solution were

performed. Potassium ferricyanide K4[Fe(CN)6] is commonly used in electrochemical

DNA detection systems as a redox reporter. Cyclic voltammetry recordings of 20µM

potassium ferricyanide solution and 1M potassium phosphate buffer (pH 7.3) have been

carried out. On-chip waveform generator was utilized to generate the CV excitation

waveform. A 500mV/sec 0.7V peak-to-peak ramp-up-ramp-down CV waveform with


100.6 100.8 101.0 101.2 101.4 101.6 101.8 102.0−0.5

0

0.5

1

TIME (μs)

AMPLITUDE (V)

1 1 0 1 0 1 1 1

(a)

−0.1

−0.05

0

0.05

0.1

TIME (μs)

AM

PL

ITU

DE

(V

)

100.6 100.8 101.0 101.2 101.4 101.6 101.8 102.0

(b)

Figure 3.18: Experimentally measured (a) Manchester-encoded input data to the UWB trans-mitter and (b) the output pulses.

a 50ms resting period was applied between a 55µm×55µm flat gold working electrode

in Fig. 3.2(b) and an off-chip Ag/AgCl reference electrode (Basi, RE-5B). The result-

ing CV curves recorded by the chopper-stabilized channel with DEM are shown in

Fig. 3.22. The phosphate buffer CV curve occurs mainly because of the charging and

discharging of the electrode-electrolyte double layer capacitance and thus has no peak.

In contrast, the potassium ferricyanide CV curve shows two distinct peaks at the re-

duction and oxidation voltages of potassium ferricyanide. Indeed, such flat electrodes,

typically produce such distinct redox peaks.

The recorded CV waveforms characteristics (redox peaks location and spacing) are

similar to those reported in the literature [141, 142]. A typical CV curve is shown in

Fig. 1.11(b). The separation between the two peak potentials, ∆Ep=Epc-Epa, can be

used determine the electrochemical reversibility for a redox couple. For a reversible

CV reaction one has [102]


80.42 80.43 80.44 80.45 80.46−0.12

−0. 1

−0.08

−0.06

−0.04

−0.02

0

0.02

0.04

0.06

0.08

TIME (μs)

AM

PL

ITU

DE

(V

)

SIMULATED

MEASURED

RINGING

Figure 3.19: Wirelessly measured UWB pulse at the distance of 5cm using a custom-built UWBreceiver.

109

−100

−90

−80

−70

−60

−50

−40

−30

108

FREQUENCY (Hz)

OU

TP

UT

PO

WE

R (

dB

m)

FCC MASK

Figure 3.20: Experimentally measured UWB transmitter output spectrum (direct output of thetransmitter driving a 50 ohm load). The output spectrum is compliant with the 0-1GHz FCCUWB band output power criteria

.


0 0.25 0.750.5 1

0

0.5

1

TIME (µS)

TRANSMITTED DATA @ 10Mb/s

MANCHESTER-ENCODED DATA

AM

PL

ITU

DE

(V

)

(a)

0 0.5 10.25 0.750

0.5

1

TIME (µS)

RECEIVED DATA (5cm)

AM

PL

ITU

DE

(V

)

(b)

Figure 3.21: (a) Manchester-encoded input data to the UWB transmitter and (b) the correspond-ing data received wirelessly at a 5cm distance.

−700 −600 −500 −400 −300 −200 −100 0−12

−10

−8

−6

−4

−2

0

2

4

6

8

V -V (mV)

CU

RR

EN

T (

nA

)

OXIDATION

VOLTAGE

RE WE

REDUCTION

VOLTAGE

FORWARD

SCAN

REVERSE

SCAN

1M POTASSIUM

PHOSPHATE BUFFER

20µM K [Fe(CN )]4 6

Figure 3.22: Experimentally recorded cyclic voltammograms of 1M potassium phosphatebuffer and 20µM potassium ferricyanide solution using the 55µm×55µm working electrodein Fig. 3.2(b)

.


−700 −600 −500 −400 −300 −200 −100 0−12

−10

−8

−6

−4

−2

0

2

4

6

8

V -V (mV)

CU

RR

EN

T (

nA

)

RE WE

20µM K [Fe(CN) ]

10µM K [Fe(CN) ]

30µM K [Fe(CN) ]

40µM K [Fe(CN) ] 4 6

4 6

4 6

4 6

OXIDATION

VOLTAGEREDUCTION

VOLTAGE

Figure 3.23: Experimentally recorded cyclic voltammograms of 10µM, 20µM, 30µM and40µM potassium ferricyanide solution using the 55µm×55µm working electrode in Fig. 3.2(b)

.

CU

RR

EN

T (

nA

)

10 20 30 40 600

2

4

6

8

K [Fe(CN) ] CONCENTRATION (µM)4 6

Figure 3.24: Calibration curve for the peak reduction current of potassium ferricyanide solutionfor the 55µm×55µm Au

working electrode in Fig. 3.2(b).


Table 3.2: Experimentally Measured Electrical Characteristics

Technology 0.13µm CMOSSupply Voltage 1.2VArea 3mm×3mmArray Dimensions 9×6 channelsChannel Size 200µm×300µmSensitivity 8.6pAPower Consumption (System)

Waveform Generator 1.1 mWSRAM 1.3µWUWB Transmitter 400µW

Power Consumption (Channel)Current Conveyer 8µWComparator 19µWBiasing 4µWDigital 11µWTotal (channel) 42µW

CURRENT (nA)

NU

MB

ER

OF

OC

CU

RA

NC

ES

6.4 6.6 6.8 7.0 7.2 7.4 7.6 7.80

3

5

7

9

11

13

MEAN= 7.10nA

SD(3σ) =0.22nA

N= 48

Figure 3.25: Experimentally recorded peak reduction current of the 40µM potassium ferri-cyanide solution recorded using the 55µm×55µm working electrode shown in Fig. 3.2(b) by48-channel on the CMOS DNA analysis SoC.

∆Ep =0.058

n, (3.1)

where n is the number of electrons transferred between the redox complex. This value

is independent of the scan rate for fast electron transfer. Increasing values of ∆Ep as a

function of increasing scan rate indicates the presence of electrochemical irreversibil-

ity. In practice, the theoretical value of 58/n mV for ∆Ep is seldom observed. In all

experiments the potassium ferrocyanide solution was diluted in 1M potassium phos-

phate buffer. This combined with the slow electron transfer kinetics present in case of

our complex multi-material electrodes have caused the peak voltage difference to de-


∆ pH

CU

RR

EN

T (

nA

)

−2 −1.5 −1 −0.5 0 0.5 1 1.5 22

3

4

5

6

7

8

9

10

11

Figure 3.26: Experimentally measured on-chip pH sensor calibration curve relative to pH of 7.A total of 60 measurements from 3 chips, 20 measurements each, have been performed. Thecorresponding 3σ error bars are shown.

viate from the theoretical 58mV value [33, 34]. In all experiments, the first four CV

curves were discarded and the fifth curve was used as the recorded data. As a result,

the peak recorded redox current is consistent for different concentrations. Other record-

ings [33, 34] using a similar DNA detection method also achieve ∆Ep higher than the

theoretical value of 58mV.

Next, CV scans of a potassium ferricyanide solution with four different concen-

trations (10µM to 40µM) using a 55µm×55µm flat gold working electrode shown in

Fig. 3.2(b) have been performed to study the effect of a change in the DNA reporter

concentration on the recorded redox current. As shown in Fig. 3.23, the peak current

at the reduction and oxidation voltages of potassium ferricyanide increases with an in-

crease in its concentration. The corresponding calibration curve is shown in Fig. 3.24.

This curve demonstrates the linear relationship between the concentration of potassium

ferricyanide and the output redox current.


CV scans of a potassium ferricyanide solution at 40µM have been conducted on

all 48 channels with 55µm×55µm flat gold working electrodes shown in Fig. 3.2(b)

to study the effect of the channel-to-channel variation on the CV recording results. A

500mV/sec 0.7V peak-to-peak ramp-up-ramp-down CV waveform with a 50ms resting

period was used in this experiment. Fig. 3.25 shows the resulting peak reduction cur-

rents recorded by the 48 channels. The mean peak reduction current is 7.02nA, and the

three-sigma variation is 0.22nA.

To validate the performance of the pH sensors, the sensitivity of the ISFET is mea-

sured in response to change in the solution pH level. A preliminary analysis of the

ISFET characteristics indicated that the pH sensors have different threshold voltages,

due to the trapped charge on the floating gates of the ISFETs. The UV radiation and

bulk substrate biasing (for 8 hours) technique was used to remove the trapped charge

and thus remove the threshold voltage mismatch among the pH sensors. Before the

pH sensor sensitivity is measured, the sensor array must be etched for 10s in a 10%

buffered hydrofluoric acid solution. Measurements made without this step are gener-

ally very noisy and result in a low sensitivity. After the threshold voltage calibration,

the sensitivity of the pH sensor is measured in a 0.1M NaCl electrolyte by adding small

quantities of hydrochloric acid to change the solution pH from five to nine. Recording

the calibrated steps in the measured current leads to the finding that the array has a lin-

ear response of 1.8nA/pH. The corresponding calibration curve with error bars (from

three chips, 20 measurements each) is shown in Fig. 3.26.

3.8 Synthetic Prostate Cancer DNA Detection

The SoC has been validated in label-free amperometric detection of synthetic prostate

cancer DNA. The DNA sequences are synthesized by Integrated DNA technology [154].


−12

−10

−8

−6

−4

−2

0

2

4

6

8

BARE GOLD ELECTRODE

GOLD ELECTRODE WITH PROBE DNA

NONCOMPLEMENTARY TARGET DNA

COMPLEMENTARY TARGET DNA C

UR

RE

NT

(n

A)

DETECTION

NO DETECTION

NO HYBRIDIZATION

K [Fe(CN) ] 64

OXIDATION VOLTAGE

(SSDNA)

−700 −600 −500 −400 −300 −200 −100 0

V -V (mV)RE WE

(DSDNA)

10

12

Figure 3.27: Experimentally measured cyclic voltammetry results of 5µM prostate cancer syn-thetic DNA detection from the 55µm×55µm flat gold working electrode in Fig. 3.2(b)

.

The following synthetic DNA sequences have been used in the experiments: DNA

probes (5ThioMC6-D/AG CGC GGC AGG AAG CCT TAT), complementary target

DNA (ATA AGG CTT CCT GCC GCG CT) and non-complementary DNA (TTT TTT

TTT TTT TTT TTT TT). All the DNA experiments were conducted at room tempera-

ture. In all the experiments a 500mV/sec 0.7V peak-to-peak ramp-up-ramp-down CV

waveform with a 50ms resting period was applied between the working electrode and a

commercially available off-chip Ag/AgCl reference electrode (Basi RE-5B) [153].

Fig. 3.27 shows cyclic voltammetry scans from an on-chip 55µm×55µm flat gold

electrode for the 5µM prostate cancer synthetic DNA cyclic voltammetry recording,

in a 40µM potassium ferricyanide solution. The CV scan rate and range were set

to 500mV/sec and 0.7V peak-to-peak, respectively, with a 40ms resting period. The


PROBE

DNA

NONCOMPLEMENTARY

TARGET DNA

COMPLEMENTARY

TARGET DNA

CU

RR

EN

T (

nA

)

−12

−10

−8

−6

−4

−2

0

AT V -V = -450mVRE WE

ABSTRACT TIME

BARE GOLD 55μm×55μm

FLAT ELECTRODE

DETECTIBLE SIGNAL

CHANGE

Figure 3.28: Experimentally measured 5µM prostate cancer synthetic DNA cyclic voltamme-try recording 3σ error bars from 3 chips 60 measurements each from 55µm×55µm flat goldworking electrodes in Fig. 3.2(b).

bare gold electrode CV scan demonstrates well-defined oxidation and reduction peaks,

whereas scans taken using 5µM single-stranded probe DNA attached to electrodes show

a reduction in the oxidation/reduction peaks. This is expected since thiolated DNA

probes create a negatively charged film on the electrode repelling the negatively charged

electrochemical reporter potassium ferricyanide as illustrated in Fig. 3.1. Further adding

a 5µM non-complementary DNA target does not change the CV signal oxidation peak

value significantly indicating that non-specific adsorption is negligible. On the other

hand, adding a 5µM complementary target single-stranded DNA onto the chip leads to

creation of double-stranded DNA on the biosensing electrode resulting in an additional

negative charge and elimination of potassium ferricyanide redox peaks. The corre-

sponding error bars (from 3 chips, 20 measurements each) are shown in Fig. 3.28. As it

can be seen, the detectible signal change in this case is 2.85nA.

The same set of experiments were repeated with the on-die nanostructured elec-

trodes to study their DNA detection capabilities. Fig. 3.29 shows the CV curves ob-


−700 −600 −500 −400 −300 −200 −100 0

V -V (mV)RE WE

−4

−3

−2

-1

0

1

2

3

4

5

6

CU

RR

EN

T (

nA

)

BARE GOLD ELECTRODE

NONCOMPLEMENTARY TARGET DNA

COMPLEMENTARY TARGET DNA

GOLD ELECTRODE WITH PROBE DNA (SSDNA)

(DSDNA)

DETECTION

NO DETECTION

NO

HYBRIDIZATION

7

Figure 3.29: Experimentally measured cyclic voltammetry results of 100aM prostate cancersynthetic DNA detection, from 2µm×2µm nanostructured working electrodes in Fig. 3.3(e)

.

BARE NANOSTRUCTURED

2μm×2μm MICROELECTRODE

PROBE

DNA

NONCOMPLEMENTARY

TARGET DNA

COMPLEMENTARY

TARGET DNA

CU

RR

EN

T (

nA

)

6

5

4

3

2

1

0

AT V -V = -200mVRE WE

ABSTRACT TIME

DETECTIBLE SIGNAL

CHANGE

Figure 3.30: Experimentally measured 100aM prostate cancer synthetic DNA cyclic voltam-metry recording 3σ error bars from 3 chips, 60 measurements each, from 2µm×2µm nanostruc-tured working electrodes in Fig. 3.3(e)

.


10−19

10−17

10−15

10−13

10−11

10−9

10−7

CONCENTRATION (M)

I -I

0

25

50

75

100

10−5

DS

DN

AS

SD

NA

SS

DN

AI

(%

)

NUMBER of DSDNA STRANDS

10 102

1040

106

108

1010

1012

1014

10aM

DETECTION

LIMIT

100fM

DETECTION

LIMIT

100nM

DETECTION

LIMITΔI

=

Figure 3.31: Experimentally measured microelectrode characteristics, detection limits and dy-namic ranges in prostate cancer synthetic DNA detection using the three electrodes types shownin Fig. 3.3(d) and (e) and Fig. 3.2(b). Error bars (3 sigma) are from 3 chips, 100 measurementseach.

tained for a nanostructured electrode grown at VRE-VWE = 0mV for 100aM prostate

cancer synthetic DNA concentration, in a 40µM potassium ferricyanide solution. As

expected, compared to the flat gold electrodes the nanostructured electrodes typically

do not exhibit the redox peaks [138]. As it can be seen from Fig. 3.29 the current level

in the presence of complementary target DNA (DSDNA) is smaller compared to the

case where only the probe DNA (SSDNA) is present. The corresponding error bars

(from 3 chips, 60 measurements each) are shown in Fig. 3.30. As it can be seen, the

detectible signal change in this case is 1.1nA.

DNA sensing experiments were conducted for the target DNA concentrations of

1aM to 10µM to study the detection limits of the on-die nanostructured electrodes and

the on-die flat gold electrode. The resulting characteristics, detection limits and dy-

namic ranges of the two nanostructured electrode types and the 55µm×55µm flat gold

electrode are given in Fig. 3.31. ∆I is computed as (IDSDNA-ISSDNA)/ISSDNA)×100,


where IDSDNA is the redox current after the hybridization and ISSDNA is the redox cur-

rent before the hybridization. All the current recordings for nanostructured electrodes

are taken at VRE-VWE of -200mV and at at VRE-VWE of -250mV for the flat gold

electrodes. The corresponding error bars (from 3 chips, 100 measurements each) are

also shown in Fig. 3.31.

The detection limit, defined as the lowest concentration for which the background-

subtracted signal is three times higher than the standard deviation at that concentra-

tion, for nanostructured electrodes grown at VRE-VWE = 0mV as shown in Fig. 3.31

is 10aM. The 10aM sensitivity achieved using the optimized on-CMOS nanostructured

electrode enables PCR-free detection for many applications. This limit corresponds to

the detection of fewer than 100 copies of the target sequence.

Another benefit of having several types of electrodes on the same chip is that dif-

ferent electrodes cover different concentration ranges. As it can be seen from Fig. 3.31,

the nanostructured electrodes grown at VRE-VWE = 0mV cover a dynamic range (de-

fined as the range at which the 3 sigma error bar of the given concentration is below 100

and above 0 on the y-axis) of 3aM to 100fM, the nanostructured electrodes grown at

VRE-VWE of 100mV cover a dynamic range of 100fM to 90pM, and the 55µm×55µm

flat gold electrodes cover a dynamic range of 1nM to 10µM. As a result, by fabricating

electrodes with different degrees of nanostructuring, we can significantly expand the

dynamic range of the CMOS DNA sensing microsystem (as wide as 140dB with these

types of nanostructured microelectrodes [138]).

Table 5.3 provides a comparative analysis of the presented design and existing

amperometric biochemical sensory microsystems. The design presented in this work

achieves the highest dynamic range and the lowest sensitivity in terms of ADC LSB.

We have shown successful detection of 20-base pair long synthetic prostate cancer DNA

from several types of on-chip Au electrodes. The 10aM detection limit is the lowest de-


tection limit reported in literature from an integrated circuit-based DNA sensor to date.


Tabl

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3.9 Chapter Summary

A 54-channel 0.13µm CMOS fast-scan cyclic voltammetry DNA analysis SoC has been

presented. The microsystem includes 600 time-multiplexed DNA sensors and 54 pH

sensors. It also includes an arbitrary waveform generator, an on-chip memory, an in-

channel low-noise chopper-stabilized frontend current conveyer with dynamic element

matching, an in-channel dual-slope ADC and a fully digital ultra-wideband transmit-

ter. Chopper stabilization achieves input-referred noise of less than 0.13pA over the

operating bandwidth. Dynamic element matching improves current conveyer accuracy

by 54 percent at the 100pA input current level. The in-channel SRAM enables in-

channel calibration which results in a 17 percent improvement in channel-to-channel

ENOB variation. Each channel occupies an area of 0.06mm2 and consumes 42µW of

power from a 1.2V supply. The presented current-to-digital channel design achieves a

combined dynamic range of 93dB with the sensitivity of 8.6pA. Two types of nanostruc-

tured microelectrodes and one type of a flat gold electrode have been characterized in

on-CMOS DNA prostate cancer detection. The on-chip nanostructured microelectrodes

achieve label-free PCR-free detection limit of 10aM, which is the lowest reported on-

CMOS detection limit.

111

Chapter 4

Impedance Spectroscopy DNA

We present a 54-channel, mixed-signal CMOS DNA analyzer that utilizes frequency

response analysis (FRA) to extract the real and imaginary impedance components of

the biosensor. Two computationally intensive operations, the multiplication and in-

tegration required by the FRA algorithm, are performed by an in-channel dual-slope

multiplying ADC in the mixed-signal domain resulting in minimal area and power con-

sumption. Multiplication of the input current by a digital coefficient is implemented

by modulating the counter-controlled duration of the charging phase of the ADC. In-

tegration is implemented by accumulating output digital bits in the ADC counter over

multiple input samples. The 1.2mm×1.6mm prototype fabricated in a 0.13µm standard

CMOS technology has been validated in prostate cancer synthetic DNA detection. Each

channel occupies an area of only 0.06mm2 and consumes 42µW of power from a 1.2V

supply.

4.1 Introduction

This Chapter presents a scalable, multi-channel, compact and low-power CMOS impedance

spectroscopy DNA analyzer. Frequency response analysis (FRA) algorithm is utilized

CHAPTER 4. IMPEDANCE SPECTROSCOPY DNA 112

to extract the real and imaginary components of the biosensor impedance [156]. The

proposed microsystem consists of a programable on-chip waveform generator and 16

impedance extraction channels. Each channel includes a current-mode input dual-slope

multiplying ADC. It efficiently performs multiplication and integration, two computa-

tionally intensive operations required to implement the FRA algorithm. Multiplication

of the input current by a digital coefficient is implemented by modulating the counter-

controlled duration of the charging phase of the ADC by that coefficient. Integration

is implemented by accumulating the output digital bits in the ADC counter. The dual-

slope multiplying ADC utilizes mostly the same circuits as a conventional dual-slope

ADC, and the multiplication and integration are achieved by modifying the ADC algo-

rithm. The rest of the chapter is organized as follows. Section II presents the principle

of DNA detection on a CMOS die. Section III presents the impedance spectroscopy

VLSI architecture. Section IV details the circuit implementation of the VLSI architec-

ture. Section V demonstrates the electrical experimental results obtained from a 0.13µm

CMOS prototype. In Section VI, results of on-chip impedance spectroscopy of DNA in

prostate cancer screening are presented.

4.2 DNA Detection Principle

The principle of the label-free DNA detection method is shown in Fig. 3.1. It employs

potassium ferrocyanide K4[Fe(CN6)] reporter. Potassium ferrocyanide is a negatively

charged redox complex with well-defined electrochemical signature exhibiting oxida-

tion and reduction currents at VWE-VRE of -450mV and -200mV respectively [140].

The maximum electron transfer between the electrode and potassium ferrocyanide is

achieved in the absence of DNA target and probe as illustrated in Fig. 3.1(a). The

oxidization current IOX drops when the Au electrode surface is hybridized with nega-


tively charged probe DNA, as illustrated in Fig. 3.1(b). When the complementary DNA

binds with the probe DNA, the surface negative charge further increases and the oxi-

dization current is further reduced, as shown in Fig. 3.1(c). This change in the current

is an indicator of the target DNA presence and concentration. In other words, the pres-

ence of negatively charged DNA on the biosensor surface is translated to a decrease

in the potassium ferrocyanide oxidation/reduction current creating a detectible signal

change [141, 142].

The DNA sensing electrodes are created by first forming passivation openings on

the top metal layer (aluminum) of the CMOS die similarly to how it is done for wire

bond pads. Electroless electroplating is then employed to deposit nickel (2µm), palla-

dium (0.2µm) and gold (0.1µm) on the exposed Al surface to form a bio-compatible

electrode surface. After the electrode fabrication, the die is wire-bonded and the bond-

ing wires are insulated with a biocompatible epoxy to enable on-chip electrochemical

experiments without damaging the bonding wires [157].

∫

∫

cos(ωt)

BIOSENSOR

sin(ωt)

FRA UNIT

REAL

IMAG

Asin(ωt+φ)

SIG

NA

L G

EN

ER

AT

OR

(Q

- O

SC

ILL

AT

OR

)

Figure 4.1: Block diagram of a frequency-response analyzer (FRA) system for biosensorimpedance spectroscopy.

4.3 Impedance Spectroscopy VLSI Architecture

A small-signal model of the electrode-electrolyte interface in an electrochemical cell is

shown in the center of Fig. 1.8. Fast fourier transform (FFT) and frequency-response an-

alyzer (FRA) are two methods widely used for characterizing the electrode impedance [156].


DUAL-SLOPE MULTIPLYING ADCBIOSENSOR

CWE

RS

RCT

SRAM1

REWE

OR

REAL

IMAG∫I-V

D=sin(n) OR D=cos(n)

I IN

DAC

sin(n)

WAVEFORM

GENERATOR

SRAM2

CLK

1.2 V

IBIAS VBIAS

CDB

MULTI-FREQUENCY

CLOCK GENERATOR

PATTERN

GENERATOR

Figure 4.2: Impedance spectroscopy microsystem functional block diagram.

Compared to the FFT, the FRA method requires simple circuitry and can be imple-

mented on a small silicon area, making it suitable for sensory array microsystems. A

functional block diagram of the FRA algorithm is shown in Fig. 4.1. The sensor is in-

terrogated with a sinusoidal voltage. Multiplication of the sensor response Asin(ωt+ϕ)

with sin(ωt) or cos(ωt) and integration of the results over one cycle of the interroga-

tion signal results in separation of the real and the imaginary components of the sensor

impedance as follows [156]

Real =

∫ T

0

Asin(ωt+ ϕ)× sin(ωt) dt =

∫ T

0

0.5× (Acos(ϕ) + cos(2ωt+ ϕ)) dt,

=0.5T

π× Acos(ϕ) (4.1)

Imag =

∫ T

0

Asin(ωt+ ϕ)× cos(ωt) dt =

∫ T

0

0.5× (Asin(ϕ) + sin(2ωt+ ϕ)) dt,

=0.5T

π× Asin(ϕ) (4.2)


where A represents the amplitude, ϕ represents the phase of the sensor impedance and

T is the period of the interrogation signal. Acos(ϕ) is the real portion and Asin(ϕ) is the

imaginary portion of the sensor impedance. These values over a wide frequency range

are sufficient to fully describe the sensor impedance.

D0

D15

SRAM1

CO

NT

RO

L L

OG

IC

MU

LT

I-F

RE

QU

EN

CY

CL

OC

K G

EN

ER

AT

OR

8-BIT

UP/DOWN

COUNTER

8-BIT

R-2R DACOFF-CHIP

DATA

ANALOG

OUTPUT

S 2-bits

00

01

10

11

ON-CHIP OFF-CHIP

Figure 4.3: Waveform generator functional block diagram.

−1−0.8

−0.6

−0.4

−0.2

0

0.2

0.40.6

0.8

1

D0

D1

D15

D14D14

D1

D0D0

-1

D1-1

D15D14 D14

-1

-1

-1

D1-1

D0-1

COUNT UP

S = 00

COUNT DOWN

S = 00

COUNT UP

S = 01

COUNT DOWN

S = 01

45 90 135 180

Angle (Degree)

0

NO

RM

AL

IZE

D A

MP

LIT

UD

E

Figure 4.4: Principle of the sine wave generation.

In this work the FRA algorithm has been chosen to implement a sensory array

impedance spectroscopy microsystem. The two key components in this system are

the multiplier and the digital integrator. Both of these operations are implemented with

an in-channel multiplying dual-slope ADC that reuses the circuits of a conventional

dual-slope ADC [158].


0.00

0.25

0.50

0.75

1.00

1.25

|REA

L R

ELA

TIV

E ER

RO

R| (

%)

4−110

010

110

210

31010

4−110

010

110

210

31010

FREQUENCY (Hz)

0.00

0.25

0.50

0.75

1.00

1.25

|IMA

G R

ELA

TIV

E ER

RO

R| (

%)

FREQUENCY (Hz)(a)

(b)

NUMBER OF INTEGRATION CYCLES ( )4

100

101

102

103

105

10

NUMBER OF INTEGRATION CYCLES ( )4

100

101

102

103

105

10

N

N

Figure 4.5: Absolute value of the relative error of the biosensor R-C model impedance as afunction of frequency due to stepwise approximation of the interrogation signal for the : (a)real, and (b) imaginary components.


4.4.1 Multi-channel System-Level Architecture

The functional block diagram of the impedance spectroscopy microsystem based on

the FRA algorithm is shown in Fig. 4.2. The microsystem is comprised of a pro-

gramable analog waveform generator, a programmable digital pattern generator and

an array of impedance extraction units. The waveform generator produces the interro-

gation waveform sin(ωt) and drives the reference electrode with it. The digital pattern

generator generates digital multiplication coefficients representing either sine wave or

cosine wave that are synchronized with the interrogation waveform. Each impedance

extraction unit consists of a dual-slope multiplying ADC (DS-MADC). The front-end


−

+

S1

S2

S2

IIN

WE

VWE

VWE

IREF+

IREF-

CFS3

VOUT

3

(S1 ,S3,S2 )

CLK

RESET

UP/

DOWN

OR

REAL

IMAG

D = sin(n) OR D = cos(n) (FROM PATTERN GENERATOR)

TIME A

VWES1

UP

/DO

WN

CO

UN

TE

R

IN/O

UT

LA

TC

H

READ_IN

READ_OUT

WRITE

6

CO

NT

RO

L

L

OG

IC

TIME B TIME C

−

+

−

+

−

+

OTA1

VOUT

S

S

UP/DOWN

READ_OUT

WRITE

READ_IN

1

2

3

DT1T

RDDT

2

RESET

A B C

S2OR

t

S

VLATCH

(a)

(b)

Figure 4.6: (a) Dual-slope multiplying ADC VLSI architecture, and (b) timing diagram of allrelevant signals.

current-to-voltage converter is an analog integrator that acquires an input current at

the low-impedance input node set to a controlled potential. The DS-MADC multiplies

the biosensor response IIN with a set of digital coefficients D representing sin(ωt) or

cos(ωt) that are synchronized with the analog sinusoidal interrogation voltage on the

reference electrode. Next the DS-MADC accumulates the results over one period of the

interrogation signal using a digital integrator (counter), thus extracting the real or the

imaginary components of the biosensor impedance.


4.4.2 Waveform Generator and Pattern Generator

The frequency response analysis implementation is the simplest when rectangular wave-

forms are used instead of sine wave for both the interrogation and the multiplication

signals. The problem is that severe systematic errors appear due to the higher order

harmonics existing in the rectangular waveforms. Stepwise approximation of the in-

terrogation waveform and the multiplication signals reduces the effect of higher or-

der harmonics and increases the measurement accuracy. It has been shown [130] that

representing both the interrogation signal and the multiplication signals by a coarsely

quantized approximation can significantly reduces the error due to the higher order har-

monics and can significantly reduce the measurement inaccuracy.

= TI IN

I REF

VOUT

TIME

PHASE I PHASE II

DT

2

T2

T1

1

1

DT

2DT

Figure 4.7: Timing diagram illustrating the ADC multiplication function.

In this work a programmable analog waveform generator is utilized to generate

the stepwise approximation of the interrogation signal. The block diagram of the pro-

grammable waveform generator is shown in Fig. 4.3. It is composed of an 8-bit R-2R

DAC, an on-chip SRAM1 and a bidirectional counter. The DAC coefficients are stored

in the on-chip SRAM1. The on-chip waveform generator generates both stepwise sinu-

soidal and cyclic voltammetry (CV) ramp waveforms. It provides a wide range of user-

controlled rate and amplitude parameters with a maximum CV scan range of 1.2V, and

scan rate ranging between 0.1mV/sec to 400V/sec and the sinusoidal frequency range


from 100mHz to 50kHz. An off-chip clock generator provides the variable frequency

clock signal enabling the waveform generator to generate the interrogation signal at dif-

ferent frequencies. The DAC occupies an area of 0.012mm2 and dissipates 1.1mW of

power from a 1.2V supply when driving a load of 5nF at 50kHz. At low frequencies the

interrogation sine wave is represented by 64 samples per period. The DAC coefficients

for the first 16 samples (D0 to D15) of the waveform generator output waveform and for

the corresponding digital sin/cos multiplication coefficients are stored in two on-chip

global SRAM banks (SRAM1 and SRAM2, respectively), as shown in Fig. 4.2. The

SRAMs occupy an area of 0.028mm2 and dissipate 0.9µW of power when clocked at

50kHz. By symmetry, 64 samples in one period are generated from the 16 samples

stored on-chip. As shown in Fig. 4.4, for a sine wave, the first 16 samples stored in the

SRAM1 generates the first quadrant of the sine wave. In this case the counter control-

ling the SRAM1 is counting up and S is set to 00, in Fig. 4.3. In the next quadrant the

counter controlling the SRAM1 counts down thus reversing the order of the samples

and generating the second quadrant of the sine wave. S is set to 00 in this case. The

third and forth quadrants are generated in the same manner but in this case the polarities

of the samples fed into the DAC are reversed (by setting S to 01). To generate a cosine,

the cycle starts from the second quadrant instate of the first one. If a CV waveform is

required the up/down counter is directly interfaced to the waveform generator DAC (by

setting S to 11).

As the interrogation frequency increases, the number of samples representing the

interrogation and multiplication signals decreases and at 10kHz both signals are repre-

sented by three samples. This greatly reduces the ADC speed requirement while the

error caused by the reduction in number of samples is kept low by averaging the results

over multiple cycles.

An ideal model of the impedance spectroscopy microsystem has been constructed


using verilog-A model components (ideal analog integrator and multiplying ADC) to

study the effect of step-wise approximation of the interrogation signal on the overall

system accuracy. First a simulation was performed where a sinusoidal voltage stimulus

(8-bit accurate generated by an ideal DAC) with the frequency swept from 0.1Hz to

10kHz was applied to the biosensor model shown in Fig. 4.2. The value of RS was set

to 1GΩ, CWE was set to 500pF, CDB was set to 300pF and RCT was set to 1MΩ. The

sensor response was recorded with the ideal analog integrator and multiplied, using an

ideal multiplying dual-slope ADC, by an 8-bit accurate digital multiplication coefficient

representing sine or cosine in the FRA algorithm in Fig. 4.1. Next, the same set of sim-

ulations were performed in which both digital multiplication coefficients and the analog

sinusoidal voltage interrogation signal are represented by 64 samples per period at low

frequencies. In this simulation as the interrogation frequency increases, the number of

samples representing the interrogation and multiplication signals decreases. Also, in

this simulation the results are averaged over multiple cycles (N) to reduce the errors

caused by reduction in the number of samples. The absolute value of the relative error

of the biosensor impedance computed using 8-bit accurate interrogation signal and 8-bit

accurate multiplication coefficients versus the simulated biosensor impedance obtained

from stepwise approximation of these signals is shown in Fig. 4.5. The absolute value

of the relative error stays below 0.9% and 1.0% for the real and imaginary component

of the biosensor respectively.

4.4.3 Dual-slope Multiplying ADC Channel

The VLSI architecture of one channel of the integrated spectrum analyzer is depicted in

Fig. 4.6(a). Each channel consists of an integrating amplifier with an on-chip 10pF ca-

pacitor CF , a high-speed latched comparator and digital blocks. The integrator switches

are implemented with low-leakage switches as shown.


The conventional dual-slope ADC operates in two phases as depicted in Fig. 4.7.

In phase I the integrating capacitor CF is charged for a predetermined period of time

T1. Next, during the second phase of the operation, the capacitor is discharged to zero

by a DC reference current. By counting the time T2, a digital representation of IIN

can thus be obtained as (T2/T1)×IREF . To implement multiplication of the input cur-

rent by a digital sin/cos coefficient (D1 to D15) as needed by the FRA algorithm, the

duration of phase I is scaled with a constant coefficient D<1 as shown in Fig. 4.7. In

this case by counting the time DT2, a digital representation of DIIN can be obtained as

D×(T2/T1)×IREF .

To extract the real and imaginary components of the biosensor impedance, the input

current IIN is multiplied by the reference sine or cosine digital coefficient denoted as D

(stored in SRAM2, in Fig. 4.2) and the results are integrated over one period by a 16-bit

counter as follows

Real =

∫ T

0

IIN × sin(ωt)dt =

N/2∑1

IIN × |sin[n]| −N∑

N/2+1

IIN × |sin[n]|(4.3)

Imag =

∫ T

0

IIN × cos(ωt)dt =

N/2∑1

IIN × |cos[n]| −N∑

N/2+1

IIN × |cos[n]|(4.4)

where N is the number of samples in one period and sin[n] and cos[n] are the digital

multiplication coefficients D that are stored in the SRAM2. The multiplying dual slope

ADC performs the multiplication required by equations (5.3) and (4.4), thus the need

for 16 digital multipliers is eliminated.

The timing diagram of the ADC for a typical conversion cycle is shown in Fig. 4.6(b).

First, the integrating counter is reset. At the same time, the sin/cos multiplication coeffi-


VIN+

VIN-

VBIASN

VCASCP

CASCNVV

OUT

M1 M2

M3 M4

M5 M6

M7 M8

M9 M10M11

chopM

Figure 4.8: Chopper-stabilized folded-cascode OTA1 in the analog integrator.

VIN+ IN-

V

VLATCH

V

OUT+V

OUT-

VLATCH VLATCH

VOUT- VOUT+

VIN+

VIN-

VBIAS

(a) (b)

M1

M2

M3

M4 M

5M

6 M7

M8 M9

M10

M11 M12

M13 M14M15 M16

Figure 4.9: Comparator circuit diagram. (a) One of the three identical gain stages, and (b) thehigh-speed latch.

cient, D, is loaded into the in-channel input latch (time A). Next, the in-channel counter

counts up from zero to time DT1 and the input current is integrated onto capacitor CF .

After time DT1, the voltage on the capacitor is held constant for a fixed time interval

TRD. During this time the content of the output latch (zero for the first conversion cycle)

is loaded into the counter (time B). During time DT2, depending on the comparator out-

put, the integrating capacitor is discharged using the appropriate current source, IREF+

or IREF−. During time DT2 the counter counts up or down depending on the sign of

the input current in phase I and the final value of the counter is written into the output


Table 4.1: OTA1 and Comparator Transistor Sizing

OTA1 ComparatorTransistor W/L (µm) Transistor W/L (µm)

M1,2 8× 3/1 M1,2 12× 12/0.2M3,4 2× 1/4 M3 3× 2/4M5,6 4× 1/1 M4,5 2× 1.5/0.4M7,8 8× 1/1 M6,7 2× 1.5/0.4M9,10 4× 1/4 M8,9 6× 6/0.5M11 8× 1/2 M10 3× 2/4

– – M11,12 6× 6/0.5– – M13,14 2× 1.5/1

latch (time C). This part of the ADC conversion cycle performs the summation which

implements integration required by the FRA algorithm. This process is repeated for one

full cycle of the sinusoidal stimulation waveform and the final value stored in the output

latch corresponds to the real or imaginary component of the biosensor impedance, for

sin and cos multipliers respectively.

10−2

10−1

100

101

102

103

104

−140

−130

−120

−110

−100

−90

−80

−70

FREQUENCY (Hz)

CHOPPER ON

CHOPPER OFF


rms

1/2

Figure 4.10: Simulated input-referred noise spectrum of the analog integrator with the chopperon and off (clocked at 100kHz) from 0.01Hz to 10kHz.

The analog integrator amplifier shown in Fig. 4.8 is a folded-cascode transconduc-

tance amplifier consuming 6µW from a 1.2V supply. Impedance spectroscopy is per-

formed over the frequency range of 1Hz to 10kHz. To reduce the effect of flicker noise,

the amplifier utilizes input PMOS devices with a high aspect ratio and internal chop-


Table 4.2: Analog Integrator Noise Summary

Transistor Noise Total Noise, Total Noise,Source Chopper OFF Chopper ON

M3,4 Flicker 26.4% 11.02%M1,2 Flicker 22.1% 10. 3%M3,4 Thermal 21.2% 23.4%M1,2 Thermal 18.3% 22.1%M9,10 Flicker 5.25% 17.50%M9,10 Thermal 3.26% 8.26%M5,6 Flicker 1.1% 3.38%M7,8 Flicker 1.04% 1.19%M5,6 Thermal 0.9% 1.0%M7,8 Thermal 0.8% 0.9%

M1

M2

M3

M4

M5

M6

M7M8

M9 M10

M11 M

12

M13 M

14

M15

M16

M19

M20

M21

M22

M23

M24

M17

M18

RC

CC

RC

CC

Vin+

Vin-

VbiasN

VbiasP

VbiasP

VbiasP2

VbiasN2

OUTV

M26

M27

(a) (b)

RE

−

+

−

+

V

R1

R2

WAVEFORM

GENERATOR

CE

WE

BIOSENSOR

OTA2

OTA3

WE

Figure 4.11: (a) Circuit architecture of the three-electrode regulation loop, and (b) OTA2 andOTA3 circuit schematic.

ping. As shown in Fig. 4.8, the chopper switches are placed at the input of the OTA.

Another set is placed between the cascoded NMOS current mirrors. This significantly

reduces the flicker noise and DC offsets caused by the input pair transistors and the

NMOS current mirror transistors.

The simulated input-referred noise of the integrator for the cases where the chop-

per is disabled and enabled is shown in Fig. 4.10. The chopping improves the noise

spectrum by 5dB at 10Hz. The relatively low improvement due to the chopping is due

to the noise contribution from the PMOS current sources that are not chopped in this

design. A fully differential design would enable chopping at the input and output of the

op-amp thus minimizing the fliker noise contribution from the PMOS current sources.


The integrated input-referred noise from 0.01Hz to 10kHz is 0.12pA when the chopper

is disabled and 0.07pA when the chopper is enabled. The integrator transistor sizes are

shown in Table 4.1. Contribution of each transistor to the total input-referred noise is

shown Table 4.2. When the chopper is disabled the main contributions are from the

OTA current mirror transistors M3 and M4 and the input pair transistors M1 and M2.

When the chopper is enabled, the current mirror transistors M9 and M10 are the main

contributors to the input-referred noise.

The comparator is implemented with three pre-amplifier stages (Fig. 4.9(a)), with

a total gain of 60dB, and a high-speed output latch as shown in Fig. 4.9(b). The com-

parator is verified in simulation to operate at up to 40MHz. The full channel consumes

42µW of power from a 1.2V supply when clocked at 10MHz. The comparator transistor

sizes are shown in Table 4.1.

4.4.4 Three-electrode Regulation Loop

The functional block diagram of the regulation loop is shown in Fig. 4.11(a). A three-

electrode potentiostat configuration [68] is employed as opposed to the basic two-

electrode configuration. In a two-electrode configuration the current flowing between

the two electrodes alters the voltage drop across CDB due to the non-zero impedance

of the electrolyte solution and the electrode-electrolyte interface. In the three-electrode

configuration, a third, counter electrode (CE), rather than the reference electrode is

placed in the current path. The reference electrode does not sink/source an electric

current. Thus the voltage difference between the working and reference electrode is

maintained at a known voltage independent of the electrochemical cell impedance. The

three-electrode regulation loop is composed of the two OTAs as shown in Fig. 4.11(a).

To allow for wide input and output swings class-AB OTAs with both NMOS and PMOS

input differential pairs are used for the potentiostat as shown in Fig. 4.11(b). OTA tran-


Table 4.3: OTA2 and OTA3 Transistors Sizing

Transistor W/L (µm) Transistor W/L (µm)M1,2 6× 2/6 M16 1× 1.2/1M3,4 2× 1.5/0.4 M17 1× 1.2/1M5,6 2× 1.5/0.4 M18 1× 1.2/1M7,8 3× 2/0.12 M19,20 1× 1.2/1M9,10 3× 2/0.12 M21,22,23,24 3× 2/0.12M11 1× 1.2/1 M25 3× 2/0.12

M12,13,14,15 3× 2/0.12 M26 1× 1.2/1M9,10 3× 2/0.12

sistor sizing is given in Table 4.3. Assuming that VWE is fixed by the analog integrator,

the regulation loop configuration ensures that VRE-VWE tracks the voltage difference

between the output of the waveform generator and the positive terminal of the OTA3.

4.5 Electrical Experimental Results

1Х6 BANK OF CROSS-VALIDATION CHANNELS

UWB TX

WAVEFORM

GENERATOR & SRAMCONTROL

LOGIC

9×6 ARRAY OF

CHANNELSRECE

WE

2X2μm2

55X55μm

ISFET

5X5μm2X2μm

2

2

2ONE CHANNEL

8Х8 ARRAY OF

2Х2μm WE2

ARRAY OF

WEs

Figure 4.12: Die micrograph of the 54-channel integrated impedance spectrum analyzer fabri-cated in a 0.13µm standard CMOS technology.


0

10100f 1p 10p 100p 1n 10n 100n

140 dB

f =100 f =1000s s

INPUT CURRENT (A)

DIG

ITA

L O

UT

PU

T C

OD

E

1μ

f =500s

1

102

103

104

SNDR=48.2dB

LSB=320fA

SNDR=54.1dB

LSB=12.1pA

SNDR=60.1dB

LSB=0.82nA

Figure 4.13: Experimentally measured transfer characteristics of the impedance spectrum ana-lyzer channel for three sampling frequencies.

0

0 10 20 30 40

−100

−80

−60

−40

−20

0

FREQUENCY(Hz)

NO

RM

AL

IZE

D P

OW

ER

(d

B)

2nd HARMONIC

3rd HARMONIC

SFDR = 60.1dB

SNR = 78.2dB

50

Figure 4.14: Experimentally measured output spectrum of the ADC for a 10Hz sinusoidal input.

The 16-channel 1.2×1.6mm2 integrated impedance spectrum analyzer prototype

was implemented in a 0.13µm CMOS process with a 1.2 V supply. The die micro-

graph is shown in Fig. 4.12. Each channel consists of 4 WEs, with different WE sizes

(55µm×55µm, 5µm×5µm, and two 2µm×2µm) to cover a wide combined dynamic

range. Each column of four working electrodes shares a 45µm-wide Au reference and

counter electrodes driven by the on-chip waveform generator. Each channel consists of

the analog integrator, dual-slope multiplying ADC and in-channel bias and clock gen-

erator. Each channel WE can be set to a specific voltage independently, allowing for


10−3

10−2

10−1

100

0

10

20

30

40

50

60

MULTIPLICATION FACTOR (D)

SNDR (dB)

Figure 4.15: Experimentally measured MADC SNDR vs. the multiplication factor.

0 32 64 96 128 160 192 224 256

INPUT CODE

DN

L (

LS

B)

-1

-0.5

0

0.5

1

1.5

SIGNAL GENERATOR OUTPUT

Figure 4.16: Waveform generator DNL.

simultaneous detection of different DNA strands.

The digital output of one channel for the input current swept between 100fA and

400nA is shown in Fig. 4.13. The static input dynamic range is 140dB cumulatively for

the three sampling frequency settings, or 55.4dB at 1MHz clock. The dynamic range

for each setting is computed by taking the ratio of the maximum signal that saturates

the ADC to the LSB for a given sampling frequency setting. Dynamic performance

of an entire channel was measured by applying a 10Hz full scale (400nA) sinusoidal

input current with the ADC clocked at 10MHz. Fig. 4.14 shows the 65536-point FFT

of the measured ADC output. The strong second harmonic is due to the single-ended

architecture of the ADC. The resulting effective number of bits (ENOB) is 9.3.


0 32 64 96 128 160 192 224 256

−2

−1. 5

−1

−0.5

0

0.5

1

1.5

2

INL

(L

SB

)

INPUT CODE

SIGNAL GENERATOR OUTPUT

Figure 4.17: Waveform generator INL.

10−1

100

101

102

103

104

105

36

38

40

42

44

46

48

FREQUENCY (Hz)

THD (dB)

Figure 4.18: Total harmonic distortion of waveform generator DAC.

The effect of multiplication of the ADC analog input by a digital coefficient on

the SNDR is studied by measuring the ADC SNDR as the multiplication coefficient is

varied from D = 1 to 0. Fig. 4.15 shows the SNDR of the ADC versus the multiplication

coefficient. The ADC SNDR stays above 30dB down to the multiplication coefficient

of 1/70.

The DNL and INL of the waveform generator are shown in Fig. 4.16 and Fig. 4.17

respectively. The waveform generator achieves an static accuracy of 8 bits. Dynamic

performance of the waveform generator was characterized by measuring the total har-

monic distortion of the DAC versus frequency. Fig. 4.18 shows the THD of the mea-

sured DAC output. The DAC archives a THD of greater than 42dB over the operating

frequency range.


Table 4.4: Experimentally Measured Characteristics

Technology 0.13µm CMOSSupply Voltage 1.2VArea 1.2mm×1.6mmArray Dimensions 4×4 channelsChannel Size 300µm×200µmSensitivity 320fAChannel SNR (fclk=10MHz) 78.2dBChannel SFDR (fclk=10MHz) 60.1dBPower Consumption

DAC 1.1mWSRAM 1.3µWAnalog Integrator 8µWComparator 19µWBiasing 4µWDigital 11µWTotal (channel) 42µW

The electrode model shown in Fig. 1.9 was used first to emulate the biosensor. The

value of RS was set to 1GΩ, CWE was set to 500pF, CDB was set to 300pF and RCT

was set to 1MΩ. To verify the impedance extraction capability of the microsystem, a

sinusoidal voltage stimulus (generated by the on-chip DAC) with the frequency swept

from 0.1Hz to 10kHz was applied to the biosensor model. Figs. 4.19 (a) and (b)

demonstrate that the fabricated prototype tracks the theoretical model well over the

full range of frequencies. The absolute value of the relative error of the ideal biosen-

sor impedance verses the measured biosensor impedance as a function of frequency is

shown in Fig. 4.20. The absolute value of the relative error stays below 8.4% and 7.5%

for real and imaginary component of the biosensor respectively.

Table 4.4 provides a summary of experimentally measured characteristics of the

integrated impedance spectroscopy microsystem.


RE

AL

IMP

ED

AN

CE

(Ω

)

10 5

10 6

10 7

10 8

10 9

IDEAL

MEASUREDIM

AG

IMP

ED

AN

CE

(Ω

)

10 4

10 5

10 6

10 7

10 8

10 9

10 10

IDEAL

MEASURED

500pF1GΩ

1MΩ

WERE300pF

500pF1GΩ

1MΩ

WERE300pF

−110

010

110

210

3

FREQUENCY (Hz)

10 104

(a)

−110

010

110

210

3

FREQUENCY (Hz)

10 104

(b)

Figure 4.19: Off-chip biosensor model impedance as a function of frequency experimentallymeasured by the impedance spectroscopy microsystem: (a) real, and (b) imaginary components.

4.6 Analog vs Digital vs Mixed-Signal Multiplication

Multiplication of the input current by a digital coefficient is implemented by modulat-

ing the counter-controlled duration of the charging phase of the ADC, as described in

Section 4.4.3.

The number of extra logic gates required for implementing in-ADC multiplication

in this design is 23. They are not required to be clocked at the ADC clock speed. The

digital gates required for the multiplications make a state transition only twice during

one entire ADC conversion cycle. For example, if the ADC is generating 10ksps while

being clocked at 10MHz, then the digital logic will be clocked at 20kHz. As a result, the

extra logic cells do not have a significant contribution to the overall power consumption

of the recording channel.


0

2

4

6

8

10

|REA

L R

ELA

TIV

E ER

RO

R| (

%)

4−110

010

110

210

31010

4−110

010

110

210

31010

FREQUENCY (Hz)

0

2

4

6

8

10

|IMA

G R

ELA

TIV

E ER

RO

R| (

%)

FREQUENCY (Hz)(a)

(b)

Figure 4.20: Absolute value of the relative error of the off-chip biosensor model impedance asa function of frequency: (a) real, and (b) imaginary components.

Mixed-signal multiplication approaches presented here can be compared to the ana-

log and digital multiplication approaches in term of area, power and linearity.

Analog multiplication requires a frontend circuit that performs the multiplications

before the signal is passed to the ADC. This will result in added noise, linearity degra-

dation and area requirement which otherwise do not exist in the mixed-signal or digital

approach. Compared to the analog design shown in Fig. 1.16, our approach results in

32µm2 area saving (0.5 percent of total area) and 6.73µW power saving (15.2 percent

of total power) per recording channel. Also, in terms of non-linearity degradation, the

purely analog multiplication approach in Fig. 1.16 adds 4.2 percent linearity degrada-

tion to the channel. This would be in addition to the degradation of the ADC ENOB

(shown in Fig. 4.15) due to the reduction of the signal amplitude vs multiplication


−1

100

101

102

103

103

104

105

106

10 7

FREQUENCY (Hz)

IMA

G I

MP

ED

AN

CE

(Ω

)

10

103

104

105

106

107

104

−1

100

101

102

103

FREQUENCY (Hz)

10 104

RE

AL

IM

PE

DA

NC

E (

Ω)

(a)

(b)

1.0mM K [Fe(CN) ]3 6

0.1mM K [Fe(CN) ]3 6

1.0mM K [Fe(CN) ]3 6

0.1mM K [Fe(CN) ]3 6

Figure 4.21: Potassium ferricyanide solution impedance as a function of frequency experi-mentally measured by the impedance spectroscopy microsystem: (a) real, and (b) imaginarycomponents.

coefficients.

The main disadvantage of the digital multiplication vs mixed-signal multiplication

approach is the excessive area required by the digital multiplier. For example the digital

multiplier shown in Fig. 1.17 requires 436 logic cells. This results in an area of 0.015

mm2 (adding up the area required by the digital cells listed in Fig. 1.17 scaled to CMOS

0.13µm technology as listed in Table. 4.5 ). The total channel area in this design is 0.06

mm2, thus the area required by the digital multiplier (0.015 mm2) would correspond to

25 percent of the channel area. By comparison the digital overhead (23 gates with a

total area of 0.0009mm2) required by the ADC for multiplication in the mixed signal

approach corresponds to 1.5 percent of the total channel area.

The power consumption of the digital multiplier shown in Fig. 1.17 is 550pJ/MMPS.


Table 4.5: Serial Digital Multiplier Area Breakdown

Gate Type Number of Gates Area/Gate (µm2) Total Gate Area (µm2)AND12 64 1.74107E − 11 1.11429E − 09AOI21 37 3.48214E − 11 1.28839E − 09AOI22 16 5.22321E − 11 8.35714E − 10DFF 32 1.67143E − 10 5.34857E − 09

MUX21 32 1.85714E − 11 5.94286E − 10NAND21 28 2.90179E − 11 8.125E − 10NOR21 54 3.71429E − 11 2.00571E − 09OAI21 27 3.76071E − 11 1.01539E − 09OAI22 15 5.64107E − 11 8.46161E − 10XOR21 32 1.67143E − 11 5.34857E − 10XNOR21 64 1.85714E − 11 1.18857E − 09

TotalMultiplierArea(mm2) 1.55844E-08

Simulations shows that the ADC logic required for digital multiplication consumes

40pJ/MMPS. The total power consumed by the fully digital multiplier or by the extra

logic required by the MADC is not a significant portion of total channel power con-

sumption. The main advantage of the digital multiplier vs the mixed-signal multiplier

is that the multiplication is done after the ADC and the multiplication does not degrade

the ADC ENOB. In our application the degradation of the ADC ENOB does not re-

sult in a significant error in the system level performance. As shown in Figs. 4.20, in

the impedance extraction experiment, the absolute value of the relative error due to the

ADC ENOB degradation stays below 8.4% and 7.5% for real and imaginary component

of the biosensor respectively.

A comparison of the mixed-signal vs analog vs digital multiplication is shown in

Table 4.6. The mixed-signal multiplication approach presented in this work results in

small area and power requirements compared to the fully analog and fully digital multi-

plication approach. The mixed signal approach does not result in linearity degradation

as in the case of the analog multiplication approach. The fully digital approach provides

better system accuracy at the cost of excessive area requirement.


Table 4.6: Comparison of Mixed-Signal vs Analog vs Digital Multiplication

Performance Metric Mixed-Signal Analog DigitalArea Small Small Large

Power Small High SmallLinearity Degradation No Yes NoENOB Degradation Yes Yes No

4.7 Electrochemical Experimental Results

A set of electrochemical experiments have been conducted to validate the function-

ality of the impedance spectroscopy microsystem. In these tests a 3-electrode setup

has been utilized with on-chip gold-plated 55µm×55µm working electrode, on-chip

gold plated counter electrode and an off-chip Ag-AgCl reference electrode. The refer-

ence and counter electrodes are driven by the on-chip three-electrode regulation loop as

shown in Fig. 4.11(a) and all the excitation waveforms are generated using the on-chip

waveform generator.

First, in order to validate the design, impedance spectroscopy recordings of 0.1mM

and 1mM potassium ferricyanide in 1µM potassium phosphate buffer (pH 7.3) were

carried out. A 9mV 0.1Hz to 10kHz sine wave was applied between the WEs and an

off-chip Ag-AgCl reference electrode. The real and imaginary impedance results ob-

tained from the two concentrations of the potassium ferricyanide solution are shown in

Figs. 4.21 (a) and (b). An increase in the concentration of the potassium ferricyanide

results in a decrease in value of RS and RCT and increase in value of CWE . The mea-

surements of potassium ferricyanide validate impedance spectroscopy microsystem in

DNA sensing applications.


Tabl

e4.

7:C

ompa

rativ

eA

naly

sis

ofR

epor

ted

Am

prom

etri

cSe

nsor

yM

icro

syst

em

Syst

emIS

SCC

08JS

SC08

JSSC

09IS

SCC

10IS

SCC

10T

BIO

CA

S07

Thi

sW

ork

[20]

[26]

[84]

[131

][9

7][6

9]Te

chno

logy

(CM

OS)

0.18

µm

0.25

µm

0.5µ

m0.

35µ

m0.

6µm

0.5µ

m0.

13µ

mPo

wer

25m

W16

0mW

0.6m

W84

.5m

W—

1.2m

W0.

35m

WSu

pply

Volta

ge5.

0V2.

5V3.

0V3.

3V3.

3V3.

0V1.

2VC

hip

Are

a11

.2m

m2

15m

m2

2.25

mm

24m

m2

25.8

mm

29m

m2

1.92

mm

2

Ele

ctro

deC

ount

576

4810

010

040

1664

Cha

nnel

Cha

nnel

Cou

nt24

1610

010

040

1616

Type

Of

2D2D

2D2D

—O

ff2D

Ele

ctro

des

Poly

mer

Gol

dG

old

Gol

d—

Chi

pG

old

Sens

ing

Prot

ocol

CV

CV

ISIS

CV

CA

ISPo

wer

—10

mW

6µW

0.84

mW

—4.

3µW

42µ

WD

ynam

icR

ange

—60

dB50

dB97

dB80

dB—

140d

B(3

-mod

e)C

onve

rsio

nR

ate

10H

z10

kHz

10kH

z—

1Hz

10H

z10

kHz

Sens

itivi

ty(L

SB)

97pA

240p

A—

330p

A25µ

V—

320f

AE

NO

B11

bits

9bi

ts8

bits

No

AD

C12

bits

8bi

ts9.

3bi

tsW

avef

orm

Gen

erat

or—

—Y

esY

es—

—Y

esTy

pe—

—Sq

uare

Wav

eSq

uare

Wav

e—

—8-

bitP

rogr

amm

able

Freq

uenc

y—

—10

KH

z50

MH

z—

—10

kHz

Pow

er—

——

——

—1.

1mW

(5nF

Loa

d)


The SoC was also extensively validated in DNA analysis for detection of prostate

cancer synthetic DNA. Fig. 4.22(a) shows real and imaginary impedance components

of an on-chip Au electrode hybridized with 5µM single-stranded prostate cancer syn-

thetic probe DNA. Adding a 5µM noncomplementary target DNA does not significantly

change the real and imaginary components of the electrode impedance indicating that

non-specific adsorption is negligible. Adding a 5µM complementary prostate cancer

synthetic target DNA leads to significant reduction of potassium ferrocyanide redox

current, resulting in increase in both real and imaginary components of the electrode

impedance. This is due to the additional negative surface charge resulting from forma-

tion of double-stranded DNA on the electrode surface. The 3-sigma error bars (from 3

chips) with the real and imaginary impedance detection noise margins of approximately

18.3KΩ and 20.9MΩ respectively are shown in Fig. 4.22(b).

Table 4.7 provides a comparative analysis of the presented design and existing

amperometric biochemical sensory microsystems. The design presented in this work

achieves the highest dynamic range and the lowest sensitivity in terms of LSB. Also,

the design presented in this Chapter requires less time to extract real and imaginary

components of the biosensor compared to the FRA implementations in which a square

waveform is utilized for multiplication and integration. The stepwise approximation of

the interrogation waveform and multiplication coefficients significantly reduces the sys-

tematic error caused by higher order harmonics present in the rectangular waveforms,

thus eliminating the need for excessive averaging that is time consuming at the lower

frequency range. This enables real-time monitoring and analysis of DNA hybridization.


104

105

106

107

108

109

1010

10−1

100

101

102

103

104

FREQUENCY (Hz)

RE

AL

IM

PE

DA

NC

E (

Ω)

PROBE

NONCOMPLEMENTARY TARGET COMPLEMENTARY TARGET

106

107

RE

AL

IM

PE

DA

NC

E (

Ω)

COMPLEMENTARY

TARGET DNA

NONCOMPLEMENTARY

TARGET DNA

PROBE

DNA

COMPLEMENTARY

TARGET DNA

NONCOMPLEMENTARY

TARGET DNA

PROBE

DNA

DETECTION

NOISE MARGIN

DETECTION

NOISE MARGIN

104

105

106

107

108

109

1010

107

108

IMA

G I

MP

ED

AN

CE

(Ω

)

AT 10 HzAT 10 Hz

(a)

(b)

IMA

G I

MP

ED

AN

CE

(Ω

)

Figure 4.22: (a) Impedance spectrum of 5µM prostate cancer synthetic DNA probe, comple-mentary, non-complementary targets and (b) the corresponding 3-sigma error bars (from 3 chips,20 measurements each).

4.8 Chapter Summary

A 54-channel, mixed-signal CMOS impedance spectroscopy DNA analyzer is pre-

sented. It consists of a programable waveform generator, on-chip memory and multiple

impedance extraction units. Multiplication and integration, two operations required

for frequency response analysis (FRA) algorithm, are efficiently performed by the in-

channel current-mode input dual-slope multiplying ADC with negligible resources over-

head. The ADC combines impedance extraction and analog-to-digital conversion into a

single conversion cycle. The impedance spectroscopy microsystem was implemented in


a CMOS 0.13µm technology. Each channel occupies an area of 0.06mm2 and consumes

42µW of power from a 1.2V supply.

140

Chapter 5

Temperature Regulation

A 9×6-arrayed-cell mixed signal CMOS thermal controller for on-die DNA sensing is

presented. The SoC reuses the circuits for impedance spectroscopy and cyclic voltam-

metry to also perform temperature regulation. The on-chip, in-cell heating and temperature-

sensing elements are implemented in standard CMOS without any post-processing.

Using feedback proportional-integral-derivative (PID) control the temperature can be

regulated to within 0.5C of the desired value. The two computationally intensive oper-

ations, the multiplication and subtraction required by the PID algorithm, are performed

by the in-cell dual-slope multiplying ADC in the mixed-signal domain, resulting in

small area and low power consumption. Specifically, multiplication of the input current

by a digital coefficient is implemented by modulating the counter-controlled duration of

the charging phase of the ADC. Subtraction is implemented by the ADC counter. The

3mm×3mm prototype fabricated in a 0.13µm CMOS technology has been fully ex-

perimentally characterized. Each channel occupies an area of 0.06mm2 and consumes

42µW from a 1.2V supply.

CHAPTER 5. TEMPERATURE REGULATION 141

5.1 Introduction

In this Chapter we present a 0.13µm CMOS distributed temperature regulator with an

on-chip mixed-signal PID controller. The hybrid analog-digital nature of the PID con-

troller yields a compact implementation with a low power dissipation overhead. This

enables distributed localized temperature control in large-scale temperature-dependent

parallel sensing applications such as on-CMOS amperometric electrochemical DNA

analysis essays.

The presented temperature controller is a part of a highly-integrated multi-functional

SoC that performs amperometric electrochemical DNA analysis. The DNA analysis

SoC consists of 54 recoding channels and 600 gold microelectrodes. It performs label-

free DNA analysis and pH sensing for prostate cancer detection. We previously reported

on the DNA sensing functionality of this SoC [137]. Accurate DNA analysis requires

distributed temperature control. We reported a brief preliminary summary of how the

DNA sensing circuits and the temperature control circuits are integrated and shared on

the same die [87]. Here we present a detailed report on the VLSI architecture and circuit

implementation of the distributed temperature regulator that extends on [87] and offer

an in-depth treatment of the algorithm design and the circuit implementation, as well as

extensive experimental results illustrating temperature regulation characteristics.

The distributed temperature controller consists of a spatial array of 54 heating ele-

ments and 54 temperature-sensing elements, one such pair for each DNA sensing chan-

nel. The heaters and temperature sensors are interfaced to a distributed mixed-signal

PID controller that regulates the temperature of the chip. The PID controller re-uses

key circuits from the DNA-sensing channel, thus significantly saving silicon area. The

re-used DNA sensing circuits are reconfigured to perform both temperature sensing and

computation needed by the PID temperature control algorithm. This architecture allows

for independent thermal regulation of each channel. Channels can be further thermally


isolated from each other using a CMOS post-processing technique as described in [170].

In that method a combination of thin-film metal deposition, photolithography and wet

etching is utilized to etch out the dielectric and the bulk silicon around the area of the

interest and thus thermally isolating it from the rest of the chip.

+u(k) = u(k-1) + c e(k)

+ c e(k-1) + c e(k-2)

DESIRED TEMP, t(k)

+

e(k)y(k)

u(k)(c ,c ,c )

0 1 2 PWM

ACTUATOR

PID CONTROLLER

0

1 2

RE

TEMPERATURE

SENSOR

RE

HEATER

TEMPERATURE CONTROLLER

Figure 5.1: Block diagram of PID implementation of the temperature regulation loop.

The rest of the Chapter is organized as follows. Section II provides background on

temperature regulation principles. Section III describes the temperature-sensing princi-

ple. Section IV presents the VLSI architecture of the temperature controller. Section V

details the circuit implementation of the VLSI architecture. Section VI demonstrates

the experimental results obtained from the 0.13µm CMOS prototype.

5.2 Thermal Control Principles

5.2.1 Temperature regulation

The block diagram of the temperature regulation loop is shown in Fig. 5.1. It consists

of a heater, a temperature sensor, a temperature controller, and an actuator as the feed-

back element. In this work, a proportional-integral-derivative (PID) algorithm has been

chosen for the temperature controller, since this class of algorithms is effective in the

regulation of thermal systems [162, 163]. As shown in Fig. 5.1, a discrete-time PID

controller calculates the error value, e(k), as the difference between a measured input

signal, y(k), and a desired input signal, t(k). The controller attempts to minimize the


error by adjusting the input to the feedback element (actuator), u(k), using the process

control inputs (c0, c1, c2). The PID controller algorithm involves three separate con-

stant parameters, namely, the proportional, the integral, and the derivative components,

which are denoted as P, I, and D, respectively. The proportional component depends on

the present error value, the integral component depends on the past error value, and the

derivative component is a prediction of the future error value. The weighted sum of the

three components can be used to adjust the heat, and thus regulate the temperature of

the microsystem [164].

IREF

IN

VBE1

R1

ᾳ∆VBE

R2

DIG

ITA

L

TE

MP

ER

AT

UR

E

VBE1

VBE2

∆VBE

ᾳ∆VBE

R2

VBE1

R1

V

T T

I

T

I

VBE1

R1

ᾳ∆VBE

R2

(b) (c) (d)

(e)

CHANNEL

I

CTAT

PTAT

(a)

E

C

B

VBE2+

-E

C

B

VBE1

+

-IE1

IE2

IC1

IC2

Q 1

Q 2

IB1

IB2

∆VBE

+ -

Figure 5.2: (a) A pair of BJTs for generating ∆VBE and VBE , (b) Temperature dependenceof the base-emitter voltage, (c) generation of PTAT and CTAT current, (d) generation of tem-perature dependent current and (e) utilization of a current-to-digital channel for temperaturemeasurements.

In this work the temperature of the sensing site is measured using an on-chip temper-

ature sensor. The output of the temperature sensor, y(k), is subtracted from the desired

temperature, t(k), and the resulting error value, e(k), is fed to the PID controller. The

output of the PID controller, u(k), sets the duty cycle of a digital pulse that is gener-

ated by the on-chip digital PWM controlling the on-chip heater. The PID controller is

implemented as a recursive filter [136]. The continuous-time representation of the PID


controller is given by

u(t) = Kpe(t) +Ki

∫ t

0

e(t)dt+Kdd

dte(t) (5.1)

where Kp is the proportional gain, Ki is the integral gain, Kd is the derivative gain, e(t)

is the error value, and u(t) is the actuation value. A discrete-time representation of (5.1)

in Z-domain is given by

D(z) =u(z)

e(z)= Kp +Ki

Ts

2

z + 1

z − 1+

Kd

Ts

z − 1

z(5.2)

where Ts is the sampling period. Equation (5.2) can be implemented numerically as

follows

u(k) = u(k − 1) + c0e(k) + c1e(k − 1) + c2e(k − 2) (5.3)

where c0, c1, and c2 are the PID coefficients, and k is the discrete time index.

5.2.2 Temperature sensing

Temperature can be measured using two signals linearly dependent on temperature.

Such a temperature sensor measures the temperature by taking the ratio of these two

signals. In a CMOS technology, these signals can be derived from the base-emitter

voltage of substrate pnp transistors [165].

These transistors consist of the p-substrate as the collector, the p+-diffusion as the

emitter, and a n-well as the base. Two signals linearly dependent on temperature can be

generated from the base-emitter voltage (VBE) of a single pnp transistor and from the

difference between the base-emitter voltages (∆VBE) of two pnp transistors biased at

different collector current levels as shown in Fig. 5.2(a). The temperature dependence


of the VBE can be derived from the exponential relation between the collector current

IC and the base emitter voltage as follows

VBE(T ) = Vg0(1−T

Tr

) +T

Tr

VBETr

− nKT

qln

T

Tr

+KT

qln

ICT

ICTr

(5.4)

where Vg0 is the extrapolated bandgap voltage at 0 Kelvin, n is a process-dependent

constant, K is Boltzmann constant, q is the electron charge, and Tr is an arbitrary refer-

ence temperature [159]. As illustrated in Fig. 5.2(b), VBE is complementary to absolute

temperature (CTAT), with a typical slope of − 2 mV/K. The nonlinearity, or curvature,

is represented by the last two terms of (5.4). These nonlinearities are negligible in the

temperature range of 20oC to 90oC that is required by the DNA analysis SoC.

The slope of the base-emitter voltage depends on the absolute value of the collec-

tor current. This dependence can be used to generate a voltage that is proportional to

absolute temperature (PTAT), as shown in Fig. 5.2(b). The difference between the base-

emitter voltages of two pnp transistors biased at two different collector currents can be

expressed as follows

∆VBE(T ) = VBE2(T )− VBE1(T ) =KT

qln(

IC2

IC1

) (5.5)

where IC1 and IC2 are the collector currents of the two pnp transistors. As can be seen

in (5.5), if the collector current ratio is constant, then the difference between the base-

emitter voltages of the two pnp transistors is proportional to the absolute temperature

(PTAT).

A typical temperature sensor measures the temperature by determining the ratio of

the PTAT voltage to a temperature-independent reference voltage. The temperature-

independent reference voltage can be generated by adding VBE to a scaled version of


∆VBE [159–161]. The scaling factor α is used to match the temperature dependance

(with an opposite sign) of VBE and ∆VBE . In this work a current-mode temperature sen-

sor is utilized. The PTAT and CTAT voltages are converted to the current domain, using

resistors R1 and R2 as shown in Fig. 5.2(c). Also, instead of generating a temperature-

independent reference signal, the ratio of α∆VBE/R2 to VBE/R1 is determined, which

has a linear temperature dependence, as shown in Fig. 5.2(d). This eliminates the need

for a temperature-independent reference current generator.

In this work a current-to-digital recording channel is utilized for temperature sens-

ing. The channel operates by digitizing the ratio of the input current to the reference

current. As shown in Fig. 5.2(e), the CTAT current is used as the input to the channel,

and the PTAT current is used as the reference of the channel. The current-to-digital

channel digitizes the ratio of these two currents. The factor α, the resistance R1 and R2

are set such that both the input current and the reference current utilize the full input

dynamic range of the channel over the operating temperature range.


I PTA

T

I CTA

T×5

4V

BE+

-

×5

4

CC

I RE

F

I IN S

CA

LE

D D

ES

IRE

D T

EM

PE

RA

TU

RE

PW

M

H

EA

TE

R

u(k

)

×5

4

CT

AT

c e

(k)

n

(c t

(k),

c t

(k),

c t

(k))

01

2

B0

Z-1

C y

(k)

n

CT

AT

CU

RR

EN

T

S

OU

RC

E

PT

AT

CU

RR

EN

T

S

OU

RC

E

SE

NS

OR

×5

4

B1

B2

B3

B4

B5

B6

MA

DC

CH

AN

NE

L

Q3 Q

1Q

2

×5

4

PT

AT

SE

NS

OR

M

UL

TIP

LIC

AT

ION

CO

EF

FIC

IEN

TS

( c

, c

, c

)

01

2

7-B

IT S

RA

M

Figu

re5.

3:To

p-le

velV

LSI

arch

itect

ure

ofte

mpe

ratu

rere

gula

tion

loop

(sha

ded

boxe

sar

ein

-cha

nnel

).


B0

B1

B2

B3

B4

B5

B6

c e(0)0

c e(1)0

c e(2)0

c e(0) 1

c e(2) 1

c e(0)2

c e(1) 1

c e(1)2

c e(2)2

u(0) u(1) u(2)

PID CYCLE

ADC CYCLE

TIME

Figure 5.4: Timing diagram of the temperature regulation loop.

5.2.3 Temperature Regulation

The block diagram of the wireless electrochemical DNA analysis SoC configured for

temperature regulation is shown in Fig. 5.1. In this mode, the SoC utilizes the 54

current-to-digital recording channels to measure temperature. The on-chip PID con-

troller is used to regulate each cell temperature. On-chip SRAMs are utilized to store

PID coefficients and multiplication coefficients. The channel measures temperature by

taking the ratio of a CTAT to PTAT currents as described in Section II. The CTAT

current is used as the input and the PTAT current is used as the reference to the current-

to-digital recording channel. The CTAT and PTAT currents are generated using the

in-cell BJTs. The measured temperature is fed to the on-chip PID controller. The PID

controller regulates the 2D chip temperature profile by modulating the in-cell heaters.

Two computationally intensive operations, the multiplication and subtraction required

by the PID algorithm, are performed by the in-channel MADC in the mixed-signal do-

main, yielding small area and power consumption, as described next.


S1

S1

CM

VCM

ICTAT

IPTAT

C

S2

CCV

VC

(c t(k), c t(k), c t(k))0 1 2

S2

CURRENT

CONVEYER

DUAL-SLOPE MULTIPLYING

ADC

INT

CALIBRATION

SRAM

2(S

1,S ,S

2)

CLK

RESET

UP/DN

UP/DOWN

COUNTERLOAD

CO

NT

RO

L

L

OG

IC

(c ,c ,c ) 0 1 2

c e(k) n

CLK

OUT_EN

PID_ENPID LATCH

0

1

S 2

D_IN

D_OUT

D_IN

RESET

S 3

S3

3

Figure 5.5: Top-level VLSI architecture of one temperature sensing channel.

5.3 VLSI Architecture

The top-level VLSI architecture of the temperature regulation loop is shown in Fig. 5.3.

The regulation loop consists of CTAT and PTAT BJTs, PTAT and CTAT current sources,

current-to-digital channel, a 7-bit SRAM bank with an adder, a 12-bit digital PWM, and

an in-cell heater. The in-channel heaters are implemented by a PMOS transistors with

width and length of 100µm and 0.13µm and a 15Ω load, as shown in Fig. 5.6. The

in-cell BJTs are interfaced sequentially to the current sources that generate IPTAT and

ICTAT . ICTAT is used as the input to the channel, and IPTAT is used as the reference

current. The channel determines the ratio of ICTAT to IPTAT . Both currents are scaled

such that their magnitude fits within the channel dynamic range over the operating tem-

perature range. The ratio of ICTAT to IPTAT results in a linear transfer characteristic

(versus temperature) [165]. The channel dual-slope multiplying ADC performs three

tasks. First, it computes the ratio of ICTAT to IPTAT . Next, it multiplies this ratio by the

PID coefficients, (c0, c1, c2), and finally it subtracts the results from the scaled version

of the desired temperature, (c0t(k), c1t(k), c2t(k)). The result of these three operations

is the computation of cne(k), as shown in Fig. 5.3. Next, these values are stored in the

on-chip 7-bit SRAM bank. An on-chip adder adds the appropriate values (according to

equation (3)) and computes u(k), which sets the duty cycle of the signal generated by


VIN 100μm/0.13μm

15Ω

Figure 5.6: In-channel heater schematic.

the digital PWM. The duty cycle of this signal sets the amount of time that the in-cell

heater is on, thus regulating the chip temperature.

The timing diagram of the PID regulation loop is shown in Fig. 5.4. The channel

dual-slope multiplying ADC performs three conversion cycles during one cycle of the

PID controller, thus eliminating the need for three parallel ADCs. As shown in Fig. 5.4,

in the first conversion cycle of the PID controller, the ADC computes c0e(0), c1e(0), and

c2e(0). Next, u(0) is calculated from these values, and the duty cycle of the digital PWM

is set accordingly. Consecutive values of c0e(n), c1e(n), c2e(n), and u(n) are generated

in the same manner.


5.4.1 Current-to-Digital Channel

The top-level VLSI architecture of the current-to-digital channel is shown in Fig. 5.5.

Each channel consists of a chopper-stabilized current conveyer [166], a dual-slope mul-

tiplying ADC [167], control logic, and on-chip memories and a counter. The dual-slope

multiplying ADC performs both multiplication and subtraction, as required by the PID

algorithm.

The timing diagram of the ADC for a typical conversion cycle is shown in Fig. 5.7.

The conventional dual-slope ADC digitizes the input signal in two phases: charging


VC

1

c T1

TRD

c T2

RESET

Time

S

n n

2

S 3

S

Figure 5.7: Timing diagram of the dual-slope multiplying ADC.

phase T1 and discharging phase T2, of the integrating capacitor. The ratio of the duration

of the charging phase to the discharging phase represents the input signal level. As

shown in Fig. 5.7, to implement multiplication of the input current by a digital PID

coefficient cn, as required by the PID algorithm, the duration of the charging phase is

scaled with a constant coefficient cn<1 [167]. In this case, by counting the time cnT2,

a digital representation of cnICTAT can be obtained.

As shown in Fig. 5.7, first the ADC counter is reset. At the same time, the PID

multiplication coefficient, cn, is loaded into the in-channel PID latch (S3=1). There is

a large channel-to-channel variation between the BJT current output. In this work to

compensate for this variation, the channel counter is pre-loaded with a digital calibra-

tion value (stored in the in-channel calibration SRAM) at the rising edge of the RESET

signal. Next, the in-channel counter counts up from this value to time cnT1, and the in-

put current ICTAT is integrated onto the capacitor CINT . After time cnT1, the voltage on

the capacitor is held constant for a fixed time interval, TRD. During this time, the digital

representation of the desired temperature that is scaled by the PID coefficients, cnt(k),

is loaded into the counter. During time cnT2, the integrating capacitor is discharged

with the PTAT current source. During time cnT2, the counter counts down, and the final

value of the counter is available at the falling edge of S2 signal. By counting down from

cnt(k) in the second phase of the conversion cycle, the ADC performs subtraction of


Table 5.1: OTA Transistor Sizing

Transistor W/L (µm)

M1,2 9× 4/1M3,4 4× 1.5/4M5,6 5× 1.5/1M7,8 10× 1.5/1M9,10 4× 1.5/4M11 11× 1.5/2

cny(k) from cnt(k) (as shown in Fig. 5.3) thus computing the error signal cne(k).

The front end bidirectional current conveyer is implemented by a PMOS and an

NMOS transistors connected in the feedback of an OTA [88]. The negative feedback

ensures a known potential at the working electrode is set by the voltage at the non-

inverting input of the OTA. It also enables the current conveyer to source and sink an

input current without the need for a DC offset current. The OTA is implemented as

a folded-cascode amplifier with PMOS input pair. Chopper-stabilization is utilized to

reduce the OTA flicker noise and offset and dynamic element matching is utilized to

reduce the output current mirrors mismatch [88].

The ADC comparator is implemented with three stages of pre-amplifiers, with a

total gain of 60dB and the last stage with a high-speed latch. The first stage of the

comparator is implemented as a cross-coupled diode-connected gain stage to provide a

moderate gain and a high frequency bandwidth. Chopper-stabilization suppresses the

input offset and ensures 9-bit accuracy. The second and third stages are identical to the

first one but with no chopping. The high-speed latch is implemented with an NMOS

input pair gain stage and a NMOS-PMOS cross-coupled load to provides high accuracy,

low offset and a high frequency bandwidth [144].


−

+

R1

IPTAT

6

VBEtrim

COEFFICIENTS

TRIMMING V P

10:1 DEM

R

VBEtrim

1

VBE

+

-

Q1

×54

CTAT

SENSOR

M 0 M 9 M 10

OTA1

PRBS GEN

Figure 5.8: Circuit diagram of the CTAT sensor and CTAT current source

V IN+

VIN-

V BIASN

V CASCP

CASCNVVOUT

M1 M2

M3 M4

M5 M6

M7 M8

M9 M10M11

CHOPM

Figure 5.9: Chopper-stabilized folded-cascode OTA in the CTAT (OTA1) and PTAT (OTA2)current sources.

5.4.2 CTAT Current Source

A simplified circuit diagram of the CTAT current source is shown in Fig. 5.8. To gen-

erate a digitally programmable base-emitter voltage VBEtrim (compensating for chip-

to-chip variations), a PTAT current (generated by a separate on-chip biasing circuit)

is passed through a 6-bit programmable resistor connected in series with the diode-

connected pnp transistor Q1. The current VBEtrim/R1 is generated by a voltage-to-

current converter, as shown in Fig. 5.8. A large resistor (R1>100MΩ) is required to

ensure that the CTAT current is kept relatively low (20nA) and stays within the dy-

namic range of the ADC. This would result in a large integration area. To reduce the


10−2

10−1

100

101

102

103

−130

−120

−110

−100

−90

−80

−70

FREQUENCY (Hz)

CHOPPER ON

CHOPPER OFF


rms

1/2

Figure 5.10: Simulated input-referred noise spectrum of the OTA with chopper enabled anddisabled from 0.01Hz to 1kHz.

size of the resistor, a current mirror with a 10:1 ratio is utilized. A 10:1 dynamic ele-

ment matching is used to reduce the effect of the current mirror mismatch. In this work

R1 is set to 1.5MΩ.

To reduce the effect of flicker noise, OTA1 amplifier in Fig. 5.8 utilizes input PMOS

devices with a high aspect ratio and internal OTA chopping. As shown in Fig. 5.9, the

chopper switches are placed at the input of the OTA. Another set is placed below the

cascoded NMOS transistors in the current mirror. This placement reduces the flicker

noise and DC offsets caused by the input pair transistors and the NMOS current mirror

transistors.

The simulated input-referred noise of OTA1 in the cases where the chopper is dis-

abled and enabled is shown in Fig. 5.10. The integrated input-referred noise from

0.01Hz to 1kHz is 0.11pA when the chopper is disabled and 0.06pA when the chopper

is enabled. The OTA1 transistor sizes are given in Table 5.1. The contribution of each

transistor to the total input-referred noise is shown in Fig. 5.11. When the chopper is

disabled, the main contributions to the input-referred flicker noise are from the OTA

current mirror transistors M3 and M4, and from the input pair transistors M1 and M2.


M (T)3,4

M (T)1,2

M (T)5,6

M (T)7,8

M (T)9,10

CHOPPER ON

CHOPPER OFF

0

0.005

0.010

0.015

0.020

0.025

0.030

INT

EG

RA

TE

D I

NP

UT

NO

ISE

( p

A )

(a)

M (F)3,4

M (F)1,2

M (F)5,6

M (F)7,8

M (F)9,10

0

0.005

0.010

0.015

0.020

0.025

0.030

INT

EG

RA

TE

D I

NP

UT

NO

ISE

( p

A )

0.035

CHOPPER ON

CHOPPER OFF

(b)

Figure 5.11: Simulated OTA noise summary: (a) thermal noise contribution, and (b) flickernoise contribution.

When the chopper is enabled, the current mirror transistors M9 and M10 are the main

contributors to the input-referred flicker noise.

5.4.3 PTAT Current Source

A simplified circuit diagram of the PTAT current source is shown in Fig. 5.12. The

∆VBE is generated by determining the difference between the VBE of the two substrate

pnp transistors that are biased at 3:1 collector current ratio. The voltage-to-current con-

verter consisting of OTA2 and the NMOS feedback transistor generates ∆VBE across

R2 (split into R2/3 and 2R2/3). In this work R2 is set to 100kΩ. This results in the

desired output current ∆VBE/R2. The resistor R2/3 is added in series with the base of

the Q3 transistor in order to ensure that the base current of the Q3 does not affect the

output current.

The main causes of inaccuracy in the PTAT current source are the input offset of

OTA2, OTA2 flicker noise, and the mismatch in the bias current mirror. Internal OTA


−

+

IBAIS

3 I I ∆VBE

R2

Q2 Q3

×54

PTAT

SENSOR

R13

R232 2

OTA2

DEM

DEM

∆VBE+ -

Figure 5.12: Circuit diagram of the PTAT sensor and PTAT current source.

chopping is utilized to reduce the effect of flicker noise and input offset voltage. OTA2

has the same amplifier topology and transistor size as those used in the CTAT current

source. Dynamic element matching at 100Hz, by the means of the chopper switches in

series between the current sources and the resistors, is utilized to improve the matching.

5.4.4 Digital Pulse Width Modulator

In the temperature regulation loop, the digital pulse width modulator (PWM) sets the

duty cycle of the pulse that is controlling the in-channel heater. The achievable discrete

set of the duty cycle settings of the pulse depends on the digital PWM resolution. In

this work, a 12-bit digital PWM architecture is selected. The design is based on a hy-

brid delay-line/counter architecture in [168]. The block diagram of the PWM is shown

in Fig. 5.13. In this architecture, a 7-bit counter and 32-stage ring oscillator are used to

achieve the 12-bit resolution. At the beginning of a switching cycle, the output set-reset

(SR) flip-flop is set, and the PWM output pulse goes high. The pulse that propagates

through the ring at the oscillation frequency serves as the clock for the counter. At the

time when the counter output matches the top most significant bits of the digital input


(nc), and a pulse reaches the tap selected by the least significant bits (nd), the output

flip-flop is reset, and the output pulse goes low [168].

OUT

32 :1

.

32-STAGE RING OSCILLATOR7-BIT

COUNTER

R

SQ7-BIT

COMP

7-BIT

COMPInput [11:0]

n [4:0]d

n [6:0]c

n [6:0]f

. .

n [6:0]f

Figure 5.13: Block diagram of the digital pulse width modulator.

5.5 Experimental Results

9×6 ARRAY OF

CELLS

CONTROL

LOGIC

PULSE WIDTH

MODULATOR

PID

CONTROLLER

PTAT & CTAT

CURRENT SOURCES

CALIBRATION

SRAM

BIAS

CURRENT

CONVEYERCOMPARATOR

HEATER

CONTROL

LOGIC

PID SRAM

COUNTER BJTs

ADDER

PID LATCH

Figure 5.14: Die micrograph of the 54-cell wireless temperature regulated DNA analysis SoC.


0 5 10 15 20 25 30 35 40 45 50−140

−120

−100

−80

−60

−40

−20

0

2nd Harmonic3rd Harmonic

SFDR = 58.7 dB

NORMALIZED POWER(dB) SNR = 56.9 dB

FREQUENCY(Hz)

I = 350nAin

ADC

OUTPUT

SPECTRUM

Figure 5.15: Experimentally measured output spectrum of the ADC for a 15Hz sinusoidal input.

The die micrograph of the mixed-signal CMOS thermally-controlled DNA-sensing

SoC is shown Fig. 5.14. The 54 cells are arranged in a 9×6 array on a 3mm×3mm

0.13µm CMOS die. Each cell consists of a current conveyer, a dual-slope multiplying

ADC, an in-cell bias and clock generator, a pH sensor, a heater, and temperature-sensing

BJTs. The PID controller, CTAT and PTAT current sources, and the digital PWM are

located in the top section of the die.

0 500 1000 1500 2000 2500 3000 3500 40000

20

40

60

80

100

PWM INPUT CODE

OU

TP

UT

DU

TY

RA

TIO

(%)

Figure 5.16: Experimentally measured transfer function of the pulse-width-modulator.

The dynamic performance of the entire cell was measured by applying a 15Hz full-

scale (400nA) sinusoidal input current, with the ADC clocked at 10MHz. Fig. 5.15


20 30 40 50 60 70 80200

400

600

800

TEMPERATURE ( C)

OU

TP

UT

DIG

ITA

L C

OD

E

o

90

Figure 5.17: Experimentally measured digital output of the dual-slope ADC vs temperature(in-air).

shows the 65536-point FFT of the measured ADC output. The strong second harmonic

occurs due to the single-ended architecture of the ADC. The resulting effective number

of bits (ENOB) is 9.3.

The digital PWM is characterized by sweeping the digital input code from 0 to

4095 and clocking the digital PWM at 2.5MHz. The experimental results of Fig. 5.16

show the measured duty ratio of the output pulses as a function of the 12-bits digital

input. The minimum (4%) and the maximum (96%) duty ratios are set by design. The

digital PWM achieves a linear transfer function over the programable digital input code

with a 0.82 percent maximum error. The digital PWM achieves a minimum duty cycle

resolution of 0.1µS.

The TP04390A ThermoStream [169], as shown in Fig. 5.18, is utilized to charac-

terize the temperature response of the CMOS temperature sensor. In these experiments,

an external temperature sensor is placed on top of the CMOS chip. The TP04390A

ThermoStream uses the temperature reading from the external temperature sensor to set

the chip temperature to a desired value. In these experiments the CMOS chip is ther-

mally isolated using a rubber thermal insulation pad (DF-32 from Liwung) to ensure a

consistent temperature gradient.

The output digital code of the ADC versus temperature, from 20 to 90 degrees


Figure 5.18: TP04390A ThermoStream [169] unit used for temperature sensor characterization.

Celsius, is shown in Fig. 5.17. The temperature sensor archives a linear transfer function

over the operating temperature range. Fig. 5.19 shows the measured temperature error of

seven dies from one wafer, which operated at a supply voltage of 1.2V. Their inaccuracy,

after calibration in the temperature range of 20 degrees Celsius to 90 degrees Celsius,

is less than 0.5 degrees Celsius.


0.0

0.30

ER

RO

R (

C

)o

30 40 50 60 70 80

TEMPERATURE ( C)

90o

0.15

0.45

-0.15

-0.30

-0.45

Figure 5.19: Temperature sensor error experimentally measured on seven dies from one wafer(in-air).

Table 5.2: Experimentally Measured Characteristics

Technology 0.13µm CMOSSupply Voltage 1.2VArea 3mm×3mmArray Dimensions 9×6 channelsChannel Size 300µm×200µmSensitivity 8.6pAPower Consumption(System)

SRAM 1.3µWPID Controller 21µWTemperature Core 12µWHeater 270mW @ 90 degree

Power Consumption(Channel)Current Conveyer 8µWComparator 19µWBiasing 4µWDigital 11µWTotal (channel) 42µW


NU

MB

ER

OF

OC

CU

RA

NC

ES

49.6 49.7 49.8 49.9 50.0 50.1 50.2 50.30

2

4

6

8

10

12

MEAN = 49.83 C

SD(3σ) = 0.20 C

N= 54

o

o

TEMPERATURE ( C)o

Figure 5.20: Experimentally measured temperature from 54 channels in one chip (in-air).


Tabl

e5.

3:C

ompa

rativ

eA

naly

sis

ofR

epor

ted

The

rmal

lyR

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Bio

sens

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stem

s

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emIS

SCC

08B

IOC

AS

09T

BIO

CA

S07

ISSC

C10

ISSC

C09

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sW

ork

[20]

[135

][1

32]

[25]

[97]

Tech

nolo

gy(C

MO

S)0.

18µ

m0.

5µm

0.5µ

m0.

13µ

m0.

6µm

0.13

µm

Pow

er25

mW

30m

W—

165m

W—

0.35

mW

Supp

lyVo

ltage

5.0V

5V5.

0V1.

2V3.

3V1.

2VC

hip

Are

a11

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m2

3mm

26m

m2

7.5m

m2

25.8

mm

29m

m2

Ele

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deC

ount

576

91

1640

600

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54Ty

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des

2D2D

2D2D

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Gol

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agne

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Yes

Yes

Yes

No

No

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espH

No

No

No

No

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TIS

FET

Tem

pera

ture

Sens

ing

Yes

Yes

Yes

Yes

Yes

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No

PID

PID

PID

PID

PID

—(A

nalo

g)(A

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l)(M

ixed

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nal)

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C


An external heater was utilized to fix the chip temperature at 50 degrees Celsius and

the temperature at each recording channel was recorded to study the effect of channel-

to-channel mismatch. Experimentally measured temperature from 54 channels in one

chip is shown in Fig. 5.20. The mean digital output temperature and the corresponding

standard deviation are 49.83 and 0.20 degree Celsius, respectively.

An example of the temperature regulation cycle in liquid (5mL 1M potassium phos-

phate buffer solution), with steps at 35, 45, 55, and 65 degrees Celsius, is shown

in Fig. 5.21. In these experiments the PWM was clocked at 5kHz and the ADC was

clocked at 10MHz to provide 3 samples (as needed by the timing diagram shown

in Fig. 5.7) in each PID cycle. The solid line is the chip temperature regulated by the

on-chip temperature regulator and the dashed line is the desired temperature. It takes

roughly 10 seconds to achieve a 5 degree increase in the chip temperature. Measured

absolute value of the relative error of the PID regulation loop over the operating tem-

perature range is shown in Fig. 5.22. The absolute value of the error stays below 0.75

degree Celsius over the operating temperature range. In these experiments the CMOS

chip was thermally isolated using a silicone rubber thermal insulation pad.

0 10 20 30 9025

35

45

55

65

75

TIME (S)

TE

MP

ER

AT

UR

E (

C)

o

40 50 60 70 80

DESIRED

TEMPERATURE

REGULATED

TEMPERATURE

Figure 5.21: Temperature regulation cycle with steps at 35, 45, 55 and 65 degree Celsius (in-liquid).

Table 5.2 provides a summary of the experimentally measured characteristics of the

integrated impedance spectroscopy microsystem.


20 30 40 50 60 70 800.10

0.30

0.60

0.90

TEMPERATURE ( C)o

90

0.15

0.45

0.75

|REL

ATI

VE

ERR

OR

| ( C

)

o

Figure 5.22: Measured absolute value of the relative error of PID regulation loop (in-air).

Table 5.3 provides a comparative analysis of the presented design and existing ther-

mally regulated biochemical sensory microsystems. The design presented in this work

utilizes a mixed signal temperature regulation loop that reuses components from the

DNA-sensing recording channel [137] to perform temperature regulation, thus saving

power and per-channel area as compared to the designs presented in Table 5.3.

5.6 Chapter Summary

A 54-channel, mixed-signal CMOS thermal-controlled, DNA-sensing SoC was devel-

oped. The DNA-sensing SoC shares circuitries to perform impedance spectroscopy,

cyclic voltammetry, and temperature regulation. The on-chip, in-channel heating and

temperature-sensing elements were implemented in standard CMOS without any post-

processing. Using feedback proportional-integral-derivative (PID) control, the temper-

ature can be regulated to within 0.5C of the desired point, a process that enables precise

control of on-chip DNA hybridization during characterization or sensing. Two compu-

tationally intensive operations, the multiplication and subtraction required by the PID

algorithm, are performed by an in-channel, dual-slope multiplying ADC in the mixed-

signal domain, and the result is minimal area and power consumption. The PID regu-

lation loop reuses circuitries from the DNA-sensing channel, and the result is minimal


area. Each channel occupies an area of only 0.06mm2 and consumes 42µW of power

from a 1.2V supply.

167

Chapter 6

Conclusions and Future Work

6.1 Contributions and Related Publications

This thesis describes a wireless thermally regulated label-free DNA analysis SoC with

nanostructured on-die electrodes. The summary of all contributions by chapter is pre-

sented next.

• Chapter 2

Chapter 2 describes the design, analysis and experimental results of two low-noise

bidirectional current acquisition circuits for interfacing with electrochemical ampero-

metric biosensor arrays. The first design is a switched-capacitor transimpedance am-

plifier (TIA) and the second design is a current conveyer (CC) with regulated-cascode

current mirrors. Both designs were fabricated in 0.13µm CMOS technology. The elec-

trical and electrochemical recording properties of both circuits have been characterized.

It is shown that the current conveyer exhibits superior performance in low-concentration

electrochemical catalytic reporter sensing. This work was presented at the 2011 IEEE

International Symposium on Circuits and Systems [87]. The results from this chapter

are published in IEEE Transactions on Circuits and Systems I [88].

CHAPTER 6. CONCLUSIONS AND FUTURE WORK 168

The main contribution of this chapter is the comparison of the switched-capacitor

transimpedance amplifier to the current conveyer approach for use as a current sens-

ing frontend and providing experimental proofs that the current conveyer (CC) is more

suitable for DNA sensing applications. It is shown through on-chip electrochemical

recording that the CC results in lower charge injection into the working electrode. The

charge injection can disturb the charge balance at the electrode-electrolyte interface

thus affecting the electrochemical reaction. It is shown through extensive electrochem-

ical recordings that the CC is more suitable for DNA sensing from small (2µm×2µm)

working electrodes.

• Chapter 3

Chapter 3 describes the fully integrated 54-channel wireless label-free DNA anal-

ysis SoC configured for cyclic voltammetry and pH sensing. The SoC includes 600

nanostructured DNA sensors and 54 pH sensors, and reuses key circuits for cyclic

voltammetry and pH sensing. The 3mm×3mm prototype fabricated in a 0.13µm stan-

dard CMOS technology has been validated in prostate cancer DNA detection. Each

channel occupies an area of only 0.06mm2 and consumes 42µW of power from a 1.2V

supply. This work was presented in 2012 Symposia on VLSI Technology and Circuits.

The main contribution of this paper is the fabrication of the nanostructured micro-

electrodes (NME) for the first time on a CMOS die. Also, we have characterized the

NME electrodes for the first time for on-CMOS DNA sensing. The detection limit of

10aM is the lowest reported in literature by a CMOS DNA detection system. Another

contribution of this chapter is the design and experimental demonstration of an elec-

trochemical recording channel that achieves the highest sensitivity and dynamic range

compared to designs previously reported in literature and also integration of a fully

digital UWB transmitter. The frontend current conveyer employs chopper stabilization

and dynamic element matching to achieve input-referred noise of less 0.13pA over the


operating bandwidth and to improve the current conveyer accuracy by 60 percent at the

100nA input current level. The in-channel SRAM allows for in-channel gain calibra-

tion (by adjusting the timing of the ADC) which results in a 17 percent improvement in

channel-to-channel ENOB variation. The UWB transmitter design is adopted from [77]

and the design presented here is optimized for our application by significantly simplify-

ing the previously reported implementations, thus achieving small area and low power

consumption.

• Chapter 4

Chapter 4 presents the fully integrated 54-channel wireless, label-free DNA analysis

SoC configured for impedance spectroscopy. The SoC utilizes frequency response anal-

ysis (FRA) to extract the real and imaginary impedance components of the biosensor.

Two computationally expensive operations, the multiplication and integration required

by the FRA algorithm, are performed by an in-channel dual-slope multiplying ADC

in the mixed-signal domain resulting in high accuracy, small area and low power con-

sumption. Multiplication of the input current by a digital coefficient is implemented

by modulating the counter-controlled duration of the charging phase of the ADC. In-

tegration is implemented by accumulating output digital bits in the ADC counter over

multiple input samples. The 1.2mm×1.6mm prototype fabricated in a 0.13µm standard

CMOS technology has been validated in prostate cancer DNA detection. Each channel

occupies an area of only 0.06mm2 and consumes 42µW of power from a 1.2V supply.

This work was presented in 2012 IEEE Biomedical Circuits and Systems Conference

(BioCAS) [158] and published in IEEE Transactions on Biomedical Circuits and Sys-

tems [144].

The main contribution of this chapter is the design and implementation of a low-

power mixed-signal VLSI approach to implement frequency response algorithm (FRA)

on-chip for DNA impedance extraction. Multiplication, which is the most area-consuming


and power-intensive operation required by the FRA algorithm is implemented by modi-

fying the timing of the in-channel dual-slope ADC resulting in a low-power, small-area

implementation of the FRA algorithm.

• Chapter 5

Chapter 5 presents the fully integrated 54-channel wireless, label-free DNA anal-

ysis SoC configured for temperature regulation. The on-chip, in-channel heating and

temperature sensing elements were implemented in standard CMOS without any post-

processing using feedback proportional-integral-derivative (PID) control. The tempera-

ture can be regulated to within 0.5C of the desired point, which enables precise control

of on-chip DNA hybridization during characterization or sensing. Two computationally

expensive operations, the multiplication and subtraction required by the PID algorithm,

are performed by the in-channel, dual-slope multiplying ADC in the mixed-signal do-

main, similar to which is described in Chapter 4, and the result is minimal area and

power consumption. The 3mm×3mm prototype fabricated in a 0.13µm standard CMOS

technology has been validated in a portion of the PCR protocol. The PID regulation loop

reuses circuits from the DNA-sensing channel resulting in minimal area. Each channel

occupies an area of only 0.06mm2 and consumes 42µW from a 1.2V supply. This work

was presented at 2012 Symposia on VLSI Technology and Circuits [87].

The main contribution of this chapter is the efficient mixed-signal VLSI implemen-

tation of the PID algorithm for temperature regulation. The multiplication required by

the PID algorithm which is the most area-consuming and power-hungary operation is

efficiently implemented by the in-channel dual-slope ADC. This results is small area

and power consumption and enables a large channel count.


6.2 Future Work

6.2.1 Power Harvesting and Wireless Communication Chip

An additional 6mm2 chip has been designed and fabricated in 0.13µm CMOS tech-

nology that consists of a wireless inductive power receiver, a command data receiver

and two high-speed transmitters. The chip was co-designed with Karim Abdelhalim, in

order to provide wireless power transfer and wireless data command transfer capabili-

ties to the DNA analysis SoC. The fabricated prototype die micrograph is depicted in

Fig. 6.1.

The power harvesting unit of the chip consists of a low-drop-out rectifier capable

of operating at up to 10MHz, two low-dropout regulators providing 2.5V and 1.2V

supply voltages with a maximum load current of 10mA and an on-chip programable

DAC generating up to 8 bias voltages. A 4MHz amplitude shift keying (ASK)/ on-off

keying (OOK) wireless data receiver was implemented to enable wireless transmission

of configuration and command data to the DNA analysis SoC. Finally, a 915MHz PLL-

based FSK transmitter and a UWB transmitter were also included in the CMOS chip.

This chip will be integrated with the presented SoC in the near future.

6.2.2 Improvements to the SoC

The presented SoC prototype contains a large number of input/output (I/O) and debug

pins and also requires some off-chip current and voltage bias sources. In the next gen-

eration of the SoC the bias voltages and current sources should be implemented on the

chip using on-chip band-gap voltage and current sources. Also, the I/O and debug pins

number needs to be reduced. One approach to reduce the debug pins number is to mux

all the analog test points into a single node and the same should be done for the digital

I/O pins resulting in only one or few analog and one digital I/O pins.


915MHz OOK TX

3.1-10.6GHz

UWB TX

10MHz

COMMAND

RX

4MHz

POWER

RX

Figure 6.1: Die micrograph of the 2mm×3mm power harvesting and wireless communicationchip. The chip was fabricated in a 0.13µm standard CMOS technology.

Analog improvements to the SoC design include implementation of a differential

front end current acquisition circuitry to reduce the second harmonic interference and

common-mode noise. Another improvement is reduction of flicker noise in the front-

end current acquisition circuits.

Testing improvements include design of a better and more durable packaging for

the CMOS chips. Currently a fill-and-dam technique is used to protect the chip bond-

wires from the electrolyte solution under test. This technique is only protecting the

chip for a limited time and after roughly one hour of testing a chip may become non-

functional. There is a high chance that the pins are shorted due to leakage. Another

improvement to the testing is integration and miniaturization of the Ag/AgCl electrode

on the CMOS chip. Currently a somewhat bulky off-chip Ag/AgCl is used for all the

testing. A number of miniaturized Ag/AgCl electrode fabrication methods have been

reported in literature [78, 170]. These reports can be used as a starting point for full

integration of the reference electrode on the CMOS chip.

173

References

[1] B. Albert, A. Johonson, J. Lewis, M. Raff, K. Robert and P. Walter, Molecular Biology of the cell,

5th ed. New York, NY: Garland Sceince, 2008.

[2] M. Schena, Mircorarray Analyis. hoboken, NJ: Wiley, 2003.

[3] B. A. Flusberg, D. R. Webster, J. H Lee, K. J. Travers, E. C. Olivares, T. A. Clark, J. Korlach,

and S. W. Turner, “Direct Detection of DNA Methylation During Single-Molecule, Real-Time

Sequencing,” Nature Methods, vol. 7, pp. 461−465, 2010.

[4] G. Liu, Y. Wan, V. Gau, J. Zhang, L. Wang, S. Song and C. Fan, “An Enzyme-Based E-DNA Sensor

for Sequence-Specific Detection of Femtomolar DNA Targets,” Journal of American Chemical

Society, vol. 130, no. 21, pp. 6820−6825, 2008.

[5] B. Malorny, C. Bunge, R. Helmuth, “A Real-Time PCR for the Detection of Salmonella Enteritidis

in Poultry Meat and Consumption Eggs,” Journal of Microbiological Methods, vol. 70, no. 2,

pp. 245−251, 2007.

[6] A. Nouri, and C. F. Chyba, “DNA Synthesis Security,” Methods in Molecular Biology, Vol. 852,

Part. 4, pp. 285−296, 2012.

[7] N. A. Straus, T. I. Bonner, “Temperature dependence of RNA-DNA hybridization kinetics,”

Biochimica et Biophysica Acta (BBA) - Nucleic Acids and Protein Synthesis., Vol. 277, pp. 87−95,

1972.

[8] A. Hassibi, H. Vikalo and A. Hajimiri, “On Noise Processes and Limits of Performance in Biosen-

sors,” Journal of Applied Physics., 102, 2007.

[9] N. Levine, Quantum Chemistry, 5th ed. Prentice-Hall, New York, 1999 .

[10] N. G. Van Kampen, Stochastic Processes in Physics and Chemistry North-Holland, Amsterdam,

1981 .

REFERENCES 174

[11] J. I. Steinield, J. S. Fransisco, and W. L. Hase, Chemical Kinetics and Dynamics, 2nd ed. Prentice-

Hall, New York, 1998 .

[12] J. Hubble, “Monte Carlo simulation of biospecific interactions,” Biotechnology Letters, Volume

22, Issue 18, pp. 1483−1486, 2000.

[13] K. Linneti and M. Kondratovich, “Partly nonparametric approach for determining the limit of

detection,” Clin Chem. Volume 50, Issue 4, pp. 732−740, 2004.

[14] P. M. Levine and G. W. Roberts, “High-resolution flash time-to-digital conversion and calibration

for system-on-chip testing,” Embedded Microelectronic Systems: Status and Trends, B. M. Al-

Hashimi, Ed. London: IEE Press, 2006.

[15] S. Cagnin, M. Caraballo, C. Guiducci, P. Martini, M. Ross, M. S. Ana, D. Danley, T. West, and

G. Lanfranchi, “Overview of Electrochemical DNA Biosensors: New Approaches to Detect the

Expression of Life,” Sensors, vol. 9, no. 4, pp. 3122−3148, 2009.

[16] O. A. Loaiza, S. Campuzano, M. Pedrero, M. I. Pividori, P. Garci, and J. M. Pingarro, “Disposable

Magnetic DNA Sensors for the Determination at the Attomolar Level of a Specific Enterobacteri-

aceae Family Gene,” Anal. Chem., vol. 80, no. 21, pp. 8239−8245, 2008.

[17] C. Stagni, C. Guiducci, L. Benini, B. Ricc, S. Carrara, B. Samor, C. Paulus, M. Schienle, M.

Augustyniak, and R. Thewes, “CMOS DNA sensor array with integrated A/D conversion based on

label-free capacitance measurement, IEEE J. Solid-State Circuits, vol. 41, no. 12, pp. 29562964,

Dec. 2006.

[18] X. C. Zaho, L. Q. Huang, S. F. Li, “Microgravimetric DNA Sensor Based on Quartz Crystal

Microbalance: Comparison of Oligonucleotide Immobilization Methods and the Application in

Genetic Diagnosis,” Biosens Bioelectron, vol. 16, no.2, pp. 85−95, 2001.

[19] A. Sassolas, B. D. Leca-Bouvier, and L. J. Blum, “DNA Biosensors and Microarrays,” Chem. Rev.,

Vol. 108, pp. 109−139, 2008.

[20] F. Heer, M. Keller, G. Yu, J. Janata, M. Josowicz, and A. Hierlemann, “CMOS Electro-Chemical

DNA-Detection Array with On-Chip ADC,” IEEE Int. Solid-State Circuits Conf. (ISSCC),

pp. 168−169, 2008.

REFERENCES 175

[21] M. Schienle, C. Paulus, A. Frey, F. Hofmann, B. Holzapfl, P. S. Bauer, and R. Thewes, “A Fully

Electronic DNA Sensor with 128 Positions and In-Pixel A/D Conversion,” IEEE J. Solid-State

Circuits, vol. 39, no. 12, pp. 2438−2445, 2004.

[22] A. Hassibi, T. H. Lee, “A Programmable Electrochemical Biosensor Array in 0.18µm Standard

CMOS,” IEEE Int. Solid-State Circuits Conf. (ISSCC), pp. 168−169, 2005.

[23] A. Hassibi, H. Vikalo, J. L. Riechmann, and B. Hassibi, “Real-time DNA microarray analysis,”

Nucl. Acids Res., pp. 1−12, Aug. 2009.

[24] B. Jang, P. Cao, A. Chevalier, A. Ellington, and A. Hassibi, “A CMOS fluorescence-based biosen-

sor microarray,” IEEE Int. Solid-State Circuits Conf. (ISSCC), pp. 436−437, Feb. 2009.

[25] H. Wang, Y. Chen, A. Hassibi, A. Scherer, and A. Hajimiri, “A frequency-shift CMOS magnetic

biosensor array with single-bead sensitivity and no external magnet,” IEEE Int. Solid-State Circuits

Conf. (ISSCC), pp. 438−439, Feb. 2009.

[26] P. M. Levine, P. Gong, R. Levicky, K.L. Shepard, “Active CMOS Sensor Array for Electrochemical

Biomolecular Detection,” IEEE J. Solid-State Circuits, vol. 43, no. 8, pp. 1859−1871, 2008.

[27] T. G. Drummond, M. G. Hill, J. K. Barton, “Electrochemical DNA sensors,” Nature Biotechnology,

vol. 21, no. 10, pp. 1192−1199, Oct. 2003.

[28] C. Yang, Y. Huang, B.L. Hassler, R.M. Worden, and A.J. Mason, “Amperometric electrochemical

microsystem for a miniaturized protein biosensor array,” IEEE Transaction on Biomedical Circuits

Systems, vol. 3, no. 3, pp. 160−168, 2009.

[29] M. schena, R. A. Heller, T. P. Theriault, K, Konrad, E. Lachenmeier, and R. W Davis, “Microar-

rays: biotechnology dicovery platform for functional genimics,” Trends in Biotechnology, vol. 16,

no. 7, pp.301–306, july 1998.

[30] D. Baran, M. Aktan and V. G. Oklobdzija, “Multiplier Structures for Low Power Applications in

Deep-CMOS, IEEE Int. Symposium on Circ. and Systems (ISCAS), pp. 3562-3565, May 2010.

[31] http://trojanuv.com/

[32] P. F. Macgregor and J. A squire, “Application of microarrays to the analysis of gene expression in

cancer,” Clinical Chemistary, vol. 48, no.8, pp.1170-1177, 2002.

[33] M. J. Cunningham, “Genomics and proteomics: The new millennium of drug discovery and devel-

opment,” Journal of Pharmacological and Toxiocological Methos, vol. 44, pp. 291-300, 2000.

REFERENCES 176

[34] M. Schena, “Microarray Aalysis,” Hoboken, NJ: Wiley, 2003.

[35] http://www.affymetrix.com

[36] http://www.agilent.com

[37] http://www.spartanbio.com/

[38] http://www.earlywarninginc.com/

[39] http://www.thasar.com/

[40] http://www.metrohm.com/

[41] http://www.iontorrent.com/

[42] www.geneonyx.com/

[43] http://www.idetekt.com/

[44] http://dnae.co.uk/

[45] C.R. Tamanaha et al, “Magnetic Method for DNA Detection on an Arrayed Solid State Device,

Micro Total Analysis Systems, Kluwer Academic, pp. 444-446, 2001.

[46] G. Li et al., “Detection of Single Micron-Sized Magnetic Bead and Magnetic Nanoparticles Using

Spin Valve Sensors for Biological Applications, J. Appl. Phys., vol. 93, no. 10, pp. 7557-7559,

May 2003.

[47] S. Han et al., “A High-Density Magnetoresistive Biosensor Array with Drift-Compensation Mech-

anism, ISSCC Dig. Tech. Papers, pp. 168-169, Feb. 2007.

[48] Y. Liu et al., “CMOS Mini Nuclear Magnetic Resonance System and its Application for Biomolec-

ular Sensing, ISSCC Dig. Tech. Papers, pp. 140-141, Feb. 2008.

[49] P. Besse et al., “Detection of a Single Magnetic Microbead Using a Miniaturized Silicon Hall

Sensor, Appl. Phys.Letters, vol 80, no. 22, pp. 4199-4201, June 2002.

[50] R. Singh, D. Ho, A. Nilchi, R. Genov, G. Gulak, “A Hybrid Thin-Film/CMOS Fluorescence Con-

tact Imager, IEEE Int. Symp. on Circuits and Systems (ISCAS’2009), pp. 2437-2440, May 2009.

[51] A. Romani, R. Guerrieri et. al., “Capacitive sensor localization of bioparticles in CMOS lab-on-a-

chip, ISSCC Proceedings of the International Solid-State Circuits Conference, vol. 1, 2004.

[52] M. Coln, “Chopper stabilization of MOS operational amplifers using feed forward techniques,

IEEE J. Solid-State Circuits, vol. SC-16, no. 6, pp. 745748, Dec. 1981.

REFERENCES 177

[53] L. Toth and Y. P. Tsividis, “Generalization of the principle of chopper stabilization, IEEE Trans.

Circuits Syst. I, Fundam. Theory Appl., vol. 50, no. 8, pp. 975983, Aug. 2003.

[54] C. C. Enz and G. C. Temes, “Circuit techniques for reducing the effects of op-amp imperfec-

tions: autozeroing, correlated double sampling, and chopper stabilization, Proc. IEEE, vol. 84, pp.

5841614, Nov. 1996.

[55] J. Correia, G. Graaf, M. Bartek, and R. Wolffenbuttel, “A single-chip CMOS optical microspec-

trometer with light-tofrequency converter and bus interface, IEEE J. Solid-State Circuits, vol. 37,

no. 10, pp. 13441347, Oct 2002.

[56] X. Fang, V. Hsiao, V. Chodavarapu, A. Titus, and A. Cartwright, “Colorimetric porous photonic

bandgap sensors with integrated CMOS color detectors, IEEE Sensors Journal, vol. 6, no. 3, pp.

661667, Jun 2006.

[57] H. Ishii, Y. Maruyama, H. Takao, M. Ishida, and K. Sawada, “Improvement in filter-less fuores-

cence sensor capability by optimization of potential distribution, 4th Asia-Pacific Conf. on Trans-

ducers and Micro-Nano Tech., pp. 6871, Jun 2008.

[58] R. Singh, L. Leng, A. Guenther, R. Genov, “A Hybrid CMOS-Microfluidic Contact Imaging Mi-

crosystem, SPIE Optics and Photonics Aug. 1−6, 2009.

[59] D. Ho, G. Gulak, R. Genov, “CMOS Electric Field-Modulated Color Sensor, IEEE Custom Inte-

grated Circuits Conference (CICC2011), Sept. 2011.

[60] D. Ho, M. O. Noor, U. J. Krull, G. Gulak, R. Genov, “Single-Filter Multi-Color CMOS Fluorescent

Contact Sensing Microsystem, IEEE Int. Symp. on Circuits and Systems (ISCAS’2012), 2012.

[61] D. Ho, G. Gulak, R. Genov, “CMOS 3-T Digital Pixel Sensor with In-Pixel Shared Comparator,

IEEE Int. Symp. on Circuits and Systems (ISCAS’2012), 2012.

[62] Saiki, R.; Scharf, S.; Faloona, F.; Mullis, K.; Horn, G.; Erlich, H.; Arnheim, “Enzymatic ampli-

fication of beta-globin genomic sequences and restriction site analysis for diagnosis of sickle cell

anemia”. Science 230 (4732): 13501354.

[63] Cheng, S.; Fockler, C.; Barnes, W. M.; Higuchi, “Effective Amplification of Long Targets from

Cloned Inserts and Human Genomic DNA”. Proceedings of the National Academy of Sciences 91

(12): 56955699.

REFERENCES 178

[64] Pavlov, A. R.; Pavlova, N. V.; Kozyavkin, S. A.; Slesarev, “Recent developments in the optimiza-

tion of thermostable DNA polymerases for efficient applications”. Trends in Biotechnology 22 (5):

253260.

[65] Herman JG, Graff JR, Myhnen S, Nelkin BD, Baylin SB (1996). ”Methylation-specific PCR:

a novel PCR assay for methylation status of CpG islands”, Proc Natl Acad Sci USA 93 (13):

98219826

[66] J. Wang, “Electrochemical biosensors: Towards point-of-care cancer diagnostics,” Journal of

Biosensors and Bioelectronics, vol 21, pp. 18871892, 2006.

[67] R. F. B. Turner, D. J. Harrison, and H. P. Baltes, “A CMOS potentiostat for amperometric chemical

sensors,” IEEE Journal of Solid-State Circuits, vol. 22, 1987.

[68] L. Li, X. Liu, W. A. Qureshi, and A. J. Mason, “CMOS Amperometric Instrumentation and Pack-

aging for Biosensor Array Applications,” IEEE Transactions on Biomedical Circuits and Systems,

vol. 5, no. 5, pp. 439−448, 2011.

[69] R. Genov, M. Stanacevic, M. Naware, G. Cauwenberghs, and N. Thakor, ”16-channel Integrated

Potentiostat for Distributed Neurochemical Sensing,” IEEE Transactions on Circuits and Systems

I, vol. 53, no. 11, pp. 2371−2376, Nov. 2006.

[70] M. Nazari, H. Jafari, L. Leng, A. Guenther, R. Genov, “CMOS Neurotransmitter Microarray:

96-Channel Integrated Potentiostat with On-die Microsensors,” to appear, IEEE Transactions on

Biomedical Circuits and Systems, 2012.

[71] M. Roham, M.P. Garris, and P. Mohseni, “A Wireless IC for Time-Share Chemical and Electrical

Neural Recording,” IEEE J. Solid-State Circuits, vol. 44, no. 10, pp. 3645−3658, 2009.

[72] K. Abdelhalim, R. Genov, 915-MHz Wireless 64-Channel Neural Recording SoC with Pro-

grammable Mixed-Signal FIR Filters, IEEE European Solid-State Circuits Conference (ESS-

CIRC2011), Sept. 2011.

[73] M. Jamal Deen, Munir M. El-Desouki, Hamed M. Jafari and Saman Asgaran, Low-Power In-

tegrated CMOS RF Transceiver Circuits for Short-Range Applications, 50th IEEE International

Midwest Symposium on Circuits and Systems (MWSCAS 2007) and 5th IEEE International North-

east Workshop on Circuits and Systems (NEWCAS 2007), Montreal, Canada, (5-8 August 2007).

REFERENCES 179

[74] R. R. Harrison, P. T. Watkins, R. J. Kier, R. O. Lovejoy, D. J. Black, B. Greger, and F. Solzbacher,

A low-power integrated circuit for a wireless 100-electrode neural recording system, IEEE Journal

of Solid- State Circuits, vol. 42, no. 1, pp. 123133, January 2007.

[75] S. Lee, H. Lee, M. Kiani, U. Jow, and M. Ghovanloo, An inductively powered scalable 32-

channel wireless neural recording system-on-a-chip for neuroscience applications, IEEE Interna-

tional Solid-State Circuits Conference, pp. 120121, Feb. 2010.

[76] M. S. Chae, W. Liu, Z. Yang, T. Chen, J. Kim, M. Sivaprakasam, and M. Yuce, A 128-channel

6mW wireless neural recording IC with on-the-fly spike sorting and UWB transmitter, IEEE Inter-

national Solid-State Circuits Conference Digest of Technical Papers, pp. 146603, February 2008.

[77] S. Cho and A. P. Chandrakasan, A 6.5-Ghz energy-efficient BFSK modulator for wireless sensor

applications, IEEE Journal of Solid-State Circuits, vol. 39, no. 5, pp. 731739, May 2004.

[78] M.W. Shinwari, D. Zhitomirsky, I.A. Deen, P.R. Selvaganapathy, M.J. Deen, and D. Landheer,

“Microfabricated Reference Electrodes and their Biosensing Applications,” Sensors Special Issue

on ISFET Sensors, Vol. 10(3), pp. 16791715.

[79] Roberge, P.R. Handbook of Corrosion Engineering, McGraw-Hill: New York, NY, USA, 2000.

[80] Bockris, J.O’M.; Devanathan M.A.V.; Reddy, A.K.N., “The anodic formation of calomel films

on mercury electrodes-an ellipsometric-galvanostatic study,” Proc. R. Soc. Lond., Vol. 279, pp.

327345, 1964.

[81] Kita, H. Periodic variation of exchange current density of hydrogen electrode reaction with atomic

number and reaction mechanism. Trans. Symp. Elec. Proc.1966, 113, 7990.

[82] Appelby, A.J.; Chemla, M.; Kita, H.; Bronoel, G. Encyclopedia of the Electrochemistry of Ele-

ments; Marcel Dekker: New York, NY, USA, 1982; Volume IX, p. 383.

[83] N. Trombly, A. Mason, Post-CMOS Thermally Controlled Electrochemical Electrodes, IET Elec-

tronics Letters, vol. 44, no. 1, p. 2930, Jan. 2008.

[84] C. Yang, S. R. Jadhav, R. M. Worden, and A. J. Mason, “Compact low-power impedance-to-

digital converter for sensor array microsystems,” IEEE J. Solid-State Circuits, vol. 44, no. 10, pp.

2844−2855, 2009.

[85] J. Kafka, O. Pnke, B. Abendroth, and F. Lisdat, “A label-free DNA sensor based on impedance

spectroscopy,” Electrochimica Acta, vol. 53, Issue 25, pp. 7467−7474, October 2008.

REFERENCES 180

[86] Y. T. Long, C. Z. Li, H. B. Kraatz, and J. S. Lee, “AC Impedance Spectroscopy of Native DNA

and M-DNA,” Biophys J, vol.84, Issue 5, pp. 3218−3225, May 2003.

[87] H. M. Jafari, L. Soleymani, K. Abdelhalim, E. H. Sargent, S. O. Kelley and R. Genov, “Nanos-

tructured CMOS Wireless Ultra-Wideband Label-free DNA Analysis SoC,” IEEE Symposium on

VLSI Circuits, Dig. of Tech. Papers, pp. 6970, June 2012.

[88] H. Jafari, R. Genov, “Chopper-Stabilized Bidirectional Current Acquisition Circuits for Electro-

chemical Amperometric Biosensors,” accepted for publication, IEEE Transactions on Circuits and

Systems I, 2012.

[89] P. Kim, N. Kohli, A. Mason, R. M. Worden, and R. Ofoli, An Electrochemical Interface for Inte-

grated Biosensors, IEEE Int. Conf. on Sensors, Toronto, Canada, October 2003.

[90] A. Mason, Y. Huang, C. Yang, J. Zhang, Amperometric Readout and Electrode Array Chip for

Bioelectrochemical Sensors, IEEE Int. Symposium on Circ. and Systems (ISCAS), pp. 35623565,

May 2007.

[91] M. Ahmadi and G. Jullien, “Current-Mirror-Based Potentiostats for Three-Electrode Amperomet-

ric Electrochemical Sensors,” IEEE Transactions on Circuits and Systems I, vol. 56, pp. 13391348,

2009.

[92] H. S. Narula, J. G. Harris, “A Time-Based VLSI Potentiostat for Ion Current Measurement,” IEEE

Sensors Journal, vol. 6, pp. 239247, 2006.

[93] S. Hwang and S. Sonkusale, “CMOS VLSI Potentiostat for Portable Environmental Sensing Ap-

plications,” IEEE Sensors Journal, vol. 10, pp. 820−821, 2010.

[94] S. Ayers, K. D. Gillis, M. Lindau and B. A. Minch, “Design of a CMOS Potentiostat Circuit for

Electrochemical Detector Arrays,” IEEE Transactions on Circuits and Systems I, vol. 54, no. 4,

pp. 736−744, April 2007.

[95] A. Zeki and H. Kuntman, “Accurate and High Output Impedance Current Mirror Suitable for

CMOS Current Output Stages,” IEEE Sensors Journal, vol. 6, pp. 1380−1388, 2006.

[96] M. Augustyniak, C. Paulus, R. Brederlow, N. Persike, G. Hartwich, D. Schmitt-Landsiedel, R.

Thewes, “A 24×16 CMOS-Based Chronocoulometric DNA Microarray,”IEEE Int. Solid-State Cir-

cuits Conf. (ISSCC), pp. 168−169, 2006.

REFERENCES 181

[97] D. M. Garner, H. Bai, P. Georgiou, T. G. Constandinou, S. Reed, L. M. Shepherd, W. Wong, K. T.

Lim, C. Toumazou , “A Multichannel DNA SoC for Rapid Point-of-Care Gene Detection,” IEEE

J. Solid-State Circuits, vol. 43, no. 8, pp. 1859−1871, 2010.

[98] C. W. Huang, Y. J. Huang, P. W. Yen, H. T. Hsueh, C. Y. Lin, M. C. Chen, C. Ho, F. L. Yang, H. H.

Tsai, H. H. Liao, Y. Z. Juang, C. K. Wang, C. T. Lin, and S. S. Lu, “A Fully Integrated Hepatitis

B Virus DNA Detection SoC based on Monolithic Polysilicon Nanowire CMOS Process,” IEEE

Symposium on VLSI Circuits, Dig. of Tech. Papers, pp. 124125, June 2012.

[99] A. Hassibi, T. H. Lee, “A Programmable Electrochemical Biosensor Array in 0.18µm Standard

CMOS,” IEEE Int. Solid-State Circuits Conf. (ISSCC), pp. 168−169, 2005.

[100] B. Razavi, “Design of analog CMOS integrated circuits,” Boston, MA: McGraw-Hill, 2001.

[101] H.-S. Wong and M. White, “A self-contained CMOS integrated pH sensor,” Proc. IEEE Int. Elec-

ton Devices Meeting, pp.658661 1988

[102] Allen J. Bard, Larry R. Faulkner, ” Electrochemical Methods: Fundamentals and Applications”,

2nd Edition ISBN: 978471-04372-0

[103] G. Han and E. S. anchez-Sinencio, “CMOS Transconductance Multipliers: A Tutorial,” IEEE

TRANSACTIONS ON CIRCUITS AND SYSTEMSII: ANALOG AND DIGITAL SIGNAL PROCESS-

ING, VOL. 45, NO. 12, DECEMBER 1998.

[104] K. Bult and H. Wallinga, “A four-quadrant analog multiplier, IEEE J. Solid-State Circuits, vol.

SC-21, no. 3, pp. 430435, Jun. 1986.

[105] O. Changyue, C. Peng, and X. Yizhong, “Study of switched capacitor multiplier, in Proc. Int.

Conf. Circuits Syst, Jun. 1991, pp. 234237.

[106] Z.Wang, “A four-transistor four-quadrant analog multiplier using MOS transistors operating in the

saturation region, IEEE Trans. Instrum. Meas., vol. 42, no. 1, pp. 7577, Feb. 1993.

[107] H. R. Mehrvarz and C. Y. Kwok, “A large-input-dynamic-range multiinput floating gate MOS

four-quadrant analog multiplier, in Proc. IEEE Int. Solid-State Circuits Conf., Feb. 1995, pp. 6061.

[108] S. Liu and C. Chang, “CMOS subthreshold four quadrant multiplier based on unbalanced source

coupled pairs, Int. J. Electron. , vol. 78, pp. 327332, Feb. 1995.

[109] B. Maundy and P. Aronhime, “Useful multipliers for low-voltage applications, in Proc. IEEE Int.

Symp. Circuits Syst., vol. 1, May 2002, pp. 2629.

REFERENCES 182

[110] U. Gatti, F. Maloberti, and G. Torelli, “CMOS triode-transistor transconductor for high-frequency

continuous-time filters, Proc. IEE Circuits, Devices, Syst., vol. 141, no. 6, pp. 462468, Dec. 1994.

[111] S. Liu and Y. Hwang, “CMOC four-quadrant multiplier using bias feedback techniques, IEEE J.

Solid-State Circuits, vol. 29, pp. 750752, Jun. 1994.

[112] S. I. Liu, “Low voltage CMOS four-quadrant multiplier, Electron. Lett., vol. 30, pp. 21252126,

Dec. 1994.

[113] C. Chen, Z. Li , “A Low-Power CMOS Analog Multiplier , IEEE TRANSACTIONS ON CIRCUITS

AND SYSTEMSII: EXPRESS BRIEFS, VOL. 53, NO. 2, FEBRUARY 2006.

[114] A. Naderi., A. Khoei,K. Hadidi,H. Ghasemzadeh, “ A new high speed and low power four-

quadrant CMOS analog multiplier in current mode , Int. J.Electron.Commun.(A)63(2009)769775.

[115] Po-lun. Chang, Fei-hu. Shieh, Horng. lin, “A novel digit serial dual basis GF(2m) multiplier,IEEE

Industrial Engineering and Engineering Management (IEEM), pp. 329333, Feb. 2010.

[116] A. D. Booth, “A Signed Binary Multiplication Technique, Quarterly J. Mechanical Applications

in Math., vol. 4, part 2, pp. 236240, 1951.

[117] Q. L. Macsorley, “High Speed Arithmetic in Binary Computers, IRE Proc., vol. 49, pp. 6791, Jan.

1961.

[118] L. Dadda, “Some schemes for Parallel Multipliers, Alta Frequenza, vol. 34, pp. 349356, Mar.

1965.

[119] Y. Seo, and D. Kim, “ A New VLSI Architecture of Parallel MultiplierAccumulator Based on

Radix-2 Modified Booth Algorithm, IEEE TRANSACTIONS ON VERY LARGE SCALE INTE-

GRATION (VLSI) SYSTEMS, VOL. 18, NO. 2, FEBRUARY 2010

[120] G. Goto, T. Sato, M. Nakajima, and T. Sukemura, A 54 by 54 regular structured tree multi-

plier,IEEE J. Solid-State Circuits, vol. 27, no. 9, pp. 12291236, Sep. 1992.

[121] J. Fadavi-Ardekani, M..N Booth encoded multiplier generator using optimizedWallace trees,IEEE

Trans. Very Large Scale Integr. (VLSI) Syst., vol. 1, no. 2, pp. 120125, Jun. 1993.

[122] N. Ohkubo, M. Suzuki, T. Shinbo, T. Yamanaka, A. Shimizu, K. Sasaki, and Y. Nakagome, A 4.4

ns CMOS 54..54 multiplier using pass-transistor multiplexer, IEEE J. Solid-State Circuits, vol. 30,

no. 3, pp. 251257, Mar. 1995.

REFERENCES 183

[123] A. Tawfik, F. Elguibaly, and P. Agathoklis, New realization and implementation of fixed-point IIR

digital filters, J. Circuits, Syst., Comput., vol. 7, no. 3, pp. 191209, 1997.

[124] A. Tawfik, F. Elguibaly, M. N. Fahmi, E. Abdel-Raheem, and P. Agathoklis, High-speed area-

efficient inner-product processor, Can. J. Electr. Comput. Eng., vol. 19, pp. 187191, 1994.

[125] B. Parhami, Computer arithmetic algorithms and hardware designs, Oxford Univ. Press, 2000.

[126] C. S. Wallace, “A Suggestion for a Fast Multiplier, IEEE Trans. Computers, vol. 13, no. 2, pp.

1417, Feb. 1964.

[127] L. Dadda, “ Some schemes for Parallel Multipliers, Alta Frequenza, vol. 34, pp. 349356, Mar.

1965.

[128] Baran, D; Aktan, M; Oklobdzija, V.G;, “Energy-Efficient Implementation of Parallel CMOS Mul-

tipliers with Improved Compressors, 16th ACM/IEEE International Symposium on Lowpower

Electronics and Design , pp. 147152 , August 2010.

[129] Zeydel, B.R.; Baran, D.; Oklobdzija, V.G.; , ”Energy-Efficient Design Methodologies: High-

Performance VLSI Adders,” Solid-State Circuits, IEEE Journal of , vol.45, no.6, pp.12201233,

June 2010.

[130] M. Min and T. Parve, “Improvement of lock-in bio-impedance analyzer for implantable medical

devices,” IEEE Trans. Instrum. Meas., vol. 56, pp. 968−974, 2007.

[131] A. Manickam, A. Chevalier, M. McDermortt, A. D. Ellington, and A. Hassibi, “A CMOS Elec-

trochemical Impedance Spectroscopy Biosensor Array for Label-Free Biomolecular Detection,”

IEEE Int. Solid-State Circuits Conf. (ISSCC), pp. 130−131, 2010.

[132] J. B. Christen, A. G. Andreou, “Design, Fabrication and Testing of a Hybrid CMOS/PDMS Mi-

crosystem for Cell Culture and Incubation,”IEEE Transactions on Biomedical Circuits and Sys-

tems, vol. 1, no. 1, pp. 2−14, 2007.

[133] D. Kim, B. Goldstein, W. Tang, F. Sigworth, E. Culurciello, “Noise Analysis and Performance

Comparison of Low Current Measurement Systems for Biomedical Applications” IEEE Transac-

tions on Biomedical Circuits Systems, vol. 99, no. 1, 2012.

[134] J. Xu, F. Q. Fan, J. H. Huijsing, C. V. Hoof, R. F. Yazicioglu, K. A. A. Makinwa, “Measure-

ment and Analysis of Input Current Noise in Chopper Amplifiers,” European Solid-State Circuits

Conference (ESSCIRC), pp. 8184 , 2012.

REFERENCES 184

[135] L. Xiaowen, Li. Lin, A.J. Mason, “Thermal control microsystem for protein characterization and

sensing,”IEEE BioCAS Conf., pp. 277−280,2009.

[136] U. Frey, M. Graf, S. Taschini, K. Kirstein, A. Hierlemann, “A Digital CMOS Architecture for a

Micro-Hotplate Array,”IEEE J. Solid-State Circuits, vol. 42, no. 2, pp. 441−450, 2007.

[137] H. M. Jafari, L. Soleymani, K. Abdelhalim, E. H. Sargent, S. O. Kelley and R. Genov, “Nanos-

tructured CMOS Wireless Ultra-Wideband Label-free DNA Analysis SoC,” IEEE J. Solid-State

Circuits, submited 2012.

[138] L. Soleymani, Z. Fang, E. H. Sargent and S. O. Kelley, “Programming Nucleic Acids De-

tection Sensitivity Using Controlled Nanostructuring,” Nature Nanotechnology, vol. 4, no. 12,

pp. 844−848, 2009.

[139] J. Zhang, H. P. Lang, G. Yoshikawa, and C. Gerber, “Optimization of DNA Hybridization Effi-

ciency by pH-Driven Nanomechanical Bending,” Langmuir, 2012, 28 (15), pp 64946501.

[140] M. Alubaidy, L. Soleymani, K. Venkatakrishnan, and B. Tan “Femtosecond Laser Nanostructuring

for Femtosensitive DNA detection,” Biosensors and Bioelectronics, vol. 33, no. 1, pp. 83−87,

March. 2012.

[141] L. Aorga, A. Rebollo, J. Herrn, S. Arana, E. Bandrs, and J. Garca-Foncillas, “Development of a

DNA Microelectrochemical Biosensor for CEACAM5 Detection,” IEEE Sensor Journal, vol. 10,

no. 8, pp. 1368−1374, Aug 2010.

[142] Y. S. Choi, K. S. Lee, and D. H. Park, “A Genomic Detection Using an Indicator-Free DNA on a

DNA Chip Microarray,” Applied Physics, vol. 6, pp. 772−776, 2006

[143] M. J. Milgrew, “A 16×16 CMOS Proton Camera Array for Direct Extracellular Imaging of

Hydrogen-Ion Activity,” IEEE Int. Solid-State Circuits Conf. (ISSCC), pp. 590−638, 2008.

[144] H. M. Jafari, L. Soleymani, R. Genov, “16-Channle CMOS Impedance Spectroscopy DNA Ana-

lyzer with Dual-Slope Multiplying ADC,” IEEE Transactions on Biomedical Circuits and Systems,

to appear, 2012.

[145] A. J. Chen, “Multibit Delta-Sigma Modulator With Noise-Shaping Dynamic Element Matching,”

IEEE Transactions on Circuits and Systems I, vol. 56, no. 6, pp. 1125−1133, June 2009.

[146] H. Kim and D. P. and Y. Joo, ”All-digital low-power CMOS pulse generator for UWB sys-

tem,”Electron. Lett., vol. 40, pp. 15341535, Nov. 2004.

REFERENCES 185

[147] H. Kim, Y. Joo, and S. Jung, ”Tunable CMOS UWB pulse generator,”in Proc. IEEE Int. Conf.

Ultra-Wideband, Waltham, MA, Sep. 2427, 2006, pp. 109112.

[148] D. D. Wentzloff and A. P. Chandrakasan, ”Gaussian pulse generators for subbanded ultra-

wideband transmitters, ” IEEE Trans. Microwave Theory Tech., vol. 54, pp. 16471655, Jun. 2006.

[149] Wentzloff, D. D., R. Blazquez, F. S. Lee, B. P. Ginsburg, J. Powell and A. P. Chandrakasan, ”Sys-

tem Design Considerations for Ultra-Wideband Communication,” IEEE Communications Maga-

zine, vol. 43, no. 8, pp. 114-121, August 2005.

[150] P. P. Mercier, D. C. Daly, and A. P. Chandrakasan, “An Energy- Efficient All-Digital UWB Trans-

mitter Employing Dual Capacitively-Coupled Pulse-Shaping Drivers,” IEEE J. Solid-State Cir-

cuits, vol. 44, no. 6, pp. 1679−1688, 2009.

[151] Y. Park and D. Wentzloff, “An All-Digital 12 pJ/Pulse IR-UWB Transmitter Synthesized from a

Standard Cell Library,” IEEE J. Solid-State Circuits, vol. 46, no. 5, pp. 1147−1157, 2011.

[152] P. P. Mercier, M. Bhardwaj, D. C. Daly, and A. P. Chandrakasan, “A Low-Voltage Energy-

Sampling IR-UWB Digital Baseband Employing Quadratic Correlation,” IEEE J. Solid-State Cir-

cuits, Vol. 45, no. 6, 2010.

[153] http://www.basinc.com/products/ec/ref.php

[154] http://www.idt.com/

[155] H. M. Jafari, R. Genov, “Chopper-Stabilized Bidirectional Current Acquisition for Electrochemi-

cal Amperometric Biosensors,” IEEE Transactions on Circuits and Systems I, 2012.

[156] E. Barsoukov, J. R. Macdonald, Impedance Spectroscopy: Theory, Experiment, and Applications,

2nd ed., John Wiley, Hoboken, NJ, 2005.

[157] M. Nazari, H. M. Jafari, R. Genov,“192-Channel CMOS neurochemical microarray,”IEEE Custom

Integrated Circuits Conference (CICC), pp. 121−125, Sept. 2010.

[158] H. M. Jafari, and R. Genov, “CMOS Impedance Spectrum Analyzer with Dual-Slope Multiplying

ADC,” IEEE Biomedical Circuits and Systems Conference, San Diego, Nov. 2011.

[159] M. K. Law, A. Bermak, H. C. Luon, “A Sub-µW Embedded CMOS Temperature Sensor for RFID

Food Monitoring Application,” IEEE J. Solid-State Circuits, vol. 45, no.6, pp. 1246−1255, 2010.

REFERENCES 186

[160] K. Souri, Y. Chae, K. Makinwa, “A CMOS temperature sensor with a voltage-calibrated in-

accuracy of 0.15C (3sigma) from -55 to 125C,” IEEE Int. Solid-State Circuits Conf. (ISSCC),

pp. 208−210, 2012.

[161] C. Chen, C. Tsai, W. Lu, “A time-to-digital-converter-based CMOS smart temperature sensor,”

IEEE J. Solid-State Circuits, vol. 40, no. 8, pp. 1642−1648 , 2005.

[162] J. Wang, N. Zhang, “Proportional-Integral-Differential Controller for Polymerase Chain

Reaction,” International Conference on Bioinformatics and Biomedical Engineering,

pp. 1584−1587,2008.

[163] Z. Q. Niu, W. Y. Chen, S. Y. Shao, X. Y. Jia, W. P. Zhang, “DNA amplification on a PDMS-glass

hybrid microchip,” J. Micromech. Microeng vol. 16, pp. 425−433 2006.

[164] K. H. Ang, G. C. Y. Chong, Y. Li, “PID Control System Analysis, Design, and Technology,” IEEE

Trans Control Systems Tech, vol. 13, no. 5, pp. 559−576, 2005.

[165] M. A. P. Pertijs, K. Makinwa and J. Huijsing, “A CMOS Smart Temperature Sensor With a 3σ

Inaccuracy of ±0.1 oC From -55 oC to 125 oC,” IEEE Int. Solid-State Circuits Conf. (ISSCC),

pp. 436−437, Feb. 2005.

[166] H. Jafari, R. Genov, “Bidirectional Current Conveyer with Chopper Stabilization and Dynamic

Element Matching,” IEEE Int. Symp. on Circuits and Systems (ISCAS’2012), 2012.

[167] H. M. Jafari, and R. Genov, “CMOS Impedance Spectrum Analyzer with Dual-Slope Multiplying

ADC,” IEEE Biomedical Circuits and Systems Conference, San Diego, Nov. 2011.

[168] A. Prodic, A. Zirger, D. Maksimovic, “High-frequency digital PWM controller IC for DC-DC

converters,” IEEE Transactions on Power Electronics, vol. 18, no. 1, pp. 438− 446, 2003.

[169] http://www.temptronic.com/

[170] N. Trombly, A. Mason, “Post-CMOS Thermally Controlled Electrochemical Electrodes,” IET

Electronic Letters, vol. 44, no. 1, p. 29-30, jan. 2008.

187

Appendix A

Supplementary Hardware and

Software Documentation

A.1 Board Design

In order to demonstrate the DNA analysis SoC in applications a 6-layer PCB was de-

signed and fabricated as shown in Fig. A.1. The PCB provides the required digital

control sequence and DC bias voltages and currents to the SoC. The test board is com-

prised of the following components:

• FPGA FPGA: Alterra CYCLONE III FPGA 10K (256-FBGA544-2417-ND).

• FPGA EPROM: Memory to store FPGA pragrams (93LC46C-I/MS-ND)

• USB Controller: Communication link between the test board and PC (USB 768-

1024-1-ND)

• 12-bit DAC: Generating bias voltages for the DNA SoC (AD5328BRUZ-ND)

• Voltage regulator: Generating supply voltage for DNA SoC and the test board

(497-8717-1-ND)

CHAPTER A. SUPPLEMENTARY HARDWARE AND SOFTWARE DOCUMENTATION 188

• Crystal: Generating reference clock (XC1426CT-ND)

1

2

3

4

5

6

7

8

9

10

11121314

15

16

17

18

1920

21

22

23

2425 26

27

28

29

30

31

Figure A.1: PCB fabricated for characterizing the DNA analysis SoC.

The parts that are labeled on the board are as follows:

1. Channel digital supply voltage (1.2V)

2. Channel analog supply voltage (1.2V)

3. Channel common mode supply voltage (0.65V)

4. On-chip waveform generator supply voltage (1.5V)

5. On-chip SRAM supply voltage (1.2V)


6. On-chip PID supply voltage (1.2V)

7. On-chip heater supply voltage (1.5V)

8. UWB transmitter supply voltage (1.2V)

9. 5V supply voltage connector

10. On-chip DC bias voltage and current generator (from right to left: 1- channel

opamp voltage bias (300mV), 2-channel opamp voltage bias (700mV), 3- com-

parator voltage bias (500mV), 4-waveform generator voltage bias (320mV), 5-

waveform generator voltage bias (650mV), 6-channel reference current generator

(adjustable from 10pA to 100nA), 6-temperature sensor voltage bias (320mV)).

11. FPGA supply voltage (1.2V)



14. FPGA debug pins

15. USB chip supply voltage (3.3V)

16. USB chip supply voltage (1.8V)

17. USB connector

18. JTAG connector

19. Off-chip waveform generator 3.3V supply voltage

20. Off-chip counter electrode

21. Off-chip reference electrode


22. Off-chip waveform generator

23. FPGA debug pins

24. Off-chip waveform generator output

25. 3-electrode regulation loop opamps

26. UWB transmitter output

27. CMOS DNA analysis chip

28. On-chip waveform generator output

29. FPGA chip

30. USB controller chip

A.2 DNA Sensing protocol

The following protocol is utilized for preparation of 5µM DNA samples.

1. scan the electrode in 20mM ferro from 0 to 500mV at 100mV/s

2. Prepare the probes purchased from IDT DNA

• Product: 100 nmole DNA oligo

• Purification: Standard Desalting

• Sequence: /5ThioMC6-D/AG CGC GGC AGG AAG CCT TAT

Reduction of thiol modified oligos:

We recommend the use of immobilized TCEP from Pierce for the reduction of thiol

modified oligos. The immobilized TCEP has a number of advantages over alternative


reducing agents such as the ability to reduce a thiol over a large pH and temperature

range and in a variety of buffers. The protocol below is one we have used and may need

to be optimized for specific conditions/requirements.

1. Resuspend the oligo in PBS pH 7.0, or other appropriate buffer, at about 1nmole/µL.

2. Mix the TCEP slurry and take a volume equal to about 2x the volume of oligo

you want to reduce. Centrifuge and remove the supernatant. If desired you may

wash the TCEP slurry with your reaction buffer.

3. Add your oligo to the TCEP slurry.

4. Incubate at room temperature with constant mixing for 2 hours.

5. Transfer the reaction to a PALL Nanosep 100 microspin column.

6. Centrifuge for 5 min at 10000 rpm. The filtrate contains the reduced thiol oligo.

To ensure full reactivity, thiol-modified oligos should be used immediately after

reduction.

3. From the prepared probe, make a solution of 5µM probe, 25mM NaCl, 25mM

phosphate buffer and 20mM MgCl2.

4. Leave the solution in 3 on the chip for 60 minutes in the room temperature in a

humidity chamber in dark.

5. Remove solution left in 4 and wash with 25mM phosphate buffer and 25mM

NaCl.

6. Repeat the same scan as in 1.

7. Make the DNA target solution as per protocol.

• Product: 25 nmole DNA Oligo



• Sequence: ATA AGG CTT CCT GCC GCG CT

There is a protocol although much less complicated than the Thiol reduction. Here

is what we recommend:

1- Spin tube down briefly before opening. This ensures the pellet is at the bottom of

the tube and will not be lost when opening.

2- Under a hood, or in a clean environment, carefully remove cap.

3- Resuspend to desired concentration using TE buffer or DNase free H2O.

Generally, people prefer to resuspend to 100µM stock concentrations. To do this,

you would need to multiply the number of nmoles by 10 and add that many ul of water

or buffer. As an example: If you receive 18.2nm of product, you would resuspend in

182 µl of water or buffer to obtain a 100µM conc. Alternatively, you could resuspend to

any concentration you wish. It is sometimes helpful to use our resuspension calculator

which can be found here:

http://www.idtdna.com/analyzer/Applications/resuspensioncalc/

8. Make a solution of 5µM target, 100mM MgCl2, 25mM NCl, 25mM phosphate

buffer.

9. Leave a drop of solution made in 8 on chip for 60 minutes at 37 degrees in a dark

humidity chamber

10. Remove the drop and wash with 25mM phosphate buffer and 25mM NaCl.

11. Scan the electrode as 1.

12. Repeat the experiment for the other negative target

• Product: 25 nmole DNA Oligo


• Sequence: TTT TTT TTT TTT TTT TTT TT


A.3 Data Acquisition

Data acquisition from the board is done using the FTDI USB controller chip. The chip

is connected to the host PC using a USB cable and matlab is used to control and acquire

the data from the board.

Before running the matlab code the user should setup two virtual COM port using

FT-PROG software available from http://www.ftdichip.com/Drivers/VCP.htm.

The matlab code for recording CV plots and calculating the SNR is given below:1 clear all;

2 format long;

3 s1 = serial(’COM57’);

4 s2 = serial(’COM58’);

5 set(s1,’BaudRate’,915200,’DataBits’,8,’Parity’,’none’,’StopBits’,1);

6 set(s2,’BaudRate’,915200,’DataBits’,8,’Parity’,’none’,’StopBits’,1);

78 N=30000;

9 S1BUFF = N;

10 S2BUFF = 2;

1112 s1.InputBufferSize = S1BUFF;

13 s1.OutputBufferSize = S1BUFF;

1415 s2.InputBufferSize = S2BUFF;

16 s2.OutputBufferSize = S2BUFF;

1718 s1.Timeout = 600;

19 s2.Timeout = 600;

2021 fopen(s1);

22 s1.ByteOrder = ’bigEndian’;

2324 x=[4 3];

25 y=[4 4];

2627 fopen(s2);

28 s2.ByteOrder = ’bigEndian’;

29 %

30 fwrite(s2, x , ’int8’);

3132 % fwrite(s2, ’11111110’);

33 data = fread(s1);

34 values(:,1) = data;

3536 fwrite(s2, y , ’int8’);

3738 %just to check properties

39 get(s1)


40 get(s2)

4142 fclose(s1)

43 delete(s1)

44 clear s1

4546 fclose(s2)

47 delete(s2)

48 clear s2

4950 x=0;

51 y=0;

52 z=1;

53 hamed =1;

54 BB1 = 0;

55 BB2 = 0;

56 BB3 = 0;

5758 for i = 1 : S1BUFF

59 str = dec2bin(values(i),8);

60 B11 = bitget(values(i), 8:-1:1);

61 BB = B11(8)*1 +B11(7)*2;

62 for xx = 1:8

63 B(xx)=bitcmp(B11(xx),1);

64 end

65 if (BB == 0 & x==0)

66 x=1;

67 y=1;

68 end

69 if(BB == 0 & y==1 & z==1)

70 BB1 =BB1+1;

71 ii =1;

72 for ii = 1 : 6

73 p1(ii,BB1) = B(ii);

74 end

75 z=2;

76 elseif(BB == 1 & y==1 & z==2)

77 BB2 =BB2+1;

78 for ii = 1 : 6

79 p2(ii,BB2) = B(ii);

80 end

81 z=3;

82 elseif(BB == 2 & y==1 & z==3)

83 BB3 =BB3+1;

84 ii =1;

85 for ii = 1 : 6

86 p3(ii,BB3) = B(ii);

87 end

88 z=1;

89 end

9091 end

9293 for i = 1 : N/3-20


94 OUTF1(i) = p1(1,i)*2ˆ5 + p1(2,i)*2ˆ4 + p1(3,i)*2ˆ3 + p1(4,i)*2ˆ2 + p1(5,i)*2ˆ1 + p1(6,i)*2ˆ0;

95 OUTF2(i) = p2(1,i)*2ˆ11 + p2(2,i)*2ˆ10 + p2(3,i)*2ˆ9 + p2(4,i)*2ˆ8 + p2(5,i)*2ˆ7 + p2(6,i)*2ˆ6;

96 OUTF3(i) = p3(3,i)*2ˆ15 + p3(4,i)*2ˆ14 + p3(5,i)*2ˆ13 + p3(6,i)*2ˆ12;

97 if (p3(2,i) == 1)

98 OUTF(i) = OUTF1(i)+OUTF2(i)+OUTF3(i);

99 else

100 OUTF(i) = (OUTF1(i)+OUTF2(i)+OUTF3(i))*-1;

101 end

102 end

103 H=1;

104 hh=0;

105 for i = 1 : N/3-21

106 if( OUTF(i)== 65535)

107 hh = 1;

108 elseif (OUTF(i)˜=65535)

109 if(hh == 1)

110 OUTFF(H) = 65535;

111 hh = 0;

112 H=H+1;

113 else

114 OUTFF(H) = OUTF(i);

115 H=H+1;

116 end

117 end

118 end

119120 % figure(1)

121 % X1 = 10:1:H-1;

122 % y = smooth(OUTFF(10:1:H-1));

123 % plot(X1,y)

124 % xlabel(’Sample’)

125 % ylabel(’Current (A)’)

126127 % figure(1)

128129 X1 = 4:1:N/12-21

130 y1 = OUTFF(4:1:N/12-21);

131 y2 = smooth(OUTFF(4:1:N/12-21),0.03,’rloess’);

132 % for i = 1:1:(10898)

133 % if y1(i,1)>= 0

134 % y1(i,1)=y1(i,1);

135 % elseif y1(i,1)<0

136 % y1(i,1)=y1(i,1)*1.5;

137 % end

138 % end

139140 figure(1);

141 subplot(2,1,1), plot(X1,y2)

142 subplot(2,1,2), plot(X1,y1)

143 xlabel(’Sample’)

144 ylabel(’Current (A)’)

145 % plot(X1,y2)

146 N=size(y1);

147 MM = N(1);


148 MM2=MM-10;

149 M = mean(y1)

150151 k=1;

152 for i=1:length(y1)

153 if y1(i)==65535

154 time_stamp(k)=i;

155 k=k+1;

156 end;

157 end;

158159 for i=1:length(time_stamp)-1

160 m=(time_stamp(i+1) + time_stamp(i))/2;

161 d1=y2(time_stamp(i):m)’;

162 d2=flipdim( y2(m:time_stamp(i+1)),1)’ ;

163 NN=min([length(d1),length(d2)]);

164 hamed_data(i,1,:) = d1(1:NN);

165 hamed_data(i,2,:) = d2(1:NN);

166 end;

167168169 figure(2);

170 for i=1:length(time_stamp)-1

171 plot(-1*[1:NN],-1*squeeze(hamed_data(i,1,:)),-1*[1:NN],-1*squeeze(hamed_data(i,2,:)));

172 hold on;

173 end

174175 fid = fopen(’C:\Users\hamed\Desktop\CV_july_0002_h1.txt’,’w’);

176 fprintf(fid, ’%d’, hamed_data(i,1,:));

177 fclose(fid);

178 fid = fopen(’C:\Users\hamed\Desktop\CV_july3_0002_h2.txt’,’w’);

179 fprintf(fid, ’%d’, hamed_data(i,2,:));

180 fclose(fid);

181 fid = fopen(’C:\Users\hamed\Desktop\CV_july3_0002_y1.txt’,’w’);

182 fprintf(fid, ’%d’, y1);

183 fclose(fid);

184 fid = fopen(’C:\Users\hamed\Desktop\CV_july3_0002_y2.txt’,’w’);

185 fprintf(fid, ’%d’, y2);

186 fclose(fid);

187188 % fs=100;

189 % T = 1/fs;

190 % t = MM2 * T;

191 % w= window(@blackmanharris,MM);

192 % data2=y1;

193 % y1 = y1.*w;

194 % out_fft = fft(y1);

195 % psd = abs(out_fft).ˆ2;

196 %

197 % % scale and plot

198 % psd(1:5,1)=psd(10,1);

199 % psd(end-5:end,1)=psd(end-10,1);

200 % [max_value fbin] = max (psd);

201 % P = fbin


202 % fin = P/(842) * fs

203 % psd = psd./max_value;

204 % figure(2)

205 % f = fs/2000*linspace(0,1,MM2/2);

206 % % plot (f,10*log10(psd(1:MM2/2)));

207 % % plot (10*log10(psd(1:N/2)));

208 %

209 % nbins =13; % number of signal bins

210 % signal_bins = P +1+ [-(nbins-1)/2:(nbins-1)/2]

211 % inband_bins = 1:842/2;

212 %

213 % noise_bins = setdiff(inband_bins,signal_bins);

214 % noise_bins = setdiff(noise_bins,[1 2 3 4 5 6 7 8 9 10 11 12]);

215 %

216 % snrOut = 10*log10(sum(psd(signal_bins)))- 10*log10(sum(sum(psd(noise_bins))))

217 % X=psd(noise_bins);

A.4 FPGA Verilog Code

The CMOS DNA detection microsystem and the PCB board are controlled by the on-

board FPGA.

The top level FPGA verilog code which is shared between all mode of operations

(CV, IS and temperature regulation) is given below:

A.4.1 Top Level Module

1 // Copyright (C) 1991-2011 Altera Corporation

2 // Your use of Altera Corporation’s design tools, logic functions

3 // and other software and tools, and its AMPP partner logic

4 // functions, and any output files from any of the foregoing

5 // (including device programming or simulation files), and any

6 // associated documentation or information are expressly subject

7 // to the terms and conditions of the Altera Program License

8 // Subscription Agreement, Altera MegaCore Function License

9 // Agreement, or other applicable license agreement, including,

10 // without limitation, that your use is for the sole purpose of

11 // programming logic devices manufactured by Altera and sold by

12 // Altera or its authorized distributors. Please refer to the

13 // applicable agreement for further details.

1415 // PROGRAM "Quartus II 64-Bit"

16 // VERSION "Version 11.0 Build 157 04/27/2011 SJ Full Version"

17 // CREATED "Fri Jun 21 22:26:40 2013"

18


19 module CHEM_PRO_TOP(

20 OSC_CLK,

21 TXE_USB,

22 RXF_USB_IN,

23 DATA_IN,

24 DIN_USB,

25 CS,

26 WR,

27 PD,

28 GAIN,

29 CLR,

30 ldac1,

31 sync1,

32 ldac2,

33 sync2,

34 din_out2,

35 din_out1,

36 SING,

37 C1,

38 VLATCH,

39 P1,

40 EXT_RESET,

41 BS_A,

42 BS_P,

43 Q,

44 LDAC,

45 DAC_CLK1,

46 DAC_CLK2,

47 CLK_SH1,

48 I_STAT_PAD,

49 CLK_SH2_TEMP,

50 IN_E_PAD_TEMP,

51 CLK_SHIFT_PAD,

52 I_SHIFT_PAD,

53 RESET_CH,

54 CLK_CH,

55 CLK_CV,

56 UP_DOWN_CV,

57 IMP_CV,

58 RESET_SRAM_LOGIC,

59 W,

60 CLK_COEF,

61 OFF_ON_CHIP,

62 COS,

63 OFF_CHIP_DAC_P,

64 ELECTRODE_OFFON,

65 READ_PWM,

66 CLK_SHFIT_RING,

67 CLK_RING,

68 READ_ON_OFF_PWM,

69 RESET_RING,

70 WRITE_PAD,

71 EXT_RING,

72 ON_OFF_RING,


73 B_PID,

74 A_PID,

75 ADC_PID,

76 ADC_MEM_READ,

77 MAINE_READ,

78 RESET_,

79 WP0,

80 WP1,

81 WP2,

82 WP3,

83 RESET_P,

84 CLK_P,

85 PAR_P,

86 RESET_PS,

87 CLK_PS,

88 LOAD_SHIFT,

89 RESET_SHIFT,

90 IN_COEF_PAD,

91 CLK_SH3_PAD,

92 IN_SHIFT_RING_PAD,

93 CLK_TEMP,

94 CHAMBER_RESET,

95 CHAMBER_CLK,

96 WR_USB,

97 RD_USB,

98 SIWUA,

99 SIWUB,

100 B,

101 DATA_OUT,

102 DATA_OUT_USB

103 );

104105106 input wire OSC_CLK;

107 input wire TXE_USB;

108 input wire RXF_USB_IN;

109 input wire [16:0] DATA_IN;

110 input wire [7:0] DIN_USB;

111 output wire CS;

112 output wire WR;

113 output wire PD;

114 output wire GAIN;

115 output wire CLR;

116 output wire ldac1;

117 output wire sync1;



120 output wire din_out2;


122 output wire SING;

123 output wire C1;

124 output wire VLATCH;

125 output wire P1;

126 output wire EXT_RESET;


127 output wire BS_A;

128 output wire BS_P;

129 output wire Q;

130 output wire LDAC;

131 output wire DAC_CLK1;


133 output wire CLK_SH1;

134 output wire I_STAT_PAD;

135 output wire CLK_SH2_TEMP;

136 output wire IN_E_PAD_TEMP;

137 output wire CLK_SHIFT_PAD;

138 output wire I_SHIFT_PAD;

139 output wire RESET_CH;

140 output wire CLK_CH;

141 output wire CLK_CV;

142 output wire UP_DOWN_CV;

143 output wire IMP_CV;

144 output wire RESET_SRAM_LOGIC;

145 output wire W;

146 output wire CLK_COEF;

147 output wire OFF_ON_CHIP;

148 output wire COS;

149 output wire OFF_CHIP_DAC_P;

150 output wire ELECTRODE_OFFON;

151 output wire READ_PWM;

152 output wire CLK_SHFIT_RING;

153 output wire CLK_RING;

154 output wire READ_ON_OFF_PWM;

155 output wire RESET_RING;

156 output wire WRITE_PAD;

157 output wire EXT_RING;

158 output wire ON_OFF_RING;

159 output wire B_PID;

160 output wire A_PID;

161 output wire ADC_PID;

162 output wire ADC_MEM_READ;

163 output wire MAINE_READ;

164 output wire RESET_;

165 output wire WP0;




169 output wire RESET_P;

170 output wire CLK_P;

171 output wire PAR_P;

172 output wire RESET_PS;

173 output wire CLK_PS;

174 output wire LOAD_SHIFT;

175 output wire RESET_SHIFT;

176 output wire IN_COEF_PAD;

177 output wire CLK_SH3_PAD;

178 output wire IN_SHIFT_RING_PAD;

179 output wire CLK_TEMP;

180 output wire CHAMBER_RESET;


181 output wire CHAMBER_CLK;

182 output wire WR_USB;

183 output wire RD_USB;

184 output wire SIWUA;

185 output wire SIWUB;

186 output wire [9:0] B;

187 output wire [16:0] DATA_OUT;

188 output wire [7:0] DATA_OUT_USB;

189190 wire CLK_DAC;

191 wire [17:0] DIN;

192 wire idel;

193 wire idel2;

194 wire idel3;

195 wire idel4;

196 wire start_DAC;

197 wire SYNTHESIZED_WIRE_25;








205 wire [16:0] SYNTHESIZED_WIRE_10;









214215 assign DAC_CLK1 = SYNTHESIZED_WIRE_23;


217 assign CLK_CH = SYNTHESIZED_WIRE_24;

218 assign RESET_SHIFT = SYNTHESIZED_WIRE_27;

219 assign CLK_TEMP = SYNTHESIZED_WIRE_24;

220 assign DATA_OUT = SYNTHESIZED_WIRE_10;

221222223224225 Reference_generator b2v_inst(

226 .clkIn(OSC_CLK),

227 .CS(CS),

228 .WR(WR),

229 .LDAC(LDAC),

230 .PD(PD),

231 .GAIN(GAIN),

232 .CLR(CLR),

233 .stop(SYNTHESIZED_WIRE_12),

234 .counter(B));


235236237 CH_TIMING b2v_inst1(


239 .RESET_MASTER(SYNTHESIZED_WIRE_25),

240 .SING(SING),

241 .C1(C1),

242 .VLATCH(VLATCH),

243 .P1(P1),

244 .EXT_RESET(EXT_RESET),

245 .BS_A(BS_A),

246 .BS_P(BS_P),

247 .Q(Q));

248249250 Inv b2v_inst10(

251 .a(idel2),

252 .y(SYNTHESIZED_WIRE_21));

253254255 andgate b2v_inst11(

256 .a(SYNTHESIZED_WIRE_1),

257 .b(SYNTHESIZED_WIRE_2),

258 .y(CLK_SHIFT_PAD));

259260261 Inv b2v_inst12(

262 .a(idel3),


264265266 Channel_Control b2v_inst13(


268 .RESET_IN(SYNTHESIZED_WIRE_25),

269 .RESET_CH(RESET_CH),

270 .CLK_CH(SYNTHESIZED_WIRE_24));




276 .RESET_CH(CHAMBER_RESET),

277 .CLK_CH(CHAMBER_CLK));

278279280 SHIFT_REG_RESET b2v_inst15(


282 .RESET_SHIFT(SYNTHESIZED_WIRE_27));

283284285 PID_CONT b2v_inst17(


287 .READ_PWM(READ_PWM),

288 .CLK_SHIFT_RING(CLK_SHFIT_RING),


289 .IN_SHIFT(IN_SHIFT_RING_PAD),

290 .CLK_RING(CLK_RING),

291 .READ_ON_OFF_PWM(READ_ON_OFF_PWM),

292 .RESET_RING(RESET_RING),

293 .WRITE_PAD(WRITE_PAD),

294 .EXT_RING(EXT_RING),

295 .ON__OFF_RING(ON_OFF_RING),

296 .B_PID(B_PID),

297 .A_PID(A_PID),

298 .ADC_PID(ADC_PID),

299 .ADC_MEM_READ(ADC_MEM_READ),

300 .MAINE_READ(MAINE_READ),

301 .RESET_(RESET_),

302 .WP0(WP0),

303 .WP1(WP1),

304 .WP2(WP2),

305 .WP3(WP3),

306 .RESET_P(RESET_P),

307 .CLK_P(CLK_P),

308 .PAR_P(PAR_P));

309310311 TX b2v_inst18(


313 .RESET_PS(RESET_PS),

314 .CLK_PS(CLK_PS),

315 .LOADSHIFT(LOAD_SHIFT));




321 .y(CLK_SH3_PAD));

322323324 Voltage_DAC_cont b2v_inst2(


326 .start(start_DAC),

327 .clkout(CLK_DAC),

328 .DIN(DIN));

329330331 Inv b2v_inst20(

332 .a(idel4),


334335336 Datap b2v_inst21(

337 .clkIn(SYNTHESIZED_WIRE_26),

338 .start(SYNTHESIZED_WIRE_8),

339 .TXE(TXE_USB),

340 .reset(SYNTHESIZED_WIRE_9),

341 .DIN(SYNTHESIZED_WIRE_10),

342 .WR(WR_USB),


343 .SIWUA(SIWUA),

344 .DATA_OUT(DATA_OUT_USB));

345346347 IO_B b2v_inst22(



350 .DATA_IN(DATA_IN),

351 .DATA_OUT(SYNTHESIZED_WIRE_10));

352353354 Startp b2v_inst23(


356 .RXF(RXF_USB_IN),

357 .DIN(DIN_USB),


359 .RD(RD_USB),

360 .SIWUB(SIWUB),

361 .reset(SYNTHESIZED_WIRE_9)

362 );

363364365 DummyData b2v_inst24(

366 .clkIn(SYNTHESIZED_WIRE_26)

367 );

368369370 divider_usb b2v_inst25(


372 .clk_out(SYNTHESIZED_WIRE_26));

373374375 Shift_Reg_COEFF b2v_inst29(


377 .RESET_SHIFT(SYNTHESIZED_WIRE_27),

378 .CLK_SH3(SYNTHESIZED_WIRE_6),

379 .I_COEFF_PAD(IN_COEF_PAD),

380 .IDEL_COEFF(idel4));

381382383 Shift_Reg_TEMP b2v_inst3(




387 .I_STAT_PAD(IN_E_PAD_TEMP),

388 .idel_TMEP(idel2));

389390391 SRAM b2v_inst30(


393 .CLK_CV(CLK_CV),

394 .UP_DOWN_CV(UP_DOWN_CV),

395 .IMP_CV(IMP_CV),

396 .RESET_SRAM_LOGIC(RESET_SRAM_LOGIC),


397 .W(W),

398 .CLK_COEF(CLK_COEF),

399 .OFF_ON_CHIP(OFF_ON_CHIP),

400 .COS(COS),

401 .OFF_CHIP_DAC_P(OFF_CHIP_DAC_P),

402 .ELECTRODE_OFFON(ELECTRODE_OFFON));

403404405 Voltage_DAC b2v_inst4(

406 .clkIn(CLK_DAC),


408 .Data(DIN),

409 .clkDAC(SYNTHESIZED_WIRE_23),

410 .din1(din_out1),

411 .ldac1(ldac1),

412 .sync1(sync1),


414 .ldac2(ldac2),

415 .sync2(sync2));

416417418 Shift_Reg_TEMP_STAT b2v_inst5(



421 .CLK_SHIFT_PAD(SYNTHESIZED_WIRE_2),

422 .I_SHIFT_PAD(I_SHIFT_PAD),

423 .idel_TMEP_STAT(idel3));

424425426 Shift_Reg_POT b2v_inst6(



429 .clk_POT(SYNTHESIZED_WIRE_20),

430 .I_STAT_PAD_POT(I_STAT_PAD),

431 .idel_POT(idel));

432433434 RESET_CONT b2v_inst64(


436 .IDEL1(idel),

437 .IDEL2(idel2),

438 .IDEL3(idel3),

439 .IDEL4(idel4),

440 .RESET_OUT(SYNTHESIZED_WIRE_25));




446 .y(CLK_SH1));

447448449 Inv b2v_inst8(

450 .a(idel),






457 .y(CLK_SH2_TEMP));

458459460 endmodule

This code must be run prior to each measurements.

The Verilog code for each of the submodules are given below:

A.4.2 USB Controller

1 ‘timescale 1ns / 1ps

2 ////////////////////////////////////////////////////////////////////////////////

3 // Company: U of T

4 // Engineer: Hamed

5 //

6 // Create Date: 2:44pm 12/08/2010

7 // Module Name: Voltage_DAC_cont

8 // Target Device: EP3C10-FBGA256-2

9 //

10 ////////////////////////////////////////////////////////////////////////////////

11 module Datap(clkIn, start, DIN, TXE, WR, DATA_OUT, SIWUA, reset);

1213 input clkIn;

14 input start;

15 input TXE;

16 input [16:0] DIN; //16

17 output WR;

18 output [7:0] DATA_OUT;

19 output SIWUA;

20 input reset;

2122 //----------------------------------------------------------------------------

23 reg [29:0] count = 0;

24 reg WR = 0;

25 reg clkout;

26 reg [4:0] timer = 5’b00000;

27 reg [16:0] DATA_MID; //16

28 reg [7:0] DATA_OUT;

29 reg [7:0] DATAA;

30 reg [7:0] DATAB;

31 reg [7:0] DATAC;

32 reg [2:0] run = 3’b000;

33 reg SIWUA;

34 //----------------------------------------------------------------------------

35


3637 always @ (posedge clkIn)

38 begin

39 if (count < 40) //200

40 begin

41 clkout <= 1;

42 count <= count+1;

43 end

44 else if (count < 79) //399

45 begin

46 clkout <= 0;


48 end

49 if (count == 79) //399

50 count <= 0;

51 end

5253 //----------------------------------------------------------------------------

5455 always @ (posedge clkout)

56 begin

5758 SIWUA <= 1;

5960 // if (reset == 0)

61 // begin

62 // timer <= 0;

63 // end

64 if (!start)

65 begin

66 DATAA <= 8’b00000000;

67 DATAB <= 8’b00000001;

68 DATAC <= 8’b00000010;

69 end

7071 if (˜TXE && start)

72 begin

73 case (timer)

74 0:

75 begin

76 DATA_MID <= DIN;

77 end

78 5:

79 begin

80 WR <= 1;

81 end

82 10:

83 begin

84 if (run == 0)

85 begin

8687 DATAA[7:2] <= DATA_MID[5:0];

88 // DATAA[7] <= DATA_MID[0];







94 DATA_OUT <= DATAA[7:0];

95 run <= run +1 ;

96 end

97 else if (run == 1)

98 begin

99100 DATAB[7:2] <= DATA_MID[11:6];

101 // DATAA[7] <= DATA_MID[6];

102 // DATAA[6] <= DATA_MID[7];

103 // DATAA[5] <= DATA_MID[8];

104 // DATAA[4] <= DATA_MID[9];

105 // DATAA[3] <= DATA_MID[10];

106 // DATAA[2] <= DATA_MID[11];

107 DATA_OUT <= DATAB[7:0]; //[15:8]

108 run <= run +1 ;

109 end

110 else if (run == 2)

111 begin

112113 DATAC[6:2] <= DATA_MID[16:12];

114 // DATAA[6] <= DATA_MID[12];

115 // DATAA[5] <= DATA_MID[13];

116 // DATAA[4] <= DATA_MID[14];

117 // DATAA[3] <= DATA_MID[15];

118 // DATAA[2] <= DATA_MID[16];

119 DATA_OUT <= DATAC[7:0]; //[16:9]

120 run <= 0;

121 end

122 end

123 15:

124 begin

125 WR <= 0;

126 end

127 endcase

128 end

129130 if (timer < 16 && ˜TXE && reset)

131 timer <= timer + 1;

132 else if (timer == 16 || ˜reset)

133 timer <= 0;

134135136 end

137 endmodule

A.4.3 Master Clock Divider


2 ////////////////////////////////////////////////////////////////////////////////




5 //

6 // Create Date: 2:44pm 12/08/2010

7 // Module Name: Channel_Control


9 //

10 ////////////////////////////////////////////////////////////////////////////////

11 module Divider(clkIn, clkout);

12 ///////////////////////////////////////////////////////////////////////////////

13 input clkIn;

14 output clkout;

15 ///////////////////////////////////////////////////////////////////////////////

16 reg clkout;

17 ///////////////////////////////////////////////////////////////////////////////

18 reg [29:0] count = 0;

19 ///////////////////////////////////////////////////////////////////////////////


21 begin

22 if (count < 4)

23 begin

24 clkout <= 1;


26 end

27 else if (count < 7)

28 begin

29 clkout <= 0;


31 end

32 if (count == 7)

33 count <= 0;

34 end

35 ///////////////////////////////////////////////////////////////////////////////

3637 endmodule

38 ///////////////////////////////////////////////////////////////////////////////

A.4.4 USB Clock Divider


2 ////////////////////////////////////////////////////////////////////////////////



5 //

6 // Create Date: 2:44pm 12/08/2010



9 //

10 ////////////////////////////////////////////////////////////////////////////////

11 module divider_usb(clkIn, clk_out);

12 ///////////////////////////////////////////////////////////////////////////////

13 input clkIn;

14 output clk_out;


15 ///////////////////////////////////////////////////////////////////////////////

16 reg clk_out;

1718 reg [9:0] timer = 10’b0000000000;

19 ///////////////////////////////////////////////////////////////////////////////

20 reg [29:0] count = 0;

21 ///////////////////////////////////////////////////////////////////////////////


23 begin

24 if (count < 8)

25 begin

26 clk_out <= 1;


28 end


30 begin

31 clk_out <= 0;


33 end

34 if (count == 15)

35 count <= 0;

36 end

37 endmodule

38 ///////////////////////////////////////////////////////////////////////////////

A.4.5 One Channel FIFO Controller

1 // megafunction wizard: %FIFO%

2 // GENERATION: STANDARD

3 // VERSION: WM1.0

4 // MODULE: dcfifo

56 // ============================================================

7 // File Name: FIFO_ONE_CH.v

8 // Megafunction Name(s):

9 // dcfifo

10 //

11 // Simulation Library Files(s):

12 // altera_mf

13 // ============================================================

14 // ************************************************************

15 // THIS IS A WIZARD-GENERATED FILE. DO NOT EDIT THIS FILE!

16 //

17 // 10.0 Build 218 06/27/2010 SJ Full Version

18 // ************************************************************

192021 //Copyright (C) 1991-2010 Altera Corporation

22 //Your use of Altera Corporation’s design tools, logic functions

23 //and other software and tools, and its AMPP partner logic

24 //functions, and any output files from any of the foregoing

25 //(including device programming or simulation files), and any

26 //associated documentation or information are expressly subject


27 //to the terms and conditions of the Altera Program License

28 //Subscription Agreement, Altera MegaCore Function License

29 //Agreement, or other applicable license agreement, including,

30 //without limitation, that your use is for the sole purpose of

31 //programming logic devices manufactured by Altera and sold by

32 //Altera or its authorized distributors. Please refer to the

33 //applicable agreement for further details.

343536 // synopsys translate_off

37 ‘timescale 1 ps / 1 ps

38 // synopsys translate_on

39 module FIFO_ONE_CH (

40 aclr,

41 data,

42 rdclk,

43 rdreq,

44 wrclk,

45 wrreq,

46 q,

47 rdempty,

48 rdfull,

49 wrempty,

50 wrfull);

5152 input aclr;

53 input [16:0] data;

54 input rdclk;

55 input rdreq;

56 input wrclk;

57 input wrreq;

58 output [16:0] q;

59 output rdempty;

60 output rdfull;

61 output wrempty;

62 output wrfull;

63 ‘ifndef ALTERA_RESERVED_QIS


65 ‘endif

66 tri0 aclr;

67 ‘ifndef ALTERA_RESERVED_QIS


69 ‘endif

7071 wire sub_wire0;

72 wire sub_wire1;

73 wire [16:0] sub_wire2;

74 wire sub_wire3;

75 wire sub_wire4;

76 wire wrempty = sub_wire0;

77 wire wrfull = sub_wire1;

78 wire [16:0] q = sub_wire2[16:0];

79 wire rdempty = sub_wire3;

80 wire rdfull = sub_wire4;


8182 dcfifo dcfifo_component (

83 .rdclk (rdclk),

84 .wrclk (wrclk),

85 .wrreq (wrreq),

86 .aclr (aclr),

87 .data (data),

88 .rdreq (rdreq),

89 .wrempty (sub_wire0),

90 .wrfull (sub_wire1),

91 .q (sub_wire2),

92 .rdempty (sub_wire3),

93 .rdfull (sub_wire4),

94 .rdusedw (),

95 .wrusedw ());

96 defparam

97 dcfifo_component.intended_device_family = "Cyclone III",

98 dcfifo_component.lpm_numwords = 131072,

99 dcfifo_component.lpm_showahead = "OFF",

100 dcfifo_component.lpm_type = "dcfifo",

101 dcfifo_component.lpm_width = 17,

102 dcfifo_component.lpm_widthu = 17,

103 dcfifo_component.overflow_checking = "ON",

104 dcfifo_component.rdsync_delaypipe = 8,

105 dcfifo_component.underflow_checking = "ON",

106 dcfifo_component.use_eab = "ON",

107 dcfifo_component.write_aclr_synch = "ON",

108 dcfifo_component.wrsync_delaypipe = 8;

109110111 endmodule

112113 // ============================================================

114 // CNX file retrieval info

115 // ============================================================

116 // Retrieval info: PRIVATE: AlmostEmpty NUMERIC "0"

117 // Retrieval info: PRIVATE: AlmostEmptyThr NUMERIC "-1"

118 // Retrieval info: PRIVATE: AlmostFull NUMERIC "0"

119 // Retrieval info: PRIVATE: AlmostFullThr NUMERIC "-1"

120 // Retrieval info: PRIVATE: CLOCKS_ARE_SYNCHRONIZED NUMERIC "0"

121 // Retrieval info: PRIVATE: Clock NUMERIC "4"

122 // Retrieval info: PRIVATE: Depth NUMERIC "131072"

123 // Retrieval info: PRIVATE: Empty NUMERIC "1"

124 // Retrieval info: PRIVATE: Full NUMERIC "1"

125 // Retrieval info: PRIVATE: INTENDED_DEVICE_FAMILY STRING "Cyclone III"

126 // Retrieval info: PRIVATE: LE_BasedFIFO NUMERIC "0"

127 // Retrieval info: PRIVATE: LegacyRREQ NUMERIC "1"

128 // Retrieval info: PRIVATE: MAX_DEPTH_BY_9 NUMERIC "0"

129 // Retrieval info: PRIVATE: OVERFLOW_CHECKING NUMERIC "0"

130 // Retrieval info: PRIVATE: Optimize NUMERIC "1"

131 // Retrieval info: PRIVATE: RAM_BLOCK_TYPE NUMERIC "0"

132 // Retrieval info: PRIVATE: SYNTH_WRAPPER_GEN_POSTFIX STRING "0"

133 // Retrieval info: PRIVATE: UNDERFLOW_CHECKING NUMERIC "0"

134 // Retrieval info: PRIVATE: UsedW NUMERIC "1"


135 // Retrieval info: PRIVATE: Width NUMERIC "17"

136 // Retrieval info: PRIVATE: dc_aclr NUMERIC "1"

137 // Retrieval info: PRIVATE: diff_widths NUMERIC "0"

138 // Retrieval info: PRIVATE: msb_usedw NUMERIC "0"

139 // Retrieval info: PRIVATE: output_width NUMERIC "17"

140 // Retrieval info: PRIVATE: rsEmpty NUMERIC "1"

141 // Retrieval info: PRIVATE: rsFull NUMERIC "1"

142 // Retrieval info: PRIVATE: rsUsedW NUMERIC "0"

143 // Retrieval info: PRIVATE: sc_aclr NUMERIC "0"

144 // Retrieval info: PRIVATE: sc_sclr NUMERIC "1"

145 // Retrieval info: PRIVATE: wsEmpty NUMERIC "1"

146 // Retrieval info: PRIVATE: wsFull NUMERIC "1"

147 // Retrieval info: PRIVATE: wsUsedW NUMERIC "0"

148 // Retrieval info: LIBRARY: altera_mf altera_mf.altera_mf_components.all

149 // Retrieval info: CONSTANT: INTENDED_DEVICE_FAMILY STRING "Cyclone III"

150 // Retrieval info: CONSTANT: LPM_NUMWORDS NUMERIC "131072"

151 // Retrieval info: CONSTANT: LPM_SHOWAHEAD STRING "OFF"

152 // Retrieval info: CONSTANT: LPM_TYPE STRING "dcfifo"

153 // Retrieval info: CONSTANT: LPM_WIDTH NUMERIC "17"

154 // Retrieval info: CONSTANT: LPM_WIDTHU NUMERIC "17"

155 // Retrieval info: CONSTANT: OVERFLOW_CHECKING STRING "ON"

156 // Retrieval info: CONSTANT: RDSYNC_DELAYPIPE NUMERIC "8"

157 // Retrieval info: CONSTANT: UNDERFLOW_CHECKING STRING "ON"

158 // Retrieval info: CONSTANT: USE_EAB STRING "ON"

159 // Retrieval info: CONSTANT: WRITE_ACLR_SYNCH STRING "ON"

160 // Retrieval info: CONSTANT: WRSYNC_DELAYPIPE NUMERIC "8"

161 // Retrieval info: USED_PORT: aclr 0 0 0 0 INPUT GND "aclr"

162 // Retrieval info: USED_PORT: data 0 0 17 0 INPUT NODEFVAL "data[16..0]"

163 // Retrieval info: USED_PORT: q 0 0 17 0 OUTPUT NODEFVAL "q[16..0]"

164 // Retrieval info: USED_PORT: rdclk 0 0 0 0 INPUT NODEFVAL "rdclk"

165 // Retrieval info: USED_PORT: rdempty 0 0 0 0 OUTPUT NODEFVAL "rdempty"

166 // Retrieval info: USED_PORT: rdfull 0 0 0 0 OUTPUT NODEFVAL "rdfull"

167 // Retrieval info: USED_PORT: rdreq 0 0 0 0 INPUT NODEFVAL "rdreq"

168 // Retrieval info: USED_PORT: wrclk 0 0 0 0 INPUT NODEFVAL "wrclk"

169 // Retrieval info: USED_PORT: wrempty 0 0 0 0 OUTPUT NODEFVAL "wrempty"

170 // Retrieval info: USED_PORT: wrfull 0 0 0 0 OUTPUT NODEFVAL "wrfull"

171 // Retrieval info: USED_PORT: wrreq 0 0 0 0 INPUT NODEFVAL "wrreq"

172 // Retrieval info: CONNECT: @aclr 0 0 0 0 aclr 0 0 0 0

173 // Retrieval info: CONNECT: @data 0 0 17 0 data 0 0 17 0

174 // Retrieval info: CONNECT: @rdclk 0 0 0 0 rdclk 0 0 0 0

175 // Retrieval info: CONNECT: @rdreq 0 0 0 0 rdreq 0 0 0 0

176 // Retrieval info: CONNECT: @wrclk 0 0 0 0 wrclk 0 0 0 0

177 // Retrieval info: CONNECT: @wrreq 0 0 0 0 wrreq 0 0 0 0

178 // Retrieval info: CONNECT: q 0 0 17 0 @q 0 0 17 0

179 // Retrieval info: CONNECT: rdempty 0 0 0 0 @rdempty 0 0 0 0

180 // Retrieval info: CONNECT: rdfull 0 0 0 0 @rdfull 0 0 0 0

181 // Retrieval info: CONNECT: wrempty 0 0 0 0 @wrempty 0 0 0 0

182 // Retrieval info: CONNECT: wrfull 0 0 0 0 @wrfull 0 0 0 0

183 // Retrieval info: GEN_FILE: TYPE_NORMAL FIFO_ONE_CH.v TRUE

184 // Retrieval info: GEN_FILE: TYPE_NORMAL FIFO_ONE_CH.inc FALSE

185 // Retrieval info: GEN_FILE: TYPE_NORMAL FIFO_ONE_CH.cmp FALSE

186 // Retrieval info: GEN_FILE: TYPE_NORMAL FIFO_ONE_CH.bsf TRUE FALSE

187 // Retrieval info: GEN_FILE: TYPE_NORMAL FIFO_ONE_CH_inst.v TRUE

188 // Retrieval info: GEN_FILE: TYPE_NORMAL FIFO_ONE_CH_bb.v TRUE


189 // Retrieval info: LIB_FILE: altera_mf

A.4.6 FIFO Controller

1 // megafunction wizard: %FIFO%


3 // VERSION: WM1.0

4 // MODULE: scfifo

56 // ============================================================

7 // File Name: FIFO_OUT.v


9 // scfifo

10 //


12 // altera_mf

13 // ============================================================

14 // ************************************************************


16 //


18 // ************************************************************

















39 module FIFO_OUT (

40 clock,

41 data,

42 rdreq,

43 wrreq,

44 empty,

45 full,

46 q);

4748 input clock;

49 input [16:0] data;


50 input rdreq;

51 input wrreq;

52 output empty;

53 output full;

54 output [16:0] q;

5556 wire sub_wire0;

57 wire sub_wire1;


59 wire empty = sub_wire0;

60 wire full = sub_wire1;

61 wire [16:0] q = sub_wire2[16:0];

6263 scfifo scfifo_component (

64 .clock (clock),

65 .data (data),

66 .rdreq (rdreq),

67 .wrreq (wrreq),

68 .empty (sub_wire0),

69 .full (sub_wire1),

70 .q (sub_wire2),

71 .aclr (),

72 .almost_empty (),

73 .almost_full (),

74 .sclr (),

75 .usedw ());

76 defparam

77 scfifo_component.add_ram_output_register = "ON",

78 scfifo_component.intended_device_family = "Cyclone III",

79 scfifo_component.lpm_numwords = 256,

80 scfifo_component.lpm_showahead = "OFF",

81 scfifo_component.lpm_type = "scfifo",

82 scfifo_component.lpm_width = 17,

83 scfifo_component.lpm_widthu = 8,

84 scfifo_component.overflow_checking = "ON",

85 scfifo_component.underflow_checking = "ON",

86 scfifo_component.use_eab = "ON";

878889 endmodule

9091 // ============================================================


93 // ============================================================

94 // Retrieval info: PRIVATE: AlmostEmpty NUMERIC "0"

95 // Retrieval info: PRIVATE: AlmostEmptyThr NUMERIC "-1"

96 // Retrieval info: PRIVATE: AlmostFull NUMERIC "0"

97 // Retrieval info: PRIVATE: AlmostFullThr NUMERIC "-1"

98 // Retrieval info: PRIVATE: CLOCKS_ARE_SYNCHRONIZED NUMERIC "0"

99 // Retrieval info: PRIVATE: Clock NUMERIC "0"

100 // Retrieval info: PRIVATE: Depth NUMERIC "256"

101 // Retrieval info: PRIVATE: Empty NUMERIC "1"

102 // Retrieval info: PRIVATE: Full NUMERIC "1"



104 // Retrieval info: PRIVATE: LE_BasedFIFO NUMERIC "0"

105 // Retrieval info: PRIVATE: LegacyRREQ NUMERIC "1"

106 // Retrieval info: PRIVATE: MAX_DEPTH_BY_9 NUMERIC "0"

107 // Retrieval info: PRIVATE: OVERFLOW_CHECKING NUMERIC "0"

108 // Retrieval info: PRIVATE: Optimize NUMERIC "1"

109 // Retrieval info: PRIVATE: RAM_BLOCK_TYPE NUMERIC "0"


111 // Retrieval info: PRIVATE: UNDERFLOW_CHECKING NUMERIC "0"

112 // Retrieval info: PRIVATE: UsedW NUMERIC "0"

113 // Retrieval info: PRIVATE: Width NUMERIC "17"

114 // Retrieval info: PRIVATE: dc_aclr NUMERIC "0"

115 // Retrieval info: PRIVATE: diff_widths NUMERIC "0"

116 // Retrieval info: PRIVATE: msb_usedw NUMERIC "0"

117 // Retrieval info: PRIVATE: output_width NUMERIC "17"

118 // Retrieval info: PRIVATE: rsEmpty NUMERIC "1"

119 // Retrieval info: PRIVATE: rsFull NUMERIC "0"

120 // Retrieval info: PRIVATE: rsUsedW NUMERIC "0"

121 // Retrieval info: PRIVATE: sc_aclr NUMERIC "0"

122 // Retrieval info: PRIVATE: sc_sclr NUMERIC "0"

123 // Retrieval info: PRIVATE: wsEmpty NUMERIC "0"

124 // Retrieval info: PRIVATE: wsFull NUMERIC "1"

125 // Retrieval info: PRIVATE: wsUsedW NUMERIC "0"


127 // Retrieval info: CONSTANT: ADD_RAM_OUTPUT_REGISTER STRING "ON"


129 // Retrieval info: CONSTANT: LPM_NUMWORDS NUMERIC "256"

130 // Retrieval info: CONSTANT: LPM_SHOWAHEAD STRING "OFF"

131 // Retrieval info: CONSTANT: LPM_TYPE STRING "scfifo"

132 // Retrieval info: CONSTANT: LPM_WIDTH NUMERIC "17"

133 // Retrieval info: CONSTANT: LPM_WIDTHU NUMERIC "8"

134 // Retrieval info: CONSTANT: OVERFLOW_CHECKING STRING "ON"

135 // Retrieval info: CONSTANT: UNDERFLOW_CHECKING STRING "ON"

136 // Retrieval info: CONSTANT: USE_EAB STRING "ON"

137 // Retrieval info: USED_PORT: clock 0 0 0 0 INPUT NODEFVAL "clock"

138 // Retrieval info: USED_PORT: data 0 0 17 0 INPUT NODEFVAL "data[16..0]"

139 // Retrieval info: USED_PORT: empty 0 0 0 0 OUTPUT NODEFVAL "empty"

140 // Retrieval info: USED_PORT: full 0 0 0 0 OUTPUT NODEFVAL "full"

141 // Retrieval info: USED_PORT: q 0 0 17 0 OUTPUT NODEFVAL "q[16..0]"

142 // Retrieval info: USED_PORT: rdreq 0 0 0 0 INPUT NODEFVAL "rdreq"

143 // Retrieval info: USED_PORT: wrreq 0 0 0 0 INPUT NODEFVAL "wrreq"

144 // Retrieval info: CONNECT: @clock 0 0 0 0 clock 0 0 0 0

145 // Retrieval info: CONNECT: @data 0 0 17 0 data 0 0 17 0

146 // Retrieval info: CONNECT: @rdreq 0 0 0 0 rdreq 0 0 0 0

147 // Retrieval info: CONNECT: @wrreq 0 0 0 0 wrreq 0 0 0 0

148 // Retrieval info: CONNECT: empty 0 0 0 0 @empty 0 0 0 0

149 // Retrieval info: CONNECT: full 0 0 0 0 @full 0 0 0 0

150 // Retrieval info: CONNECT: q 0 0 17 0 @q 0 0 17 0

151 // Retrieval info: GEN_FILE: TYPE_NORMAL FIFO_OUT.v TRUE

152 // Retrieval info: GEN_FILE: TYPE_NORMAL FIFO_OUT.inc FALSE

153 // Retrieval info: GEN_FILE: TYPE_NORMAL FIFO_OUT.cmp FALSE

154 // Retrieval info: GEN_FILE: TYPE_NORMAL FIFO_OUT.bsf TRUE FALSE

155 // Retrieval info: GEN_FILE: TYPE_NORMAL FIFO_OUT_inst.v FALSE

156 // Retrieval info: GEN_FILE: TYPE_NORMAL FIFO_OUT_bb.v TRUE



A.4.7 16-bits Parallel to Serial Converter


2 ////////////////////////////////////////////////////////////////////////////////



5 //

6 // Create Date: 2:44pm 12/08/2010

7 // Module Name: parallel to serial


9 //

10 ////////////////////////////////////////////////////////////////////////////////

1112 module parallel_to_serial (

13 clkIn,

14 DATA_IN,

15 start,

16 din1,

17 clkUSB

18 );

1920 // Inputs

21 input clkIn;

22 input [16:0] DATA_IN; // Incoming data

23 input start;

2425 //Wires

26 output wire din1;

27 output wire clkUSB;

28293031 // Registers

32 reg [4:0] timer = 5’b00000;

33 reg [29:0] count = 0;

34 reg [16:0] DATA_MID;

35 reg clkout;

3637 //----------------------------------------------------------------------------

383940 //always @ (posedge clkIn)

41 // begin

42 // if (count < 4) //200

43 // begin

44 // clkout <= 1;

45 // count <= count+1;

46 // end

47 // else if (count < 7) //399

48 // begin

49 // clkout <= 0;

50 // count <= count+1;

51 // end


52 // if (count == 7) //399

53 // count <= 0;

54 // end

5556 //----------------------------------------------------------------------------


60 begin

61 case (timer)

62 0:

63 begin

64 if (start)

65 DATA_MID <= DATA_IN;

66 end

67 17:

68 begin

69 timer <=0;

70 end

71 endcase

72 if (timer < 17 & start)


7475 if (timer < 15 & start)

76 DATA_MID <= DATA_MID[15:0] , DATA_MID[16];

777879 end

8081 assign din1 = DATA_MID[16];

82 assign clkUSB = clkIn && start;

8384 endmodule

A.4.8 Timing PLL

1 // megafunction wizard: %ALTPLL%


3 // VERSION: WM1.0

4 // MODULE: altpll

56 // ============================================================

7 // File Name: PLL1.v


9 // altpll

10 //


12 // altera_mf

13 // ============================================================

14 // ************************************************************


16 //



18 // ************************************************************

















39 module PLL1 (

40 inclk0,

41 c0);

4243 input inclk0;

44 output c0;


47 wire [0:0] sub_wire4 = 1’h0;

48 wire [0:0] sub_wire1 = sub_wire0[0:0];

49 wire c0 = sub_wire1;

50 wire sub_wire2 = inclk0;

51 wire [1:0] sub_wire3 = sub_wire4, sub_wire2;

5253 altpll altpll_component (

54 .inclk (sub_wire3),

55 .clk (sub_wire0),

56 .activeclock (),

57 .areset (1’b0),

58 .clkbad (),

59 .clkena (61’b1),

60 .clkloss (),

61 .clkswitch (1’b0),

62 .configupdate (1’b0),

63 .enable0 (),

64 .enable1 (),

65 .extclk (),

66 .extclkena (41’b1),

67 .fbin (1’b1),

68 .fbmimicbidir (),

69 .fbout (),

70 .fref (),

71 .icdrclk (),


72 .locked (),

73 .pfdena (1’b1),

74 .phasecounterselect (41’b1),

75 .phasedone (),

76 .phasestep (1’b1),

77 .phaseupdown (1’b1),

78 .pllena (1’b1),

79 .scanaclr (1’b0),

80 .scanclk (1’b0),

81 .scanclkena (1’b1),

82 .scandata (1’b0),

83 .scandataout (),

84 .scandone (),

85 .scanread (1’b0),

86 .scanwrite (1’b0),

87 .sclkout0 (),

88 .sclkout1 (),

89 .vcooverrange (),

90 .vcounderrange ());

91 defparam

92 altpll_component.bandwidth_type = "AUTO",

93 altpll_component.clk0_divide_by = 10,

94 altpll_component.clk0_duty_cycle = 50,

95 altpll_component.clk0_multiply_by = 1,

96 altpll_component.clk0_phase_shift = "0",

97 altpll_component.compensate_clock = "CLK0",

98 altpll_component.inclk0_input_frequency = 20000,

99 altpll_component.intended_device_family = "Cyclone III",

100 altpll_component.lpm_hint = "CBX_MODULE_PREFIX=PLL1",

101 altpll_component.lpm_type = "altpll",

102 altpll_component.operation_mode = "NORMAL",

103 altpll_component.pll_type = "AUTO",

104 altpll_component.port_activeclock = "PORT_UNUSED",

105 altpll_component.port_areset = "PORT_UNUSED",

106 altpll_component.port_clkbad0 = "PORT_UNUSED",

107 altpll_component.port_clkbad1 = "PORT_UNUSED",

108 altpll_component.port_clkloss = "PORT_UNUSED",

109 altpll_component.port_clkswitch = "PORT_UNUSED",

110 altpll_component.port_configupdate = "PORT_UNUSED",

111 altpll_component.port_fbin = "PORT_UNUSED",

112 altpll_component.port_inclk0 = "PORT_USED",

113 altpll_component.port_inclk1 = "PORT_UNUSED",

114 altpll_component.port_locked = "PORT_UNUSED",

115 altpll_component.port_pfdena = "PORT_UNUSED",

116 altpll_component.port_phasecounterselect = "PORT_UNUSED",

117 altpll_component.port_phasedone = "PORT_UNUSED",

118 altpll_component.port_phasestep = "PORT_UNUSED",

119 altpll_component.port_phaseupdown = "PORT_UNUSED",

120 altpll_component.port_pllena = "PORT_UNUSED",

121 altpll_component.port_scanaclr = "PORT_UNUSED",

122 altpll_component.port_scanclk = "PORT_UNUSED",

123 altpll_component.port_scanclkena = "PORT_UNUSED",

124 altpll_component.port_scandata = "PORT_UNUSED",

125 altpll_component.port_scandataout = "PORT_UNUSED",


126 altpll_component.port_scandone = "PORT_UNUSED",

127 altpll_component.port_scanread = "PORT_UNUSED",

128 altpll_component.port_scanwrite = "PORT_UNUSED",

129 altpll_component.port_clk0 = "PORT_USED",

130 altpll_component.port_clk1 = "PORT_UNUSED",





135 altpll_component.port_clkena0 = "PORT_UNUSED",






141 altpll_component.port_extclk0 = "PORT_UNUSED",




145 altpll_component.width_clock = 5;

146147148 endmodule

149150 // ============================================================


152 // ============================================================

153 // Retrieval info: PRIVATE: ACTIVECLK_CHECK STRING "0"

154 // Retrieval info: PRIVATE: BANDWIDTH STRING "1.000"

155 // Retrieval info: PRIVATE: BANDWIDTH_FEATURE_ENABLED STRING "1"

156 // Retrieval info: PRIVATE: BANDWIDTH_FREQ_UNIT STRING "MHz"

157 // Retrieval info: PRIVATE: BANDWIDTH_PRESET STRING "Low"

158 // Retrieval info: PRIVATE: BANDWIDTH_USE_AUTO STRING "1"

159 // Retrieval info: PRIVATE: BANDWIDTH_USE_PRESET STRING "0"

160 // Retrieval info: PRIVATE: CLKBAD_SWITCHOVER_CHECK STRING "0"

161 // Retrieval info: PRIVATE: CLKLOSS_CHECK STRING "0"

162 // Retrieval info: PRIVATE: CLKSWITCH_CHECK STRING "0"

163 // Retrieval info: PRIVATE: CNX_NO_COMPENSATE_RADIO STRING "0"

164 // Retrieval info: PRIVATE: CREATE_CLKBAD_CHECK STRING "0"

165 // Retrieval info: PRIVATE: CREATE_INCLK1_CHECK STRING "0"

166 // Retrieval info: PRIVATE: CUR_DEDICATED_CLK STRING "c0"

167 // Retrieval info: PRIVATE: CUR_FBIN_CLK STRING "c0"

168 // Retrieval info: PRIVATE: DEVICE_SPEED_GRADE STRING "Any"

169 // Retrieval info: PRIVATE: DIV_FACTOR0 NUMERIC "10"

170 // Retrieval info: PRIVATE: DUTY_CYCLE0 STRING "50.00000000"

171 // Retrieval info: PRIVATE: EFF_OUTPUT_FREQ_VALUE0 STRING "5.000000"

172 // Retrieval info: PRIVATE: EXPLICIT_SWITCHOVER_COUNTER STRING "0"

173 // Retrieval info: PRIVATE: EXT_FEEDBACK_RADIO STRING "0"

174 // Retrieval info: PRIVATE: GLOCKED_COUNTER_EDIT_CHANGED STRING "1"

175 // Retrieval info: PRIVATE: GLOCKED_FEATURE_ENABLED STRING "0"

176 // Retrieval info: PRIVATE: GLOCKED_MODE_CHECK STRING "0"

177 // Retrieval info: PRIVATE: GLOCK_COUNTER_EDIT NUMERIC "1048575"

178 // Retrieval info: PRIVATE: HAS_MANUAL_SWITCHOVER STRING "1"

179 // Retrieval info: PRIVATE: INCLK0_FREQ_EDIT STRING "50.000"


180 // Retrieval info: PRIVATE: INCLK0_FREQ_UNIT_COMBO STRING "MHz"

181 // Retrieval info: PRIVATE: INCLK1_FREQ_EDIT STRING "100.000"

182 // Retrieval info: PRIVATE: INCLK1_FREQ_EDIT_CHANGED STRING "1"

183 // Retrieval info: PRIVATE: INCLK1_FREQ_UNIT_CHANGED STRING "1"

184 // Retrieval info: PRIVATE: INCLK1_FREQ_UNIT_COMBO STRING "MHz"


186 // Retrieval info: PRIVATE: INT_FEEDBACK__MODE_RADIO STRING "1"

187 // Retrieval info: PRIVATE: LOCKED_OUTPUT_CHECK STRING "0"

188 // Retrieval info: PRIVATE: LONG_SCAN_RADIO STRING "1"

189 // Retrieval info: PRIVATE: LVDS_MODE_DATA_RATE STRING "Not Available"

190 // Retrieval info: PRIVATE: LVDS_MODE_DATA_RATE_DIRTY NUMERIC "0"

191 // Retrieval info: PRIVATE: LVDS_PHASE_SHIFT_UNIT0 STRING "deg"

192 // Retrieval info: PRIVATE: MIG_DEVICE_SPEED_GRADE STRING "Any"

193 // Retrieval info: PRIVATE: MIRROR_CLK0 STRING "0"

194 // Retrieval info: PRIVATE: MULT_FACTOR0 NUMERIC "1"

195 // Retrieval info: PRIVATE: NORMAL_MODE_RADIO STRING "1"

196 // Retrieval info: PRIVATE: OUTPUT_FREQ0 STRING "100.00000000"

197 // Retrieval info: PRIVATE: OUTPUT_FREQ_MODE0 STRING "0"

198 // Retrieval info: PRIVATE: OUTPUT_FREQ_UNIT0 STRING "MHz"

199 // Retrieval info: PRIVATE: PHASE_RECONFIG_FEATURE_ENABLED STRING "1"

200 // Retrieval info: PRIVATE: PHASE_RECONFIG_INPUTS_CHECK STRING "0"

201 // Retrieval info: PRIVATE: PHASE_SHIFT0 STRING "0.00000000"

202 // Retrieval info: PRIVATE: PHASE_SHIFT_STEP_ENABLED_CHECK STRING "0"

203 // Retrieval info: PRIVATE: PHASE_SHIFT_UNIT0 STRING "deg"

204 // Retrieval info: PRIVATE: PLL_ADVANCED_PARAM_CHECK STRING "0"

205 // Retrieval info: PRIVATE: PLL_ARESET_CHECK STRING "0"

206 // Retrieval info: PRIVATE: PLL_AUTOPLL_CHECK NUMERIC "1"

207 // Retrieval info: PRIVATE: PLL_ENHPLL_CHECK NUMERIC "0"

208 // Retrieval info: PRIVATE: PLL_FASTPLL_CHECK NUMERIC "0"

209 // Retrieval info: PRIVATE: PLL_FBMIMIC_CHECK STRING "0"

210 // Retrieval info: PRIVATE: PLL_LVDS_PLL_CHECK NUMERIC "0"

211 // Retrieval info: PRIVATE: PLL_PFDENA_CHECK STRING "0"

212 // Retrieval info: PRIVATE: PLL_TARGET_HARCOPY_CHECK NUMERIC "0"

213 // Retrieval info: PRIVATE: PRIMARY_CLK_COMBO STRING "inclk0"

214 // Retrieval info: PRIVATE: RECONFIG_FILE STRING "PLL1.mif"

215 // Retrieval info: PRIVATE: SACN_INPUTS_CHECK STRING "0"

216 // Retrieval info: PRIVATE: SCAN_FEATURE_ENABLED STRING "1"

217 // Retrieval info: PRIVATE: SELF_RESET_LOCK_LOSS STRING "0"

218 // Retrieval info: PRIVATE: SHORT_SCAN_RADIO STRING "0"

219 // Retrieval info: PRIVATE: SPREAD_FEATURE_ENABLED STRING "0"

220 // Retrieval info: PRIVATE: SPREAD_FREQ STRING "50.000"

221 // Retrieval info: PRIVATE: SPREAD_FREQ_UNIT STRING "KHz"

222 // Retrieval info: PRIVATE: SPREAD_PERCENT STRING "0.500"

223 // Retrieval info: PRIVATE: SPREAD_USE STRING "0"

224 // Retrieval info: PRIVATE: SRC_SYNCH_COMP_RADIO STRING "0"

225 // Retrieval info: PRIVATE: STICKY_CLK0 STRING "1"

226 // Retrieval info: PRIVATE: SWITCHOVER_COUNT_EDIT NUMERIC "1"

227 // Retrieval info: PRIVATE: SWITCHOVER_FEATURE_ENABLED STRING "1"


229 // Retrieval info: PRIVATE: USE_CLK0 STRING "1"

230 // Retrieval info: PRIVATE: USE_CLKENA0 STRING "0"

231 // Retrieval info: PRIVATE: USE_MIL_SPEED_GRADE NUMERIC "0"

232 // Retrieval info: PRIVATE: ZERO_DELAY_RADIO STRING "0"



234 // Retrieval info: CONSTANT: BANDWIDTH_TYPE STRING "AUTO"

235 // Retrieval info: CONSTANT: CLK0_DIVIDE_BY NUMERIC "10"

236 // Retrieval info: CONSTANT: CLK0_DUTY_CYCLE NUMERIC "50"

237 // Retrieval info: CONSTANT: CLK0_MULTIPLY_BY NUMERIC "1"

238 // Retrieval info: CONSTANT: CLK0_PHASE_SHIFT STRING "0"

239 // Retrieval info: CONSTANT: COMPENSATE_CLOCK STRING "CLK0"

240 // Retrieval info: CONSTANT: INCLK0_INPUT_FREQUENCY NUMERIC "20000"


242 // Retrieval info: CONSTANT: LPM_TYPE STRING "altpll"

243 // Retrieval info: CONSTANT: OPERATION_MODE STRING "NORMAL"

244 // Retrieval info: CONSTANT: PLL_TYPE STRING "AUTO"

245 // Retrieval info: CONSTANT: PORT_ACTIVECLOCK STRING "PORT_UNUSED"

246 // Retrieval info: CONSTANT: PORT_ARESET STRING "PORT_UNUSED"

247 // Retrieval info: CONSTANT: PORT_CLKBAD0 STRING "PORT_UNUSED"

248 // Retrieval info: CONSTANT: PORT_CLKBAD1 STRING "PORT_UNUSED"

249 // Retrieval info: CONSTANT: PORT_CLKLOSS STRING "PORT_UNUSED"

250 // Retrieval info: CONSTANT: PORT_CLKSWITCH STRING "PORT_UNUSED"

251 // Retrieval info: CONSTANT: PORT_CONFIGUPDATE STRING "PORT_UNUSED"

252 // Retrieval info: CONSTANT: PORT_FBIN STRING "PORT_UNUSED"

253 // Retrieval info: CONSTANT: PORT_INCLK0 STRING "PORT_USED"

254 // Retrieval info: CONSTANT: PORT_INCLK1 STRING "PORT_UNUSED"

255 // Retrieval info: CONSTANT: PORT_LOCKED STRING "PORT_UNUSED"

256 // Retrieval info: CONSTANT: PORT_PFDENA STRING "PORT_UNUSED"

257 // Retrieval info: CONSTANT: PORT_PHASECOUNTERSELECT STRING "PORT_UNUSED"

258 // Retrieval info: CONSTANT: PORT_PHASEDONE STRING "PORT_UNUSED"

259 // Retrieval info: CONSTANT: PORT_PHASESTEP STRING "PORT_UNUSED"

260 // Retrieval info: CONSTANT: PORT_PHASEUPDOWN STRING "PORT_UNUSED"

261 // Retrieval info: CONSTANT: PORT_PLLENA STRING "PORT_UNUSED"

262 // Retrieval info: CONSTANT: PORT_SCANACLR STRING "PORT_UNUSED"

263 // Retrieval info: CONSTANT: PORT_SCANCLK STRING "PORT_UNUSED"

264 // Retrieval info: CONSTANT: PORT_SCANCLKENA STRING "PORT_UNUSED"

265 // Retrieval info: CONSTANT: PORT_SCANDATA STRING "PORT_UNUSED"

266 // Retrieval info: CONSTANT: PORT_SCANDATAOUT STRING "PORT_UNUSED"

267 // Retrieval info: CONSTANT: PORT_SCANDONE STRING "PORT_UNUSED"

268 // Retrieval info: CONSTANT: PORT_SCANREAD STRING "PORT_UNUSED"

269 // Retrieval info: CONSTANT: PORT_SCANWRITE STRING "PORT_UNUSED"

270 // Retrieval info: CONSTANT: PORT_clk0 STRING "PORT_USED"

271 // Retrieval info: CONSTANT: PORT_clk1 STRING "PORT_UNUSED"





276 // Retrieval info: CONSTANT: PORT_clkena0 STRING "PORT_UNUSED"






282 // Retrieval info: CONSTANT: PORT_extclk0 STRING "PORT_UNUSED"




286 // Retrieval info: CONSTANT: WIDTH_CLOCK NUMERIC "5"

287 // Retrieval info: USED_PORT: @clk 0 0 5 0 OUTPUT_CLK_EXT VCC "@clk[4..0]"


288 // Retrieval info: USED_PORT: c0 0 0 0 0 OUTPUT_CLK_EXT VCC "c0"

289 // Retrieval info: USED_PORT: inclk0 0 0 0 0 INPUT_CLK_EXT GND "inclk0"

290 // Retrieval info: CONNECT: @inclk 0 0 1 1 GND 0 0 0 0

291 // Retrieval info: CONNECT: @inclk 0 0 1 0 inclk0 0 0 0 0

292 // Retrieval info: CONNECT: c0 0 0 0 0 @clk 0 0 1 0

293 // Retrieval info: GEN_FILE: TYPE_NORMAL PLL1.v TRUE

294 // Retrieval info: GEN_FILE: TYPE_NORMAL PLL1.ppf TRUE

295 // Retrieval info: GEN_FILE: TYPE_NORMAL PLL1.inc FALSE

296 // Retrieval info: GEN_FILE: TYPE_NORMAL PLL1.cmp FALSE

297 // Retrieval info: GEN_FILE: TYPE_NORMAL PLL1.bsf TRUE FALSE

298 // Retrieval info: GEN_FILE: TYPE_NORMAL PLL1_inst.v FALSE

299 // Retrieval info: GEN_FILE: TYPE_NORMAL PLL1_bb.v TRUE


301 // Retrieval info: CBX_MODULE_PREFIX: ON

A.4.9 Waveform Generator


2 ////////////////////////////////////////////////////////////////////////////////



5 //

6 // Create Date: 2:44pm 12/08/2010

7 // Module Name: Reference_generator


9 //

10 ////////////////////////////////////////////////////////////////////////////////

11 module Reference_generator(clkIn,counter,CS,WR,LDAC,PD,GAIN,CLR,stop);

1213 input clkIn;

1415 output [9:0] counter;

16 output CS,WR,LDAC,PD,GAIN,CLR,stop;

1718 reg [9:0] counter = 0;

19 reg [9:0] counter2 = 0;

20 reg upward = 1;

21 reg [20:0] clk_count = 0;

22 reg [29:0] count = 0;

23 reg [29:0] count2 = 0;

24 reg [9:0] up = 430;

25 reg [9:0] pause_time = 300;

26 reg [9:0] base = 231;

27 reg [29:0] pause = 0;

28 reg stop = 0;

29 reg clkout;

3031 reg clkout2;

323334 //----------------------------------------------------------------------------

3536 assign CS = 0;


37 assign WR = clkout;

38 assign PD = 1;

39 assign GAIN = 0;

40 assign CLR = 1;

41 assign LDAC = 0;

424344 //----------------------------------------------------------------------------


47 begin

48 if (count < 250) //200

49 begin

50 clkout <= 1;


52 end

53 else if (count < 499) //399

54 begin

55 clkout <= 0;


57 end

58 if (count == 499) //399

59 count <= 0;

60 end

6162 //----------------------------------------------------------------------------


65 begin

66 if (count2 < 100000) //200

67 begin

68 clkout2 <= 1;

69 count2 <= count2+1;

70 end

71 else if (count2 < 199999) //399

72 begin

73 clkout2 <= 0;

74 count2 <= count2+1;

75 end

76 if (count2 == 199999) //399

77 count2 <= 0;

78 end

798081 always @ (negedge clkout2)

82 begin

8384 // clk_count <= clk_count + 1;

85 counter <= base;

86 counter2 <= base; //276

87 if ( (counter2 < up+1) & (upward == 1) ) //563

88 begin

89 counter <= counter - 1;

90 counter2 <= counter2 + 1;


91 stop <=0;

92 end

93 if ( (counter2 == up) & (upward == 1) ) //562

94 begin

95 upward <= 0;

96 stop <=0;

97 end

98 if ( (counter2 < up+2) & (upward == 0) ) //564

99 begin

100 counter <= counter + 1;

101 counter2 <= counter2 - 1;

102 stop <=0;

103 end

104 if ( (counter2 == base+1) & (upward == 0) ) //277

105 if (pause < pause_time) //200

106 begin

107 pause <= pause+1;

108 counter <= base+1;

109 counter2 <= base+1;

110 stop <=0;

111 end

112 else //399

113 begin

114 pause <= 0;

115 // clk_count <= 0;

116 upward <= 1;

117 counter <= base;

118 counter2 <= base;

119 stop <=0;

120 end

121 end

122123 endmodule

A.4.10 Master Reset


2 ////////////////////////////////////////////////////////////////////////////////



5 //

6 // Create Date: 2:44pm 12/08/2010

7 // Module Name: CH_TIMING


9 //

10 ////////////////////////////////////////////////////////////////////////////////

11 module RESET_CONT(clkIn, IDEL1, IDEL2, IDEL3, IDEL4, RESET_OUT);

12 ///////////////////////////////////////////////////////////////////////////////

13 input clkIn, IDEL1, IDEL2, IDEL3, IDEL4;

14 output RESET_OUT;

15 ///////////////////////////////////////////////////////////////////////////////

16 reg clk;

17 reg RESET_OUT = 0;


18 reg [29:0] count = 0;

19 ///////////////////////////////////////////////////////////////////////////////


21 begin

22 if (count < 4)

23 begin

24 clk <= 1;


26 end


28 begin

29 clk <= 0;


31 end

32 if (count == 7)

33 count <= 0;

34 end

35 ///////////////////////////////////////////////////////////////////////////////


37 begin

3839 if ( !IDEL1 && !IDEL2 && !IDEL3 && !IDEL4)

40 RESET_OUT <= 1;

41 else

42 RESET_OUT <= 0;

43 end

44 endmodule

A.4.11 Seven Bit Counter


2 ////////////////////////////////////////////////////////////////////////////////



5 //

6 // Create Date: 2:44pm 12/08/2010

7 // Module Name: TX


9 //

10 ////////////////////////////////////////////////////////////////////////////////

11 module Seven_bit_counter(clkIn,

12 Timer, start

13 );

14 ///////////////////////////////////////////////////////////////////////////////

15 input clkIn;

16 output [16:0] Timer;

17 output start;

18 ///////////////////////////////////////////////////////////////////////////////

19 reg [16:0] Timer =0;

20 reg clk;

21 reg start;

22 ///////////////////////////////////////////////////////////////////////////////

23 reg [29:0] count = 0;


24 ///////////////////////////////////////////////////////////////////////////////


26 begin

27 if (count < 8)

28 begin

29 clk <= 1;


31 end


33 begin

34 clk <= 0;


36 end

37 if (count == 15)

38 count <= 0;

39 end

40 ////////////////////////////////////////////////////////////////////////////////

41 always @ (negedge clk)

42 begin

43 start <= 1;

44 Timer <= Timer +1;

45 end

46 endmodule

A.4.12 Chip Timing Coefficients


2 ////////////////////////////////////////////////////////////////////////////////



5 //

6 // Create Date: 2:44pm 12/08/2010

7 // Module Name: Shift_Reg_TEMP


9 //

10 ////////////////////////////////////////////////////////////////////////////////

11 module Shift_Reg_COEFF(clkIn, RESET_SHIFT, CLK_SH3, I_COEFF_PAD, IDEL_COEFF);

12 ///////////////////////////////////////////////////////////////////////////////

13 input clkIn;

14 input RESET_SHIFT;

15 output CLK_SH3, IDEL_COEFF;

16 output wire I_COEFF_PAD;

17 ///////////////////////////////////////////////////////////////////////////////

18 reg CLK_SH3;

19 reg IDEL_COEFF =1;

20 reg [4:0] timer = 5’b00000;

21 reg [29:0] count = 0;

22 reg [29:0] DATA = 30’b000001000000000000011111111111; //30’b000001000000000001111111111111; ’

b000001000000011111111111111011; 30’b00000100000000001100000000000;30’b000001000000000000111111111111;30’

b000001000000000000011111111111;

23 reg Dataout;

24 ///////////////////////////////////////////////////////////////////////////////



26 begin

27 if (count < 10000)

28 begin

29 CLK_SH3 <= 1;


31 end


33 begin

34 CLK_SH3 <= 0;


36 end

37 if (count == 19999)

38 count <= 0;

39 end

40 ///////////////////////////////////////////////////////////////////////////////

4142 always @ (negedge CLK_SH3)

43 begin

44 case (timer)

45 0:

46 begin

47 IDEL_COEFF = 1;

48 Dataout <= DATA[29];

49 end

50 endcase

5152 if (timer < 30 && RESET_SHIFT)

53 begin


55 DATA <= DATA[28:0] , DATA[29];


57 end

58 else if (timer == 30)

59 begin

60 IDEL_COEFF = 0;

61 Dataout <= 0;

62 end

63 end

64 assign I_COEFF_PAD = Dataout;

65 endmodule

A.4.13 SRAM Timing Coefficients


2 ////////////////////////////////////////////////////////////////////////////////



5 //

6 // Create Date: 2:44pm 12/08/2010



9 //

10 ////////////////////////////////////////////////////////////////////////////////


11 module Shift_Reg_COEFF_SRAM(clkIn, RESET_SHIFT, CLK_SH3, I_COEFF_PAD, IDEL_COEFF, clk_SRAM);

12 ///////////////////////////////////////////////////////////////////////////////

13 input clkIn;


15 input clk_SRAM;

16 output CLK_SH3, IDEL_COEFF;

17 output wire I_COEFF_PAD;

18 ///////////////////////////////////////////////////////////////////////////////

19 reg CLK_SH3;

20 reg IDEL_COEFF =1;

21 reg [4:0] timer = 5’b00000;

22 reg [4:0] SRAM = 5’b00000;

23 reg [4:0] SRAMIDEL = 5’b00000;

24 reg timerX = 0;

2526 reg [29:0] count = 0;

2728 reg [29:0] DATA1 = 30’b000001000000000000000000011111; //000001000000000000000000011111

000001000000001111111110011111

29 reg [29:0] DATA2 = 30’b000001000000000000000000011111;

30 reg [29:0] DATA3 = 30’b000001000000000000000000011111;

31 reg [29:0] DATA4 = 30’b000001000000000000000000011111;

32 reg [29:0] DATA5 = 30’b000001000000000000000000011111;

33 reg [29:0] DATA6 = 30’b000001000000000000000000011111;

34 reg [29:0] DATA7 = 30’b000001000000000000000000011111;

35 reg [29:0] DATA8 = 30’b000001000000000000000000011111;

36 reg [29:0] DATA9 = 30’b000001000000000000000000011111;

37 reg [29:0] DATA10 = 30’b000001000000000000000000011111;

38 reg [29:0] DATA11 = 30’b000001000000000000000000011111;

39 reg [29:0] DATA12 = 30’b000001000000000000000000011111;

40 reg [29:0] DATA13 = 30’b000001000000000000000000011111;

41 reg [29:0] DATA14 = 30’b000001000000000000000000011111;

42 reg [29:0] DATA15 = 30’b000001000000000000000000011111;

43 reg [29:0] DATA16 = 30’b000001000000000000000000011111;

4445 reg Dataout;

46 ///////////////////////////////////////////////////////////////////////////////


48 begin

49 if (count < 50)

50 begin

51 CLK_SH3 <= 1;


53 end


55 begin

56 CLK_SH3 <= 0;


58 end

59 if (count == 99)

60 count <= 0;

61 end

62 ///////////////////////////////////////////////////////////////////////////////

63 always @ (negedge clk_SRAM)


64 begin

65 SRAMIDEL = SRAMIDEL +1;

66 if (RESET_SHIFT && SRAMIDEL > 4)

67 begin

68 timerX <= 1;

69 SRAM <= SRAM +1;

70 end

71 if (SRAM == 16)

72 begin

73 SRAM <= 5’b00000;

74 timerX <= 0;

75 end

76 end

77 ///////////////////////////////////////////////////////////////////////////////


79 begin

8081 if (timer == 0)

82 begin

83 IDEL_COEFF = 1;

8485 if (SRAM == 0)

86 Dataout <= DATA1[29];

87 else if (SRAM == 1)






























117


118 end


122 begin


124125 if (SRAM == 0)

126 begin

127 DATA1 <= DATA1[28:0] , DATA1[29];


129 end

130 if (SRAM == 1)

131 begin

132 DATA2 <= DATA2[28:0] , DATA2[29];


134 end

135 if (SRAM == 2)

136 begin

137 DATA3 <= DATA3[28:0] , DATA3[29];


139 end

140 if (SRAM == 3)

141 begin

142 DATA4 <= DATA4[28:0] , DATA4[29];


144 end

145 if (SRAM == 4)

146 begin

147 DATA5 <= DATA5[28:0] , DATA5[29];


149 end

150 if (SRAM == 5)

151 begin

152 DATA6 <= DATA6[28:0] , DATA6[29];


154 end

155 if (SRAM == 6)

156 begin

157 DATA7 <= DATA7[28:0] , DATA7[29];


159 end

160 if (SRAM == 7)

161 begin

162 DATA8 <= DATA8[28:0] , DATA8[29];


164 end

165 if (SRAM == 8)

166 begin

167 DATA9 <= DATA9[28:0] , DATA9[29];


169 end

170 if (SRAM == 9)

171 begin


172 DATA10 <= DATA10[28:0] , DATA10[29];


174 end

175 if (SRAM == 10)

176 begin

177 DATA11 <= DATA11[28:0] , DATA11[29];


179 end

180 if (SRAM == 11)

181 begin

182 DATA12 <= DATA12[28:0] , DATA12[29];


184 end

185 if (SRAM == 12)

186 begin

187 DATA13 <= DATA13[28:0] , DATA13[29];


189 end

190 if (SRAM == 13)

191 begin

192 DATA14 <= DATA14[28:0] , DATA14[29];


194 end

195 if (SRAM == 14)

196 begin

197 DATA15 <= DATA15[28:0] , DATA15[29];


199 end

200 if (SRAM == 15)

201 begin

202 DATA16 <= DATA16[28:0] , DATA16[29];


204 end

205 end


207 begin

208 IDEL_COEFF = 0;

209 Dataout <= 0;

210 end

211 if (timerX == 1)

212 timer <= 0;

213 end

214 assign I_COEFF_PAD = Dataout;

215216 endmodule

A.4.14 Channel Range Controller Coefficients


2 ////////////////////////////////////////////////////////////////////////////////



5 //


6 // Create Date: 2:44pm 12/08/2010

7 // Module Name: Shift_Reg_POT


9 //

10 ////////////////////////////////////////////////////////////////////////////////

11 module Shift_Reg_POT(clkIn, RESET_SHIFT, clk_POT, I_STAT_PAD_POT, idel_POT);

12 ///////////////////////////////////////////////////////////////////////////////

13 input clkIn;


15 output clk_POT, idel_POT;

16 output wire I_STAT_PAD_POT;

1718 ///////////////////////////////////////////////////////////////////////////////

19 reg clk_POT;

20 reg idel_POT =1;

21 reg [4:0] timer = 5’b00000;

22 reg [29:0] count = 0;

23 reg [17:0] DATA = 18’b000001101000010011; //1101000000011 1100000001011 b000001101000010011;

24 reg Dataout;

25 ///////////////////////////////////////////////////////////////////////////////


27 begin

28 if (count < 10000)

29 begin

30 clk_POT <= 1;


32 end


34 begin

35 clk_POT <= 0;


37 end

38 if (count == 19999)

39 count <= 0;

40 end

41 ///////////////////////////////////////////////////////////////////////////////

4243 always @ (negedge clk_POT)

44 begin

45 case (timer)

46 0:

47 begin

48 idel_POT = 1;


50 end

51 endcase


54 begin


56 DATA <= DATA[16:0] , DATA[17];


58 end



60 begin

61 idel_POT = 0;

62 Dataout <= 0;

63 end

64 end

65 assign I_STAT_PAD_POT = Dataout;

66 endmodule

A.4.15 Impedance Spectroscopy SRAM Coefficients


2 ////////////////////////////////////////////////////////////////////////////////



5 //

6 // Create Date: 2:44pm 12/08/2010



9 //

10 ////////////////////////////////////////////////////////////////////////////////

11 module SHIFT_REG_RESET(clkIn, RESET_SHIFT);

12 ///////////////////////////////////////////////////////////////////////////////

13 input clkIn;

14 output RESET_SHIFT;

15 ///////////////////////////////////////////////////////////////////////////////

16 reg RESET_SHIFT = 0;

17 reg clk;

18 reg [9:0] timer = 10’b0000000000;

19 ///////////////////////////////////////////////////////////////////////////////

20 reg [29:0] count = 0;

21 ///////////////////////////////////////////////////////////////////////////////


23 begin

24 if (count < 10000)

25 begin

26 clk <= 1;


28 end


30 begin

31 clk <= 0;


33 end

34 if (count == 19999)

35 count <= 0;

36 end

37 ///////////////////////////////////////////////////////////////////////////////

38 always @ (negedge clk)

39 begin

40 if (timer < 500)

41 begin


43 RESET_SHIFT <= 0;


44 end

45 else

46 RESET_SHIFT <= 1;

47 end

48 endmodule

49 ///////////////////////////////////////////////////////////////////////////////

A.4.16 Temperature Regulation SRAM Coefficients


2 ////////////////////////////////////////////////////////////////////////////////



5 //

6 // Create Date: 2:44pm 12/08/2010



9 //

10 ////////////////////////////////////////////////////////////////////////////////

11 module Shift_Reg_TEMP(clkIn, RESET_SHIFT, CLK_SH2, I_STAT_PAD, IDEL_TMEP);

12 ///////////////////////////////////////////////////////////////////////////////

13 input clkIn;


15 output CLK_SH2, IDEL_TMEP;


17 ///////////////////////////////////////////////////////////////////////////////

18 reg CLK_SH2;

19 reg IDEL_TEMP =1;

20 reg [4:0] timer = 5’b00000;

21 reg [29:0] count = 0;

22 reg [15:0] DATA = 16’b0000000010000000;

23 reg Dataout;

24 ///////////////////////////////////////////////////////////////////////////////


26 begin

27 if (count < 10000)

28 begin

29 CLK_SH2 <= 1;


31 end


33 begin

34 CLK_SH2 <= 0;


36 end

37 if (count == 19999)

38 count <= 0;

39 end

40 ///////////////////////////////////////////////////////////////////////////////


43 begin

44 case (timer)


45 0:

46 begin

47 IDEL_TEMP = 1;


49 end

50 endcase


53 begin


55 DATA <= DATA[14:0] , DATA[15];


57 end


59 begin

60 IDEL_TEMP = 0;

61 Dataout <= 0;

62 end

63 end

64 assign I_STAT_PAD = Dataout;

65 endmodule

A.4.17 Test Channel Controller

1 ////////////////////////////////////////////////////////////////////////////////



4 //

5 // Create Date: 2:44pm 12/08/2010

6 // Module Name: Channel_control test bench


8 //

9 ////////////////////////////////////////////////////////////////////////////////

1011 module TEST_Channel_Control;

1213 reg clkIn, RESET_IN ;

1415 wire RESET_CH, CLK_CH;

1617 Channel_Control dut (clkIn, RESET_CH, RESET_IN, CLK_CH);

1819 initial // Clock generator

20 begin

21 clkIn=0;

22 RESET_IN = 1;

23 forever #10 clkIn = !clkIn;

24 end

2526 initial

27 $monitor($stime,, clkIn,, RESET_CH,, RESET_IN,, CLK_CH);

28 endmodule


A.5 Example CV Recording Verilog Code

The following code generates a 0 to 700mV CV waveform with scan rate of 100mV/s

and simultaneously records the resulting current on 54 channels.1 // Copyright (C) 1991-2011 Altera Corporation















17 // CREATED "Sun Jul 21 20:24:15 2013"


20 OSC_CLK,

21 TXE_USB,

22 RXF_USB_IN,

23 DATA_IN,

24 DIN_USB,

25 CS,

26 WR,

27 PD,

28 GAIN,

29 CLR,

30 ldac1,

31 sync1,

32 ldac2,

33 sync2,

34 din_out2,

35 din_out1,

36 SING,

37 C1,

38 VLATCH,

39 P1,

40 EXT_RESET,

41 BS_A,

42 BS_P,

43 Q,

44 LDAC,

45 DAC_CLK1,

46 DAC_CLK2,

47 CLK_SH1,


48 I_STAT_PAD,

49 CLK_SH2_TEMP,

50 IN_E_PAD_TEMP,

51 CLK_SHIFT_PAD,

52 I_SHIFT_PAD,

53 RESET_CH,

54 CLK_CH,

55 CLK_CV,

56 UP_DOWN_CV,

57 IMP_CV,


59 W,

60 CLK_COEF,

61 OFF_ON_CHIP,

62 COS,

63 OFF_CHIP_DAC_P,

64 ELECTRODE_OFFON,

65 READ_PWM,

66 CLK_SHFIT_RING,

67 CLK_RING,

68 READ_ON_OFF_PWM,

69 RESET_RING,

70 WRITE_PAD,

71 EXT_RING,

72 ON_OFF_RING,

73 B_PID,

74 A_PID,

75 ADC_PID,

76 ADC_MEM_READ,

77 MAINE_READ,

78 RESET_,

79 WP0,

80 WP1,

81 WP2,

82 WP3,

83 RESET_P,

84 CLK_P,

85 PAR_P,

86 RESET_PS,

87 CLK_PS,

88 LOAD_SHIFT,

89 RESET_SHIFT,

90 IN_COEF_PAD,

91 CLK_SH3_PAD,


93 CLK_TEMP,

94 CHAMBER_RESET,

95 CHAMBER_CLK,

96 WR_USB,

97 RD_USB,

98 SIWUA,

99 SIWUB,

100 B,

101 DATA_OUT,


102 DATA_OUT_USB

103 );






111 output wire CS;

112 output wire WR;

113 output wire PD;










123 output wire C1;


125 output wire P1;




129 output wire Q;
















145 output wire W;














































191 wire [17:0] DIN;

192 wire idel;

193 wire idel2;

194 wire idel3;

195 wire idel4;

196 wire start_DAC;


























226 .CS(CS),

227 .WR(WR),

228 .LDAC(LDAC),

229 .PD(PD),

230 .GAIN(GAIN),

231 .CLR(CLR),


233 .counter(B));




239 .SING(SING),

240 .C1(C1),


242 .P1(P1),


244 .BS_A(BS_A),

245 .BS_P(BS_P),

246 .Q(Q));

247248249 Inv b2v_inst10(

250 .a(idel2),






258259260 Inv b2v_inst12(

261 .a(idel3),


263


























295 .B_PID(B_PID),

296 .A_PID(A_PID),





301 .WP0(WP0),

302 .WP1(WP1),

303 .WP2(WP2),

304 .WP3(WP3),


306 .CLK_P(CLK_P),

307 .PAR_P(PAR_P));

308309310 TX b2v_inst18(














327 .DIN(DIN));

328329330 Inv b2v_inst20(

331 .a(idel4),





338 .TXE(TXE_USB),



341 .WR(WR_USB),

342 .SIWUA(SIWUA),


344345346 IO_B b2v_inst22(








356 .DIN(DIN_USB),


358 .RD(RD_USB),

359 .SIWUB(SIWUB),


361 );



366 );





372373374 Inv16 b2v_inst26(

375 .DATA_IN(SYNTHESIZED_WIRE_28),

376 .DATA_OUT(DATA_OUT));













393394395 SRAM b2v_inst30(






401 .W(W),



404 .COS(COS),














418 .B_PID(B_PID),

419 .A_PID(A_PID),





424 .WP0(WP0),

425 .WP1(WP1),


426 .WP2(WP2),

427 .WP3(WP3),


429 .CLK_P(CLK_P),

430 .PAR_P(PAR_P));

431432433 TX b2v_inst18(













450 .DIN(DIN));

451452453 Inv b2v_inst20(

454 .a(idel4),





461 .TXE(TXE_USB),



464 .WR(WR_USB),

465 .SIWUA(SIWUA),


467468469 IO_B b2v_inst22(








479 .DIN(DIN_USB),



481 .RD(RD_USB),

482 .SIWUB(SIWUB),


484 );




489 .Data(DIN),



492 .ldac1(ldac1),

493 .sync1(sync1),


495 .ldac2(ldac2),

496 .sync2(sync2));

497498 endmodule

A.6 Example IS Recording Verilog Code

The following code generates a 50mV peak-to-peak sinusoid waveform from 0.1Hz to

1kHz and simultaneously extract the on-chip electrode impedance from 54 channels.1 // Copyright (C) 1991-2011 Altera Corporation















17 // CREATED "Sun Jul 21 20:31:33 2013"


20 OSC_CLK,

21 TXE_USB,

22 RXF_USB_IN,

23 DATA_IN,

24 DIN_USB,


25 CS,

26 WR,

27 PD,

28 GAIN,

29 CLR,

30 ldac1,

31 sync1,

32 ldac2,

33 sync2,

34 din_out2,

35 din_out1,

36 SING,

37 C1,

38 VLATCH,

39 P1,

40 EXT_RESET,

41 BS_A,

42 BS_P,

43 Q,

44 LDAC,

45 DAC_CLK1,

46 DAC_CLK2,

47 CLK_SH1,

48 I_STAT_PAD,

49 CLK_SH2_TEMP,

50 IN_E_PAD_TEMP,

51 CLK_SHIFT_PAD,

52 I_SHIFT_PAD,

53 RESET_CH,

54 CLK_CH,

55 CLK_CV,

56 UP_DOWN_CV,

57 IMP_CV,


59 W,

60 CLK_COEF,

61 OFF_ON_CHIP,

62 COS,

63 OFF_CHIP_DAC_P,

64 ELECTRODE_OFFON,

65 READ_PWM,

66 CLK_SHFIT_RING,

67 CLK_RING,

68 READ_ON_OFF_PWM,

69 RESET_RING,

70 WRITE_PAD,

71 EXT_RING,

72 ON_OFF_RING,

73 B_PID,

74 A_PID,

75 ADC_PID,

76 ADC_MEM_READ,

77 MAINE_READ,

78 RESET_,


79 WP0,

80 WP1,

81 WP2,

82 WP3,

83 RESET_P,

84 CLK_P,

85 PAR_P,

86 RESET_PS,

87 CLK_PS,

88 LOAD_SHIFT,

89 RESET_SHIFT,

90 IN_COEF_PAD,

91 CLK_SH3_PAD,


93 CLK_TEMP,

94 CHAMBER_RESET,

95 CHAMBER_CLK,

96 WR_USB,

97 RD_USB,

98 SIWUA,

99 SIWUB,

100 B,

101 DATA_OUT,

102 DATA_OUT_USB

103 );






111 output wire CS;

112 output wire WR;

113 output wire PD;










123 output wire C1;


125 output wire P1;




129 output wire Q;

















145 output wire W;














































191 wire [17:0] DIN;

192 wire idel;

193 wire idel2;

194 wire idel3;

195 wire idel4;

196 wire start_DAC;


























227 .CS(CS),

228 .WR(WR),

229 .LDAC(LDAC),

230 .PD(PD),

231 .GAIN(GAIN),

232 .CLR(CLR),


234 .counter(B));




240 .SING(SING),


241 .C1(C1),


243 .P1(P1),


245 .BS_A(BS_A),

246 .BS_P(BS_P),

247 .Q(Q));

248249250 Inv b2v_inst10(

251 .a(idel2),






259260261 Inv b2v_inst12(

262 .a(idel3),















283284285286 PID_CONT b2v_inst17(












297 .B_PID(B_PID),

298 .A_PID(A_PID),





303 .WP0(WP0),

304 .WP1(WP1),

305 .WP2(WP2),

306 .WP3(WP3),


308 .CLK_P(CLK_P),

309 .PAR_P(PAR_P));












321 .B_PID(B_PID),

322 .A_PID(A_PID),





327 .WP0(WP0),

328 .WP1(WP1),

329 .WP2(WP2),

330 .WP3(WP3),


332 .CLK_P(CLK_P),

333 .PAR_P(PAR_P));

334335336 TX b2v_inst18(









347348






353 .DIN(DIN));

354355356 Inv b2v_inst20(

357 .a(idel4),





364 .TXE(TXE_USB),



367 .WR(WR_USB),

368 .SIWUA(SIWUA),


370371372 IO_B b2v_inst22(








382 .DIN(DIN_USB),


384 .RD(RD_USB),

385 .SIWUB(SIWUB),


387 );

388389 TX b2v_inst18(














406 .DIN(DIN));

407408409 Inv b2v_inst20(

410 .a(idel4),





417 .TXE(TXE_USB),



420 .WR(WR_USB),

421 .SIWUA(SIWUA),


423424425 IO_B b2v_inst22(








435 .DIN(DIN_USB),


437 .RD(RD_USB),

438 .SIWUB(SIWUB),


440 );



445 );

















467468469 SRAM b2v_inst30(






475 .W(W),



478 .COS(COS),






486 .Data(DIN),



489 .ldac1(ldac1),

490 .sync1(sync1),


492 .ldac2(ldac2),

493 .sync2(sync2));













510




514 .IDEL1(idel),

515 .IDEL2(idel2),

516 .IDEL3(idel3),

517 .IDEL4(idel4),





524 .y(CLK_SH1));

525526527 Inv b2v_inst8(

528 .a(idel),






536 SRAM b2v_inst30(






542 .W(W),



545 .COS(COS),






553 .Data(DIN),



556 .ldac1(ldac1),

557 .sync1(sync1),


559 .ldac2(ldac2),

560 .sync2(sync2));














577578579 endmodule

A.7 Example Temperature Regulation Verilog Code

The following code generate the results shown in Fig. 5.21.1 // Copyright (C) 1991-2011 Altera Corporation















17 // CREATED "Sun Jul 21 20:36:38 2013"


20 OSC_CLK,

21 TXE_USB,

22 RXF_USB_IN,

23 DATA_IN,

24 DIN_USB,

25 CS,

26 WR,

27 PD,

28 GAIN,

29 CLR,

30 ldac1,


31 sync1,

32 ldac2,

33 sync2,

34 din_out2,

35 din_out1,

36 SING,

37 C1,

38 VLATCH,

39 P1,

40 EXT_RESET,

41 BS_A,

42 BS_P,

43 Q,

44 LDAC,

45 DAC_CLK1,

46 DAC_CLK2,

47 CLK_SH1,

48 I_STAT_PAD,

49 CLK_SH2_TEMP,

50 IN_E_PAD_TEMP,

51 CLK_SHIFT_PAD,

52 I_SHIFT_PAD,

53 RESET_CH,

54 CLK_CH,

55 CLK_CV,

56 UP_DOWN_CV,

57 IMP_CV,


59 W,

60 CLK_COEF,

61 OFF_ON_CHIP,

62 COS,

63 OFF_CHIP_DAC_P,

64 ELECTRODE_OFFON,

65 READ_PWM,

66 CLK_SHFIT_RING,

67 CLK_RING,

68 READ_ON_OFF_PWM,

69 RESET_RING,

70 WRITE_PAD,

71 EXT_RING,

72 ON_OFF_RING,

73 B_PID,

74 A_PID,

75 ADC_PID,

76 ADC_MEM_READ,

77 MAINE_READ,

78 RESET_,

79 WP0,

80 WP1,

81 WP2,

82 WP3,

83 RESET_P,

84 CLK_P,


85 PAR_P,

86 RESET_PS,

87 CLK_PS,

88 LOAD_SHIFT,

89 RESET_SHIFT,

90 IN_COEF_PAD,

91 CLK_SH3_PAD,


93 CLK_TEMP,

94 CHAMBER_RESET,

95 CHAMBER_CLK,

96 WR_USB,

97 RD_USB,

98 SIWUA,

99 SIWUB,

100 B,

101 DATA_OUT,

102 DATA_OUT_USB

103 );






111 output wire CS;

112 output wire WR;

113 output wire PD;










123 output wire C1;


125 output wire P1;




129 output wire Q;

















145 output wire W;













































191 wire [17:0] DIN;

192 wire idel;


193 wire idel2;

194 wire idel3;

195 wire idel4;

196 wire start_DAC;



































233 .B_PID(B_PID),

234 .A_PID(A_PID),





239 .WP0(WP0),

240 .WP1(WP1),

241 .WP2(WP2),

242 .WP3(WP3),


244 .CLK_P(CLK_P),

245 .PAR_P(PAR_P));

246


247248 TX b2v_inst18(













265 .DIN(DIN));

266267268 Inv b2v_inst20(

269 .a(idel4),





276 .TXE(TXE_USB),



279 .WR(WR_USB),

280 .SIWUA(SIWUA),


282283284 IO_B b2v_inst22(








294 .DIN(DIN_USB),


296 .RD(RD_USB),

297 .SIWUB(SIWUB),


299 );

300




304 .CS(CS),

305 .WR(WR),

306 .LDAC(LDAC),

307 .PD(PD),

308 .GAIN(GAIN),

309 .CLR(CLR),


311 .counter(B));




317 .SING(SING),

318 .C1(C1),


320 .P1(P1),


322 .BS_A(BS_A),

323 .BS_P(BS_P),

324 .Q(Q));

325326327 Inv b2v_inst10(

328 .a(idel2),






336337338 Inv b2v_inst12(

339 .a(idel3),
















360361362363 PID_CONT b2v_inst17(











374 .B_PID(B_PID),

375 .A_PID(A_PID),





380 .WP0(WP0),

381 .WP1(WP1),

382 .WP2(WP2),

383 .WP3(WP3),


385 .CLK_P(CLK_P),

386 .PAR_P(PAR_P));

387388389 TX b2v_inst18(













406 .DIN(DIN));

407408


409 Inv b2v_inst20(

410 .a(idel4),





417 .TXE(TXE_USB),



420 .WR(WR_USB),

421 .SIWUA(SIWUA),


423424425 IO_B b2v_inst22(








435 .DIN(DIN_USB),


437 .RD(RD_USB),

438 .SIWUB(SIWUB),


440 );



445 );










461462








469470471 SRAM b2v_inst30(






477 .W(W),



480 .COS(COS),






488 .Data(DIN),



491 .ldac1(ldac1),

492 .sync1(sync1),


494 .ldac2(ldac2),

495 .sync2(sync2));















516 .IDEL1(idel),


517 .IDEL2(idel2),

518 .IDEL3(idel3),

519 .IDEL4(idel4),





526 .y(CLK_SH1));

527528529 Inv b2v_inst8(

530 .a(idel),

















549 .B_PID(B_PID),

550 .A_PID(A_PID),





555 .WP0(WP0),

556 .WP1(WP1),

557 .WP2(WP2),

558 .WP3(WP3),


560 .CLK_P(CLK_P),

561 .PAR_P(PAR_P));

562563564 TX b2v_inst18(





569570










581 .DIN(DIN));

582583584 Inv b2v_inst20(

585 .a(idel4),





592 .TXE(TXE_USB),



595 .WR(WR_USB),

596 .SIWUA(SIWUA),


598599600 IO_B b2v_inst22(








610 .DIN(DIN_USB),


612 .RD(RD_USB),

613 .SIWUB(SIWUB),


615 );

616 Inv b2v_inst8(

617 .a(idel),


















636 .B_PID(B_PID),

637 .A_PID(A_PID),





642 .WP0(WP0),

643 .WP1(WP1),

644 .WP2(WP2),

645 .WP3(WP3),


647 .CLK_P(CLK_P),

648 .PAR_P(PAR_P));

649650651 TX b2v_inst18(





656657 endmodule