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1118 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 48, NO. 6, DECEMBER 2001
Harmonic Filtering of High-Power 12-Pulse RectifierLoads With a Selective Hybrid Filter System
Duro Basic, Victor S. Ramsden, and Peeter K. Muttik , Member, IEEE
Abstract—Current distortion of 12-pulse rectifier loads is sig-nificantly lower compared to six-pulse rectifier loads. However, inpassive filtering of the lowest anddominant characteristic 11th and13th harmonics the use of 5th and 7th filters is often required inorder to prevent possible parallel and series resonance betweenpassive filter and source impedance which can be excited by sourcebackground distortion or by load current residual noncharacter-istic harmonics at the 5th and 7th harmonic frequencies. In hy-brid filter systems, an active filter (AF) can be added in series withthe passive filter in order to isolate the source and load. In mostproposed hybrid filter systems, AF control is based on the detec-tion of total current distortion and high-frequency inverters. Witha selective AF control system and voltage-controlled inverter, theAF can be controlled to isolate the load at the critical frequen-cies only while at all other frequencies the passive filter functionis preserved so that lower switching frequency and AF rating is re-quired. In this paper, we present a selective AF filter control systemand simple hybrid filter topology suitable for the compensation of high-power 12-pulse rectifier loads. Harmonic current controllersbased on the second-order infinite-impulse response digital reso-nant filters are used, as they can be considered as simple digitalalgorithms for more complex double cascaded synchronous-refer-ence-frame-based proportional plus integral controllers. They arecentered to the targeted harmonic frequencies by using an adap-tive fundamental frequency tracking filter. This approach givesgood results, even if the reference waveform (in our case, a loadvoltage) is highly distorted or unbalanced and no separate phase-locked loop is required. Test results for a laboratory model of thissystem and stability analysis are presented and the importance of
delay-time compensation is discussed.
Index Terms—Active filters, harmonics, hybrid filters.
I. INTRODUCTION
T HE lowest harmonics in the source current spectrum of a
12–pulse rectifier are theoretically the 11th and 13th har-
monics, but some residual noncharacteristic 5th and 7th har-
monics can be present. Normally, filtering of 11th and 13th
characteristic harmonics is required to reduce voltage distortion
at the point of common coupling. However, 5th and 7th har-
monic filters are often required in order to prevent possible par-
Manuscript received February 17, 1999; revised June 1, 2001. Abstract pub-lished on the Internet October 24, 2001.
D. Basic is with the Centre for Electrical Machines and Power Elec-tronics, University of Technology, Sydney, N.S.W. 2007, Australia (e-mail:[email protected]).
V. S. Ramsden, retired, was with the Electrical Engineering Group, Facultyof Engineering, University of Technology, Sydney, N.S.W. 2007, Australia.He is now at 13 Bareena Rd., Avalon, N.S.W. 2107, Australia (e-mail:[email protected]).
P. K. Muttik is with Transmission and Distribution Systems, ALSTOMAustralia Ltd., Milperra, N.S.W. 2214, Australia (e-mail: [email protected]).
Publisher Item Identifier S 0278-0046(01)10280-7.
allel and series resonance between the passive filter and source
impedance which can be excited by the source background dis-
tortion or by load noncharacteristic 5th and 7th harmonics.
Hybrid filters with a shunt passive filter and a small-rating
active filter (AF) in series with the supply [1] or in series with a
passive filter [2], [3] have been proposed for harmonic isolation
of large rectifier loads with a simple control strategy based on
a proportional controller and detection of total source current
distortion (obtained after subtraction of the fundamental com-
ponent). In both cases, the AF behaves as a resistor at harmonic
frequencies in series with the supply providing harmonic iso-
lation. A proportional controller, however, cannot provide sat-isfactory attenuation of source current harmonics if the passive
filter is not properly tuned at the dominant load harmonics, and
a broad-band high-frequency AF inverter is required.
For large 12–pulse rectifiers, a selective AF control system
has been proposed [4] with full isolation at 5th and 7th har-
monic frequencies achieved with square-wave voltage injection
into dominant harmonic (11th and 13th) passive filters. For the
detection and control of 5th and 7th harmonics, low-pass fil-
ters and proportional plus integral (PI) controllers were applied
in reference frames rotating synchronously with corresponding
harmonic space vectors. This technique was successfully used in
vector-controlled ac drives for many years and later applied for
AFs [5]. However, a single synchronous reference frame (SRF)is appropriate for balanced three-phase systems only because
it can track only positive- or negative-sequence vectors. For
tracking both sequence harmonic vectors in unbalanced three-
phase systems, double cascaded SRFs have been proposed [6],
resulting in complex AF control systems, especially if tracking
several spectral components is required.
In this paper, we propose and examine a selective hybrid filter
system with a voltage-controlled inverter suitable for the har-
monic isolation of high-power 12-pulse rectifier loads at the
critical frequencies. The AF is connected in parallel with the
load through a simple tuned passive filter created by a power-
factor-correction capacitor and the AF matching transformer
leakage inductance. The selective AF control system is based onsource current detection and second-order infinite-impulse re-
sponse (IIR) digital notch and resonant filters. These filters can
be considered as a simple digital algorithm for double cascaded
SRF notch filters or PI controllers [7], [12] and they are suitable
for tracking multiple harmonics [8]. Estimation of the funda-
mental and targeted harmonic frequencies is based on an adap-
tive notch filter so that an additional phase-locked loop (PLL)
is not required. Experimental results and stability analysis are
presented and the importance of delay-time compensation is dis-
cussed.
0278–0046/01$10.00 © 2001 IEEE
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Fig. 1. (a) System configuration. (b) Equivalent block diagram of the system.
II. SYSTEM CONFIGURATION AND PRINCIPLE OF OPERATION
A simplified drawing of the system configuration is shown in
Fig. 1(a). A series combination of a single tuned filter and AF
filter is connected in parallel with a 12-pulse rectifier load. If the
nonlinear load can be considered as a harmonic current source
generator, thesystem of Fig. 1(a) can be represented at harmonic
frequencies by the block diagram shown in Fig. 1(b), where
is voltage background distortion, and is AF voltage injec-
tion.
The load and passive filter can be isolated from the source at
targeted harmonic frequencies by using a closed-loop system
which will shape the AF voltage harmonic injection to pro-
duce AF current exactly equal to the targeted load cur-
rent harmonics . The source current will be zero even in
presence of disturbance . Thus the load current harmonics
should be detected ( ) and used as the current reference .
This reference is compared with detected AF current ( )
and the error is corrected by the controller ( ) and AF in-
verter ( ) as shown in Fig. 2(a). This scheme can be usedwith a simple hysteresis controller resulting in a current con-
trolled voltage inverter or with P controller and voltage-con-
trolled inverter [9]. Instead of separate detection of the load and
AF currents, the error signal can be directly retrieved from the
sourcecurrentdistortion ( ) asshownin Fig.2(b).The block
diagram of Fig. 2(b) can be rearranged as shown in Fig. 2(c)
so that the contributions to the source current distortion of the
load current and source voltage are clearly visible. It is obvious
than the source current harmonics with the AF will be lower
by the factor than with the passive filter only,
where is the transfer function of the AF control system
[see Fig. 2(c)].
Fig. 2. Block diagrams showing closed-loop control of the AF current. (a)With separate detection of the load and AF currents.(b) With direct error signal(source current) detection. (c) Modified block diagram of Fig. 2(b).
As mentioned earlier, we have selected a system based on
the source current detection and voltage-controlled AF inverter
[Fig. 2(c)]. This method requires fewer current sensors, and theselective AF harmonic voltage injection targets several critical
source current harmonics only, while at all other frequencies the
passive filter function is mostly preserved.
III. AF CONTROL SYSTEM
Two banks of harmonic controllers in Fig. 3 track corre-
sponding harmonics in the source current and adjust AF voltage
until their full cancellation is achieved. Only two controller
banks are necessary in a three-wire three-phase power system
and they can be designed in the phase domain or, as in ourcase, in the domain (block ). Load voltage signals
and an adaptive notch filter are used to track the fundamental
frequency. The retrieved voltage fundamental components are
used to estimate the passive filter fundamental current neces-
sary for the AF inverter dc voltage control by balancing the
total active power flowing into the AF dc-link capacitor. From
the estimated fundamental frequency the frequencies of
the targeted harmonics are calculated and transferred to the
harmonic current controllers. Load voltage harmonics can be
retrieved also and used as a feedforward compensation of the
disturbance signals in Fig. 2(c). Delay-time compensation is
introduced at the harmonic controller outputs to stabilize the
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1120 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 48, NO. 6, DECEMBER 2001
Fig. 3. Selective active filter control system with tracking and controlling four critical harmonics.
AF control system. All blocks in Fig. 3 will be described in
more detail.
A. Fundamental Frequency Notch Filter
In nonselective AF control systems based on the detection of
total current distortion [1]–[3], the removal of the fundamental
componentis practically theonly and most importantsignal pro-
cessing task. However, a selective control system may not be
sensitive to the fundamental component and the fundamental
component notching can be omitted., but, to provide the pos-
sibility of using a proportional gain ( ), useful if a broader
range of harmonics should be attenuated, a fundamental com-
ponent notch filter has been implemented. A notch filter can be
constructed by using SRF notch filters [5] or by using the
theory [1]–[3]. However, neither of these methods can provide
full notching of the fundamental component in an unbalanced
situation. SRF-based notch filters will completely pass the neg-
ative-sequence components and -based notch filters will in-
troduce on top of that an additional harmonic distortion (mainly,3rd harmonic). To solve these problems, two cascaded SRFs can
be used for separate notching of the positive- and negative-se-
quence components (Fig. 4). The transfer function of double
SRF notch filters given in (1) can be transformed by using the
bilinear transformation into a discrete form that will result in an
IIR second-order digital notch filter [7], [12] (2)
(1)
(2)
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Fig. 4. Double-cascaded-SRF-based notch filters.
where ( , and are sampling time and
frequency), and
(3)
The notch width is dependent on the cutoff frequency of the
SRF-based low-pass filter. If the location of the poles in (2) isvery close to unity , a very narrow notch filter can be
created. The center frequency of the notch is determined by the
center frequency parameter .
B. Harmonic Controllers
A PI controller (4) is commonly used for the tracking of
dc signals because it produces zero steady-state error in a
closed-loop system
(4)
Three-phase current components at arbitrary frequency can
be converted into dc signals by filtering the currents transformed
into SRF coordinates ( , block in Figs. 4 and 5). Thus,
SRF-based PI controllers can be used for tracking sinusoidal
currents [Fig. 5(a)] and this technique is in common use in ac
motor drives. Our simulation results showed that, because of
an additional phase shift introduced by the low-pass filters in
Fig. 5(a), small PI gains are required for a stable closed-loop
system, causing a poor transient response. However, these fil-
ters can be omitted because of the low-pass nature of a PI con-
troller (in this case, the fundamental frequency notch filter is re-
quired). A single SRF PI controller of Fig. 5(a) can be used for
controlling either a positive- or a negative-sequence componentat synchronous frequency. In unbalanced three-phase three-wire
systems, both sequences are present and a PI controller based on
a single SRF cannot provide full compensation of critical har-
monics, which can be a problem because even a very small un-
compensated harmonic current can be amplified or can create
a high-voltage distortion in a resonant power system. To over-
come this problem, we can use double cascaded SRF controllers
[Fig. 5(b)].
In this case, the resultant transfer function is
(5)
Fig. 5. (a) Single and (b) double cascaded SRF PI controllers.
This transfer function can be transformed into a digital form
by using the bilinear transformation [7], [12], giving a simple
second-order IIR digital resonant filter (6)
(6)
The filter parameters are related to PI controller parameters as
follows:
(7)
The controller bandwidth depends on the P and I gains ( and
) of the analog prototype. If the location of the zeros in (6) is
very close to unity , a narrow bandwidth controller can
be created. At the center frequency, this resonant filter produces
infinite gain and no phase shift.
C. Frequency Tracking
An adaptive IIR notch tracking filter is used for fundamental
frequency component estimation. By this technique, an addi-
tional PLL is not necessary. All other harmonically related com-
ponents can be retrieved if the fundamental frequency is known.
The direct notch filter form (2) can be transformed into latticeform as a numerically reliable alternative [10]
(8)
with the complementary (band-pass) transfer function
(9)
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1122 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 48, NO. 6, DECEMBER 2001
Fig. 6. Adaptive lattice second-order IIR notch filter for fundamentalfrequency tracking.
The center frequency is defined by the parameter
(10)
The parameter defines the filter bandwidth. It is related to
as follows:
(11)
The filter algorithmis shown in Fig. 6 in lattice form where
is the filter input, and are the filter state variables,
and and the notch filter output and the complemen-
tary band-pass filter output, respectively. This notch filter can
be used to track the fundamental frequency. The simplest way
is to use an adaptive algorithm which seeks a minimum point
of the cost function (expected value of the squared filter outputs
) by changing the parameter in the negative gradient di-
rection (gradient-descent algorithm). To minimize the noise-in-duced term of the cost function, a narrow bandwidth is required.
In this situation, the gradient-descent adaptive algorithm can ex-
hibit very slow convergence and a simplified adaptation algo-
rithm for the lattice form can improve the convergence speed
[10]. This version of the lattice algorithm is
(12)
where instead of using and calculating the gradient ,
the filter internal state is used. The minimum is normally
achieved when the fundamental component is notched
and, thus,
(13)
Center frequency parameters for all harmonics can be directly
calculated from (14)
(14)
To reduce the computation burden, we adopted a recursive
technique. This technique is based on the fact that a dis-
crete value of a cosine function at sampling instants
can be found recursively
from two last samples and and the angular
increment
(15)
Combining(14) and (15), a recursive equation can be derived for
the center frequency parameters of all harmonics
(16)
D. Delay-Time Compensation
Because of synchronous sampling of the inputs and updating
of the outputs, a pure delay of a sample is introduced. Be-
cause of zeroth-order sample–hold ( delay) and AF in-
verter deadtime ( ), an additional delay of approximately
is introduced so the total time delay in-
troduced in the closed loop of the system in Fig. 5 is nearly .Phase shift caused by the system delay can be considerable at
the harmonic frequencies and, as will be shown in Section V, it
can lead to instability if the compensation of higher order har-
monics is required. For example, for a sampling frequency of
kHz, the phase lag at the 11th and 13th harmonics will
be approximately 45 and 117 .
In a selective AF control system, it is possible to predict an
individual cosinesignal one samplein advance by using (15), as-
suming that the amplitude of the signal will stay approximately
the same
(17)
From (17), the prediction two samples in advance will be
(18)
Thus, with the correction given in (19) applied at outputs of
all harmonic controllers (delay time compensation blocks), it
is possible to compensate for the phase lag at the targeted fre-
quencies introduced in the closed loop due to system delay time
(19)
E. AF Inverter DC Voltage Control
AF dc capacitor voltage is maintained at the reference
value by a separate PI controller which corrects the AF ref-
erence so that AF filter introduces an additional voltage at fun-
damental frequency in phase with the passive filter fundamental
current
(20)
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Fig. 7. Impedance seen from source side (Z ), load side ( Z ), and currentmagnification factor ( K ) for system of Fig. 1(a) with the passive filter only.
The total active power flow into the AF is controlled in both di-
rections at the fundamental frequency with the regulator output
so that the dc capacitor voltage remains constant without anadditional active source at the dc side. The filter fundamental
current is estimated from the retrieved voltage signals and the
passive filter impedance [(20)].
IV. PERFORMANCE ANALYSIS OF AN EXAMPLE SYSTEM
Firstly, we will analyze the performance of a laboratory
model of the system of Fig. 1 without the AF and with a single
tuned passive filter. In our experimental setup, the system
impedance was 4.5% ( mH, V, A).
The passive filter tuned frequency was adjusted approximately
to the 11th harmonic. The passive filter capacitance was 90
F and the tuning inductor was the AF matching transformer( ) leakage inductance with the AF side short circuited.
For these system parameters, the impedance seen from the load
side ( ) and source current magnification ( )
(21)
are shown in Fig. 7. They have a parallel resonance peak be-
tween 5th–7th harmonics that can cause an excessive voltagedistortion due to load current harmonics. The impedance seen
from the load side ( , Fig. 7) has a series res-
onance minimum between 5th–7th harmonics that can cause an
excessive source current flowing into the passive filter due tobackground source voltage distortion.
Several experiments were carried out to illustrate problems
related to the applications of passive compensation of 12-pulse
loads. The voltage and current waveforms and source current
spectrum with the load only are shown in Fig. 8(a). The most
emphasized harmonics of the load current are characteristic 11th
and 13th harmonics, but small residual 5th and 7th harmonics
are present [Fig. 8(a)]. The load voltage waveform is highly
distorted with typical notches caused by the thyristor commu-
tation. In the following test, only the passive filter was con-
nected without the load. The passive filter current [Fig. 8(b)]
is highly distorted because of the series resonance and back-
ground voltage distortion. In our laboratory the dominant back-
ground distortion (Table I) was at 3rd and 7th harmonics. Un-
expectedly, the 5th harmonic was relatively small and, thus, the
harmonic spectrum of the passive filter current in Fig. 8(b) in-
dicates strong 7th harmonic component only. The source cur-
rent and voltage waveforms and source current spectrum with
the load and passive filter connected are shown in Fig. 8(c). In
comparison to the previous results, the source current is nowmore distorted because of the additional distortion created by
the magnification of the residual 5th and 7th load current har-
monics. All higher frequency components, including the domi-
nant characteristic 11th and 13th harmonics of the load current,
are attenuated well and, consequently, the commutation notches
on the voltage waveform have disappeared.
From these results, we can conclude that although the most
emphasized characteristic harmonics are attenuated well, we
cannot apply passive filters for characteristic harmonics only
without additional 5th and 7th harmonic filters in order to pre-
vent parallel resonance. In this case, a selective AF filter can be
used very effectively to prevent the problems at 5th and 7th har-
monics and to improve the passive filter performance at the 11thand 13th harmonics.
Finally, we will show the results when the AF is applied. For
the AF control system a digital signal processor (DSP) board
with TMS320C32-60Mz processor was used. Two source
currents, two load, and dc capacitor voltages were detected by
on-board 16-bit A/D converters. The AF inverter was controlled
directly through on-board digital outputs because space-vector
pulsewidth modulation (PWM) was implemented as a part of
the AF control program. The sampling frequency was set at
kHz and the AF control system was programmed to
target 5th, 7th, 11th, and 13th harmonics. All necessary signal
processing tasks were executed by second-order IIR digital
filter blocks (22)
(22)
The parameters of the fundamental frequency notch and
tracking filters and harmonic controllers are given in Table II.
The source current and voltage waveforms shown in Fig. 8(d)
are considerably improved because all targeted harmonics are
reduced to negligible levels and the resonance phenomena at 5th
and 7th harmonics are prevented. The AF current and voltage
waveforms are also shown in Fig. 8(d). It can be noticed that
the AF voltage is much lower than the load voltage. High-fre-quency harmonics are filtered out by the passive filter so that the
required AF frequency band is restricted to the 13th harmonic.
Thus, the AF voltage rating and switching frequency in this hy-
brid filter topology can be much lower than with a shunt AF of
similar performance.
V. STABILITY ANALYSIS
For the stability analysis, the block diagram of Fig. 9 will be
used. The source current is sampled and processed by the AF
digital controller [transfer function ]. Because of syn-
chronous sampling of the inputs and outputs the control signal
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1124 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 48, NO. 6, DECEMBER 2001
(a)
(b)
(c)
(d)
Fig. 8. Current and voltage waveforms and harmonic spectra. (a) With load only. (b) With the passive filter only. (c) With load and passive filter. (d) With load,passive, and active filters.
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1126 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 48, NO. 6, DECEMBER 2001
Fig. 9. Block diagram of the AF closed-oop system with digital controller used in the stability analysis.
Fig. 10. (a) Modulus and argument and (b) polar plot of the AF controllertransfer function G ( j ! ) .
Without the compensation, the system is not stable at higher
frequencies for inductive , while in the region where is
capacitive, the compensation may not be required. An expanded
view of the rectangle in Fig. 12(a) is shown in Fig. 12(b), con-
firming that the closed-loop system with the delay-time com-
pensation is stable as the Nyquist diagram does not encircle
the point ( 1, ). However, the stability margin may be low
if low sampling frequency is used and can be considerably
improved if higher sampling frequency is used.
Fig. 11. (a) Modulus and argument of the AF control loop pulse transferfunction G ( j ! ) without the delay time compensation and (b) Nyquist plotin this case.
VI. CONCLUSION
A hybridfilter systemwith selective AF control allows theuse
of a low-rating AF with reduced switching frequency that is par-
ticularly advantageous in high-power applications. The effec-
tiveness of a small-rating AF connected in series with a low-cost
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BASIC et al.: HARMONIC FILTERING OF HIGH-POWER 12-PULSE RECTIFIER LOADS 1127
Fig. 12. (a) Nyquist plot for the AF system with the delay time compensationand (b) an expanded view around the critical point ( 0 1, j 0 ).
power-factor-correction capacitor is experimentally verified in
the filtering of the dominant harmonics of a 12–pulse rectifier
load and preventing series and parallel resonance conditions
by targeting several critical harmonics. Harmonic current con-
trollers based on IIR second-order digital resonant filters are
centered to the targeted harmonic frequencies by using an adap-
tive fundamental frequency tracking filter. This approach gives
good synchronization, even if the reference waveform (in our
case, a load voltage) is highly distorted and no separate PLL is
required. Stability analysis was carried out and it showed the
importance of the system delay-time compensation.
REFERENCES
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[2] H.Fujita andH. Akagi, “Apracticalapproachto harmonic compensationin power systems—Series connectionof passive and active filters,” IEEE Trans. Ind. Applicat., vol. 27, pp. 1020–1025, Nov./Dec. 1991.
[3] , “Design strategy for the combined system of shunt passive andseries active filters,” in Conf. Rec. IEEE-IAS Annu. Meeting, vol. 1,Sept./Oct. 1991, pp. 898–903.
[4] P. T. Cheng, S. Bhattachaarya, and D. M. Divan, “Application of domi-nant harmonic active filter system with 12 pulse nonlinear loads,” IEEE Trans. Power Delivery, vol. 14, pp. 642–647, Apr. 1999.
[5] S. Bhattacharya and D. Divan, “Synchronous frame based controllerimplementation for a hybrid series active filter system,” in Conf. Rec.
IEEE-IAS Annu. Meeting, vol. 3, Oct. 1995, pp. 2531–2540.[6] J. Hafner, M. Aredes, and K. Neumann, “A shunt active power filter
applied to high voltage distribution lines,” IEEE Trans. Power Delivery,vol. 12, pp. 266–272, Jan. 1997.
[7] D. Basic, V. S. Ramsden, and P. Muttik, “Digital implementation of thesynchronous reference frame controller for a selective hybrid filter con-trol system,” in Proc. AUPEC/EECON’99, Sept. 1999, pp. 473–478.
[8] W. Zhang, A. J. Isaksson, and A. Ekstorm, “Analysis on the controlprinciple of the active DC filter in the Lindome converter station of the Konti–Skan HVDC link,” IEEE Trans. Power Syst., vol. 13, pp.374–381, May 1998.
[9] F. Z. Peng, H. Akagi, and A. Nabae, “Compensation characteristics of the combined system of shunt passive and series active filter,” IEEE Trans. Ind. Applicat., vol. 29, pp. 144–152, Jan./Feb. 1993.
[10] A. Regalia, Adaptive IIR Filtering in Signal Processing and Con-trol. New York: Marcel Dekker, 1995.
[11] C. L. Phillips and H. T. Nagle, Digital Control System—Analysis and Design. Englewood Cliffs, NJ: Prentice-Hall, 1990.
[12] D. Basic, V. S. Ramsden, and P. K. Muttik, “Hybrid filter controlsystem with adaptive filters for selective elimination of harmonics andinterharmonics,” Proc. IEE—Elect. Power Applicat., vol. 147, no. 4,pp. 295–303, July 2000.
Duro Basic received the Dipl.Eng. degree fromthe University of Novi Sad, Novi Sad, Yugoslavia,the M.E. degree from the University of Belgrade,Belgrade, Yugoslavia, and the Ph.D. degree fromthe University of Technology, Sydney, Australia, in1981, 1993, and 2001, respectively, all in electricalengineering.
He is currentlya Research Officerworkingon con-trol of electrical drives at the Centre for ElectricalMachines and Power Electronics (CEMPE), Univer-sity of Technology. His research interests are power
electronics, active filters, power quality, and control of electrical drives.
Victor S. Ramsden graduated in electrical engi-neering in 1964 and received the Master’s degreein 1965 from Melbourne University, Melbourne,Australia, and received the Ph.D. degree from theUniversity of Aston, Birmingham, U.K.
He spent one year with ASEA in Sweden and oneyear with GEC Stafford. In 1972, he joined the Uni-versity of Technology, Sydney, Australia (UTS), ob-taining a Professorship in Electrical Engineering in1993. Beginning in 1988, he lead a collaboration onpermanent-magnet machine design between UTS and
CSIRO Telecommunicationsand Industrial Physics, where he worked part time.His research interests include ac motor control, electrical machine design, ironlosses, renewable energy, and medical applications. He retired in 2000 and re-mains an Emeritus Professor with UTS.
Peeter K. Muttik (S’78–M’79) received the B.Sc.,B.E. (Hons.), and Ph.D. degrees from the Universityof Adelaide, Adelaide, Australia, in 1973, 1974, and1980, respectively.
He currently holds the position of Chief Engineer,Transmission and Distribution Systems, in theproject sector of ALSTOM Australia Ltd., Milperra,Australia, which he joined in 1980. He has wideexperience in power system analysis, static varcompensators and other high-power electronicsturnkey projects, and in harmonic filter design,
commissioning, and testing.Dr. Muttik is a member of the Institution of Engineers, Australia.