Abstract—This paper proposes a family of magnetic coupling non-isolated bidirectional dc-dc converters, which achieve soft-switching operation in both power flow directions with a simple auxiliary circuit. Compared with the conventional zero-voltage-transition (ZVT) converters, the additional resonant inductor is eliminated and magnetic core number is reduced. Therefore, both high efficiency and low cost are achieved. In the paper, a general zero-voltage-switching (ZVS) magnetic coupling structure is firstly proposed for bidirectional buck/boost converter, based on which 13 different ZVS topologies with different connections of auxiliary circuit are derived. It is very beneficial in practical industrial applications because engineers can choose an optimal one according to converter performance characteristics, and the topology derivation methodology can be easily extended to other bidirectional dc-dc converters. Secondly, operation principle and performance characteristics are simultaneously obtained for all proposed topologies based on the general structure, which can facilitate topologies comparison and selection. Finally, in order to verify the effectiveness of theoretical analysis, design considerations and experiment results of a 500W prototype circuit are demonstrated.

Index Terms—Bidirectional dc-dc converters, coupled inductor, topology derivation, zero-voltage-switching (ZVS).

I. INTRODUCTION

on-isolated bidirectional dc-dc converters (BDCs) which are able to transfer power in both directions, have been

widely employed in various industrial applications, such as uninterruptable power supplies, electric vehicles and so on [1-9]. BDCs can be simply derived from unidirectional dc-dc converters by replacing diodes with switches and thus in the hard-switching BDCs, high switching losses are similarly caused by hard-switching operation. Moreover, with increase of switching frequency, the losses further deteriorate. In order to effectively minimize switching losses, many soft-switching techniques including zero-voltage-switching (ZVS) converters with large inductor ripple current [10-12], active-clamping ZVS converters [13-16], zero-voltage-transition (ZVT) converters [17-24] and ZVS converters with coupled inductors [25-38], have been proposed.

N

In [10-11], multiphase interleaved bidirectional converters with large inductor ripple current were proposed, which can achieve ZVS operation for switches without additional

components. However, resulting from the large inductor ripple current, conduction losses significantly increase in each converter and the efficiency consequently decreases, especially at light load condition. Although adaptive frequency modulation was presented in [12] to reduce the conduction losses, penalty on control complexity is introduced. Moreover, the ZVS realization method is not suitable for non-interleaved applications with only single converter because large input/output filters are required.

In order to achieve soft-switching operation and eliminate the stringent requirement of filters as well, an active-clamping converter was proposed in [13], in which both main switches and auxiliary switch are ZVS. As a result, low switching losses are obtained. Unfortunately, large current circulates in the auxiliary circuit and voltage stresses of switches increase, resulting in high conduction losses. In [14, 15], an auxiliary resonant circuit is employed to achieve low circulating current. But the auxiliary switch is nearly hard-switching.

Recently, ZVT bidirectional converters have been very attractive because of its soft-switching operation, low voltage stresses and high efficiency [17-23]. The auxiliary circuit is removed out of the main circuit and is only activated in a short interval, hence additional conduction losses are small. Besides, the auxiliary switches are zero-current turned on and zero-voltage turned off in [17-19], which are further improved in [20-23] with zero-current turn-on as well as turn-off. Nevertheless, additional magnetic cores are needed in ZVT converters to implement auxiliary resonant inductors, which result in increased cost.

The additional magnetic cores are eliminated in the ZVS dc-dc converters with coupled inductors [25-38], which utilize the leakage inductor to function as the auxiliary resonant inductor. The operation is similar with that of ZVT converters, and thus soft-switching operation as well as improved efficiency are also attained. In [25], a family of magnetic coupling bidirectional dc-dc converters was derived, which can achieve ZVS operation in both power flow directions. Unfortunately, two extra diodes are required in series connection with two auxiliary switches to implement unidirectional switches. The extra diodes are eliminated in the ZVS magnetic coupling bidirectional converters presented in [26]. Meanwhile, advantages of soft-switching operation and reduced magnetic core remain. However, only one specific ZVS magnetic coupling topology has been proposed for different dc-dc converters in [26], while optional topologies should be

111Equation Chapter 1 Section 1A Family of Zero-Voltage-Switching Magnetic Coupling Non-

isolated Bidirectional DC-DC Converters

1

provided for engineers to choose an optimum one according to the requirement in different practical applications. Therefore, this paper aims to explore and provide more alternative topologies. Based on the topology derivation methodology in [27], 13 different viable ZVS magnetic coupling bidirectional buck/boost topologies are firstly obtained in the paper, including the converter presented in [26]. With different auxiliary circuit connections, the converter performance characteristics are diverse and thus are preferred in different applications. In order to simplify the selection of an optimal topology for a specific application, generalized analysis is also provided in the paper, from which the performance characteristic of each converter can be easily obtained.

The paper is organized as follows. A family of ZVS magnetic coupling bidirectional buck/boost converters are proposed in Section II. In section III and IV, operation principle and performance characteristics effective for all proposed converters are introduced in detail, based on the general structure. Then design considerations of a 500W prototype circuit and experimental verifications are presented in section V. Finally, conclusions are drawn in Section VI.

II.ZVS MAGNETIC COUPLING BIDIRECTIONAL BUCK/BOOST CONVERTERS

General structure of the proposed ZVS magnetic coupling bidirectional buck/boost converters is shown in Fig. 1(a). A simple auxiliary circuit comprised of an additional secondary winding and a pair of switches Sa1~Sa2 is added to achieve ZVS operation for switches S1~S2 in both buck and boost power flow directions. Moreover, the auxiliary switches Sa1~Sa2 are zero-current-switching (ZCS) and are only activated in a short interval. Therefore, high efficiency can be obtained. Furthermore, no extra resonant inductor is needed due to the exploitation of leakage inductor and thus lower cost is achieved in the proposed converters when compared with ZVT converters in [17~23]. It is noted that the auxiliary circuit in Fig. 1 (a) also can be easily employed to achieve ZVS operation for other non-isolated bidirectional converters besides the buck/boost converter, e.g. bidirectional buck-boost/buck-boost, Cuk/Cuk and zeta/sepic converters. Likewise, with different auxiliary circuit connections (p, q), different viable topologies also can be derived for each converter.

V1

S1

S2 V2

Sa2 Sa1q p

a b c

d

T1

(a)

V1

S1 T1

S2 V2

a b c

dSa2 Sa1q p

iLr Lr

is2

is1

ILm Lm

vLr+ -

ia+-vs2

+vs1-

+vpq-

vLm+ -

i2

(b)Fig. 1. Proposed ZVS magnetic coupling bidirectional buck/boost converters: (a) general structure and (b) equivalent circuit.

From the general structure in Fig. 1 (a), 13 different ZVS

magnetic coupling bidirectional buck/boost converters are obtained in Fig. 2 with different connections of (p, q) to any two nodes among {a, b, c, d}, including one already presented in [26] and other 12 topologies firstly proposed in the paper. With more optional topologies whose performance characteristics are different provided for engineers, an optimum one can be chosen for a specific application.

V1

S1 T1

S2

V2

a b c

d

Sa1 Sa2

p q

V1

S1 T1S2

Sa1 Sa2

V2

a b c

d

p q

V1

S1 T1

S2

V2

a b c

d

Sa1

Sa2

p

q

(a) (b) (c)

V1

S1 T1

S2

V2

a b c

d

Sa2 Sa1

q p

V1

S1 T1

S2

V2

a b c

d

Sa2Sa1

qp

V1

S1 T1

S2 V2

a b c

d q

Sa1

Sa2

p

(d) (e) (f)

V1

S1 T1S2

Sa2 Sa1

V2

a b c

d

q p

V1

S1 T1

S2

V2

a b c

d

Sa1Sa2

pq

V1

S1 T1

S2

V2

a b c

d q

Sa1

Sa2

p

(g) (h) (i)

V1

S1 T1

S2

V2

a b c

d

Sa2

Sa1

q

p

V1

S1 T1

S2 V2

a b c

d p

Sa2

Sa1

q

V1

S1 T1

S2

V2

a b c

d p

Sa2

Sa1

q

(j) (k) (l)

V1

S1

S2 V2

a b c

d

T1

Sa2 Sa1q p

(m)[26] Fig. 2. Proposed ZVS magnetic coupling bidirectional buck/boost converters with different connections of (p, q): (a) (a, b), (b) (a, c), (c) (a, d), (d) (b, a), (e) (b, c), (f) (b, d), (g) (c, a), (h) (c, b), (i) (c, d), (j) (d, a), (k) (d, b), (l) (d, c), and (m) (d, d)[26].

III. OPERATION PRINCIPLE

Equivalent circuit of the proposed general ZVS magnetic coupling bidirectional buck/boost converter is shown in Fig. 1 (b). The coupled inductor T1 is modelled as a magnetizing inductor Lm, a leakage inductor Lr and an ideal transformer with turns ratio of Np:Ns=1:n1. Voltage across the auxiliary circuit vpq for different connections (p, q) in Fig. 2 is derived in 2, in terms of voltage V1, V2 and drain-to-source voltage vs2(t). Values of coefficients k1-k3 for different connections are determined in Table 1. Besides, the relationship between drain-to-source current is1~is2, auxiliary current ia, leakage inductor current iLr, magnetizing current ILm and port current i2

are obtained in 3~5.

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Table 1. Values of k1, k2 and k3 for different topologies in Fig. 2.

Fig. a b c d e f g h i j k l m

2

2k1 1 1 1 -

1 0 0 -1 0 0 -1 0 0 0

k2 0 -1 0 0 -1 0 1 1 1 0 0 -1 0

k3 -1 0 0 1 1 1 0 -1 0 0 -1 0 0To simplify the operation principle, several assumptions are

made: 1) Lm is large and the magnetizing current ILm is assumed to be constant; 2) Lr is much less than Lm; 3) the sum of two switches parasitic capacitances are constant during the switching process, which is denoted as Cs.

Steady-state waveforms of the proposed general ZVS bidirectional buck/boost converters with different power flow directions are respectively shown in Fig. 3(a) and (b). It is noted that Fig. 3 is based on the converter Fig. 2(e), and the steady-state waveforms of other proposed converters in Fig. 2 are similar with Fig. 3. The drive signals vg1 and vg2 of switches S1 and S2 are complementary. In the buck mode, the auxiliary switch Sa1 is turned on φ1 in advance before the turn-off of switch S2 and auxiliary switch Sa2 is always off. The duty-cycle of Sa1 is equal to that of S1. On the other hand, in the boost mode, auxiliary switch Sa1 is always off and auxiliary switch Sa2 is turned on φ2 in advance before the turn-off of switch S1. The duty-cycle of Sa2 is equal to that of S2. Owing to the similarity in buck and boost mode, only operation in the buck mode is detailed introduced in the paper. According to Fig. 3(a), the operation of buck mode in a switching period comprises 7 stages and corresponding equivalent circuits are illustrated in Fig. 4.

Prior to t0, as a result of the previous stage, S1 is on and the auxiliary circuit is active. The auxiliary current ia is decreasing while the leakage inductor current iLr is increasing.

Stage 1 (t0-t1): At t0, ia decays to zero, thus the auxiliary circuit turns to be inactive and the auxiliary switch Sa1 is ZCS turned off in this stage. From 4, the leakage inductor current iLr reaches the magnetizing current ILm. Therefore, the leakage inductor Lr and magnetizing inductor Lm are connected in series and charged by V1-V2. The magnetizing inductor voltage is given in 7 with the assumption that leakage inductance Lr is much smaller than magnetizing inductance Lm. From 2, value of voltage vpq in this stage is derived in 8 with vs2(t)=V1. Then the turn-off voltage of auxiliary switch Vsa2 is obtained in 9 , which should be larger than zero.

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Stage 2 (t1-t2): At t1, S1 is turned off. The parasitic capacitor of S1 is charged by is1 and that of S2 is discharged by -is2, as shown in 10. Therefore, the drain-to-source voltage vs2

decreases linearly.

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Stage 3 (t2-t3): At t2, vs2 drops to zero and current is2 flows through the body diode of S2. In this stage, switch S2 is ZVS turned on. The series connection of leakage inductor Lr and magnetizing inductor Lm are discharged by -V2. From 2, value

t

t

t

t

is2

vs1

vs2

tis1

t

ia

t

tvLr

t0 t5t1 t4 t7t2 t3 t6

iLr

vg1vg2

vgsa1

1

ILm

tvgsa2

Dtd1

(a)

t

t

t

t

vs1vs2

ia

t

tvLr

t7t4t0 t3 t6t1 t2 t5

iLr

vg1vg2

vgsa2

2

ILm

tvgsa1

D

is1 t

tis2

td2

(b)Fig. 3. Steady-state waveforms of the proposed ZVS magnetic coupling bidirectional buck/boost converters: (a) buck mode and (b) boost mode.

V1

S1 T1

S2 V2

a b c

dSa2 Sa1q p

Lr Lm

V1

S1 T1

S2 V2

a b c

dSa2 Sa1q p

Lr Lm

(a) (b)

3

V1

S1 T1

S2 V2

a b c

dSa2 Sa1q p

Lr Lm

V1

S1 T1

S2 V2

a b c

dq p

Lr Lm

Sa2 Sa1

(c) (d)

V1

S1 T1

S2 V2

a b c

dq p

Lr Lm

Sa2 Sa1

V1

S1 T1

S2 V2

a b c

dq p

Lr Lm

Sa2 Sa1

(e) (f)Fig. 4. Equivalent circuits of the general ZVS topology in different stages: (a) stage 1(t0-t1), (b) stage 2(t1-t2), (c) stage 3(t2-t3), (d) stage 4(t3-t4), (e) stage 5(t4-t5), and (f) stage 6(t5-t6), 7(t6-t7).of voltage vpq in this stage is derived in 14 with vs2(t)=0. Then the turn-off voltage of auxiliary switch Vsa1 is obtained in 15, which also should be larger than zero.

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From 9 and 15, 16 is derived.

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Stage 4 (t3-t4): At t3, the auxiliary switch Sa1 is turned on and hence magnetizing voltage vLm is clamped to -Vpq2/n1. From Fig. 4(d), the leakage inductor current iLr is derived in 17. Combining with 3 and 4, the auxiliary current ia and the drain-to-source current is2 are obtained in 18 and 19, respectively. According to 15 and 16, iLr decreases while ia and is2 increase. In this stage, is2 turns from negative to positive.

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Stage 5 (t4-t5): At t4, the current is2 is larger than zero and switch S2 is turned off. In this stage, the leakage inductor Lr

resonates with parasitic capacitors of switches. The drain-to-source voltage vs1 and leakage inductor current iLr are derived in 20 with parameters Vc, A, ω and θ in 21. Then the interval t5-t4 also can be obtained, as shown in 22.

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Stage 6 (t5-t6): At t5, vs1 drops to zero. The current is1 flows through the body diode of S1 and thus ZVS operation is achieved for S1. In this stage, the magnetizing voltage vLm is clamped to -Vpq1/n1. Similar with stage 4, the leakage inductor current iLr, the auxiliary current ia and the drain-to-source current is1 are obtained in 23~25. From 9 and 16, ia decreases while iLr and is1 increase. The interval t6-t5 is given in 26.

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Stage 7 (t6-t7): At t6, is1 reaches to zero. The operation in this stage is the same as that of stage 6 except that the drain-to-source current is1 is larger than zero. The switching period ends at t7 when the auxiliary current ia decays to zero.

According to Fig. 3(a) and (b), operation of the proposed ZVS magnetic coupling buck/boost converters in boost mode is similar with that of buck mode, and thus its detailed operation is not illustrated in this paper.

IV. PERFORMANCE CHARACTERISTICS

In order to simultaneously obtain the performance characteristics of all proposed ZVS bidirectional buck/boost converters in Fig. 2, generalized analysis based on the general ZVS topology in Fig. 1(b) is performed, which is helpful in choosing the preferred converter according to practical requirement.

4

A. Voltage Transfer RatioOwing to the flux balance, the average primary voltage of

coupled inductor T1 in a switching period is zero. Therefore, with the neglect of switching process, the relationship between V1 and V2 is derived in 27 from Fig. 3(a) and (b). According to 27, the voltage transfer ratio of all proposed converters is the same as the conventional buck/boost converter.

2727\*MERGEFORMAT ()where D is the duty-cycle of switch S1.

B. Turns Ratio n1

From above analysis, 9 and 15 must be satisfied to guarantee normal operation of the proposed converters. From 9 and 15, constraint of turns ratio n1 is obtained in 28, whose minimum value is illustrated in Table 2 for all proposed converters in Fig. 2 with k1~k3 in Table 1. From Table 2, the minimum values of n1 of converters in Fig. 2 (d)~(i), (h) are constant, while it vary with voltages V1 and V2 in the rest converters.

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C. Voltage StressThe turn-off voltage of switches S1 and S2 in the proposed

ZVS magnetic coupling bidirectional buck/boost converters are clamped by V1, which is the same as the conventional bidirectional buck/boost converter. According to 8, 9, 14 and 15, voltage stresses of the auxiliary switches Sa1~Sa2 are obtained in 29, which are also illustrated in Table 2 for all proposed converters in Fig. 2 with k1~k3 in Table 1. With different connections of (p, q), the voltage stresses of Sa1~Sa2

are different due to different values of k1~k3 and turns ratio n1.

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Table 2. Minimum n1 and voltage stress of Sa1~Sa2 of all proposed converters in Fig. 2.

Fig. 2 n1,min Vsa1 Vsa2

(a) V1/V2 -V1+n1V2 n1V2

(b) (V1-V2) /V2 -V1+(n1+1)V2 (n1+1)(V1-V2)(c) V1/V2 -V1+n1V2 (n1+1)V1- n1V2

(d) 0 V1+n1V2 n1(V1-V2)(e) 0 (n1+1)V2 (n1+1)(V1-V2)(f) 0 n1V2 (n1+1)V1- n1V2

(g) 1 V1+(n1-1)V2 (n1-1)(V1-V2)(h) 1 (n1-1)V2 (n1-1)(V1-V2)(i) 1 (n1-1)V2 n1V1-(n1-1)V2

(j) V1/(V1-V2) V1+n1V2 (n1-1)V1- n1V2

(k) V1/(V1-V2) n1V2 (n1-1)V1- n1V2

(l) V2/(V1-V2) (n1+1)V2 n1V1-(n1+1)V2

(m) 0 n1V2 n1(V1-V2)

D. Average Auxiliary Current and Magnetizing CurrentFrom Fig. 3(a), the average auxiliary current Ia,buck in the

buck mode is derived in 30 , combining with 18 and 24. Then according to 4 and 5, the average magnetizing current ILm,buck is obtained in 31. Also, according to Fig. 1 (a), the circulation loss of auxiliary circuit in the buck mode is given in 32.

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3232\*MERGEFORMAT ()Where Vd is the diode forward voltage and Vigbt is the collector-emitter voltage of IGBT

Likewise, the average auxiliary current Ia,boost, average magnetizing current ILm,boost and auxiliary circuit circulation loss Ploss_aux_boost in the boost mode are respectively derived in 33, 34 and 35.

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E. Soft-switching CharacteristicsIn the buck mode, the parasitic capacitor of switch S2 is

discharged by magnetizing current in stage 2, thus ZVS turn-on is easily achieved. In order to eliminate the reverse recovery problem of S2 at turn-off, current is2 should turn from negative to positive at t4 and thus 36 should be satisfied. Meanwhile, according to 20, the value of Vc-A should be less than zero to achieve ZVS operation for switch S1 and the requirement is derived in 37. Moreover, as shown in 38, the dead-time td1 between switches S1 and S2 also must be larger than the maximum value of t5-t4 and the intrinsic turn-off time toff of switches, and must be smaller than the minimum value of t6-t4, otherwise the ZVS operation of S1 would be lost.

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Likewise, in the boost mode, 39, 40 and 41 should be satisfied to achieve soft-switching operation for switches S1~ S2.

5

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From above, soft-switching operation of switches S1 and S2

can be achieved if only 36~41 could be satisfied. Moreover, from Fig. 3(a) and (b), the auxiliary switches Sa1 and Sa2 are ZCS turned off in buck and boost mode because the auxiliary current ia has already decayed to zero before their turn-off. And the increasing ratio of auxiliary current ia is limited by the leakage inductor. Thus ZCS operation is achieved for Sa1~ Sa2.

F. Comparison In the proposed and other ZVS converters in [10, 13, 17, 20,

25], ZVS operation is achieved for the main circuit switches. The circuit in [10] is very simple since no extra components are required. However, the inductor ripple current and conduction losses are large. The filter requirement is lessened

in the active-clamping converter [13], but with the penalty on increased voltage stresses. In addition, the conduction losses are still high in [13], which are reduced in the ZVT converters in [17] and [20] with the auxiliary circuit removing from the main circuit. Moreover, the auxiliary switches in [20] are ZCS. Similar with the ZVT converters, low conduction losses and desirable soft-switching characteristics of both main switches and auxiliary switches are obtained in the ZVS magnetic coupling converters proposed in [25] and in this paper. Besides, magnetic core number is also reduced in comparison with the ZVT converters in [17, 20] because the additional resonant inductor is eliminated. Furthermore, in comparison with [25], the proposed ZVS magnetic coupling bidirectional converters eliminate the two extra diodes and only needs two auxiliary switches, contributing to reduced cost. For ZVT converters in [17, 20] and ZVS magnetic coupling converters in [25] and this paper, comparison results in terms of efficiency, power density, cost and control difficulty are summarized in Table 4. Their efficiency and control difficulty are similar because of the similar performance characteristics. The power density of ZVS magnetic coupling converters in [25] and this paper is higher than ZVT converters in [17, 20] because the number of magnetic core is reduced. Moreover, the cost is reduced in the proposed topologies when compared with the converter proposed in [25] since additional diodes are eliminated.

Table 3. Comparison between the proposed and other ZVS converters.

ConvertersZVS

Realization

Inductor RippleCurrent

Conduction

Losses

MagneticComponen

t

Auxiliary Switches and diodes

number operation

Large inductorripple current [10]

Yes Large High 1 -------- --------

Active-clamping [13] Yes Small High 2 1 ZVS

ZVT [17] Yes Small Low 2 2ZCS turn-on,ZVS turn-off

ZVT [20] Yes Small Low 2 2 ZCSMagnetic coupling

[25]Yes Small Low 1 4 ZCS

Proposed Yes Small Low 1 2 ZCS

Table 4. Comparison results among ZVS bidirectional converters in [17, 20, 25] and this paper.

ConvertersEfficienc

yPower

DensityCost

Control Difficulty

ZVT [17] HighMediu

mHigh Medium

ZVT [20] HighMedium

High Medium

Magnetic coupling [25]

High High High Medium

Proposed High High Medium Medium

Table 5. Comparison of different converters in Fig. 2 under experiment application.

Fig. 2 (a) (b) (c) (d) (e) (f) (g) (h) (i) (j) (k) (l) (m)n1 >2.5 >1.5 >2.5 >0 >0 >0 >1 >1 >1 >2.5 >2.5 >1.5 >0

Vsa1,sa2 — — — >V1,max <V1,max >V1,max >V1,max <V1,max <V1,max — — — <V1,max

Ia, ILm — — — — Small — — Large Large — — — Medium

V.DESIGN CONSIDERATIONS AND EXPERIMENT VERIFICATION

6

A. Design ConsiderationsIn this section, a prototype circuit for the application with

voltages V1=200~300V (V1,min=200V, V1,max=300V) and V2=120V, maximum output power Po,max=500W and switching period Ts=10μs is built. With different connections of auxiliary circuit, the performance characteristics of 13 proposed converters in Fig. 2 are different, which are preferred in different applications with different requirements. In order to choose a preferred converter, performance comparison should be conducted. Thanks to the unified performance analysis in section IV which are suitable for all converters, comparison results can be easily obtained. Firstly, minimum turns ratio of different converters can be derived according to Table 2. Secondly, from 36~41, minimum phase shift φ1 and φ2 as a function of leakage inductance Lr and turns ratio n1 which ensure soft-switching operation over 10%-100% load range in buck and boost modes for different converters in Fig. 2, are derived (Fig. 5 shows the calculation results of converter Fig.2(e) with V1=200V and 300V, as an example). Then, minimum average magnetizing current ILm and average auxiliary current Ia as a function of Lr and n1 at full load conditions in different converters can be obtained from 30~34. (Likewise, Fig. 6 and Fig. 7 respectively show the calculation result of minimum ILm

and Ia in converter Fig. 2(e), as an example) After that, voltage stresses of auxiliary switches in different converters are calculated according to 29. Finally, comparisons in terms of turns ratio n1, voltage stresses of auxiliary switches Vsa1,sa2 and current stresses Ia, ILm, are conducted between different converters to choose an optimal one for the application. The comparison results for the above application are shown in Table 5. From Table 5, turns ratio n1 of converter Fig. 2(a)~(c), (j)~(l) are large, which requires larger magnetic core and results in higher secondary coil resistances as well as conduction losses. Therefore, these six converters are not suitable for the application. Among the other seven converters, voltage stresses Vsa1,sa2 in Fig. 2(d), (f) and (g) are larger than maximum input voltage V1,max, which is not preferred. Finally, current stresses Ia, ILm are compared among the remaining four converters Fig. 2(e), (h), (i) and (m). The converter Fig. 2 (e) is finally chosen in this paper due to its relatively small current stresses.

For converter Fig. 2 (e), k1=0, k2=-1 and k3=1 can be obtained from Table 1. Then restrictions n1>0 is derived from 28, and voltage stresses of auxiliary switches are given in 42 according to 29. From 30 and 33, the average auxiliary current Ia as a function of the leakage inductance Lr and turns ratio n1

in the buck and boost modes is shown in 43. Likewise, the average magnetizing current ILm is also derived in 44 from 31 and 34. In order to achieve ZVS operation, minimum phase shift φ1 and φ2 as a function of Lr and n1 in the converter Fig.2(e) are depicted in Fig. 5 from 36~41. And according to 43 and 44, Fig. 6 and Fig. 7 respectively give the minimum ILm

and Ia as a function of Lr and n1 at full load condition. Combing with ILm in Fig. 6, Ia in Fig. 7, and voltage stresses Vsa1~ Vsa2 in 29, turns ratio and leakage inductance of coupled inductor is designed as 1:n1=1:0.55 and Lr=10μH, which can guarantee ZVS realization and small current/voltage stresses simultaneously. In order to achieve the desired design, ferrite EE42 is used as magnetic core, the turns of primary and

secondary windings are respectively 49 and 27, and the air gap is 0.64mm. Primary/secondary windings enwind in five layers and in a sandwich type. Then φ1=1.8o, φ2=4.68o are decided for V1=200V and φ1=4.68o, φ2=1.8o are decided for V1=300V according to Fig. 5 to achieve soft-switching operation over 10% ~100% load range in both buck and boost modes. And current as well as voltage stresses are obtained from Fig. 6, Fig. 7 and 42 that Mosfet IPW60R190E6 is chosen as main switches while IGBT IKA15N65H5 is selected as auxiliary switches. The reason why auxiliary switches use IGBT instead of Mosfet is that auxiliary switches are ZCS and thus lower switching losses can be achieved. It is noted that IGBT suffers from tailing current process when it is turned off and thus it is not suitable for applications with very narrow duty-cycle as well as very high frequency.

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200V300V

200V300V

(a) (b)Fig. 5. Minimum φ1 and φ2 as a function of leakage inductance Lr and turns ratio n1 to achieve soft-switching operation for buck and boost modes over 10%~100% load range: (a) φ1 and (b) φ2.

I Lm

(A)

200V300V

-ILm

(A)

200V300V

(a) (b)Fig. 6. Minimum average magnetizing current ILm as a function of the leakage inductance Lr and turns ratio n1 at full load conditions under different modes: (a) buck and (b) boost.

7

I a(A

)

200V300V

-Ia(

A)

200V

300V

(a) (b)Fig. 7. Minimum average auxiliary current Ia as a function of the leakage inductance Lr and turns ratio n1 at full load conditions under different modes: (a) buck and (b) boost.

The control diagram is also shown in Fig. 8. Average value of current i2 in Fig. 1(b) is sampled (denoted as I2,samp) and compared with reference current Iref to generate the duty-cycle D1 for switches S1 and S2. From Fig. 8, when I2,samp is smaller than Iref, D1 increases so as to enlarge current i2. According to the above analysis, the required phase shift φ1 in buck mode and φ2 in boost mode depend on the value of voltage V1. Therefore, a look-up table is provided for φ1 and φ2. Then according to the direction of current i2, drive signals of Sa1 and Sa2 can be obtained. When I2,samp is larger than zero, the converter works in buck mode. Then the auxiliary switch Sa1 is turned on φ1 in advance before the turn-off of switch S2 and auxiliary switch Sa2 is always off. The duty-cycle of Sa1 is equal to that of S1. When I2,samp is smaller than zero, the converter works in boost mode. Then the auxiliary switch Sa1

is always off and auxiliary switch Sa2 is turned on φ2 in advance before the turn-off of switch S1. The duty-cycle of Sa2

is equal to that of S2.

Iref+-

vg1

I2,samp

PWMPI vg2

Look-UpTableV1 Modulation

I2,samp vg1 vg2

vgsa1

vgsa2

φ1

φ2

Fig. 8. Control diagram

B. Experiment VerificationThe steady-state waveforms of drive signals, leakage

inductor current and auxiliary switches current at buck and boost modes under full-load condition are shown in Fig. 9, which are in well coincidence with the theoretical analysis. Because the active interval of auxiliary circuit in both buck and boost mode is very short, additional conduction losses are low and thus high efficiency can be achieved. The soft-switching characteristics of switches are illustrated in Fig. 10 ~ Fig. 15.

vgs1(20V/div)

vgs2(20V/div)

vgsa1(20V/div)

iLr(5A/div)

isa1(5A/div) 2μs/div

vgs1(20V/div)

vgs2(20V/div)

vgsa2(20V/div)

iLr(5A/div)

isa2(5A/div) 2μs/div

(a) (b)Fig. 9. Waveforms of drive signals, leakage inductor current and auxiliary switch current at full load condition for different mode: (a) buck and (b) boost.

As shown in Fig. 10, in the buck mode, S1 achieves ZVS operation under both 100% and 10% load conditions since the drain-to-source current is1 is negative and flows through its body diode when it is turned on. From Fig. 11, ZVS operation is also obtained for S2 over wide load conditions. According to Fig. 12, the auxiliary switch Sa1 is ZCS turned off as the current isa1 has already decayed to zero before its turn-off. Therefore, switching losses are significantly reduced over wide load range.

vgs1(20V/div)

vds1(200V/div)

is1(5A/div)ZVS 2μs/div

vgs1(20V/div)

vds1(200V/div)

is1(5A/div)ZVS 2μs/div

(a) (b)Fig. 10. Waveforms of S1 at buck mode under different load conditions: (a) Po,max and (b) 0.1 Po,max.

vgs2(20V/div)

is2(5A/div) RRE

vds2(200V/div)

2μs/div

vgs2(20V/div)

vds2(200V/div)

is2(5A/div)2μs/divRRE

(a) (b)Fig. 11. Waveforms of S2 at buck mode under different load conditions: (a) Po,max and (b) 0.1 Po,max.

vgsa1(20V/div)

vdsa1(200V/div)

isa1(5A/div)ZCS 2μs/div

vgsa1(20V/div)

vdsa1(200V/div)

isa1(5A/div)ZCS 2μs/div

(a) (b)Fig. 12. Waveforms of Sa1 at buck mode under different load conditions: (a) Po,max and (b) 0.1 Po,max.

In the boost mode, ZVS operation and reverse-recovery problem elimination are similarly achieved for S1-S2 and ZCS turn-off is also obtained for the auxiliary switch Sa2 over wide load range as illustrated in Fig. 13 ~ Fig. 15, contributing to reduced switching losses and higher efficiency.

vgs1(20V/div)

vds1(200V/div)

is1(5A/div)

2μs/divRRE

vgs1(20V/div)

vds1(200V/div)

is1(5A/div)

2μs/divRRE (a) (b)Fig. 13. Waveforms of S1 at boost mode under different load conditions: (a) Po,max and (b) 0.1 Po,max.

8

vgs2(20V/div)

vds2(200V/div)

is2(5A/div)

ZVS 2μs/div

vgs2(20V/div)

vds2(200V/div)

is2(5A/div)

ZVS 2μs/div

(a) (b)Fig. 14. Waveforms of S2 at boost mode under different load conditions: (a) Po,max and (b) 0.1 Po,max.

vgsa2(20V/div)

vdsa2(200V/div)

isa2(5A/div)

ZCS 2μs/div

vgsa2(20V/div)

vdsa2(200V/div)

isa2(5A/div)

ZCS 2μs/div

(a) (b)Fig. 15. Waveforms of Sa2 at boost mode under different load conditions: (a) Po,max and (b) 0.1 Po,max.

The measured efficiency of the proposed ZVS magnetic coupling and conventional bidirectional buck/boost converter is shown in Fig. 16. In the conventional bidirectional buck/boost converter, switches also use Mosfet IPW60R190E6 and switching period remains 10μs. From Fig.16, owing to the reduced switching losses, efficiency of the proposed ZVS converter is obviously higher than that of the conventional bidirectional buck/boost converter over the whole load range, which reaches a maximum value of 97.50% and 97.56% in the buck and boost modes, respectively. Photograph of the prototype circuit is shown in Fig. 17.

Eff

icie

ncy(

%)

output power Po(W)

buck modeboost mode

-500 -400 -300 -200 -100 100 200 300 400 500080828486889092949698

100

Proposed

Conventional

Fig. 16. Measured efficiency vs output power in both power directions.

S1&S2

Sa1&Sa2

CoupledInductor

Fig. 17. Photograph of the prototype circuit.

VI.CONCLUSION

With the utilization of leakage inductor to achieve soft-switching operation, reduced magnetic core and high efficiency can be obtained in the proposed ZVS magnetic

coupling bidirectional dc-dc converters. With different auxiliary circuit connections, several different viable topologies were derived, among which an optimal one can be chosen according to the application requirements. In order to simplify characteristic comparison and topology selection, the theoretical analysis was performed based on the general structure, which is effective for all proposed converters. Finally, experimental results of an example ZVS magnetic coupling bidirectional buck/boost converter, were demonstrated to verify the favorable advantages. Soft-switching operation of switches were achieved in both buck and boost modes over wide load ranges, and maximum efficiency of 97.50% and 97.56% were respectively attained.

REFERENCES

[1] M. A. Abusara, J. M. Guerrero, and S. M. Sharkh, “Line-interactive UPS for microgrids,” IEEE Trans. Ind. Electron., vol. 61, no. 3, pp. 1292–1300, Mar. 2014.

[2] D. Han, J. Noppakunkajorn and B. Sarlioglu, "Comprehensive Efficiency, Weight, and Volume Comparison of SiC- and Si-Based Bidirectional DC-DC Converters for Hybrid Electric Vehicles," IEEE Trans. Veh. Technol., vol. 63, no. 7, pp. 3001-3010, Sep. 2014.

[3] H. Ardi, A. Ajami, F. Kardan, and S. Nikpour, "Analysis and Implementation of a Non-Isolated Bidirectional DC-DC Converter with High Voltage Gain," IEEE Trans. Ind. Electron., vol. 63, no. 8, pp. 4878- 4888, Aug, 2016.

[4] H. Wu, K. Sun, L. Chen, L. Zhu, and Y. Xing, "High Step-Up/Step-Down Soft-Switching Bidirectional DC–DC Converter With Coupled-Inductor and Voltage Matching Control for Energy Storage Systems," IEEE Trans. Ind. Electron., vol. 63, no. 5, pp. 2892-2903, May. 2016.

[5] Y. Hsieh, J. Chen, L. Yang, C. Wu, and W. Liu, "High-Conversion-Ratio Bidirectional DC–DC Converter With Coupled Inductor," IEEE Trans. Ind. Electron., vol. 61, no. 1, pp. 210-222, Jan. 2014.

[6] L. Yang and T. Liang, "Analysis and Implementation of a Novel Bidirectional DC – DC Converter," IEEE Trans. Ind. Electron., vol. 59, no. 1, pp. 422-434, Jan. 2012.

[7] S. Dusmez, A. Hasanzadeh and A. Khaligh, "Comparative Analysis of Bidirectional Three-Level DC-DC Converter for Automotive Applications," IEEE Trans. Ind. Electron., vol. 62, no. 5, pp. 3305-3315, May. 2015.

[8] K. Jin, M. Yang, X. Ruan and M. Xu, "Three-Level Bidirectional Converter for Fuel-Cell/Battery Hybrid Power System," IEEE Trans. Ind. Electron., vol. 57, no. 6, pp. 1976-1986, Jun. 2010.

[9] P. J. Grbovic, P. Delarue, P. Le Moigne, and P. Bartholomeus, "A Bidirectional Three-Level DC–DC Converter for the Ultracapacitor Applications," IEEE Trans. Ind. Electron., vol. 57, no. 10, pp. 3415-3430, Oct. 2010.

[10] J. Zhang, J. Lai, R. Kim, and W. Yu, "High-Power Density Design of a Soft-Switching High-Power Bidirectional dc-dc Converter," IEEE Trans. Power Electron., vol. 22, no. 4, pp. 1145-1153, Jul. 2007.

[11] O. Garcia, P. Zumel, A. de Castro, P. Alou, and J. A. Cobos, "Current Self-Balance Mechanism in Multiphase buck Converter," IEEE Trans. Power Electron., vol. 24, no. 6, pp. 1600-1606, Jun. 2009.

[12] J. Baek, W. Choi and B. Cho, "Digital Adaptive Frequency Modulation for Bidirectional DC–DC Converter," IEEE Trans. Ind. Electron., vol. 60, no.11, pp. 5167-5176, Nov. 2013.

[13] P. Dos Santos Garcia Giacomini, J. S. Scholtz and M. Mezaroba, "Step-Up/Step-Down DC–DC ZVS PWM Converter With Active Clamping," IEEE Trans. Ind. Electron., vol. 55, no. 10, pp. 3635-3643, Oct. 2008.

[14] P. Das, B. Laan, S. A. Mousavi, and G. Moschopoulos, "A Nonisolated Bidirectional ZVS-PWM Active Clamped DC–DC Converter," IEEE Trans. Power Electron., vol. 24, no. 2, pp. 553-558, Feb. 2009.

[15] S. Dusmez, A. Khaligh and A. Hasanzadeh, "A Zero-Voltage-Transition Bidirectional DC/DC Converter," IEEE Trans. Ind. Electron., vol. 62, no. 5, pp. 3152-3162, May. 2015.

[16] D. B. Costa and C. M. C. Duarte, “The ZVS-PWM active-clamping CUK converter,” IEEE Trans. Ind. Electron., vol. 51, no. 1, pp. 54–60, Feb. 2004.

9

[17] J.-W. Yang and H.-L. Do, “High-efficiency bidirectional dc–dc converter with low circulating current and ZVS characteristic throughout a full range of loads,” IEEE Trans. Ind. Electron., vol. 61, no. 7, pp. 3248–3256, Jul. 2014.

[18] K.-H. Chao and C.-H. Huang, “Bidirectional dc–dc soft-switching converter for stand-alone photovoltaic power generation systems,” IET Power Electron., vol. 7, no. 6, pp. 1557–1565, Jun. 2014.

[19] I. Lee, J. Kim, T. Lee, Y. Jung, and C. Won, "A new bidirectional DC-DC converter with ZVT switching," in Proc. VPPC, 2012, pp. 684 - 689.

[20] J. H. Lee, D. H. Yu, J. G. Kim, Y. H. Kim, S. C. Shin, D. Y. Jung, Y. C. Jung and C. Y. Won, “Auxiliary switch control of a bidirectional soft-switching dc/dc converter,” IEEE Trans. Power Electron., vol. 28, no. 12, pp. 5446–5457, Dec. 2013.

[21] M. Pavlovsky, G. Guidi, and A. Kawamura, “Buck/boost dc–dc converter topology with soft switching in the whole operating region,” IEEE Trans. Power Electron., vol. 29, no. 2, pp. 851–862, Feb. 2014.

[22] K. In-Dong, P. Seong-Hwan, A. Jin-Woo, N. Eui-Cheol, and K. Jong-Sun, "New Bidirectional ZVS PWM Sepic/Zeta DC-DC Converter," in Proc. IEEE ISIE, 2007, pp. 555–560.

[23] M. Song, Y. Son and K. Lee, "Non-isolated Bidirectional Soft-switching SEPIC/ZETA Converter with Reduced Ripple Currents," J. Power Electron., vol. 14, no. 4, pp. 649-660, Jul. 2014.

[24] E. Adib and H. Farzanehfard, "Zero-Voltage-Transition PWM Converters With Synchronous Rectifier," IEEE Trans. Power Electron., vol. 25, no. 1, pp. 105-110, Jan. 2010.

[25] M. R. Mohammadi and H. Farzanehfard, "New Family of Zero-Voltage-Transition PWM Bidirectional Converters With Coupled Inductors," IEEE Trans. Ind. Electron., vol. 59, no. 2, pp. 912-919, Feb. 2012.

[26] M. R. Mohammadi and H. Farzanehfard, "A New Family of Zero-Voltage-Transition Nonisolated Bidirectional Converters With Simple Auxiliary Circuit," IEEE Trans. Ind. Electron., vol. 63, no. 3, pp. 1519-1527, Mar. 2016.

[27] G. Chen, Y. Deng, Y. Tao, X. He, Y. Wang and Y. Hu, "Topology Derivation and Generalized Analysis of Zero-Voltage-Switching Synchronous DC-DC Converters with Coupled Inductors," IEEE Trans. Ind. Electron., vol. 63, no. 8, pp. 4805-4815, Aug. 2016.

[28] H. Do, "Nonisolated Bidirectional Zero-Voltage-Switching DC–DC Converter," IEEE Trans. Power Electron., vol. 26, pp. 2563-2569, 2011.

[29] J. Lei, C. C. Mi, L. Siqi, Z. Mengyang, Z. Xi, and Y. Chengliang, "A Novel Soft-Switching Bidirectional DC-DC Converter With Coupled Inductors," IEEE Trans. Ind. Appl., vol. 49, no. 6, pp. 2730-2740, Nov /Dec. 2013.

[30] L. Sung-Sae, R. Seung-Wu and M. Gun-Woo, "Coupled Inductor Incorporated boost Half-Bridge Converter With Wide ZVS Operation Range," IEEE Trans. Ind. Electron., vol. 56, no. 7, pp. 2505-2512, Jul. 2009.

[31] C. Hung-Liang and L. Cheng-Wei, "Design and Implementation of a High-Power-Factor LED Driver With Zero-Voltage Switching-On Characteristics," IEEE Trans. Power Electron., vol. 29, no. 9, pp. 4949-4958, Sep. 2014.

[32] W. Li, J. Xiao, J. Wu, J. Liu, and X. He, "Application Summarization of Coupled Inductors in DC/DC Converters," ” in Proc. IEEE Appl. Power Electron. Conf. Expo., Washington DC, 2009, pp. 1487–1491.

[33] Z. Yingqi and P. C. Sen, "A new soft-switching technique for buck, boost, and buck-boost converters," IEEE Trans. Ind. Appl., vol. 39, no. 6, pp. 1775-1782, Nov /Dec. 2003.

[34] G. Chen, Y. Deng, X. He, Y. Wang and J. Zhang, "Zero-voltage-switching Buck Converter with Low-voltage Stress using Coupled Inductor," IET Power Electron., vol. 9, no. 4, pp. 719-727, Apr. 2016.

[35] R. Y. Duan and J. D. Lee, "High-efficiency bidirectional DC-DC converter with coupled inductor," IET Power Electron., vol. 5, no. 1, pp. 115-123, Jan. 2012.

[36] A. Rahimi and M. R. Mohammadi, "Zero-Voltage-Transition Synchronous DC–DC Converters with Coupled Inductors," J. Power Electron., vol. 16, no. 1, pp. 74-83, Jan. 2016.

[37] N. Lakshminarasamma and V. Ramanarayanan, "A Family of Auxiliary Switch ZVS-PWM DC-DC Converters With Coupled Inductor," IEEE Trans. Power Electron., vol. 22, no. 5, pp. 2008-2017, Sep. 2007.

[38] S. L. Sung and M. Gun-Woo, "High-Efficiency Boost Half-Bridge Converter With Fast Boost Current Transferring ZVS Operation," IEEE Trans. Power Electron., vol. 23, no. 4, pp. 1822-1829, Jul. 2008.

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