IEEE TRANSACTIONS ON POWER ELECTRONICS 1

Using Magnetic Regulators for the Optimizationof Universal Ballasts

Marina S. Perdigao, Student Member, IEEE, J. Marcos Alonso, Senior Member, IEEE,Marco A. Dalla Costa, Student Member, IEEE, and Eduardo Sousa Saraiva, Member, IEEE

Abstract—Nowadays, there are few commercial types of ballaststhat are able to control fluorescent lamps with different power rat-ings. One of the main issues is how to get optimum operation foreach lamp in this condition. This paper tries to demonstrate howto accomplish this objective using a magnetic regulator, i.e., usinga variable inductor. This variable inductor is controlled by a dccurrent delivered by a forward converter directly supplied fromthe ballast dc bus voltage. This control current allows changing theresonant tank and adapting it to the working parameters imposedby each lamp, ensuring near-resonance working conditions. Designcriteria and practical verifications are included to confirm this newtechnique. Theoretical predictions are verified with the experimen-tal results for three TLD Philips fluorescent lamps, from 18 to 58 W.

Index Terms—Fluorescent lamps, half-bridge inverter, magneticregulators, resonant tank, universal ballasts, variable inductance.

I. INTRODUCTION

IN LESS than 20 years, electronic ballasts have revolution-ized the way lighting systems are designed and operated.

Since they have become the standard for both new constructionand renovation projects, there are always new demands with re-spect to their working possibilities [1]–[12]. Multiple lamp oper-ation has also become popular [13]–[15]. However, ballasts areusually designed for operating one or several fluorescent lampsof a particular power rating. In a commercial or an industrial flu-orescent system, lamp power typically ranges from 18 to 70 W.Ballast manufacturers usually offer a range of ballasts adaptedfor this particular use. Nonetheless, large infrastructures, suchas airports, shopping centers, train stations, underground rail-way systems, etc., use a large variety of lighting devices withdifferent power ratings. Therefore, an important commercial

Manuscript received March 13, 2008; revised July 14, 2008. This work wassupported in part by the Foundation for Science and Technology, Ministry forScience, Technology and Higher Education (FCT-MCTES), Government of thePortuguese Republic, under Research Grant PTDC/EEA-ENE/66859/2006 andin part by the Government of the Spanish Kingdom, Education and ScienceOffice, under Research Grant DPI-2007-61267. Recommended for publicationby Associate Editor P. Jain.

M. S. Perdigao is with the Instituto de Telecomunicacoes, P-3030-290 Coim-bra, Portugal, also with the Department of Electrical Engineering (DEE), Insti-tuto Superior de Engenharia de Coimbra (ISEC), 3030-199 Coimbra, Portugal,and also with the University of Coimbra, 3030-393 Coimbra, Portugal.

J. M. Alonso is with the Department of Electrical and Electronics Engineer-ing, Universidad de Oviedo, 33204 Gijon, Spain.

M. A. Dalla Costa is with the Universidade de Caxias do Sul, 95020-972Caxias do Sul, Brazil.

E. S. Saraiva is with the Instituto de Telecomunicacoes, Coimbra, Portugal,and also with the Department of Electrical Engineering and Computers (DEEC),University of Coimbra, 3030-393 Coimbra, Portugal.

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2008.2005060

objective is the ability to provide an electronic ballast systemcapable of driving any lamp within a range of nominal powerratings, allowing lighting distributors and fixture manufacturersto stock far fewer types of ballasts in inventory and obviouslyreducing costs.

The availability of low-cost microcontrollers offers, by settingsoftware parameters only, the possibility to adjust automaticallythe ballast output power to a specific lamp, executing variousroutines to start, run, and dim that particular lamp. The sameballast can thereby accommodate a gas discharge lamp within arelatively wide wattage range. Presently, some types of “univer-sal” ballasts, capable of operating as described before, alreadyexist [13], [14]. Usually, the ability to supply different lamps isbased on controlling each lamp current by doing a shift in theoperating frequency. Using the same resonant circuit involves alarge range of operating frequencies, which may not be feasible,due to high switching losses and electromagnetic interference(EMI) stresses. Other techniques suggest adapting the switch-ing frequency and the bulk voltage that supplies the resonantconverter together, or selectively changing simultaneously boththe switching frequency and the duty ratio of the inverter signal.The first one causes an increase of complexity and cost, sinceanother converter with step-up and step-down characteristics isneeded. Also, supplying the lamp with an asymmetrical cur-rent waveform by changing the duty ratio may cause prematureaging of the lamp.

In this paper, some new developments regarding the optimiza-tion of “universal ballasts,” considering the magnetic regulatorconcept, are presented. This application represents an efficientalternative to enhance the performance of these types of ballasts.

II. STANDARD DESIGN ISSUES

Typically, the most common topology used as a high-frequency inverter for electronic ballasts design is the half-bridge inverter connected to a parallel-loaded resonant tank,as presented in Fig. 1. The dc-blocking capacitance is repre-sented in Fig. 1 as CB ∼ ∞. This dc-blocking capacitance ishigh enough so that its ac voltage ripple is negligible, avoidingdc current flowing through the lamp. With this topology, softstarting and safe operation for the lamp can be provided. Thebest method for soft starting is to control the inverter switch-ing frequency so that the lamp voltage and current are alwaysunder control. During the heating process, the operating fre-quency should be adjusted to a value higher than the naturalresonant frequency of the resonant tank. In this way, the heat-ing current can be adjusted to the necessary value, maintainingthe lamp voltage much lower than the starting voltage. After a

0885-8993/$25.00 © 2008 IEEE

2 IEEE TRANSACTIONS ON POWER ELECTRONICS

Fig. 1. Magnetically controlled electronic ballast: half-bridge resonant in-verter with LC tank.

short period, the operating frequency is reduced until the start-ing voltage is obtained, igniting the lamp. The final operatingpoint, at steady-state conditions, is adjusted to be at a switchingfrequency equal to or near the natural frequency, ensuring stableoperation for the lamp.

Like standard electronic ballasts, conventional “universal”ballasts proposed up until now also use the inverter switchingfrequency as the main control parameter, ensuring that eachlamp is operated at nominal power. Moreover, they exhibit lampdetection techniques that identify the lamp power, hence allow-ing operating the inverter at the appropriate switching frequency.After ignition, the lamp voltage falls to a lower operating volt-age. If the lamp is not dimmed, this lamp voltage will remainmore or less constant under steady-state conditions. In [14], acomparison of lamp characteristics of major brand names is pre-sented. It is stated that for all popular standard T8 fluorescentlamps with the same rated power, the lamp voltage as well aslamp currents are almost identical. Yet, for Philips’ lamps of 18and 30 W, the lamp current is the same, so they cannot be dif-ferentiated by means of current sensors. Similar current valuesalso occur for 58- and 70-W lamps from GEC and Thorn, whichleads to the same conclusion. The same does not happen to thelamp voltage. It is demonstrated that for T8 lamps, the largestvoltage variance among different manufacturers is below 3.5%,which represents a theoretically acceptable tolerance. Similarobservations apply to tubular lamps of equal diameter. So, dif-ferent power or types of fluorescent lamps are distinguishableby detection of lamp voltage through a voltage sensor. Nor-mally, after detection, the lamp voltage is then stepped downand modified as a control signal to determine the switching fre-quency of the inverter. In order to select and vary the switchingfrequency, a voltage-controlled oscillator, together with somemapping functions, is used [13].

In another possible arrangement, a microcontroller could beused to detect the lamp voltage and command the ballast to sup-ply each lamp according to its rated power. In this case, the pro-grammable system-on-chip (PSoC) technology from Cypressseems to be a good possibility for providing both flexibility andreliability at a reasonably low cost. PSoC technology incorpo-rates both analog and digital modules together with a micropro-cessor, so that many external circuitries that would be necessary

when using traditional microcontrollers can now be avoided. Inthis way, as shown in Fig. 1, the PSoC device would be used toboth drive the half-bridge inverter and the forward converter viadigital pulsewidth modulation. Besides, the closed-loop opera-tion of the power converters could be incorporated in the PSoCdevice by using an analog-to-digital (A/D) converter and digitalregulator implemented in the circuit microprocessor circuit. Inorder to detect the lamp type and provide lamp power regulation,lamp voltage and current can be measured by using the othertwo A/D converters available inside the PSoC device.

Obviously, the design methodology of the resonant invertercan be very different depending on the lamp type and character-istics, inverter topology, and design goals. Since each lamp hasits own starting and working voltage, using the same resonanttank with the switching frequency as the only control variableinvolves a large range of operating frequencies and does notensure optimum working conditions.

To overcome these limitations, it is suggested in [14] to adaptthe bulk voltage that supplies the resonant converter to eachattached lamp. Another converter with step-up and step-downcharacteristics is needed to perform this operation. This con-verter must handle the total circuit power, thus decreasing theoverall efficiency. Another alternative presented in [15] con-siders selectively changing simultaneously both the switchingfrequency and the duty ratio of the inverter signal. Varying theduty ratio of the switches allows controlling the width of thevoltage waveform at the output of the inverter, which meanscontrolling its rms value and, consequently, providing a currentwith variable rms to each lamp. The main disadvantage of thiscontrol method resides in the eventual asymmetrical lamp cur-rent waveform, which causes the premature aging of the lampdue to a mechanism called nonlinear electrophoresis.

This paper presents an alternative technique that uses themagnetic regulator concept. One of its possible applicationswas already exposed in previous presented studies [16]–[20],as one way of improving the performance of dimmable elec-tronic ballasts. Here, near-resonance working conditions can beplanned for each attached lamp if the typical resonant induc-tor is replaced by the variable inductor, as shown in Fig. 1.Using magnetically controlled ballasts, it is possible to obtainone additional DOF that can be used to optimize the workingconditions for each lamp [21].

III. DESIGN PROCEDURE

The purpose of this implementation is to present an electronicballast system capable of driving three lamps within a rangeof nominal power ratings, guaranteeing stable and optimumoperation for each lamp.

The variable inductor used in this application is the core ofthis new technique. This regulator, as shown in Fig. 1, consists ofone main winding, which corresponds to the resonant inductor,and two bias dc windings [22]. A dc current circulating throughthe bias dc windings allows modifying the inductance of themain winding; the more the dc current, the less the inductance.By controlling this dc current, the ability to change and adaptthe resonant tank to the working parameters imposed by each

PERDIGAO et al.: USING MAGNETIC REGULATORS FOR THE OPTIMIZATION OF UNIVERSAL BALLASTS 3

Fig. 2. Design procedure for obtaining the suitable working parameters foreach lamp.

TABLE ILAMP CHARACTERISTICS AT HIGH FREQUENCY

lamp is granted. Furthermore, near-resonant working conditionscan be expected for each lamp.

The selected lamps are three TLD lamps from Philips, withdifferent power ratings: 18, 36, and 58 W. The flowchart of thisdesign procedure is presented in Fig. 2. The design procedurebegins by obtaining each lamp’s electrical parameters at nominalpower, as shown in Table I. After that, one of these lamps isselected as the reference lamp.

For the half-bridge inverter connected to a parallel-loadedresonant tank, the maximum voltage gain can be approximatedby the Q at the natural frequency, Q being the normalized load,as represented in (2) [23]. Using the input voltage given by (1)and the lamp voltage and current, from (2), the base impedance

TABLE IIRESONANT TANK PARAMETERS FOR THE DIFFERENT LAMPS

Zb can be calculated. Subsequently, the desired reference lampoperating frequency is selected. Afterward, the values for in-ductance and capacitance of the resonant tank are calculatedusing (4). For the rest of the lamps, the inductance is a controlparameter that can be modified, but the resonant capacitancemust remain the same. With the capacitance value obtained inthe first iteration, and using an iterative process, the new valuesfor the inductance and resonance frequencies, fr , for the otherlamps are also calculated. This procedure is carried out untilall the lamps are analyzed. During the design procedure, onemust ensure that the obtained frequencies are in an acceptableworking range, normally between 45 and 100 kHz. If not, theprocedure should be repeated.

For example, choosing the 58-W lamp as the reference lamp,the normalized load and the base impedance are calculated us-ing the nominal lamp characteristics presented in Table I. Ifthis is the first iteration, the operating frequency must be se-lected, for instance at 100 kHz. The values for L and C are thencalculated, resulting in 0.37 mH and 6.4 nF, respectively. Theclosest available commercial value for this capacitor is 6.8 nF.Maintaining this value for the next iterations with the remaininglamps, the process is repeated. The normalized load and baseimpedance are calculated, and finally, the resonance frequencyand the inductance value for each lamp are determined.

Following previous procedure, the calculated values for eachlamp are shown in Table II, considering a bulk voltage, VDC , of310 V. With this technique, near-resonance operation for eachlamp can be obtained

Vin =4 × (VDC/2)

π ×√

2(1)

Q =Vlamp

Vin=

Rlamp

Zb(2)

Zb =

√L

C(3)

L =Zb

2πfr, C =

12πZbfr

. (4)

IV. VARIABLE INDUCTOR

The regulator is implemented using a double E core, as shownin the schematic of Fig. 3 [23]. The basic idea of this device is tocontrol the main inductance value by establishing different per-meance regions in the core. In order to implement this regulator,two equal bias windings N1 and N2 are mounted in the lateralarms of the core, which generate dc magnetomotive forces ϕ1and ϕ2 . An additional winding N3 that determines the ac induc-tance Lac is placed in the air-gapped middle arm of the core. Adc control current circulating through the bias windings allows

4 IEEE TRANSACTIONS ON POWER ELECTRONICS

Fig. 3. Magnetic structure of the variable inductor: definitions of fluxes ϕ1and ϕ2 due to the dc bias current and ϕ3 due to ac excitation of the center leg.

the modification of the Lac inductance associated to the mag-netic flux ϕ3 established in the center arm. In order to cancel theac voltage induced on the dc control circuit due to ϕ1 and ϕ2 , thebias windings must be serially connected in opposite polarity.Even though assuming symmetrical bias windings, a nonlinearcoupling between the ac winding and the bias windings remains,due to the nonlinearity of the magnetic material [22]. The in-jected bias current will produce a dc magnetic flux in the outerarms that will bring these sections into the nonlinear region ofthe B(H), i.e., it pushes the operating point on the B(H) curvetoward saturation [24]. This device gives the ability to changethe Lac inductance value as required by the working parametersfor each lamp, thus permitting an optimum operation for theballast.

The regulator design was done considering that it should havean inductance variation between 1 and 0.25 mH. The prototypewas built using an EF25 core with 3C85 material from Phillips.The following design method is applied to determine the numberof turns of each winding and the airgap value. The maximuminductance value will be obtained for a zero value of the dccontrol current. Using the basic equation of an inductor, theanalytical expression for the ac inductance can be determinedas in (5)

Lacmax =N 2

3

R=

N 23

�e/µAe + g/µ0Ae(5)

where N is the number of turns in the central arm, R is thereluctance of the magnetic circuit, Ae is the effective magneticcross section, �e is the effective magnetic path length, g is theair gap, and µ and µ0 are the magnetic permeabilities of thecore material and vacuum. A technical aspect relates to the factthat the ac component of the magnetic flux density B must besmaller than the saturation value, which represents the maximumattainable flux density. Typically

B =BSAT

10. (6)

Thereby, for a zero value of the control current, the ac com-ponent of the magnetic flux density can be calculated, using the

Fig. 4. Variable inductor prototype. (a) Core and auxiliary windings. (b) Finalappearance of the variable inductor.

ac value of the inductance current, I , as follows

B =N3

RAeI. (7)

Solving the system formed by (5) and (7), the number of turnsof the main winding and the airgap value can be determined.Finally, the number of turns of each control winding, Ncontrol ,can be estimated, considering that the external path of the EF25core must reach saturation, for a stipulated maximum value ofthe dc control current

Ncontrol =12

BSAT�ext

µIdc max(8)

where �ext is the length of the external path of the core andIdc max is the maximum value of the dc control current forwhich the minimum inductance value is obtained.

The preceding equations represent a brief summary of themagnetic regulator design method. This is, in fact, a simplifiedand approximated method. For increasing the performance ofthe magnetically controlled electronic ballast, it is important tounderstand the real behavior of this magnetic element. How-ever, its theoretical analysis is a nontrivial task, and in order tocarry out a more accurate design of the regulator, one must com-prehend the theoretical and practical aspects of this nonlinearcontrollable reactor [22]. The authors are currently working onthis subject.

According to the design estimation, the bias windings shouldhave each 35 turns and the main winding 68 turns. An airgap of0.3 mm was calculated. Fig. 4 shows the built prototype.

The characterization of the magnetic regulator was done usinga variable dc source and an impedance analyzer, as representedin Fig. 5. Each level of dc voltage corresponds to a differentlevel of dc current in the control winding, which gives a specificinductance value. Table III presents the obtained values for thedc control current and inductance considering several operatingpoints.

It must be noted that the measurements shown in Table IIIhave been taken under small-signal conditions for the ac wind-ing. However, in normal operation, as part of the resonant tankof the electronic ballast, the inductor will be operating withlarger current values that could lead to changes in the inductancevalue [24]. In order to evaluate the inductance variation underlarge-signal operation, the inductance value can be measuredusing the same electronic ballast arrangement and measuringthe power delivered to a resistive load. This has been made by

PERDIGAO et al.: USING MAGNETIC REGULATORS FOR THE OPTIMIZATION OF UNIVERSAL BALLASTS 5

Fig. 5. Characterization of the variable inductor.

TABLE IIIVARIABLE INDUCTOR CHARACTERISTICS

the authors in a previous study [25] and experiments showedthat the deviation is approximately 10% from the small-signalvalue.

Regarding the selection of the core material, a soft saturationmaterial would be preferred. Ferrites have high permeability,and thus, can be used to generate high inductance, and thereexists a variety of ferrites for minimal power dissipation in var-ious frequency bands. Yet, ferrite cores saturate rather abruptly;moreover, the saturation flux density is a function of tempera-ture. For molyperm (MPP) cores, the reduction in permeabilityas a function of flux is very gradual, so MPP is a soft saturationmaterial. Nevertheless, one of the problems with MPP coresis that they have much higher losses than ferrites. Powderediron cores saturate slightly harder than MPP, and while a vari-ety of permeabilities are available, they are typically lower thanMPPs. This means a powered iron regulator would be largerthan a device having the same inductance and current capacitybut built on an MPP core. Nevertheless, powdered iron coresare cheaper than molyperm cores. So, in fact, all of this must beconsidered when choosing the core material. The authors haveinitially opted for ferrites since they are made by a wide vari-ety of vendors, in the necessary shapes, gapped E core, whichwas not the case for the other materials. Even so, other variableinductor structures are currently being investigated in order totest these other materials.

In order to evaluate the power handled by the bias windingsat each dc current level, the measurement of its voltage andresistance was also done. Those measures are also presented inTable III. The control winding must be supplied by means of asmall dc-to-dc converter supplied from the ballast bus voltage. Itshould be noted that this converter only handles the bias windingpower, and that ideally this power would be null. Nevertheless,

Fig. 6. Variable inductor characterization. (a) Experimental value of the in-ductance as a function of the dc control current. (b) Power handled by the controlwinding as a function of the dc control current.

Fig. 7. Forward converter used for supplying the dc control current to thevariable inductor.

even though low power levels are handled by the control circuit,a higher efficiency for the entire ballast circuit may be obtained.In fact, from the analysis of Table III, it is observed that verylow power levels are managed by the control winding. In reality,this power does not exceed 1 W.

Fig. 6(a) shows the experimental value of the regulator asa function of the dc control current, and Fig 6(b) the powerhandled by the control winding of the regulator as a function ofthe dc control current.

In Fig. 7, a forward converter has been selected for supplyingthe dc control current. The forward converter will be operated in

6 IEEE TRANSACTIONS ON POWER ELECTRONICS

TABLE IVDESIGN PARAMETERS OF THE FORWARD CONVERTER

continuous current mode [26]. The turns ratio Nd/Np limits thepeak voltage seen by the switch. With an equal number of turnsfor the primary and the demagnetizing windings (Np = Nd ),the maximum duty cycle is limited to 0.5. Under this condition,the switch must block 2VDC during turn-off. After taking intoaccount switching delays, this duty cycle falls to 0.45.

The LC filter is calculated so that a very low ripple in theoutput current is assured. This is achieved by using high valuesof both the inductor and capacitor, and operating the forwardconverter in closed loop by regulating the dc output current.In this prototype, an inductance of 200 µH and a capacitor of220 µF have been used and a current ripple lower than 20 mAwas attained.

The design of the forward converter was done consideringthat it should have the following specifications: VDC = 310 V,D = 0.3, Idc = 1.5 A. The relation between the windings,Ns/Np , was calculated to be approximately 0.017. The designparameters are presented in Table IV.

The output of the forward converter is regulated by meansof a feedback control, which employs a PWM controller, theSG3524. This type of control allows the adjustment of the out-put current, giving an adequate value for the variable inductor,according to the working parameters imposed by each lamp.

Fig. 7 shows that the output dc current of the converter issensed, through Rs , and then amplified, using the LM358 opera-tional amplifier. This output signal is fed to the transconductanceerror amplifier of the SG3524. Theoretically, a transconductanceamplifier is an equivalent voltage-controlled current source. Itmultiplies the difference of input voltage by a certain gain andgenerates a current into the output node [27]. So, this error am-plifier produces an error feedback signal, which is proportionalto the difference between the reference input voltage and theamplified sensed current. Its compensation network was cho-sen to be a shunt capacitor, acting as an integrator, since thereis no need of a fast response. Finally, the sawtooth waveformis then compared with the error amplifier output in an internalcomparator at the selected switching frequency, to determinethe duty cycle of the switch.

V. EXPERIMENTAL RESULTS

The electronic ballast prototype was implemented as shownin Fig. 8. From this figure, it is clearly shown that the controlwinding of the variable inductor is supplied by means of theforward converter supplied directly from the ballast bus volt-age. The lamp detection and control circuit detects which lampis present and gives its working parameters: inverter frequencyand inductance values. For instance, as an example representedin Fig. 8, it gives the adequate value for the reference voltagein order to supply the correct dc control current, to obtain theright inductance value for the resonant tank. Taking into account

Fig. 8. Electronic ballast circuit prototype.

its working parameters, each lamp was started as follows: theoperating frequency of the inverter was set at the previously cal-culated value. Then, the dc control current was adjusted in orderto get the maximum inductance value. Afterward, the value ofthe inductance was lowered, by increasing the dc control cur-rent, until the lamp was ignited. Then, the frequency was finallyadjusted in order to get near-resonance working conditions andstable operation.

Other important issues deal with the power control of theelectronic ballast: First, it implies the study of the phase angleof the input current of the resonant tank for each case. It is wellknown that a delayed current must be used in order to achievezero-voltage switching (ZVS) in the bridge transistors. In orderto perform this analysis, the phase angle of the input resonanttank current was also verified. Besides, this phase analysis isimportant to ensure near-resonance conditions. At the moment,the prototype works in open loop, which means that after theµP detection of the lamp power and setting of the lamp workingparameters, the phase angle of the input resonant tank currentis not verified by the system; only experimental verificationwas performed. Once the inductance has been adjusted for eachlamp, there is no control loop to assure near-resonance working.This mode of operation is similar to the one found in typicalcommercial ballasts where, usually, there is no phase detectionto adjust the operation point. The inclusion of phase detectionin this prototype is also possible but it would increase cost andcomplicate the system. Experimental study has shown that there

PERDIGAO et al.: USING MAGNETIC REGULATORS FOR THE OPTIMIZATION OF UNIVERSAL BALLASTS 7

Fig. 9. Experimental results for the TLD 18 W. (a) Waveforms of the lampvoltage and resonant current (50 V/div, 1 A/div, 5 µs/div). (b) Waveforms of theinverter output voltage and resonant current (100 V/div, 1 A/div, 5 µs/div).

is no excessive variation of inductance in the variable inductorprovided that the dc current is regulated, which leads to theconclusion that there is no need for a closed-loop operation. Onthe other hand, the microcontroller will perform a closed-loopoperation of the lamp power by measuring the lamp voltage andcurrent. This would also compensate possible perturbations inthe circuit parameters.

Figs. 9–11 show the obtained experimental results for thelaboratory prototype, for the three different fluorescent lamps,respectively, 18 W, 36 W, and 58 W. In Figs. 9(a), 10(a), and11(a) the resonant current and lamp voltage waveforms for eachlamp are shown. Figs. 9(b), 10(b), and 11(b) present the wave-forms of the inverter output voltage and the input resonant tankcurrent for each lamp. The measurements performed were lamppower, which means arc power plus electrode power, the bal-last input power, and the dc value of the control current. Thesemeasurements are presented in Table V.

As an initial analysis, stable operation for each lamp can berecognized from Figs. 9(a), 10(a), and 11(a). This was the mainobjective of this implementation. For the 18-W lamp, the for-ward converter was OFF, since the necessary inductance value

Fig. 10. Experimental results for the TLD 36 W. (a) Waveforms of the lampvoltage and resonant current. (b) Waveforms of the inverter output voltage andresonant current (100 V/div, 1 A/div, 5 µs/div).

was maximum. For the 58-W lamp, some distortion can be ob-served in the resonant current; nevertheless, this did not resultin any working instability. Table V shows the dc current levelin the control winding for each lamp. The obtained current lev-els comply with the estimated working parameters in terms ofinductance value, guaranteeing near-resonance working condi-tions for each lamp. A graphical view of the experimental re-sults is presented in Fig. 12. The obtained operating frequenciesagree with the calculated values. The upper traces in Figs. 9(a),10(a), and 11(a) correspond to the product of the lamp voltagewaveform and the resonant current waveform; in fact, these up-per traces represent the instantaneous lamp power. Similarly, forFigs. 9(b), 10(b), and 11(b), the upper traces correspond to theinstantaneous power delivered by the inverter.

From the analysis of Figs. 9(b), 10(b), and 11(b), whichpresent the waveforms of the inverter output voltage and the in-put resonant tank current for all three lamps, it can be observedthat the current lags the voltage, thus ensuring zero-voltageswitching (ZVS) in the bridge transistors. The second issue isthe analysis of the overall efficiency. In Table V, the calculatedefficiency is shown as the ratio between the lamp power and the

8 IEEE TRANSACTIONS ON POWER ELECTRONICS

Fig. 11. Experimental results for the TLD 58 W. (a) Waveforms of the lampvoltage and resonant current. (b) Waveforms of the inverter output voltage andresonant current (100 V/div, 1 A/div, 5 µs/div).

TABLE VLAMP POWER MEASUREMENTS

Fig. 12. Experimental results for the operating points of each lamp, in termsof inductance value and control current.

input power. It is noted that this measured input power includesalso the power absorbed by the forward converter. From theanalysis of Table V, it can be concluded that high efficiency isguaranteed with this technique.

VI. CONCLUSION

A recent domain in ballasts applications seems to be the de-velopment of “universal” ballasts, able to detect and operatelamps with different power ratings. In this paper, it was demon-strated that the use of magnetically controlled electronic ballastsallows changing and adapting the resonant tank to the workingparameters imposed by each lamp. This can be accomplishedusing a variable inductor together with different operating fre-quencies. With this new technique, it is not necessary to adaptthe dc bus voltage, neither changing simultaneously both theswitching frequency and duty ratio of the inverter control sig-nal, with their respective disadvantages. It was also shown thatthe control winding of the variable inductor can be suppliedby means of a forward converter powered from the ballast busvoltage. Since this converter only manages low power levels,high efficiency for the complete system is obtained. With thisnew technique, besides getting soft lamp starting, it is allowedto operate each lamp near resonance, guaranteeing stable andoptimum operation.

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Marina S. Perdigao (S’06) was born in Coimbra,Portugal, in 1978. She received the M.Sc. degree inelectrical engineering in 2004 from the University ofCoimbra, Coimbra, Portugal, where she is currentlyworking toward the Ph.D. degree.

Since 2001, she has been a Researcher at the In-stituto de Telecomunicacoes, Coimbra. Since 2002,she has also been an Assistant Professor in the De-partment of Electrical Engineering, Instituto Supe-rior de Engenharia de Coimbra, Coimbra. Her currentresearch interests include high-frequency electronic

ballasts, discharge lamp modeling, power-factor-correction topologies, high-frequency switching converters, and computer simulation applications.

Ms. Perdigao is a member of the Order of Engineers of Portugal.

J. Marcos Alonso (S’94–M’98–SM’03) received theM.Sc. and Ph.D. degrees in electrical engineeringfrom the Universidad de Oviedo, Gijon, Spain, in1990 and 1994, respectively.

From 1990 to 1999, he was an Assistant Professorin the Department of Electrical and Electronics En-gineering, Universidad de Oviedo, where he was anAssociate Professor from 1999 to 2007, and has beena Full Professor since October 2007. He is the pri-mary author of more than 60 journal and internationalconference papers in power and industrial electron-

ics and a coauthor of more than 130 papers. He holds four Spanish patents,

with two under review. His current research interests include high-frequencyelectronic ballasts, discharge lamp modeling, power converters for ozone gen-eration, power converters for electrostatic applications, power-factor-correctiontopologies, and high-frequency switching converters.

Prof. Alonso is a member of the International Ozone Association. He hasbeen an Associate Editor of the IEEE TRANSACTIONS ON POWER ELECTRONICS

in the field of lighting applications since October 2002. He was also a GuestEditor for the IEEE TRANSACTIONS ON POWER ELECTRONICS Special Issue onLighting Applications published in May 2007. He usually collaborates as atransactions paper Reviewer and as the Conference Session Chairman, amongother positions, in the IEEE. He was the recipient of numerous awards, includ-ing the Early Career Award of the IEEE Industrial Electronics Society in 2006,the Second Prize Paper Award from the Production and Application of LightCommittee of the 2005 IEEE Industry Applications Society Meeting, and theIEEE Industrial Electronics Society Meritorious Paper Award in 1996.

Marco A. Dalla Costa (S’03) was born in SantaMaria, Brazil, in 1978. He received the B.S. andM.Sc. degrees in electrical engineering from the Fed-eral University of Santa Maria, Santa Maria, in 2002and 2004, respectively, and the Ph.D. degree from theUniversity of Oviedo, Gijon, Spain, in 2008.

Since 2008, he has been an Associate Professorat the Universidade de Caxias do Sul, Caxias do Sul,Brazil. He is currently a Researcher and is engagedin the development of electronic systems for lightingand high-intensity discharge (HID) lamp modeling.

His current research interests include dc/dc converters, power-factor correctionstages, dimming systems, high-frequency electronic ballasts, discharge lampmodeling, and electronic starters for HID lamps.

Eduardo Sousa Saraiva (S’78–M’78) received theDegree in electrical engineering from the Universityof Oporto, Oporto, Portugal, in 1970, the Ph.D. de-gree in electrical engineering from the University ofLondon, London, U.K., in 1979, and the Aggrega-tion in electrical engineering from the University ofCoimbra, Coimbra, Portugal, in 1985.

From July 1973 to July 1975, he was an AssistantEventual with the Faculty of Science and Technology,University of Coimbra, where he was an Assistantfrom July 1975 to July 1979, an Auxiliary Professor

from July 1979 to November 1979, an Associate Professor from December1979 to November 1986, and has been a Full Professor since November 1986.He is the author or coauthor of about a hundred papers in national and interna-tional journals and conferences. His current research interests include electricmachines, power electronics, the influence of the electromagnetic field in thehuman body, and more recently, electronic ballasts for fluorescent lamps.

Prof. Saraiva is a member of the Portuguese Ordem dos Engenheiros, andwas elected for positions in the Central Region and in National Boards. He isone of the founders of the IEEE Portugal Section. He usually collaborates as aReviewer for the IEEE.