The Application of the Half Bridge Dual Converter to Photovoltaic

7/30/2019 The Application of the Half Bridge Dual Converter to Photovoltaic

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THE APPLICATION OF THE HALF BRIDGE DUAL CONVERTER TO PHOTOVOLTAIC

APPLICATIONS

Quan Li, Peter Wolfs and Steven Senini

Central Queensland UniversityAbstract

The half bridge dual converter is a current sourced converter topology that has been previouslydeveloped from the half bridge converter. This paper examines some of the unique properties ofthis converter which make it suitable for lower voltage higher current applications. Theseconditions often arise in photovoltaic applications. An example of the use of this topology as an85W, low voltage photovoltaic input, 360Vdc output converter for use in an inverter application is

presented.

1. INTRODUCTIONPhotovoltaics (PVs) are currently being produced at

approximately 200 MWp annually, as impressive asthe figure may sound, momentary reflection showsthat this is still a small part of the global energy need.Fortunately photovoltaic usage continues to growrapidly. If the current growth rates of 20% and betterannually continue, photovoltaics will be a significantenergy source within a few decades, [1].

While the traditional markets of remote area power(RAPS) and telecommunications remain important, anew area of growth is grid interactive PV, [2]. Whilelarge centralised systems have been constructed,smaller systems appear to be growing in importance.

These tend to be modestly sized arrays, typically a fewkilowatts that generate directly into the power system.A recent local example is the solar school program,

promoted by Stanwell Corporation that aims to place5kW grid interactive arrays into 200 Queenslandschools.

Smaller grid interactive systems can make use of atleast two technologies:

Arrays with medium string voltages and gridinteractive inverters in the low kilowatt range

Panel sized inverters, or module integratedconverters/inverters (MICs).

The panel and module integrated inverters have subkilowatt ratings, typically 50-200W, and convert theoutput of a module directly into AC. A grid interactivesystem of any size is developed by simply connectingthe integrated modules and inverters to the ACsystem, [3].

While this approach brings the benefits of a simplifiedmodular design, the designer of the module integratedinverter faces significant challenges in developing a

small rating, but high efficiency, high reliability, cost

effective inverter design. To achieve a compact highefficiency design, many designers opt to use a DC-DCconverter to translate the output voltage of a module,which is typically between 12Vdc and perhaps 48Vdc,to a level that allows direct inversion to produce AC,typically 360Vdc. The DC-DC converter represents asignificant design challenge, especially in regard toobtaining efficiencies in the mid to high 90% range.

This paper will examine a DC-DC converter topologythat is expected to offer some advantages in thisapplication. Encouraging experimental results have

been achieved with a hard-switched 100kHz versionof the converter. Several opportunities for furtherdevelopment are highlighted.

2. THE HALF BRIDGE DUAL CONVERTERThe half bridge dual converter is a current sourcedconverter topology which was developed from themore conventional voltage sourced half bridgeconverter using duality theory, [4]. Figures 1 and 2respectively show the half bridge converter and itsdual topology.

E/2

E

E/2

Q1

Q2

C

C

R V

Figure 1. The Half Bridge Converter

As the half bridge dual converter has been developedusing duality theory, every attribute of the half bridgeconverter exists, but in a dual form in its descendent,[4]. Some key points are:

The half bridge converter uses two capacitors todivide the supply voltage into two. Alternate


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closures of the switches apply either +E/2 or E/2across the transformer primary. The dualconverter uses two inductors to divide theavailable supply current into two. Alternateswitch openings direct either +I/2 or I/2 to flow

through the primary.

At least one switch is open at all times in the halfbridge coverter, in its dual at least one switch isclosed.

The closure of both switches is destructive for thehalf bridge, causing a short circuitof the voltage

source input, in the dual converter opening bothswitches is destructive causing a open circuit ofthe input current source.

The half bridge is buckderived, the dual is boost

derived.

The output rectifier stage in the half bridgeconverter uses a current stiff inductive filter, thedual filter uses a voltage stiff capacitive filter.

While the half bridge converter is suited to higherDC bus voltages, the dual converter is suited tohigher DC bus currents.

I

I/2 I/2L L

Q1 Q2

R V

Figure 2. The Half Bridge Dual Converter

The last point is of significance in photovoltaicapplications where the voltages tend to be low. Animportant implementation point is that it is generally

possible to replace the input current source with a

voltage source in a practical application. The twoinductors act to make a voltage source appear currentstiff.

3. COMPETING TECHNOLOGIESUntil now, the reflexive design choice for a 50-100WDC-DC converter, operating from a low voltage

photovoltaic source would have been the centre tappedpush-pull converter as shown in Figure 3.

EQ1 Q2R V

Figure 3. The Push-Pull Converter

The relative advantages offered by the dual converterinclude lower primary side average and RMS switchcurrents and less stress for the output rectifiers.

For the dual converter, the primary switches carryeither the input current I or I/2 when both devicesoverlap. The transformer turns ratio is selected to give50% duty cycle at the highest input voltage. The boostnature of the dual converter allows lower voltage

inputs to be covered by increasing the Mosfet overlaptime. The peak device voltage stress is twice themaximum input voltage.

In the push-pull circuit the transformer turns rationeeds to be selected to cover the lowest allowableinput voltage. Under normal conditions the Mosfetsmust work at less than 50% duty cycle, this raises the

peak current further increasing the switch currentratings. A two to one input voltage range, for exampleresults in peak stresses that are twice the average inputcurrent. The voltage stresses are equivalent to the dualconverter case, twice the maximum input voltage.

The output rectifiers of the push-pull converter arestressed by two factors, the high turns ratio needed if awide input voltage range must be designed for, andreverse recovery. The output rectifiers for the dualconverter work into a capacitive filter, this clamps thereverse voltage to 360Vp, under all conditions. Noadditional snubbers are required to cope with thereverse recovery of the rectifier diodes.

A bonus is that the dual design typically allows, fasterrecovery 600V devices to be used while a push-pulldesign typically requires slower 1000V devices, once

allowances for voltage regulation and reverse recoverysnubbers are made. The 1000V devices bringadditional penalties in switching loss and forwardconduction drop.

A disadvantage suffered by the dual converter is ahigher sensitivity to transformer primary to secondaryleakage inductance. The energy stored in thisinductance is dissipated in hard-switched versions ofthe converter. Push-pull converters are not completelyimmune however. Successful operation does requiretightly coupled primary windings.


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4. HARD SWITCHED CONVERTERSIMULATION RESULTS

In order to provide a starting point, this paperconsiders the design optimisation of a 100kHz hard

switched converter intended to be used with acustomised solar module using 44 BP solar cells andhaving a nominal 85W, 17Vdc, 5Adc rating. Theintention is to progressively examine soft switchingtechniques for this converter and raise both efficiencyand switching frequency.

Figure 4 shows the actual experimental circuit. Thesimulation is performed with SIMULINK and is basedon an idealised version of the experimental circuit.The device models do not include parasiticcomponents and the primary side voltage limitingsnubber, consisting of two diodes, a capacitor and

zener diode is omitted from the simulation model.

176H

Q1 Q2

R

V17V

176H

1F

20V

Figure 4. Experimental Circuit

The results of the simulation are shown in Figure 5.The leading edges of the Mosfet current waveformsshown in Figure 5 part (a) sometimes show a transientcurrent spike. This is an artefact of the simulationmodels used in the SIMULINK package. Aninteresting result is that the Mosfet peak currents showvery little ripple, in comparison to the primarycurrents. It can be shown that the 50% duty ratio caseresults in zero input current ripple for this converter.The output diode voltage waveforms are exceptionallyclean, no reverse recovery voltage is seen.

5. EXPERIMENTAL RESULTSFigure 6 contains the experimental waveforms thatmatch the simulated results. Good agreement isobtained at the cycle by cycle time scale. As expectedsmall variations exist at the switching time scale as thesimulation model does not yet include detailedswitching models. The Mosfet current, part (a)confirms that the leading edge spike noted in thesimulation is indeed an artefact. The current edge iscomplicated, however by oscillatory currents relatedto the device capacitances. Figure 6 part (b) shows theMosfet turn off voltage. This is limited by the primary

side snubber network that absorbs the leakage

inductance energy. A relatively low level oftransformer leakage inductance, 0.5H referred to the

primary side, was obtained by interleaving the primaryand secondary windings, [5]. The losses in the snubberare a concern and some lossless solutions will be

examined later. However for this topology the lossescould be reduced by raising the clamping voltage butthis would require a higher Mosfet drain rating. Forthe same die area this would cause some additionalconduction loss due to higher Rdson. Temperature rise

based estimates place the snubber losses at1.5W

Figure 6 part(c) shows the primary current, during theoverlap times, when both Mosfets conduct, someoscillations related to the device capacitances andtransformer leakage inductance are observed.Ultimately we hope to develop lossless snubbernetwork, or resonant switching methods that will

make better use of both of these components. Figure 6part (e) confirms that the output diodes are free fromany over voltages due to reverse recovery.

As the hard switched converter was intended to be aforerunner for higher frequency, low loss converters,significant effort was put into identifying thesignificant sources of loss. The converter was operatedat slightly above rated load, with an input power of97W and an output power of 90W. The measured loss,7W, is in good agreement with the total thermal lossestimate of 6.5W. The measured efficiency was 93%under these conditions. Table 1 contains a list of the

losses estimated by allowing the converter to reachthermal equilibrium and observing temperature rises.

Ultimately, however the target efficiency for thisapplication must be in the high ninety percent range.Further increases in efficiency can be gained in thesnubber network. Elimination of the loss would raisethe converter efficiency to above 94%. The othermajor loss areas are the magnetic components. Someimprovement can be gained by selecting a betterferrite grade. The transformer used Philips 3C80, [6],significantly better grades exist.

Component TemperatureRise (K) EstimatedPower Loss(W)Transformer 17 2.1

Inductor 13 1.1Mosfet 11 0.2Zener 45 1.5Diode 5 0.08Total N/A 6.5

Table 1. Converter Losses-Room Temperature: 23C


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0 0.2 0.4 0.6 0.8 1

x 10-4

-20

-10

0

10

20

30

40

50

60

0 0.2 0.4 0.6 0.8 1

x 10-4

-10

-5

0

5

10

Time (s)

MosfetQ1Voltage(V)

Time (s)

(a)

(b)

MosfetQ1Current(A)

0 20 40 60 80 100

0 20 40 60 80 100

Figure 5. Simulated Waveforms (a) Mosfet Q1 current(b)Mosfet Q1 Voltage

6. SOFT SWITCHING OPTIONSThe experimental results have highlighted the effects

produced by the leakage inductance and other parasiticeffects. Our future goals are to raise both switching

frequency and efficiency. Three possible methods maybring improvements. These are:

The use of improved construction techniques suchas surface mount technology or co-axialtransformer designs, [7], to reduce leakageinductance.

Resonant switching, these techniques may be ableto reduce switching losses and make better use ofthe parasitic elements.

0 0.2 0.4 0.6 0.8 1

x 10-4

-400

-350

-300

-250

-200

-150

-100

-50

0

0 0.2 0.4 0.6 0.8 1

x 10

-4

-80

-60

-40

-20

0

20

40

60

80

0 0.2 0.4 0.6 0.8 1

x 10-4

-10

-5

0

5

10

Time (s)

(c)

Time (s)

(d)

Time (s)

(e)

DiodeVoltage(V)

TransformerPrimaryVoltage(V)

TransformerPrimaryCurrent(A)

0 20 40 60 80 100

0 20 40 60 80 100

0 20 40 60 80 100

Figure 5. Simulated Waveforms (continued)(c) Primary Current (d) PrimaryVoltage (e) Diode Voltage


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1 >1 >1 >1 >

1) Ch 1: 10 Volt 10 us

(a)

(b)

2 >2 >2 >2 >

2) Ch 2: 2.5Amp 10 us

Figure 6. Experimental Waveforms(a) Mosfet Q1 current (b) MosfetQ1 Voltage

Lossless snubbers, these can recover energystored in the leakage inductance and reduceswitching losses, [4,8].

The resonant switching approach can help in reducingswitching losses, [9], but is problematic in that the

switching device current and/or voltage stress maysignificantly increase. It is often difficult to maintainthe resonance conditions across a wide load range. Inthe medium term some lossless snubber solutions may

be better. Figure 7 shows a variation of the converterthat includes lossless or non-dissipative snubbers thatrecover the leakage inductance and reduce the turn offswitching losses. The operation sequence is asfollows. Assume Cs is initially charged to E:

2 >2 >2 >2 >

2) Ch 2: 2.5Amp 10 us

(c)

(d)

1 >1 >1 >1 >

1) Math: 20 Volt 10 us

(e)

2 >2 >2 >2 >

2) Math: 50 Volt 10 us

Figure 6. Experimental Waveforms (continued)(c) Primary Current (d) PrimaryVoltage (e) Diode Voltage


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R

L L

VE

Q1

Cs Ds

+Vcs Dr

LrQ2

Figure 7. Converter with non-dissipative snubber

Q1 turns off, and its drain current transfers to theloop, Cs, Ds and the supply E, given the initialcharge on Cs, Q1 turns off with zero voltage.

Cs linearly charges until the normal off statevoltage is reached, normally this is above +E.

The stored energy in the leakage inductancecontinues to drive a reducing current into Cs, thecurrent falls to zero once the leakage inductanceenergy is recovered, the capacitor, and the drainvoltage, is driven above the normal off statevoltage.

Ds blocks in the reverse, the capacitor remains inthis state until Q1 conducts once more.

At Q1s next conduction, a resonant current isestablished in Lr, Dr and Cs, this current reversesthe polarity of Cs, once a voltage of E is reached,Ds conducts allow Lr to discharge into the inputsource E.

While this approach is theoretically lossless it hassome disadvantages that need attention. The circuit isrelatively complex and additional components arerequired. It can be practically difficult to construct theloop Q1, Cs, Ds and the supply E. This loop needs tohave very low inductance if adequate voltage

clamping is to be maintained. The optimisation of thisand other snubber networks is an ongoing project.

7. CONCLUSIONThis paper has presented the dual converter anattractive candidate for low voltage input, high voltageoutput DC-DC conversion in PV systems. It offers anatural advantages in terms of conduction loss in lowvoltage systems and is much easier on the highvoltage rectifier diodes that can be so problematic inthe conventional designs. One area of relativedisadvantage is a higher level of sensitivity to

transformer leakage inductance. While this can be

controlled by construction, to a degree, furtherdevelopment is required to develop energy recoverymethods, such as non-dissipative snubbers. Whilereasonable performance has been obtained with ahard switched converter, the topology is under active

study with the aim of developing variations that aresuited to higher frequency, higher density and low lossapplications.

8. REFERENCES[1] Photovoltaic Energy for the New Millennium:

the US National Photovoltaics Program Plan,SunWorld, Vol. 24, No. 1, pp. 4-5, March/April2000.

[2] D. Butler, Domestic Rooftop Systems A CostEffective Answer to Greenhouse Emissions or a

Political Sop to the Green Lobby, Energy inCentral Queensland Facing the environmentalChallenge, IEAust. Central Region Conference,Central Queensland University, Jun. 23-25, 2000.

[3] M. Meinhardt, T. ODonnell, H. Schneider, J.Flannery, C. Mathuna, P. Zacharias and T.Krieger, Miniaturised Low Profile moduleIntegrated Converter for PhotovoltaicApplications with Integrated MagneticComponents, Proc. 1999 IEEE 14th Applied

Power Electronics Conference, 1999.

[4]

P. J. Wolfs, A Current-Sourced Dc-Dc ConverterDerived via the Duality Principle from the Half-Bridge Converter, IEEE Trans. Industrial

Electronics, Vol. 40, No. 1, pp. 139-144, Feb.1993.

[5] E. C. Snelling, Soft Ferrites, Properties andApplications. London: Butterworths , 1988, pp335-338.

[6] Soft Ferrites, Philips 1996 Data HandbookMA01, Netherlands, Aug. 30, 1995, pp. 94-95.

[7] M. H. Kheraluwala, D. W. Novotny and D. M.

Divan, Coaxially Wound Transformers for High-Power High-Frequency Applications, IEEETrans. Power Electronics, Vol. 7, No. 1, pp. 54-62, Jan. 1992.

[8] J. D. Van Wyk and J. A. Ferreira, TransistorInverter Design Optimization in the FrequencyRange above 5 kHz up to 50 kVA, IEEE Trans.

Industry Applications, Vol. IA-19, No. 2, pp. 296-302, March/April 1983.

[9] S. Freeland and R. D. Middlebrook, A UnifiedAnalysis of Converters with Resonant Switches,

Proc. IEEE PESC Conf. Rec. 1987, pp. 20-30.

The Application of the Half Bridge Dual Converter to Photovoltaic

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