Software Defined Radio Part 4

8/11/2019 Software Defined Radio Part 4

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20 Mar/Apr 2003

8900 Marybank Dr.Austin, TX [email protected]

A Software Defined Radiofor the Masses, Part 4

By Gerald Youngblood, AC5OG

We conclude this series with a description of a dc-60 MHz

transceiver that will allow open-software experimentation

with software defined radios.

It has been a pleasure to receivefeedback from so many QEXread-ers that they have been inspired

to experiment with software-definedradios (SDRs) through this article se-ries. SDRs truly offer opportunities toreinvigorate experimentation in the

service and attract new blood from theranks of future generations of com-puter-literate young people.1It is en-couraging to learn that many readerssee the opportunity to return to a loveof experimentation left behind becauseof the complexity of modern hardware.With SDRs, the opportunity again ex-

ists for the experimenter to achieveresults that exceed the performanceof existing commercial equipment.

Most respondents indicated an in-terest in gaining access to a completeSDR hardware solution on which theycan experiment in software. Based on

this feedback, I have decided to offerthe SDR-1000 transceiver described inthis article as a semi-assembled,three-board set. The SDR-1000 soft-ware will also be made available inopen-source form along with supportfor the GNU Radio project onLinux.2

Table 1 outlines preliminary specifi-cations for the SDR-1000 transceiver.I expect to have the hardware avail-able by the time this article is in print.

The ARRL SDR Working Group in-cludes in its mission the encourage-ment of SDR experimentation througheducational articles and the availabil-

ity of SDR hardware on which to ex-periment. A significant advance to-ward this end has been seen in thepages of QEXover the last year, andit continues into 2003.

This series began in Part 1 with ageneral description of digital signal

processing (DSP) in SDRs.3Part 2 de-scribed Visual Basic source code toimplement a full-duplex, quadratureinterface on a PC sound card.4Part 3described the use of DSP to make thePC sound-card interface into a func-tional software-defined radio.5It alsoexplored the filtering technique calledFFT fast-convolution filtering. In thisfinal article, I will describe the SDR-1000 transceiver hardware includingan analysis of gain distribution, noisefigure and dynamic range. There isalso a discussion of frequency controlusing the AD9854 quadrature DDS.

1Notes appear on page 28.


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Mar/Apr 2003 21

To further support the interestgenerated by this series, I have est-ablished a Web site at home.earthlink.net/~g_youngblood. Asyou experiment in this interestingtechnology, please e-mail suggested en-hancements to the site.

Is the Tayloe DetectorReally New?

In Part 1, I described what I knewat the time about a potentially new ap-proach to detection that was dubbedthe Tayloe Detector. In the same is-sue, Rod Green described the use ofthe same circuit in a multiple conver-sion scheme he called the Dirodyne.6

The question has been raised: Is thisnew technology or rediscovery of priorart? After significant research, I haveconcluded that both the Tayloe De-tector and the Dirodyne are simplyrediscovery of prior art; albeit littleknown or understood. In the Septem-ber 1990 issue of QEX, D. H. vanGraas, PADEN, describes TheFourth Method: Generating and De-tecting SSB Signals.7The three pre-vious methods are commonly calledthephasing method, thefilter methodand the Weaver method. The TayloeDetector uses exactlythe same con-cept as that described by van Grasswith the exception that van Grass usesa double-balanced version of the cir-cuit that is actually superior to the sin-gly-balanced detector described byDan Tayloe8 in 2001.

In his article, van Graas describes

how he was inspired by old frequency-converter systems that used ac motor-generators called selsyn motors. Theselsyn was one part of an electric axleformerly used in radar systems. Hiscircuit used the CMOS 4052 dual 1-4multiplexer (an early version of themore modern 3253 multiplexers ref-erenced in Part 1 of this series) to pro-vide the four-phase switching. Thearticle describes circuits for bothtransmit and receive operation.

Phil Rice, VK3BKR, published anearly identical version of the vanGraas transmitter circuit inAmateurRadio (Australia) in February 1998,which may be found on the Web.9

While he only describes the transmitcircuitry, he also states, . . . the switch-ing modulator should be capable ofacting as a demodulator.

Its the Capacitor, Stupid!So why is all this so interesting?

First, it appears that this truly is afourth method that dates back to atleast 1990. In the early 1990s, there wasa saying in the political realm: Its theeconomy, stupid! Well, in this case, itsthe capacitor, stupid! Traditional com-mutating mixers do not have capacitors(or integrators) on their output. Thecapacitor converts the commutatingswitch from a mixer into a samplingdetector (more accurately a track-and-hold) as discussed on page 8 of Part 1(see Note 3). Because the detector op-

erates according to sampling theory, themixing products sum aliases back to thesame frequency as the difference prod-uct, thereby limiting conversion loss. Inreality, a switching detector is simply amodified version of a digital commutat-ing filter as described in previous QEXarticles.10, 11, 12

Instead of summing the four ormore phases of the commutating fil-ter into a single output, the samplingdetector sums the 0and 180phasesinto the in-phase (I) channel and the90and 270phases into the quadra-ture (Q) channel. In fact, the math-ematical analysis described in MikeKossors article (see Note 10) appliesequally well to the sampling detector.

Is the Dirodyne Really New?

The Dirodyne is in reality the sam-pling detector driving the samplinggenerator as described by van Graas,forming the architecture first de-scribed by Weaver in 1956.13 TheWeaver method was covered in a se-

ries of QEXarticles14, 15, 16 that areworth reading. Other interesting read-ing on the subject may be found on theWeb in a Phillips Semiconductors ap-plication note17 and an article inMicrowaves & RF.18

Peter Anderson in his Jul/Aug 1999letter to the QEX editor specificallydescribes the use of back-to-back com-mutating filters to perform frequency

shifting for SSB generation or recep-tion.19 He states that if, on the outputof a commutating filter, we can, adda second commutator connected to thesame set of capacitors, and take theoutput from the second commutator.Run the two commutators at differentfrequencies and find that the inputpassband is centered at a frequencyset by the input commutator; the out-put passband is centered at a fre-quency set by the output commutator.Thus, we have a device that shifts thesignal frequency, an SSB generator orreceiver. This is exactly what the

Dirodyne does. He goes on to state,The frequency-shifting commutatingfilter is a generalization of the Weavermethod of SSB generation.

So What Shall We Call It?

Although Dan Tayloe popularizedthe sampling detector, it is probablynot appropriate to call it the Tayloedetector, since its origin was at least10 years earlier, with van Graas.Should we call it the van Graas De-tector or just the Fourth Method?Maybe we should, but since I dontknow if van Graas originally inventedit, I will simply call it the quadrature-sampling detector (QSD) or quadra-ture-sampling exciter (QSE).

Dynamic RangeHow Much is Enough?

The QSD is capable of exceptionaldynamic range. It is possible to designa QSD with virtually no loss and 1-dBcompression of at least 18 dBm(5 VP-P). I have seen postings on e-mail

Table 2Acceptable Noise Figure

for Terrestrial Communications

Frequency Acceptable

(MHz) NF (dB)

1.8 453.5 374.0 2714.0 2421.0 2028.0 1550.0 9144.0 2

Table 1SDR-1000 Preliminary Hardware Specifications

Frequency Range 0-60 MHzMinimum Tuning Step 1 HzDDS Clock 200 MHz,


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22 Mar/Apr 2003

Fig 1SDR-1000 receiver/exciter schematic.


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reflectors claiming measuredIP3in the+40 dBm range for QSD detectors us-ing 5-V parts. With ultra-low-noise au-dio op amps, it is possible to achieve ananalog noise figure on the order of 1 dBwithout an RF preamplifier. With ap-propriately designed analog AGC andcareful gain distribution, it is theoreti-cally possible to achieve over 150 dB oftotal dynamic range. The question is

whether that much range is needed fortypical HF applications. In reality, theanswer is no. So how much is enough?

Several QEXwriters have done anexcellent job of addressing the sub-ject.20, 21, 22Table 2 was originally pub-lished in an October 1975 ham radioarticle.23It provides a straightforwardsummary of the acceptable receivernoise figure for terrestrial communi-cation for each band from 160 m to2 m. Table 3 from the same article il-lustrates the acceptable noise figuresfor satellite communications on bandsfrom 10 m to 70 cm.

For my objective of dc-60 MHz cov-erage in the SDR-1000, Table 2 indi-cates that the acceptable noise figureranges from 45 dB on 160 m to 9 dB on6 m. This means that a 1-dB noise fig-ure is overkill until we operate near the2-m band. Further, to utilize a1-dB noise figure requires almost 70 dBof analog gain ahead of the sound card.This means that proper gain distribu-tion and analog AGC design is criticalto maximize IMD dynamic range.

After reading the referenced articlesand performing measurements on theTurtle Beach Santa Cruz sound card, Idetermined that the complexity of ananalog AGC circuit was unwarrantedfor my application. The Santa Cruz cardhas an input clipping level of 12 V (RMS,34.6 dBm, normalized to 50 ) whenset to a gain of 10 dB. The maximumoutput available from my audio signalgenerator is 12 V (RMS). The SDR soft-ware can easily monitor the peak sig-nal input and set the correspondingsound card input gain to effectively cre-ate a digitally controlled analog AGCwith no external hardware. I measuredthe sound cards 11-kHz SNR to be in

the range of 96 dB to 103 dB, depend-ing on the setting of the cards inputgain control. The input control is ca-pable of attenuating the gain by up to60 dB from full scale. Given the largesignal-handling capability of the QSDand sound card, the 1-dB compressionpoint will be determined by the outputsaturation level of the instrumentationamplifier.

Of note is the fact that DVD salesare driving improvements in PC soundcards. The newest 24-bit sound cardssample at a rate of up to 192 kHz. TheWaveterminal 192X from EGO SYS is

Table 3Acceptable Noise Figure for Satellite Communications

Frequency Galactic Noise Acceptable (MHz) Floor (dBm/Hz) NF (dB)

28 125 850 130 5144 139 1220 140 0.7432 141 0.2

Fig 2QS4A210insertion lossversus frequency

one example.24 The manufacturerboasts of a 123 dB dynamic range, butthat number should be viewed withcaution because of the technical diffi-culties of achieving that many bits oftrue resolution. With a 192-kHz sam-pling rate, it is possible to achieve real-time reception of 192 kHz of spectrum(assuming quadrature sampling).

Quadrature Sampling Detector/Exciter Design

In Part 1 of this series (Note 3), Idescribed the operation of a single-bal-anced version of the QSD. When the cir-cuit is reversed so that a quadratureexcitation signal drives the sampler, aSSB generator or exciter is created. Itis a simple matter to reverse the SDRreceiver software so that it transformsmicrophone input into filtered, quadra-ture output to the exciter.

While the singly-balanced circuitdescribed in Part 1 is extremely simple,I have chosen to use the double-bal-

anced QSD as shown in Fig 1 becauseof its superior common mode and even-harmonic rejection. U1, U6 and U7 formthe receiver and U2, U3 and U8 formthe exciter. In the receive mode, theQSD functions as a two-capacitor com-mutating filter, as described by ChenPing in his article (Note 11). A commu-tating filter works like a comb filter,

wherein the circuit responds to harmon-ics of the commutation frequency. As henotes, . . . it can be shown that signalshaving harmonic numbers equal to anyof the integer factors of the number ofcapacitors may pass. Since two capaci-tors are used in each of the I and Qchannels, a two-capacitor commutatingfilter is formed. As Ping further states,this serves to suppress the even-order

harmonic responses of the circuit. Theoutput of a two-capacitor filter is ex-tremely phase-sensitive, therefore al-lowing the circuit to perform signal de-tection just as a CW demodulator does.When a signal is near the filters cen-ter frequency, the output amplitudewould be modulated at the difference(beat) frequency. Unlike a typical filter,where phase sensitivity is undesirable,here we actually take advantage of thatcapability.

The commutator, as described inPart 1, revolves at the center frequencyof the filter/detector. A signal tuned ex-

actly to the commutating frequencywill result in a zero beat. As the signalis tuned to either side of the commuta-tion frequency, the beat note output willbe proportional to the difference fre-quency. As the signal is tuned towardthe second harmonic, the output willdecrease until a null occurs at the har-monic frequency. As the signal is tuned


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further, it will rise to a peak at the thirdharmonic and then decrease to anothernull at the fourth harmonic. This cyclewill repeat indefinitely with an ampli-tude output corresponding to thesin(x)/xcurve that is characteristic ofsampling systems as discussed in DSPtexts. The output will be further attenu-ated by the frequency-response char-acteristics of the device used for the

commutating switch. The PI5V331multiplexer has a 3-dB bandwidth of150 MHz. Other parts are availablewith 3-dB bandwidths of up to 1.4 GHz(from IDT Semiconductor).

Fig 2 shows the insertion loss ver-sus frequency for the QS4A210. Theupper frequency limitation is deter-mined by the switching speed of thepart (1 ns = T

on/ T

off, best-case or 12.5

ns worst-case for the 1.4-GHz part)and the sin(x)/xcurve for under-sam-pling applications.

The PI5V331 (functionally equiva-lent to the IDT QS4A210) is rated for

analog operation from 0 to 2 V. TheQS4A210 data sheet provides a drain-to-source on-resistance curve versusthe input voltage as shown in Fig 3.From the curve, notice that the on re-sistance (R

on) is linear from 0 to 1 V

and increases by less than 2 at 2 V.No curve is provided in the PI5V331data sheet, but we should be able toassume the two are comparable. Infact, the PI5V331 has aR

onspecifica-

tion of 3 (typical) versus the 5 (typical) for the QS41210. In the re-ceive application of the QSD, the Ronis looking into the 60-Minput of theinstrumentation amplifier. This meansthat Ron modulation is virtuallynonexistent and will have no materialeffect on circuit linearity.25 Unliketypical mixers, which are nonlinear,the QSD is a linear detector!

Eq 1 determines the bandwidth ofthe QSD, whereRantis the antenna im-pedance, CS is the sampling capacitor

value and nis the total number of sam-pling capacitors (1/nis effectively theswitch duty cycle on each capacitor). Inthe doubly balanced QSD, nis equal to2 instead of 4 as in the singly balanced

circuit. This is because the capacitor isselected twice during each commutationcycle in the doubly balanced version.

Santdet

1

CRnBW

=(Eq 1)

A tradeoff exists in the choice of QSDbandwidth. A narrow bandwidth suchas 6 kHz provides increased blockingand IMD dynamic range because of the

very high Qof the circuit. When de-signed for a 6-kHz bandwidth, theresponse at 30 kHzone decade fromthe 3-kHz 3-dB pointeither side of the

center frequency will be attenuated by20 dB. In this case, the QSD forms a6-kHz-wide tracking filter centered atthe commutating frequency. This meansthat strong signals outside the pass-band of the QSD will be attenuated,thereby dramatically increasing IP3and blocking dynamic range.

I am interested in wider bandwidthfor several reasons and therefore will-

ing to trade off some of the IMD-re-duction potential of the QSD filter. InSDR applications, it is desirable inmany cases to receive the widest band-width of which the sound card is ca-pable. In my original design, that is44 kHz with quadrature sampling.This capability increases to 192 kHzwith the newest sound cards. Not onlydoes this allow the capability of ob-serving the real-time spectrum of upto 192 kHz, but it also brings the po-tential for sophisticated noise and in-terference reduction.26

Further, as we will see in a moment,

the wider bandwidth allows us to re-duce the analog gain for a given sen-sitivity level. The 0.068-F samplingcapacitors are selected to provide a

QSD bandwidth of 22 kHz with a50- antenna. Notice that any vari-ance in the antenna impedance willresult in a corresponding change in thebandwidth of the detector. The onlyway to avoid this is to put a buffer infront of the detector.

The receiver circuit shown in Part 1used a differential summing op ampafter the detector. The primary advan-

tage of a low-noise op amp is that it canprovide a lower noise figure at low gainsettings. Its disadvantage is that theinverting input of the op amp will be at

virtual ground and the non-invertinginput will be high impedance. Thismeans that the sampling capacitor onthe inverting input will be loaded dif-ferently from the non-inverting input.Thus, the respective passbands of thetwo inputs will not track one another.This problem is eliminated if an instru-mentation amplifier is used. Anotheradvantage of using an instrumentationamplifier as opposed to an op amp is

that the antenna impedance is removedfrom the amplifier gain equation. Thesingle disadvantage of the instrumen-tation amplifier is that the voltage noise

Fig 3QS4A210 Ronversus VIN.

Table 4INA 163 Noise Data at 10 kHz

Gain (dB) e n

in

NF (dB)

20 7.5 nV/ Hz 0.8 pA/ Hz 12.440 1.8 nV/ Hz 0.8 pA/ Hz 3.060 1.0 nV/ Hz 0.8 pA/ Hz 1.3


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and thus the noise figure increaseswith decreasing gain.

Table 4 shows the voltage noise,current noise and noise figure for a200- source impedance for the TIINA163 instrumentation amplifier.Since a single resistor sets the gain ofeach amplifier, it is a simple matter toprovide two or more gain settings withrelay or solid-state switching.

Unlike typical mixers, which arenormally terminated in their character-istic impedances, the QSD is a high-im-pedance, sampling device. Within thepassband, the QSD outputs are termi-nated in the 60-M inputs of the in-strumentation amplifiers. The IDT datasheet for the QS4A210 indicates thatthe switch has no insertion losswithloads of 1 kor more! This coincideswith my measurements on the circuit.If you apply 1 V of RF into the detector,you get 1 V of audio out on each of thefour capacitorsa no-loss detector.Outside the passband, the decreasing

reactance of the sampling capacitorswill reduce the signal level on the am-plifier inputs. While it is possible to in-sert series resistors on the output of theQSD, so that it is terminated outsidethe passband, I believe this is unneces-sary. For receive operation, filter reflec-tions outside the passband are not veryimportant. Further, the terminationresistors would create an additionalsource of thermal noise.

As stated earlier, the circuitry of theQSD may be reversed to form aquadrature sampling exciter (QSE). Todo so, we must differentially drive theI and Q inputs of the QSE. The TexasInstruments DRV135 50- differen-tial audio line driver is ideally suitedfor the task. Blocking capacitors on thedriver outputs prevent dc-offset varia-tion between the phases from creat-ing a carrier on the QSE output. Car-rier suppression has been measuredto be on the order of 48 dBc relativeto the exciters maximum output of+10 dBm. In transmit mode, the out-put impedance of the exciter is 50 so that the band-pass filters are prop-erly terminated.

Conveniently, T/R switching is asimple matter since the QSD and QSEcan have their inputs connected in par-allel to share the same transformer.Logic control of the respective multi-plexer-enable lines allows switchingbetween transmit and receive mode.

Level Analysis

The next step in the design processis to perform a system-level analysisof the gain required to drive the soundcard A/D converter. One of the betterreferences I have found on the subjectis the book by W. Sabin and E.

Schoenike, HF Radio Systems andCircuits.27The book includes anExcelspreadsheet that allows interactiveexamination of receiver performanceusing various A/D converters, samplerates, bandwidths and gain distribu-tions. I have placed a copy of the SDR-1000 Level Analysis spreadsheet (bypermission, a highly modified versionof the one provided in the book) for

download from ARRLWeb.28

Anotherexcellent resource on the subject is theDigital Receiver/Exciter Design chap-ter from the bookDigital Signal Pro-cessing in Communication Systems.29

Notice that the former referencehas a better discussion of the mini-mum gain required for thermal noiseto transition the quantizing level asdiscussed here. Neither text deals withthe effects of atmospheric noise on thenoise floor and hence on dynamicrange. This isin my opinionamajor oversight for HF communica-tions since atmospheric noise will

most likely limit the minimumdiscernable signal, not thermal noise.

For a weak signal to be recovered,the minimum analog gain must be greatenough so that the weakest signal tobe received, plus thermal and atmo-spheric noise, is greater than at leastone A/D converter quantizing level (theleast-significant usable bit). For the

A/D converter quantizing noise to beevenly distributed, several quantizinglevels must be traversed. There are twoprimary ways to achieve this: Out-of-band dither noise may be added andthen filtered out in the DSP routines,or in-band thermal and atmosphericnoise may be amplified to a level thataccomplishes the same. While the firstapproach offers the best sensitivity atthe lowest gain, the second approach issimpler and was chosen for my appli-cation.HF Radio Systems and Circuitsstates, Normally, if the noise isGaussian distributed, and the RMSlevel of the noise at the A/D converteris greater than or equal to the level of asine wave which just bridges a singlequantizing level, an adequate numberof quantizing levels will be bridged to

guarantee uniformly distributed quan-tizing noise. Assuming uniform noisedistribution, Eq 2 is used to determinethe quantizing noise density,N0q:

W/Hz6

2

s

2pp

0qRf

V

Nb

=(Eq 2)

whereVP-P= peak-to-peak voltage rangeb= number of valid bits of resolutionfs= A/D converter sampling rateR= input resistanceN0q= quantizing noise density

The quantizing noise decreases by3 dB when doubling the sampling rateand by 6 dB for every additional bit ofresolution added to the A/D converter.Notice that just because a converteris specified to have a certain numberof bits does not mean that they are allusable bits. For example, a convertermay be specified to have 16 bits;but in reality, only be usable to 14-bits.

The Santa Cruz card utilizes an 18-bit A/D converter to deliver 16 usablebits of resolution. The maximum sig-nal-to-noise ratio may be determinedfrom Eq 3:

dB75.102.6 += bSNR (Eq 3)

For a 16-bit A/D converter havinga maximum signal level (without in-put attenuation) of 12.8 VP-P, the mini-mum quantum level is 70.2 dBm.Once the quantizing level is known,we can compute the minimum gain re-quired from Eq 4:

BWNFcatmospheri

NFanalogkTBlevelquantizingGain

10log10++=

(Eq 4)

where:

dBm001.050

22

707.0

log10

2

10

=

b

ppV

levelquantizing

kTB/ Hz= 174 dBm/Hz

analog NF = analog receiver noise fig-

ure, in decibelsatmospheric NF = atmospheric noisefigure for a given frequency

BW= the final receive filter bandwidthin hertz

Table 5, from the SDR-1000 LevelAnalysis spreadsheet, provides thecascaded noise figure and gain for thecircuit shown if Fig 1. This is wherethings get interesting.

Fig 4 shows an equivalent circuitfor the QSD and instrumentation am-plifier during a respective switch pe-riod. The transformer was selected to

have a 1:4 impedance ratio. Thismeans that the turns ratio from theprimary to the secondary for eachswitch to ground is 1:1, and thereforethe voltage on each switch is equal tothe input signal voltage. The differen-tial impedance across the transformersecondary will be 200 , providing agood noise match to the INA163 am-plifier. Since the input impedance ofthe INA163 is 60 M, power lossthrough the circuit is virtually nonex-istent. We must therefore analyze thecircuit based on voltage gain, notpower gain.


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26 Mar/Apr 2003

Table 5 Cascaded Noise Figure and Gain Analysis from the SDR-1000 Level Analysis Spreadsheet

BPF T1-4 PI5V331 INA163 ADC

dB Noise Figure 0.0 0.0 0.0 3.0 58.6dB Gain 0.0 6.0 0.0 40.0 0.0Equivalent Power Factor Noise Factor 1.00 1.00 1.00 1.99 720,482Equivalent Power Factor Gain 1 4 1 10,000 1Clipping Level Vpk 1.0 13.0 6.4Clipping Level dBm 10.0 32.3 26.1

Cascaded Gain dB 0.0 6.0 6.0 46.0 46.0Cascaded Noise Factor 1.00 1.00 1.00 1.25 19.06Cascaded Noise Figure dB 0.0 0.0 0.0 1.0 12.8Output Noise dBm/Hz 174.0 174.0 174.0 173.0 161.2

Fig 4Doubly balanced QSD equivalentcircuit.

Table 6Atmospheric Equivalent Noise Figure By Band

Band (Meters) Ext Noise Ext NF

(dBm/Hz) (dB)

160 128 4680 136 3840 144 30

30 146 2820 146 2817 152 2215 152 2212 154 2010 156 186 162 12

That means that we get a 6-dBdifferential voltage gain from the in-put transformerthe equivalent of a0-dB noise figure amplifier! Further,there is no loss through the QSDswitches due to the high-impedanceload of the INA. With a source imped-ance of 200 , the INA163 has a noisefigure of approximately 12.4 dB at20 dB of gain, 3 dB at 40 dB of gainand 1.3 dB at 60 dB of gain.

In fact, the noise figure of the ana-log front end is so low that if it werenot for the atmospheric noise on the HFbands, we would need to add a lot ofgain to amplify the thermal noise to thequantizing level. The textbook refer-ences ignore this fact. In addition to theham radioarticle (Note 23) and PeterChadwicks QEX article (Note 20), JohnStephenson in his QEXarticle30aboutthe ATR-2000 HF transceiver providesfurther insight into the subject.Table 6 provides a summary of the ex-ternal noise figure for a by-band quietlocation as determined from Fig 1 in

Stephensons article. As can be seenfrom the table, it is counterproductiveto have high gain and low receiver noisefigure on most of the HF bands.

Tables 7 and 8 are derived from theSDR-1000 Level Analysisspreadsheet(Note 28) for the 10-m band. Thespreadsheet tables interact with oneanother so that a change in an as-sumption will flow through all theother tables. A detailed discussion ofthe spreadsheet is beyond the scopeof this text. The best way to learn howto use the spreadsheet is to plug invalues of your own. It is also instruc-

tive to highlight cells of interest to seehow the formulas are derived. Basedon analysis using the spreadsheet, Ihave chosen to make the gain settingrelay-selectable between INA gain set-tings of 20 dB for the lower bands and40 dB for the higher bands.

It is important to remember that mynoise and dynamic-range calculationsinclude external noise figure in addition

to the thermal noise figure.This is muchmore realistic for HF applications thanthe typical lab testing and calculationsyou see in most references. With theINA163 gain set to 40 dB, the cascadedanalog thermal NF is calculated to be

just 1 dB at the input to the sound card.If it were not for the external noise,nearly 70 dB of analog gain would berequired to amplify the thermal noise

Table 7SDR-1000 Level Analysis Assumptions for the 10-Meter Bandwith 40 dB of INA Gain

Receiver Gain Distribution and Noise PerformanceTurtle Beach Santa Cruz Audio CardBand Number 9Band 10 MetersInclude External NF? (True=1, False=0) 1External (Atmospheric) Noise Figure 18 dBA/D Converter Resolution (bits) 16 bits (98.1 dB)A/D Converter FullScale Voltage 6.4 V-peak (26.1 dBm)A/D Converter Quantizing Signal Level 70.2 dBmQuantizing Gain Over/(Under) 7.2 dBA/D Converter Sample Frequency 44.1 kHzA/D Converter Input Bandwidth (BW1) 40.0 kHzInformation Bandwidth (BW2) 0.5 kHzSignal at Antenna for INA Saturation 13.7 dBmNominal DAC Output Level 0.5 V peak (4.0 dBm)

AGC Threshold at Ant (40 dB Headroom) 51.4 dBmSound Card AGC Range 60.0 dB


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Mar/Apr 2003 27

to the quantizing level or dither noisewould have to be added outside thepassband. Fig 6 illustrates the signal-to-noise ratio curve with externalnoise for the 10-m band and 40 dB ofINA gain. Fig 5 shows the same curvewithout external noise and with INAgain of 60 dB. This much gain wouldnot improve the sensitivity in the pres-ence of external noise but would re-

duce blocking and IMD dynamic rangeby 20 dB. On the lower bands, 20 dBor lower INA gain is perfectly accept-able given the higher external noise.

Frequency Control

Fig 7 illustrates the Analog DevicesAD9854 quadrature DDS circuitry fordriving the QSD/QSE. Quadrature lo-cal-oscillator signals allow the elimi-nation of the divide-by-four Johnsoncounter, described in Part 1, so thatthe DDS runs at the carrier frequencyinstead of its fourth harmonic. I havechosen to use the 200-MHz version of

the part to minimize heat dissipation,and because it easily meets my fre-quency coverage requirements of dc-60 MHz. The DDS outputs are con-nected to seventh-order ellipticallow-pass filters that also provide a dcreference for the high-speed compara-tors. The AD9854 may be controlledeither through a SPI port or a paral-lel interface. There are timing issuesin SPI mode that require special carein programming. Analog Devices havedeveloped a protocol that allows thechip to be put into external I/O updatemode to work around the serial

Table8SDR-1000LevelAnalys

isDetailforthe10-MeterBandwith40dBofINAGain

Antenna

INA

Antenna

Total

SoundCard

Noiseat

A/D

Noisein

Quantiz

ing

Total

Output

D

igital

Signal

Output

Overload

Analog

AGC

A/DInput

Signal

A/DInput

Noise

of

Noise

S/NRatio

G

ain

Level

Level

Level

Gain

Reduction

inBW1

Level

inBW2

A/DinB

W2

inBW2

inBW2

Required

(dBm)

(dBm)

(dBm)

(dB)

(dB)

(dBm)

(dBm)

(dBm)

(dBm

)

(dBm)

(dB)

(dB)

128

82

46.

0

0.

0

63.

0

82.

0

82.

0

88.4

81.

1

0.

9

86.

0

118

72

46.

0

0.

0

63.

0

72.

0

82.

0

88.4

81.

1

9.

1

76.

0

108

62

46.

0

0.

0

63.

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About Intel Performance

Primitives

Many readers have inquired aboutIntels replacement of its Signal Pro-cessing Library (SPL) with the IntelPerformance Primatives (IPP). TheSPL was a free distribution, but theIntel Web site states that IPP re-quires payment of a $199 fee after a30 day evaluation period. A fullyfunctional trial version of IPP may bedownloaded from the Intel site atwww.intel.com/software/prod-ucts/global/eval.htm. The authorhas confirmed with Intel ProductManagement that no license fee isrequired for amateur experimenta-tion using IPP, and there is no limiton the evaluation period for suchuse. Intel actually encourages thistype of experimental use. Payment ofthe license fee is required if and onlyif there is a commercial distribution ofthe DLL code.Gerald Youngblood


9/12

28 Mar/Apr 2003

Fig 5Output signal-to-noise ratio excluding external(atmospheric) noise. INA gain is set to 60 dB. Antenna signal levelfor saturation is 33.7 dBm.

Fig 6Output signal-to-noise ratio for the 10-m band includingexternal (atmospheric) noise. INA gain is set to40 dB. Antenna signal level for INA saturation is 13.7dBm.

timing problem. In the final circuit, Ichose to use the parallel mode.

According to Peter Chadwicks ar-ticle (Note 20), phase-noise dynamicrange is often the limiting factor in re-ceivers instead of IMD dynamic range.The AD9854 has a residual phase noiseof better than 140 dBc/Hz at a 10-kHzoffset when directly clocked at 300 MHzand programmed for an 80-MHz out-

put. A very low-jitter clock oscillator isrequired so that the residual phasenoise is not degraded significantly.

High-speed data communicationstechnology is fortunately driving theintroduction of high-frequency crystaloscillators with very low jitter specifi-cations. For example, Valpey Fishermakes oscillators specified at less than1 ps RMS jitter that operate in thedesired 200-300 MHz range. Accordingto Analog Devices, 1 ps is on the orderof the residual jitter of the AD9854.

Band-Pass Filters

Theoretically, the QSD will work justfine with low-pass rather than band-pass filters. It responds to the carrierfrequency and odd harmonics of the car-rier; however, very large signals at halfthe carrier frequency can be heard inthe output. For example, my measure-ments show that when the receiver istuned to 7.0 MHz, a signal at 3.5 MHzis attenuated by 49 dB. The measure-ments show that the attenuation of thesecond harmonic is 37 dB and the thirdharmonic is down 9 dB from the 7-MHzreference. While a simple low-pass fil-

ter will suffice in some applications, Ichose to use band-pass filters.Fig 8 shows the six-band filter de-

sign for the SDR-1000. Notice that only

the 2.5-MHz filter has a low-pass char-acteristic; the rest are band-pass filters.

SDR-1000 Board Layout

For the final PC-board layout, I de-cided on a 34-inch form factor. Thereceiver, exciter and DDS are locatedon one board. The band-pass filter anda 1-W driver amplifier are located ona second board. The third board has a

PC parallel-port interface for control,and power regulators for operationfrom a 13.8-V dc power source. Thethree boards sandwich together intoa small 342-inch module with rear-mount connectors and no interconnec-tion wiring required. The boards useprimarily surface-mount components,except for the band-pass filter, whichuses mostly through-hole components.

Acknowledgments

I would like to thank David Bran-don and Pascal Nelson of Analog De-vices for their answering my questionsabout the AD9854 DDS. My apprecia-tion also goes to Mike Pendley,WA5VTV, for his assistance in designof the band-pass filters as well as hisongoing advice.

Conclusion

This series has presented a practi-cal approach to high-performanceSDR development that is intended tospur broad-scale amateur experimen-tation. It is my hopeand that of theARRL SDR Working Groupthatmany will be encouraged to contrib-

ute to the technical art in this fasci-nating area. By making the SDR-1000hardware and software available tothe amateur community, software ex-

tensions may be easily and quicklyadded. Thanks for reading.

Notes1M. Markus, N3JMM, Linux, Software Radio

and the Radio Amateur, QST, October2002, pp 33-35.

2The GNU Radio project may be found atwww.gnu .org /so f tware /gnurad io / gnuradio.html.

3G. Youngblood, AC5OG, A Software De-fined Radio for the Masses: Part 1, QEX,

Jul/Aug 2002, pp 13-21.4G. Youngblood, AC5OG, A Software De-fined Radio for the Masses: Part 2, QEX,Sep/Oct 2002, pp 10-18.

5G. Youngblood, AC5OG, A Software De-fined Radio for the Masses: Part 3, QEX,Nov/Dec 2002, pp 27-36.

6R. Green, VK6KRG, The Dirodyne: A NewRadio Architecture? QEX, Jul/Aug 2002,pp 3-12.

7D. H. van Graas, The Fourth Method: Gen-erating and Detecting SSB Signals, QEX,Sep 1990, pp 7-11.

8D. Tayloe, N7VE, Letters to the Editor,Notes on Ideal Commutating Mixers (Nov/Dec 1999), QEX, Mar/Apr 2001, p 61.

9

P. Rice, VK3BHR, SSB by the FourthMethod? ironbark.bendigo.latrobe.edu.au/~rice/ssb/ssb.html .

10M. Kossor, WA2EBY, A Digital Commu-tating Filter, QEX, May/Jun 1999, pp 3-8.

11C. Ping, BA1HAM, An Improved SwitchedCapacitor Filter, QEX, Sep/Oct 2000,pp 41-45.

12P. Anderson, KC1HR, Letters to the Edi-tor, A Digital Commutating Filter, QEX,Jul/Aug 1999, pp 62.

13D. Weaver, A Third Method of Generationof Single-Sideband Signals, Proceedingsof the IRE, Dec 1956.

14P. Anderson, KC1HR, A Different Weaveof SSB Exciter, QEX, Aug 1991, pp 3-9.

15P. Anderson, KC1HR, A Different Weave of

SSB Receiver, QEX, Sep 1993, pp 3-7.16C. Puig, KJ6ST, A Weaver Method SSB

Modulator Using DSP, QEX, Sep 1993,pp 8-13.


10/12

Mar/Apr 2003 29

Fig 7SDR-1000 quadrature DDS schematic.


11/12

30 Mar/Apr 2003

Fig 8SDR-1000 six-band filter schematic.


12/12

Mar/Apr 2003 31

17R. Zavrell Jr, New Low Power SingleSideband Circuits, Phillips Semiconduc-tors Application Note AN1981, Oct 1997,www.semiconductors.philips.com/ac-robat/applicationnotes/AN1981.pdf .

18M. Vidmar, Achieve Phasor Rotation WithAnalog Switches, Microwaves & RF, Feb2000, www.mwrf.com/Articles/Index.cfm?ArticleID=10069&extension=pdf .

19P. Anderson, KC1HR, Letters to the Edi-tor, A Digital Commutating Filter, QEX,

Jul/Aug 1999, pp 62.20P. Chadwick, G3RZP, HF Receiver Dy-namic Range: How Much Do we Need?QEX,May/Jun 2002, pp 36-41.

21J. Scarlett, KD7O, A High-PerformanceDigital Transceiver Design, Part 1, QEX,Jul/Aug 2002, pp 36-39.

22U. Rohde, KA2WEU/DJ2LR/HB9AWE,Theory of Intermodulation and ReciprocalMixing: Practice, Definitions and Mea-surements in Devices and Systems, Part1, QEX, Nov/Dec 2002, pp 3-15.

23J. Fisk, Receiver Sensitivity, Noise Figureand Dynamic Range, ham radio maga-zine, Oct 1975, pp 8-25. Ham radiomaga-zine is available on CD from ARRL.

24The Waveterminal 192X is manufacturedby EGO SYS and may be found on theWeb at www.esi-pro.com.

25J. Wynne, Ron Modulation in CMOSSwitches and Multiplexers; What It is andHow to Predict its Effect on Signal Distor-tion, Application Note AN-251, AnalogDevices, pp 1-3; www.analog.com/UploadedFiles/Application_Notes/413221855AN251.pdf .

26L. sbrink, SM5BSZ, Linrad: New Possi-bilities for the Communications Experi-menter, Part 1, QEX, Nov/Dec 2002, pp37-41. More information on the Web atwww.nitehawk.com/sm5bsz/linuxdsp/linrad.htm.

27W. Sabin and E. Schoenike, Editors, HF

Radio Systems & Circuits, (Atlanta, Geor-gia: Noble Publishing Corporation, ISBN1-884932-04-5) pp 349-355 and 642-644.

28 You can download this package from theARRL Web http://www.arrl.org/qexfiles/.Look for 0303Youn.ZIP.

29M. Frerking, Digital Signal Processing inCommunication Systems(Boston, Massa-chusetts: Kluwer Academic Publishers,ISBN 0-442-01616-6), pp 305-391.

30J. Stephenson, KD6OZH, The ATR-2000:A Homemade High-Performance HF Trans-ceiver, Pt 1, QEX, Mar/Apr 2000, pp 3-4.

Software Defined Radio Part 4

Documents

Transcript of Software Defined Radio Part 4