Research Article Considerations of Physical Design and...

Research ArticleConsiderations of Physical Design and Implementation for5 MHz-100 W LLC Resonant DC-DC Converters

Akinori Hariya,1 Ken Matsuura,2 Hiroshige Yanagi,3 Satoshi Tomioka,3

Yoichi Ishizuka,1 and Tamotsu Ninomiya4

1Nagasaki University, Graduate School of Engineering, 1-14 Bunkyo-machi, Nagasaki-shi, Nagasaki, Japan2TDK Corporation, 2-15-7 Higashiowada, Ichikawa-shi, Chiba, Japan3TDK-Lambda Corporation, 2704-1 Settaya, Nagaoka-shi, Nigata, Japan4Green Electronics Research Institute, Kitakyushu, 1-1 Jyonai, Kokurakita-ku, Kitakyushu-shi, Fukuoka, Japan

Correspondence should be addressed to Akinori Hariya; [email protected]

Received 3 June 2016; Accepted 29 August 2016

Academic Editor: Hongyang Zhao

Copyright © 2016 Akinori Hariya et al.This is an open access article distributed under the Creative Commons Attribution License,which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Recently, high power-density, high power-efficiency, and wide regulation range isolated DC-DC converters have been required.This paper presents considerations of physical design and implementation for wide regulation range MHz-level LLC resonant DC-DC converters. The circuit parameters are designed with 3–5MHz-level switching frequency. Also, the physical parameters andthe size of the planar transformer are optimized by using derived equations and finite element method (FEM) with Maxwell 3D.Some experiments are done with prototype LLC resonant DC-DC converter using gallium nitride high electronmobility transistors(GaN-HEMTs); the input voltage is 42–53V, the reference output voltage is 12 V, the load current is 8 A, the maximum switchingfrequency is about 5MHz, the total volume of the circuit is 4.1 cm3, and the power density of the prototype converter is 24.4W/cc.

1. Introduction

Recently, high power-density and high power-efficiency iso-lated DC-DC converters have been required in informationand communication technology (ICT) facilities. Therefore,high switching frequency and low power loss technique havebeen studied. In particular, to achieve these requirements,the isolated DC-DC converters with GaN-HEMTs have beenstudied by some research institute in recent years [1–7].

Furthermore, the regulated MHz-level operated isolatedDC-DC converters have been studied with wide input volt-age: 60–120V input voltage, 20V output voltage, 2.6MHzswitching frequency, and 50W output power, series resonantDC-DC converter [8]; 36–72V input voltage, 12 V outputvoltage, 5MHz switching frequency, and 30W output power,flyback DC-DC converter [9]; 110–250V input voltage, 15 Voutput voltage, 2MHz switching frequency, and 45W outputpower, quasi-square-wave zero-voltage switching three-levelhalf-bridge DC-DC converter [10]; 60–120V input voltage,

15 V output voltage, 3–4.5MHz switching frequency, and40Woutput power, zero-voltage switching (ZVS) half-bridgeDC-DC converter [11]. These isolated DC-DC convertersachieve high power efficiency. However, higher power effi-ciency and higher output power are required in MHz-levelfrequency isolated DC-DC converter.

LLC resonant DC-DC converters have been known toachieve high power efficiency with high switching frequency.In case of this topology, the regulated converters of 1MHz-level switching frequency have been studied so far [12,13]. Also, the unregulated LLC resonant DC-DC convertersoperated at 5MHz have been considered so far [14]. In orderto achieve the miniaturization of the converter, LLC resonantDC-DC converters with wide input voltage and high powerefficiency are needed at higher switching frequency.

In this paper, the designs of high power-density and wideregulated LLC resonant DC-DC converter have been consid-ered. The circuit parameters have been calculated for widerregulation atMHz-level switching frequency.The target input

Hindawi Publishing CorporationActive and Passive Electronic ComponentsVolume 2016, Article ID 4027406, 11 pageshttp://dx.doi.org/10.1155/2016/4027406

2 Active and Passive Electronic Components

Transformer

+

−

+

−

+

−

+ −

Ladd Llk1

Q1

Q2

�DS2

D1

D2

�D2

+ −�D1

𝛼 : 1 : 1Vo

RL

Co

LmLr

ir1

Cr2

Cr1

Vi

Figure 1: LLC resonant DC-DC converter circuit topology.

Table 1:The target specification of the prototype DC-DC converter.

Input voltage range: 𝑉i 42–53V (48V)Reference output voltage: 𝑉o 12 VOutput power: 𝑃o 100WPower efficiency: 𝜂 More than 90%Volume 3.3 cm3

Power density 30W/cc

voltage is 42–53V, output voltage is 12 V, output power is100W, and power efficiency is more than 90%. This designprocedure includes planar transformer design with FEM ofMaxwell 3D. To evaluate the performance of the prototypeboard, the experiment has been conducted with open loop.

In Section 2, the consideration of LLC resonant DC-DC converter circuit parameter is described. In Section 3,the physical design considerations of planar transformerare revealed. In Section 4, the experimental results aredemonstrated.

2. The Consideration ofCircuit Parameter Design

The circuit topology is based on a LLC resonant DC-DCconverter, as shown in Figure 1. The target specification isshown in Table 1. The primary side is a half-bridge topology.𝑄1and𝑄

2are driven in 50% duty ratio, alternatively. 𝐶r1 and

𝐶r2 are the resonant capacitances which have the same capac-itance. Also, they make averaged voltage of 𝐶r2 to one half ofthe input. The magnetic transformer is composed of leakageinductance 𝐿 lk1 and magnetizing inductance 𝐿m. 𝐿add isprimary-side additional inductance.The secondary side is thecenter-tap topology with diodes D

1and D

2. To achieve target

input voltage range, transformer turn ratio is set to be 3.LLC resonant DC-DC converters have two resonant

frequencies. One is the series resonant frequency 𝑓0which

is series resonance between resonant capacitance 𝐶r andinductance 𝐿 r. The other is the resonant frequency 𝑓

1which

is resonance between𝐶r, 𝐿 r andmagnetizing inductance 𝐿m.LLC resonant DC-DC converter can operate in the region of

i1 i2

�1

+

−

+

−

�2

n : 1

𝛼 : 1

L12 = L21

L11 L22

Llk1

Lm

Figure 2: The equivalent circuit of the magnetic transformer.

above resonance 𝑓s > 𝑓0 or below resonance 𝑓1< 𝑓s < 𝑓0. In

case of above resonance, ZVS of primary-side switches can beachieved. In this region, the converter behaves as buck-modeoperation. In case of below resonance, ZVS of primary-sideswitches and zero-current switching (ZCS) of secondary-sidediodes can be achieved. In this region, the converter behavesas boost-mode operation. LLC resonant DC-DC converterused in this paper is operated in below resonance at ratedcurrent.

In order to achieve the target specification, the design ofLLC resonant DC-DC converter is considered. At first, theequivalent circuit of the magnetic transformer can be drawn,as shown in Figure 2 [15]. Also, the following equations canbe obtained:

𝐿 lk1 = (1 − 𝑘2) 𝐿11,

𝐿m = 𝑘2𝐿11,

𝛼 = 𝑘𝑛 = 𝑘√𝐿11

𝐿22

,

(1)

where 𝐿 lk1 is the leakage inductance, 𝐿m is the magne-tizing inductance, 𝑘 is the coupling coefficient, 𝐿

11is the

primary-side self-inductance, 𝐿22is the secondary-side self-

inductance, and 𝑛 is the turn ratio.

Active and Passive Electronic Components 3

In LLC resonant DC-DC converter, for achieving ZVSoperation of primary-side switches, the magnetizing induc-tance has to be designed small enough. The magnetizinginductance can result in the following equation [16, 17]:

𝐿m ≤𝑇s𝑡dead16𝐶oss

, (2)

where 𝑇s is the switching period, 𝑡dead is the dead time ofthe primary-side switches, and 𝐶oss is the output capacitanceof the primary-side switches. From this equation, when theparameters are set as 𝑇s = 250 ns, 𝑡dead = 10 ns, and 𝐶oss =654 pF at𝑉i =42V, the value of𝐿m < 238.9 nH can be derived.Therefore, the magnetizing inductance can be set at 220 nH.Also, LLC resonantDC-DC converter has been analyzedwithfundamental harmonic approximation (FHA) [16, 17]. Thedefinitions for this analysis are as follows:

𝜔s = 2𝜋𝑓s, (3)

𝜔0= 2𝜋𝑓

0=

1

√𝐿 r𝐶r, (4)

𝐹 =𝑓s𝑓0

=𝜔s𝜔0

, (5)

𝑄 =𝑍0

𝑅ac=

1

𝑅ac√𝐿 r𝐶r, (6)

𝐿11= 𝐿 lk1 + 𝐿m, (7)

𝐿 r = 𝐿add + 𝐿 lk1, (8)

𝐾 =𝐿m𝐿 r

, (9)

𝐶r = 𝐶r1 + 𝐶r2, (10)

𝑅ac =8

𝜋2𝑅L, (11)

where 𝑓s is the switching frequency, 𝑓0is the resonant fre-

quency, 𝑄 is the quality factor of the circuit, 𝑅ac is the equiv-alent resistance, 𝐶r is the resonant capacitance, 𝐿 r is the res-onant inductance, 𝐿add is the additional inductance for opti-mizing 𝐿 r, and 𝐾 is the ratio of the inductances. The equiva-lent circuit of LLC resonant DC-DC converter can be drawn,as shown in Figure 3. From this equivalent circuit, the ratioof the input and output voltage𝑀 can be obtained as follows:

𝑀 =VacVs

=2𝛼𝑉o𝑉i

=1

√(1 + (1/𝐾) (1 − 1/𝐹2))2+ 𝑄2 (𝐹 − 1/𝐹)

2

.

(12)

When 𝑘 is set to 0.94, 𝛼 is decided at 2.82. The minimuminput voltage is 42V; thus, the required peak ratio𝑀p can be

+

−

+

−

�acRac

Lm

Lr

Cr

�s

Figure 3: The equivalent circuit of LLC resonant DC-DC converterin FHA.

0.20 0.30 0.400.10Q

K = 3

K = 4

K = 5

K = 6

K = 7

K = 8

K = 9

1.00

1.25

1.50

1.75

2.00

2.25

2.50

2.75

3.00

Mp

Figure 4: The relationship between peak ratio 𝑀p and the qualityfactor 𝑄.

derived about 2 from (12).This result includes 20%margin of𝑀. Figure 4 can result from (12). The limitation of 𝑄 in each𝐾 can be found from this graph. B82801B manufactured byEPCOS is used as not only current measurement but alsoadditional inductance. The short inductance of the currenttransformer is 7 nH. Therefore, from Figure 5, when 𝐾 =6, 𝐿 r is decided about 36.7 nH. Also, from Figure 4, 𝑄 <

0.23. Then, 𝐶r and 𝑓0can be decided from (6) and (4),

respectively; 𝐶r = 12.66 nF and 𝑓0= 7.4MHz.

3. The Physical Design Considerations ofPlanar Transformer

In general, Nickel Zinc (NiZn) ferrite core has been usedin MHz-level switching frequency operation. In terms ofthe core of this prototype isolated DC-DC converter, SY22NiZn ferrite core manufactured by TDK is used [18]. Therelative permeability 𝜇i of the core material is 80. This corematerial can be used from 5 to 15MHz. The good featuresare guaranteed in MHz-level frequency. The core shape is EI

4 Active and Passive Electronic ComponentsL

add

(nH

)

0

2

4

6

8

10

12

14

16

18

20

200 250 300 350 400 450 500150Lm (nH)

K = 3

K = 4

K = 5

K = 6

K = 7

K = 8

Figure 5:The relationship between the magnetizing inductance 𝐿mand the additional inductance 𝐿add.

core. In this paper, the physical parameters of this EI core areoptimized for obtaining good performance.

For the prototype LLC resonant DC-DC converter, 8-layer printed circuit board (PCB) is used. The windingarrangement of the prototype planar transformer is shownin Figure 6. In terms of planar transformer, layers 1, 2, 7,and 8 (L1, 2, 7, and 8) are assigned to primary-side windings,while layers 3, 4, 5, and 6 (L3, 4, 5, and 6) are assigned tosecondary-side windings. The primary-side turn number is3 turns, 1 layer has 2 turns, and parallel number is 2. Also,the secondary-side turn number is 1 turn, 1 layer has 1 turn,and parallel number is 2. Each diode is connected with eachsecondary-side winding, as shown in Figure 1.

Figure 7 shows the definitions for each physical parameterof magnetic transformer which have EI core. From thesedefinitions, the following equation can be obtained:

𝑤2= 2𝑤1+ 𝑔,

𝑝 = 𝑤2+ 2𝑔,

𝑏 =𝑙e2− ℎ − 𝑝,

𝑐 = 2 (𝑏 + 𝑝) ,

𝑉e = 𝑎𝑏 ⋅ 2 (𝑏 + 𝑝 + ℎ) = 𝐴e𝑙e,

(13)

where 𝑙e is effective magnetic path length, 𝐴e is effective corecross-sectional area, and 𝑉e is volume of core material. Inaddition, based on magnetic circuit of EI core transformer,the following equation can be derived:

𝐴e =(𝐶c𝐿11 (2𝜇s𝑙g + 𝑙e))

(𝜇0𝜇s𝑁p2)

, (14)

L1 (P)L2 (P)L3 (S)L4 (S)

L5 (S)L6 (S)L7 (P)L8 (P)

Core

Core

Figure 6: The winding arrangement of the prototype planar trans-former.

where𝐶c is correction coefficient, 𝜇0is absolute permeability,

𝑙g is air gap, 𝜇s is relative permeability of corematerial, and𝑁pis number of primary-side turns.

The restriction of this converter is considered. The phys-ical features of each component are shown in Figure 8.Target depth, width, height, and volume are written inthis figure. From these physical features, the limitation ofphysical parameters in magnetic transformer can be derivedas follows: 0 < 𝑎 < 15 − 2𝑝 and 𝑐 ≤ 16.5.

The relationship between 𝐵m and temperature can bedrawn from past experimental data, as shown in Figure 9. 𝐵mcan be derived as the following equation from experimentalresults:

𝐵m =(Vm𝑡m)(2𝑁p𝐴e)

. (15)

From this data, when the allowable temperature of magneticcore is set to 80∘C, 𝐵m is restricted from 32 to 36mT.

From Figures 10(a) and 10(b), the length of the windingin the primary-side and secondary-side can be calculated asfollows:

𝑙p = 4𝑎 + 3.5𝑏 + 15𝑤1 + 21.5𝑔,

𝑙s = 2𝑎 + 𝑏 + 8𝑤2 + 8𝑔.(16)

Each layer of resistance is shown here:

𝑅lay 𝑖 = 𝑟lay 𝑖𝑙p,

𝑅lay 𝑗 = 𝑟lay 𝑗𝑙s,(17)

where 𝑖 = 1, 2, 7, 8 and 𝑗 = 3, 4, 5, 6. Also, 𝑟lay 𝑖 and 𝑟lay 𝑗 arewinding resistance per unit length. The winding resistanceper unit length 𝑟lay is obtained from simplified Maxwell 3Dwinding model, as shown in Figure 11. The solver of thesimulation is eddy current. The thickness of L1 and L8 is0.073mm, that of L2 and L7 is 0.055mm, and that of L3–6 is


a

b p

h

c

Primary winding

p

g g g

Secondary winding

b/2

b/2

Magnetic core

x

yz

x

yz

x yz

w1w1

w2

5 100(mm)

3.5 70(mm)

3.5 70

(mm)

le

Ae

lg

Figure 7: The definitions for each physical parameter of magnetic transformer.

CoreGaN-HEMT:

Driver:

Current transformer:Output capacitor:

Input capacitor:

Resonant capacitor:

Winding

Target

Volume: 3.3 cm3

Height: 7.0 mmWidth: 28.4 mmDepth: 16.5 mm

0.8 mm × 1.6 mm

1.6 mm × 3.2mm

1.25 mm × 2mm

8.1 mm × 7.8mm

4mm × 4mm

4mm × 1.6 mm

Diode: 3.9mm × 6.4mm

Figure 8: The physical features of each component.

36mT35.5 mT34mT32mT

40

50

60

70

80

90

100

Tem

pera

ture

(∘C)

30 35 4025Bm (mT)

fs = 4.55MHzfs = 5.00 MHz

fs = 3.85 MHzfs = 4.17 MHz

Figure 9: 𝐵m versus temperature derived by past experimental data (core: SY22, 𝑁p = 2 turns, 𝑉e = 1159.62mm3, 𝐴e = 30.8mm2, and 𝑙g =0.5mm).


a

b

50 10

(mm)

lp

(a) The primary-side length

a

b

0 2010

(mm)

ls

(b) The secondary-side length

Figure 10: The length of the winding.

x y

z

x y

z

3 60

(mm)

(a)

L1 (P)

L2 (P)

L3 (S)

L4 (S)

L5 (S)

L6 (S)

L7 (P)

L8 (P)

x y

z Current L3_2Current L4_2

Current L5_2Current L6_2

2 40

(mm)

(b)

Figure 11: Simplified Maxwell 3D winding model.

0.090mm.These thicknesses are the real copper foil thicknessof PCB board. Also, each material is copper where bulkconductivity is 46 × 106 S/m.The frequency of the simulationis 5MHz. The current direction of this simulation is shownin Figure 11(b). The results of the simulation are shown inFigure 12. These results indicate the relationship betweenthe winding width 𝑤 and the winding resistance per unit

length 𝑟lay. From the approximate of the plot, the relationshipequation can be obtained as follows:

𝑟lay 1,8 = 0.7047𝑤−0.605

,

𝑟lay 2,7 = 0.9195𝑤−0.667

,

𝑟lay 3,4,5,6 = 0.8040𝑤−0.635

.

(18)


L1, 8L2, 7L3–6

0.000.100.200.300.400.500.600.700.800.901.00

r lay

(mΩ

/mm

)

2 3 4 51w (mm)

Figure 12: The results of the simulation: the relationship betweenthe winding width𝑤 and the winding resistance per unit length 𝑟lay .

As mentioned above, layers 1, 2, 7, and 8 are assigned toprimary-side windings, while layers 3, 4, 5, and 6 are assignedto secondary-sidewindings.Theprimary-side turn number is3 turns, 1 layer has 2 turns, and parallel number is 2. Also, thesecondary-side turn number is 1 turn, 1 layer has 1 turn, andparallel number is 2. Thus, each winding resistance is

𝑅pri =(𝑅lay 1 + 𝑅lay 2) (𝑅lay 7 + 𝑅lay 8)

(𝑅lay 1 + 𝑅lay 2) + (𝑅lay 7 + 𝑅lay 8),

𝑅sec =𝑅lay 3𝑅lay 4

𝑅lay 3 + 𝑅lay 4=

𝑅lay 5𝑅lay 6

𝑅lay 5 + 𝑅lay 6.

(19)

The copper loss of planar transformer𝑃copper can be estimatedas follows:

𝑃copper = 𝑅pri𝐼12+ 2𝑅sec𝐼2

2, (20)

where 𝑅pri and 𝑅sec are winding resistance of primary-side and secondary-side winding, respectively. 𝐼

1and 𝐼2are

root mean square (RMS) current of the primary-side andsecondary-side winding.

Core loss of planar transformer 𝑃core and total power loss𝑃sum can be derived as follows:

𝑃core = 𝐶m𝑓s𝛼𝐵m𝛽𝑉e,

𝑃sum = 𝑃copper + 𝑃core.(21)

From these derived equations and FEMwithMaxwell 3D,the relationship between primary-side winding width𝑤

1and

𝑃sum can be obtained, as shown in Figure 13. The minimumtotal power loss can be found in this figure. From this figure,

1.2 1.4 1.6 1.8 2.01.0w1

3.63.73.83.94.04.14.24.34.44.54.6

Psu

m(W

)

lg = 0.0 mmlg = 0.5 mmlg = 0.6 mmlg = 0.7 mm

lg = 0.8 mmlg = 0.9 mmlg = 1.0 mm

Figure 13: 𝑤1versus 𝑃sum (fixed parameters: 𝐿

11= 250 nH, 𝜇s = 80,

𝑁p = 3, 𝜇0= 4𝜋 × 10

−7H/m, ℎ = 2mm, and 𝑔 = 0.5mm).

1.0 2.0 3.0 4.0 5.00.0a/b

3.4

3.6

3.8

4.0

4.2

4.4

4.6

4.8

5.0P

sum

(W)

lg = 1.0 mmlg = 0.8 mmlg = 0.6 mm

Figure 14: 𝑎/𝑏 versus 𝑃sum (fixed parameters: 𝐿11= 250 nH, 𝜇s = 80,

𝑁p = 3, 𝜇0= 4𝜋 × 10

−7H/m, ℎ = 2mm, and 𝑔 = 0.5mm).

𝑤1and air gap 𝑙g can be decided. The ratio of 𝑎 and 𝑏 can

be decided from Figure 14. It can be found that 𝐵m is lessthan the limitation of 𝐵m from Figure 15. From these figuresand equations, the physical parameters of planar transformercan be obtained as follows: 𝑎 = 5.15mm, 𝑏 = 3.7mm, 𝑐 =16mm, ℎ = 2mm,𝑝= 4.3mm,𝑔= 0.5mm,𝑤

1= 1.4mm,𝑤

2=

3.3mm, 𝑙g = 1mm, 𝑙e = 20mm, and 𝐴e = 19.07mm2.

4. Experimental Results

Some experiments are done with parameters and compo-nents, as shown in Table 2. The prototype MHz-level LLCresonant DC-DC converter with GaN-HEMTs is shown inFigure 16. In this prototype circuit, to suppress the power lossat diodes and use the board area effectively, diodes D

1and


1.0 2.0 3.0 4.0 5.00.0a/b

Less than Bm = 36mT

lg = 0.6 mmlg = 0.8 mmlg = 1.0 mm

18202224262830323436

Bm

(mT)

Figure 15: 𝑎/𝑏 versus 𝐵m (fixed parameters: 𝐿11= 250 nH, 𝜇s = 80,𝑁p = 3, 𝜇

0= 4𝜋 × 10

−7H/m, ℎ = 2mm, and 𝑔 = 0.5mm).

Winding

Diode

GaN-HEMT

Driver

Current transformer

Output capacitor

Input capacitor

Resonantcapacitor

Core

21mm

29mm

Height: 6.7 mm, volume: 4.1cm3

Figure 16: The prototype MHz-level isolated DC-DC converter with GaN-HEMTs.

Table 2: The parameters and components.

Name Value

Primary-side GaN-HEMT (EPC2001C)ON resistance: 𝑅DS(ON) 7mΩOutput capacitance: 𝐶oss 430 pF (𝑉DS = 50V)Reverse conduction voltage: 𝑉SD 1.8 V

Secondary-side diode (PDS1040L)Forward voltage: 𝑉F 0.3 VDiode resistor: 𝑅d 10mΩJunction capacitance: 𝐶j 1 nF

Transformer (core: SY22)

Magnetizing inductance: 𝐿m 195 nHLeakage inductance: 𝐿 lk1 40.82 nHTurns ratio: 𝑛 3Coupling coefficient: 𝑘 0.9095

Current transformer (B82801B) Secondary-side short inductance: 𝐿CT 7 nHResonant capacitor (C1608C0G2A152J) Capacitance: 𝐶r1, 𝐶r2 6 nF

Driver (LM5113)


(20V/div.)

�D2

(10A/div.)ir1

(50V/div.)�DS2

(20V/div.)

�D1

H: 40 ns/div.

(a) 𝑓s = 3.85MHz, 𝑉i = 42V, 𝐼o = 8.3 A, and 𝑉o = 12.647V

Dead time

(20V/div.)

�D2

(10A/div.)ir1

(50V/div.)�DS2

(20V/div.)

�D1

H: 40 ns/div.

(b) 𝑓s = 4.31MHz, 𝑉i = 48V, 𝐼o = 8.3 A, and 𝑉o = 12.333V

(20V/div.)

�D2

(10A/div.)ir1

(50V/div.)�DS2

(20V/div.)

�D1

H: 40 ns/div.

(c) 𝑓s = 4.81MHz, 𝑉i = 53V, 𝐼o = 8.3 A, and 𝑉o = 12.187V

Figure 17: The voltage and current waveforms of each switch with various input voltages.

D2are 2 parallel connections, respectively. The volume of the

prototype converter is 4.1 cm3.From Figure 17, the ZVS of the primary-side switches at

turn-on seems to be achieved during the dead time even ifthe input voltage is changed. The temperature distributionsof the prototype DC-DC converter are exhibited, as shownin Figure 18. Each core temperature can be found lower than60∘C.These resultsmean low flux density𝐵m. Each calculated𝐵m is as follows: 𝐵m = 28.17mT under𝑉i = 42V and 𝐼o = 8.3 Acondition, 𝐵m = 27.41mT under𝑉i = 48V and 𝐼o = 8.3 A con-dition, and𝐵m = 26.90mTunder𝑉i = 53V and 𝐼o = 8.3 A con-dition. Also, at minimum input voltage, the temperature ofwinding and diode seems to be higher than other conditions.

From these experimental results and datasheets of devices[18, 19], each power loss analysis is conducted, as shown inFigure 19. It can be seen that the target efficiency is achievedat𝑉i = 48 and 53V. On the other hand, it can be seen that thetarget efficiency is not achieved at worst condition 𝑉i = 42V.According to this analysis, the major reason of this problemis considered 6W of copper loss.

Also, from these experimental results, it can be found thatthe power density of the prototype converter is 24.4W/cc.

5. Conclusions

In this paper, the considerations of physical design and imple-mentation for wide regulation range LLC resonant DC-DCconverter with GaN-HEMTs are revealed. Also, the physicalparameters and size of the planar transformer are optimizedby using derived equations and FEM with Maxwell 3D.

The prototype LLC resonant DC-DC converter usingGaN-HEMTs is tested with open loop. The specificationsof this converter are that the input voltage is 42–53V, thereference output voltage is 12 V, the load current is 8 A,the maximum switching frequency is about 5MHz, and thetotal volume of the circuit is 4.1 cm3. ZVS operation of theprimary-side switches at turn-on can be achieved duringthe dead time even if the input voltage is changed. Theprototype DC-DC converter is obtained more than 90% ofpower efficiency at𝑉i = 48 and 53V, 𝐼o = 8A. Also, the powerdensity of the prototype converter is 24.4W/cc.

Competing Interests

The authors declare that they have no competing interests.


AB

D E (106∘C)C (102∘C)

120.0

108.1

96.2

84.3

72.5

60.6

48.7

36.8

25.0

(a) 𝑓s = 3.85MHz, 𝑉i = 42V, 𝐼o = 8.3 A, and 𝑉o = 12.647V

AB

D E (91.3∘C)C (88.2∘C)

120.0

108.1

96.2

84.3

72.5

60.6

48.7

36.8

25.0

(b) 𝑓s = 4.31MHz, 𝑉i = 48V, 𝐼o = 8.3 A, and 𝑉o = 12.333V

AB

D E (88.4∘C)C (85.0∘C)

120.0

108.1

96.2

84.3

72.5

60.6

48.7

36.8

25.0

(c) 𝑓s = 4.81MHz, 𝑉i = 53V, 𝐼o = 8.3 A, and 𝑉o = 12.187V

Figure 18: The temperature distributions (A: GaN, B: driver, C: winding, D: core, and E: diode).

5.93 5.07

4.08

0.971.15

1.40

3.713.27

3.46

(89.89%)11.82 W

(90.94%)10.08 W(90.62%)

10.59W

Switch conduction lossSwitch driving lossSwitch body diode lossDiode conduction loss

Diode reverse lossCore lossCopper loss

0

1

2

3

4

5

6

7

8

9

10

11

12

Plo

ss(W

)

Vi = 48V Vi = 53VVi = 42V

Figure 19: The power loss analysis of the experiment at 𝑃o = 100W.


References

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