RCAGLOBAL

Materials, Junctions, and Devices 3 Bipolar Transistors 22 MOS Field-Effect Transistors 39 Thyristors 56 Silicon Rectifiers 77 Other Solid-state Diodes 83 Receiver Tuner-Circui t Applica t iions 88 Low-Frequency Amplification 112 R F Power Amplification and Generation 133 T V Deflection 175 Power Switching and Control 189 DC Power Supplies 228 Testing and Mounting 237 RCA SK-Series Solid-State Replacement Devices 2 5 8Symbols 270 Selection Charts 276 Interpretation of Data 282 Technical Data for Small-Signal Bipolar Transistors 284 Technical Data for MOS Field-Effect Transistors 363 Technical Data for LOW- and Medium-Frequency Power

Transistors 389 Technical Data for RF Power Transistors 538 Technical Data for Thyristors 586 Technical Data for Silicon Itectifiers and Other Solitl-State

Diodes 652 Char t of Discontinued Transistors 659 Outlines 6 6 6Mounting Hardware 684 Circuits 6911 Other RCA Technical Manuals 758 Index to RCA Solitl-State Devices 759 Index 766

Information furnished by RCA is b e l i e v e dLo be accurate a n dr e l i a b l e111, \ \~- cvrr, nu responsibility is nsm~mrrl by HCA for it4 u s e ; nor for a n y iufrince- men& of patents or oLhcr rirrhtn o i thi~-cl parties which may re3ralt from its u a c No license is ~ r n n t e r l I,y implicalion 01. otherwise under any pntcnt air

pntent rights of R C h . .

Materials, Junctions, and Devices

S OIJD-STATE devices a re small but versatile units that can per-

form an amazing variety of control functions in electronic equipment. Lilte other electron devices, they have the ability to control almost instantly the movement of charges of electricity. They a re used a s rectifiers, de tec tors , ampl i f ie r s , osci l la tors , e lectronic swi tches , mixers , a n d modulators.

In addition, solid-state devices have many , important advantages over other types of electron devices. They a re very small and light in weight (some a r e less than an inch long and weigh just a fraction of a n ounce). They have no filaments o r heaters, and therefore require no heating power or warm-up time. They consume very little poweiJThey a re solid in construction, extremely rugged, f ree from microphonics, and can be made impervious to many severe environmental conditions. The circuits required f o r their operation are usually simple.

i SEMlCONDUCTOR MATERIALS ' Unlike other electron devices, which

' depend for their functioning on the ! How of electric charges through a

vacuum or a gas, solid-state devices make use of the flow of current

I in a solid. In general, all materials I may be classified in three major I categories-conductors, semiconduc-

tors, and insulators-depending upon their ability to conduct a n electric

current. As the name indicates, a semiconductor material has poorer conductivity than a conductor, but better conductivity than an insulator3

The materials most often used in semiconductor devices a r e germanium and silicon. Germanium has higher electrical conductivity (less resistance to current flow) than silicon, and is used in devices intended for applications tha t require low voltage drops at high currents and in some small-signal transistors. Silicon is more suitable for high-power devices than germanium One reason is that i t can be used a? much higher temperatures. I n general, silicon is preferred over germanium because processing techniques yield more economical devices. As a result, today, silicon tends to supersede germanium in almost every type of application, including the small-signal area, unless a very low device voltage drop is required.

Resistivity The ability of a material to con-

duct current (conductivity) is directly proportional to the number of free (loosely held) electrons in the material. Good conductors, such a s silver, copper, and aluminum, have large numbers of f ree electrons; their resistivities a r e of the order of a few millionths of a n ohm-centimeter. Insulators such a s glass, rubber, and mica, which have very few loosely held electrons, have resistivities of several million ohm-centimeters.

4 RCA Transistor, Thyristor, & Diode lvlarlual

Semiconductor materials lie in the range bctwcen thcse two cs t ren~es , as shown in Fig. 1. Pure germanlum has a resistivity of 60 ohm-centimeters. Pure silicon has a considerably h i ~ l i e r resistivity, in the order of 60,000 ohm-centimeters. As used in semicontluctor devices, l~owever, these nlaterials contain carefully controlled amounts of certain impurities

INCREASING RESISTIVITY - IO-ti I lo3 lo6

OHM-CM I-{

COPPER GERMANIUM SILICON GLASS - INCREASING CONDUCTIVITY

Fi.?. I-Rcsis/i~.ity o f typical corrdrrc:or, ss~rticortdrrcrors, and ir~sulator.

which redwe their resistivity to about 2 ohm-centimeters a t room temperature (this resistivity decreases rapidly as temperature rises).

1- Carefully prepared semiconductor n~a te r ia l s hgve a crystal structure. In this typc of structure, which is called a latticc, the outer or valence ~ lec t rons of individual atoms a re tightly 1)ound to tlie electrons of ad- jaccnt atoms in electron-pair bonds,) as shown in Fig. 2.- Because such a

.ECTRON -PAIR BONDS

Fig. 2-Cr)~srol 1arr;te srrrrcrrtre. d 1. ' ( structure has no l o sely hcld electrons, semiconductor materials a r e poor conductors ander normal conditions. In order to separate the electron-pair bonds and provide f ree electrons for electrical conduction,

it would be necessary to apply h i ~ h tcniperatures o r s t rong electric fields.

Another way t o alter the lattice structure and thereby obtain free electrons, however, is t o add small nnlounts of other elements having a difFerent atomic structure. By the addition of almost infinitesimal amounts of such other elements, called "impurities", the basic electrical properties of pure semiconductor materials can be modified and controlled. The ratjo of impurity to the scmiconductor material is usually extremely small, in the order of one par t in ten million. -.

When the impurity clenlents a re added t o the semiconductor material, impurity atoms take the place of semiconductor atoms in the lattice structure.3 If the impurity atoms added have the same number of valence electrons a s the atoms oL the original semiconductor material, they fit neatly into the lattice, forming the required number of electron-pair bonds with semiconductor atoms. In this case, the electrical properties of the material a r e essentially unchanged.

When the inlpurity atom Kas onc more valence electron than the semiconductor atom, however, this extra electron cannot form a n clcctron- pair bond because no adjacent vtr- lcnce electron is available. The exccss electron is then held very loosely by the atom, a s shown in Pig. 3, and

Materials, Junctions, and Devices 5

rcquires only slight excitation to break away. Consequently, the presence of such excess electrons makes the material a better conductor, i.e., i t s res i s tance t o - c u r r e n t f low i s reduced. " Impurity elements which a r e added to germanium and silicon crystals to provide excess electrons include ar- senic and antimony. When tliese elements a re introduced, the resulting material is called n-type because the excess free electrons have a negative charge. ( I t should be noted, however, that the negative charge of the electrons is balanced by a n equivalent positive charge in the centcr of the impurity atoms. Therefore, the net electrical charge of the semiconductor material is not changed.)

A diffcrcrit e r e c t is produced when an impurity atom having one less valence electron than the semiconductor atom is substituted in the lattice structure. Although all the valence electrons of the impurity atom form electron-pair bonds with electrons of neighboring semiconductor atoms, one of the bonds ill the lattice structure cannot be completed because the impurity atom lacks the final valence electron. As a result, a vacancy or "hole" exists in the lattice, a s shown in Fig. 4. An electron from a n adjacent electron-pair bond may then absorb enough energy to brealc i ts bond and move through the lattice to fill the hole. A s in the

IMPURITY/ VACANCY ATOM (HOLE)

Fig. 4- lattice strrrcrttre o f p-type nra terial.

case of excess electrons, the presence of "holes" encourages the flow of electrons in the semiconductor material; consequently, the conductivity is increased and the resistivity is I reduced. I

The vacancy or hole in the crystal 1 s tructure is considered to have a positive electrical charge because i t represents the absence of a n electron. (Again, however, the net charge of the crystal is unchanged.) Semi- conductor material which contains these "holes" or positive charges is called p-type material. P-type materials a re formed by the addition of aluminum, gallium, or indium.

Although the difference in the chen~ical composition of n-type and p-type materials is slight, the differences in the electrical characteristics of the two types a r e substantial, and a re very important in the operation of solid-state devices.

P-N JUNCTIONS When n-type and p-type materials

a re joined together, a s shown in Fig. 5, an unusual but very important phenomenon occurs a t the interface

p-n JUNCTIOH

- - - -

SPACE-CHARGE REGION

Fig. 5-Z~rteracrion o f Iroles and electrons at p-n jurrcliot~. I

where the two materials meet (called the p-n junction). An interaction takes place between the two types

.'of material a t the junction a s a result of the holes in one material and the excess electrons in the other.

When a p-n junction is formed, some of the free electrons from the n-type material diffuse across the junction and recombine with holes in

6 RCA Transistor, Thyristor, & Diode Manual

the lattice structure of the p-type niaterial; similarly, some of the holes in the p-type material diffuse across the junction and recombine with f ree electrons in the lattice structure of the n-type material. This interaction or diffusion is brought into equilibrium by a small space-charge region (sonletimes called the transition region or drplction layer). The p-type material thus acquires a slight negative charge and the n-type material

>acquires a slight positive charge. I Tliernlal energy causes charge car- 7 , : riers (electrons ant1 holes) to diffuse

side of the p-n junction to side; th i s flow of charge

carriers is called dimusion current. AS a result of the diffusion process, however, a potential gradient builds up across the space-charge region. This potential gradient can be represented, a s shown in Fig. 6, by a n imaginary battery connected across the p-n junction. (The battery ,symbol

JUNCTION

j IMAGINARY - + SPACE-CHARGE EQUIVALENT

BATTERY

Fis. 6-Pofrtttial grnllieltt across space- clrarge region.

-his used merely to illustrate internal effects; the potential i t represents is not directly measurable.) The potential gradient causes a flow

ELECTRON .FLOW _3

(a) REVERSE BlAS

of charge carriers, referred to a s drift current, in the opposite direction to the diffusion current. Under equilibrium conditions, the diffusion current is exactly balanced by the drift current so t h a t the net current across the p-n junction is zero. In other words, when no external current or v o l t a ~ e is applied to the p-n junction, the pote~ltinl gradient fornis an enerRy barrier t h a t prevents further dilfusion of charge carriers across the junction. In effect, elcc- trons from the n-type material tha t tend to diffuse across the junction a re repelled by the slight negative charge induced in the p-type material by the potential gradient, and holes from the p-type material a r e repelled by the slight positive charge induced in the n-type material. The potential gradient (or energy barrier, as i t is sometimes called), therefoce, prevents total interaction between the two types of matetials, and thus preserves the differences in their characteristics.

CURRENT FLOW When a n external battery is con-

nected across a p-n junction, the amount of current flow is determined by the polarity of the applied voltage and i ts effect on the space-charge region. In Fig. 7 ( a ) , the positive terminal of the battery is connected t o the n-type niaterial and the negative terminal t o the p-type material. I n this arrangement, the f ree electrons in the n-type material a r e attracted toward the positive terminal of the battery and away from the junction. At the same time, holes from the

ELECTRON FLOW

f-

(b) FORWARD BlAS

Fig. 7-Elecrrotr current flow CI biased p-rc jrrncfions.

Materials, Junctions, and Devices

p-type material a re attracted toward the negative terminal of the battery tMILLIAMPERES, and away from the junction. A s a result, the space-charge region a t the junction bccolnes effectively wider, and the potential gradient increases -REVERSE BIAS until i t approaches the potential of FORWARD BIAS--+ the external battery. Current flow is then extremely small because no voltage difference (electric field) exists across either the p-type or the 1

REVERSE CURRENT

n-type region. Under these condi- (MICROAMPERESI tions, the p-n junction is said to be reverse-biased. Fig. 8-Voltage-crrrretrt characteris~ic for

In Fig. 7(b) , the positive terminal a p-tr jrtncliorr. of the external battery is connected to the p-type material and the negative terminal to the n-type material. In this arrangement, electrons in the p-type material near the positive terminal of the battery break their electron-pair bonds and enter the battery, creating new holes. At the same time, electrons from the negative terminal of the battery enter the n-type material and diffuse toward the junction. As a result, the space- charge region becomes effectively narrower, and the energy barrier decreases to an insignificant value. Ex- cess electrons from the n-tvne material can then penetrate tkk space- charge region, flow across the junction, and move by way o£ the holes in the p-type material toward the positive terminal of the battery. This electron flow continues a s long a s the external voltage is applied. Un- der these conditions, the junction is said to be forward-biased.

The generalized voltage-current characteristic for a p-n junction in Fig. 8 shows both the reverse-bias and forward-bias regions. In the forward-bias region, current rises rapidly a s the voltage is increased and is quite high. Current in the reverse-bias region is usually much lower. Excessive voltage (bias) in either direction should be avoided in normal applications because excessive currents and the resulting high temperatures may per~nanently damage the solid-state device.

TYPES OF DEVICES The simplest type of solid-state

device is the diode,' which is represented by the symbol shown in Fig. 9. Structurally, the diode is basically a p-n junction'similar to those shown in Fig. 7.;The'n-type material which serves as'. the negative electrode is referred to a s the cathode, and the p-type material which serves a s the positive electrode is referred to a s the nnode.",The arrow symbol used for the anode represents the direction of "conventional current flow";

Fig. 9-Schettrafic sytnbol for a solid- sfale diode.

electron current flows in a direction opposite to the arrow.

%ecause the junction diode conducts current more easily in one direction than in the other, i t is an effective rectifying device. If an ac signal is applied, a s shown in Fig. 10, electron current flows freely during the positive half cycle, but little o r no current flows during the negative half cycle.

One of the most widoly used types of solid-state diode is the silicon rectifier. These devices a re available in a wide range of current

R C A Transistor, Thyristor, & Diode Manual

INPUT SIGNAL mLoAD 'L'

Fi,q. 10-Sirrrple diode recrilyirrg circliil.

capabilities, ranging from tenths of a n ampere to several hundred amperes or more, and a r e capal)le of operation a t voltages a s high a s 1000 volts or more. Parallel and series arrangements of silicon rectifiers permit even fur ther extension of current and voltage limits: Characteris- tics and applications of these devices a r e discussed in detail in the section on Silicorl llectifiers.

Several variations of the basic junction diode structure have been dcvcloped for use in special applications. One of the most important of these developments is the tunnel diode, which is used for amplification, switching, and pulse generation, This diode and other special types c vnractor, volt:tjie-reference, and compensating diodes) a r e tle- scribetl in the section on Other Solid-State 1)iodes.

When another Inyer is addcd to a semicorldnctor diode to form three l:lyers (two junctions), n device is produced xvhich provides power or voltage amplification. The resulting device is called a bipolar tr;u~sistor. The three regions of the device a re called the cmitter, the b:ise, and the col lectoj a s shown in Fig. I l ( a ) . In normal operation, the emitter-to- base junction is biased in the forward direction, and the collector-to- base junction in the reverse direction.

Uiffcrellt symbols a re usetl for n-p-n ant1 p-n-p transistors to show the tlifFercnce in the direction of current flow in the two types of devices. In the n-p-n transistor shown in Fig. l l ( b ) , electrons flow from the emitter to the collector. In the p-n-p transistor shown in Fig. l l ( c ) , electrons flow from the collector to the emitter. In other wonls, the direction of

/ \A - ,G, 1

dc electron current is always ol)l)o- site to that of the arrow on the emitter lead. (As in the case of scmi- contluctor tliodes, the arrow intlicntes the direction of "conventional current flow" in the circuit.)

The first two letters of the n-p-n and p-n-p designations intlicate the respective polarities of the voltages applied to the emitter and the collector in normal operation. In

l a 1 FUNCTIONAL DIAGRAM

EMITTER COLLECTOR

(. b\ n - p - n TRANSISTOR

t c l p -n - p TRANSISTOR

Fig. 11-F~rrictiorral rlingrarrl arid rclie- ~riatrc syrrrbols f o r hipolar trori~istots.

an n-p-n transistor the emitter is made negative with respect to b o K f l i e c o l t o r and the base, and the collector is made positive with respect to both- the emitter and the base. In a p-n-p transistor, the emitter is made positive with respect to both the collector and the base, and the collector is made negative with respect to both emitter and base.

The transistor, which is a three- element device, can be used for a wide variety of control functions, including amplification, oscillation, and frequency conversion. A conlplete description of the fabrication, electrical characteristics, and basic circuits of 1)ipolar transistors is given in the section on Bipolar Transistors.

; The field-effect transistor (I'ET) 1 is another type of solid-state de- 1 vice that is becoming increasingly

popular in electronic circuits. Func- I tionally, this type of transistor dif- / -

\ '


fcrs fro111 the bipolar transistor in that current flow through the device is controlled by variation of the electric field cstahlished by a control voltage rather than by vari-

- ation of the current injected into the base terminal. Field-effect transistors exhibit many of the electrical characteristics of electron tubes, but still retain the inherent advantages of solid-state devices (e.g., small size, low power consumption, and mechanical ruggedness). On the basis of structural and functional differences, these devices a r e classified a s either junction-gate field-effect transistors ( J F E T ) o r metal-oxide- semiconduclor field-effect transistors (MOS/FET). Although in both types the conduction current is controlled by a n electric field, the electrical characteristics of these devices differ significantly.

Fig. 12 shows the schematic symbols for both n-channel and p- channel ction-gate field-effect tmnsistors!?The gate, source, and

DRAIN 0

DRAIEI

0

SOURCE b n- CHANNEL p -CHANNEL

Fix. 12-.Ycl1cr11ntic SJ.III~OIE for jr111cti011- nore field-eflect trtrfuistors (JFET) .

drain electrodes of these devices a re c!quivalent to the base, emitter, and collector electrodes, respectively, of bipolar transistors. A signal volttige applied to the gate electrode controls the conductivity of the semiconductor layer iminedialely below the gate, between the source and drain terminals. The n-channel type, which is analogous to an n-p-n bl- polar transistor, is operated with

the drain a t a positive potential with respect to the source terminal. In the schematic symbol, this type is indicated by an arrow in the gate lead tha t points into the device. The drain potential f o r the p- channel type, which i s analogous t o a p-n-p bipolar transistor, is negative with respect t o the source terminal. In the schematic symbol fo r this type, the arrow in the ga te lead points away from the device.

Fig. 13 shows the schematic symbols f o r both n-channel and p-channel versions of the basic classes of MOS field-effect transistors, i.e., enhancement types and depletion types. The arrow used in the schematic symbol to indicate whether a device is a n n-channel type (points inward) o r a p-channel type (points outward) is shown in the lead from the substrate terminal. The sub- s trate terminal is connected to the semiconductor substrate (also referred to us the active "bulk") on which the transistor is fabricated.

The technology f o r MOS field- effect transistors is more versatile than tha t f o r junction-gate types. Specific categories of MOS field- effect transistors have been designed with unique characteristics t h a t make them ideal f o r linear (analog) and digital applications. F o r example, tho depletion type is frequently used in linear applications, and the enhancement type is ideal f o r digital applications. An enhancement type of MOS field-effect t ransistor is equivalent to a "normally open" switch, a s indicated in the schematic symbol by the gaps in the source-to-drain path. The depletion type, however, is normally conductive and its source-to-drain path is shown continuous in the schematic symbol. The enhancement-MOSI FET technology is being used increasingly in the fabrication of integrated circuits fo r digital application, particularly for large-scale- integration (LSI) circuits. A comprehensive description of MOSJFET devices is given in the section on MOS Field-Effect Transistors.

RCA Transistor, Thyristor, & Diode ManuaI

@RAIN GATE @ AIN

SUBSTRATE SUBSTRATE GATE

SOURCE SOURCE

n- CHANNEL p- CHANNEL

OEPLETION TYPES

Q DRAIN 0 DRAIN

SUBSTRATE SUBSTRATE GATE GATE

6 SOURCE

n-CHANNEL

ENHANCEMENT TYPES

Fig. 13-Sclrc~rtrotic sytrrhols for r~ce~nl-oxiiie-sertricot~cl~ictor field-elfec! tvctrrsjsrors (MOSIFET).

,' ,-' When alternate layers of 11-type ! and n-type semiconductor materials I are arranged in a series array, vari-

ous types of thyristors can be produced. The tern1 thyristor is the

t generic name for solitl-state devices tha t have electrical characteristics similar to tllosc of thyratron tubes. The three l~asic types of thyristors a r e the bidirectional trigger diode called the tliac, the reverse blocking triode called the silicon controlled rectificr o r SCR, and the bidirectional triode thyristor, called the triac. The diac, shown in Fig. 14, is a two-electrode, three-layer device having the satne doping level a t both junctions and a "floating" base. The device conducts current in

&I

Fi,?. I4-Jrotcriorr dingrarir (a) nrrcl sche- ~rtaric s)~rrrbo[ ( 6 ) for a iliac.

either direction af ter the applied voltage exceeds a certain value called the "breakover voltage." The SCR is a three-electrode, four-layer device, a s shown in Fig. 15. The SCR

ccR(FGATE ANODE (CASE)

Fig. 15-J~irrctiorr diagrar~r (a ) arrd sclre- rrratic syrrrbol ( b ) for a silicor~ coritrolled

rectifier or SCR.

behaves as a conventional rectifier to bloclr current flow in the reverse direction and a s a transistor switch in the forward direction to first block current and then conduct / through the device when a current


pulse of suflicient magnitude is ap- for either direction of voltage applied to the gate electrode. The plied to the main terminals. The triac i s a three-electrode, five-layer schematic symbols for these thyris- device, a s shown in Fig. lG, which tor devices a re also shown in Figs. exhibits the forward-blocking- 14, 15, and 16. A complete descrip- forward-conducting voltage-current tion of these devices is given in the characteristic of the SCR structure section on Thyristors.

MAIN

(a)

PMAIN TERMINAL I

Fia6-Jutrc f ion diagram ( a ) and sche- nratic symbol (b) for a triac.

Bipolar Transistors

*A p-11 jur~ct ion I)insctl in the re- tlirection to provitlc a low- rc s i s t a~~ce verse direction is cquivalent to i npu t circuit, and t h e rigllt-11alld

a high-resistance ele lnrnt (low (p- n ) junction is biased ill the rc- cu r r en t f o r a given apl)lied volt- verse clirection to provide a high- a ~ o ) , while a junction biased in t h e resistance ou tpu t cii.cuit. fortvard direction i s equivalent t o &? Electrons flow easily f rom the left- a lot\,-resistance c lement ( h i ~ h cur- hantl n- type region to tlle celitcr p- r e n t f o r a given a111)lied voltage). t ype region a s a resul t of t he forwartl - Because the polver devcl01)ed by a biasing. Most of thcs i r .c lcc t~~ons tlif- given cu r r en t is g r e a t e r in a high- f u s e through the th in p-type region, resistnnce element t h a n in a low- ho\~rever, and a r e a t t r ac t c t [ I,y the resistance element (P = I"), ),OXVer positive potential of the l)at- ga in can be obtained in a s t ruc tu re tery across tllc riRllt-llantl junction. containing two such resistance ele- 1, ,,ractical devices, appl.oxilllately men t s if t he cu r r en t flow is not $15 to 99.5 cent of t l lc c ~ ~ c c ~ r o l l rnntcrially reduccd. A tlcvice con- current reacllcs tllc right-llallct 11-

t a i n i n r two p-n junc l io l~s biased ill type region. ~ l ~ i ~ high of opposite directions is called a junc- cu r r en t penetration provitlcs power tion o r bipolar t rans is tor . ga in in the high-resistance ou tpu t

Such a two-junction tlcvice i s circuit and is t he basis f o r t rans is tor shown in Fig . 17. The thiclc entl l aye r s amplification capability. a r e made of t he s ame type of mate- #? - The operation of p-n-p tlcvices is r ia l (n - type ill this case) , and a r e similar to that shown fo r the n-p-n separatctl by a very t.11i1i layer of t h e device, except t h a t t h e bias-voltage opposite lnaterial (p-'ype in a r e ~*e\rersc([, and electron- t he (levice s l ~ o w n ) . BY means of t h e current flOIV is i n opposite dircc-

tion. (Rlany tliscussions of semicon- OUTPUT tluctor theory a s sume t h a t t he "holes"

in semicontluctor mater ia l cons t i tu te t hc cha rge carr iers in p-n-p tlevices, and tliscuss "hole currents" f o r these

ELECTRON devices ant1 "electron currents" f o r n-p-n devices. Other t ex t s discuss neither hole cu r r en t no r electron current , bu t r a the r "conventional cu r r en t flow", which i s assumed to travel

Fi:. 17-AII 11-p-11 .rtrlrcllrrc biased for poivcr gairi. through a circuit in a tlirection frorn

t h e posiLive terminal of the external e s t e r ~ l n l hntterics, t h e lcft-hand (n-p) ba t t e ry back t o i t s negat ive terminal. junction is biased i n t he fo rward F o r t he s a k e of sinlplicity, this tlis-

Bipolar Transistors 13

cussion will be restricted to the con- BASE CONTACT cept of electron current flow, which travels from a negative to a positive SOLDER

terminal.) COLLECTOR

SOLOE- CONTACT EMITTER .CONTACT REGION

DESIGN AND FABRICATION

The ultimate aim of all t ransistor fabrication techniques is the construction of two parallel p-n junctions with co~ltrolled spacing hetween the junctions and co~ltrolled impurity levels on both sides of each junction. A variety of structures has been dcvcloped in the course of transisto

r

evolution. The earliest transistors made were

of the point-contact type. I n this type of structure, two pointed wires were placed next to each other on a n n-type block of semiconductor material. The p-n junctions were formed by electrical pulsing of the wires. This type has been superseded by junction transistors, which a re fabricated by various+loy, diffusion, and crystal-growth techniques.

I11 grown-junction transistors, the impurity content of the semiconductor material is changed during the growth of the original crystal ingot to provide the p-n-p or n-p-n regions. Thc grown crystal is then sliced into a large number of small-area devices, and contacts a r e made t o each region of the devices. Fig. 18(a ) shows a cross-section of a grown- junction transistor.

In alloy-junction transistors, two small "dots" of a p-type or n-type impurity element a r e placed on opposite sides of a thin wafer of 11-type or p-type sen~iconductor n~nter ial , ~.espectively, a s sho\vn in Fig. 18(b) . After proper heating, the impurity "tlots" alloy with the scmiconductor material to form the regions for the emitter and collector junctions. The base connection in this structure is made to the original semiconductor wafer. ,.;

The drift-field transistor is a modified al loy-junct io~~ device in which the in~pur i ty concentration in the base wafer is diffused or graded, a s

(a) GROWN-JUNCTION TYPE

ORIGINAL SEMICONDUCTOR MATERIAL-BASE

EMITTER DOT

(b) ALLOY.JUNCTION TYPE

DIFFUSED

CONTACT . . - . b - 7 \ 7

SOLDER UNDIFFUSED BASE REGION-ORIGINAL

COLL&TOR SEMICONDUCTOR DOT MATERIAL

(c) DRIFT-FIELD T Y P E Fig . 18-Cross-sectioru of jrcrrclion Iran-

sistors.

shown in Fig. 18(c). Two advantages a r e derived from this structure: (a ) the resultant built-in voltage or "drift field" speeds current flow, and (,b) the ability to use a heavy impurity concentration in the vicinity of the emitter and a light concentration in the vicinity of the collector makes i t possible to minimize capacitive charging times. Both these advantages lead to a substantial extension of the frequency performance over the alloy-junction device.

T h e diffused- junct ion t r a n s i s t o r represents a major advance in tran- ,sister technology because increased control over junction spacings and impurity levels makes possible significant inlprovements in transistor performance capabilities. A cross- section of a single-diffused "hometaxial" structure is shown in Fig. 19 (a) . Hometaxial transistors a r e fabricated by sinlultaneous diffusion of impurity from each side of a homo- geneously doped base wafer. A mesa or flat-topped peak is etched on one side of the wafer in a n intricate design to define the transistor emitter

RCA Transistor, Thyristor, & Diode Manual

and expose the base region for connection of metal contacts. Large amounts of heat can be dissipated from a hometaxial structure through the highly conductive solder joint between the semiconductor material and the device package. This structure provides a very low collector resistance.

Double-diffused transistors have a n additional degree of freedom for selection of the impurity levels and junction spacings of t h e base, emitter, and collector. This structure provides high voltage capability through a lightly doped collector region without compromise of the junction spacings which determine device frequency response and other important characteristics. Fig. 19(b) shows a typical double-diffused transistor; the emitter and base junctions a r e diffused into the same side of the original sen~iconductor wafer, which serves a s the collector. A mesa is usually etched through the base region to reduce the collector arca a t the base-to-collector junction and to provide a stable sen~iconductor surface.

Double-diffused planar transistors provide the added advantage of protection or passivation of the emitter- to-base and collector-to-base junction surfaces. Fig. 19(c) shonrs a typical double-diffused planar transistor. The base and emitter regions terminate

EMITTER METAL

BASE CO?TACT \ ,EMITTER CONTACT

.DIFFUSED EMITTER DIFFUSED .-8-----

BASE' C

r PACKAGE SOLDER

DIFFU&D COLLECTOR UNDiFFUSED BASE (LOW RESISTANCE) (HOMOGENOUS)

(a1 SINGLE-DIFFUSED "HOMETAXIAL" TYPE

coNTAcT EMITTER CONTACT ;DIFFUSED EMITTER

METAL FILM^, - -

-

'ONTACT UNDIFFUSED SOLDER PACKAGE COLLECTOR

(b) DOUBLE-DIFFUSED TYPE

Fig. 19-Cross-secrioirs

a t the top surface of the semiconductor wafer under the protection of a n insulating layer. Photolitho- graphic and masking techniques a r e used to provide f o r diffusion of both base and emitter impurities in selective areas of the semiconductor wafer.

In triple-diffused transistors, a heavily doped region diffused from the bottom of the semiconductor wafer effectively reduces the thickness of the lightly doped collector region to a value dictated only by electric-field considerations. Thus, the thickness of the lightly doped or high-resistivity portion of the collector is minimized to obtain a low collector resistance. A section of a triple-diffused planar structure is shown in Fig. 19(d).

Epitaxial transistors diner from diffused structures in the mafiner in which the various regions a re fabricated. Epitaxial structures a re grown on top of a semiconductor wafer in a high-temperature reaction chamber. The growth proceeds atom by atom, and is a perfect extension of the crystal lattice of the wafer on which i t is grown. In the epitaxial-base transistor shown in Fit. 20(a) a lightly doped base region is deposited by epitaxial techniques on a heavily doped collector wafer of opposite-type dopant. Photolitho- graphic and masking techniques and

METAL FILM

SILICON DIOXIDE

DIFFUSED UNDIFFUSED BASE

(c ) DOUBLE-DIFFUSED PLANAR TYPE

SILICON DIOXIDE UNDIFFUSED

BASE PACKAGE DIFFUSED

COLLECTOR HEAVILY DOPED

(d) TRIPLE-DIFFUSED PLANAR TYPE


DIFFUSED EMITTER through a typical overlay emitter CONTACT METAL region.

HEAVILY After fabrication, individual tran- DOPED sistor chips a re mechanically sepa-

COLLECTOR PACKAGE rated and mounted on individual SOLDER headers. Connector wires a re then ,

'r (0) EP ITA X IA L- BAS E T YP E bonded to the metalized regions, and DIFFUSED EMITTER each unit is encased in plastic or a

EPITAXIAL . hermetically sealed enclosure. In COLLECTOR ,,,,,,,, power transistors, the wafer is us-

DOPED ually soldered or alloyed t o a solid PACKAGE

SOLDER COLLECTOR metal header to provide for high (b) D UA L- EP ITA X IA L- LA YE R T Y P E thermal conductivity and low-resis-

SILICON tance collector contacts, and low- resistance contacts a re soldered or metal-bonded from the emitter o r

,,,,,,,,, base metalizing contacts t o the ap- EMITTER propriate package leads. This pack-

aging concept results in a simple BASE structure t h a t can be readily attached

R to a variety of circuit heat sinks and can safely withstand power dissipa-

(c) "OVERLAY" T Y P E tions of hundreds of wat t s and cur- Flg. 20-Cross-seclio,rs of el,i~axial Iran- rents of tens of amperes.

slslors. - a single in~pur i ty diffusion a re used

(BASIC CIRCUITS ? to define the emitter region. This Bipolar transistors are structure offers the advantages of amplifiers. When a small signal low collector resistance and easy.$ i s applied to the input ternli- control of impurity spacings and J a bipolar transistor, a n am- emitter geometry. A variation of reproduction of this signal this structure uses two epitaxial a t the output layers. A thin lightly doped e p i - , r ~ l t h o u g h there a re six possible ways taxial layer used for the collector - of connecting the input signal, only is deposited over the original heavilyi three useful circuit configurations doped sen~iconductor wafer prior to exist for current or power amplifi- the epitaxial deposition of the base? cation: common-base, common-emit- region. The collector epitaxial layer \ter, and common-collector. In the1 is of opposite-type dopant to the common-base (or grounded-base) con- epitaxial base layer. This structure, nection shown in Fig. 21, the signal shown in Fig. 20(b), has the added is introduced into the emitter-base advantage of higher voltage ratings circuit and extracted from the collec- provided by the epitaxial collector tor-base circuit. (Thus the base layer. element of the transistor is common

The overlay transistor is a double- to both the input and output cir- diffused epitaxial device which em- cuits). Because the input or emitter- ploys a unique emitter structure. A base circuit has a low impedance : large number of separate emitters (resistance plus reactance) in the ' a r e tied together by diffused and order of 0.5 to 50 ohms, and the metalized regions t o increase the output o r collector-base circuit has ' emitter edge-to-area ratio and reduce a high impedance in the order of the charging-time constants of the 1000 ohms to one megohm, the transistor without compromise of voltage o r power gain in this type current- and power-handling capa- of configuration may be in the order bility. Fig. 20(c) shows a section of 1500. -


Fig. 21-Cottrrrron-bn.re circrtif corlfigrtra- liorf.

The direction of the arrows in Fig. 21 indicates eleclron current flow. As stated previously, most of the current from the emitter flows to the collector; the remainder flows through the base. In practical transistors, from 95 to 99.5 per cent of the emitter current reaches the collector. The current gain of this configuration, therefore, is always less than unity, usually in the order of 0.95 to 0.995.

'The waveforms in Fip. 21 represent the input voltage produced by the signal generator e. and the output voltage developed across the load resistor RT.. When thc input voltage is positive, a s shown n t AB, it opposes the forward hias produced by the base-emitter battery, and thus reduces current flow through the

v n-p-n transistor. The reclucetl elec- ' tron current flow through RI. then

the top point of the resistor : / f less negative (or more positive) with respect to the lower point, a s shown a t A'B' on the output waveform. Conversely, when the

' input signal is negative, a s a t CD{ the output signal is also negative, as a t C'D'. Thus, the phase of the signal remains u~ichanyred in this circuit, i.e., there is no voltage phase reversal between the input and the output of a common-base amplifier.

I n t h e c o m m o n - e n l i t t e r ( o r grounded-emitter) connection shown in Fig. 22 the signal is introduced into the base-emitter circuit and extracted from the collector-emitter circuit. This configuration has more moderate input and output impedances than the common-base circuit. The input (base-emitter) impedance

is in the range of 20 to 5000 ohms, and the output (collector-enlitter) impedance is nbout 60 to 50,000 ohms. I'ower gains ill thc ortlcr of 10,000 (or approximately 40 dB) can be realized with this circuit because i t provides both current gain and voltage gain.

Current gain in the common- emitter configuration is measured between the base and the collector, ra ther than between the emitter and the collector a s in the common-base circuit. Because a very small change in base current produces a relatively large changc in collector current, the current gain is always greater than unity in a common-emitter circuit; a typical value is about 50.

The input signal voltage undergoes a phase reversal of 180 degrees in a common-enlitter hmplifier, a s shown by the wavefor~lls in Fig. 22.

Fig. 22-Corrt~rtorf-crrrittcr circctir corl- figftmlio~r.

*a

\\"hen the input voltage is positive, a s shown a t AB, i t increases the forward bias across the base-emitter junction, and thus increases the total current flow through the transistor. The increased electron flow through RI. then causes the output voltage to become negative, a s shown a t A'B'. During the second half-cycle of the v ~ v e f o r m , the process is re- 6. versed, ~,e.', when the input signal is negative, the output signal is posi- F tive (as shown a t CD and C'D'.)

The third type of connection, sho\vn in Fig. 23, is the common-collector (o r grounded-collector) circuit. In this configuration, the signal is introduced into the base-collector circuit and extracted from the emitter- collector circuit. Because the input


impedance of the transistor is high and the output impedance low in this connection, the voltage gain is less than unity and the power gain is usually lower than that obtained in either a common-base or a com- nlon-emitter circuit. The common- collector circuit is used primarily a s

a n impetlance-matching device. As in the case of the common-base circuit, there is no phase reversal of the s ~ g - nal between the input and the output.

The circuits shown in Figs. 21 through 23 are biased for n-p-n transistors. When p-n-p transistors a r e used, the polarities of the batteries must be reversed. The voltage phase relationsliips, however, remain the same.

CHARACTERISTICS f

HE term "characteristic" is used T . . to ~dcnt l fy the tlistinguisl~ing electrical features and values of a transistor. These values may be shown in curve for111 or they may be tabu- lated. When the characteristics values a re given in curve form, the curves may be used for the determination of transistor performance and the calculation of additional transistor parameters.

Characteristics values a r e obtained from electrical measurements of transistors in various circuits under certain definite conditions of current and voltage. Static characteristics a re ob- taillet1 with dc potentials applied to the transistor electrodes. Dyntuuic characteristics a re obtained with an ac voltage on one electrode under various conditions of dc potentials

on all the electrodes. The dynamic characteristics, therefore, a re indica- tive of the performance capabilities of the transistor under actual working conditions. u

Published data for transistors include both electrode characteristic curves and transfer characteristic curves. These curves present the same information, but in two differen t forms t o provide more useful data. Because transistors a r e used most often in the common-emitter configuration, characteristic, curves a re usually shown f o r the collector or output electrode. The collector- characteristic curve is obtained by varying collector-to-emitter voltage and measuring collector current fo r dilrercnt values of base current. The transfer-characteristic curve is obtained by varying the base-to-emitter (bias) voltage or current a t a specified or constant collector voltage, ant1 measuring collector current. A t collector-characteristic family of curves is shown in Fig. 24. Fig. 25 shows transfer-characteristic curves fo r the same transistor.

UI W n W

5 500 - 4 4 1 4 0 0 I - u - 300 -

I- z k! 200

3 u u 100 0

6 W d 0 2 4 6 8 1 0 0 0 COLLECTOR-TO-EMITTER VOLTS (VCEl

9268-123277

Fig. ,74-Collec1or-chnr~c1crisric clrrves.

A measure of the current gain of a transistor is its forward current- transfer ratio, i.e., the ratio of the current in the output electrode to the current in the input electrode. Because of the different ways in which transistors may be connected in circuits, the forward current- transfer ratio is specified for a


C typical electrode currents in a 'x common-emitter circuit (a ) under no- $ signal conditions and (b) with n 'I one-microampere signal applied to d 1 400 the base. The signal current of one A microampere in the base causes a U t! 300 change of 49 microamperes (147-98) G in the collector current. Thus the ac g 200 beta fo r the transistor is 49. I) 0

- The cutoK frequency of a tmnsis- e too tor is defined a s the frequency a t

I- 0 W - Pa -1 J 0 0.2 0.4 0.6 0.8 1.0 1.2 o BASE-TO-EMITTER VOLTS (VBE)

92CS-I232(lT

Fig. 25-Trarrsfer-c\raracterislic cltrves. 'F;T+ NO SIGNAL 3 particular circuit configuration. The ELECTRON

IE=lOOpA common-base forward current-trans- f e r ratio is often called alpha (o r a), and the common-emitter for- + - I. - ward current-transfer ratio is often (a)

called beta (o r 8). In the common-base circuit shown

in Fig. 21 the emitter is the input electrode and the collector is the output eIectrode. The dc alpha, therefore, is the ratio of the steady-state collector current I r to the steady- 6

IB=15OpA s ta te emitter current IE:

I c 0.98 1 a=-=-= 0.98 .,- . -

In 1 (b) 4 ln the common-e ,n i t~er circui t Fig. 26-Elcc/rotlc crrrrc.trts rrrrtfer ( n ) 110-

shown in Fig. 22, the base is the sigrrcrl mrll ( 6 ) s ig~ml corrditior~s.

input electrode and the collector is the of alplla (for a the output electrode. The dc beta, conlmol,-~ase circuit) or beta (for a therefore, is the ratio of the steacly- ,olll,l,on~en,i~tcr circuit) drops to state current It' 0.707 times i ts 1-ItIIz vnluc. The? steady-state base current 111: gain-bnndwitlt 11 ~ ~ r o d o c t &he fre-

l r 0.98 I quency a t ~vhich the colnmot~- N = - ' = = 4 9 I,, 0 0 2 1

e m i t t e r f o r w ~ r d cul . rent- t ral isfel~ ratio (beta) is equal to unity. These

Because the ratios given :~bove a re charactc!ristics 1)rovitle an a])l>roxi- based on ~ t e a d y - ~ t a t ~ c~lrrcnts , they mate indication of the useful fie- a r e properly called dc alpha and ql lcl lc~ range of the device, and dc beta. I t is more conlnlon, how- help to determine the niost suitable ever, for tile curl.ent-trnnsfer ratio circuit c~llfigul~atioll for a particular to be given in tcrms of the ratio application. Fig. 27 shows typical of signal currents in the input and curves of all)lia and beta a s functions output electrodes, o r the ratio of of frequency. a change in the output current to y E x t r i r ~ s i c transcondnctnnce may the inpilt signal current which be defined as the quotient of a small causes the change. Fig. 26 shows change in collector current tlivided

.: Bipolar Transistors

cAlN-aAk9WtOTH specified electrodes at which t h e crystal structure changes and current begins to rise rapidly. The voltage then remains relatively constant over a wide range of electrode currents. Breakdown voltages may be measured with the third electrode open, shorted, o r biased in either the forward or the reverse direction. F o r example, Fig. 28 shows a series of collector-characteristic curves f o r different base-bias conditions. It can

FREQUENCY- Hz

Fig. 27-Forward crcrretrt-transfer ratio as a fu~rcriott of frequency.

1b))) o < 1 by the small change in emitter-to- base voltage producing it, under the condition that other voltages remain unchanged. Thus, if a n emitter-to- base voltage change of 0.1 9 causes a collector-current change of 3 milliamperes (0.003 ampere) with other voltages constant, the transconductance is 0.003 divided by 0.1, o r 0.03 mho. ( A "rnho" is the unit of conductance, and was named by spelling "ohm" bacltward.) F o r convenience, a millionth of a mho, or a micro- mho (pmho), is used t o express transconductance. Thus, in the example, 0.03 mho is 30,000 ~nicromhos.

,..a Cutoff currents a r e small steady- ? state reverse currents which flow

when a transistor is ljiased into '<, non-conduction. They consist of ( .) Icakage currents, which a r e related ' to the surface characteristics of the

sen~icontluctor material, and saturation currents, which a r e related to the impurity concentration in the material and which increase with increasing temperatures. Collector- cutoff current is the steady-state current which flows in the reverse- biased collector-to-base circuit when the emitter-to-base circuit is open. Emitter-cutoff current is the current which flows in the reverse- biased emitter-to-base circuit when the collector-to-base circuit is open.

Transistor breakdown voltages define the voltage values between two

~(BRI'CEO jv(0i c ~ s l YBRICER V(BR)CEV

COLLECTOR-TO-EMITTER VOLTAGE

Fig. 28-Typical collector-characteristic crtrves sltowitig locntiotr of various break-

dorvrr vol~anes.

be seen that the collector-to-emitter brealcclown voltage increases as the base-to-emitter bias decreases from the normal forward values through zero to reverse values. The symbols. shown on the abscissa a r e sometimes used to designate collector-to-emitter breakdown voltages with the base open V,nn,c~o, with external base-to- emitter resistance VqonIc~n, with the base shorted to the emitter V,nR,cas, and with a reverse base-to-emitter voltage Vcnn,cer.

As the resistance in the base-to- enlitter circuit decreases, the collector characteristic develops two breakdown points, a s shown in Fig. 28. After the initial breakdown, the collector-to-emitter voltage decreases with increasing collector current


until another hrenktlo\vn occurs a t a lower voItage. This 111iuin111nl collcctor-to-emitter l)realtdotv~~ voltage is called the x u s t a i n i n ~ voltage.

111 large-nrcn polvcr Lr:~nsistors, there is a l i~ni t ing ~ncchanism referred to a s "second breakdown". This cont l i t io~~ is not a volLage 1)renli- c low~~, but rather an rlcctrically and tl~crmally regenerative process in which curr(,nt is focusctl in a very small area of the order of the diam- etcr of a human hair. The very high current, together with the voltage across the transistor, causes a localized heating tha t may melt a n~ inu te hole from the collcctor to the emitter of the transistor and thus cause a short circuit. This regenerative process is not initiated unless certain high voltages and currents a re coincident for certain finite l e n ~ t h s of time.

In conventional transistor structures, the l in~i t ing effects of second brcaktlown vary directly with the amplitude of the applied voltage and inversely with the width of the base region. These effects a re most severe in power transistors in which narrow base structurcs a r e used to achieve fiood high-frequency response. In RCA "overlay" power transistors, a special emitter configuration is used to provide greater current-handling capability and minimize the possibility of "hot spots" occurring a t the emitter-base junction. This new design extends the range of power and frequency over which transistors can be operated before second breakdown begins to limit performance.

The curves a t thc left of Fig. 28 show typical collcctor characteristics under normal fornfnrd-l~ias conili- tions. For a given basc input rurrent, the collector-to-emitter saturation voltnge is the n~in in lun~ voltngc required to maintain the transistor in f111l conduction (i.e., in thc saturation region). Under saturation conditions, a further increase in forward bias produces no corresponding in- crcnse in collector current. Saturation voltages a re very ilnportnnt in s\vitch-

ing applications, and are usually spccificd for several conclitior~s of electrode currents and ambient temperatures.

llracll-throug11 (or ~ ) u ~ ~ c h - t I ~ r o u g l ~ ) voltage defines the v o l t a ~ e valuc a t which the depletion region in the collector r e g i o ~ ~ pass's coml)lctely t.hrough thc Ijase rerion ant1 maltcs contact a t sonlc point wilh thc emitter region. This "reach-through" phenomenon results in a rclativcly low-resistance path betwcerl the emitter ant1 the collector, and causcs a sharp increase in currcnt. Punch- througll voltagc docs not result in permanent daniage to a transistor, provided there is sulncient impetlance in the power-supply source to limit transistor dissipation to safe values.

BIASING 1

For most non-switching applications, the operating point for n ~ a r - ticular transistor by the quiescent (dc, 110-s i~ t~a l ) values of collector voltage ant1 emitter current. In general, a transistor- - ns a current-operatcmcl cle- vice, i.e., the current flowing i n the emitter-base circuit the current flowing in the collcctor circuit. The voltage a~l t l current values selected, a s well a s the particular binsing arrangement used,

both the transistor chnrartcr- l s t ~ c s and the specific require~ncnts of the application.

As mentionctl previously, biasing of a transistor for most applications o f forlvard bias across the emit te~,- l~asc junction ancl reverse bias across the collector-base junction. In Figs. 21, 22, and 23, two ljatterics to estnblish I~ias of the correct po1:lrit.y for an n-p-n transistor in the common-hasc, conlmon-emitter, and common-collector circuits, respectively. hIany varin- tions of these basic c i r c u i t ~ a l s o %-. (In these simplifietl tlc circuits, inductors and transformer- - only by their series re- sistancc.)

Bipolar Transistors 2 1

A sin~plifictl biasing arrangement for the common-l)ase circuit is shown in Fig. 29. Bias for both the collector- base jrlnction and the emitter-base

n-P-n

( b ) Fin. 29-Uinsirrg rretw~ork /or4611r1rrorr-bass

circrrit for (n) 11-p-rr nrrd (h ) p-11-p lrolrsistors.

junction is obtained from the single battery through the voltage-divider networlc consisting of resistors E and It,. ( F o r the n-p-n transistor shown in Fig. 29(a) the emitter-base junction is forward-biased because the emitter is negative with respect to the base, and the collector-base junction is reverse-biased because the collector is positive with respect to the base, a s shown. For the p-n-p transistor shown in Fig. 29 (b) , the polarity of the battery and of the electrolytic bypass capacitor Ct is reversed.) The electron current I from the battery and through the voltage tlivicler causes a voltage drop across resistor R1 which biases the base. The proper amount of current then flows through Rt so that the correct emitter potential is established to provide forward bias relative to the base. This emitter current establishes the amount of collector current which, in turn, causes a voltage drop across R,. Simply stated, the voltage divider consisting of R? and Ra establishes the base potential; the base potential essentially establishes the emitter potential; the enlitter poten-

tial and 13esistor HI establish the emitter current; the emitter current establishes the collector current; and the collector current and RI establish the collector potential. RE is bypassed with capacitor C, so tha t the base is effectively grounded for ac signals.

A single battery can also be used to bias the common-emitter circuit. The simplified arrangement shown in Fig. 30 is commonly called "fixed bias". In this case, both the base and the collector a re made positive with respect to the emitter by means of the battery. The base resistance Rs is then selected to provide the desired base current In fo r the transistor (which, in turn, establishes the desired emitter current I.), by means of the following expression:

vsu - Vos Ro =

ID where V n n is the battery supply voltage and Vns is the base-to-emitter voltage of the transistor.

I11 the circuit shown, f o r example, the battery voltage is six volts. The

Fig. 30-"Fixed-bias" urrungcrnerrt for co~nrnorr-emitter circrril.

value of Itll was selected to provide a base current of 27 microamperes, as follows:

6 - 0.6 Rn = 27 x 10-3 = 200,000 ohms

The fixed-bias arrangement shown in Fig. 30, however, is not a satisfactory method of biasing the base in a comn~on-emitter circuit. The critical base current in this type of circuit is very difficult to maintain untler fixed-bias conditions because of variations between transistors and the sensitivity of these devices

22 RCA Transistor, Thyristor, & Diode Manua l

to temperature changes. This problem is partially overcome in the "self- bias" arrangement shown in Fig. 31.

Fig. 31-"Sell-bias" nrrarrgerrrerrt /or cotrt- rrrorr-errfilter circriif,

In this circuit, the base resistor is tied directly to the collector. This connection helps to stabilize the operating point because a n increase or decrease in collector current produces a corresponding decrease or increase in base bias. The value of RI, is then determinetl as described above, except that the collector voltage VCR is used in place of the supply voltage Vnn:

The arrangement shown in Fig. 31 overcomes many of the disadvantages of fixed bias, although i t reduces the effective gain of the circuit.

In the bias method shown in Fig. 32 the voltage-divider network composed of R, and K provides the

Fig. 32-Bias rtet~cork risirrg voltage- divider arrarrgenrerrt for irrcreascd

smbilitp.

required forward bias across the base-emitter junction. The value of

the base bias voltage is determined by the current through the voltage divider. This type of circuit provides less gain than the circuit of Fig. 31, but is commonly used because of its inherent stability.

The common-emitter circuits.shown in Figs. 33 and 34 may be used to provide stability and yet minimize loss of gain. In Fig. 33, a resistor

Fig. 33-Bias rrefwork rrsirlg e~~r i t t e r sla- bitizirrg resistor.

R, is added to the emitter 'circuit, and the base resistor R2 is returned to the positive terminal of the battery instead of to the collector. The emitter resistor R,: provides additional stability. I t is bypassed with capacitor CE. The value of Ce depends on the lowest frequency t o be amplified.

In Fig. 34 the R,R3 voltage-divider network is split, and all ac feedbaclc currents through R3 are shunted to ground (bypassed) by capacitor C,.

Fix. 34-Bias ~retwork tr.rirrr: .~plir ~,olta,oe- divider network.

The value of R, is usually larger than t h e value of R,. The total resistance of R? and R2 should equal the resistance of RI in Fig. 32.

In practical circuit applications. any combination of the arrangements shown in Figs. 31, 32, 33, and 34 may be used. However, the stability of Figs. 31, 32, and 34 may be


poor unless the voltage drop across the load resistor Rr, is a t least one- third the value of the supply voltage. The determining factors in the selection of the biasing circuit a re usually gain and bias stability (which is discussed later).

In many cases, the bias network may include special elements to compensate for the effects of variations in ambient temperature or in supply voltage. F o r example, the thermistor (temperature-sensitive resistor) shown in Fig. 35(a) is used to compensate for the rapid increase of collector current with increasing

Fig. 35-Bins trrr~vorks irrclrrditrg ( a ) a ~lter~riistor arid ( 6 ) a voltage-cott~pcnsa!ing :

diode.

temperature. Because the thermistor resistance decreases a s the temperature increases, the emitter-to-base bias voltage is reduced and the collector current tends to remain constant. The addition of the shunt and series resistances provides most effective compensation over a desired temperature range.

The diode biasing network shown in Fig. 35(b) stabilizes collector current fo r variations in both temperature and supply voltage. The forward-biased diode current determines a bias voltage which establishes the transistor idling current (collector

current under no-signal conclitions). As the temperature increases, this bias voltage decreases. Because the transistor characteristic also shifts in the same direction and magnitude, however, the idling current remains essentially independent of temperature. Temperature stabilization with a properly designed diode network is substantially better than t h a t provided by most thermistor bias networks. Any temperature-stabilizing element should be thermally close to the transistor being stabilized.

In addition, the diode bias current varies in direct proportion with changes in supply voltage. The resultant change in bias voltage is small, however, so tha t the idling current also changes in direct proportion to the supply voltage. Sup- ply-voltage stabilization with a diode 'biasing network reduces current variation to about one-fifth tha t obtained when resistor o r thermistor bias is used f o r a germanium transistor and one-fifteenth f o r a silicon transistor.

The bias networks of Figs. 30 through 34 a r e generally used in class A circuits. Class B circuits normally employ the bias networks shown in Fig. 35. The bias resistor values fo r class B circuits a re generally much lower than those f o r class A circuits.

4*, . . : dy BIAS STABILITY

Because transistor currents tend to increase with temperature, i t is necessary in the design of transistor circuits to include a "stability factor" to keep the collector-current variation within tolerable values under the expected high-temperature operating conditions. The bias stability factor SF is expressed a s the ratio between a change in steady- s tate collector current and the corresponding change in steady-state collector-cutoff current.

For a given set of operating voltages, the stability factor can be calculated for a maximum permissible rise in steady-state collector current


f r o ~ n the room-ten~l)erat~t re valucx, a s follows:

where Ic , and Icntll arc measured a t 85"C, Irno? is measured a t the maximum expected ambient (o r junction) temperature, and Ic,,,,. is the maximum pern1issil)le collector current fo r the specified collector-to-emitter voltage a t the n~axilnunl expected ambient (or junction) temperature (to keep transistor dissipation within ratings).

The calculated values of S F can then be used, together with the appropriate values of beta and rb' (base- connection resistance), to determine suitable resistance values fo r the transistor circuit. Fig. 36 shows equations fo r S F in terms of resistance values f o r three typical circuit confi~urations. The maximum value which S F can assume is the value of beta. Although this analysis was originally made f o r germanium transistors, in which the collector saturation current Ic0 is relatively large, the same type of analysis may be applied to interchangeability with beta for silicon transistors.

COUPLING

Thrce basic methods are uscd to couple t r a n s i s t o r s t a g e s : t r a n s - former, resistance-capacitance, and direct coupling. P- The major advantage of trans. former coupling is tha t it permits power to be transferred from one-' impedance level to another. f A transformcr-coupled comn~on-emitter n-p-n stage is shown in Fig. 37. The voltage step-down transformer T, couples the signal from the collector of the preceding stage to the base of the common-emitter stage. The voltage loss inherent in this transformer is not significant in transistor circuits because, a s mentioned previously, the transistor is a current- operated device. Although the voltage is stepped down, the available current is stepped up.@I?he chahge in base current resulting from the presence of the signdl causes an alternating collector current to flow in the primary winding of transformer T?, and a power gain is obtained between TI and Tn., 7

This use of a voltage step-down transformer is similar t o that in the output stage of an audio amplifier, where a step-down transformer is

Fig. 36-Bins-sfabi1il)-factor rqrtnfions lor tlrree typical circrrit configrtratio~rs.


nornlally used to drive thc loudspeaker, which is also a current- operated device.

The voltage-divider network consisting of resistors R, and Ib in Fig. 37 provides bias fo r the transistor.

Fig. 37-Tra11sJorn1er-cori~~led conlmon- etnilter slage.

The voltage divider is bGPassed by capacitor C, to avoid signal attenua- t ion.qhe stabilizing emitter resistor Ra permits normal variations of the transistor and circuit elements to be compensated for automatically without adverse effects:"~his resistor RI: is bypassed by capacitor C,. The voltage supply Vnn is also bypassed, by capacitor C3, to prevent feedback in the event that ac signal voltages are developed across the power supply. Capacitors C, and Ca may nor- m a l l y be r e p l a c e d b y a s i n g l e capacitor connected between the emitter and the bottom of the secondary winding of transformer TI with little change in performance.

The use of resistance-capncitance coupling usually permits some economy of circuit costs and reduction of size, with some accompanying sacrifice of gain. This method of coupling is particularly desirable in low-level, low-noise audio amplifier stages to minimize hum pickup from stray magnetic fields. Use of resistance-capacitance (RC) coupling in battery-operated equipment is usually limited to low-power operation. The frequency response of an RC- coupled stage is normally better than that of a transformer-coupled stage.

Fig. 38 shows a two-stage RC- coupled circuit using n-p-n transistors in the common-emitter configuration. The method of bias is similar to t h a t used in the transformer- coupled circuit of Fig. 37. The major additional components a r e the collector load resistances R L ~ and RL) and the coupling capacitor C,. The value of C, must be made fairly large, in the order of 2 to 10 microfarads, because of the small input and load resistances involved. ( I t should be noted t h a t electrolytic capacitors a r e normally used for coupling in transistor audio circuits. Polarity must be observed, therefore, to obtain proper circuit operation. Occasionally, excessive leakage current through an electrolytic coupling capacitor may adversely affect transistor operating currents.)

Impedance coupling is a modified form of resistance-capacitance coupling in which inductances a re used

Fig. 38-T~t.o-s1q~e resislnt~cc-cnpnciln~rce cartpled chcrril. C


to replace the load resistors. This type of coupling is rarely used except in special applications where supply voltages are low and cost is not a significant factor.

1)irect cor~pling is usetl primarily when cost is an important factor. ( I t should be noted that direct- coupled amplifiers a re not inherently dc amplifiers, i.e., that they cannot always amplify dc signals. Low- frequency response is usually limited by other factors than the coupling network.) In the direct-coupled amplifier shown in Fig. 30, resistor R, serves a s both the collector load resistor for the first stage and the bias resistor fo r the seco~ld stage. Resistors R1 and R? provide circuit stability similar to that of Fig. 32 1)ecause the emitter voltage of transistor Qz and the collector voltage of transistor Q , are within a few tenths of a volt of each other.

a pronounced effect on the gain and power-output capabilities of transistors. A s a result, physical as- pects such a s layout, type of chassis, shielding, and heat-sink considerations a r e important in the design of high-frequency amplifiers and oscillators.

General Considerations In general, high-frequency circuits

a r e constructed on material such a s brass o r alun~inunl which is either silver-plated or machined to increase conductivity. The input and output circuits are "compartmentalized" by use of a milling operation. Copper- clad laminated or printed circuit boards facilitate soldering operations, and have been used satisfactorily a t frequencies up to 400 MHz when the entire copper surface was kept intact and dsed for the ground plane.

Because so few circuit par ts a re ~.equired in the direct-coupled amplifier, maximum economy can be achieved. However, the number of stages which can be directly coupled is limited. Temperature variation of the bias current in one stage may be amplified by all the stages, and severe temperature instability may result.

Because even a short lead provides a large impedance a t high frequencies, i t is necessary to keep all high-frequency leads a s short a s possible. This precaution i s especially important f o r ground connections and for all connections to bypass capacitors and hi~h-frequency filter capacitors. I t is recommended that a conlmon ground return be used for each s tare . and that short. direct - ,

connections be made to the common HIGH-FREQUENCY OPERATION ground point. The emitter lead es-

At freauencies of 100 RIHz or veciallv should be kevt a s short a s more, the' effects of s t ray capaci- possibie. tances and inductances, ground In many cases, problems of oscil- paths, and feedback coupling have lation and regenerative feedback a re


caused by unwanted ground currents (i.e., ground-circuit feedback currents). An effective solution is to isolate the ac signal path from the dc path so that the signal does not pass through the power supply by way of the power leads. I n a multi- s t a ~ e amplifier, the power leads should enter the circuit a t the highest power stage to minimize the amount of signal on the common power path. Lower-frequency oscillations can be minimized by use of a large capacitor across the power- supply terminals. High-quality feedthrough capacitors should also be used a s the power-lead connections.

Particular care should be taken with the lead dress of the input output circuits of h i g h - f r e q u e g stages so that the possibility of s t ray coupling is minimized. Unshielded leads connected to shielded components should be dressed close to the chassis. ( In high-gain audio amplifiers, these same precautions should be taken to minimize the possibility of self-oscillation.)

Feedback effects may occur in radio or television receivers a s a result of c o u p l i ~ ~ g between stages through coninlon voltage-supply circuits. Fil- ters find an important use in minimizing such effects. They should be placed in voltage-supply leads to each transistor to provide isolation between stages.

Capacitors used in transistor r f circuits, particularly a t high frequencies, should be mica or ceramic. F o r audio bypassing, electrolytic capacitors a re required.

In high-frequency stages having high gain, undesired feedback may occur and produce harmful effects on circuit performance unless shielding is used. The output circuit of each stage is usually shielded from the input of the stage, and each high- frequency stage is usually shielded from other high-frequency stages. I t is also desirable to shield separately each unit of the high-frequency stages. For example, each if and r f coil in a superheterodyne receiver

may be mounted in a separate shield can. Baffle plates may be mounted on the ganged tuning capacitor to shield each section of the capacitor from the other section.

The shielding precautions required in a circuit depend on the design of the circuit and the layout of the parts. When the metal case of a transistor i s grounded a t the socket terminal, the grounding connection should be a s short a s possible t o minimize lead inductance. Many transistors have a separate lead connected to the case and used a s a ground lead; where present, these leads a re indicated in the outline diagrams.

Transistor Requirements The important performance cri-

teria in rf power-amplifier circuits a re power output, power gain, and efficiency. Transistors to be used for power amplification mLustaGliviif power efficiently with sufficient gain in the frequency range of interest.

Power Output-The power-output capability of a transistor i s determined by the current- and voltage-handling capabilities of the device in the frequency range of interest. The current-handling capability of the transistor is limited by its emitter periphery and the resistivity of the epitaxial layer. The voltage-handling capability of the device is limited by the breakdown voltages which are, in turn, limited by the resistivity of the epitaxial layer and by the penetration of the junction.

Fig. 40 shows a typical family of dc collector characteristics with base current a s a parameter. The highest breakdown voltage is tha t of the collector-to-base junction Vcnn)cn'o; the lowest voltage is tha t of the collector-to-emitter junction with the base open V,nn)r~:o. Breakdown voltages may vary anywhere between these two values depending on how the base is biased with respect to the emitter or on the resistance between the emitter and the base. The


I In general, all rf power transistors have operating voltage restrictions, and only current-handling

I- capability differentiates power transistors from small-signal units. A t high current levels, the emitter current of a transistor is concentrated a t the emitter-base edge; therefore, transistor current-handling capabil-

0 i ty can be increased by the use

u of emitter geometries which have high emittcr-periphery-to-enlitter- area ratios and by the use of im-

VA proved methods of glowing collector COLLECTOR'TO-EMITTER VOLTAGE substrate material. Transistors in-

Fig. 40-Collector clrrretrl n~ n fltrlcriorr tended for large-signal applications 01 collector-lo-crtlillcr 1.ollnge Jor a SllOuld be designed so the

lypicol rf Ir-nrr.ristor. currents do not cause base widen-

static V v ~ n and V(:IIO values are related by the follo\\ring equation:

Vann VCEO =

(1 + h~l:)""

~vhere hre is the static forward- current transfer ratio and n is a n empirical number t h a t varies from 2.5 to 4 fo r n-p-n silicon transistors. When rf input is applied, the breakdown voltage is substantially higher than the dc or static value observed in the VCIW mode. Substitution of f ~ / f fo r hrr: in the equation f o r Vceo yields the following result:

where f r is the dynamic gain-bandwidth product and f is the frcqt~ency of operation. This equation indicates an increase in the breakdown characteristic f rom thc V~130 value under dc conditions to a value t h a t approaches Vcno a t operating fre-

ing, a condition tha t would limit thc current-handlina capability of the device. Base widening is severe in transistors in which the collector side of the collector-base junction has a lower carrier concentration and higher resistivity. than the base side of the junction. I-Io~vever, the need for low-resistivity material in the collector to handle high currents without base widening severely limits the breakdown voltages. As a result, epitaxial layers of differ- e n t resistivity a r e often used for different operational voltages.

Large-Signal Power Gain-The power gain of a transistor power amplifier i s cletermined by the dynamic f ~ , the dynamic input impedance. and the collector load imned- ance; the collector load inlpcdance depends on the required power output and the collector voltage swing. The power gain, P.G., of a transis- to r power amplifier may bc expressed in many forms. The simplest one i s a s follows:

quencies equal to o r -greater than f ~ . P.G. = (f.r/f)' Rr.

Another parameter which limits 4 R. (Z,.) the potver-handling capability of the transistor is the saturation volt- where RI. is the real par t of the col- RRC. The rf value of the saturation lector parallel-equivalerrt-load im- voltage Vcc,srT, is significantly pedance determined by the required greater than the dc va l~ ie because power output, and ZI. is the dynamic the active area is less a t high fre- input impedance when the collector quencies than a t dc. load impedance is Zr..

Bipolar Transistors

The equation for power gain shows tha t fo r high-gain operation large-signal or power transistors should have a high current gain which remains constant a s the large- signal current level is varied. In other words, transistors suitable fo r large-signal operation must provide current gain under large-current- swing conditions. Constant current gain for varied current level can be achieved with shallow diffusion techniques.

The dynamic input impedance of the transistor pellet varies considerably under large-signal operation a s compared to small-signal operation. The resistive par t of the input impedance is inversely proportional to the a rea of the transistor and, therefore, to the power output of the device. The package parasitic inductance has a significant effect on the input impedance. A simple representation of a common-emitter equivalent transistor input circuit a t uhf and microwave frequences is shown in Fig. 41. The large-signal R,. and LI, a re different from the small- signal values; therefore, their exact quantitative analysis is difficult. The

Zin -4

Fig . 41-Eqrrivalerr~ irrpttl circrtir of arr rf power Irurrsi,~llor.

gain, a s indicated by the following relation:

( f ~ / f )' RL P.G. =

4 (rb + W T L ~ )

The effect of the emitter parasitic inductance is to reduce the power gain.

f- Efficiency-Transistor efficiency is determined with the device operating unde signal-bias conditions. The collectu&o-base junction is reverse- biased, and the emitter-to-base junction is forward-biased partially with the input drive signal. The collector efficiency of a transistor rf amplifier is defined a s the ratio of the rf power output a t the frequency of interest to the dc input power. Therefore, high efficiency implies t h a t circuit loss is minimum and tha t the ratio of the transistor output, the parallel equivalent resistance, and its collector load resistance a r e maximum. Thus, the transistor parameter which limits the collector efficiency is output admittance. The output admittance of a transistor pellet consists of two parts: a n output capacitance C, and a n equivalent parallel output resistance which approaches l/oT C, a t microwave frequencies under small-signal conditions. In a common-emitter circuit, C u b is essentially the output capacitance because the impedance level a t the base is low relative to the impedance level a t the transistor output. The output capacitance represents effectively the transistor junction capacitance in series with a resistance. If the collector re- sistivitv is increased. the effective

input impedance as follows:

outputcapacitance and the collector- Z ~ r l can be expressed base breakdown voltage a r e both

increased. In a vower transistor. I , variations in junction and epitaxial

WT thickness cause variations in C o b Z ' n = ('b + " T ~ ~ ) ' j ( wLc - ; ) with Ven. a s shown in Fie. 42. Thus. '

the dynamic output capacitance is where WT = 2 ~ f T , w = 2;f, and L, is a function of voltage swing and the emitter parasitic inductance. power level. I t can be shown tha t

The parasitic emitter inductance the average C. under maximum volt- also has a significant effect on power a g e swing is equal t o 2 C o b , where


Cab is measured a t the voltage value of Vrn. F o r a first approxinlation, the large-signal output resistance can be assumed to be inversely proportional to C,,.. Because the ratio of the transistor output resistance to i ts collector load resistance determines the collector eflicicncy, a transistor with high output resistance and, therefore, low C,.I, is essential.

w L Fig. 42-Collccror-to-base capucitotrce as Q f~tirc~iorz of collector-to-base volta,yc /or

n typical rf ponper trarrsttor.

Another transistor parameter that affects the efficiency of the device is the dissipation capability. The maxi~nunl power t h a t can be dissipated before thermal runway occurs depends on how well internal transistor heat is ~.emoved. The amount of heat removed by conduction i s a n inverse function of the thermal resistance. The total thermal resistance is equal to the sum of several thermal drops in series: from the collector junction to the back of the pellet, a t the pellet-solder interface, a t the solder connection to the case, f rom the case to the heat sink, and from the heat sink to the atmosphere o r ain- hient. These drops a rc usually divided into two major groups, junction-to-case t h e r n ~ a l resistance 01-c and case-to-ambient thermal resistance en-.,. Generally, power transistors a r e designed for mini-

mum junction-to-case thermal resistance. The thernlal resistance 19, expressed in degrees C per wat t of dissipation, may he calculated for the various sections of total heat- flow path a s follows:

where L is the distance tha t the heat travels in inches, A is the area of the path in square inches, and K is the material constant in W / " C - inches. K is equal to 2.12 for silicon, 6.2 fo r beryllium oxide, 9.7 f o r copper, and 3.1 for aluminum. For a given length and width, the thermal resistance can thus be calculated for most geometries. I t has been conlmon practice to characterize the transistor heat dissipation by t h ~ average device thernlal resistance. The average junction-to-pmlbient thermal resistance 01-., of a device can be expressed as . follows:

One of the problems in power dissipation is t h a t of complete mounting of the pellet so that there is no discontinuity in the bond between pellet and mounting. Consid- erable care must be usecl in selection of the mounting system. A t present, microwave power transistors a rc mounted with gold-silicon niountinfi. systems. I t should be pointed out that the dissipation of a microwave power transistor is .considerably higher under rf operation than under dc operation. The junction temperature a t radio frequencies is more a function of the average device dissipation than of the peak dissipation. The dissipation of a n~icrolvave power transistor is also a function of the thermal time constant.

SWITCHING

Transistor switching applications a r e usually characterized by large- signal nonlinear operation of the devices. The switching transistor is generally required to operate in

Bipolar Transistors

either of two states: on or off. In transistor switching circuits, the common-emitter configuration is by f a r the most widely used.

Typical output characteristics fo r an n-p-n transistor in the common- emitter configuration a r e shown in Fig. 43. These characteristics a re divided into three regions of operation, i.e., cutoff region, active region, and saturation region.

Fig. 43-Typical collector chnrac!eristic o f an 11-p-11 tra~rsistor .rho$ving lllree princi-

pal regions involved in switching.

In the cutoff region, both the emitter-base and collector-base junctions a re reverse-biased. Under these conditions, the collector current is very small, and is comparable in magnitude t o the leakage current ICE", ICRP, or IcII~,, depending on the type of base-emitter biasing used.

Fig. 44 is a sketch of the minority- carrier concentration in an n-p-n transistor. For the cutoff condition, the concentration is zero a t both junctions because both junctions a re reverse-biased, a s shown by curve 1 in Fig. 44.

In the active region, the emitter- base junction is forward-biased and the collector-base junction is reverse- biased. Switching from the cutoff region to the active region is accomplished along a load line, a s indicated in Fig. 43. The speed of transition through the active region is a function of the frequency-response characteristics of the device.

EMITTER- COLLECTOR- BASE JUNCTION BASE JUNCTION

V) Z

I 0 F a F t w a 0 I- zn z s;k! Z

S

Fig. 44- firtority-carrier concentrations in an P I - 2 n transistor: (1) in c~ l to f l region, ( 2 ) 'in active region at edge o f satu-

ratiotr region, (3) in saturation region.

The minority-carrier concentration for the active region is shown by curve 2 in Fig. 44.

The remaining region of operation is the saturation region. In this region, the emitter-base and collector-base junctions a r e both forward- biased. Because the forward voltage drop across the emitter-base junction under this condition [Vs.(sat)] is greater than tha t across the collector-base junction, there is a net collector-to-emitter voltage referred to a s Vce(sat). I t is evident tha t any series-resistance effects of the emit- t e r and collector also enter into determining V ~ ~ ( s a t ) . Because the collector is now forward-biased, additional carriers a r e injected into the base, and some into the collector. This minority-carrier concentration is shown by curve 3 in Fig. 44.

A basic saturated-transistor switching circuit is shown in Fig. 45. The voltage and current waveforms for this circuit under typical

Fiz. 45-Basic sattirated transistor switching circ~rit.


base-drive conditions a r e shown in Fig. 46. Prior t o the application of the positive-going input pulse, the elllitter-base junction is reverse- biased by a voltage -V131:(off) = VIIB. Because t h e t rans is tor is in the cutoff region, t he 1)asc cu r ren t 111 is the reverse lealiage cu r ren t Iner, which i s negligible compared wi th In,, and the collector cu r ren t Ic i s the reverse lealiage cu r ren t Irrsv, which i s negligible compared wi th Vrc/Rc. When the positive-going input pulse V, i s applied, the base current In imnlediately goes positive.

INPUT 7 1 ' " g PULSE

COLLECTOR

EMITTER VOCTAGE

Fi.?. 46- Voltage nrtd crrvrcrtt w'avcforr)~~ for sat!rraletI s~vi1clli11,q circltit showtr ilr

Fig. 45.

The collector current , however, does not begin to incrcase until some time later. This delay in the flow of collector cu r ren t (t,!) resul ts because the emi t t e r and collector capacitances (lo not allow the elnit- ter-base junction to I)ecome forward- 1)iased instantaneously. These capacitances m u s t be charged f rom their original negative potential [-Vnp:(oK)] t o a forward bias sufficient to cause the t rans is tor to conduct appreciahly. Af t e r the emitter-base junction is suff~ciently forward-biased, therc is a n additional delay caused by the t ime required f o r minority carr iers which a r e injected into the basc to diffuse across the basc and he coll'ccted a t the collector. This delay is usually negligible compared with the delay

introduced by the capacitive component. The collector and emit ter capacitances vary wi th the collcctor- base and emitter-base junction voltages , and increase a s the voltage Vnn goes positive. An accurate determination of to ta l delay time, therefore, requires Itnowledge of the nonlinear characterist ics of these capacitances.

When t h e collector cu r ren t IC be- g ins to increase, the t rans is tor h a s made the transit ion f r o m the cutoff region into the active region. The collector cu r ren t talces a finite t ime to reach i t s final value. Th i s t ime, called r ise t ime ( t , ) , is determined by the gain-bandwidth product ( f r ) , t h e collector-to-emitter capacitance (CC), arid the s ta t ic forward.current- t r ans fe r ra t io (hYE) of the transistor . A t high collector cu r ren t s a n d / o r low collector voltages, t he effect of th is capacitance on rise t ime is negligible, and the rise t ime of collector cu r ren t i s inversely proportional t o f ~ . A t low currents a n d / o r high voltages, t he effect of gain- bandwidth product is negligible, and t h e rise t ime of collector current i s directly proportional t o the product RcCc. A t intermediate currents and voltages, the rise t ime is propor- t ional t o the sun1 ( $5-fT) $- R&c. Under a n y of the above conditions, t h e collector cu r ren t responds exponentially to a s t ep of base current. I f a turn-on base current (11,~) is applied t o the device, and the product I ~ l l h ~ l ~ i s less than VcrIRc, t he collector cu r ren t rises exponentially until i t reaches the steady-state value Iltshr;~?. If Ill,liv,: is g rea t e r than VCVIR,., the collector cu r ren t rises toward t h e value 1lllhP,:. The t r a n - s is tor becomes sa tu ra t ed when IC reaches the value I,., (=: VrcIRc). At this point, 1,: is effectively clamped a t the value Vrr /Rr .

The r ise time, therefore, depends on a n exponential function of the ra t io IC,;/II~, : hpr:. Because the values of ~ F I : , fT, and CC a r e no t constant, bu t va ry with collector voltage and cu r ren t a s the t rans is tor is switching, t he rise t ime a s well a s the


delay t i n ~ c is depcnclerit on nonlinear transistor characteristics.

After the collector current of the transistor has reached a steady-state value Its, the minority-charge distribution is that shown by curve 3 in Fig. 44. When the transistor i s turned off by returning the input pulse to zero, the collector current does not change immediately. This delay is caused by the excess charge in the base and collector regions, which tends to maintain the collector current a t the Ica value until this charge decays to an amount equal to tha t in the active region a t the edge of saturation (curve 2 in F ig 44). The time required f o r this charge to decay is called the storage time (t.). The rate of charge decay is determined by the minority- carrier lifetime in the base and collector regions, on the amount of reverse "turn-off" base current (In.), and on the overdrive "turn-on" current (In,) which determined how deeply the transistor was driven into saturation. ( I n non-saturated switching, there is no excess charge in the base region, so tha t storage time is negligible.)

When the stored charge (Qs) has decayed to the point where i t is equal to tha t a t the edge of saturation, the transistor again enters the active region and the collector current begins t o decrease. This fall- time portion of the collector-current characteristic is similar to the rise- time portion because the transistor is again in the active region. The fall time, however, depends on In?, whereas the rise time was dependent on IN,. Fall time, like rise time, also depends on f~ and Cc.

The approximate values of Im, I[,?, and Its f o r the circuit shown in Fig. 45 a re given by:

VG - Vnn - Vn!:(sat) IRI =

Rn

Vun + Vue(sat) I,,? =

R e

Vcc - VI:E(sat) Ics =

Hc

b w i t c h i n g Characteristics

The electrical characteristics for a switching transistor, in general, differ f rom that fo r a linear-amplifier type of transistor in several respects. The static forward current- transfer ratio ~ F E and the saturation voltages V c ~ ( s a t ) and Vne(sat) a re of fundamental importance in a switching transistor. The static forward current-transfer ratio determines the maximum amount of current an~plification t h a t can be achieved in any given circuit, saturated or non-saturated. The saturation voltages a re necessary for the proper dc design of saturated circuits. Consequently, ~ F E is always specified f o r a switching transistor, generally a t t ~ v o or more values of collector current. Vc~(sa t ) and Vne(sat) a r e specified a t one or more current levels for saturated transistor applications. Control of these three characteristics determines the performance of a given transistor type over a broad range of operating conditions. F o r non- saturated applications, Vce(sat) and Vm(sat) need not be specified. For such applications, i t is important to specify Vne a t specific values of collector current and collector-to-emit- t e r voltage in the active region. - -- Because the collector and emitter capacitances and the gain-bandwidth product influence switching time, these characteristics a re specified for most switching transistors. The collector-base and emitter-base junction capacitances a r e usually measured a t some value of reverse bias and a re designated Cob and Cib, respectively. The gain-bandwidth product (fT) of the transistor is the frequency a t which the small-signal forward current-transfer ratio (hre) is unity.& Because this characteristic falls off a t 6 dB per octave above the corner frequency, f~ is usually controlled by specifying the hr. a t a fixed frequency anywhere from 112 t o 1/10 f ~ . Because Cob, Clb, and fT vary nonlinearly over the

34 RCA Transistor, Thyristor, & Diode Manual j i

operating range, these characteris- transistor is exceeded under "off" ' tics a re generally more useful a s conditions, the following require- f igi~rcs of merit than a s controls fo r nlents must be met: determining switch in^ speeds. When The miniinuin emitter-to-base thc switching specds in a particular breakdown voltage V,I,I:,EIws must be application a re of major importance, greater than Vnn(0ff). i t is preferable to specify the re- Tile n~ini l l lu~a collector-to-base quircd switching s~)ecds in the de- breakdown voltage V,lrl{,c.ll<v must be sired switching circuit rather than greater than Vc.(: + Vsl.;(on). C,.I,, CII., and f.r. The n~inimum collector-to-emitter :

The storage time ( t - ) of a tran- breakdown voltage V,IIII,I.I:III. must be sistor is dependent on the stored greater than vC,:. charge ((2s) and on the driving cur- Vtllll,l:l,l, and Vll,,l,,.l,,, a re al\vays : rent enlploj~ed to switch the tran- specified for switching transistor. . sistor between cutoff and saturation. The collector-to-cnlitter brealrdown , Consccluentl~, either the stored voltage V,nloon($ is usually specified r charge or the storage time under under open-base conditions. l-he heavy overdrive conditions should breal<do\\rn voltage BV~.~ . .~~ , , ( the sub- j be specified. Most recent transistor script "RLU indicates a resistive :

specilications require that storage load in the collector circuit) is gen- . time be specified. erally higher than V(nlt,e*l:,,. The re-

of the dependence of the quirement that V ~ I I I I ) ~ . R ~ be greater ! switching times on current and volt- than v~ .~ . is overly pessimistic. ~h~

levels, these times a re deter- recluirement that~Vtl,ll,(.l.;lll, be greater nlinetl by the v o l t a ~ e s and currents than v ~ . ~ be used ,vherever t employed in circuit operation. applicable.

Coupled with the breakdown volt- ; Current, and ages nre the collector-to-en~ittcr and 1

Voltage Ratings base-to-enlitter transistor leakage : Up to this point, no mention has

l~ecn made of cl iss i~tat io~~, current, and voltage ratings for a switching transistor. The inaximum continuous ratings fo r dissipation and current a re determined in the same manner a s f o r any othcr transistor. In a switching applic-ation, however, the peak dissipation and current may be permitted to exceed these continuous ratings dcper~dinz on the ~tulse duration, on the duty factor, and on the thermal time constant of the transistor.

Voltc~ge ratings for switching transistors are mol e complirated. In the basic s~vitchinlr circuit shown in Fig. 45, three l)real<down voltages 117ust be consitlered. Whcn the transistor is turned off, the emitter-base junction is reverse-biased by the voltage Vllr. ( o f f ) , ( i.e., Vltn), the collector-base junction 1)s Vrc + Vl,", and the emitter-to-collector junction by + Vrr. To assure tha t none of the voltage ratings fo r the

currents. These leakage currents (1cv:r and IIWX) are particularly ini- !

portant considerations a t high oper- ' at ing temperatures. The subscript , "V" in these symbols indicates that these lealcage currents a r c specified !

a t a given emitter-to-base voltage i (either forward or reverse). In the ' basic circuit of Fig. 41, these currents a r e determined by the follow- '

ing conditions: I

I,.,:,. 1 v,., = v,, 11t1:rj V,,O = VI,I:(OIT) = -VItl, I

In a s~vitching tri~nsistor, these leak- ! age currents are usually controlled not only a t r o o n ~ temperature, but , also a t some higher operating tern- ,

perature near the upper operational limit of the transistor. 1

Inductive Switching 1 t

Most inductive switching circuits [ can be represented by the basic j equivalent circuit shown in Fig. 47. ;


This type of circuit requires a rapid and if t h e series resistance of the in- t ransfer of energy f r o m the switched ductor can be ignored, then the en- inductance to the switching mechan- e rgy to be dissipated is '/z LIZ. Th i s

type of r a t ing f o r a t rans is tor i s "CC called "reverse-bias second break-

down." The energy capabili ty of a t rans is tor varies with the load inductance and base-emitter reverse bias. A typical s e t of r a t ings which now appea r s in RCA published da ta is shown on Fig. 48.

TRANSISTOR UNDER TEST

* /'in. 47-Bnsic aqrri~~alenr rircrrir Jor it,-

~ lrrcr i i 'e stvi tchi~rg circtrit.

ism, which m a y be a relay, a t ransistor, a conlmutating diode, o r some other device. Often a n accurate calculation of the energy to be dissipated in the switching device is required, particularly if t h a t device is a transistor. If t he supply voltage is lo\\r compared t o the sustaining breakdown voltage of the t rans is tor

- 3 2

V) w I LL W

a 0 10 20 30 4 0 5 E X T E R N A L BASE-TO-EMITTER L. RESISTANCE-OHMS z 3 a LL

2

2 8 I -1 -8 -6 -4 -2 0

BASE-TO-EMITTER VOLTAGE-V u Y 4

: 3 a 2

I

0 100 200 300 400 INDUCTANCE- p H

(c

SAFE-OPERATING-AREA RATINGS

During normal circuit operation, power t rans is tors a r e often required to sus ta in high current and high voltage simultaneously. The capability of a t rans is tor to wi ths tand such conditions i s normally shown by use of a safe-operating-area r a t ing curve. This type of r a t ing curve defines, f o r both steady-state and pulsed operation, t he voltage- current boundaries t h a t resul t f rom the combined l in~i ta t ions imposed by voltage and cu r ren t ratings, the n ~ a x i m u m allowable dissipation, and the second-breakdown (Is/),) capabilities of the transistor.

If t h e safe opera t ing a rea of a power t rans is tor i s linlited within a n y portion of the voltage-current characterist ics by thermal fac tors ( thermal impedance, maximum junction temperatures , o r opera t ing case temperature) , th is l imiting is defined by a constant-power ( I = KV-') which can be represented on the log-log voltage-current curve by a s t r a igh t line t h a t h a s a slope of -1.

The energy level a t which secon breakdown occurs in a pourer t ransistor increases a s the t ime duration of the applied voltage and current decreases. The power-handling capability of the t rans is tor also increases with a decrease in pulse duration because the thermal mass of the power-transistor chip and associated n ~ o u n t i n g hardware impar t s an inherent thermal delay to a rise in junction temperature.

! RCA Transistor, Thyristor, & Diode Manual i

!

Fig. 49 sI1on.s a forward-bias safe-area r a t i n g c h a r t f o r a typical silicon power t rans is tor , t h e RCA- 2N3585. The boundaries defined by the curves in t h e safe-area c h a r t indicate, f o r both continuous-wave and nonrepetitive-pulse operation, the maximum cu r ren t ra t ings , t he max imum collector- to-emitter forward-bias avalanche breakdown- voltage r a t i n g [VnBI = 1, which i s usually approximated I)y V ~ ~ , ~ ( s u s ) ] , and the thermal and second-breakdown ra t ings of t h e t rans is tors .

A s shown in Fig . 49, t h e t he rma l (dissipation) l imiting of t h e 2N3585 ceases when the collector-to-emitter voltage r i ses above 100 volts du r ing dc operation. Beyond th is point, t h e safe opera t ing a r e a of t h e t rans istor i s limited by the second-breakdown ra t ings . Dur ing pulsed opera-

If a t r ans i s to r is- to- I,c operated ! a t a pulse dura t ion t h a t differs f rom those shown on the safe-area cha r t , t h e boundaries provided I)y thc safe-

i a r e a curve f o r t h e next h igher pulse dura t ion mus t be used, o r t he t r a n - s i s tor manufacturer should be con- sulted. Moreover, a s indicated in ' Fig. 49, safe-area r a t i ngs a r e norm- ' al ly given f o r single nonrepeti t ive pulse operation a t a case tempera- t u r e of 25°C and mus t be tlerated f o r operation at higher r a s e tcm- pe ra tu re s and under repeti t ive-pulse o r continuous-wave conditions.

F ig . 50 shows t empera tu re de ra t - i ng curves f o r t h e 2N3585 safe-area c h a r t of Fig. 49. These curves show t h a t t he rma l r a t i ngs a r e affected f a r more by increases in case tem- pe ra tu re t han a r e second-breakdolvn

Fi:. 49-Sole-orcn rcrtirr: clrtrrt l o r rlre 2iV35Sj silicorr ~ ~ o w c r lrtrrrsis/or.

tion, t he thernlal l imiting extends t o h ighe r va lues of collector-to- emi t t e r voltagc hefore t h e second- brealtdo~vn region is reached, and a s t he pulse dura t ion decreases, the thernlal-limited region increases.

CASE TEMPERATURE-°C

Fig. 50-Strfe-orca tcr~~~c~totrtrc-dcrntirrg curves l o r file 2N3585 silico~r poivcr

Irarrsislor.

ra t ings . The the rma l (dissil)ation- l imited) de ra t ing curve decreases l inearly t o zero a t the maximum junction t empera tu re of the t r an- s is tor [ T , ( m a s ) = 200"Cj . "he scc- ontf-breakdown (Is/ , , - l i initcd) tem- pe ra tu re dc ra t ing curvr , ho\vever. is less scvcre becausc t h e incrrase in t h e format ion of the high current concentrations t h a t cause second brealcdown is Icss than the increasc in dissipation fac tors a s t he l e m pernt111.e increases.


Because the thermal and second- brealtdown dcratings a r e different, it may be necessary to use both curves to determine the proper derating factor fo r a voltage-current point t h a t occurs near the breakpoint of the thermal-limited and second-breakdown-limited regions on the safe-area curve. F o r this condition, a derating factor is read from each derating curve. For one of the readings, however, either the thermal-limited section of the safe- area curve must be extrapolated upward in v o l t a ~ e or the second- Iweakdown-limited section must be extrapolated downward in voltage, depending upon which side of the voltage breakpoint the voltage- current point is located. The smaller of the collector-current values obtained from the thermal and second- breakdown deratings must be used a s the safe rating.

F o r pulsed operation, the derating factor shown in Fig. 50 must be applied to the appropriate curve on the safe-area rat ing chart. F o r the derating, the effective case temperature Ta(eff) may be approximated by the average junction temperature T,(av). The average junction tem- pernture is determined a s follows:

This approach results in a conserva- tive rating for the pulsed capability of the transistor. A more accurate determination can be made by com- putation of actual instantaneous junction temperatures. (For more detailed information on safe-area ratings and temperature derating the reader should refer to the RCA I'ower Circuits Manual, Technical Series SP-51, pp. 94 to 105.)

HANDLING CONSIDERATIONS

The generation of static charge in tlry weather is harmful to all t ransistors, and can cause permanent damage or catastrophic failure in

the case of high-speed devices. The most obvious precaution against such damage i shumidi ty control in storage and operating areas. I n addition, i t is desirable t h a t transistors be stored and transported in metal t rays rather than in polystyrene foam "snow". During testing and installation, both the equipment and the operator should be grounded, and all power should be turned off when the device is inserted into the socket. Grounded plates may also be used for stockpiling of transistors prior to o r a f te r testing, o r f o r use in testing ovens o r on operating life racks. Further protection against static charges can be provided by use of partially conducting floor planes and non-insulating footwear fo r all personnel.

Environmental temperature also affects performance. Variations of a s little a s 5 per cent can cause changes of a s much a s 50 per cent in the saturation current of a transistor.

. Some test operators can cause marked changes in measurements of saturation current because the heat of their hands affects the transistors they work on. Precautions against temperature effects include air- conditioning systems, use of finger cots in handling of transistors (or use of pliers o r "plug-in boards" to eliminate handling), and accurate monitoring and control of temperature near the devices. Prior t o testing, i t is also desirable t o allow sufficient time (about 5 minutes) fo r a transistor to stabilize if i t h a s been subjected to temperature much higher o r lower than normal room temperature (25°C).

Although transient rf fields a re not usually of sufficient magnitude to cause permanent damage to transistors, they can interfere with accurate measurement of characteristics a t very low signal levels o r a t high frequencies. For this reason, i t is desirable to check for such radiation periodically and t o eliminate i t s causes. I n addition, sensitive measurements should be made in shielded screen rooms if possible.


Care mus t also be taken to avoid the exposure of t rans is tors to o the r ac o r magnet ic fields.

firany t r a~ i s i s to r c l~aractcr is t ics a r e sensitive to variations in tcmpcra- ture, ant1 may change enough a t high operat ing tcmpcraturcs to alFcct circuit performance. Fig. 51 il lustrates the effect of increasing temperature on the common-emitter forward cur-

rent- t ransfer r a t io (be ta ) , the dc collector-cutoff current , and the in- p u t and output impedances. T o avoid undesired changes in circuit opcration, i t is rccolnmcntlcd t h a t t r an- s is tors be located a w a y f rom hea t sources in equipment, and also t h a t provisions be made f o r adequate h e a t dissipation and, if necessary, f o r temperature compensation.

FORWARD CURRENT! TRANSFER RATIO

MOS Field-Eff ect S'

\ \ ' Transistors

! sistors a r e classified, on the basis of TYPES OF : their control-gate construction, a s ' FIELD-EFFECT TRANSISTORS ; either junction-gate types or metal-

I F I E L D - E F F E C T transistors represent a unique and important

category of electronic components. These devices combine many of the desirable characteristics of electron tubes with small size, low power

/ oxide-semiconductor tyaes.7Althou~h

by variation o f a n electric field established by application of a voltage to a control electrode referred to a s the gate. In contrast, current flow in bipolar transistors i s controlled by variation of the current injected

Field-effect transistors (FET's) i both types operate on-t-hedasic pr$- derive their name from the fac t tha t 1 ciple tha t current conduction is con- current flow in them is controlled ',trolled by variation of a n electric

consumption, mechanical rugged- into the base terminal. Moreover, ness, and other advantages inherent L ~ ~ = p e r f o r r n a n c e of bipolar transis- in solid-state devices. ;For example, ors depends on the interaction of these devices can provide a square- two types of charge carriers (holes law transfer characteristic tha t is / and electrons). Field-effect transis- especially desirable fo r amplification 1 tors, however, a r e unipolar devices; of multiple signals in rf amplifiers I a s a result, their operation i s basic- that a r e required to exhibit excep- ally a function of only one type tionally low cross-modulation eRects. 1 of charge carrier, holes in p-

In this section, the basic opera- ' channel devices and electrons in n-

i tion and structure of the various channel devices. types of field-effect transistors a re A charge-control concept can be briefly described and compared. The used to explain the basic operation main emphasis, however, i s placed on ,'metal-oxide-semiconductor field- effect transistors, which a r e becbm- ing increasily popular in electronic- circuit applications, particularly in receiver rf-amplifier and mixer circuits. The fabrication, ,electrical characteristics, biasing, and basic

of field-effect transistors. A charge on the gate (control electrode) in- duces a n equal, but opposite, charge in a semiconductor layer, referred to a s the channel, located directly beneath the gate. The charge induced in the channel controls the conduction of current through the

circuit configurations of these de- channel and, therefore, between the vices a r e discussed, and the integral source and drain terminals which gate-protection system developed f o r a r e connected to opposite ends of the dual-gate types is explained.-/ i channel. fi

ri I Discrete-device field-effect tran-


fieltl, the significant difference in their ga te construction results in uniqtlc characteristics and advantages fo r each type. - *

r.

(- Jpnction-Gate Types C / / / / ' ' ; ) . , <{;" , . ' ~ u n c t i o ~ i - g a t e field-effect transistors, wl~ich are com~uor~lg refrrrctl to a s JFET's or, in popular parlance, a s JUG-FET's, may be either n- channel or p-channel tlevices. F ~ E . 52 shows the structure of a n n-channel junction-gate field-effect transistor, together with the sche~uatic symbols fo r both n-channel and p-channel versions of tlicse devices. The structure for a p-channel device is identical to that of an n-channel device \vith the exception tha t n- and p- type semiconductor ~uater inls a re replaccd by p- and n-type materials, respectively.

In both types of junction-gate devices, a thin cl~anncl under the gate provides a conductive path between the source and drain termi-

GATE TERMINAL

] ,DRAIN TERMINAL

DRAIN

P

n - CHANNEL p-CHANNEL

Fin. 52-Jrrrrctiorr-gntc field-cflcc! tr.rrrtsis- tor (JFEI'): In) .side-ijie~v cro.rs sectiorr of u r ~ rr-clrnrrrtel device; (h) scl~crrtrrtic syrrtbols

for I[- o~rrl p-cltarrrrel de~~ices.

nals with zero gate-bias voltage. A p-n junction is formed a t the interface of the gate and the source-to-drain layer. When this junction is reverse-biased, current conduction in the channel between the source and drain terminals is controlled by the magnitude of reverse-bias voltage, which if suflicient can virtually cut off the flow of current through the channel. If the junction becomes forward-biased, the input resistance (i.e., resistance between the gate and the source- to-drain layer) decreases sharply, and an appreciable amount of gate current flows. Under such conditions, the gate loading reduces the amplitude of the input signal, and a significant reduction in power gain resultQJThis characteristic is a major disadvantage of junction-gate field-effect transistors. Another yn- desirable feature of these devices is that the leakage currents across the reverse-biased p-n junction can vary marltedly with changes in ambient temperature. This latter factor tends to complicate circuit design considerations. Nonetheless, the junction-gate field-effect transistor is a very useful device in many small-signal-amplifier and chopper applications.

Metal-Oxide-Semiconductor TY pes

Figs. 53 and 54 show the structures and schematic symbols for both enhancement and depletion types of metal-oxide-semiconductor field- effect transistors (RIOSIFET'S). In these devices, the nletallic gate is electrically insulated from the semiconductor surface by a thin layer of silicon dioxide. These devices, which a re commonly referred to a s MOS field-effect transistors or, more simply, a s MOS transistors, derive their name from the tri-layer construction of metal, oxide, and sen~iconductor material. Another name sometimes used for them is IGFET, which is a n acronym for insulated-gate field- effect transistor. Insulation of the gate from the remainder of the

MOS Field-Effect Transistors 41

n - CHANNEL (EXISTS ONLY WHEN

GATE IS SUFFICIENTLY OXIDE INSULATION

s O U R ~ s ' T I V \ E ) B G h I N n L / DRAIN TERMINAL (METAL) TERMINAL

Q DRAIN

SUBSTRATE G-ATE

6 SOURCE

0 DRAIN

SUBSTRATE GATE

Fig. 53-Err/tnrlcrrrrerr1-f~~pe rtrelol-oxide- sc~r~rjcorrrl~rcror field-enect rrnr~sistor (MOSI FET): (a ) side-vietv cross src~iort o f nrt 11-clltrrrrrel (levice; ( 6 ) sclterrrolic s)lrrr6ols

of 11- nrtd p-clrar~rrel devices.

transistor structure results in a n exceedingly high input resistance (i.e., in the order of 10" ohms). I t should bc realized that the metal gate and the semiconductor channel form a capacitor in which the oxide layer serves as the dielectric insulator.

T h e marked differences in the construction of enhancement and depletion types of MOS field-effect transistors, a s is apparent from a comparison of Figs. 53(a) and 54(a), results in significant differences in the characteristics of these devices and, therefore, in the applications in which they a re normally employed. (The differences in the

OXIDE INSULATION

SOURCE GATE TERMINAL I DRAIN

TERMINAL (METAL) TERMINAL

Q DRAIN

SUBSTRATE GATE

6 SOURCE

n-CHANNEL

Q DRAIN

SUBSTRATE

SOURCE

Fir. 54-Dep1erior1-type r~tetal-oxide-semi- corrdrrcror field-eflecf fmrrsislor (MOSI FET): (n) side-view cross seclion of art 11-cl~utlrrel device; ( b ) schcnmtic syrrtbols

for n- arid p-charrrrel devices.

42 RCA Transistor, Tllyristor, & Diode Manua l

ctiaractcl.istics of the two types of RIOS t rans is tors a r e discussed sub- sequently in tlie section on Electri- cal Characteristics.)

I ~ n h a ~ ~ c e n i e ~ i t - T y l , r 1)cviccs-As indicntrtl I)y t h e in t r r rupt ions in the channel line of t h e schematic syln- 1)ols slionrn in F i g 53 (b ) , enhance- nicrit-type RlOS field-effect transistors a r e characterized by the f a c t t h a t they have a "normally open" channel so t h a t no useful channel conductivity exists f o r e i ther zero or reverse g a t e hias. Consequently, this type of device is ideal fo r use in digital and switching applications. The g a t e of t he e ~ ~ h a n c e r n c n t t ype of RIOS field-cffect t rans is tor mus t be forward-biased with rcsl)ect t o the source to produce the active charge carr iers in the cliannel required f o r contluction. When s u n - cient for\\.ard-hias (posit ive) volt- a g e is applied to the g a t e of a n n-channel device, tlie region under the g a t r changes f rom p-type to n- type and provides a contluction pa th between the n- type source and dra in regions. Sinlilarly, in p-channel devices, application of suficient ncga- tive g a t e voltage d raws holes into the region helow the ate so t h a t this channel region changes f rom n-type to p-type to provide a source- to-drniri coriduction path .

The technology fo r enliancement- type &IOS field-effect t rans is tors i s nialting i t s g rea t e s t inipact in t h e fa l~r ivat ion of in tegra ted circuits f o r digital applications, particularly in large-scale-integration ( L S I ) circuits.

1)epletion-Type Devices-Deple- t ion-type nlOS field-effect transis- to r s a r e characterized by the f a c t that , wi th zero g a t e bias, the thin channel under the ga te rcgion provides a condrlctive path 1,etwecn the source a n d drain trrmin:lls. In the sclirmatic sy1111)ols f o r tllrsc devices, sl~o\vn in Fig. 51(1)), the c11:lnncl line is clra\vn continuous to indicate th i s "nornially on" condition. IVlien the gate is reverse-biased (n rga t ive with respect to the source fo r n-

channel dcvices, o r positive with respect to the source f o r p-channel devices), the channel can I)e depleted of cha rge carr iers ; conduction in the channel, therefore, can be c u t off if the g a t e potential i s sufTicirntly high.

A unioue characterist ic of del~le- tion-type' hIOS t rans is tors is {hat additional cha rge carr iers can he produced in the channel and, tliere- fore , conduction in the channel can be increased by application of forward bias to the gate. No reduction in power gain occurs under these conditions, a s i s the case in junction- ga te field-effect t rans is tors , because the oxide insulation between the g a t e ant1 the source-to-drain layer blocks the flow of g a t e cu r ren t even when the g a t e i s forward- l~iascd.

The diagram shown in Fig. 5 4 ( a ) i l lustrates the s t ructure of a sinqle- g a t e depletion-type RlOS field-effect transistor. Depletion-type MOS field- effect t rans is tors t h a t have t ~ v o independent insulated g a t e electrodes a r e also available. These devices of- f e r unique advantages and rcpre- sent t he most impor t an t category of MOS field-effect transistors.

F ig . 5 5 ( a ) shows a cross-sectional cliagrani of an n-channel depletion- type dual-gate RIOS field-eKect t r an - sistor. The t rans is tor includes three terminat ing (n-diffused) regions connected by two conductive channels, each of which is controlled by i t s own independent g a t e terminal. F o r convenience of explanation, the t r an - s is tor i s sho\vn divided into two units. Unit No. 1 consists of the source, g a t e No. 1, channel No. 1, and t h e centra l n-region which functions a s dra in No. 1. These elements a c t a s a conventional single-gate depletion-type MOS field-effect t ransistor f o r \vliich unit No. 2 functions a s a load resistor. Un i t No. 2 consists of the centra l n-region, which functions a s source No. 2, ga t e No. 2, channcl No. 2, and the drain. This unit m a y also be used a s a n independent single-gate t rans is tor fo r which unit No. 1 ac t s a s a source resistor. Fig. 55 (b ) shows the sche-


GATE NO I GATE N0.2 TERMINAL TERMINAL (METAL) (METAL)

SOURCE OXIDE DRAIN 'EyM'NAL 1 INSULATION TERMIYAL

UNlT N0.I ] UNlT N0.2 ( 0 )

SOURCE (SUBSTRATE AND CASE)

(b)

Fig. 55-Dirtil-gale 11-chanriel depletion- type ~rieral-oxide-senricottdi4ctor field-effect rati is is tor (MOSIFET): (a ) side-view cross

sectiorr; (b) schematic syntbol.

matic symbol for a n n-channel dual- gate MOS field-effect transistor. ,

Equivalent-circuit representations of the two units in a dual-gate MOS transistor a re shown in Fig. 56.

R9 UNIT NO. 2 16 _ _ _ _ _ _ _ _ _ ---- - UNIT N O I f

- - - ( 0 ) (b)

Fin. 56-Eqirivalerrt-circrti( represerttalio~i of tlre two irrrits ir i a drral-gate MOS

field-effect transistor.

Current can be cut off if either ga te i s sufficiently reverse-biased with respect to the source. When one ga te i s biased to cutoff, a change in the voltage on the other gate is equivalent to a change in the value of a resistor in series with a cut-off transistor.

The dual-gate MOS field-effect transistor is analogous t o a multi- grid electron tube in i ts versatility fo r circuit applications. The independent pair of gates makes this device attractive f o r use in rf amplifiers, gain-controlled amplifiers, mixers, and demodulators. In a gain- controlled amplifier, the signal is applied to gate No. 1, and the gain- control voltage is applied t o gate No. 2. This arrangement is recommended because the forward transconductance obtained with gate No. 1 is higher than tha t obtained with gate No. 2. Moreover, unit No. 2 is very effective fo r isolation of the drain and gate No. 1. This unit provides sufficient isolation so t h a t the dual- gate devices can be operated a t f re- quencies into the uhf range without the need for neutralization. Ex- amples of the use of dual-gate MOS field-effect transistors in circuit applications a r e shown in t h e Circuits section of this Manual.

A gate-protection system which can be incorporated a s a n integral par t of the transistor structure has been developed f o r dual-gate .MOS transistors. In devices t h a t include this system, a set of back-to-back diodes is diffused directly into the semiconductor pellet and connected between each insulated gate and the source. (The low junction capacitance of the small diodes represents a relatively insignificant addition to the total capacitance that shunts the gate.) Fig. 57 shows a cross-sectional diagram and the schematic symbol f o r an n-channel dual-gate-protected depletion-type RlOS field-effect transistor.

The back-to-back diodes do not conduct unless the gate-to-source voltage exceeds + 10 volts typically. The transistor, therefore, can handle


I-DRAIN 2- GATE 2 3-GATE I 4-SOURCE

{SUBSTRATE A N D CASE)

DIODES DIODES

(b)

Fig. 57-Drrnl-gore-protectc(I 11-clr(o11rr1 tleplctiorl-type MOS firld-cflccl tra~rsistor: ( (1 ) side- vic~r, cross scc/iorr; ( b ) sclrcrtraric sy11r6ol.

a very wide dynamic s ignal swing described in more detail in the fol- without significant conductivc shunt- lowing section on In t eg ra l Ga te ing et'fects by thc diodes ( leakage Protection). through the "nonconductive" diodes Dual-gate-protected hlOS transis- i s very low). If the potential on to r s can be connected so t h a t func- either g a t e exceeds + 10 volts typi- t ionally they a r e directly equivalent cally, t h e upper diode [shown in Fig . t o a single-gate type with ga te pro- 57(b)] of the pai r associated with tection. Th i s ~ n c t h o d of connection i s t h a t par t icular g a t e hecomes con- shown in Fig. 58. tluctive in the forward direction and the lower diode breaks down in the INTEGRAL GATE PROTECT~ON backward (zener) direction. In this way, t he back-to-back diode pai r The advent of an integral sys- provitles a pa th to shun t excessive t em of gate-protection in M O S ficld- positive chargc fro111 the gnLc to the efl'ect t rans is tors has rcsultetl in a source. Similarly, if the potential on class of solid-state devices t h a t cx- either g a t c exceeds -10 volts typi- hibits ruggedness on a p a r with , cally, t he loiver cliodc l )econ~es con- o the r solitl-state dcviccs t h a t pro- ductive in the forward tlircction and vide comparable performance. The thc uppcr diode breaks tlown in the gate-protection sys tem mentioned in reverse dircction to provide a shun t the preceding section offers protec- pa th f o r cxcessive negativc charjie t ion aga ins t s ta t ic discharge dur ing f rom t h e g a t e t o the source. ( T h e handling operations without the need diode gate-protection technique is f o r external shor t ing mechanisms.

MOS Field-Effect Transistors

Fig. 58-Cor~tlectiott o f a drtal-gate-protected MOS field-effect transistor (a) so tllat it is frrrtcrionally equivalet~t to a single-gate-protected MOS field-effect transis-

tor (6).

This system also guards against potential damage from in-circuit transients. Because the integral gate- protection system has provided a major impact on the acceptability of MOS field-effect transistors fo r a broad spectrum of applications, i t is pertinent to examine the rudi- ments of this system.

Fig. 59 shows a simple equivalent circuit fo r a source of static electricity tha t can deliver a potential e,, to the gate input of a n MOS

Fig. 59-Eqrtivalcrrt circrtit for a sorrrce o f static electricity.

transistor. The static potential ER stored in an "equivalent" capacitor C,, must be discharged through an internal generator resistance Rs. Laboratory experiments indicate that the human body acts a s a static (storage) source with a capacitance CI, ranging from 100 to 200 picofarads and a resistance Rn greater than 1000 ohms. Although the upper limits of accumulated static voltage can be very high, measurements sug- gest that the potential stored by the

human body is usually less than 1000 volts. Experience has also indicated tha t the likelihood of damage t o an MOS transistor a s a result of static discharge is greater during handling than when the device is installed in a typical circuit. In a n rf application, for example, static potential discharged into the antenna must traverse an input circuit that normally provides a large degree of attenuation to the static surge before i t appears a t the gate terminal of the MOS transistor. The ideal gate-protection signal-limiting circuit is a configuration that allows for a signal, such a s t h a t shown in Fig. 60(a), to be handled without clipping or distortion, but limits the amplitude of all transients tha t exceed a safe operating level, a s shown in Fig. 60(b). An arrangement of back-to-back diodes, shown in Fig. 60 (c), meets these requirements f o r protection of the gate insulation in MOS transistors.

Ideally, the transfer characteristic of the protective signal-limiting diodes should have a n infinite slope a t limiting, a s shown in Fig. 61(a). Under these conditions, the static potential across Cn in Fig. 61(b) discharges through i ts internal impedance Rs into the load represented by the signal-limiting diodes. The ideal signal-limiting diodes, which have a n infinite transfer slope, would then


Fig. 61-Trnrrsjer clrnrocrcrisric of , prolec- The fabrication techniques used to rive cliorles (n) , arrd resrrltirrg n~o~vforr,rs irr produce hfOS transistors a re similar

eqrrivnletrt circrtir (b). to those used for modern high-speed

IR drop across the internal imped- (+) -----...------ ( . ) - - - ante of the source R., i.e., e. = E. - ' i n * J__[z- e , ~ where E is the potential in the

(-1 (-1 - - - - - - - - - - - - source of static electricitv and c , ~ is the diode voltage drop. The instan-

( a ) P A S S 'IGNAL ( b ' C ~ k P E ~ ~ ~ ~ , " , " N taneous value of the diode current is

(t) AND ( - ) IN then equal to e./R.. During physical AMPLITUDE handling, practical peak values of

DRAIN currents produced by static-electricity discharges range from several nlilliamperes to several hundred milliamperes.

Fig. G2 shows a typical transfer characteristic curve measured on a typical set of back-to-back diodes used to protect the gate insulation in a n MOS field-effect transistor that is nominally rated for a gate-to- source breakdown voltage of 20 volts.

SOURCE J a w

( c ) BACK-TO-BACK DIODES PROTECT Z GATE INSULATION 5 5

1.2-.8;

i-? 5 5 C1 W [r

- - 5 5 -.fu

: : I - .. 4 12 2 0 28 3;

PEAK GATE VOLTS

--

Fig. 60-bfOS gore-prorecfiorr rcqrrircrrrerrts -.

nrrd a solrttiorr. 0.8 - - -. lirnit the voltage present a t the gate 0 . 4

terminal to its knee value, e ,~. The -32 - 2 4 - 1 6 -8 differcnce voltage e, appears a s an

I -0,4..

-0.8

- 10

I I

E -1.2- 1 + 10 - I I -a Fi,q. 62-Typicol rliodc trrrrtsjer cltnroctcr- I I isric rrteo.rrrrrr1 ivith I-rrricrosecorrrl prrlsc

ividrlr a! o drrry factor o f 4 x lo-".

The transfer-characteristic curves show that the diodes will constrain

(a ) a transient impulse to potential values well below the k 2 0 volt limit, even when the source of the transient surge is capable of delivering several hundred nlilliamperes of current. (These data were measured with 1-n~icrosecond pulses applied to the protected gate a t a duty- factor of 4 x 10.").

I :

(b) FABRICATION ' \ , . r

M O S Field-Effect Transistors 47

silicon bipolar transistors. The s tar t- ing material fo r a n n-channel transistor is a lightly doped p-type silicon wafer. (Reversal of p-type and n-type materials referred to in this description produces a p-channel transistor.) After the wafer is polished on one side and oxidized in a furnace, photolithographic tech- , niques a re used to etch- away- the oxide coat in^ and expose bare silicon in the source and drain regions. The source and drain regions a re then formed by diffusion in a furnace containing a n n-type impurity (such a s phosphorus). If the transistor is to be a n enhancement-type device, no channel diffusion is required. If a depletion-type transistor is desired, a n n-type channel is formed to bridge the space between the diffused source and drain.

The wafer is then oxidized again to cover the bare silicon regions, and a second photolithographic and etching s tep is performed t o remove the oxid;-in the contact regions. After metal is evaporated over the entire wafer, another photolithographic and etching s tep removes all metal not needed f o r the ohmic contacts to the source, drain, and gate, . The individual transistor chips a r e then mechanically separated and mounted on individual headers, connector wires a re bonded to the metalized regions, and each unit is hermetically sealed in i ts case in a n inert atmosphere. After testing, the external leads of each device a r e physically shorted together to prevent electrostatic damage to the gate insulation during branding and shipping.

ELECTRICAL CHARACTERISTICS

The basic current-voltage relationship fo r a n MOS transistor is shown in Fig. 63. With a constant gate-to- source voltage (e.g., VC;S = O), the resistance of the channel is essentially constant, and current varies directly with drain-to-source voltage (Vl,%), a s illustrated in region A-B.

The flow of drain current (I,,) produces an IR drop along the channel. The polarity of this drop is such a s to oppose the field produced within the gate oxide by the gate bias. As the drain voltage is increased, a point is reached a t which the IR drop becomes sufficiently high so tha t

.the capability of the gate field to at t ract enough carriers into the channel to sustain a higher drain- current is nullified. When this condition occurs (in the proximity of point B in Fig. 63), the channel is essentially depleted of carriers (i.e., becomes "constricted"), and drain current increases very much more slowly with fur ther increases in drain-to-source voltage VllP. This condition leads to the description of region B-C a s the "pinch-off" region because the channel "pinches off" and the drain current (11,s) tends to saturate a t a constant value. Beyond point C, the transistor enters the "breakdown" region (also known a s the "punch-through" region), in which unrestricted current flow and damage to the transistor result if current flow is not limited by the external circuit.

A B C DRAIN- TO-SOURCE VOLTAGE ( V D S l

Fig. 63-Basic clrrrerrt-voltage relalionsl~ip for art MOS transislor.

MOS transistors a r e especially useful in high-impedance voltage amplifiers when they a re operated in the "pinch-offJ' region. The direct variation in their channel resistance (Region A-B in Fig. 63) makes them very attractive fo r use in voltage-


controlled rcsistor applicatior~s, such a s the chopper circuits used in con- ncction with some typcs of dc amplifiers.

Typical o u t p u t c h a r a c t e r i s t i c curves fo r n-channel RIOS transistors a re shown in Fig. 64. The rcsem- blance of these curves to the hasic curve shown in Fig. 63 should be noted. (For p-channel transistors, the polarity of the voltages and the direction of the current a re reversed.) Typical transfer charxctcr- istics fo r n-channel single-gate MOS transistors a r e shown in Fig. 65. (Again, voltage polarities and current direction would be reversed for p-channel devices.) The threshold voltage (V711) shown in connection with the enhancement-type transistor illustrates the "normally-open"

DRAIN- TO- SOURCE VOLTAGE (VDS)

1 DEPLETION T Y P E

DRAIN- TO- SOURCE VOLTAGE ( VDS)

_ 1 E N H A N C E M E N T TYPE

H - /-- I-

(CONSTANT

I

GATE- TO- SOURCE VOLTAGE (VGS)

GATE-TO-SOURCE VOLTAGE (VGSl

Fig. 65-Typical trarr.rfer clraracteristics for rr-charrrrel MOS trartsistors.

source-drain characteristic of the device. In these transistors, conduction does not begin until Vcs is increased to a particular value. Fig. G G shows typical drain-current curves

GATE-NO. I - TO- SOURCE VOLTAGE -V

Fig. 66-Drailr crrrrort of n clrrcrl-gore MOS trtrrrsislor n.7 a frrrrc~iorr of ,gate-No. I-to-

ri.g. 64--T].picn/ orr l~~rrt -c /~or-nct~~ri .~t ic vorrrce ~.o/rogc for se130.ir/ ~'o/rrrs o f gale- c r r ~ - ~ ~ , s for 11-clrrr~rrrc~l hIOS t~orrsislors. N o . 2-to-sorrrce ~ ~ o l l a g c .


fo r a dual-gate device a s a function conductance. The source voltage VS, of ga te No. 1-to-source voltage for the source resistance Ra, and the dc several values of gate No. 2-to- supply voltage Von can then be readi- source voltage. ly calculated, a s follows:

BIASING TECHNIQUES FOR SINGLE-GATE

MOS TRANSISTORS

The bias required for operation of a single-gate MOS transistor can be supplied by use of a self-bias (source-bias) arrangement, from a supply of fixed bias, or, preferably, by a combination of these methods. Fig. 67 illustrates each of the three biasing techniques.

The design of a self-bias circuit is relatively simple and straight- forward. For example, if a 3N128 MOS transistor is to be operated with a drain-to-source voltage VDS of 15 volts and a small-signal transconductance gr, of 7400 micromhos,

Vx = Va - V o s = 1.1 volts Ra = Vs/Io = 1.115 = 220 ohms

Voo = Vns + Vs = 15 + 1.1 = 16.1 volts

The self-bias arrangement is satisfactory for some applications. A particular source resistance, however, must be selected for each device if a specified drain current is required because the drain-current characteristics of individual devices can vary significantly from the typical values. The dashed-line curves in Fig. 68(b) define the "high" and "low" limits f o r the characteristics of the 3N128 MOS transistor. F o r example, the zero-bias drain current Ioas can vary from a low value of 5 milliamperes

Fig. 67-Rinsitrg nrmrr,qorrerr!s for sirrgle-gnte MOS trnrrsislors: (a) self-bias circrtit; ( 6 ) fixed bins slipply; fc) conlbitrntion o f self bias and fixed bias.

the drain current 11, required for the specified value of transconductance is first obtained from published curves, such a s those shown in Fig. G8(a). Next, the gate-to-source voltage required for this value of drain current is determined from another published curve, such a s the solid- line curve shown in Fig. 68(b) . These curves indicate that the drain current should be 5 milliamperes and t h a t the gate-to-source voltage should be -1.1 volts for the specified values of drain-to-source voltage and trans-

to a high value of 25 milliamperes, a range of 20 milliamperes. Use of a source resistor of 220 ohms, a s calculated in the preceding example, reduces the range of the drain current between "high" and "low" 3N128 transistors operated in self- bias circuits from 20 milliamperes to about 4 milliamperes. A reduction of about 5 to 1 in the range of Ioss values among individual devices can be achieved, therefore, by a judicious choice of the proper value of source resistance.


COMMON-SOURCE C I R C U I T

I -

P

g - 4 -3 -2 - 1 0 I GATE-TO-SOURCE VOLTAGE (VGS)-VOLTS

(b)

Fig. 68-0pcrntirr.q clrnrnctrrir~ir.r for /Ire KCA-3N17R MOS trarr.ristor: l o ) for~~,ctr.d Irntrscorrtlrrctflrfce ns a frtrrcfiorr o f tlroirr cfrrrerrf; (h ) [irflirr crrrrotf rrs o f~trrctiorr o f

gotc-to-sorrrce vo/lo,qc.

Fixed-bias-supply sys tems, such a s t h a t shown in Fig. G8(b), a r e generally unaLtl.activc fo r usc with R40S transis tors f o r two main reasons. Fi rs t , this type of systeni is undesirable because i t requires the use of a separa te , negative-voltage power supply. Second, a s shown by the curves in Fig. F 8 ( b ) , f o r a fixed bias supply of 1.1 volts, dra in current ~vould be 14 mill iamprres f o r a "high" 3N128 t rans is tor and would be cut off f o r a "low" device. Con- sequently, if a n external bias sys- tctn i s used provisions mus t be

made f o r adjus tnient of the bias voltage if a specific dra in cu r ren t i s required fo r a par t icular device.

The combination bias s y s t e n ~ shown in Fig . G7(c) is the most effective a r r angemen t when a n application requires a specific dra in cu r ren t despite the r anne of drain- c u r ~ . e n t cha~~ac te r i s t i c s encountcl.etl a m o n r individual devices. Fig. 60 shows two families of characterist ic curves developetl empirically f o r the combination bias systern shown in Fig . G7(c). The family of curves on the lef t i s pertinent fo r operation a t a dra in cu r ren t of 5 milliamperes. F o r operation a t a dra in cu r ren t of 10 n~i l l iamperes , the family of curves on the r igh t should be used.

If a dra in cu r ren t of 5 mi l l i a~n- peres is desired, t he per t inent curves in Fig . G9 show tha t , f o r a soufce resistance of 1000 ohms, a bias sys- t em can provide this value of cur- r e n t within l nlilliampefe ( a s indicated by projections of lines a and h to the abscissa), despite a ranEe of 5 to 25 mill iamperes in the value of 11,sa f o r individual devices. A drain cu r ren t 11) of 5 milliamperes, however, develops a self bias of -5 volts across the 1000-ohm source resistor Rs, and the t rans is tor will I)e cu t off unless sufficient positive bias is applied across the input resistors (RL and R?) to establish the correct opera t ing point. The positive bias voltage can be obtained f rom the positive dra in supply Vll l l SO t h a t there i s no need f o r a separa te bias supply. F o r a drain-to-source volt- a g e VIBS of 15 volts, a drain cu r ren t 11, of 5 milliamperes, a gate-to-source voltage VGF of -1.1 volts, and a source resistance RE of 1000 ol i~ns , the circuit parameters f o r t he coni- bination bias system shown in Fig. G7(e) can be calculated a s follows:

vs = 11,Rs = (0.005) (1000) = 5 volts

vc = \',:s + vs = -1.1 + 5 = 3.0 volts

Vno = Vns +' Vs = 15 + 5 = 20 volts

V I , I , / ~ O = (R, + Re) I I L = 2013.0 = 5.12


DRAIN CURRENT (ID)-MILLIAMPERES

Fig. 69-Drairr curreirt I D as a frrrrctiotr of zero-bias drain clrrrent ID, , for several valrres of source resislance R,.

The lower limits fo r the values of the input resistors RI and R2 a r e determined on the basis of the maximum pern~issible loading of the input circuit. The resistance tha t corresponds to this value is set equal to the equivalent value of the parallel combination of the two resistors. For e x a m ~ l e . if the total resistance

RIR?/(RI + R2) = 50,000 (R1 + Rz) /Ra = 5.12

Therefore, RI = 256,000 ohms and Ra = 62,000 ohms.

In rf-circuit applications, the effects of input-circuit loading can be circumvented by use of the circuit arrangement shown in Fig. 70.

in shunt with the input circuit is to be no less than 50,000 ohms, the BIASING TECHNIQUES values of Rl and R3 are calculated FOR DUAL-GATE as follows: MOS TRANSISTORS

The following example illustrates the techniques used to provide the bias required for operation of a dual- gate MOS transistor. This example assumes a typical application in which a 3N140 dual-gate MOS transistor is required to operate with a drain-to-source voltage VI,R of 15 volts and a forward transconduct-

Rs ;cCS ance gf , of 10,500 micromhos. (The techniques described f o r the 3N140 transistor a re also applicable to dual- gate-protected MOS transistors.)

- - - The characteristic curves fo r the 3N140, shown in Fig. 71(a), indicate that the desired value of transcon-

f i g . 70-Circrtic rrsed lo e!i,,riirnte input- ductance can be obtained for a gate circrrit loodiirg irr rf-ar~rplifier applicatioirs. NO. 1-to-source voltage Val* of -0.45


volt and a g a t e No. 2-to-source volt- a g e V(;2s of +4 volts. Tlie curves in Fig. 71 (b ) show t h a t f o r these conditions the dra in cu r ren t 11, i s 10 milliamperes.

m

COMMON-SOURCE CIRCUIT AMBIENT TEMPERATURE (TA)= 25°C

Fig . 72-Typical hinsitrg circrrit /or drrol- gntc MOS field-eflect rrtrtr.risrors.

GATE NO.-TO-SOURCE VOLTAGE ( V G I ~ ) - - - V 0 ~ ~ ~

( a )

P GATE N0. I - TO- SOURCE VOLTAGE (VGIS ) - VOLTS

( b )

Fi,v. 71-Oprratitr~ rlrartrclerislic..~ for llre RCA-31V140 rlrrul-golc M O S 1,-crtr.ristor: ( ( 1 ) jont.ard rratrscotrdr~clntrce trs n frrtrcliotr of xnfe-hro. I-to-sorrrce rolrojie; (1)) tlroirr crtrreril ar n frtrrcriotr o/ ficrrr-No. I-lo-

sorrrce rollrtjie.

Fig. 72 sliows a b i a s i n ~ nrrange- nlent t h a t can be used fo r ciual-gate bIOS field-effect transistors. Fo r the application being considered, the

(REVERSE) !::c$+),~~

shun t resistance f o r g a t e No. 1 is assumed to be 25,000 ohms. Gate No. 2 i s operated a t rf ground (by mcpns of adequate bypassing) and is I~iascd wi th a fixed dc potential. Empirical experience with dual-gate hlOS t ransistors has shown t h a t a source resistance of approx in~a te ly 270 ohms provides adequate self-bias fo r the t rans is tor f o r operation f rom the proposed dc supply v o l t a ~ e . For this value of source resistance, the remaining pa ramete r s of the 1)ias circui t a r e obtained f rom the following calculations:

Vs = It,Rs = (0.010) (270) = +2.7 volts

V,;, = V,;,, + Vs = (-0.45) + (+2.7) = +2.25 volts

V,;? = V,;zs + Vs = (+4.0) 4- (+2.7) = +6.7 volts

V,,t, = Vl,s + Vs = (+15) 1- ( f2 .7) = f17.7 volts

The values of the voltage-divider resistances required to provide the appropr ia te voltage a t each g a t e a r e determined in a manner similar to t h a t described fo r single-gate hIOS transistors. The value calculated fo r R1 i s 197,000 ohms, t h a t f o r R , is 28,600 ohms, and the ra t io R , /R , is 11.67.

The circuit shown in Fig. 73 is normally used in rf amplifier applications. I n this circuit, t he sijinal


voltage is applicd a t point "a" through appropriate input circuitry. If the agc fcature is not employed, (e.g. in mixer circuits), the resistor R,,, is disconnected a t point "b." In a mixcr application, the local oscillator signal is injected a t point "b."

GENERAL CIRCUIT CONFIGURATIONS

There a re three basic single-stage amplifier configurations f o r MOS transistors: common-source, common-gate, and common-drain. Each of these configurations provides certain advantages in particular applications.

The common-source arrangement shown in Fig. 73 is most frequently used. This configuration provides a

Fig. 73-Bosic con~~~lorr-sortrce circzril /or MOS field-eflcct ~rorrsislors.

h i ~ h input impedance, medium to high output impedance, and voltage gain greater than unity. The input signal is applied between gate and source, and the output signal is taken between drain and source. The voltage gain without feedback, A, for the common-source circuit may be determined a s follows:

RII ru r Rr. A =-

r.. + Rr. where gf, is the gate-to-drain forward transconductance of the t ransistor, r.. is the common-source output resistance, and Rr, is the effective load resistance. The addition of a n unbypassed source resistor to the circuit of Fig. 73 produces negative voltage feedback proportional to the output current. The voltage

gain with feedback, A', f o r a common-source circuit i s given by

gf. ro. RI. A'=

r,. + (g~. r.. + 1) RS + RL where Rs is the total unbypassed source resistance in series with the source terminal. The common-source output impedance with feedback, Z,, is increased by the unbypassed source resistor a s follows:

The common-drain arrangement, shown in Fig. 74, is also frequently referred to a s a source-follower. I n this configuration, the in- pu t impedance is higher than i n the common-source configuration, the output impedance is low, there is no polarity reversal between input and output, the voltage gain is always less than unity, and distortion is low. The source-follower is used in applications which require reduced input-circuit capacitance, downward impedance transformation, o r increased input-signal-handling capability. The input signal is effectively injected between gate and drain, and the output is taken between source and drain. The circuit inherently has 100-per-cent negative

Fig. 74-Basic co~nmorz-drain (or source- Jollower) circrril lor MOS lransislors.

voltage feedback; i ts gain A' is given by

Because the amplification factor ( p ) of a n MOS transistor is usually much greater than unity, the equation f o ~


gain in the source-follower can be simplified a s follows:

For example, if i t is assumed tha t the gate-to-drain forward transconductance gt, is 2000 micromltos (2 x lo-' mho) and the unhypassed source resistance RS is 600 ohms, the stage gain A' is 0.5. If the same source resistance is used with a transistor having a transconductance of 10,000 micromhos ( 1 x 10.' mho), the stage gain increases to 0.83.

When the resistor Ra is returned to ground, as shown in Fig. 74, the input resistance 1t1 of the source- follower is equal to Ro. If Ro is returned to the source tcrn~innl , however, the effective input resistance RI' is given by

n, R,' = -

1 - A'

where A' is the voltage amplification of the stage with feedback. For example, if Ra is one nlegollm and A' is 0.5, the effective resistance RI' is two megohms.

If the load is resistive, the effective input capacitance CI' of the source-followcr i s reduced by the inherent voltage feedbnclc and is given b y

where c,.~ and c,, are the intrinsic gatc-to-drain and gate-to-source capacitances, respectively, of the RlOS tmnsistor. For example, if a typical RIOS transistor having a c,,~ of 0.3 picofarad and a c,, of 5 picofarads is used, and if A' is equal to 0.5, thcn CI' is reduced to 2.8 picofarads.

The effective output resistance R,,' of the source-follower stage is ~ i v e n by

r,,. Rs R,' =

(gr. r,,. + 1) Rs + r,. where r,,, is the transistor common- source output resistance in ohms. F o r example, if a unit having a gate-to-drain forward transconduc-

tance g f , of 2000 micronlhos and a common-source output resistance r.. of 7500 ohms is used in a source- follower s tage with a n unbypassed source resistance Ra of 500 ohms, the efective output resistance R,,' of the source-follower s tage is 241 ohms.

The source-follower output capacitance C,' may be expressed a s follows:

where c,~. and c,, a re the intrinsic drain-to-source and gate-to-source capacitances, respectively, of the D10S transistor. If A' is equal to 0.5 (as assumed for the sample input-circuit calculations), C,' is reduced to the sum of c,~. and c,..

The common-gate circuit, shown in Fig. 75, is used to transf6rm from a low input impedance to a

high output impedance. The input inlpetlnnce of this configuration has approximately the same value a s the output impedance of the source-follower circuit. The common-gate circuit is also a desirable configul.ation for high-frequency applications because i ts relatively low voltage gain makes neutralization unnecessary in most cases. The common-gate voltage gain, A, is given by

(gt. rum + 1) RI, A = -

(gr. r,, + 1 ) Rn + r,. f Rr.

where RG is the resistance of the input-signal source. F o r a typical MOS transistor (gr. = 2000 micromhos, r,. = 7500 ohms) and with RI. = 2000 ohms and Rn = 500 ohms, the common-gate voltage gain


is 1.8. If the value of Rn is doubled, the voltage gain is reduced to 1.25.

TECHNICAL FEATURES

I t is apparent f rom the tliscussions tha t MOS field-effect transistors exhibit a number of technical features t h a t result in unique performance advantages in circuit applications such a s mixers, product detectors, remote gain-control circuits, bnlaneed modulators, choppers, clippers, and gated amplifiers. These features include:

1. An extremely high input resistance and a low input capacitance-as a result, MO transistors impose virtually no loading on ,an agc voltage source (i.e., virtually no agc power is required) and have a wide agc range capability.

2. A wide dynamic range-MOS transistors, therefore, can handle positive and negative input-signal excursions without diode-current loading.

3. Cross-nlodulation effects and spurious response tha t a re substantially less than those of other types of electronic devices-the cross- modulation characteristics of dual- gate transistors actually improve a s the device approaches cutoff.

4. Zero offset voltage-this feature is cspecinlly desirable for chopper applications.

5. An exceptionally high forward tmnsconductance.

6. Negative temperature coeffi- cier~t for the drnin current-"thermal

runaway," therefore, is virtually im- possible.

7. A very low gate leakage current t h a t is relatively insensitive to temperature variations.

8. Very low osciIlator feedthrough in dual-gate mixer circuits.

9. Dual-gate transistors can provide good gain in common-source amplifiers into the uhf range without neutralization.

HANDLING CONSIDERATIONS

MOS field-effect transistors, like high-frequency bipolar transistors. can be damaged by exposure to excessive voltages. The gate oxide insulation is susceptible to puncture when subjected to voltage in excess o f ' t h e rated value. The very high i resistance of the oxide insulation. i imposes a negligible load on electro- :

statically generated potentials and, i therefore, provides a n ineffective discharge path fo r sources of static ,; electricity. As discussed earlier, the integral gate-protection system incorporated into some types of dual- gate MOS transistors is highly effective in the protection of these devices against the effects of electrostatic charges. Special precautions, however, must be taken in the handling and application of other types of MOS transistors tha t do not contain the integral gate protection. '

The tliscussion of blOS Transistors in the section on Testing and Mount- ing outlines the special handling pro- ; cedures recommended for such devices.

Thyristors 57

I

Thyristors

T HE term thyristor is the generic name for solid-state devices

t h a t have characteristics similar to those of thyratron tubes. Rasically, this group includes bistal~lc solid- s tate devices tha t have two or more junctions (three or more semiconductor layers) and that can be switched between conducting s tates (from O F F to ON or from ON to O F F ) within a t least one quad- r a n t of the principal voltage-current characteristic: Ileverse-blocking triode thyristors, comnionly called silicon controlled rectifiers ( SCR's) , and bidirectional triode thyristors, usually referred to a s triacs, have three electrodes and are switched between states by a current pulse applied to the gate terminal. The bidirectional trigger diode, commonly called a diac, has only two electrodes. This device has no gate electrode hut nlay be switched from an OFF s ta te to an ON state fo r either polarity of applied voltage. \The discussions in this section deal pri~narily with the SCR and the triac, their operation, electrical characteristics, and ratings. A brief description is also given of the operation of the diac and its chief function in trinc phase: control circuits. \ , ,

, SILICON CONTROLLED ' , ,,. RECTIFIERS

!,?I? I,;,.!\.

1 h*si~i$on controlled rectifier (SCR) is basically a four-layer p-n-p-n device that has three clectrodes ( a cathode, an anode, and a_-control electrode called the gate), Fig. 76 shows the junction diagralii, principal voltage-current charactevistic, and schematic symbol for an SCR.

the device is switched to the ON state a t the instant desired.

After the SCR is triggered by the gate signal, the current through the device is independent of the gate voltage or gate current. The SCR remains in the ON state until the principal current i s reduced to a level below tha t required to sustain conduction.

Construction details of a typical SCR pellet a re shown in Fig. 78.

A CATHODE ELECTRODE

rent increases rapidly and the SCR switches to the ON state.;This value of voltage is called the breakover voltage. :When the SCR is in the ON state, the forward current is limited primarily by the. impedance of the external circuit.

Under reverse bias (anode negative with respect to cathode), the SCR exhibits a very high internal impedance, and only a small amount of current, called the reverse blocking current, flows through the device. This current remains very small and the device remains in this O F F state unless the reverse voltage exceeds the reverse-breakdown-voltage limitation, A t this point, the reverse current increases rapidly, and the SCR undergoes thermal runaway, a condition tha t norlnally causes irreversible damage

i to the device. The value of reverse breakdown voltage differs for individual SCR types, but is approximately 100 volts greater than the forward breakover voltage for most types. Under forward-bias conditions, the breakover voltage of the SCR can be controlled or varied by application of a current pulse to the

.gate electrode,%s shown in Fig. 77. ! As the amplitude of the gate current pulse is increased, the breakover voltage for the SCR decreases until

i,\\\ the curve _closely resembles that of a rectifie-_fn normal operation, the SCR is operated with critical values well below the breakover voltage and is made to switch on by gate signals , of sufficient magnitude to assure that

I

TERMINAL TERMINAL

1 QUADRANT I

A N ~ D E ELECTROM

Fig. 78-Cross-section of a typical SCR pellet.

The shorted-emitter construction used in RCA SCR's can be recognized by the metallic cathode electrode in direct contact with the p-type base layer around the periphery of the pellet. The gate, a t the center of the pellet, also makes direct metallic contact to the p-type base so that the portion of this layer under the n-type emitter acts a s an ohmic path fo r current flow between gate and cathode. Because this ohmic path is in parallel with the n-type emitter junction, current preferentially takes the ohmic path until the IR drop in this path reaches the junction threshold voltage of about 0.8 volt. When the gate voltage exceeds this value, the junction current increases rapidly, and injection of electrons by the n- type emitter reaches a level high enough to turn on the device.

~ E ~ E R ~ E ~ ~ ~ C " K : ~ ~ ~ K O V E R STATE \

7

\ REVERSE BREAKDOWN

VOLTAGE

QUADRANT IIt

ANODE (t)

HOLDING CURRENTlh

-- ---------- ;OLT~-GE v

%FF~?TATE

(b)

.with respect to cathode) the SCR has two states. ! ~ t low values of forward bias, the SCR exhibits a very high impedance;,' in this forward-blocking or O F F state, a snla11 for\vard current, called the forward OFF-state current. flows tllr0uiZh the device. As t h e forward bias is increased,..howdver, a voltage point is reached a t which the forward cur- -

ANODE(-)

CATHODE

ANODE (CASE)

Fi,?. 76-(0) Jrtrrcrio,r diopr-orrr, f h ) l~ritrci- '

pol voltage-c~rrrerrr chnl-octrristic, alrcf f c ) 1, sclier~iatic s).rrfhol for art SCR fltwisfor.

Fig. 76(b) shows that' under forward-bias conditions (anode positive

I

/ I I I ; ; " Ig4> I g 3 > I g 2 > I g l = 0

Fin. 77-Crrn,rs slro~virtg the jor~t-ard-~,olr- rigc clrorocteristics o f a tltgristor for dif-

ferctrt valtces o f gate currer~t. I

In addition to I,rovitling :L precisely controlled gate current, tlic shorted- emitter construction also improves the high-temperature and dvldt (niaxim~um allowable rate of rise of OFF-state voltage) capa1)ilities of the device.

The center-gate construction of the SClt pellet provitlcs fas t turn-on and high di ldt capa1)ilities. In :In SCR, conduction is initiated in the cathode region im'nlediatcly adjacent t o the gate contact and must then propagate to the more remote regions of the cathode. Switching losses are influenced by the rate of propagation of conduction and the distance conduction must propagate from tlie gate. With a central gate, all regions of the cathode are in close proximity to the initially conducting region so that propagation distancc is significantly decreased; a s a result, smitch- ing losses a r e minimized.


Fig. 79 shows the junction tlia- gram, voltage-current cliaracteris- tic, and schematic symbol for a triac. The triac, liltc the SC11, has three electrodes; they are designated a s main tcrminal No.1, maill terminal No.2, and the 'gate. As sho\vn in Fig. 'iD(b), the triac exhibits tlie same forward-blocl~ing, forward- conducting voltage-current characteristic of the SCR, \)ut for either polarity of voltage applicd to the main terminals. Untlcr forw;lrd bias (main terminal No.2 ~ ~ o s i t i v e with respect to main ter~ninal No.1) or reverse bias (main tcrminal No.2 negativc with respect to mnin terminal No.l), the triac cxliibits first a forward-bloclting ( O F F ) state, then a forward-conducting (ON) state. The point a t which thc device switches states is the brcal io~er voltage. Again like the SCIt, the breakover voltage of tlie triac can be controlled or varicd by application of a positive or negative current pulse to the gate electrode. As the amplitude of tlie current ~ ~ u l s e is

Thyristors MAlN

TERMINAL I T E F % ? ~ F ~ ~ ~ ~ ~

TERMINAL

(01

'I I QUADRANT I

MAIN TERMINAL 2(*)

"ON' STATE n

MAlN TERMlNAL l 0

, , --

MAIN TERMINAL 2 (CASE)

Fig. 794") Jrrrrctiort rIio,qr(trrt, (1)) />rirrci- pol vol~oge-crrrrer~t clrorocrc~risfic, or~rl (c)

schcrrrntic syrrrbol /or n trim th),ri.rtor.

increased, the brealcover point of the triac is dccrcased. The triac can therefore be considerctl a s two SCR's connected in parallel and oriented in opposite directions, a s shown in Fig. 80.

MAlN TERMINAL 2 P

()MAIN TERMINAL I

Fi,q. 60-A tr.;nc rqrti~olr~rr~ circrrit: / I ! , O SCR's irl parallrl nrrd or-ierrted irr opposite

rlir.ectiorrs.

Construction of a typical RCA triac pellet is sho\vn in Fig. 81. 111

this device, the main-terminal-No. 1 electrode makes ohmic contact to a p-type emitter a s well a s to an n-type emitter. Similarly, the main-terminal- No. 2 electrode also makes ohmic contact to both types of emitters, but the p-type emitter of the main- terminal-No. 2 side is located opposite the n-type emitter of the main- terminal-No. 1 side, and the main- terminal-No. 2 n-type emitter is opposite the main-terminal-No. 1 p-type emitter. The net result is two four- layer switches in parallel, but oriented in opposite directions, in one silicon pellet. This type of construction makes i t possible for a triac either to block o r to conduct current in either direction between main terminal No. 1 and main terminal No. 2,

n-TYPE

EMITTER I - WIN-TERMINAL-NO 2

ELECTRODE

Fig. ~ ~ - C ~ - O S S - S C C I ~ O I ~ of o typical triac pcllet.

A diac is a two-electlsode, three- layer bidirectional avalanche diode which can be switched from the O F F state to the ON state for either polarity of applied voltage. Fig. 82 shows the junction diagram, voltage- current characteristic, and schematic symbol fo r a diac.

This three-layer trigger diode is similar in construction to a bipolar

6 Fig. 82-(aJ Jrorctiorr diagranr, (bJ voltage-cfirrent characteristic, and (cJ sclle.

nlatic syr~lbol /or a diac.

transistor. A diac differs from a bipolar transistor in tha t the doping concentrations a t the two junctions are approximately the same and there is no contact made to the base layer. The equal doping levels result in a symmetrical bidirectional switching characteristic, a s shown in Fig. 82(b). When a n increasing positive o r negative voltage is applied across the terminals of the diac, a minimum (leakage) current I,Iw,, flows through the device until the voltage reaches the breakover point Vcno,. The reverse-biased junction then undergoes avalanche breakdown and, beyond this point, the device exhibits a negative-resistance characteristic, i.e., current through the device increases substantially with decreasing voltage.

Diacs are primarily used a s triggering devices in triac phase-control circuits used for light dimming, universal motor-speed control, heat control, and similar applications. Fig. 83 shows the general circuit diagram for a diac/triac phase- control circuit. The magnitude and

RCA Transistor, Thyristor, & Diode Manual I Thyristors

Fig. 83-Gerferfll circftit diflgrflttl for a diacltriac phase-conrrol circriit.

duration of the current pulse applied to the gate of the trinc a r e determined by the value of phase- shift capacitance C, the change. in voltage across and the dynamlc Im-

pedance of the diac, and the triac gate impedance. The interaction of all circuit in~pedances and the phase- shift capacitance can best be represented by the curve of peak current a s a function of the capacitance shown in Fig. 84.

SCR AND TRlAC GATE .'

CHARACTERISTICS

Silicon controlled rectif ers and triacs a r e ideal for switching applications. When the working voltage of the thyristor is below the breakover point, the device is essentially an ope11 s\\fitch; above the brcaltovcr voltngc, thc thyristor

switches to the ON s t a b and is effectively a closed switch. The breakover voltage can be varied or controlled by injections of a signal a t the gate terminal.

The manufacturer's specifications indicate the magnitude of gate current and voltage required to turn on these devices. Gate characteristics, however, vary from device to device even among devices within the same family. For this reason, manufacturer's specifications on gating characteristics provide a range of values in the form of characteristic diagrams. A diagram such a s that shown in Fig. 85 is given to define the limits of gate currents and voltages that may be used t o trigger any given device of a specific family. The boundary lines of 'maximum mid minimum ga te impedance on this characteristic diagram representsthe loci of all possible triggering points fo r thyristors in this family. The curve OA represents the gate characteristic of a specific device tha t is triggered within the shaded area.

The magnitude o f gate current and voltage required to trigger a thyristor varies inversely with junction temperature. As the junction temperature increases, the level of gate signal required to trigger the thyristor becomes smaller. Worst-case triggering conditions occur, therefore, a t

ALLUNITS AT THESE TEMPERATURES

4 I

GATE CURRENT-&

the minimum operating junction temperature.

The gate nontrigger voltage V,, is the maximum dc gate voltage that may be applied between gate and cathode of the thyristor f o r which the device can maintain its rated blocking voltage. This voltage is usually specified a t the rated operating temperature (100°C) of the thyristor. Noise signals in the gate circuit should be maintained below this level t o prevent unwanted triggering of the thyristor.

When very precise triggering of a thyristor is desired, the thyristor gate must be overdriven by a pulse of current much larger than t h e dc gate current required to trigger t h e device. The use of a large current pulse reduces variations in turn-on time, minimizes the effect of temperature variations on triggering characteristics, and makes possible very short switching times.

The coaxial gate structure and the "shorted-emitter" construction techniques used in RCA thyristors have greatly extended the range of limiting gate characteristics. A s a result, the gate-dissipation ratings of RCA thyristors a r e compatible with the power-handling capabilities of other elements of these devices. Advantage can be taken of the higher peak- power capability of the gate t o improve dynamic performance, increase di/dt capability (maximum allowable rate o f -r i se of .ON-state current), minimize interpulse jitter, and reduce switching losses. This higher peak-power capability also allows greater interchangeability of thyristors in high-performance applications.

The forward gate characteristics for thyristors, shown in Fia. 86, indicate the maximum allowable pulse widths for various peak values of gate input power. The pulse width is determined by the relationship that exists between gate power input and the increase in the temperature of the thyristor pellet tha t results from the application of gate power.

The curves shown in Fig. 86(a) a r e fo r RCA SCR's t h a t have relatively small current ratings (2N4101, 2N4102, and 40379 families), and the curves shown i n Fig. 86(b) a r e f o r RCA SCR's t h a t have larger current rat ings (2N4103, 2N3873, and 2N3899 families). Because

GATE CURRENT-A (b)

Fig. 86-Forward gate characteristics for pulse triggering of RCA SCR's: (R) low-

currot1 lypes: (b) high-current types.

the higher-current thyristors have larger pellets, they also have greater thermal capacities than the smaller- current devices. Wider ga te trigger pulses can therefore be used on these devices f o r the same peak value of gate input power.

Because of the resistive nature of the "shorted-emitter" construction, similar volt-ampere curves can be constructed for reverse ga te voltages and currents, with maximum allowable pulse widths f o r various peak- power values, a s shown in Fig. 87. These curves indicate t h a t reverse dissipations do not exceed the maximum allowable power dissipation for the device.

The total average dissipation caused by gate-trigger pulses is the sum of the average forward and re-

verse dissipations. This total dissipation should be less than the Rlaxi- mum Gate Power Dissipation P G ~ I shown in the published data for the selected SCR. If the average gate dissipation exceeds the niaximum published value, a s the result of high forward gate-trigger pulses and transient or steady-state reverse gate biasing, the maximum allowable forward-conduction-current rating of the device must be reduced to co~npensate for the increased rise of junction temperature caused by the increased gate power


dissipation The triac can be triggered in any

of four operating modes, a s sunima- rized in Table I. The quadrant designations refer to the operating quadrant on the principal voltage- current characteristics, shown in Fig. 79 (either I or 111), and the polarity

Thyristors

symbol represents the gate-to-main- terminal-No. 1 voltage.

Table I-Triac Triggering Modes

Gate-to-Main- Main-Terminal-No. 2-to- Operating Terminal-No. 1 Main-Terminal-No. 1 auadrant

Vol ta le Voltage

Positive Positive I ( + )

Negative Positive I(--)

Positive Negative I11 (+)

Negative Negative 111 (-)

The gate-trigger requirements of the triac are different in each operating mode. The I ( + ) mode (gate positive with respect to main terminal No. 1 and main terminal No. 2 positive with respect to main terminal No. l ) , which is comparable to equivalent SCR operation, is usually the most sensitke. The smallest gate current is required to trigger the triac in this mode. The other three operating modes require larger gate- trigger currents. For RCA triacs, the maximum trigger-current rating in the published data is the larxest value of gate current that is required to trigger the selected device in any operating mode.

Gate Trigger Circuits

The gate signal used to trigger an -0.5 -0.4 -03 -02 -01 0

REVERSE GATE CURRENT-A SCR or triac must be of sufficient strength to assure sustained forward conduction. Triggering requirements are usually stated in terms of dc voltage and current. Be- cause i t is comtnon practice to pulse-fire thyristors, i t is also necessary to consider the duration of

. firing pulse required. A trigger pulse that has an amplitude just equivalent to the dc requirements must be applied for a relatively long period of time (approximately 30 microseconds) to ensure that the gate signal

REVERSE GATE CURRENT-A is ~ rov ided durinrr the full turn-on . .

(b) Fiq. 8 7 - R ~ v e r ~ p gare rl~ar~rcrrristic~ o f

period of the thyFistor. As the am-

RCA S C R ~ ~ ~ (a ) ~o lu-rrrrre~ t r types; ( b ) plitude of the gate-triggering signal high-crrrrerrt t)'pes. is increased, the turn-on time of the

thyristor is decreased, and the width of the gate pulse may be reduced. When highly inductive loads a r e used, the inductance controls the current-rise portion of the turn-on time. For this type of load, the width of the gate pulse must be made long enough to assure that the principal current rises to a value greater than the latching-current level of the device. The latching current of RCA thyristors is always less than twice the holding current.

The application usually determines whether a simple o r somewhat sophis t i ca ted t r i g g e r i n g c i rcu i t should be used to trigger a given thyristor. Triggering circuits can be a s numerous and a s varied a s the applications in which they a re used; this text discusses the basic types only.

Many applications require t h a t a thyristor be switched full ON or full OFF in a manner similar to the operation of a relay. Although higher currents a re handled by the thyristor, only small trigger or gate currents are required from the control circuit or switch. The simplest method of accomplishing this type of triggering is illustrated in Fig. 88.

Each circuit shows a variable resistor in the gate circuit to control the conduction angle of the thyristor.

I .A 90"- y Bc

4-7 i ; MIN 4 ..

I I*._.' 8 1.

I I I I '.

O C - Y +90e BC': k 9 0 ' MIN

( 0 ) MIN

(b)

Fig. 88-Degree o f corlfrol over cot~drtc- lion ar~gles ~vhetr ac resisfive tterwork is

zcsrd f o trigger SCR's arld friacs.

The waveforms indicating the degree of control exercised by the variable resistance a re also shown in Fig. 88. With maximuni resistance in either circuit, the thyristor i s

OFF. As the resistance is reduced in the SCR circuit, a point is reached a t which sufficient gate trigger current is provided a t the positive peak of the voltage wave (90 degrees) to trigger the SCR ON. The SCR conducts from the 90-degree point to the 180-degree point for a total conduction angle of (180 - go), or 90 degrees. In the triac circuit, a s the resistance is reduced, the gate current increases until the triac is triggered a t both the peak positive (90 degrees) and peak negative (270 degrees) points on the voltage wave. The triac then conducts between 90 degrees and 180 degrees, and between 270 degrees and 360 degrees for a total conduction angle of 180 degrees. The conduction angles of both the SCR and the triac can be increased by further reduction of the resistance in the gate circuits. For the SCR, the firing point is moved back from 90 degrees toward zero for a total conduction angle approaching 180 degrees. The triac firing points can also be moved back from 90 degrees toward zero for .the positive half-cycle and from 270 degrees toward 180 degrees for the negative half-cycle to obtain a total conduction angle approaching 360 degrees. The resistor in the gate circuit assures tha t the gate current decreases to a negligible value af ter the thyristor is fired.

An easier method of obtaining a phase angle greater than 90 degrees for half-wave operation is to use a resistance-capacitance triggering network. Fig. 89 shows the simplest form of such networks for use with an SCR and a triac. The thyristor is in series with the load and in parallel with the RC network. A t the beginning of each half-cycle (positive half-cycle only for the SCR), the thyristor is in the O F F state. As a result, the ac voltage appears across the thyristor and essentially none appears across the load. Because the thyristor is in parallel with the potentiometer and capacitor, the voltage across the thyristor drives current through the

RCA Transistor, Thyristor, & Diode Manual I Thyristors

Fig. 89-RC tri,q,cerirrp rrc.tu~orks rrscd lor phase-cotztrol trigpri11.g of tlryrislors.

potentionleter and charges the capacitor. When thc capacitor voltage reaches the brealtover voltage of the thyristor, the capacitor discharges through the gate circuit and turns the thyristor on. At this point, the ac voltage is transferred from the thyristor to the load Rr, fo r the remainder of the half-cycle. If the potentiometer resistance is reduced, the capacitor charges more rapidly, and the breakover voltage is reached earlier in the cycle; a s a result, the power applied to the load is increased.

The gate trigger voltage can be more closely controlled in simple resistance or resistance-capacitance circuits by use of a variety of special t r i g ~ e r i n g devices. These triggering devices, including the diac, have a sn~al ler range of cllaracteristics, and are less temperature-sensitive. Basically, a thyristor triggering device exhibits a negative resist-ance a f te r a critical voltage is reached, so t h a t the gate-current rcquirc- ment of the thyristor can be oh- tained a s a pulse from the discharge of the phase-shift capacitor. Be- cause the gate pulse need 1)e only microseconds in durnlion, the g:rte- pnlse energy and the size of the t r i g ~ e r i n ~ co~nponenls :ire relatively small. Triggering circuits of this

type employ elc~nents such as ncon bulbs, diacs, unijunction transistors, and two-transistor switches.

Fig. 90 shows a l i g h t - d i m ~ ~ l i n ~ circuit in which a diac is used to trigger a triac. The voltage-current

LINE VOLTAGE

Fig. 90-A I i ~ / ~ t - d i ~ t r r ~ ~ e r circ~rit it! % I ~ / I ~ c / I a diac is used to trigger a triac.

characteristic fo r the diac in this circuit is shown in Fig. 01. The magnitude and duration of the gate-current pulse a re determined

NEGATIVE RESlSTANCE

CURRENT jI" ,,p+ + I

VOLTAGE

Vp- - \ -i NEGATIVE

RESISTANCE

Fig. 9/-Volm,~e-cztrrrr1t chamctc!ri.ilic for triggerirrg device sl~ott~rr irr Fig. 90.

by the interaction of the capacitor C,, the diac characteristics, and the impedance of the thyristor gate. Fig. 92 shows the typical shape of the gate-current pulse that is produced.

T I M E

Fig. 92-Ty/>ir~rl ,gc~rc,-rtrrrcrrt ~t'rrl~?Jorrrr for circrrit slrott~rr irt Fig. 90.

SWITCHING CHARACTERISTICS

The ratings of thyristors a r e based primarily upon the amount of heat generated within the device pellet and the ability of the device package to transfer the internal heat t o the external case. F o r high-frequency applications in which the peak-to-average current ratio is high, or f o r high- periormance applications t h a t require large peak values but narrow current pulses, the energy lost during the t u n - o n process may be the main cause of heat generation within the thyristor. The switching properties of the device must be known, therefore, to determine power dissipation which may limit the device performance.

When a thyristor is triggered by a gate signal, the turn-on time of the device consists of two stages, a delay time t r and a rise time t,, a s shown in Fig. 93. The total turn-on time tKt is defined a s the time interval between the initiation of the gate signal and the time when the resulting current through the thyristor reaches 90 per cent of its maximum value with a resistive load. The delay time t , ~ is defined a s the time interval between the 10-per-cent point of the leading edge of the gate-trigger voltage and the 10-per-cent point of the

3 I I

ANODE CURRENT

GATE TRIGGER "GT PULSE

L- '<IfOlNL--- - Fie. 93-Gate-crtrrerir arrd voltage tro.t~-or2

~r~avcf~~rtrrs jor a thyrrstor.

resulting current with a resistive load. The rise time t, is the time interval required for the principal current t o rise from 10 t o 90 per ccnt of its maximum value. The total turn-on time, thercfore, is the

sum of both the delay and rise times of the thyristor.

Although the turn-on time is affected to some extent by the peak OFF-state voltage and the peak ON- state current level, i t is influenced primarily by the magnitude of the gate-trigger current pulse. Fig. 94 shows the variation in turn-on time with gate-trigger current f o r the RCA-2N3873 SCR.

0 l I I I I I 0.1 0.3 0.5 0.7 0.9 1.1

GATE CURRENT-& F ~ K . 94-Rarlge of trrrn-OIZ tinte as a f~trtc- tion o f gate crcrrerrt for the 2N3873 SCR.

To guarantee reliable operation- and provide guidance f o r equipment designers in applications having short conduction periods, the voltage drop across RCA thyristors, a t a given instantaneous forward current and a t a specified time af ter turn-on from a n OFF-state condition, is given in the published data. The waveshape for the initial ON-state voltage for the RCA-2N3873 SCR is shown in Fig. 95. This initial voltage, together with the time required for reduction of the dynamic forward voltage drop during the spreading time, is a n indication of the current- switching capability of the thyristor.

When the entire junction area of a thyristor is not in conduction, the current through t h a t fraction of the pellet area in conduction may result in large instantaneous power losses. These turn-on switching losses a re proportional to the current and the voltage from cathode t o anode of the device, together with the repetition rate of the gate-trigger pulses. The instantaneous power dissipated in a

RCA Transistor, Thyristor, & Diode Manua l

may exceed the maximunl oper:lting x temperature given in the manufac-

" ( ~ 0 1 ~ turer's data; in this case, the thyristor should not be required to block voltages immediately af ter the conduction interval. If the thyristor

--t--- - --f , must block voltages immediately following the conduction interval, the

"~(11 I junction-temperature rating must not

I be exceeded.

I The turn-off time of an SCR also consists of two stages, a reverse-

T recovery time and a gate-recovery time, a s shown in Fig. 97. When the

Fig. 95-Irritiol orr-slate volto~?c nrlrl crfr- rerlt ~r.ovelorrrls /or the 2N3873 SCK.

v;i9

\ i I w;- thyristor under such conditions is d i r d t 1 I I *

shown in Fig. 96. The curves shown I in this figure indicate that the pealc I

I

LC-A-1 r ig . 97-~ircrtit-corrrtr111101(~[1 I tq -4 trtrrr-on \,nt/- "RM age arrd ctrrrerrt wavejorr~ls /or a tlr~.ris/or.

I I

power dissipation occurs in thc short interval in~nlcdiately after the device s ta r t s to conduct, usually in the first microsecond. During this time interval, the peak junction temperature

forward current of an SCR is rcduced to zero a t the end of a conduction period, application of reverse voltage between the anode and cathode terminals causes reverse current to flow in the SCR until the reverse-blocking junction establishes a depletion region. The time interval between the application of reverse voltage and the time tha t the reverse current passes its peak value to a steady- state level is called the reverse- recovery time t.,. A second recovery period, called the gate-recovery time t,,, must then elapse for the forward- blocking junction to establish a forward-depletion region so that forward-blocking voltage can be re- applied and successfully blocked by the SCR.

Thyristors

The gate-recovery time of a n SCR is usually much longer than the reverse-recovery time. The total time from the instant reverse-recovery current begins to flow to the s ta r t of the re-applied forward-blocking voltage is referred to a s the circuit commutated turn-off time t,. The turn-off time is dependent upon a number of circuit parameters, including the ON- state current prior to turn-off, the rate of change of current during the forward-to-reverse transition, the reverse-blocking voltage, the rate of change of the re-applied forward voltage, the gate trigger level, the gate bias, and the junction temperature. The junction temperature and the ON-state current, however, have a more significant effect on turn-off time than any of the other factors. Because the turn-off time of a n SCR depends upon a number of circuit parameters, the manufacturer's turn- off time specification is meaningful only if these critical parameters are listed and the test circuit used for the measurement is indicated.

Thyristors must be operated within the maximum ratings specified by the manufacturer to assure best results in terms of performance, life, and reliability. These ratings define .limiting values, determined on the basis of extensive tests, that represent the best judgment of the manufacturer of the safe operating capability of the device.

VOLTAGE RATINGS

The voltage ratings of thyristors are given for both steady-state and transient operation and for both forward- and reverse-blocking conditions. For SCR's, voltages a re considered to be in the forward or positive direction when the anode is positive with respect to the cathode. Negative voltages for SCR's a re referred to a s reverse-blocking voltages. For triacs, voltages a r e considered to be positive when main terminal No. 2 is positive with respect to main terminal No. 1. Alter-

natively, this condition may be referred to a s operation in the first quadrant.

OFF-State Voltages

The repetitive peak OFF-state voltage V,, t l , f is the maximum value of OFF-state voltage, either transient or steady-state, tha t the thyristor should be required to block under the stated conditions of temperature and gate-to-cathode resistance. If this voltage is exceeded, the thyristor may switch to the ON state. The circuit designer should insure t h a t the Vlmx rating is not exceeded to assure proper operation of the thyristor.

Under relaxed conditions of temperature or gate impedance, or when the blocking capability of the thyristor exceeds the specified rating, i t may be found that a thyristor can block voltages f a r in excess of its repetitive OFF-state voltage rating VI3nv. Because the application of an excessive voltage to a thyristor may produce irreversible effects, a n absolute upper limit should be imposed on the amount of voltage tha t may be applied to the main terminals of the device. This voltage rating is referred to a s the peak OFF-state voltage Vl,,!. I t should be noted tha t the peak OFF-state voltage has a single rat ing irrespective of the voltage grade of the thyristor. This rating is a function of the construction of the thyristor and of the surface properties of the pellet; i t should not be exceeded under either continuous or transient conditions.

Reverse Voltages (SCR's only)

Reverse voltage ratings a re given for SCR's to provide operating guidance in the third quadrant, or reverse-bloclting mode. There a re two voltage ratings for SCR's in the reverse-bloclting mode: repetitive peak reverse voltage (Vnnxl) and nonrepetitive peak reverse voltage (VllS,,).

Tlie rcpctitivc pcali revcrsc voltage is the maximum allowable value of rcvcrse voltage, including a11 repetitive transient voltages, that may be applied to the SCR. Because reverse power dissipation is sinall a t this voltage, the rise in junction temperature because of this reverse dissipation is very slight and is ac- coyntcd for in the rating of the SCR.

The nonre~et i t ive peak reverse voltage is the maxinium allowable value of any nonrepctitive transient reverse voltage which may be applied to the SCR. These nonrepetitive transient voltages a r e allowcd to exceed the steady-state ratings, even though thc instantaneous power dissipation can be significant. While the transient voltage is applied, the junction temperature may increase, bu t removal of the transient voltage in a specified time allows the junction teinperature t o return t o its steady- s tate operating temperature before a thermal runaway occurs.


ON-State Voltages

Thyristors

IVIicn a thyristor is in n high- corlduction state, the voltage drop across the device is no diflerent in nature from the forward-conduction voltage drop of a scmiconductor diode, although the magnitude may be slightly higher. As in diodes, the ON-state voltage-drop characteristic is the major source of power losses in the operation of the thyristor, and the teniperatures produced become a limiting feature in the rating of the device.

CURRENT RATINGS The current ratings fo r SCR's a i d

triacs define maximum values for normal or rcpctitive currents and for surge or nonrepetitive currents. These maximum ratings a re determined on the basis of thc maxi- mu111 junction-temperature rating, the junction-to-case thermal resistance, the internal pourer dissipation tha t results frotii the current

flow tlirougll thc thyristor, and t l ~ c ambient temperature. The effect of these factors in the determination of current ratings is illustrated by the following example.

Fig. 98 shows curves of the niaximum average forward power dissipation for the RCA-2N3873 SCR a s a

Fit!. 98-P~r~~rr-dissiporio, l rntirrg rhnrf {or the 2N3873 SCR.

function of average forward current for dc operation and for various conduction angles. For the 2N3873, the junction-to-case thcrmal resistance el-c is 0.9Z0C per wat t and the maximum operating junction temperature T j is 100°C. If the maximum case temperature Trc,n.r, is assumed to be 65"C, the maximum average forward powcr dissipation can be determined as follows:

TlglllnX) -TC~~,,..> PAFG(~. .~) =

01-0

= 38 watts

The maximum average forward current rating for the specified conditions can then be determined from the rating curves shown in Fig. 98. For example, if a conduction angle of 180 degrees is assumed, the average forward current rat ing f o r a maximum dissipation of 38 watts is found to be 22 amperes.

These calculations assume tha t the temperature is uniform throughout the pellet and the case. The junction temperature, however, increases and decreases under conditions of transient loading or periodic currents, depending upon the instantaneous power dissipated within the thyristor. The current rat ing takes these variations into account.

The ON-state current ratings f o r a thyristor indicate the maximum values of average, rms, and peak (surge) current that should be allowed to flow through the main terminals of the device, under stated conditions, when the thyristor is in t h e ON state. F o r heat-sinlc-mounted thyristors, these maximum ratings a r e based on the case temperature; fo r lead-mounted thyristors, the ratings a r e based on the ambient temperature.

The maximum average ON-state current rating is usually specified for a half-sine-wave current a t a particular frequency. Fig. 99 shows curves of the maximum allowable average ON-state current I ' r ~ c n r p ) fo r the RCA-2N3873 SCR family a s a function of case temperature. Be- cause peak and rms currents may be high for small conduction angles, the curves in Fig. 99 also show maximum allowable average currents a s a function of conduction angle. The maximum operating junction temperature f o r the 2N3873 is 100°C. The rating curves indicate, f o r a given case temperature, the maximum average ON-state current fo r which the average temperature of the pellet will not exceed the maximum allowable value. The rat ing curves may be used for only resistive or inductive loads. When capacitive loads a re used, the currents produced

Fig. 99--C~crrerrt ratirtg chart for the 2N3873 SCR.

by the charge or discharge of the capacitor through the thyristor may be excessively high, and a resistance should be used in series with the capacitor to limit the current to the rating of the thyristor.

The ON-state current rat ing for a triac is given only in r m s values because these devices normally conduct alternating current. Fig. 100 shows a n rms ON-state current rat ing curve

Fig. 100-Cttrrertf rafitrg curve /or a typical RCA triac,


for a typical triac a s a function of case temperature. As with the SCR, the triac curve is deratcd to zero loo

current when the case tenlperature 5; rises to the mnximunl o p c r a t i n ~ junc- $; tion temperature. Triac cul,rcbnt rat- Y$g ings a r e given for full-wave conduc- Fz tion under resistive o r inductive $? loads. I'rccautions slioultl be talcen to ;;I liniit the pealc current to tolerable L': levels when capacitive loads a r e used. 26 o ~ ~ ~ , " ~ ~ , S T E A O r - s T A T E RATEo/--l / j 1

The surge ON-state current rat inc I 2 4 6 B i 0 2 4 6B1& 2 4 6~~~~

ITP,.,,.~,) indicates the maximutn peak sURGE CmEM WRATW-FULL CYCLES

value of a short-duration current Fig. I U ~ - S I I I , ~ ~ - C I I ~ ~ ~ I I ~ rati~rg crfrvc lor a pulse that should be allowed to flow typical Iriac.

through a thyristor during one ON- s tate cycle, under stated conditions. cal triac. For triacs, the rat in^ curve This rating is for any shows peak values fo r a full-sine- rated load condition, During normal wave current a s a function of the operation, the junction tenlperature nulnber of cycles of overload dura- of a tllyristor rise to the maxi- tion. Multicycle surge curves a rc the nlllln allowallle value; if tile surge basis for the selection of circuit occulos a t this timc, the maximum breakers and fuses tha t a re ~ i s e d to limit is exceeded. For this reason, a prevent damage to the thyristor in thyristor is not rated to l)loclt OFF- the event of accidental -short-circuit s ta te voltage immediately following of the device. The number of surges the occurrence of a current surge. permitted over the life of the thy- Sufficient time must be allowed to ristor should be limited to prevent permit the junction temperature to device return to tlie normal operating value 1)cforc nate control is restored to the t11~risto.r. Fig. 101 shows a surge- current rating curve for the 2N3873 RATE OF OF

ON-STATE CURRENT (dildt)

Fi,r. 101-Srcypc,-crtrr.e,rr r.nri11.q crtrre for. rhc 2N3873 SCR.

In an SCR or triac, the load current is initially concentrated in the small area of the pellet where load current first begins to flow. This small area ef~ectivcly limits the a n ~ o u n t of current tha t the device can handle and results in a high voltage drop across the pellet in the first microsecond af ter the thyristor is t r i g ~ e r e d . If the rate of rise of current is not maintained within the rat ing of the thyristor, localized hot spots may occur within the pellet and permanent d a ~ n a a e to the device mav result. Thc wave-

SCR. This curve shows pealc values shape for testing tlie di ldt cap- of half-sine-wave forward (ON-state) ability of the RCA 2N3873 is current a s a function of overload shown in Fig. 103. The critical rate duration measured in cyclrs of the of rise of ON-state current is clepen- GO-Hz current. Fig. 102 shows a dent upon the size of the cathode surge-current rating curvc for n typi- area that begins to conduct initially,

Thyristors

and the size of this area is increased f o r larger values of gate trigger current. For this reason, the d i ld t rating is specified f o r a specific value of gate trigger current.

Fin. 103-Voltage atrd crcrretrt wavejon~rs rrsed to dcrerttritre di/dt ratitrg of the

2N3873 SCR.

HOLDING AND LATCHING CURRENTS

After an SCR or triac has been switched to the ON-state condition, a certain n~inimum value of anode current is required to maintain the thyristor in this low-impedance state. If the anode current is reduced below this critical holding- current value, the thyristor cannot maintain regeneration and reverts to the O F F or high-impedance state. Because the holding current (111) is sensitive to changes in temperature (increases a s temperature decreases), this rating is specified a t room temperature with the gate open.

The . latching-current rat ing of a thyristor specifies a value of anode current, slightly higher than the holding current, which is the minimum amount required to sustain conduction immediately af ter the thyristor is switched from the O F F state to the ON state and the gate signal is removed. Once the latching current (Ir.) is reached, the thyristor remains in the ON, or low-impedance, s ta te until i ts anode current is decreased below the holding-current value. The latching-current rat ing is

an important consideration when a thyristor is to be used with a n inductive load because t h e inductance limits the ra te of rise of the anode current. Precautions should be taken to insure that, under such conditions, the gate signal is present until the anode current rises to the latching value so t h a t complete turn-on of the thyristor is assured.

CRITICAL RATE OF RISE OF OFF-STATE VOLTAGE (dv/dt)

Because of the internal capacitance of a thyristor, the forward-blocking capability of the device is sensitive to the ra te a t which the forward voltage is applied. A steep rising voltage impressed across the main terminals of a thyristor causes a capacitive charging current t o flow through the device. This charging current (i = Cdvtdt) is a function of t h e ra te of rise of t h e OFF-state voltage.

I f the rate of rise of the forward voltage exceeds a critical value, the capacitive charging current may become large enough t o trigger the thyristor. The steeper the wavefront of applied forward voltage, the smaller the value of the thyristor breakover voltage becomes.

The use of the shorted-emitter construction in SCR's has resulted in a substantial increase in the dvldt capability of these devices by providing a shunt path around the gate-to- cathode junction. Typical units can withstand rates of voltage rise up to 200 volts per microsecond under worst-case conditions. The dv ld t capability of a thyristor decreases a s the temperature rises and is increased by the addition of a n external resistance from gate t o reference terminal. The dvtdt rating, therefore, is given f o r the maximum junction temperature with the ga te open, i.e., f o r worst-case conditions.

TRANSIENT PROTECTION

Voltage transients occur in electrical systems when some disturb-


A Ion ance disrupts t h e normal opera t ' of the system. These disturbances may be produecd by various sources (such a s lighting surges, cncrgiz- ing transformers, and load switching) ant1 may generate volt:~gc!s which exceed the rating of tlic thyristors. In addition, transients gcnerally have a i a s t ra te of rise t11;lt is usually greater than the critical value for the rate of rise of the thyristor OFF-state voltage (static dvldt).

If transient voltages have mngui- tudes f a r greater than the device rating, the thyristor. may switch from the O F F state to the ON state, and energy is then transferred from the thyristor to the load. Because the internal resistance of the thyris- to r is high during the O F F state, t h e transients may cause considerable energy to be dissipated in the thyristor before hreakover occurs. In such instances, the transient colt- a g e exceeds the ~ n a x i m u n ~ allowable voltage ratinp, and irreversible damage to the thyristor nlay occur.

Even if the magnitude of a transient voltage is within tlre ~nns imum allowahle voltnre rating of t h e thy-

cause of overvoltage or because the thyristor dv ld t capability is exceeded.

One of the obvious solutions to insure that transients do not exceed the maximum allown1)le volt.- agc rating is t o provide a thyristor \\.it11 :. voltage rating greater than thc highest transient voltage expected in a system. This technique, however, does not represent a n economical solution because, in most cases, the transient magnitude. which is dependent on the source of transient generation, i s not easily defined. Transient voltages a s high a s 2G00 volts have resulted from lighting disturbances on a 120-volt residential power line. Usually, the best solution is to specify devices tha t can withstand voltage from 2 t o 3 times t h e steady-state value. This technique provides a reason- able safety factor. The effects of voltage transients can fur ther be minimized by use of external circuit elements, such a s RC snubber networks across t h e thyristor terminals, a s shown in Fig. 104. The ra te a t which the voltage rises a t the thyris- to r terminal is a function of the load

. ~ - . .

ristor, the rate-of rise'-of the transient niay exceed the static dvfdt capability of the thyristor and cnuse the device to switch from thc O F F state to the ON state. This condition also results in transfer of energy from tlre thyristor t o the load. In this case, thyristor switching from the O F F state to the O N s ta te does not occur because the nlaximunl allowable voltage is exceeded but, instead, occurs because of the f a s t ra te of rise of OFF-state voltafre (dvldt) and the thyristor capacitance, which result in a turn-on current i = Cdvldt. Thyristm switching produced in this may is free from high-energy dissipation, and turn-on is not destructive provided tha t the current that rcsults from the energy transfer is within the device capability.

In either case, transient suppression techniques a re eml~loyed to minimize the effects of turn-on be-

impedance and the values of the resistor R and the capacitor C in the snubber network. Because the load impedance is usually variable, the preferred approach is to assume a worst-case condition for the load and, through actual transient measurement, to select a value of C that provides the minimum rate of rise

Thyristors

a t the thyristor terminals. The snubber resistance should be selected to minimize t h e capacitor discharge currents during turn-on.

F o r applications in which it is necessary to minimize false turn- on because of transients, the addition of a coil in series with the load, a s shown in Fig. 105, is very effective for suppression of transient rise times at the thyristor terminals. For example, if a transient of

POWER INPUT

Fig. 105-Sz~~pressiotz of trattsicrrt rise firnes at tlre terinirrnls of a thyristor By

nrcalrs of a coil 61 series with the load.

infinite rise time i s assumed to occur a t the input terminals and if the effects of the load impedance a re neglected, the rise time of the transient a t the thyristor terminals is npproxin~ately equal to E , ~ / ~ / L C . If the value of the added inductor L is 100 nlicrohenries and the value of the snubber capacitor C is 0.1 microfarad, the infinite rate of rise of the transient a t the thyristor terminals is reduced by a factor of 3. For a filter network consisting of L = 100 microhenries, C = 22 microfarads, and R = 47 ohms, a 1000-volt-per-microsecond transient that appears at the input terminals is suppressed by a factor of 6 a t the thyristor terminals.

COMMUTATING CAPABILITY dv/dt

In ac power-control applications, n triac must switch from the conducting s tate to the blocking s ta te

a t each zero-current point, or twice each cycle, of the applied a c power. This action i s called commutation. If the triac fails to block the circuit voltage ( turn off) following the zero-current point, this action is not damaging to the triac, b u t control of the load power is lost. Commu- tation f o r resistive loading presents no special problems because the voltage and current a r e essentially in phase. F o r inductive loading, however, the current lags the voltage so that, following t h e zero- current point, a n applied voltage opposite to the current and equal to the peak of the ac line voltage occurs across the thyristor. The maximum ra te of rise of this voltage which can be blocked without the triac reverting to the ON state is termed the critical ra te of rise of commutation voltage, o r the commutating dv ld t capability, of the triac.

SCR's do not experience commutation lilnitations because tun-on is not possible f o r the polarity of voltage opposite to current flow. The commutating dv td t i s a major

operating characteristic used to describe the performance capability of a triac. The characteristic can be more easily understood if the triac pellet, shown in Fig. 106, is considered to be divided into two halves.

GATE

Fig. 106-1~~1criot1 diagrant for a friac pellet.

One half conducts current in one direction, the other half conducts in the opposite direction. The main blocking junctions and a lightly doped n-type base region in which charge can be stored a re common to both halves of the t r iac pellet. (The base region i s the section shown between the dotted lines in Fig. 106.)


Charge is stored in the base when current is conducted in either direction. The amount of chargc stored a t the end of each half-cgclc of conduction depends on the co~nniut:~ting dildt, i.e., the rate of decrease of load current a s commutation is approached. The junction capacitance of the triac a t commutation is a function of the remaining c h a r ~ e a t tha t time. The greater the dildt, the more remaining charge, and the greater the junction capncitance. When the voltage changes direction, the remaining charge diffuses into the opposite half of the triac structure. The rate of rise of t!lis voltage (commutating dvldt) in conjuncticn with the junction capacitance results in a current flow ~vhich, if large enough, can cause the triac to re- vert to the conducting s tate in the absence of a gate signal.

The commutating dvldt capability is specified in volts per microsecond for the following conditions:

1. the maximuni rated on-state current [I-r(RRIS)];

2. the ~naximunl case tenipcrature for the rated value of on-state current;

3. the maxinlum rated off-state voltage (V~wm,) ;

4. the maximum comniutating di ldt (where di ldt = I,,t sin ot and w = 2sf).

I t is apparent, therefore, that the frequency ( f ) of the applied ac power is a n important factor in determination of the co~il~nutat ing dvldt capability of a triac.

Fig. 107 indicates how the commutating dv ld t capability of a triac depends on current and frequency. A particular triac has a spedfic commutating dv ld t capability :it the rated GO-IIz on-state current. If this GO-IIz on-state current is reduced (dashed-line), then its associated commutating dvldt capability is increased. I t should be noted tha t although the sine-wave current is decreased in magnitude, the commutating di ldt is also decreased. For a

400-Hz on-state current of the same magnitude, it is evident tha t the commutating di /dt is much greater than a t (i0 Hz and, thcrcforc, the commutating dv ld t capability is greatly reduced. These relationships indicate tha t a triac capable of 400- Hz operation must have an extrenie- ly high commutating capability.

COMMUTATING dl/dt

Fig. 107-Deperidorce o f triac corrtntrttat- irrg cupabili~y orr cttrrerit artd /reqrrertcy.

RCA offers a complete line of triacs rated for 400-Hz operation. Appli- cations of such devices a r e described in the section on I'ower Switching and Control.

I t should be evident tha t 400 Hz is not an upper liniit on frequency capability fo r triacs; 400 IIz is a characterization point simply because i t is a standard operating frequency. Figs. 108 and 109 indicate how the frequency capability of a typical RCA 400-Hz triac can be increased. Fig. 108 shows tha t reduction of load current increases frequency capability. Maximuni rated junction temperature and minimum ratcd commutating dv ld t a r e held constant fo r this test of capability. Fig. 109 shows the effects of junction temperature on frequency capability. F o r this test, rated current and minimum rated dv ld t are held constant. Therefore, if a typical

Thyris tors

400-Hz triac is used a t less than i ts maxim.um rated junction temperature and less than its rated current, its frequency capability i s greatly enhanced.

LZ LL I I I

0 50 100

PERCENTOFRATEOCURRENT

Fig. 108-Frcqtret~cy capability o f a 400-Hz ~r iac as u f~t~criorr O/ load crtrrertt.

One other factor that greatly affects commutating capability is temperature. All commutating characteristic data a re specified for maximum operating case temperature a t maximum rated steady-state current. If the operating case temperature is below the rated value, the commutating capability is increased.

I I

500 W 3

LZ I I

70 100 JUNCTION TEMPERATURE-*C

F$. 109-Freqrrertcy capability o f a 400-Hz rrinc as a jtrrictiott o/ jrrrtctiorr

Ierrlperatrtre.

RADIO-FREQU ENCY INTERFERENCE

The fast switching action of triacs when they turn on into resistive loads causes the current to rise to the instantaneous value determined by the load in a very short period of time. Triacs switch from the

high- to the low-impedance s tate within 1 or 2 microseconds; the current must rise from essentially zero to full-load value during this period. This f a s t switching action produces a current step which is largely composed of higher-harmonic frequencies of several megahertz tha t have an -. amplitude varying inversely a s the frequency. In phase-control applications, such a s light dimming, this current step is produced on each half-cycle of the input voltage. Be- cause the switching occurs many times a second, a noise pulse is generated into frequency-sensitive devices such a s AM radios and causes annoying interference. The amplitude of the higher frequencies in the current step is of such low levels that they do not interfere with television o r F M radio. In general, the level of radio-frequency interference (RFI) produced by the triac i s well below tha t produced by most acldc brush-type electric motors; however, some type of R F I suppression network is usually added.

There a r e two basic types of radio-frequency interference (RFI) associated with the switching action of triacs. One form, radiated RFI, consists of the high-frequency energy radiated through the air from the equipment. In most cases, this radiated RFI is insignificant unless the radio is located very close to the source of the radiation.

Of more significance is conducted RFI which i s carried through the power lines and affects equipment attached to the same power lines. Because the composition of the current waveshape consists of higher frequencies, a simple choke placed in series with the load increases the current rise time and reduces the amplitude of the higher harmonics. To be effective, however, such a choke must be quite large. A more effective filter, and one tha t has been found adequate fo r most light-dimming applications, is shown in Fig. 110. The LC filter provides adequate attenuation of the high- frequency harmonics and reduces


Fig. 111 sho\rs a triac control circuit that includes RFI suppression for the purpose of minimizing high-

120 VAC OR frequcncy interference. The values

indicated a re typical of those used 60 Hz

120 VAC OR

240 VAC 60 Hz Q + 1 O R I

CONTROL CIRCUIT

Fig. 110-RFI-.rrrppressio,t rrc'l~.o~ k.r (C = 0.1 PF, 200 V at I20 V ac; 0.1 PF, 400 V

at 240 V ac).

the noise interference to a low lcvcl. The capacitor connected across the cntirc nctworlt bypasscs high-frequency signals so tha t thcy are not connected to any external circuits Fig. 111-Lni~tp-co~trrol circctir i~rcorpotnf- through the power lines. iris RFI srrppressior~.

Silicon Rectifiers

S ILICON rectifiers a re essentially , cells containing a simple p-n ' jdnction. As a result, they have low

resistance to current flow in one (forward) direction, but high resistance to current flow in the opposite (reverse) direction. They can be operated a t ambient temperatures up to 200°C and a t current levels a s high a s hundreds of amperes, with voltage levels greater than 1000 volts. In addition, they can be used in parallel or series arrangements t o provide higher current or voltage capabilities.

Because of their high forward-to- reverse current ratios, silicon rectifiers can achieve rectification efficien- cies greater than 99 per cent. When properly used, they have excellent lifc characteristics which a re not affected by aging, moisture, or temperature. They a re very small and light-weight, and can be made im- ! pervious to shock and other severe

' environmental conditions.

b I

THERMAL CONSIDERATIONS

1 Although rectifiers can operate a t h i ~ h temperatures, the thermal ca- pacity of a silicon rectifier is quite low, and the junction temperature rises rapidly during high-current operation. Sudden rises in junction temperature caused by either high currents or excessive an~bient-tcm- perature conditions can cause failure. ( A silicon rectifier is considered to have failed when either the forward voltage drop or the reverse current has increased to a point where the crystal structure or surrounding material brealts down.) Consequently,

temperature effects a re very important in the consideration of silicon rectifier characteristics.

REVERSE CHARACTERISTICS When a reverse-bias voltage is ap-

plied to a silicon rectifier, a limited amount of reverse current (usually measured in microamperes, a s compared to milliamperes or amperes of forward current) begins to flow. As shown in Fig. 112, this reverse current flow increases slightly a s the bias voltage increases, but then tends

VOLTAGE

Fig. 112-T)~pical reverse characteris!ics DI a silicon rectifier.

to remain constant even though the voltage continues to increase significantly. However, an increase in operating temperature increases the reverse current considerably for a given reverse bias.

At a specific reverse voltage (which varies fo r different types of diodes), a very sharp increase in reverse current occurs. This voltage is called the breakdown or avalanche (or zener) voltage. In many applications, rectifiers can operate safely a t the avalanche point. If the reverse voltage is increased beyond this point, however, or if the ambient temperature is raised sufficiently (for ex-

anll)lc, a rise fro111 25 to 150°C increases the current by a factor of several hundred), "thermal runaway" rcsults and the diode may 1)e destroy ed.

FORWARD CHARACTERISTICS

A silicon rectifier usually requires n forward voltane of 0.4 to 0.8 volt (depending upon the temperature and the impurity concentration in the p-type and n-type materials) before significant current flow occurs. As shown in Fig. 113, a s l i ~ h t rise in voltage beyond this point increases the forward current sharply. Because of the small mass of the silicon rectifier, the forward voltage tlrop n ~ u s t be carefully controlled so that the specified maximum value of dissipation for the device is not exceeded. Othenvise, the diode may be seriously damaged o r destroyed.

Fig. 113 shows the effects of a n increase in temperature on the forward- current characteristic of a silicon


VOLTS Fig. 113-Tl,p;col jor~clord c\torncterisrics

i11 a si/icorr recrifier.

Silicon Rectifiers

rectifier. In certain applications, close control of ambient temperature is required for satisfactory operation. Close control is not usually requi~ed, however, in power circuits.

RATINGS

Ratings for silicon rectifiers a r e determined by the rna~~ufncturcr o n the basis of extensive reliability testing. One of the most important ratings is the nlaximunl peak reverse voltage (PRV), i.e., the highest a ~ n o u n t of reverse voltage which can be applied to a specific rectifier before the avalanche breakdo\vn point

is reached. PRV ratings range from about 50 volts to a s high a s 1000 volts fo r some single-junction diodes. As will be discussed later, several junction diodcs can be conricctcd in series to obtain the PRV values required for very-high-voltage power- supply applications.

Because the current through a rcc- tifier i s normally not dc, current ratings a r e usually given in terms of average, rms, and peak values. Tine waveshapes showrl in Fig. 114 and 115 help to illustrate the relationships among these r a t i n ~ s . F o r example, Fig. 114 shows the current variation with time of a sine wave.

Fig. I14-Vnrioriorr of crtrrorr of n sitre wove wit11 riflle.

tha t has a peak current I,vnk of 10 amperes. The area under the curve can be translated mathematically into a n equivalent rectangle t h a t indicates the average value I,, of the sine wave. The relationship between the average and peak values of the total sine-wave current is then given by

I,, = 0.637 I p v , r

or I,,,t = 1.57 1%"

However, the power P consunled by a device (and thus thc heat generated within i t ) is equal to the square of the current through it times its finite electrical resistance R (i.e., P = IR). Therefore, the power is proportional to the square of the current rather than t o the peak or average value. Fig. 115 shows the square of thc current for the sine wave of Fig. 114. A horizon- t a l line drawn through a point half- way up the I' curve indicates the average (or mean) of the squares, and the square root of the 1-value

. - Fig. 115-Variatio~z o f the sqrcore of sit~e-

wave clrrrrflr with ti~rre.

a t this point is the root-mean-square (rrns) value of the current. The relationship between r m s and peak current is given by

I,,. = 0.707 Ipe.t - or .

I,..r = 1.414 I.,.,. Because a single rectifier cell

passes current in one direction only, it conducts fo r only half of each cycle of an a c sine wave. Therefore, the second half of the curves in Figs. 114 and 115 is eliminated. The average current I.. then becomes half of the value determined for full-cycle conduction, and the r m s current I.,. is equal to the square root of half the mean-square value f o r full-cycle conduction. I n terms of half-cycle sine-wave conduction (as in a single- phase half-wave circuit), the relationships of the rectifier currents can be shown a s follows:

I,.,t = rr x I., = 3.14 I.. I.. = ( 1 1 ~ ) Ipc.k = 0.32 IrBpat I,,. = ( ~ 1 2 ) I.. = 1.57 I,. I.. = (21s) I,,. = 0.64 I,,,,. I,..L = 2 I,",. I r m a = 0.5 1p.m~

For different con~binations of rectifier cells and different circuit con-

figurations, these relationships are, of course, changed again. Current (and voltage) relationships have been derived f o r various types of rectifier applications and a r e given in the section on DC Power Supplies.

Published data f o r silicon rectifiers usually include maximum ratings fo r both average and peak forward current. As shown in Fig. 116, the maximum average forward current is the maximum average value of current which is allowed t o flow in the forward direction during a full ac cycle a t a specified ambient or case temperature. Typical average current outputs range from 0.5 ampere to a s high a s 100 amperes fo r single silicon diodes. The peak recurrent forward current is the maximum repetitive instantaneous forward current permitted under stated conditions.

SURGE OR FAULT CURRENT

PEAK REPETITIVE CURRENT

- AVERAGE FORWARD CURRENT

Fig. 116-Representarion of rectifier currents.

In addition, ratings a re usually given for non-repetitive surge, o r fault, current. In rectifier applications, conditions may develop which cause momentary currents tha t a r e considerably higher than normal operating current. These increases (current surges) may occur from time to time during normal circuit operation a s a result of normal load variations, o r they may be caused by abnormal conditions or faul ts in the circuit. Although a rectifier can usually absorb a limited amount of additional heat without any effects other than a momentary rise in junction temperature, a sufficiently high '

surge can drive, the junction temperature high enough to destroy the rectifier. Surge ratings indicate t h e amount of current overload or surge that the rectifier can withstand without detrimental effects. Fig. 117 shows universal surge


rating charts f o r families of recti- ~ i v e n circuit can be deternlined by fiers having average current ratings use of a coordination chart such as up to 40 amperes. The rnls currents that shown in Fig. 118. Two charac- shown in these charts are incrcmen- tcristics a re plotted on the coordi- ta l values which ndtl to the normal nation chart initially: ( A ) the surge rms forward current during surge rating curve for the rcctificr, and periotls. The charts indicate maxi- co mum current increnlents tha t can be zoo-] safely handlcd by the rectifiers f o r given lengths of time. These charts can bc used by designers to de- tcrnline whclhcr circuit motlifica- tions a re necessary to protect the rectifiers. If the value and duration of expected currcnt surges a re . .

p e a t e r than the ratings for the rec- $ -'"' SURGE DURATION-SECONDS

tifier, inlpedance should be to Fi.,,. ,,R-TYpicn, coort,~rlnfioll c/l(lr-, lor circuits Or fuses Or ~ ~ c ~ ~ c r f f l i f l ~ f l s /rrsiflg r ~ q ~ t i ~ c ~ ~ l e r r ~ ~ ( A =

circuit breakers to variable-load cir- s,lrsc-rafipll: chart lor 20-arrrpere recfificr; cuits fo r surge protection. U = expected s~tr ,~c crorott irr Irn!j-11@Ore

The fusing requirements fo r a circlrir: c = operrirrg clrnracrcrisrics ol

.- SURGE DURATION - SECONDS

( R ) the maximum surge (fault current) expected in the circuit. In Fig. 118, curve A is the surge rating curve for a 20-ampere rectifier, and curve B is the maximum surge expected to occur in a single-phase Iialf-wave rectifier circuit t h a t has an input voltage of 600 volts and is subject to overload conditions in which the load resistance can decrease to 2 ohms. The maximum rms current which can flow under these conditions is given by

I,,,,. = E1,,/2Rr. = 60014 = 150 anlpcres

The incremental portion of this current is determined by subtracting the normal rms current of thc 20- ampere rcctificr (I ,,,,. = 1.57 1." = 1.57 x 20 = 31.4 amperes; I.,,,,. = 150 - 31.4 = 118.6 nmpcres). The straight line of curve I3 is then drawn a t a n mms value of 118.G am- pcl.es in F ig 118.

The intersection of curves A 2nd 13 indicates that thc 20-aml)crf! rcc- tifier can safely support an Incre- mental rnis surge current of 118.6 anlperes fo r a lnasinlunl duration of allout 40 niilliscconds. Thcrcior~. , the

. - f l ; . / , I / I o r circuit nlust be lllodificd to include RC.4 rcrr i f icrs. a protective element tha t has an

Silicon Rectifiers

"opening" characteristic t h a t falls below the rectifier surge rat ing curve for all times greater than 40 n~illisecontls. The opening characteristic of such a protective element is shown in Fig. 118 a s curve C. Surge current in the modified circuit is then limited by the circuit resistance for periods up to 40 milliseconds and by the protective elenlent f o r surges of longer duration, a s shown by curve D.

Surge currents generally occur when the equipment is first turned on, or when unusual voltage transients are introduced in the a c supply line. Protection against excessive currents of this type can be provided in various ways, a s will be discussed later.

Because these maximum current ratings a re all affected by thermal variations, ambient-temperature conditions must be considered in the application of silicon rectifiers. Tem- perature-rating charts a r e usually provided to show the percentage by which maximum currents must be decreased for operation a t temperatures higher than normal room temperature (25°C). -

OVERLOAD PROTECTION ' In the application of silicon recti-

fiers, i t is necessary to guard against both over-voltage and over-current (surge) conditions. A voltage surge in a rectifier arrangement can be caused by dc switching, reverse recovery transients, transformer switching, inductive-load switching, and various other causes. The effects of such surges can be reduced by the use of a capacitor connected across the input or the output of the rectifier. In addition, the magnitude of the voltage surge can be reduced by changes in the switching elements or the sequence of switching, or by a rcduction in the speed of current in- terruption by the switching elements.

In all applications, a rectifier having a more-than-adequate peak reverse voltage rat ing should be used. The safety margin for reverse volt-

age usually depends on the application. F o r a single-phase half-wave application using switching of the transformer primary and having no transient suppression, a rectifizr having a peak reverse voltage three or four times the expected working voltage should be used. F o r a full- wave bridge using load switching and having adequate suppression of transients, a margin of 1.5 to 1 i s generally acceptable.

Because of the small size of the silicon rectifier, excessive surge currents a r e particularly harmful to rectifier operation. Current surges may be caused by short circuits, capacitor inrush, dc overload, o r failure of a single cell in a multiple arrangement. I n the case of low-power cells, fuses o r circuit breakers a r e often placed in the a c input circuit to the rectifier to interrupt the faul t current before i t damages the rectifier. When circuit requirements a r e such tha t service mus t be continued in case of failure of a n individual diode, a number of cells can be used in parallel, each with i t s own fuse. Ad- ditional fuses should be used in the ac line and in series with the load for protection against dc load faults. In high-power cells, an arrangement of circuit breakers, fuses, and series resistances is often used to reduce the amplitude of the surge current. Fus- ing requirements can be determined by use of coordination charts f o r the particular circuits and rectifiers used.

SERIES AND PARALLEL ARRANGEMENTS

Silicon rectifiers can be arranged in series o r in parallel to provide higher voltage o r current capabilities, respectively, a s required for specific applications.

A parallel arrangement of rectifiers can be used when the maximum average forward current required i s larger than the maximum current rat ing of a n individual rectifier cell. In such arrangements, however, some means must be provided to assure proper division of current


through the parallel rectifier cells. Parallel rectifier arrangements a re not in general use. Designers normally use a polyphase arrangement to provide higher currents, or simply substitute the readily available higher-current rectifier types.

Series arrangements of silicon rectifiers a r e used when the applied reverse voltage is expected to be greater than the maximum peak reverse voltage rating of a single silicon rectifier (o r cell). F o r example, four rectifiers having a maximum reverse voltage rat ing of 200 volts each could be connected in series to handle a n applied reverse voltage of 800 volts.

In a series arrangement, the most important consideration is tha t the applied voltage be divided equally across the individual rectifiers. I f the instantaneous voltage is not uni- formly divided, one of the rectifiers may be subjected to a voltage greater than its specified maximum reverse voltage, and, a s a result, may be destroyed. Uniform voltage division can usually be assured by connection of either resistors or capacitors in parallel with individual cells. Shunt

resistors a r e used in steady-state applications, and shunt capacitors in a,pplications in which transient voltages a re expected. Both resistors and capacitors should be used if the circuit is to be exposed to both dc ant1 ac components. When only a few diodes a r e in series, multiple transformer windings may be used, each winding supplying its own assembly consisting of one series diode. The outputs of the diodes a re then connected in series fo r the desired voltage.

RCA rectifier stacks (CR101, CR201, and CR301 series) a re designed to provide equal reverse voltage across the individual rectifier cells in the assembly under both steady-state and transient conditions. The CRlOl and CR301 series stacks include a n integral resistance- ca~ac i tance network to equalize the' re jerse voltage across the series- connected rectifier cells. Thd. CR201 series stacks use precisely matched rectifier cells fo r internal voltage equalization. Extended life tests have shown tha t these rectifier stacks a re capable of operating for many thou- sands of hours without noticeable degradation of performance.

Other Solid-State Diodes

I N addition to the silicon rectifiers, described in the preceding section,

a number of other types of solid- s tate diode devices a r e available f o r use in a broad variety of circuit applications.' ' F o r example, low-level rectifying diodes a r e widely used in signal-mixing, detector, and balanced-modulator applications. Such diodes, although they have significantly lo\ver voltage and current ratings, operate essentially the same a s the silicon rectifiers and a r e not discussed further. The emphasis in this section is on specialized types (i.e., tunnel, varactor, voltage-reference, and compensating diodes) tha t a re used primarily to provide functions other than rectification.

TUNNEL DIODES

A tunnel diode is a slnall p-n junction device having a very high concentration of impurities in the p-type and n-type semiconductor materials. This high impurity density makes the junction depletion region (or space-charge region) so narrow t h a t electrical charges can transfer across the junction by a quantum-mechanical action called "tunneling." This tunneling effect provides a negative-~esistance region on the characteristic curve of the device that makes it possible to achieve amplification, pulse generation, and rf-energy generation.

Characteristics Typical current-voltage character-

istics for a tunnel diode a r e shown in Fig. 119. Conventional diodes do

not conduct current under conditions of reverse bias until the breakdown voltage is reached; under forward bias they .begin to conduct a t approximately 300 millivolts. I n tunnel diodes, however, a small reverse bias

Fig. 119-Tj~pical crtrretrt-volrage charac- rerisric of a tutrnel diode.

causes the valence electrons of semiconductor atoms near the junction t o "tunnel" across the junction from the p-type region into the n-type region; as a result, the tunnel diode is highly conductive f o r all reverse biases. Similarly, under conditions of small forward bias, the electrons in the n-type region ''tunnel" across the junction to the p-type region and the tunnel-diode current rises rapidly to a sharp maximum peak I,. A t intermediate values of forward bias, the tunnel diode exhibits a negative- resistance characteristic and the current drops to a deep minimum valley point I,.. A t higher values of forward bias, the tunnel diode exhibits the diode characteristic associated with conventional semiconductor current flow. The decreasing current with increasing forward bias in the nega-

RCA Transistor, Thyristor, & Diode Manual I Other Solid-State Diodes

tive-resistance region of t h e characteristic provides the tunnel diode with i ts ability t o amplify, oscillate, and switch.

Equivalent Circuit In the equivalent circuit for a tun-

nel diode shown in Fig. 120, the n- type and p-type regions a r e show~l a s . -

I TRANSITION . " ! REGION I

L w r r v

C(V)

Fix. 120-Eqrrir~nletr~ circttic for a rrrrzrrel diode.

Dure resistances r, and r,. The tran-

Fig. 121-Eqviraletrr circrril for a lrr~trrel diode biased irr llte ttc~alive-resistalrce

regiorr.

(1) it is the diode gain-bandwidth product fo r circuits operating in the linear negative-resistance region of the characteristic, and (2) i ts reciprocal is the diode switching time when the device is used a s a logic element.

Operating Point sition region is represented a s a when the tunnel diode is used in voltage-sensitive resistance R ( v ) in such as amplifiers and oscil- parallel with a voltage-sensitive ca- lators, the operating point be.' pacitance C(V) because tunneling is established in the negative-resistance a function of both voltage and June- region. ~h~ dc load line, shown a s tion capacitance. This capacitance is , solid line in ~ i ~ . 122, must be very similar t o tha t of a paral le l-~late steep so tha t i t intersects the static capacitor having plates separated by characteristic curve a t only one point the transition region. A. The ac load line can be either

The dashed portion L in Fig. 120 steep with only one in te rsec t io~~ B, represents a n inductance which re- as in the case of an amplifier, or sults from the case and mounting of relatively flat with three intersee- the tunnel diode. This inductance IS tions C, D, and E, as ill the case of u n i ~ ~ l p o r t a n t fo r low-frequency di- an oscillator. l-he location of the op- odes, but becomes i~~creas ing ly im- erating point is determined by the portant a t high frequencies (above anticipated s imal swing, the 100 MHz). signal-to-noise ratio, and the operat-

Fig. 121 shows the f0rnl of the iI,g tenlperature of the device. Bias- equivalent circuit when the diode is ing at the center of the linear portion biased so that its operating point is in the negative-resistance region; dynamic characteristics of tunnel diodes a re defined with respect to this circuit. Lq represents the total series -%.

inductance, and Ra the total scrics resistance. CD is the capacitance and -R,, is the negative resistance o f ' the diode. For small signal variations, both the resisbncc RD and the capacitance C,, arc constant.

The ficrure of merit F of a tunnel -

cliorle is equal to the reciprocal of 2rRC. where R and C a r e the equiva- 'I - DC LOAD LINE --- AC L O A D LINE

lent ia lues -RD and Ca, rcspcctirely, , s l ~ o ~ v n in Fig. 121. This expression F ~ X . ~22-Ty,)icctl lour1 l i ~ r r . ~ !or ~ttr~ttrl-

a Ions: has two very useful interpret t ' diode rircrrifs.

of the nenative-resistance slope permits the greatest signal swing. For high-temperature operation, a higher operating current is chosen; f o r low noise, the device is operated a t the lowest possible bias current.

Radiation and Thermal Considerations

One of the most important features of the tunnel diode is its resistance to nuclear radiation. Experimental results have shown tunnel diodes t o be a t least ten times more resistant to radiation than transistors. Because the resistivity of tunnel diodes is so low initially, i t is not critically affected by radiation until large doses have been applied. In addition, tunnel diodes a re less affected by ioniz- ing radiation because they a r e relatively insensitive to surface changes produced by such radiation.

I n general, the tunnel-diode voltage-current characteristic is relatively independent of tenlperature. Specific tunnel-diode applications ]nay be affected, however, by the relative temperature dependence of the various circuit components. I n such applications, negative feedback or direct (circuit) compensation may be required.

TUNNEL RECTIFIERS In addition to its neaative-resist-

ance properties, the tunnel diode has an efficient rectification characteristic which call be used in many rectifier applications. When a tunnel diode is used in a circuit in such a way that this rectification property is emphasized rather than i ts negative-resistance characteristic, i t is called a tunnel rectifier. In general, the peak current for a tunnel rectifier is less than one milliampere.

The major differences in the current-voltage characteristics of tunnel rectifiers and conventional rectifiers are shown in Fig. 123. In conventional rectifiers, current flow is substantial in the forward direction, but catreniely small in the reverse direction (for signal voltages less than the breakdown voltage for the de-

vice). In tunnel rectifiers, however, substantial reverse current flows at very low voltages, while forward current is relatively small. Conse- quently, tunnel rectifiers can provide rectification a t smaller signal voltages than conventional rectifiers, although their polarity requirements a r e opposite. (For this reason, tunnel rectifiers a re sometimes called $ 7 "back diodes.") ,

'I I 'i I

I I - CONVENTIONAL 1 RECTIFIER 1 --- TUNNEL

I RECTIFIER

I I

1 J l Fig. 123-C~rrretrr-voltage characrerisrics

of frrrurel rectifier or~d conver~tional rectifier.

Because of their high-speed capability and superior rectification characteristics, tunnel rectifiers can be used to provide coupling in one direction and isolation in the opposite direction. Fig. 124 shows the use of tunnel rectifiers to provide directional coupling in a tunnel-diode logic circuit.

- Fig: 124-Logic circrrir trsitzg a rutzrrel

diode and ~Izrec rrttrnel recrifiers.

RCA Transistor, Thyristor, & Diode Manual Other Solid-State Diodes 87

VARACTOR DIODES A varactor or variable-reactance

diode is a n~icro\~ave-freque~lcy 1)-n junction solid-state device in which the depletion-layer capacitance 1,ears a nonlinear relation to the junction voltage, as sho\vn in Fig. 125(a). When biased in the reverse direction, a varactor diode can be rcpreserlted by a voltage-sensitive capacitance C(v) in series with a resistance R., a s shown in Fig. 125(b). This nonlinear capacitance and low series resistance, which permit the device to perform frequency-nlulti- plication, oscillation, and s\vitcliing functions, result from a very high impurity concentration outside the depletion-layer region and a rela- p3

VOLTAGE

Fi.?. 125-(n) C ~ p ~ r c . i ~ r r t ~ c e - ~ ~ o l t r ~ ~ ~ c ~ rrl(rtiorr- n t~d (0) egrri\,aletrt cir-crrit Jor n 1,rrrnclor

diode.

tively low concentration a t the junction. Very low noise levels are possible in circuits using varactor diodes because the dominant current across the junction is reactive and shot-noise components are absent.

Reactive nonlinearity, without an appreciable series r e s i s t a ~ ~ c e con~ponent, enables varactor diodes to generate harmonics with very high efficiency in circuits such a s the shunt- type frequency multiplier shown in Fig. 126. The circuit is driven by a- sinusoitlal voltage source V, having a fundamental frequency f and an internal impedance Z.. Bccausr the ideal input filter is an open circuit for all frcqucncics except thc fundamental frequency, only the fundamental con~ponent of current i r can flow in the input loop. A second- harmonic current i,r is generated by

FILTER

the varactor diode and flo\vs toward the load Z,.; another ideal filter is used in the output loop to bloclc the fundamental-frequency component of the input current.

Varactor diodes can amplify signals when their voltage-dependent capacitance is modulated by an alter; nating voltage a t a different f requency. This alternating,.- voltage supply, which is often referred to as the "pump", adds energy to the signal by changing the diotle capacitance in a specific phase relation with the stored signal charge so tha t potential energy is added to this charge. An "idler" circuit is gcnerally uscd to provide the proper phase relationship between the signal and the "pu111p."

VOLTAGE-REFERENCE DIODES Voltage-reference or zencr diodes

a re silicon rectifiers in which the reverse current remains small until the breakdown voltage is reached and then increases rapidly with little further increase in voltage. The brealtdown voltage is a function of the diode material and construction, and can be varied from one volt to several hundred volts for various current and power ratinas, depending on the junction area ant1 the method of cooling. A stabilized supply can deliver a constant output (voltage or current) unaffected by temperature, output load, or input voltage, within given limits. The stability provided by voltage-reference diodes makes

then1 useful a s stabilizing devices and a s reference sources capable of supplying extremely constant current loads.

COMPENSATING DIODES Excellent stabilization of collector

current for variations in both supply voltage and tenlperature can be obtained by the use of a colnpensating diode operating in the forward direction in the bias networlc of amplifier or oscillator circuits. Fig. 127 shows the transfer characteristics of a transistor; Fig. 128 shows the forward characteristics of a compensating diode. In a typical circuit, the diode is biased in the forward direction; the operating point is represented on the diode characteristics by the dashed horizontal line. The dioge current a t this point deter-

--. BASE-TO-EMITTER VOLTAGE-rnV

F i r . 127-1 rrorsler charoc~erisrics of trcrn- sistor.

mines a bias voltage which establishes the transistor idling current. This bias voltage shifts with varying temperature in the same direction and magnitude a s the transistor characteristic, and thus provides a n idling current t h a t is essentially independ- en t of temperature.

FORWARD DIODE VOLTAGE- mV

Fig. 128-Forward characteristics of cortl- perlsatir~g diode.

The use of a compensating diode also reduces the variation in transistor idling current a s a result of supply-voltage variations. Because the diode current changes in proportion with the supply voltage, the bias voltage to the transistor changes in the same proportion and idling-current changes a r e minimized. (The use of diode compensation is discussed in more detail under "Biasing" in the section on Ripolar Transistors.

Receiver Tuner-Circuit

W H E N speech, music, or video information is transmitted from

a radio or television station, the station radiates a modulated radio- frequency ( r f ) carrier. The function of a radio or television receiver IS

simply to reproduce the modulating wave from the modulated carrier.

,as. 6h"cri:wr i6'Pir: T B g L SLIIWT-

heterodyne radio receiver picks up the transmitted modula t~d rf s i ~ n a l , amplifies i t and converts i t to a niodulated inter~nediate-frequency ( i f ) signal, amplifies the niodulated if signal, separates the modulating signal from the basic carrier wave, and amplifies the resulting audio signal to a level sufficient t o produce the dcsirctl volume in a spenkcr. In addition, the iecciver usually includes solnc means of producing automatic gain control (agc) of the modulated signal before the audio information is separated from the carrier.

The transmitted rf signal picked

up by the radio receiver may contain either amplitude nlodulation (Ah[) or frequency modulation (FM).

J 3 ~ t a t i 0 ~ In either case, an>plifica+ tion prior to the detector stage is performed by tuned amplifier circuits designed for the proper frequency and bandwidth. Frequency conversion i s perfomled by mixer and oscillator circuits or by a single converter stage which performs both miser and oscillator functions. Separation of the nlodulatinrz signal is nornlally accomplished by one or more diodes in a detector o r discriminator circuit. Amplification of the audio signal is then perfor~ned by one or more audio am lifier stages. (A%tlio 'arnpfi- + 5 ~ & & l i 6 ~ u ~ a 7

&*qgcncg, ~fldi'cafitlnJ" \ The operation of a television re- &/ceiver & h 0 1 i ? i @&t&~Q*.?m is more complex

DETECTOR

A I

AUDIO AMPLIFIER

-

Fig. 129-Si111p1ific.d block dingra~r~ for a Broadrosr-bond receiver.

Receiver Tuner-Circuit Applications

AND DISCRIMINATOR AMPLIFIER SPEAKER

VIDEO AMPLIFIER PICTURE

DEFLECTlON CIRCUITS CIRCUITS

Fig. 130-Sitr1plified block diagranz for a lekl~isio~l receiver.

than that of a radjo receiver,+ ~ ~ ~ , c ~ 1 @ d i d ~ . 9 ~ ~ &13f):.. - The tuner section of the television receiver selects the proper rf signals f o r the desired channel frequency, amplifies them, and converts them to a lower intermediate frequency. As in a radio receiver, these functions are accomplished in rf-amplifier, mixer, and local- oscillator stages. The if signal is then amplified in if-amplifier stages which provide the additional gain required to bring the signal level to an amplitude suitable fo r detection.

After if amplification, the detected signal is separated into sound and picture information. The sound signal is an~plilicd and processed to provide a n audio signal which is fed t o an audio anlplifier system. The picture (video) signal is passed through a video n~nplifier (&-&he. Mifl-:.FmqBency : i$nq~li; &Q, ~vhich conveys beam-intensity information to the television picture tube and thus controls instantaneous "spot" brightness. A t the same time, deflection circuits cause the electron bcam of the picture tube to move the "spot" across the faceplate horizon- tally and vertically. Special "sync"

signals derived from the video signal assure t h a t the horizontal and vertical scanning a re timed so t h a t the picture produced on the receiver exactly duplicates the picture being viewed by the camera or pickup t u b d tThe s y h c - a d .8eflection circuits a r e -

a s c r i b e d inpthe sectioh on T V h - ~ Aection:).

In a television receiver, the video signal contains a dc component, and therefore the average carrier level varies with signal information. A s a result, the agc circuit is designed to provide a control voltage proportional to the peak modulated carrier level rather than the average modulated carrier leveI. The time constant of the agc detector circuit is made large enough so that the picture content of the conlposite video signal does not influence the magnitude of the agc voltage. In addition, a n electronic switch i s often included in t h e circuit so tha t i t can be operated only during the retrace portion of the scanning cycle. This "gated agc" technique prevents noise peaks from affecting agc operation.

DETECTION

The circuit of a radio, television, or communications receiver in which the

RCA Transistor, Thyristor, & Diode Manual I Receiver Tuner-Circuit Applications

modulation is separated from the carrier is called the demotlulator or detector stage. Trans~uit tcd rf signals nlay be modulated in cithcr of two ways. If the frequency of the carrier remains constant and its amplitude is varied, the carrier is called a n amplitude-modulated (AM) signal. If the amplitude remains essentially constant and the frequency is varied, the carrier is called a frequency-modulated (FFI) signal.

The effect of amplitude modulation (12i11) on an 1.f carrier wave is shown in Fig. 131. The audio-

UNMODULATED HF CARRIER

w AMPLITUDE-MODULATED RF WAVE

A f MODULATING WAVE

Fir. 13l--ll'n1~c/or111.r slrowi~tg t~ljcr'l o f arrr~rlirtttle ~ ~ ~ o d ~ t l a r i o ~ r 0 1 1 art r f wave.

frequency ( a f ) nlodulation can be extracted from the an~plitude-lnodu- latetl carrier by means of a siniple diode detector such a s that shown in Fig. 132(a). This circuit eliminates alternate half-cycles of the waveform, and dctects the pcaks of the remaining half-cycles to produce t h e output voltage shown in Fig. 132(11). In this figure, the rf voltage applied to the circuit is shown in light line; the output voltape across the caoacitor C is shown in heavy line.

Retween points a and I) of Fig. 132(b), capacitor C charges up to the peak value of the rI voltage. Then, a s the appIied ~f voltage falls away from i ts peak value, the capacitor holds the cathode of the diode a t a potential Inore positive than the voltage applied to the anode. The capacitor thus temporarily cuts off current through the diode. While the

; -+- ) , ,y~~"T INPUT

AMPLITUDE-MODULATED RF WAVE

Ib)

Fig. 132-(11) Uasic tliotlc tfclrr~cror. circaif nrrtl (h) bcct~~clor~rr slron~ir~r: ~lrotlrrlnreri rl irrpttl (lip111 li~re) nrl(l ortrprtt vo/ra.ee

(heavy /;tie) of diode-di>rcoor circ~tit.

diode current is cut off, the capacitor discharges from b to c through the diode load resistor R.

When the rf voltage on the anode rises high enough to exceed the potential a t which the capacitor holds the cathode, current flows again, and the capacitor charges up to the peak value of the second positive half- cycle a t d. In this way, the voltape across the capacitor follows the peak value of the applied rf voltage and reproduces the af modulating signal. The jaggedness of the curve in Fig. 132(b) , which represents an rf component in the voltage across the capacitor, is exaggcrated in the drawing. In an actual clrcuit, the rf component of the voltage across the capacitor is small. When the voltage across the capacitor is amplified, the output of the a~nplifier reproduces the speech or music that originated a t the transmitting station.

Another way to describe the action of a diode detector is to consider the circuit a s a half-wave rectifier. When the signal on the anode swings positive, the diode conducts and the rectified current flows. The dc voltage

across the capacitor C varies in ac- cordance with the rectified amplitude of the carrier and thus reproduces the af signal. Capacitor C should be large enough to smooth out rf o r if variations, but should not be so large a s to affect the audio variations. (Although two diodes can be connected in a circuit similar to a full-wave rectifier to produce full-wave detection, in practice the advantages of this connection generally do not justify the extra circuit cost and complication.)

In the circuit shown in Fig. 132(a), i t is often desirable t o forward-bias the diode almost to the point of conduction to improve performance f o r weak signal levels. I t is also desirable tha t the resistance of the ac load which follows the detector be considerably larger than the diode load resistor to avoid severe distortion of the audio waveform a t high modulation levels.

The basic diode detector may also be adapted to provide video-signal detection in black-and-white and color television receivers. Fig. 133 shows a n example of a diode type of video detector for a color television receiver.

The video detector demodulates the if signal so that the luminance, chrominance, and sync signals a r e available a t the output of the detector circuit. A crystal diode with a n

SOUND TAKE-OFF

T CARRIER

TO VIDEO

AMPLIFIER

Fig. 133-Vitieo ticrector for (1 color rrle- r v i s i o ~ ~ rccciver. I I

if filter is commonly used f o r this purpose. The video detector in a color receiver may employ a sound- carrier t rap in i ts input. This t rap attenuates the sound carrier and insures against the development of a n undesirable 920-kHz beat frequency which is the frequency difference between the sound carrier and the color subcarrier. When the sound carrier is attenuated in this manner, t h e sound take-off point is located ahead of the video detector.

The effect of frequency modulation (FM) on the waveform of a n rf carrier wave is shown in Fig. 134. I n

U N M O D U L A T E D R F CARRIER

F R E Q U E N C Y- M O D U L A T E D RF WAVE

F 6 . 134-Wnveforrrls sl~owitag effect of jreqltericy ~ttodrrlafiort on a ~ z rf wave.

this type of transmission, the frequency of the d carrier deviates from the mean value a t a rate proportional to the audio-frequency modulation and by an amount (determined in the transmitter) proportional to the amplitude of the af modulating signal. That is, the number of times the carrier frequency deviates above and below the center frequency i s a measure of the frequency of the modulating signal; the amount of frequency deviation from the center frequency i s a measure of the loudness (amplitude) of the modulating signal. For

RCA Transistor, Thyristor, & Diode Manual I Receiver Tuner-Circuit Applications

Bt r ig . 13S-Bnlnrlced pltase-sltift discrirrri~raror circtcit.

this type of modulation, a detector is required to discriminate between deviations above and below the center frequency and to translate these deviations into a voltage having a n a m ~ l i t u d e tha t varies at audio frequencies.

The FM dctector shown in Fie;. 135 is called a balanced phase-shift discriminator. In this detector, the mu- tually coupled tuned circuits in the primary and secondary windings of the transformer T are tuned to the center frequency. A characteristic of a double-tuned transformer is tha t the voltages in the primary and secondary windings a re 90 degrees out of phase a t resonance, and tha t the phase shift changes as the frequency changes from resonancc. Therefore, the signal applied t o the diodes and the RC con~binations fo r peak detection also changes with freqnency.

Because the secondary winding of the transformer T is center-tapped, the applied primary voltage E, is added to one-half the secondary voltage E:, through the cap:icitor C,. The addition of these voltages a t rcso- nancc can be representcd by the diagranl in Fig. 136; the resultant volt-

age El is the signal applied to one peak-detector network consisting of one diode and i t s RC load. When the signal freqlrency decreases (from resonance), the phase shift of E.12 becon~es greater than 90 degrees, a s shown a t ( a ) in Fig. .137, and El becomes smaller. When the signal frequency increases (above resonance), t h e phase shif t of E.12 is less than 90 degrees, a s shown at (b), and El becomes larger. The curve

EP EP (a) (b)

Fig. 137-Dinar-nrrr.r illrulrrctirrg pirose slJ;/f irr rio~rble-t~rrrcti tr(rtr~forrtrcr. fa1 h~~lo lv rcj-

orratrce ottd ( h ) abovc rrsotrartce.

of El a s a function of frequency in Fig. 138 is readily identified a s the response curve of an F M dctector.

Because the discriminator circuit shown in Fig. 135 uses a push-pdl configuration, the diotles conduct on alternate half-cycles of the signnl frequency and produce a plus-and- minus output with respect to zero rather than with respect to El. The primary advantage of this arrangement is tha t there is no output at resonance. JVhen a n F h l signal is applied to tlie input, the audio out- nu t voltare varies above and belolv 1 - - - .

i s . I - ; , , I ~ ~ I I ~ I . ~ ~ ~ ; I , i~ i f zero a s tlic instantaneolls frequcnc~ ir~ dorrble-trtrrell rrnrr.rjor.rrrcr at rcsorrorrcr. varies above and below resonance.

The frequency of this audio voltage is determined by the modulation frequency of the FM signal, and the amplitude of the voltage is proportional to tlie frequency excursion from resonance. (The resistor R2 in the circuit provides a dc return f o r the diodes, and also maintains a load impedance across the primary winding of the transformer.)

Fig. 138-Dicrgrn~rr showirrg r.esrtltar~t volt- njie E, it1 Fig. 136 os a jrrtrc/iorr of ire-

qltolcy.

One disadvantage of the balanced phase-shift discriminator shown i n Fig. 135 is that i t detects amplitude modulation ( A M ) a s well a s frequency modulation ( F M ) in the if signal because the circuit is balanced only a t the center frequency. At frequencies off resonance, any variation in amplitude of the if signal is reproduced to some extent in the audio output.

The ratio-detector circuit shown in Fig. 139 is a discriminator circuit which has the advantage of being relatively insensitive t o amplitude variations in the F M signal:In this circuit, E, i s added to E,/2 through

the mutual coupling M, (this voltage addition may be made by either mutual o r capacitive coupling). Be- cause of the phase-shift relationship of these voltages, the resultant detected signals vary with frequency variations in the same manner a s described for the phase-discriminator circuit shown in Fig. 135. How- ever, the diodes in the ratio detector a r e placed "back-to-back" (in series, ra ther than in push-pull) so that both halves of the circuit oper- a t e simultaneously during one-half of the signal frequency cycle (and a r e cut off on the other half-cycle). A s a result, the detected voltages El and Ez a r e in series, a s shown f o r the instantaneous polarities t h a t occur during the conduction half-cycle. When the audio output is taken between the equal capacitors CI and C2, therefore, the output voltage is equal to (Ez-B)/2 (for eaual resistors Rx and RI).

The dc circuit of the ratio detector consists of a path through the secondary winding of the transformer, both diodes (which a r e in series), and resistors RI and R2. The value of the electrolytic capacitor C3 is selected so that the time constant of R1, Ri, and CJ is very long compared t o the detected audio signal. A s a result, the sum of the detected voltages (El + E,) is a constant, and the AM components on the signal frequency a r e suppressed. This feature of the ratio detector provides improved AM rejection a s compared t o the phase- shift discriminator circuit shown in Fig. 135.

rr Fig. 139-Rutio-derecror circltir.


TUNED AMPLIFIERS

In radio-frequency (rf) and intermediate-frequency (if) anrpli- fiers, the bandwidth of frequencies to be amplified is usually only a small percentage of the center frequency. Tuned amplifiers a r e used in these applications to select the desired bandwidth of frequencies and to suppress unwanted frequencies. The selectivity of the amplifier is obtained by means of tuned inter- s tage coupling networks.

Resonant-Circuit Characteristics

The properties of tuned amplifiers depend upon the characteristics of resonant circuits. A simple pariillel resonant circuit (sometimes called a "tank" because it stores energy) is shown in Fig. 140. F o r practical pur- p o s ~ s , the resonant frequency of such a circuit may be consitlered inde- jlendent of thc resistance R, providcd R is small compared t o the inductive reactance XI.. The resonant frequency f, is then given by

For any given resonant frequency. the product of L and C is a constant; a t low frequencies LC is large; a t high frequencies it is small.

The Q (selectivity) of a parallel resonant circuit alone is the ratio of the current in the tank (11, or Ic) to the current in the line (I). This un-

Fig. 140-Si~~rplc prrrallel reso~rarrl circ~tif.

loaded Q, or Q,, may be expressed in various ways, for example:

Ic Xr. Rp ---- - Qo =r - R~ xc

where XI. is the inductive reactance ( = 2r fL) , X, is the capacitive reactance ( = 1/[2nfC]), and R, is the total impedance of the parallel resonant circuit (tank) a t resonance. The Q varies inversely with the resistance of the inductor Rs. The lower the resistance, the higher the Q and the greater the difference between the tank impedance a t frequencies off resonance compared t o the tank impedance a t the resonant frequency.

The Q of a tuned interstage coupling network also depends upon the impedances of the preceding and following stages. The output impedance of a transistor can be considered as consisting of a resistallcc R. in parallel with a capacitance C,, a s shown in Fig. 141. Similarly, the.input impedance can be considered a s consisting of a resistance R I in parallel with a capacitance Ct. Because +.he

OVTPUT OF INPUT OF PRECEDING COUPLING FOLLOWING

TRbNSISTOR NETWORK TRANSISTOR

2 4

Fig. 141-Eq11i~~nl~~111 ot~tprtf (111d i~rprrf cir- crrirs of Ira~rsislors corr~~rcred 6)' n cou-

pling rterwork.

tuned circuit is shunted by both the output impedance of the preceding transistor and the input impedance of the following transistor, the effective selectivity of the circuit is the loaded Q (or QI,) based upon the total impedance of the coupled network, a s follows:

total loadinp; on 1 lcoil a t resonance J

QL = XL o r XC

The capacitances C,, and CI in Fig. 141 are usually considered a s par t of

Receiver Tuner-Circuit Applications 95

the coupling network. For example, if the required capacitance between terminals 1 and 2 of the coupling network is calculated to be 500 picofarads and the value of C, is 1 0 picofarads, a capacitor of 490 picofarads is used between terminals 1 and 2 so that the total capacitance is 500 picofarads. The same method is used to allow for the capacitance CI a t terminals 3 and 4.

When a tuned resonant circuit in the primary winding of a transformer is coupled t o the nonresonant secondary winding of the transformer, a s shown in Fig. 142(a), the effect of the input impedance of the following stage on the Q of the tuned circuit can be determined by considering the values reflected (or referred) to the primary circuit by transformer action. The reflected resistance rl is equal to the resistance RI in the secondary circuit times the square of the effective turns ratio between the primary and secondary windings of the transformer T:

where NI/Nz represents the electrical turns ratio between the primary winding and the secondary winding of T. If there is capacitance in the secondary circuit (C.), i t is reflected to the primary circuit as a capacitance C.,,, and is given by

The loaded Q, o r Ql., is then calculated on the basis of the inductance L,., the total shunt resistance (R. plus r~ plus the tuned-circuit impedance Zt = Q,Xc = Q..XL), and the total capacitance (C, + C.,) in the tuned circuit.

Fig. 142(b) shows a coupling network which consists of a single- tuned circuit using mutual inductive coupling. The capacitance C, includes the effects of both the output capacitance of the preceding transistor and the input capacitance of the following transistor (referred

to the primary of transformer TI). The bandwidth of a single-tuned transformer i s determined by the half-power points on the resonance curve (-3 dB o r 0.707 down from -- -

(a

PRECEDING FOLLOWING TRANSISTOR

(b)

FOLLOWING TRANSISTOR TRbNSISTOR

(C) Fig. 142-Eqrrivalent circitits for rrans- fornier-colrplirtg networks: (a) liaving tuned pri~tlary winding; (b) usi~zg inductive cou- plirrg; (c) using tap or1 prinrary winding.

the maximum). Under these conditions, the band pass a f is equal to the ratio of the center o r resonant frequency f , divided by the loaded (effective) Q of the circuit, a s follows:

Af = f J Q , The inherent internal feedback in

transistors can cause instability and oscillation a s the gain of a n amplifier stage is increased (i.e., a s the load and source impedances a r e increased from zero t o matched conditions). A t low radio frequencies, therefore, where the potential gain of transistors is high, i t is often desirable t o keep the transistor load impedance low. Relatively high capacitance values in the tuned collector circuit can then be avoided by use of a t a p on the primary winding of the coupling transformer, a s shown i n Fig. 142(c). A t higher frequencies, the gain potential of the transistor decreases, and impedance matching is permissible. However, lead inductance becomes significant a t higher

RCA Transistor, Thyristor, & Diode Manual 1 I

frequencies, particularly in the emit- capacitance, and the typical value of / t e r circuit. All lead lengths should be device capacitance can generally be kcpt short, therefore, and especially neutralized. A t a given frequency, the emitter lead, which not only de- therefore, the niaximunl usable grades performance but is also a power gain M U G of a neutralized cir- mutual coupling to the output circuit. cuit depends on the transconductance i

g,,, and the amount of internal feed- Gain and Noise Figure back ca~ac i tance CI. In unneutralized

In the design of low-lcvel tuned rf amplifiers, careful consideration must be given to the transistor and circuit parameters which control circuit stability, a s well a s those which maintain adequate power gain. The power gain of an rf transistor must be sufficient to provide a signal t h a t will overcome the noise level of succeeding stages. In addition, if the signals to be amplified a r e relatively weak, i t is important t h a t the transistor and i ts associated circuit provide low noise figure a t the operating frequency. In communication receivers, the noise figure of the rf stage determines the absolute sensitivity of the receiver and is, therefore, one of the most important characteristics of the device used in the rf stage.

The relative power-gain capabilities of transistors a t high frequencies a re indicated by their theoretical maximum frequency of oscillation f ,,,,,. At this frequency, the unilateral- ized matched power gain, or maximum available gain RIA(;, is 0 dB. As shown in Fig. 143, the curve of MAG a s a function of frequency f o r a typical rf transistor rises approximately 6 dB per octave below f,,,,.

Because most practical rf amplifiers are not individually unilateral- ized, the power gain tha t can be obtained is somewhat lcss than the MAG because of internal fecdback in thc circuit. This fecdback is grgater in unncutralized circuits than in neutralized circuits, and therefore gain is lower when neutralization is not used. From a practical consideration, the feedback capacitance which must be considered is the total feedback capacitance bet~vcen collector and base, including 1.wth stray and soclcct capacitances. In neutralized circuits, s t ray capacitances, socket

circuits, however, both socltct and s tray capacitances a rc involved in the determination of gain and must be included in the value of Cr. The ratio of g,,, to Cr should be high to provide high power gain. Fig. 146 shows typical curves of MAG and RlUG (for both the neutralized and the unneutralized case) for a low- level rf transistor used in a common- emitter circuit.

FREQUENCY I

I I I 1 1 1 1 1 1 I I I I I L L I J ~ FREQUENCY --C

Fk. 143-Af~sirrrrrrrr oroilohlc gtri~r M A G , rrraxirrrrtrrl rrsable gnirr M U G , orrd rroise

figrrrc N F os frrrrctiorrs o f jr-eqrrericy.

The transistor requirements for high power gain and low noisc figure are essentially thc same. Published data fo r transistors intcndcd for low- level rf applications generally indicate a minimum power gain and 3

maximum noise f i ~ u r c in a circuit typical of the intended use. A curve

Receiver Tuner-Circuit Applicatio

of noise figure N F a s a function of frequency is also shown in Fig. 143. Circuit design factors fo r lowest noise figure include use of a low- noise transistor, choice of optimum bias current and source resistance, and use of low-loss input circuits. Optinlun~ low-noise bias current f o r most low-level r f transistors is about 1 milliampere, or slightly higher in the uhf range. Optimum source resistance is a function of operating frequency and bias current for a given transistor.

Although maximum theoretical power gain cannot be achived in practical circuits, the gain of MOS transistors at high frequencies closely approximates the theoretical limit except fo r some losses in the input and output matching circuits.

Power gain is essential pendent of channel width, w \ ich is a determining factor in the size of nlOS transistors. For example, if the width of the transistor is reduced by one half (and the steady- state drain current is similarly reduced to maintain a constant current density in the device), power gain remains the same because the transconductance, the input conductance, and the output conductance are all reduced by one half. Con- sequently, the frequency capability of MOS transistors can be increased by a reduction in their size.

Thc input circuit to the first stage of the amplifier should have a s little loss a s possible because such loss adds directly to the otherwise attain- able noise figure. In other words, if the loss a t the input to the first stage is 2 dB, the amplifier noise figure will be 2 dB higher than could be achieved with no loss a t the input. To minimize such loss, i t is generally desirable tha t the ratio of unloaded Q (Q,<) to loaded & (QL) of the input circuit be high and that the bias resistors be isolated from the input by chokes or tuned circuits.

In practical rf-amplifier circuits using MOS transistors, the best possible noise figures a re obtained when the input impedance of the transistor

is slightly mismatched to t h a t of the source. With this technique, noise figures a s low a s 1.9 dB have been obtained. Dual-gate MOS transistors typically exhibit a noise figure of 3.5 dB in the vhf range and of 4.5 dB in the uhf range.

In high-frequency tuned amplifiers, in which the input impedance is typically low, mutual inductive coupling may be impractical because of the small number of turns in the secondary winding. I t is extremely difficult in practice to construct a fractional p a r t of a turn. In such cases, capacitance coupling may be used, a s shown in Fig. 144. This arrangement, which is also called capacitive division, is similar to

TRANSISTOR TRANSISTOR

Fig. 144-Sii~gle-lurrcd corrpling network usirlg capacitivedivisiorl.

tapping down on a coil a t o r near resonance. Impedance transformation in this network is determined by the ratio between capacitors C1 and C,. Capac i to r C , is normal ly much smaller than C,; thus the capacitive reactance XC, is normally much larger than Xn. Provided the input resistance of the following transistor is much greater than XI.,, the effective turns ratio from the top of the coil to the input of the following transistor is (C, + C,)/C,. The total capacitance Ct across the inductance L is given by

The resonant frequency f , is then given by

Double-tuncd interstage coul~ling netu~orlcs a re often used in prcfer- ence to singlc-tunctl nctworl:~ to provitle flat.ter frcqucncg rc.sl)onse within the pass band, a sharper drop in response imnlediately adjacent to the cntls of t l ~ e pass band, o r more attenuation a t frequencies f a r removed from resonance. I n synchronous double-tuned networks, both the resonant circuit in the input of the coupling networlc and the resonant circuit in the output a r e tuned to the same rcsollant frequency. I n "stagger-tuncd" net- worlts, the two resonant circuits a r e tuned to slightly different resonant frequencies to provide a more rectangular band pass with sharper selectivity a t the ends of the pass band. Double-tuned or stagger-tuned networks nlay use capacitive, inductive, or n~utua l inductance coupling, or any combination of the three.


Automatic Gain Control


Automatic gain control (agc) i s oftcn used in rf and if an~plifiers in AM radio and television receivers t o provide lower gain f o r strong s i ~ n a l s and higher gain f o r weak s i p ~ a l s . (In radio receivers, this gain-compensntion network may also be called alltomatic volunie control o r avc.) When the signal strength a t the antenna changes, the agc circuit modi- fies the receiver gain so tha t the output of the last if-anlplifier stage remains nearly constant and consequently maintains R nearly constant speaker volume o r picture contrast.

The agc circuit usullly reduces the rf and if gain for a s t r o n ~ signal by varyinr the bias on the rf-amplifier :tnd if-amplifier staxes wllen the signal increases. A simple rcvcrsc agc circuit is shown in Fig. 145. On each positive half-cycle of the signal volt- a ~ r , when the diode anode is positive with respect to the cathode, the cliode passes current. Recause of the flow of diode current t l~rough R,, there is :I voltage drop across ItI which makes the upper end of the resistor negative with respect to ground. This

voltage drop across R, is applied, through the filter R2 and C, a s reverse

A G C VOLTAGE

DETEcTo;. IR2 AGC DIODE ::.::3 rgO"

IF STAGE

AUDIO OUTPUT

bias on the preceding stages. When the signal strength a t the antenna increases, therefore, the signal ap$!ied to the agc diode increases, the voltage drop across R I increases, the reverse bias applied to the rf and if stages increases, and the r a i n of the rf and if stages is dccreasetl. A s a result, the increase in signal strength a t the antenna does not produce a s much increase in the output of the last if-amplifier stage a s i t mould without agc.

When the sisnal strength a t the antenna decreases from a previous steady value, the agc circuit acts in the opposite direction, applying less reverse bias and thus permitting the rf and if gain to increase.

The filter composed of C and R3 prevents the agc voltage from varying a t an audio frequency. This filter is necessary brcause the voltage drop across R, varies with the nlodulation of the carrier being received. If agc voltnge were talten directly from Rr without filtering, the audio variations in agc voltage ~vould vary the receiver gain so a s to smooth out the modulation of the carrier. To avoid this effect, the agr volt- a r e is taken from the capacitor C. Because of the resistance R, in series with C, the capacitor can charge and discharge a t only a con1para- tively slow rate. The agc voltage therefore cannot vary a t f rcquencies

a s high a s the audio range, but can vary rapidly a t frequencies high enough to compensate fo r most changes in signal strength.

There are two ways in which automatic gain control can be applied to a transistor. I n the reverse agc method shown in Fig. 145, agc action is obtained by decreasing the collector o r emitter current of the transistor, and thus i ts transconductance and gain. The use of forward agc provides improved cross-modulation characteristics and better signal- handling capability than reverse agc. For forward agc operation, however, the transistor used must be specially designed so t h a t transconductance decreases with increasing enlitter current. In such transistors, the current-cutoff characteristics are designed t o be more remote than the typical sharp-cutoff characteristics of conventional transistors. (All transistors can be used with reverse agc, but only specially designed types with forward agc.)

Reverse agc is simpler to use, and provides less bandpass shift and tilt with s i~na l - s t reng th variations. The input and output resistances of a transistor increase when reverse agc is applied, but the input and output capacitances a re not appreciably changed. The change in the loading of tuned circuits is minimal, however, because considerable mismatch already exists and the additional mismatch caused by agc has little effect.

I n forward agc, however, the input and output resistances of the transistor a re reduced when the collector o r emitter current is increased, and thus the tuned circuits a r e damped. In addition, the input and output cal~acitances change drastically, and alter the resonant frequency of the tuned circuits. In a practical circuit, the bantlpass shift and tilt caused by forward agc can be compensated t o a large cxtent by the use of passive coupling circuits.

Cross-Modulation Distortion Cross-niodulation, a n important

consideration in the evaluation of

transistorized tuner circuits, is produced when a n undesired signal within the pass band of the receiver input circuit modulates the carrier of the desired signal. Such distortion occurs when third- and higher- odd-order nonlinearities a re present in a n rf-amplifier stage. In general, the severity of cross-modulation is independent of both the semieonduc- tor material and the construction of the transistor (provided gain and noise factor a r e not sacrificed). A t low frequencies, cross-modulation is also independent of the amplitude of the desired carrier, but varies a s the square of the amplitude of the interfering signal.

To measure cross-modulation distortion, it is necessary to determine the amplitude of the undesired signal which transfers one per cent of its modulation to the desired signal. In most cases, a value of 100 millivolts o r more over the complete agc range is considered good. The cross-modulation characteristics of MOS transistors a re a s good a s those of bipolar transistors in the high-attenuation region, and a r e a s much a s ten times better in the low- attenuation region (when the incoming signal is weak). This low cross-nrodulation distortion should ultimately lead to extensive use of MOS transistors in the rf s tages of all types of communications receivers.

In most r f circuits, the undesirable effects of cross-modulation can be minimized by good selectivity in the antenna and rf interstage coils. Minimum cross-modulation can best be achieved by use of the optimum circuit Q with respect to bandwidth and tracking considerations, which implies minimum loading of the tank circuits.

In rf circuits where selectivity is limited by the low unloaded Q's of the coils being used, improved cross- modulation can be obtained by mis- matching the antenna circuit ( t h a t is, selecting the antenna primary- to-secondary tu rns ratio such t h a t the reflected antenna impedance at

RCA Tr?~nsistor, Thyristor, & Diode Manual

the base of the rf anlplificr is very lo\\, compared to the input impedance). This technique is comlnonly used in nutonlobile rcceivcrs, and causes a slight degradation in noise f i ~ u l e . A t high frequencies, such a s in television, where low source impedances a re dificult to obtain because of lead inductance or the i~npracticality of putting a t ap on a coil having one or two turns, a n unl>ppassct~ emitter resistor having a lorn value of resistance (ex. , 22 ohms) may be used to obtain the same effect.

Cross-modulation may occur in the nlixer or rf amplifier, or both. Accordingly, it is important to ana- lyre tlic cntire tuner a s well a s thc indivitlual stages. Cross-modulation is nlso a function of agc. A t sensitivity conditions where the rf stage is opcratinr a t masilnum gain and the interfering s i ~ n a l is f a r rcnlovetl from the dcsirctl sirnal, cross-modulation occurs primarily in the rf stage. As the desirecl s i ~ n n l level increases and a r c is applied to the rf stagc, the rf transistor gain decleases and provides improved cross-modulation. If the interfering s i ~ n a l is closc to the dcsirctl sirnal, i t is the rf gain a t the und~sirct l signal flequcncy which dctennincs \vhcthcr the rf stage or miser stage is the prime contributor of cross-modulation. For example, i t i s possible t h a t the rf stage gain (including selectivity of tuned circuits) nt the undesired frequency i s greatcr than unity. In this case, the unt1esi1-ed signal a t the mixer input is l a ~ y e r than tha t a t thc rf input; thus the contribution of the rniscr is apprc- cial)l~. Intermediate and hicli signal conditions may bc analyzetl similarly by considering rf agc.

If adequate limiting is clnployed, cross-modulation docs not occur in an FhI signal.

Spurious-response clla~.acicristics are an important considcration in the evaluation of transistorized FRI tuner circuits. Like cross-modulation, spurious response, a n cmcct caused by the mixture of unwanted signals

with the desired carrier, can occur in either the rf s tage or the mixer. BIOS field-effect transistors a re cs- pccially suitable fo r use in FA1 rf- amplifier and mixer stages because of their inherently superior spurious- response rejection properties and signal-handling capal~ilities.

When spurious response is created in the rf amplifier, i t m:ly be removed by improved filtering I x - tween the rf amplifier and the mixer. The output of a n MOS-transistor rf amplifier is low in harmonics. As a result, the need for a double-toned rf intcrstage t ransfor~ncr is reduced and acceptable performance can usually be achieved with single-tuned circuits in both the antenna and 1.f interstage sections.

The dynamic-range c:il,al)ility of AIOS field-effect transistors is a l ~ p ~ l t 25 times greater than that of bipolar transistors. In an actual tuner circuit, this large intrinsii. dyna~nic range i s reduced by a factor proportional to the squ:\re of the circuit source impcdanccs. The net result is a practical dynamic range for RIOS tuner circuits a l~out five times tha t fo r l)ipolnr types.

With IIIOS field-elrect transistol.~, a s contrasted \vith either I~il,olar transistors or junction-gate field- effect transistors, there is 110 loading of the input signal, nor drastic change of input capacitance even under extreme overdrive contlitions.

In junction-gate field-effect transistors, a largc inconling signal can have sufficiently high positive swing to drive the gate into conduction by a momentary forward 11i;ls; power is then drawn from the input signal just a s if a resistance were p1:iced across the input circuit. In bipolar transistors, there is a gratlual change of both input impcrlnnce nnd input cap:icitance as a function of large signal excursions. Thcsc changes are undesirable hccausc they can result in detuning of tuned circuits and uridening of thc input selectivity curve.

Fig. 146 shows the basic circuit configuration for the "front-end"


Fig. 146-Circrril diagrctrrr oJ FM rrrrrer rrsi~rfi drml-gore MOS rrnirsislors ir~ /Ire r'f n~rrplifier arrd mixer stages.

stages of a n F M tuner that uses dual-gate-protected MOS field-effect transistors in both the rf-amplifier and mixer stages. A bipolar transistor is used in the local-oscillator stage. The detailed schematic diagram and functional description of a practical circuit of this type a re given in the Circuits section a t the back of this Manual.

Selection of appropriate source and load impedances for the rf stage should also taltc into consideration the fact that achievement of a low spurious response requires that the gate of the NOS transistor be tapped as f a r down on the antenna coil a s gain and noise considerations permit. This arrangement maltes possible optimum use of the available dynamic range of the MOS transistor.

The dual-gate nIOS transistor is very attractive for use in mixer service because the two signals to

be mixed a r e applied to separate gate terminals. This arrangement is a n effective technique for reduction of oscillator radiation. In the circuit shown in Fig. 146, the signal frequency is applied to gate No. 1 of the mixer transistor and the local- oscillator input to gate No. 2.

Figs. 147 and 146 show F M tuner circuits tha t use bipolar transistors only. The n-p-n silicon transistors used a re characterized by very low feedback capacitance, low noise, and high useful power gain, and feature a terminal arrangement in which the base and emitter terminals a r e interchanged to provide maximum isolation between the base and collector terminals. Although this basing configuration does not appreciably change the measured device-feedback capacitance, it does allow 'reduction of the collector-to- base capacitance due t o external circuitry.


La1)oratory results indicate t h a t although tuners using thrcc tuned circuits (including the oscillator tank) perform extremely ~vcl l with rcrard to ~ a i n , noisc, nnd rejection of ccrtain highcr-order sl,ut.ious responses, the addition of another tuncd circuit provides truly superior performance with regard to the attenuation of all spurious responses including image and the trouble- some "half-if."

Figs. 147 and 148 each show the schematic diagram of a four-coil tuner designed around bipolar transistors. The dc conditions of both circuits are identical. The rf-stage transistor operates in the common- emitter configuration a t a n emitter current of 1.5 milliamperes. This configuration offers the hichest stable gain a t FBI frequencies; the operating point specified was chosen a s a con~pron~ise between noise, gain, and spurious response rejection. The mixer transistor operates in a common-emitter configuration a t 1.5 milliamperes. The oscillator transistor operates in the common-collector configuration a t

approximately 2.5 milliamperes and provides approximately 28 n~illivolts of injection voltage to the mixer hase. The common-collector configuration was chosen because i t offers the grc!atest frccluency stal)ility with respect to changes in voltage and temperature. Also, if recom- mcndcd wiring practices a re adhered to, the use of the common-.collector oscillator minimizes higher-order spurious responses.

In Fig. 147, the antenna coil is double-tuned, and thus provides better selectivity characteristics ahead of the rf stage than a single- tuned transformer under the same impedance-matching condition. By using coils with unloaded, mounted Q's of 100, sufficient selectivity is realized so tha t a t signal levels up to 200 millivolts there a re no spurious responses within the F M frequency band. One disadvantage of double-tuned transformers is the coupling loss associated with them. Noise performance is degraded from t h a t obtained when single tuning is enlployed in the antenna coil by exactly the coupling loss of the double-tuned coil.

B ' 4-

F ~ E . 147-Fortr-coil Fbf lltrlcr with dortble-tuned nntelttta tro~rsforrrrer.


Because the I H F (Institute of H i ~ h Fidelity) sensitivity has developed into a n important requirement and because a low value of IHF sensitivity is determined in part by noise pcrform:ince, a circuit, Fia. 148, has been designed t h a t improves the noise performance and yet maintains a high degree of rejection of spurious responses. I t is

Neither of the four-coil tuner circuits shown in Figs. 147 and 148 uses a 10.7-MHz if t r ap because the need for such a t rap is eliminated with the use of the inductively tapped transformer.

A choice of first if transformer is offered. One version employs a capacitance-tapped secondary, a s shown in Figs. 147 and 148; the

*L_-Lp Fig. 148-Four-coil FM troler with doubk-tuned rj transjormer.

felt that although high selectivity ahead of the rf stage is desirable, i t is not essential. Laboratory tests indicate tha t the mixer is primarily responsible for spurious generation and that i t is more important to maintain low drive to the mixer base and to have adequate selectivity ahead of it. Because the over-all gain from antenna to mixer base must be kept low enough for spurious im- munity, and sufficiently high (10 to 15 dB) to mask mixer noise, i t is

, clear that all of the available maximum usable gain is not needed. A t a sacrifice of some gain, therefore, the selectivity characteristics of the double-tuned rf transformer can be improved by decreasing the coupling. I t is assumed tha t if harmonics a rc generated in the rf stage, they will be adequately attenuated by the rf transformer. With a single- tuncd antenna coil, circuit noise performance is improved for the reasons described.

other has a n inductively tapped secondary. Electrically, both transformers a r e identical.

A limiter circuit is essentially a n if-amplifier s tage designed t o provide clipping a t a desired signal level. Such circuits a r e used in F M receiv- e rs to remove AM components from the if signal prior to F M detection. The limiter stage is normally the last stage prior to detection, and is similar to preceding if stages. A t low input rf signal levels, i t amplifies the if signal in the same manner a s preceding stages. A s the signal level increases, however, a point is reached a t which the limiter stage i s driven into saturation (i.e., the peak currents and voltages a r e limited by the supply voltage and load impedances and increases in signal produce very little increase in collector current). A t this point, the if signal is "clipped" (or flattened) and fur ther increases in rf signal level produce no fur ther output in if signal t o the detector.


Limiter stages may be designcd to provide clipping a t various input- signal Icvels. A high-gain Fnl tuner is usually designed to limit a t very low rf input signal levels, ant1 pos- sibly even on noise signals. Atldi- tionar A M rejection may be obtained by use of a ratio detector f o r the frequency discriminator.

OSCILLATION

Ilipolar and field-effect transistor oscillator circuits are sirnilar in many respects to the tuned anlplifiers discussed previously, except that a portion of the output power is returned to the input network in phase with the starting power (regenerative or positive fcedbarl<) to sustain oscillation. DC bins-volt:~ge requirements fo r oscillators are similar to those discussed for amplifiers.

The maximum operating frcqucncy of an oscillator circuit is l in~i ted by the frcqucncy capability of the tran-

amplifier powcr gain 1)ccomcs less t h a n un i ty , osci l la t ions become sl~ialler with time (are "da~nped") until they cease to exist. In prarti- cal oscillator circuits, powcr gains greater than unity a r c rcquirctl because the power output is divided between the load and thc fced1)ack network, a s shown in Fig. 149. The feedback power must be equal to the input power plus the losses in the feedback network to sustain oscillation. ( A number of the oscillator circuits shown in the following sections on LC Resonant Feedback Oscilla- tors and Crystal Oscillators employ bIOS field-effect transistors. Al- though only single-gate. types a re shown in these circuits, the configurations a re equally applicable for use with dual-gate 'devices. In such applications, the dual-gate hlOS tra.n- sistor. is connected a s shown in Fig. 58 to provide pcrfornlance suhstan- tially equivalent t o tha t provided by the single-gate device.)

sistor used. The maximbm frequency of oscillation of a transistor is de- LC Resonant Feedback fined a s the freauency a t which the O S C ~ I I ~ ~ O ~ S

A - power gain is unity. Because some power gain is required in an oscillator circuit to overcome losses in the feedback network, the operating frequency must be some value below the transistor maximum frequency of oscillhtion.

For sustained oscillation in a transistor oscillator, the power gain of thc amplifier nctwoslc must 1)c cqual to or greater than unity. When the

The frequency-detcsn~ining elcmcnts of an oscillator circuit may consist of an inductance-capacitance (LC) network, a crystal, or a rcsist- ance-capacitance (RC) network. An LC tuned circuit may be placed in either the base circuit or the collector circuit of a com~non-emitter transistor oscillator. In the tuned-base oscillator shown in Fig. 150, one

- . I FEEDBAC

POWER I INPUT POWER - OUTPUT POWER

1 1 1 9 Fig. 149-Block tliograril of trrrtr~isfor oscillator sholvirrg divisiorr o/ olrfprrt power.


battery is used to provide all the dc operating voltages f o r the transistor. Resistors RI, R3, and R, provide the necessary bias conditions. Resistor R2 is the emitter stabilizing resistor. The conlponents within the dotted lines comprise the transis- to r amplifier. The collector shunt- feed arrangement prevents dc current flow through the tickler

L ------- J Fig. I50-Ttrned-base oscillator.

(primary) winding of transformer T. Feedback is accomplished by the mutual inductance between the transformer windings.

The tuned circuit consisting of the secondary winding of transformer T and variable capacitor C, is the frequency-determining element of the oscillator. Variable capacitor Cl permits tuning through a range of frequencies. Capacitor G couples the oscillation signal to the base of the transistor, and also blocks dc. Ca- pacitor C4 bypasses the ac signal around the emitter resistor R, and prevents degeneration. The output signal is coupled from the collector through coupling capacitor C3 to the load.

A tuned-collector transistor oscillator is shown in Fig. 151. In this circuit, resistors RI and % establish the base bias. Resistor R, is the emitter stabilizing resistor. Capaci- tors C, and C, bypass ac around resistors R1 and R?, respectively. The

Fig. 151-Tuned-collector oscillaror.

tuned circuit consists of the primary winding of transformer T and the variable capacitor Cs. Regeneration is accomplished by coupling the feedback signal from transformer winding 3-4 t o the tickler coil winding 1-2. The secondary winding of the transformer couples the signal out- pu t to the load.

Another form of LC resonant feedback oscillator is the Hartley oscillator. This oscillator makes use of split inductance to obtain feedback and may be either shunt o r series fed. In t h e shunt-fed circuit of Fig. 152, R1, Rz, and R. a r e the biasing resistors; the frequency- determining network consists of

OUTPUT "E

Fig. 152-Sl11tr1t-fcd Hartley oscillator.

variable capacitor C, in series with the windings of TI. The frequency of the oscillator is varied by CI; C2 is the dc blocking capacitor and C., is an ac bypass capacitor.

The circuit inductance fullctions in the manner of a n auto transformer and provides the regenerative feed- 1)aclc signal obtained from the voltage induced 'in the lower half of the tmnsfornler winding and coupled through C, t o the transistor base. No dc current flows through the primary of T, because the collector is shunt fed through R?.

In the series-fed Hartley circuit shown in Fig. 153, the base-emitter circuit is biased through R, and Rz;


Fig. 153-Series-fed Hartley oscillator..


the collector is biased th roag l~ the upper half of the transformer mind- ings. Again, a s in the shunt-fed circuit, C1 provides an ac bypass. Feedback in the series-fed Hartley circuit is obtained from the lower- half of the transformer windil~g and is coupled through C, to the hase of the transistor. The center-tap of the transformer \v ind i~~g is maintained a t ac ground potential by C:.

F i r . 151 shows two arrangements

circuit is dependent on the position of the t ap on the coil. Too little feedback results in a feedback s i p nal voltage a t the gate insuflicicnt

"D

to sustain oscillation; too nluch feed-. back causes the impedance between source and drain to become so low that the circuit becomes unstable. Output from these circuits can be obtained through inductive 'coupling t o the coil or through capacitive coupling to the gate.

Another form of LC resonant feedback oscillator is the transistor version of the Colpitts oscillator, shown in Fig. 155. Regenerative feedback is obtained from the tuned circuit consisting of capacitors CI

of n H:~rtley oscillator circuit using 310s field-effect transistors. Circuit ( a ) uses a bypassed source resistor to provide proper opc~,ating conditions; circuit (I)) uses a gate-leak resistor and biasing diode. The amount of feedback in cithcr

and C, in parallel with the primary winding of thc transformer, and is applied to the enlitter of the transistor. Base bias is provided by resistors F& and R2. Resistor R, is the collector load resistor. Resistor

RI develops the emitter input signal and also acts a s the emitter stabilizing resistor. Capacitors C1 and C, form a voltage divider; the voltage developed across CI is the feedback voltage. The frequency and the amount of feedback voltage can be controlled by adjustment of either or both capacitors. For minimum feedback loss, the ratio of the capacitive reactance between C, and CJ should be approximately equal t o the ratio between the output impedance and the input impedance of the transistor.

F i a 156 shows the field-effect transistor in use in two forms of the Colpitts oscillator circuit. These circuits a r e more commonly used in vhf and uhf equipment than the

Fb. I5G-Colpitts oscillator circuits rrsit~g MOS transistors.

Hartley circuits because of the mechanical difficulty involved in mak- ing the tapped coils required a t these frequencies by the Hartley circuits. Feedback is controlled in the Col- pitts oscillator by the ratio of the capacitance of C' to C".

Fig. 157, the gate-tickler-feedback oscillator circuit, and Fig. 158, the drain-tickler-feedback oscillator

~ V D ,a, '"D ( b )

Fig. 157-Gate-tickler-leedback oscillator circrtils.

circuit, have no particular advantages over the Hartley and Colpitts circuits except t h a t in some designs

( b l Fig. 158-Drain-tickler-feedback oscillator

circuits.

i t may be more economical to provide a tickler winding than the tapped coil o r capacitive divider required in . the Hartley or Colpitts circuits, respectively.

A Clapp oscillator is a modification of the Colpitts circuit shown in Fig. 155 in which a capacitor is added in series with the primary winding of the transformer to improve frequency stability. When the added capacitance is small compared to the series capacitance of C, and C,,

108 RCA Trans

the oscillator frequcncy is dcler- mined by the series LC conilination of the transformer primary and the addcd capacitor.

Crystal Oscillators A quartz crystal is often used

a s the frequency-determining element in a transistor oscillator circuit bccausc of its extremely high Q (narrow 1)andwidth) and good frequency stability over a given temperature rangc. A quartz crystal may be operatcd a s either a scries or parallel resonant circuit. As shown in Fig. 150, the clcctrical equivalcnt of the nieclianical vibrating characteristic of the crystal can be represented by a resistance R, an inductance L, and a capacitance C. in series. The lowest impedance of the crystal occurs a t tlie series resonant frequcncy of C, and L; the resonant frequcncy of the circuit is then determined only by the mechanical vibrating characteristics of the crystal.

The parallel capacitance C,, shown in Fig. 159 represents the elertro- static capacitance between the crystal electrodes. A t frequencies above the

Fig. 159-Eqiri~alort circrrir nl qltarrz crystal.

scrics resonant frequcncy, the combination of L and C. has thc cffect of a net inductance because the m- ductive reactance of L is greater than the capacitive reactance of C.. This net inductance forms a parallel rcsonant circuit with C,, antl any circu i t capacitance across the crystal. The impedance of the crystal is highest a t the parallel resonant f requency; the resonant frequency of the circuit is then deternlincd by

;istor, Thyristor, & Diode Manua l

both thc crystal and cxtcrnally connected circuit elements.

Increased frequency stability can I)c obtainctl in the turicd-collector and tuned-base oscillators discussed previously if a crystal is used in the feedback path. The oscillation frequency is then fixed by the crystal. At frequencies above and below the series resonant frequency of the crystal, the impedance of the crystal increases and the feedback is reduced. Thus, oscillation is prevented a t frequencies other than the series resonant frequency.

The parallel mode of crystal rcso- nance is used in the I'irrcc oscillator shown in Fig. 160. (If the c ~ y s t a l were replaced by its equivalent cir-

C 4

Fig. 160-Pierce-type tra~rsistor crystal oscillator.

cuit, the functioning of the oscillator would be analogous to that of the Colpitts oscillator shown in Fig. 155.) The resistances shown in Fig. 160 provide the proper bias and stabilizing conditions fo r the coninion-emit- t e r circuit. Capacitor C, is the emitter bypass capacitor. The required 180-degree phase inversion of the feedback signal i s acconiplishcd through the arrangement of the voltage-divider network C2 and Cx. The connection between the capacitors is grounded so that the voltage tle- veloped across C3 is applied betmcen base and ground antl a 180-degree phase reversal is obtai~letl. The oscillating frequcncy of thc circuit is determined by tlie crystal and the capacitors connected in parallel with it.


The field-effect transistor also operates ~vel l in crystal oscillator c.irr.uits srlch ;IS thc Picrcc-typc oscillators shown in Fig. 161. Pierce oscillator a re extremely popular because

/:is. 161-Pierce-tjpc crystal oscillrttor circrtils rtsi~rg M O S tra~~sistors.

of their si~nplicity and minimum number of components. At frequencies below 2 MHz, a capacitive voltage divider may be required across thc crystal. The connection between the voltage-divider capacitors must be grounded so that the voltage developed across the capacitors is reversed in phase by 180 degrees.

I t is frequently desirable to oper- a te crystals in communications equipment a t their harmonic o r overtone frequencies; Fig. 162 shows two circuits designed for this purpose. Additional feedback is obtained for the overtone crystal by the use of a capacitive divider a s the tuned-circuit bypass. Most third-overtone crystals operate satisfactorily \vithout this addit,ional feedback, but the extra feedback is required for the 5th and 7th har- n~onics. The tuned circuit in Figs. 1G2(a) and 1G2(b) is not fully bypassed and produces a voltage tha t aids oscillation. The crystal in both circuits is connected to the junction of the capacitors C,,' and C.,"; the ratio of these capacitors should be approxinlately 1:3.

The circuit of Fig. 1G3 operates well with low-frequency quartz bars. The crystal is located in the feedback circuit between the sources of

lbl Fi.q. 162-Crystal oscillator circuits per- ~ i ~ i ~ t i ~ l g ope ratio^^ at overtone or harmonic

jreqrte~rcies.

the two field-effect transistors and operates in t h e series mode. Capa- citor C? is normally used for precise adjustment of the frequency of the

VD -

Fig. 163-Low-jreqllency crystal oscillator circuit ~tsitlg MOS transistors.

110 RCA Tran

oscillator; a reduction in tile cnpa- citance increases the frequency slightly.

RC Feedback Oscillators

A resistance-capacitance (RC) network is sometimes used in place of an inductance-capacitance net\vork in a transistor oscillator. In the phase- shift oscillator shown in Fig. 164, the ILC network consists of three sections (C,K,, C2R2, and CJKI), each of which contributes a phase shift of 60 tlegrees a t the frequency of oscillation. Because the capacitive reactance of the network increases or decreases a t other frequencies, the 180-degree phase shift required for the common-emitter oscillator occurs only a t one frequency; thus, the output frequency of the oscillator is fixed. Phase-shift oscillators may be

Fig. 164-Trartsirtor RC plinre-s11iJf oscillator.

made variable over particular frequency ranges by the use of ganged variable cnpacitors or resistors in the RC networks. Three or more sections must be used in the phase- shifting networks to reduce feedback losses. The use of more sections contributes to increased stability.

FREQUENCY CONVERSION Transistors can be used in various

types of circuits to change the frequency of an incoming signal. In radio and television receivers, frequency conversion is used to change the frequency of the rf signal to an intermediate frequency. In communi-

isistor, Thyristor, & Diode M a n u a l

cations transmitters, frequency multiplication is often used to raise the frequency of the developed rf signal.

In a radio or television rcceiver, the oscillating and mixing functions a re performed by a nonlinear device such a s a diode or a transistor. As shown in the diagram of Fig. 165,

INTERMEDIATE-

OSCILLATOR e two voltages of different frequencies, the rf signal voltage and the voltage' generated by the oscillator, a re applied to the input of the mixer. These voltages "beat," or heterodyne, within the mixer transistor to produce a current having, in addition to the frequencies of the input voltages, numerous sum and difference frequencies.

The output circuit of the mixer stage is providcd with a tuned circuit which is adjusted to select only one beat frequency, i.e., the frequency equal to the difference between the signal frequency and the oscillator frequency. The selected output frequency is known a s the intermediate frequency, or if. The output frequency of the mixer transistor is kept constant fo r all values of signal frequency by tuning of the oscillator circuit.

In AM broadcast-band receivers, the oscillator and mixer functions a r e often accon~plished by use of a single transistor called an "auto- dyne converter". In FM receivers, stable oscillator operation is more readily obtained when a separate transistor is used for the oscillator function. In such a circuit, the oscillator voltage is applied to the niixer by inductive coupling, capacitive coupling, or a combination of the two.


AUTOMATIC FREQUENCY CONTROL

An :iutomatic frequency control (afc) circuit is often used to provide automatic correction of the oscillator frequency of a superheterodyne receiver when, fo r any reason, i t drifts from the frequency which produces the proper if center frequency. This correction is made by adjustment of the frequency of the oscillator. Such a circuit automatically compensates for slight changes-in rf carrier or oscillator frequency, a s well a s fo r inaccurate manual o r push-button tuning.

An afc system requires two sections: a frequency detector and a variable reactance. The detector section may be essentially the same a s the F M detector illustrated in Fig. 120. In the afc system, however, the output is a dc control voltage, the magnitude of which is proportional to the amount of frequency shift. This dc control voltage is used to control the bias on a transistor o r diode which con~prises the variable reactance.

Automatic frequency control i s also used in television receivers to keep the horizontal oscillator in step with the horizontal-scanning frequency a t the transmitter. A widely used horizontal afc circuit is shown in Fig. 166. This circuit, which is often referred to a s a balanced- phase-dcteetor or phase-discriminator circuit, is usually employed t o control the frequency of the horizon-

tal-oscillator circuit. The detector diodes supply a dc control voltage to the horizontal-oscillator circuit which counteracts changes in i ts operating frequency. The magnitude and polarity of the control voltages a re determined by phase relationships in the afc circuit.

The horizontal sync pulses obtained from the sync-separator circuit a re fed through a phase-inverter or phase-splitter circuit to the two diode detectors. Because of the action of the phase-inverter circuit, the signals applied to the two diode units a re equal in amplitude but 180 degrees out of phase. A reference sawtooth voltage obtained from the horizontal output circuit is also applied simultaneously to both units. The diodes a r e biased so tha t conduction takes place only during the tips of the sync pulses. Any change in the oscillator frequency alters the phase relationship betwen the reference sawtooth and the incoming horizontal sync pulses, and thus causes one of the diodes to conduct more heavily than the other so tha t a correction signal is produced. The system remains unbalanced a t all times, therefore, because momen- ta ry changes in oscillator frequency a re instantaneously corrected by the action of this control voltage. The network between the diodes and the horizontal-oscillator circuit is essentially a low-pass filter which prevents the horizontal sync pulses from affecting the horizontal-oscillator performance.

Fig . 166-Balatrccd-phase-deteclor or phase-discrir~~inator circuit for horizontal afc.

"

FROM PHASE INVERTER^ 0-I(

AN' REFERENCE-

VOLTAGE FROM

DC CORRECTION VOLTAGE TO

-HORIZONTAL OSCILLATOR

-- -- ,, L -

HORIZONTAL - OUTPUT CIRCUIT -

T IIE amplifying action of a transistor can be uscd in various ways

in electronic circuits, depending on the ~ ~ c s u l l s desired. The four recog- ~lizctl classes of amplifier service can be defined for transistor circuits a s follows:

A class 11 amplificr is an amplifier in which the basc bias and alter- ~ ~ a t i n g signal a r c such t h a t collector currcnt in a transistor flows continuously during the con~plctc electrical cycle of the signal, and even when no s i ~ n a l is present.

A class AIj aniplilier is an amplifier in which the Imse bias and alternating signal a rc such t h a t collector current in a transistor flows for np- preciahly more than half but less than the entire electrical cycle.

11 class 1% an111lifier is an amplifier In ~vhich the base is biased to ap. proximately collector-curre11 t cutoff, so that collector current is npprosi- niately zero when no signal is applied, arid so tha t collector currcnt in a transistor flows for approximately one-half of each cycle when a n alternating signal is applied.

A class C an~l~ l i l i e r is a n amplifier in which the basc is biased t o such n degree that the collcclor current in a transistor is zero when no signal is applied, and so tllnt collector currcnt in a tmnsistor flows for appreciably less tlinn onc~-half of each cycle when a n alternntii~g signal is applied.

For ratlio-frequency ( r f ) nn~plifiers which operate into sclcctive tuned circuits, such :IS Lhc Tuned Arnplificrs discussed in the section on Recciver Tuner-Circuit Apn1ic.n- tioris, 01. f o r other rnnl~1ifiei.s in which distortion is not a I)I iinc

factor, any of the above classes of amplification may be used \vith cither a single transistor or a push- pull stage. For audio-frequency ( a f ) amplifiers in which distortion is an important factor, single transistors can be usctl only in class A amplifiers. For class AB or class 13 autlio- amplifier service, a l~alancctl nmplifier stage using two transistors is required. A push-pull stage cm1 d s . ~ be used in class A audio amplifiers to obtain reduced distortion and greater power output. Class C ampli- tiers cannot be used for audio or AM applications.

AUDIO AMPLIFIERS Audio amplifier circuits are uscd

in radio ant1 television receivers, public address systems, sound re- corders ant1 reproilucers, and similar applications to amplify signals in the frequency range from 20 to 20,000 Hz. Each transistor in an audio amplifier can be consideretl a s either a currcnt amplifier or a power amplifier. The type of circuit configuration selected is dictated by the requirements of the given application. T11c output power to be supplied, the required se~rsitivity and frequency response, and the maxin~uni distortion limits, together with the capal,ilities and lin~itations of available devices, a re the niain criteria uscd to determine the circuit that will ~,rovide the desired performance most efficiently and econon~ically.

I n addition to the consideration that must bc given to the ncl~ieve- 111ent of pcrforinanc:e objectives and the selcctio~i of the optintwn c:i~,cl.lit configuration, the circuit desi!:ner

Low-Frequency Amplification 113

nus st also take steps to assure re- low-level output transducers such as liable operation of the audio ampli- microphones, hearing-aid and phono- ficr under varying conditions of graph piclrup devices, and recorder- signal level, frequency, ambient reproducer heads. temperature, load impedance, line voltage, and other factors which may Noise Figure--One of the im- subject the transistors to either portant characteristics of a low- r transietlt or steady-state high stress level amplifier circuit is i ts signal- levels. L ~ w - ~ o s t , low-power audio to-noise ratio, o r noise figure. The systenls (such a s those used in input circuit of a n amplifier inher- mobile and TV output stages), in ently contains some thermal noise which high operating emciency is contributed by the resistive elements not all important consideration, in the input device. All resistors usually employ a single-ended, class generate a predictable quantity of A, transformer-coupled output stage noise Power a s a result of thermal such a s tha t shown in Fig. 167. activity. This power is about 160 dB

below one watt f o r a bandwidth of' 10 kHz.

When an input signal is amplified, therefore, the thermal noise generated in the input circuit is also

INPUT amplified. If the ratio of signal

Oj power to noise power (S/N) is the same in the output circuit as in the

UTPIJT input circuit, the amplifier is considered to be "noiseless" and i s said to have a noise figure of unity, or zero dB.

In practical circuits. however. the ratio i f signal power to noise pbwer is inevitably impaired during amplification a s a result of the generation of additional noise in the circuit ele-

Fix. 167-Typicnl low-power a~idio-a111pli- fier circlcit.

The input to a n audio amplifier is a low-power-level audio signal from the phonograph or magnetic- tape picltup head or, in a radio receiver, from the detector stage a s indicated in Fig. 129. This signal is usually amplified through a preamplifier stage, one or more low-level (pre-driver o r driver) audio stages, and a n audio power amplifier. The system may also include frequency- selective circuits which act a s equalization nctworlts and/or tone controls.

Low-Level Audio Stages

Si~nple class A amplifier circuits are norn~ully used in low-level audio stages such a s ~~renmplifiers and drivers. Prealnplifiers usually follow

ments. A measure of the degree of impairment is called the noise figure (NF) of the amplifier, and is expressed a s the ratio of signal power to noise power a t the input (Sl/NI) divided by the ratio of signal power to noise power a t the output (So/No), a s follows:

The noise figure in dB is equal to ten times the logarithm of this power ratio. For example, a n amplifier with a I-dB noise figure decreases the signal-to-noise ratio by a factor of 1.26, a 3-dB noise figure by a factor of 2, a 10-dB noise figure by a factor of 10, and a 20-dB noise figure by a factor of 100.

In audio amplifiers, i t is desirable that the noise figure be kept low. I n general, the lowest valoe of N F is


obtained by usc of an en~i t t c r current of less than one milliampere and a colIector voltage of less than two volts for a signal-source resistance bctwecn 300 and 3000 ohms. If the input impedance of the transistor is matched to the in~pedance of the signal source, the lowest value of N F tha t can bc attained is 3 dB. Gcncr- ally, the best noise figure is oblainetl by use of a transistor input impedance approxinlately 1.5 timcs the source impedance. However, this condition is often not realizable in practice because lnnny transducers a re reactive rather than resistive. I n addition, other requ'ircments such a s circuit ra in, signal-handling capability, and reliability may not permit opt i~i~izat ion for noise.

In the simple low-level amplifier s t a r e shown in Fig.. 168, resistor R I

can I)e recordcd on a p11onogral)h record or on magnetic tape depend on several factors, including tlie composition, niechanical characteristics, and speed of the rccord or tapc, and the electrical and mcchani- cal characteristics of the recording equipment. To achieve wide frequency and dynamic range, manufacturers of con~niercial recordings use equipn~cnt which introduces a nonuniform relationship between atn- plitude and frcqucncy. This rcla- tionship i s known a s a "recording characteristic". To assure proper reproduction of a high-fidelity record in^, therefore, some part of the reproducing system nlust have a frequency-response characteristic which is the inverse of the recordinr characteristic. Most manufacturers of high-fidelity recordings usc the RIAA characteristic for discs :tnd " the NARTB characteristic for magnetic tape.

The siniplest type of equalization network is shown in Fig. 10. Be- cause the capacitor C is effectively an open circuit a t low frequencies, thc low frequencies must be passed through the resistor R and a r e attenuated. The capacitor has a lower reactance st high frequencies. how-

oped across the load resistor R=. The collector voltage and the emittcr current a rc kept relativclg low to retluce STAGE STAGE the noise figure. If the load in~pcd-

compared to R., very little \,oltage ance across the capacitor C= is low riK, ]69-Sirlrp/c~ RC ~ r c q l l c l t c l - c ~ ~ t ~ ~ ~ ~ ~ ~ ~ -

snriorr rrcrn~ork. swing results on the collector. There- fore, ac feedback through R I (loe? not receive ncgligil)le attenuation. ~ 1 1 ~ s cause much reduction in gain . ' ) the network ef(ectively "l~oosls" the -

Etlualization-In many cnscs, lo\\- high frequencies. This type of equiili- Irvel amplifier stages used :IS prc- . zation is called "attenuative." amplifiers inclutle solne typo of Some typical prca~llplificr stages frctl~rcncy-conlpe~~sotion ~retn.orlc to are shown in the Circuits section. enhance either the low-frequcncy or Thc location of the frcqucncy-com- tlic high-frequency components of pensation network or "equalizer" in the input signal. The frequency the reproducing systcni depends on range and dynamic rango':' wllicli thc types of recordings which are to * T h e dynamic rnnfic or a n nn~p l i f i c r is n mrnsurv of i ts s i cn :~ l -h ;~n# l l inc rnlrnbilits. T l ~ e

dynnn>ic r:>ngc cxprcsscs i n d l \ t l lc m t i n of t h e mnximum u:;:i\)lc ( I I I ~ I I L I ~ s i ~ n : l l I L . c ~ P ~ : I I I Y f o r n dis tor t ion of nbout 10 per crnf 1 t o tllp r n i n i l n ~ ~ l n us;lLle o u t l ~ u t s ipnn l (ccnr .r :~l ls f o r a siannl-to-n'sisc ~ n t i o I,T nbuut 20 rll31. A rlynnmic r n n m of 40 clLI is uslinlly accrptnblc: n v n l ~ ~ c of 70 $113 in rxcept ional flbr nny nutlio e s ~ t e m .

Low-Frequency Amplification

be reproduced and on the pickup devices used. All commercial pickup devices provide very low power levels to a transistor preamplifier stage.

A ceramic high-fidelity phonograph pickup is usually designed to provide proper compensation for the RIAA recording characteristic when the pickup i s operated into the load resistance specified by its manufacturer. Usually, a "matching" resis- to r is inserted in series with the.input of tlie preamplifier transistor. How- ever, this arrangement produces a fairly small signal current which must then be amplified. If the matching resistor is not used, equalization is required, but some improvement can be obtained in dynamic range and gain.

A magnetic high-fidelity phonograph pickup, on the other hand, usually has a n essentially flat frequency-response characteristic. Be- cause a pickup of this type merely reproduces the recording characteristic, i t must be followed by a n equalizer network, a s well a s by a preamplifier having sufficient gain to satisfy the input requirements of the tone-control amplifier and/or power amplifier. Many designs include both the equalizing and amplifying circuits in a s i n ~ l e unit.

A high-fidelity magnetic-tape pickup head, like a magnetic phonograph pickup, reproduces the recording characteristic. This type of pickup device, therefore, must also be Iol- lolvcd by a n equalizing network and preamplifier to provide equalization for the NARTB characteristic.

Feedback networks niay also be used for frequency compensation and for retluction of distortion. Basically, a feetlhack network returns a portion of the output signal to the input circuit of an amplifier. The feedback signal niay be returned in phase with the input sirnal (positive or regenerative feedback) or 180 degrees out of phase with the input signal (negative, inverse, or degenerative fectlbacl~). I n either case, the feedback can bc made proportional to either the output voltage or the out-

put current, and can be applied to either the input voltage or the input current. A negative feedback signal proportional t o the output current raises the output impedance of the amplifier; negative feedback proportional to the output voltage reduces the output impedance. A negative feedback signal applied to the input current decreases the input impedance; negative feedback applied to the input voltage increases the input impedance. Opposite effects a r e produced by positive feedback.

A simple negative o r inverse feedback network which provides high- frequency boost is shown in Fig. 170.

STAGE -

Fig. 170-Negative-feedback frequency- cor~~pct~sation network.

This network provides equalization comparable to that obtained with Fig. 1G9, but is more suitable fo r low-level amplifier stanes because i t does not require the first amplifier s tage t o provide high-level low frequencies. In addition, the inverse feedback improves the distortion characteristics of the amplifier.

Input Impedance-As mentioned previously, i t is undesirable to use a high-resistance signal source for a transistor audio amplifier because the extreme impedance mismatch results in high noise figure. High source resistance cannot be avoided, however, if an input device such a s a ceramic pickup is used. In such cases, the use of negative feedback to raise the input impedance of the amplifier circuit ( to avoid mismatch loss) is no solution because feedback cannot improve the signal-to-noise ratio of the amplifier. A more practical method is to increase the input impedance somewhat by operating the transistor a t the lowest practical current level and by using a transistor which has a high forward current-transfer ratio.


Yolurne nrtd Tone Controls- Somc preamplifier or lo\\.-levcl audio n~liplilier circuits inclutle vmi- ahlr resistors or pott~ntio~nctel~s which function a s volume or tone contl.ols. Such circuits shoultl be tlcsigl~ctl to minimize tllc flow of dc currc!nts t h r o u ~ l i these controls so thnt little 01. 110 uoisc nfill I)c devc1opc.d 1,s thc movable contact during the life of the circuit. 'Jolumc! controls and tllrir associatcd circuits should pcrmit variation of gain from zero to maximunl, ant1 should attcnuntc all frrqucvlc,irs equally for all positions of the varial)le arm of the control. Several cxalnplcs of volun~e cont1.01~ 311d totic controls a re shown in the Circuits srction.

A tonc control is a variable filtcr (or one in which a t lcast one element is adjustable) by means of \vhich the user may vary the frequency response of an amplifier to suit his own taste. I n radio reccivcrs and home a~ilplifiers, the tonc control usually consists of a rcsistancc-capncitn~lcc ~ictworlc in which the resistance is the varinblc clc~nent.

The simplest form of tonc conlrol is a "trel)lc cut" networlc such as that sl~own in Fig. 171. As JI, is ~natle smaller, the capacitor C? 1)s-

.])nsscs ~ n o r c of the high audio frc- quencics; t.herefore, the output of thc nctwork is dccreascd by an amount dependent upon the value of R,. The rcsistancc of R , should be very largc in comparison to the

reactance of Cs a t the highest audio frequency.

The tone-control network shown in Fig. 172 has two stages with completely separate bass and treble con- "

trols. Fig. 173 sllo\\~s simplified rcprcscntations of t h e bass control whcri the potentiometer is turned to its extreme variations (labeled COOST and CUT). A t vcry high frequencies, C, ant1 C2 are cffcctively short circuits and the network becomes the s i~nple v o l t a ~ c tlividcr It, and R,. In the bass-boost position, It., is inserted in series with R2 so that there is lcss attenuation to vcry . low frequencies than to very high frequencies. Therefore, the bass is said to be "boosted". In the bass-cut position, R. is inserted in series with Rl so tha t there is more attenuation to very low frcqucncics.

BASS TREBLE

B+

CUT

Low-Frequency Amplif ication 11 7

A BASS CUT corporates negative feedback, the 7 tone control must be inserted in a par t of the amplifier which is external to the feedback loop, or must be made a part of the feedback network. The over-all gain of a well designed tone-

Rz control network should be approxi- - mately unity. The system dynamic - range should be adequate fo r all fre-

~ i , p . /73--~irrrp/i/ieri ,rpr.esrrrrntior~s o f quencies anticipated with the tone boss-corrtrol circrrit crt rstrettle e r ~ d s o f po- controls in any position. The high-

tcr~tiortrctcr. frequency gain should not be ma- terially affected a s the bass control

Fig. 174 shows extreme positions is varied, nor should the low-fre- of the treble control. R; is generally quency gain be sensitive to the much larger than R, o r Ra and may treble control. be treated a s an onen circuit in the extrcmc positions. i n both the boost and cut positions, very low frequencies a re controlled by the voltage divider Ri and Ra. In the boost position,

TREBLE BOOST TREBLE CUT

Fir. 174--Sitrrpli/icd reprcserltafiorrs o f trcblc-cotrtrol circrrit ot cxtrctrre crrds o f

porcrrtiotrreler.

1 R , is bypassed by the high frequencies and the voltage-divider point D is placed closer to C. In the cut position, Its is bypassed and there i s

, greater attenuation of the high frequencies.

The frequencies a t which boost and cut occur in the circuit of Fig. 172 are controlled by the values of C,, C?, C,, and Cr. Both the output im-

, pedance of the driving stage (gen- , erally R 1 . , ) and the loading of the 1 driven stage affect the response

curves and must be considered. This I

tone-control circuit, like the one in Fig. 171, is attenuative. Feedback

1 tone controls may also be employed. The location of a tone-control net-

work is of co~~sidcrable importance. I n a typical preamplifier, i t may be in the collector circuit of the final low-level stagc or in the input circuit of the first stage. If the amplifier in-

Driver and Output Stages

Driver stages in audio amplifiers a re located innnediately before the power-output stage. When a single- entled class A output stage is used, the driver stage is similar to a preamplifier stage. When a push-pull output stage in which both transistors are the same type (n-p-n or p-n-p) is used, however, the audio driver must provide two output signals, each 180 degrees out of phase with the other. This phase requirement can be met by use of a tapped- secondary transformer between a single-ended driver stage and the output stage, a s shown in Fig. 175.

Fip. 175-Driver stage for plrsll-prtll oltt- p ~ t t circlrir.

The transformer T, provides the required out-of-phase input signals for the two transistors Q , and Qr in the push-pull output stage.

Transistor audio power amplifiers may be class A single-ended stages,

11 8 RCA Transistor, Thyristor, & Diode Manual

or class A, class AB, or class B push-pull stages. A simple class A single-ended power amplifier is shown in Fig. 176. Conlponent v:~lucs which will provide the desired power output can be calculated from the

TO SPEAKER

transistor characteristics and the supply voltage. For example, a n output of four watts may be desired from a circuit operating with a supply voltage of 14.5 volts (this voltage is normally available in auto- mobiles which have a 12-volt ignition system). If losses a re assumed to he negligible, the power output (Po) is equal to the peak collector voltage (e,) times the peak collector current (i.), each divided by the square root of two to obtain rms values. The peak collector current can then be determined a s follows:

- - 4 (2) li x (2)"? 14.5

= 0.55, or approximately 0.6 ampere.

to the 0.6-ampere collector current.) The current through resistor Ra

should be about 10 to 20 per cent of the collector current; a typical value i s 15 per cent of 0 5 , o r 90 milliamperes.

The voltage from base to ground is equal t o the base-to-emitter voltage (determined from the transistor transfer-characteristics curves fo r the tlesired collector or emitter current; normally about 0.4 volt f o r a germanium power transistor opernt- ing a t a n emitter current of 600 milliamperes) plus the emitter-to-ground voltage (0.G volt a s described above), o r one volt. The voltage across Rz, therefore, is 14.5 minus 1, o r 13.5 volts. The value of RI must equal 13.5 divided by 90, o r about 150 ohms.

Because the voltage drop across the secondary winding of the driver transformer TI is negligible, the vole: age drop across R, is one volt. The current throughmR, equals the current through R, (90 milliamperes) minus the base current. If the dc forward current-transfer ratio (beta) of the transistor selected has a typical value of 60, the base current equals the collector current of 600 milliamperes divided by 60, or 10 milliamperes. The current through R1 is then 90 minus 10, o r 80 milliamperes, and the value of R, is 1 volt divided by 80 milliamperes, or about 12 ohms.

The transformer requirements a re determined from the a c voltages and currents in the circuit. The peak collector voltage swing that can be used before distortion occurs as a result of clipping of the output volt- In class A service, the dc collector age is about 13 volts. The peak col- and the peak swing lector current swing available before

a re about the same- current cutoff occurs is the dc cur- lector voltage and current a re 14.5- rent of milliamperes. ~ h ~ ~ ~ ~ ~ ~ ~ , "its and ampere, the collector load impedance s h o ~ ~ l d

The the re- be 13 volts divided by 600 rnilli- sistor RE in Fig. 176 usually ranges amperes, or about 20 ohms, and the

Om3 a output transformer T, should be de- 0.6 volt can be assumed. The value of RK must equal the O.G-volt drop a 20-0hm primary divided by the 0.6-amnere emitter impedance to the desired speaker im- current, or one ohm. he emitter pedance. If a 3.2-ohm speaker is current is assumed to be nearly equal used, fo r example, the impedance


values for TI should be 20 ohms to 3.2 ohms.

The total input power to the circuit of Fig. 176 is equal to the voltage required across the secondary winding of the driver transformer T, times the current. The driver signal current is equal to the base current (10 milliamperes peak, o r 7 milliamperes rnm). The peak ac signal voltage is nearly equal to the sum of the base-to-emitter voltage across the transistor (0.4 volt a s determined above), plus the voltage across RE (0.G volt), plus the peak ac signal voltane across R, (10 milliamperes times 12 ohms, o r 0.12 volt). The input voltage, therefore, i s about one volt peak, o r 0.7 volt rms. Thus, the total a c input power required. to produce a n output of 4 watts is 0.7 volt times 7 millialnperes, o r 5 milliwatts, and the input impedance is 0.7 volt divided by 7 milliamperes, or 100 ohms.

H i g h e r power o u t p u t c a n be achieved with less distortion in class A service by the use of a push-pull amplifier. One of the disadvantages of a transistor class A amplifier (single-ended or push-pull), however, is tha t collector current flows a t all times. As a result, transistor dissipation is highest when no ac signal is present. This dissipation can be greatly reduced by use of class B push-pull operation. When two transistors a r e connected in class B push-pull, one transistor amplifies half of the signal, and the other transistor anlnlifies the other

tortion, called cross-over distortion, can be suppressed by the use of a bias voltage which permits a small collector current flow a t zero signal level. Any residual distortion can be further reduced by the use of negative feedback.

OUTPUT COLLECTOR

CURRENT

Fig., 177-Waveforms showing cause of cross-over distortion.

A typical class B push-pull audio amplifier is shown in Fig. 178. Re- sistors RE, and Rn a r e the emitter stabilizing resistors. Resistors R, and R, form a voltage-divider network which provides the bias for the transistors. The base-emitter circuit is biased near collector cutoff so tha t

half. These half-sknals are then combined in the output circuit to restore the original waveform in a n amplified state.

Ideally, transistors used in class B push-pull service should be biased to collector cutoff so tha t no power is dissipated under zero-signal conditions. At low signal inputs, however, the resulting signal would be '2 "cc distorted, a s shown in Fig. 177, be- -11 + cause of the low forward current-

very low currents. This type of dis-

& transfer ratio of the transistor a t Fig. 178-C1o.r~ B prtsl~-~rrll altdio-ar,lpli-

fier circuit.


vcry littlc collector powcr is tlissi- patcd undcr no-signal conditions. The charactcristics of thc hias nct- worlc must ltc vcry carefully clloscn so that the bias voltage will be just suf l ic ic~~t to minimize cross-over tlistortion a t low signal Icvcls. J3ecausc the collector current, collector dissl- pation, and clc operating point of a transistor vary with ambient ten]- perature, a tempcraturc-sensitive rcsistor (such a s a tl~erniistor) or a bias-compensating diode may be used in the biasing netmorlc to minimize the effect of temperature variations.

The advnntages of class R push- pull operation can be obtained without the need for an output transformer by use of a circuit such a s that shown in Fig. 179. In this circuit, the secondary wintlings of the driver transformer T, a rc pliasctl so that a negative signal from basc to cmittcr of o11c transistor is accompanied by a positive signal from

transistors slightly above cutofr undcr no-signal conditions ant1 thus minimizes cross-over distortion. The cniilter resistors R , , antl I<,, hell) to compcnsatc for ditfcrenccs 1)ctwecn trarisistors and for the elfccts of anl1)icnt-ten~pcrature variations.

T h e secondary windings of any class B driver t r a ~ ~ s f o r m c r shoultl be bililar-wound (i.e., wo~untl together) t o obtain t i g h t e r coupling and thereby minimize leakage inductance. Othcrwisc, "ringing" may occur in the cross-over region as a rcsult of the cnergy storetl in the leakage inductance.

Decause junction transistors can be made in both p-n-p and n-p-n typcs, they can bc uscd in contl)le- mentary-symmetry circuits to obtain all the atlvantages of conventional push-pull amplifiers plus direct cou- ., pling. The arrows in Fig. 180 indicate the dircction of electron current flow in the terminal leads of p-n-p and n-p-n transistors. When these

p-n-p p3

Jm Fig. 180-Electt-otr-crrrrrrlr arrd rr-p-11 tr-artsistors. floiv it1 1,-tr-I>

=VCC two transistors a r e connected in a single stage, as shown in Fig. 181,

R ~ 2 '4 the steady-state electron current path in the output circuit is conl- pleted through the collector-emitter

Fig. 179-Sirrgle-etrtled class B cir-crril. C-

base to emitter of thc other transistor. When a negative signal is applied to the base of transistor (la, - -=- Vccl

for example, Q, draws currcnt. This c u r r e n t m u s t f low t h r o u ~ h t h e load because the accompanying positive signal on the base of tmnsistor nal polarity Q2 C L I ~ S reverses, Qz off. When transistor thc s i r - Q, lGqy = + V C C ~

is cut od, \vliile Q: conducts cui-rent. - The resistive dividers R,R, and R,R, ~ l g . 181-Basic cot~rplcllletl~nrl-s~lll~trc~rry provide a dc bias which liccps the C I ~ C I ( I ~ .


circuits of the transistors. In the circuits of Figs. 179 and 181, essentially no steady-state current flows through the load rcsistor Rr.. There- fore, the voice coil of a loudspeaker can be connected directly in place of RI. without excessive speaker cone distortion.

The true complementary amplifier, shown in Fig. 182, is the simplest of all complementary circuits. I t s features include a single

Fig. I82-Tr~ce-con~plorrerltary atirplifier.

driver transistor, a single diode for bias, antl the application of turn-off drive to the output devices. Because i t requires a class A driver and both p-n-p and n-p-n output devices and has high standby current, the true- complementary design is seldom used for power-output levels in excess of 25 watts rms.

The class A driver stage shown in Fig. 182 requires the use of a large heat sink. The p-n-p power device in the complementary output stage is more expensive and has lower safe-area ratings than i t s n-p-n equivalent. Because control of base diffusion is more difficult in p-n-1) devices, these types a re generally 25-per-cent costlier than comparable n-p-n types.

One way to avoid the high cost of power p-n-p transistors is to employ a quasi-complementary circuit such a s that shown in Fig. 183. In this type of circuit, a low-current

CLASS

Fig. 183-Quasi-cor~rple,t1c~ary arriplificr.

p-n-p transistor is directly coupled to a high-current n-p-n transistor to simulate a high-current transistor, a s shown in Fig. 184.

The advantages of quasi-comple- nlentary amplifiers include improved safe area fo r the n-p-n output transistor, lower cost, and the use of class B drivers. The major disadvantages a re the need for two driver transistors and two bias diodes, and the absence of turn-off drive to the output transistors. Be- cause the advantages f a r outweigh the disadvantages for high-power amplifiers, quasi-complementary circuits a re generally used a t power levels above 25 watts rms. The high- frequency response of such circuits can be improved by use of bleeder resistors in the base circuits of the output transistors.


In both true-complement:~ry and quasi-con~plementary circuits, the output devices do not need. to be well matched for beta. These circuits are essentially voltage amplifiers used in a n emitter-follolver configuration tha t has a voltage gain of nearly unity which varies only s l i ~ h t l y with transistor beta. In the higher-power quasi-complementary amplifier, the effect of beta is even less important because a Darlington- connected stage is used. The basic requirement is tha t a minimum current gain be maintained from mininlum to maxitnun~ drive.

EFFECTIVE EMITTER ?

(O) EFFECTIVE COLLECTOR EFFECTIVE.

EMITTER P

(b) EFFECTIVE COLLECTOR

Several high-fidelity anlpliflcrs a t e shown in the Circuits section. The performance capabilities of such amplifiers are usually given in terms of frequency response, total harmonic distortion, maximurll power output, and noise level. To provide high- fidelity reproduction of audio pro- Kranl material, a n alnplificr should have a frequency response which does

not vary more than 1 dB over the entire audio spectrum. General practice i s to design the amplifier so tha t i ts frequency response is flat within 1 dB from a frequency well below the lowest to be reproduced to one well above the upper limit of the audible region.

IIarmonic distortion and intermodulation distortion p r o d u r e changes in program material which may have adverse effects on the qual- i ty of the reproduced sound. Har- monic distortion causes a change in the character of a n individual tone by the introduction of harmonics which were not originally present in the program material. F o r high- fidelity reproduction, total harmonic distortion (expressed a s a percent- a g e of the output power) should not be greater than about 0.5 per cent, a t the desired listening level.

Intermodulation distortion is a change in the waveform of a n individual tone a s a result of interaction with another tone present a t the same time in the program material. This type of distortion not only alters the character of the modulated tone, but may also result in the generation of spurious signals at frequencies equal to the sum and difference of the interacting frequencies. Intermodu- lation distortion should be less than 2 per cent a t the desired listening level. I n general, any amplifier which has low intermodulation distortion will have very low harmonic distortion.

The mnxinrum power output which a high-fidelity amplifier should deliver depends upon a complex relation of several factors, including the size and acoustical characteristics of the listening area, the desired listening level, and the eficiency of the loudspeaker system.

The noise level and maximum output power determine the range of volu~ne the amplifier is able to reproduce, i-e., the difference (usually expressed in dB) between the loudest and softest sounds in program material. Because the greatest volunle range utilized in electrical program


material a t the present time i s about 60 dB, the noise level of a high- fidelity amplifier should be a t least 60 dB below the signal level at the desired listening level.

The design of audio equipment f o r direct operation from the a c power line normally requires the use of either a power transformer o r a large voltage-dropping resistor to reduce the 120-volt ac line voltage to a level that is appropriate f o r transistors. Both of these techniques have disadvantages: The use of a transformer adds cost t o the system. The use of a dropping resistor places restrictions on the final packaging of the instrument because the resistor must dissipate power. I n addition, low- v o l t a ~ e supplies a re usually more expensive to filter than high-voltage supplies.

The use of high-voltage silicon transistors eliminates the need f o r either a power transformer or a high- power voltage-dropping resistor, and permits the use of economical eir- cuits and components in line-operated audio equipment. Several ac/dc circuits using these high-voltage transistors a r e shown in the Circuits section. The basic class A audio output stage shown in Fig. 185 is essentially of the same design a s the class A amplifier discussed previously. Be- cause the supply voltage is much higher, however, the currents a re ahout one-tenth a s high and the impedances about 100 times a s high.

The use of a voltage-dependent resistor (VDR) a s a damping resistor across the primary winding of the output transformer in Fig. 185 pro- tects the output circuit against the destructive effects of transient voltages t h a t can occur under abnormal conditions. If the VDR were not used, the peak collector voltage under transient conditions could be a s high as five to ten times the supply voltage, o r f a r in excess of the breakdown-voltage rating f o r the transistor. Because the resistance of the VDR varies directly with voltage, its use limits the transient voltage to

safe levels bu t does not degrade over- all circuit performance.

Fig. 185-Basic audio-output stage for line- operated equipment.

Fig. 186 shows another effective method for protection against transient voltages. I n this arrangement,

DRIVER

- Fig. 186-Alternate mefhod for proleclion

against transient vobages.

the output transformer is replaced by a center-tapped transformer and a silicon rectifier t h a t has a peak- reverse-voltage rat ing of 300 t o 400 volts. The peak voltage across t h e output is thus limited to a value which does not exceed twice the magnitude of the supply voltage. A s t h e collector voltage approaches a value equal to twice the supply voltage, the voltage a t the diode end of the transformer becomes sufficiently negative to forward-bias the diode and thus clamp the collector voltage. The required transformer primary impedance is generally about 10,000 ohms center-tapped; in addition, it i s recommended t h a t a bifilar winding be used to minimize leakage inductance. Because the arrangement


shown in Fig. 186 providcs more reliable protection against transients than that of Fig. 185, a higher supply voltage and a higher transfonncr i~npedance can I)e uscd.

I t sliould be noted tha t special prc- cautions a re required in thc construction of circuits f o r line-voltage opcrntion. Because these circuits operate a t h i ~ l i ac and dc voltages, special rare niust be exercised to assure that no metallic par t of tlie chassis or output transformer is exposed to touch, accitlcntal or otherwise. The circuits should be installed in non-metallic cahincts, or should he

When the input signal is positive, the output current increases and 01)- posite voltage polarities a re established across rcsisto~,s R, and It,. Thus, two output signals are pro- ducctl which are 180 dcgrccs out of phase with each other. This circuit provides the 180-dcgrce phase relationship only when each load is resistive and constant throughout the entire signal swing. I t is not suitable a s a driver stagc for a class B output stage. . :

DC AMPLIFIERS properly insulated from metallic cnl)incts. Insulated ltnobs sliould bc are

usctl for potentiometer shafts and nonnally used in transistor circuits

switches. to amplify small dc or very-low-

A phase inverter is a type of class frequency ac signals; tliey can am- A alnplifier usct] when tlVo out-of- plify a

are requiretl. zero hertz. The upper frequency limit

split-load phase-inverter stage shown , of such an anlplifier niay range"

in F ~ E . 187, the output current of from a few hundred hertz in gen- transistor Q, both the eral-purpose electrolneter applica-

tions to several rnerrahertz in other PHASE

collector load resistor R, ant1 tlie emitter load resistor R,. When the input signal is negative, thc decreased output current ca1isc.s tlie collector side of resistor R, to become more positive and thc e~i i i t tcr side of resistor R3 to become more negative with respect to ground.

applications. In general, dc amplifiers a re used to amplify the output of transducers which produce quantitative information relative to heat, vibration, pressure, speed, and distance. Other applications include the output stages of series-type and shunt-type regulating circuits, cliop- per-type circuits, difrerential alnpli- fiers, and pulse amplifiers.

Dircct-coupled amplifiers are also used in chopper-type circuits to amplify low-level dc signals, a s illustrated by the block diagranl in Fig. 188. khe dc signal niodulatcs an ac carrikr wave, usually a square wave, and the modulated wave is then amplified to a convenient level. The series of amplified pulses can then be detected and integrated into the desired dc output signal.

CHOPPER INTEGRATOR

Fig. 188-Block rlirrgro~~r s l r o ~ ~ ~ i t r , ~ ooiorr of "c1topl)cr" cir-rrrir


Chopl~cr amplifiers consist of three basic sections. The first section converts the low-level input signal into a ~l~otlulated ac signal, the second scction amplifies this ac signal, and the third section demodulates the amplified signal.

The first section of a chopper amplifier is funtlamentally a continu-.,.. ously operated ON-OFF switch. Ideally, this switch would have zero'' ON resistance, infinite O F F resistance, zero shunt capacitance, and zero switching time. I t would also require no driving power and have infinite life. In actual practice, i t is possible to achieve satisfactory performance with a switch tha t does not have these ideal characteristics.

The two basic circuit configurations fo r chopping a r e the series chopper and the shunt chopper. The shunt chopper is the more popular of the two because it can be capaci- tivcly coupled to a n ac amplifier without the need fol. either a choke or a transfornier. The series chopper has the disadvantage that i t requires a dc return path fo r the input current. This path can be provided hy an additional resistor a t the expense of over-all circuit efficiency.

The basic series chopper circuit usinn a n nlOS tl-ansistor is shown in Fig. 189. This circuit has the characteristics of a simple L-pad attenuator in which the transistor is tlie variable series resistor. In the

AC CARRIER

Fig . 189-n(tsic set-ic.s clroppcr circrtif rrsir~g or1 MOS 11-ansisfor.

ON condition, the value of the dc return resistance Rs must be large compared to the load resistance RL to minimize resistive losses; Rr., in turn, nus st be large compared to the intrinsic drain resistance rd(ON) so

that the voltage VL across the load approaches the value of the dc input voltage VG. In the O F F condition, the dc return resistance RS must be small compared to rd(0FF). Because of these restrictions, the series chopper is seldom used except when the fixed resistance RS can be made variable by replacing i t with a shunt chopper arranged to be O F F when the series chopper is ON, and vice versa.

Fig. 190 shows a shunt chopper circuit using a n MOS transistor. In

INPUT k r@q!ouiT Fig. 190-Basic sltrrnf copper circltif rrsirrg

an MOS frarlsisror.

this circuit, the intrinsic drain resistance rd of the transistor must be small compared to the load resistance RI. in the ON condition, but niust be large compared to the fixed series resistance RD in the O F F condition. The requirement fo r rd(0N) to have a very small value is minimized if RL i s the high input impedance of a n MOS transistor amplifier stage. Because of their high ON-to-OFF resistance ratio, negligible gate-leakage currents, and low feedthrough capacitance, MOS transistors considerably improve the level of solid-state chopper per-, formance.

Differential amplifiers can be used to provide voltage regulation, or to compensate f o r fluctuations in cur- rerit due to signal, component, o r temperature variations. Typical differential-amplifier circuits, such a s those shown in Fig. 191, may also include an output stage which supplies current to tlie load resistor R, and the necessary number of direct- coupled cascaded stages to provide the required amount of gain fo r

126 R C A Transistor , Thyr istor , & Diode Manual

a given condition of line-voltage or load-current regulation. The reference-voltage source V1t is placed in one of t h e cascaded stages in such a manner tha t a n error or difference signal between VR and some portion of the output voltage V n is developed and amplified. Some form of temperature compensation is usually included to insure stabil- i ty of the direct-coupled amplifier.

MOS-transistor dc anlplifiers nlay take several difrerent forms, including single-ended input to single- ended output, differential input to singlc-ended output, and difrerential . input to differential output. Normally dc amplifiers require direct coupling of all stages (no coupling capacitors). In sonie vcrsions of dc amplifiers, this requirement is c i r c u n ~ v e ~ ~ t e d by conversion of the low- or zcro-frequency input signal into a modulated ac signal, atl~plification of this sin- nal by means of capacitor-coupled stages, and then demodulation of

the amplified signal to restore i t to the original dc form. The necessary modulation may be .accomplished by a number of different techniques, including electrically actuated mechanical switches, electronic switches, photo-optical switches, magnetic modulators, and diode bridge modulators. Input devices which function a s switches a r e generally referred t o a s "choppers" because, as described above, they divide the input signal into segments in the form of square waves or pulses having a n amplitude proportional to the amplitude of the input signal.

Single-ended dc amplifiers which do not employ "choppers" have a continuous ohmic current path between the input and the output a s the result of direct coupling of all, stages (i.e., the omission of all capacitive or inductive forms of coupling). I n this configuration, the steady-state voltage a t the output of one stage appears a t the input of the next stage. In a typical cascade arrangement using MOS field-effect transistors, the signal progresses from the drain of the first unit to the gate of the next and so on to the last stage, a s shown in Fig. 102. In this circuit configuration, the source terminal is generally placed a t a potential equal to o r greater than the drain-to-source voltage of the preceding stage. In the arrangement of F~E. 192, the gate i s a t a net zero voltage or is reverse-biased relative t o the source.

Although MOS transistors a re not optimized for direct-coupled applications, they can he used in such circuits because they have low gate leakage current (typically fractions of a picoampere), total input capacitance of about 5 picofarads, and an appreciable value of forward transconductance. In addition, tight production control liniits the spread of drain current between individual transistors to a variation of approximately two to one for a high degree of interchangeability.


Fix. 192-DC n~nplifier circltit in wShich n-clilnrr~iel deplerio~r-type MOS transistors are direct-corrpled by ctse of dc level shifting.

For a fixed value of supply volt- measurement of the forward-trans- age, there are only three ways to f e r characteristic a t different am- increase the stage voltage gain A in bient t em~era tures . R single-ended a n ~ ~ l i f i e c ( 7 ) use of a transistor having a higher ratio of gate-to-drain forward transconductance gr, to drain current In; (2) use of a higher value of load resistance It,. (if R,, is less than the common- source output resistance r,.); and (3) use of a transistor having a higher value of r... The load resistance 121. can only be increased t o the noint where the ~ r o d u c t of ID and 'R,. is equal t o approximatel$ one-half the supply voltage. In general, the ratio of transconductance to drain current increases as drain current is decreased by negative gate bias. A s a result, the s tage voltage gain may be increased and power consun~ption decreased a t the same time.

The increased voltage gain of a n hIOS transistor a t reduced values of drain current may be acconlpanied by a relatively large dr if t in the operating point if there a r e wide excursions in ambient temperature. Many field-effect transistors have a point on their fol-ward-transfer characteristic which is relatively insensitive to temperature variations. If this point does not coincide with the operating point which provides the desired voltage gain, a design compromise is required. As shown in Fig. 193, the zero-temperature- coefficient point may be identified by

GATE-TO-SOURCE VOLTAGE-V

Fig. 193-Forward-trcmsfer clraracterirtics of MOS transistor at 25'C and -30'C.

VOLTAGE-CONTROLLED ATTEN UATORS

Because the drain current-voltage characteristic of MOS transistors remains linear a t low drain-to-source voltages, these devices can be used a s low-distortion voltage-controlled attenuators. The principal advantages of MOS transistors in this application are negligible gate-power requirements and large dynamic range.

Fig. 194 shows drain resistance a s a function of gate-to-source voltage f o r , a typical n-channel depletion- type insulated-gate transistor. Tran- sistors having higher pinch-off voltages accept correspondingly greater peak signal-voltage swings before wave-sha~e distortion occurs.

RCA Transistor, Thyristor, & Diode Manual I Low-Frequency Amplification

Iiowever, the I~ighcr-pinch-off-voltage t ra~lsis tors require higher gate- voltage cxcursions to cover the resistance range from minimum to

.- 0 GATE-TO-SOURCE VOLTAGE (VGSI-V

Fig. 194-Draitr resislnrrce as a frrrrc/iott of gore volrnge for typical n-chantrel deplc-

liorr-rypc MOS rrafrsislor.

maximu~n. A typical n-channel n10S transistor produces total harmonic distortion of less than two per cent in a 100-millivolt 400-Hz sine wave. Fiz. 195 sl~oms an attenuator circuit using an BIOS transistor and the output signal of the circuit a s a function of gate-to-source voltage.

GATE VOLTAGE-V

Fig. IPS-Orrrprtl sigrral as a jrcrtc~iotr o f gnrc voltage for MOS rrartsisror irz circtril

sllolvrl.

Figs. 196 and 197 show two possible attenuator circuit configurations which usc MOS transistors a s voltage-variable resistors. The circuit in Fig. 196 is desirable fo r use at I~ igh signal levels bccausc a t such lcvcls the thermal noise of thc one- ~negohm series resistor docs not degrade the si~nal-to-noise ratio of t!~e system to an objectionable dcgrec.

This circuit i s a simple L-pad configuration in which the transistor serves a s the variable-resistive element in the low side of the attenuator. The maximum attenuation obtainable i s generally between 60 and 70 dB; minimum attenuation is 1 to 2 dB. This circuit must be followed by a high-impedance load such as a common-source amplifier stage.

r---- -

CONTROL VG J

Fig. 196-Attcrlrtalor circrri! i r r ~thiclr MOS trarrsistor scrtles as i?nrinble-reristive elc-

lticnr irr low side.

The circuit shown in Fig. 197 is the inverse of that in Fig. 19G; i.e., tho t.ransistor serves ns the variable- resistive element in the high side of the attenuator. Blaxinlum attenun- tion in this circuit is also between 60 and 70 dB; mininlum attenuation is between 1 and G dB. This circuit is

Fig. 197-Atrot~caror circrri! irr ~rlriclr AfOS rrarrsistor servcs as rcrriahle-resistive cle-

rnetrt it! /rig11 side.

usually followed by a low-impedance load such a s a con~mon-emitter bipolar transistor amplifier stage.

The following dcsign considera- t i o ~ ~ s a re important fo r effective use

of MOS field-effect transistors as linear attenuators:

( a ) The gate(s) must be adequately decoupled to prevent the introduction of unwanted signals.

(b ) The transistor attenuator must be inserted a t a point in the system where the signal level is a s high as tho transistor can accept without excessive distortion.

( c ) I n a c systems, the direct-current flow through the transistor must be minimized by the use of suitable blocl t in~ capacitors.

(d) I n ac systems, proper layout must be used to minimize s t ray shunt capacitance.

(e) I n ac systems, the effects of the capacitive elements of t h e transistor must be considered.

WIDE-BAND (VIDEO) AM PLlFlERS

In television camera chains a s well a s in ac voltmeters and vertical an~plifiers fo r oscilloscopes, i t is necessary for a transistor circuit to amplify signals ranging from very low frequencies (several hertz) to high frequencies (tens of megahertz) with a minimum of frequency and time-delay distortion. In response to

frequency limits of the amplifier a r e approached.

The need f o r such compensation is evident when many identical stages of amplification a r e employed. If ten cascaded stages a r e used, a variation of 0.3 dB per stage results in a total variation of 3 dB. In a n uncompensated amplifier, this total variation occurs two octaves ( a frequency ratio of four) prior to the half-power point. Because two octaves a r e lost f rom both the high and low frequencies, the bandwidth of ten cascaded uncompensated amplifier stages is only one-sixteenth t h a t of a single amplifier stage. Fig. 198 shows the amplitude response characteristics of various numbers of identical uncompensated amplifiers.

I n general, the output of a n amplifier may be represented by a current generator iUut and a load resistance RL, a s shown in Fig. 199(a). Because the signal current is shunted by various capacitances a t high frequencies, a s shown in Fig. 199(b), there is a loss in gain a t these frequencies. If a n inductor L is placed in series with the load resistor RL, a s shown in Fig. 199(c), a low-Q circuit is formed which somewhat suppresses the ca-

. - -

Fig. 198-Airlplitrrdc response characteris!ics of vnriorcs r~unlbers ( N ) o f iderrrtcal un- cortrper?sared amplifiers.

these demands, circuit compensation pacitive loading. This method of gain techniques have been developed to compensation, called shunt peaking, minimize the amplitude and time- can be very effective for improving delay variation a s the upper o r lower high-frequency response. Fig. 199

RCA Transistor, Thyristor, & Diode Manual 1 Low-Frequency Amplification 131

FREQUENCY-Hz 9

Fig . 199-Eqrti~~~lctlr circrri~s cr~rtl /r.cqrrrrrcy rcsporr5e of rtrrcorrrperrscrrt~d nrrtl slrrrtrf- ~)colterl ar~rplifiers.

shows the frequency response for the circuits shown in Figs. 199(a) , ( b ) , and (c) . If the inductor 1, shown in Fig. 199(c) is made self-resonant approximately one octave nl~ove the 3-dB frequency of the circuit of Fig. 199(11), the amplifier response is extended by about another 30 per cent.

If tlie s t ray capacitance C shown in Fig. 199(b) is hrolccn into two parts C' and C" and an inductor L, is placed b e t ~ ~ e c n them, a heavily damped form of series resonance may be e~iiployetl for furtllcr improve~nent. This form of compensation, called series p ra l i i~~g . is slio\vn in Fig. 200(a). If C' and C:" are

within a factor of two of each other, series peaking produces an appreciable improvement in frequency response a s compared to sliunt pcaliing. A more complex form of compensation embodying I)otli self-resonant sllunt pealcing antl series peaking is shown in Fig. 200(b).

The e rec t s of various I~igh-frequency compensation systems can be tlcmonstratetl hy consitlcration of an amplifier consisting of three itlenti- cal stages. If each of tlie three s t a ~ e s is down 3 dB a t 1 hII-Iz, antl if n total ~ a i n variation of plus 1 t l I 3 and minus 3 d B is allo\vetl, the bantl- width of the anlplifier is 0.5 nIHz without compensntion. Sllurlt peal<- in^ raises the I)nntlwitlth to 1.3 hIIIz. Self-resonant sliunt pealting raises it to 1.5 RIIlz. An infinitely co11ll)lic:ltetl system could raise it to 2 ilIIIz. If the distribution of cal)acitance p~r rn i t s it, series penliirlg alonc c:ln p~.ovidr a I~andnridth of nl)out f l l l l z , \rliilc a combination of sliunt atid scrics pcaltina call provitle a l):ln(l~vitlth of approximntcly 2.8 hIIIz. If the capacitance is gcrfectly tlistributed. and if an infinitely complex networlc

of sliunt and series peaking is employed, the ultimate capability is about 4 MHz.

The frequency response of a wide. band a~ilpliiicr is inllucnced greatly by variations in component values clue to temperature efiects, variation of transistor parameters with voltage and current (normal large-signal excursions), changes of s t ray capacitance due to relocated lead wires, or other variations. A change of 20 per cent in any of the critical parameters can cause a change of 0.7 dB in gain per stage over the last half-octave of the response for the most simple case of shunt peaking. As tlie bandwidth is extended by more complex pealcing, a circuit becomes substantially more critical. (Measurement probes generally alter circuit performance because of their capacitance; this effect should be considered during frequency-response measure- nients.)

In the design of wideband amplifiers using many stages of amplification, i t is necessary to consider time- delay variations a s well a s amplitude variation. When feedback capacitance is a major contributor to response limitation, the more complex compensating netmorlts niay produce severe ringing or even sustained oscillation. If feedbaclc capacitance is treated a s input capacitance produced by the hliller effect, the added input capacitance C,' caused by the feedback capacitor Cf is given by

where VG is the input-to-output voltage gain. The gain VG, however, has a phase angle that varies with frequency. The phase angle is 180 degrees a t low frequencies, but niay lead or lag this value a t high frequencies; the magnitude of VG then also varies. In the design of very wideband alilplifiers (20 MHz or more), the phase of the transconductance g,,, must be considered.

Fig. 201(a) shows three stages of n multistage wideband amplifier.

The resistors R1 merely provide a high-impedance bias path for the col- lectors of the transistors. The ac collector current of each transistor normally flows almost exclusively into 'the relatively low impedance ofrered by the base of the next stage through the coupling capacitor C1. The resistive network R1 and R, provides a stable dc bias fo r the transistor base.

The mid-frequency gain of each stage is approximately equal to the common-emitter current-transfer ratio (beta) of the transistor if the component values are properly chosen. The high-frequency response is limited primarily by the transistor gain-bandwidth product f ~ , the transistor feedback capacitance, and sonletimes the s t ray capacitance. The low-frequency response is limited primarily by the value of the coupling capacitor C1.

Fig. 201(b) illustrates the use of high-frequency shunt peaking and low-frequency peaking a t the expense of stage gain in the three stages of the wideband amplifier t o extend the high- and low-frequency response. The emitter resistors Re are made a s small a s possible, yet large enough to mask the variation of transconductance, and thus voltage gain, a s a function of signal- current variation. F o r very small ratios of peak ac collector current to dc collector current, this variation is not substantial. The resistors Re also partially mask the effect of the intrinsic base-lead resistance rb'.

The base-bias resistors R, of Fig. 201(a) a re split into two resistors R, and R, in Fig. 20l(b) , with R, well bypassed. The mid-frequency gain is then reduced to a value approximat- ing Rb divided by Ro. At this point, however, the high-frequency response is increased by the same factor. Shunt peaking is provided by L, and Cn for additional high-f requency improvement.

When the reactance of the bypass capacitor C3 is large compared to R., the low-frequency gain is increased


because the resistor no longer heavily gain caused by C,. When the re- shunts the transistor input. Selec- actance of Cs approaches R,, how- tion of the proper value for C3 ex- ever, the low-frequency peaking is actly offsets the loss of low-frequency no longer effective.

Fig. 201-(a) Ur~conlpe~rsated and (b ) cor?rperrsatcd versiorls o f tlrrec stages of n r~rrrlri- sragc nfidcbarld ar~rplifier.

RF Power Amplification

and Generation

ECENT significant in~provements R in the design and technology for high-frequency power transistors have resulted in the increasingly widespread application of transistors in the an~plification and generation of rf power. Previously, cost considerations and performance limitations restricted the use of high- frequency power transistors to only a limited nunlber of special circuits in which small size and light weight were the overriding requirements. As a result of the progress that has been made in design and pfocessing, today, high-frequency transistors are often used in place of low- and medium-power tubes in many new equipment designs fo r operation a t frequencies up to 2000 MHz. In addition to small size and light weight, other unique circuit advantages, such a s greater reliability and significant increases in over-all circuit efficiency and bxnd~vidth capability, have made possible this penetration of transistors into a very great nunlber of different high-frequency applications.

FEATURES OF RF POWER TRANSISTORS

The performance of an rf power transistor is critically dependent on the structure and geometry of the device. Such factors a s the length

of the emitter and base peripheries, the emitter-to-collector spacing (i.e.; base width), the length of the collector-base junction, and parasitic inductances and resistive losses in the transistor package significantly affect power output, frequency response, thermal resistance, stability, and other important performance characteristics. 7

-4 Power Output

In early transistors, power outputs were in the milliwatt region, and increased power capability could be achieved only a t the expense of frequency response. The power output of a transistor is limited by the current-handling capability and dissipation of the device. The maximum dc input power to a transistor is largely determined by the current- handling ability because the dc operating voltages of power transistors have been fairly well standardized a t either 28 volts for military systems or 12.6 volts f o r mobile applications.

The current-handling ability of any transistor is proportional to the length of the edge of the emitter, i.e., the emitter periphery. The base current results in a voltage drop tha t causes the portion of the emitter most remote from the base contact to be least forward-biased. Little o r

RCA Transistor, Thyristor, & Diode Manual I RF Power Amplification and Generation

no current, therefore, is injected from this region. This condition results even when the emitter strip is exceedingly narrow. In present transistors, the emitter is only 10,000 angstronls wide, hut the emitter rurrent is still limited by the total length of the emitter edge. The current-handling capability is approximately 1 milliampere per mil of emitter length; a transistor required to handle a current of up to 1 ampere, therefore, should have a n emitter periphery of 1 inch.

narrow rectangles to maximize the spreading of the heat in the silicon, which is a reasonably good conductor of heat (about 20 per cent of the conductivity of copper). In addition, power transistors a re usually mounted on berylliun~ oxide to provide fur ther spreading of the heat and electrical insulation of the devices from the chassis. Use of these techniques allows transistor dissipation of about 10" wat t s per square centimeter.

RF Power-Transistor Packages Frequency Response The package is an integral par t of

The frequency response of a transistor is inversely proportional to the square of the emitter-to-collector spacing and to the capacitance of the transistor. F o r a given hase width, therefore, the power-output/ frequency capability is detevmined l)y the length of emitter periphery tha t can be concentrated into a given area. One figure of merit of a power-transistor design is the ratio of emitter periphery to base area. The 2N3375 transistor has a ratio of 0.82 mil of enlitter edge per square ]nil of base area and can produce 4 watts of output power a t 400 MHz. The 2N5921 transistor in which the ratio of emitter periphery to base area is increased to 3.1 mils per square niil can produce 6 \v:~tts of output power a t 2 C H a . The 2N5921 pellet uses 180 emitters only 20,000 angstroms wicle, and has a base width of approxi~nately 1200 angstroms and a n over-all length of 40 mils. . <

Thermal Resistance The thermal resistance of a tran-

sistor is proportional to the length of the collector-base junction, i.e., the base periphery. For this reason the base regions ( the heat-generation nrea) of modern power t ransistors a r e made in the form of long,

an rf power transistor. A transistor package designed for use in rf power applications Should have good thermal properties and low parasitic rg- actance. Parasitic inductances and resistive losses of the package significantly affect circuit perfornlance characteristics, such a s power gain, bandwidth, and stability. The most critical parasitics a re the emitter- and base-lead inductanccs.. Fig. 202 shows several popular coniinercially available rf power-transistor packages, and Table I1 indicates the parasitic inductances of each type. The TO-GO and TO-39 packages were first used for devices such a s the 2N3375 and the 2N38FG. The I~ase and emitter parasitic inductances of these packages a re in the order of 3 nanohenries; this value of intluct- ance corresponds to a reactance of 7.5 ohms a t 400 MHz. If the emitter is grounded internally to a TO-60 package (as in the 2N501F), the emitter lead inductance ran be reduced to 0.G nanohenry. He~,n~et ically sealed, low-inductance radial-lead pacltages, such a s the IIF-19 package introduced by RCA, employ ceramic- to-metal seals and have good rf performance characteristics. The parasitic inductances can be reduced further by use of a hermetically sealed coaxial package, such a s the HF-11, used for the 2N5470. This package has parasitic inductances in order of 0.1 nanohenry.

JEDEC 10-39

JEDEC TO-60

HF-12 HF-11 Molded-Silicone Plastic Package (Isolated Coaxial Package Electrodes)

HF-19 Hermetic Strip-Line Type Ceramic-to-Metal Package

(Isolated Electrodes)

HF-21

Coaxial Package

Studless HF-19

Package

Fifi. 202-Cor~rti~ercially available rf power-fratisisror packages.

Table 11-Summary of Packaged-Transistor Inductances Inductance (nH)

I'nckage Emitter Base TO-39 (2N386G) 3 3 TO-GO (isolated emitter) (2N3375) 3 3 TO-60 (ground emitter) (2N5016) 0.6 2 Hermetic Strip-line (2N5919) 0.4 0.6 Coaxial case (2N5470) 0.1 0.1

136 , .. ." RCA Transistor, Thyristor, & Diode Manual , , . ,, . I '- I RF Power Amplification and Generation

I

DESIGN CONSIDERATIONS FOR RF POWER AMPLIFIERS

I n thc design of silicon-transistor rf power :tniplifiers for use in tr;uis- ~n i t t ing systems, several fundamental factors nus st be consitlered. As with any rf power amplific~., thc class of operation has an important bearing on the power output, lincar- ity, and operating efficiency. The modulation requirements of transistor sf power amplifiers differ slightly from those for tube amplifiers. The matching characteristics of input and output terminations significantly affect power output and frequency stability and, therefore, a r e particularly important considerations in the design of either transistor or vacuum-tube po\xrer anlplifiers. The selection of the proper tr:unsistor for a given circuit application is also a major consideration, and thc circuit dcsigner must realize the significance of the various t ~ n n - sistor parameters to makc a valid evaluation of different types.

by use of only a slight amount of forward bias in the transistor stage. In this class of service, care must be taken to avoid thermal runaway.

In a class C transistor stage, the collector conduction angle is less than 180 degrees. The gain of the class C stage is less than that of a class A or class B stage, but is entirely usable. In addition, in the class C stage, s tandl~y drain is virtually zero, and circuit efficiency is the highest of the three classes. Be- cause of the high effic.iency, low collector dissipation, and negligible standby drain, class C operation is the most con~nionly used mode in rf power transistor applic a t' lons.

For class C operation, the base- to-cmittcr junction of the transistor must be reverse-biased so that thc collector quiescent current is zero during zero-signal conditions. Fig..' 203 shows four methods t h a t may be used to reverse-bias a transistor stage.

Fig. 203(a) shows the use of a dc supply to establish the reverse bias. This method, although effective, re-

Class of Operation quires a separate supply, which may not bc available o r may bc difficult

The class of operation of an rf to obtain in many applications. I11 amplifier is determined by the circuit addition, the bypass elements re- pet,forinance required in the given quired for the separate sup1)ly in- applications. Class A power ampli- crease the circuit conlplcxity. fiers a re uscd when extremely good Figs. 203(b) and 203 (c) show linearity is recluircd. Although powcr methods in which revcrse bias is de- gain in this class of service is con- veloped by the flow of dc base cur- siderably higher than t h a t in class ren t through a resistance. In the B or class C service, the operating case shown in Fig. 203(b) , bias is efficiency of a class A power am- developed across the basc spreading plilier is usually only about 25 per resistance. The magnitude of this cent. Moreover, the standby drain biqs is small and uncontrollable be- and thernial dissipation of a class cause of the variation in r,,~,' among A stage arc high, and care must be , different transistors. A better ap- exercised to assure thermal stal~ility. I proach, shown in Fig. 203(c) , is to

In a])plications, such a s single-side--'. develop the bias across an external band transmitters, that require good resistor Rn. Although the bias level linearity, class I3 push-pull opera- is predictable and repeatnble, the tion is usually employed hecause the size of RII must be carefully choscn transistor dissipation and staiidl~y to avoid reduction of the collector- drain a rc usually much s~nnllcr :ind to-cmittcr breakdown voltage. operating cficiency is higher. Class The best reverse-bias method is B operation is characterized by a illustrated in Fig. 203(d). In this collector conduction angle of 180 method, self-bias is developed across degrees. This conduction is obtained a n emitter resistor RE. Because no

J- - - - ( c ) ( d l

Fig. 203-Mctltods for obtairlirlg clnss C reverse bias: (a) b.v rise o f fixed dc supply V,I~,; (b ) by rrsc of dc base crtrrorr rlrrorcgl~ the base spreading resistarrce r,,,,'; (c) by rrsc of tic base crtrrerzr rl~rorr.~h art cxtcrrtal base resistarrce Rn; (d) by use of self bias

cic~.eloped across a), errlitter resistor RE:.

external base resistance i s added, the collector-to-emitter breakdown voltage is not affected. An additional advantage of this approach is that stage current nlay be monitored by ~neasurement of the voltage drop across Rlr. This technique is very helpful in balancing the shared power in parallelcd stages. The bias resistor RI: niust Be bypassed to provide a very-low-impedance rf path to ground a t the operating frequency to prevent degeneration of stage gain. In practice, emitter bypassing is difficult and frequently requires the use, of a few capacitors in paral- el to reduce the series inductance in the capacitor leads and body. Al- ternatively, the lead-inductance problem nlay be solved by formation of a self-resonant series circuit between the capacitor and i ts leads a t the operating frequency. This method is extremely effective, but may restrict stage bandwidth.

Modulation (AM, FM, SSB) Amplitude modulation of the col-

lector supply of a transistor output stage does not result in full modulation. During down-modulation, a portion of the rf drive feeds through the transistor. Better modulation characteristics can be obtained by modulation of the supply to a t least the last two stages in the transmitter chain. On the downward modulation swing, drive from the preceding modulated stages is reduced, and less feed-through power in the output results. Flattening of the rf output during up-modulation i s reduced because of the increased drive from the modulated lower-level stages.

The modulated stages must be operated a t half their normal voltage levels to avoid high collector-voltage swings that may exceed transistor collector-to-emitter breakdown ratings. R F stability of the modulated stages should be checked for the


entire cxcursion of the modulating signal.

Amplitude modulation of transistor transmitters may also I)e obtained by modulation of the lo~ver- level stages and operation of the higher-level stages in a linear mode. The lower efliciencies and higher heat dissipation of the linear stages override any advantages tha t a r e derived fro111 the reduced audio-drive requirements; a s a result, this approach is not econo~nically ~~rac t ica l .

Frequency modulation involves a shift of carrier frequency o ~ d y . C a n i c r deviations a rc usually vcry small ant1 prcscnt no problcnls in amplifier bantl\vidth. For ex:unll~lc, maximunl carrier deviations in the 50-hINx and 150-MHz mol)ile 1)ands are only 5 ItIIz. Because therc is no amplitude variation, class C rf transistor stages have no p~~oblems handling frequency madul a t ' 1011.

Single-sideband (SSR) modulation requ~res that all stages after the modulator opcrate in a lincnr mode to avoid intermodulatio~l-distorLion products near thc carricr frequency. In many SSB applications, channel spacing is close, and cxcessivc distortion results in adjacent-ch:rnncl interference. Distortion is effectivcly reduced by class B operation of the rf stages, with close attention to biasing the transistor base-to-emit- t e r junction in a near-linear region.

Characterization of Large-Signal RF Power Transistors

The vxlucs of large-signal transistor parameters, such a s the S and Y paramctcrs, a re different fl.0111 those of small-signal transistors hc- cause (1) the values of transistor ])nrameters change with po\ver levels, and ( 2 ) the harmonic-frequency c~onlpot~cnts that exist in a large- signal rf power amplifier must bc considered in addition to the funtla- mental-frcqucncy sinusoidal coml)o- ncnt in a sniall-signal amplifier. R F po~vcr-trat~sistor characterislics are t~ormally sl~ccifietl for a given circuit in x specific application.

The design of rf power-amplifier circuits involves the determination of dynamic input and load itnped- ances. Before the input circuit is designcd, the input i~npedance a t the emitter-to-base terminals of the packaged transistor must be known a t the drive-power frequency. Re- fore the output circuit is designed, the load impedance prescnted to the collector terminal must be ltnown a t the fundamental frequency. These dynamic impedances a re difficult to calculate a t nlicrowave frequencies because transistor paralneters such a s St, and S:? vary considerably under largc-signal operation and also change with the power level. Snlall- signal equations that might serve a s useful guides for transistor design cannot be applied rigorously to large- signal circuits. Because large-signal . representation of rf power transistors has not yet been developed, transistor dynamic impedances are best determined experimentally with slotted-line or vector voltmeter nlea- s u r c ~ n n t techniques.

The systcnl used for dctcrmina- r- 1 tion of transistor im~edances under

&operating conditions 's shown in Fig. 204. This system consists of a well-padded power signal generator, a directional coupler (or reflecto~ne- t e r ) fo r monitoring the input reflected power, an input triple-stub tuner, an input low-i~npedance line section, the transistor holder (or test jig), a n output line section, a bias tee, an output triple-stub tuner, another directional coupler for monitoring the output waveform or frequency, and an output powcr metcr. For a given frequency and input power level, the input ant1 output

- tuners are adjusted f o r maximu111 I)o\ver output and nlinimum input re- flccted power. Once the system has been properly tuncd, the iml)etlance across terminals 1-1 (with the transistor disconnected) is mcasurcd a t the same frequency in a slotted-line set-up or with the vector voltnleter. The conjugate of this impedance is

RF Power Amplification and Generation

pL+-] GENERATOR

I I DIRECT. TRIPLE-

PAD COUPLER STUB TUNER

POWER

22 LINE BlAS TRIPLE-

SECTION STUB DIRECT. POWER

TEE COUPLER METER 1 0 2 T U N E R

Fig. 204-Set-lip for rrieasitrerrlent of rf trartsistor dyr~arnic irrtpedat~ces.

the dynamic input impedance of the transistor. Similarly, the impedance across terminals 2-2 (with the transistor disconnected) is the collector- load impedance presented to the transistor collector. Such measurements a re performed a t each frequency and power level. I t should be noted tha t the circuit arrangement of Fig. 204 is also useful fo r testing the performance of the transistor. Thus, power output, power gain, and efficiency a r e readily de te rmined9

RF Ampiifier Circuit Design When the dynamic input imped-

ance and the load impedance of a packaged transistor have been established, either from direct measurements a s described previously, o r from the manufacturer's data, the input and output matching circuits can be properly designed.

Output-Circuit Design-When the dc sul)ply voltage and power output are specified, the circuit designer must determine the load for the collector circuit [RI. = (VrE)V2P,,]. Because an rf power amplifier is usually designed to amplify a specific frequency or band of frequencies, tuncd circuits a re nornlally used as coupling networks. The choice of the output tuned circuit must be made with due regard to proper load

matching and good tuned-circuit efficiency.

As a result of the large dynamic voltage and current swings in a class C rf power amplifier, the collector current contains a large amount of harmonics. This effect is caused primarily by the nonlinearity in the transfer characteristics of the transistor. The tuned coupling networks selected must offer a relatively high impedance to these harmonic currents and a low impedance to the fundamental current.

Class C rf power amplifiers a r e reverse-biased beyond collector-current cutoff; harmonic currents a r e generated in the collector which a r e comparable in amplitude to the fundamental component. However, if the impedance of the tuned circuit is sufficiently high a t the harmonic frequencies, the amplitude of the harmonic currents is reduced and the contribution of these harmonic currents to the average current flowing in' the collector is minimized. The collector power dissipation is therefore reduced, and the collector-circuit output efficiency is increased.

Figs. 205 and 206 illustrate the use of parallel tuned circuits to couple the load to the collector circuit. The collector electrode of the transistor i s tapped down on the output coil. Capacitor CI provides tuning


f o r tlic fundamenta l frequency, :1nd cap:lcitor C1 provides load matching of Rl, t o t he tuned circuit. The t rans- formed 11,. across t he ent i re t u~ r r t l circuit is s t cp l~cd down t o m:~tch t h e collector by tlie proper t u rns r :~ t io of i i~ductance tlic coil I,, L,. i s If chosen the value propclrly of and the & t he 1)ortion of t he output-coil in- +Vce 1

ductancc l)ct\vcen the collector and - - arourid is s~~f l ic~icnt lv higli. tlic har- (b)

k o n i c portion of t11eco11ict"r c u ~ . r c n t FOR N : I TURN RATIO in tlie tuned circuit is sniall. There- fo1.e. the contribution of the hnr- (1) R~ = $ (FOR CLASS C I monic current t o t h e dc component of cui.rcnt in t he circuit is n~iniinizcd.

N ~ R (2) XLl = +

Tlie use of a tapped-do\vn colinec- N ~ R ~ O L RL tion of t he collector to t he coil main- (31 xcl = - [I- -1 tnins t he londed Q of t he circuit and nlinimizes variation in t he 1);uitl- width of the ou tpu t cii.cuit with changes in the o u t l ~ u t ca1)acitnncc of t l ~ c t~.nnsistoi..

Altl iol~gh the circuits sllo\vn i l l

F ins . 206 : ~ n d 206 provitlc coul~li i ig of t hc lonil to t he collcctot. cii.cuit wit11 rood hnrmonic-cu~,.cnt sul)l~l.cs- sion, the tunetl-circuit 1 1 e t ~ o 1 . l i ~ 11avc a scrio'us limit:~tion a t very liigli fie- clurncies. Recause of the poor coeflicicnt of coupling in coils a t v e ~ y

- + Vce

FOR N : 1 TURN RATIO

vce2 (11 Rc = ---- (FOR CLASS C)

2 Po Rc ' - %

(2) XLl = - - OL QL

Fi.?. 206-Trrrrcd-circrrit orrlprrt corr~~lirrn rrrc~tlrorl orrtl tic.\i,<r~r c,c/rrrrtiorr.s i r r 11.Iric11 0rrtl)rrt tc> rill, Io(l1i i , ~ Ollr(rirl(~(1 jrorrr (1

crrprrcitii~e ~'olrcr~c' tli~,iiicr-.

high f~,equcncies, t he t a p position is usually established empirically so t h a t proper collector loading is achieved. Fig. 207 sho\vs sevcr:ll suitable ou tpu t coul,liiig ntxt\\orlts t h a t provide t h e requircd collector loading and also supl)rcss t he cir- culation of collector harmonic currents. These networks a r e not dc- pendent upon coupling coeficicnt f o r load-inlpcdance transforniation.

The collector ou tpu t capacitance f o r t he networks shown in Fig . 207 i s included in tlic design equations. Tlie collector output ca l~aci tance of a t rans is tor var ies considcral)ly with t he large dynamic swing of tlie collector-to-enlitter voltage ant1 is dependent upon both the collcctor sull-

'ply voltage and tlic po\vcr output . Input-Circuit 1)csign-The input

ci~.cuit of i i ~ o s t t rans is tors c:u~ I)e rcpresentcd by a resistor r~.,.' in scries with a c:~p:~citor C,,,. The input network mus t t une ou t t he capacitance CI,, and provide a purely rcsis- tive load t o t he collector of t he dr iver stage. Fig. 208 shows several nctworlts capable of coupling tlic base

RF Power Amplification and Generation 141

FOR Rl< R2 ( 0 )

RFC

"CE

- - - ( d l - -

- - - V~~ FILTER

L E T C 2 ~ " 2 COUTiR~'R2; f1'LOW FREO. CUTOFFif2:HI-FREO. CUTOFF

(1) ( f ~ - f I ) = ~ 2 a C o R ~ ( 2 ) L2'LIK - R~ l r I f 2 - f1 )

Fig. 207-Addirior~al rronsistor orctprrt-cocrpling ~~etworks ir~clrtdir~g transistor outpuf

I capacitar~ce.

RCA Transistor Manual

FOR XLI >>Xci: RI >R2'rbb1

FOR XC,>> XC;: RI >R2' rbb'

+VCE FOR R > Rp ; R2 = r b b 0 ; X L ~ " X ~ ~ "

I RF Power Amplification and Generation 143

and C-. The circuits shown in Fir. 208(a\

to the output of thc driver stage and tuning out the input capacitance CI,,. In the event tha t the transistor uscud has an intluctivc input, the reactance X,., is made equal to zero, and the base inductance is included a s part of inductor LI for networks such a s that shown in Fig. 208(a) and is included a s part of La for networks of the type shown in Fig. 208 ( c ) . In Fig. 208 ( a ) , the input circuit is formed by the T network consisting of CI, C.., nnd LI. If the value of the inductance L, is chosen so that its reactance is much greater than that of CI,,, series tuning of the base- to-emitter circuit is obtained by LI and the parallel combination of C? and (C, + C , , ) . Capacitors CI and C, 1)l.ovide the impedance matching of the resultant input resistance r1.2 to

and 208(b) reitlire the coiiector' of

,

the driving transistor to be shunt- fvd by a high-impedance rf choke. Fig. 208(c) shows a coupling network that eliminates the need for a ~ l l ~ l i ~ . In this circuit, the collector of the driving transistor is parallel tuned, and the base-to-emitter junction of the output transistor is series tuned. Fig. 200 shows several other forms of coupling networks that can he used in rf power-a~nplifier designs.

Line-section ~natching networks- 111 most micro~vave circuit applications, cithcr air-line, strip-line, or lumped-elelne~~t circuit arrangements a rc used; some useful circuit design tcchniqucs are discussed below.

Eighth-wave line sections: One of the properties of an cighth-wave sec-

the collector of the driving stage. Fig. 208(b) shows a T network in which the location of L, and C2 is chosen so that the reactance of the capacitor is much greater than khat of CI,; C1 can then be used to step. up r~.,,' to a n appropriate value across L,. The resultant parallel resistance across L, is transformed to the required collector load value by capacitors C, and C,,. Parallel resonance of the circuit is obtained by LI and

tion is that i t has a real input impedance when i t is terminated in a reactive impedance having a magnitude equal to Z,,. Therefore, fo r an eighth-wave line section, ZI. is real if the following condition is met:

I the parallel conlbination (CI + C..)

where R1. and XI. a re the real and imaginary parts of the colllplex impedance ZI.. The real impedance Zt. can be determined from a Smith chart of the following relation:

Eighth-wave transformers are use- fu l for microwave power transistor matching, a s shown in Fig. 210, because the small complex impedances of these devices can be matched directly, without the need for tuning- out mechanisms. In a typical power- amplifier circuit, the device input impedance R + jX ohms is the ter- minating impedance Z,, of the eighth- wave line section. If the characteristic impedance of the line Z,U is made equal to the magnitude of ZI., then the input impedance ZE of this line is a real impedance. Matching to the o u t ~ u t is accon~~l i shed in a similar manner.

The real impedance of an eighth- wave section of uniform line is thus predetermined by the complex ter- minating impedance. Therefore, i t is necessary to use additional trans- formations in cascade to match to a real impedance which is different from this predetermined real impedance.

Quarter-wave line sections: Quar- ter-wave lines a r e also useful a s impedance transformers between real impedances. If quarter-wave transformers are used to match a real impedance to an active device, a s shown in Fig. 211, the reactive component of the complex impedance (the admittance) of the active device must be tuned out. For example, in the input circuit of a power-transistor amplifier circuit, the quarter-


Fig. 209-Orlrer. sltitnblc rf-rrrrr~~liJi~~r ~.urrldir~,q rrer1~~or1i.s for rrrasirrrrrrrr polr*rr trnrl~Jer.


wavc transfor~ncr lnatchcs the resistive component of the complex admittance of the device. An external capacitance C., or a stub provides the necessary susceptance needed to cancel the reactive component of the device. In the output portion of the circuit, a stub or a lumped element a t the collector is used to bring the ilnpedance to a real value and then to a quarter-wave line t h a t goes to the actual load.

Direct transformation between the transistor (complex i~npedance) and a given source or load (real resistance) is also possible. The characteristic impedance Z,, and lenrth I VCC - of the transnlission line required to ~ i ~ . 211-Qttarter-wave trartsforr,lers for provide direct transformation from rJ po~ver-rrarrsisror arrrplifiers. a pure resistance Rt to a n irnnedance z,-= R, + jX, can be determined by use of the following equations:

If the impedance Z2 is a resistance (i.e., X, = O),the expression for Z. reduces to the quarter-wave transformer equation, and 1 = A/4.

MOBILE RADIO In the United States, three fre-

quency bands have been assigned to two-way mobile radio communications by the Federal Communications Commission. These frequency bands a re 25 to 50 MHz, 148 to 174 MHz, and 450 to 470 MHz. .The low-frequency band for overseas mobile con~lnunications is 66 to 88 MHz.

Frequency modulation (FM) is practiced in mobile radio communications in the United States and most overseas countries. The modulation is achieved by phase-modulation of the oscillator frecluencies (usually the 12th or 18th'subkultiple of the operating frequency). In vhf bands, the frequency deviation is e 5 ltHz and channel spacing is 25 kHz. In uhf bands, a t present, the modulation deviation is 2 1 5 kHz and

- channel spacing is 50 kHz. In the 1:: United Kingdom, AM a s well a s F M is used in mobile con~munications.

Typical mobile-transmitter power- output levels in the United States a re 50 watts in the 50-MHz band, 30 watts in the 174-MHz band, and

- - 25 watts in the 470-MHz band. Some of the transmitters used in the United States have power-output

Fig. ?IO-Eigl~r/r-~rn~.c trn,~sforr~ter irl a ratings a s high a s 100 watts. Over- typict11 rf / > o ~ t ~ ~ , r - ( r r r ~ / ) l i J i i ~ r circ~tit . seas, power-output requirements a r e


much more moderate; t he most common power-output levels a r e in t h e 10-\~:1tt range.

All-solid-state mobile t r ansmi t t e r s can be divided into two 1)asic types: t r ansmi t t e r s t h a t opera te f rom 24- to-28-volt collector supply voltages, obtained f rom dc-to-dc convcrtcrs, and t r ansmi t t e r s t h a t opera te t i i r ~ c t - ly f r o m the 12-volt electrical sys tem of a vet~icle.

Both types have atlvantagcs and disadvantages. T h e advaritagcs of 24- t o 28-volt operation inclutle h i ~ h - el. poxver ga ins pe r s tage , good t r an - sient suppression, and fa i r ly simple cu r r en t and voltage limiting. T h e disadvantages a r e t he additional cost of dc-to-dc converters and t h e some- wha t higher power consumption and increased size of t h e radio. Ilircct operation f rom a 12-volt systetn pel.- mi ts savings in cost and size, a s well a s higher efliciency. Because 12-volt operation produces less ~ a i n per s tage , however, additional rf s t ages a r e often needed. Trans ient supl)res- sion and voltage a n d cu r r en t l imiting a r e also somewhat more difficult.

Recause of t he two discrete volt- afic ranges used f o r mobile ratlios, t he t rans is tor mus t be designed spc- cifically f o r e i ther 24-to-28-volt 01)- eration o r 12-volt operation. Devices designed f o r 24-to-28-volt ope~.a t ion have su1)stantially h igher collector- I~rcaltdown voltages. In addition, al l elements a r e usually isolated f rom the case to permi t access t o t he ctnitter.

F ig . 212(a) shows a 175-RIIIz am- plilier chain t h a t opera tes directly f rom a 12-volt dc supply. An amplifier chain of th is t ype can deliver 12 wa t t s of outl)ut power with a n input of 125 milli\vntts and h a s a n over-all . elficiency of GO per cent. The chain consists of three cascaded s tages t h a t provide power outputs of 1. 4, anrl 12 wat ts , respectively. F o r np- ~) l ica t ions such a s base s ta t ions in which higher ou tpu t power le \~els a1.e iequired, three overlay power t r an - sisto1.s can be operated in ~ ,ara l le l a s shown in Fig. 2 1 2 ( b ) . 111 th is a r - rmigcnient, the t rans is tors can supply a s much :IS 35 wa t t s a t 175 illIIz

when driven f rom t h e three- s tage amplifier chain shown in Fig . 220 (a ) . Fig. 213 shows a 25-watt, 175-BIHz amplifier chain t h a t uses 2N5995 and 2N5996 stripline-package t r an- sistors. F ig . 214 shows a 6-watt , 470- MIIZ amplifier chain t h a t ernl)loys 2N2914 and 2N2915 transistors.

The requirements of r f power t rans is tors operated in n~ol?ile-radio applications a r e extremely severe. The t rans is tors lllust withstand the load-mismatch conditions created by objects n e a r t h e t ransmit t ing antenna o r by a break in t h e t rans- mission line anywhere between zero and one-half wavelength. Under such conditions, t h e t rans is tors mus t handle no t only the increased dissipation, but also sudden enerny surges t h a t can destroy the111 in jus t R f ew microseconds. The de- ,, velopnlent of t r ansmi t t e r s t h a t a r e i ~ n m u n e to these fa i lures i s a resul t of a joint effort hetween solid-state-device and mobile-radio nlanufacturers. To avoid excessive junction tempera tures , t he equip- men t manufacturer m u s t select t rans is tors of sufliciently low thermal resistance. If a t rans is tor laclts enough dissipation capal~i l i ty , two should he used-even though one could deliver t he required rf ou tpu t power. The use of adequ;~tc ly sized heat s inks i s essential to protect devices operated under high-ambient- 1enil)eraturc conditions. Cur.rent l imiting should also be employed t o prevent excessive rise in junction tempera ture under mi sn~a tched load conditions. A s a n added p ~ ~ e c a u t i o n , a t he rmos ta t can be mounted or) the hea t sink t o reduce the t r ansmi t t e r power in t he event t h a t t he tcmpera- t u re becomes excessive. - The protection of t he devices f rom "instantaneous" failure is more difficult because the t ime response of current o r voltage l imitcrs is no t f a s t enough. Fig. 215 sho\\,s a circuit which h a s a suficiently f a s t I,csponse t ime to protect t he power t rans is tors f rom "instantaneot~s" fa i lures t h a t resul t f rom mismatched-load conditions. The circuit ope r ;~ t c s on the 1)rinciple of reflected power. Under


RCA 4020,2

RCA 1 40282 I

Fig. 212-175-MHz trarlsisfor power arnplificr: (a) 3-stage brput a~nplifier; (b) ourpur stage.


Fig. 213-Tlrrc~c-s1rr,rc, '5-1t.flll, 175-MHz ar~rplifirr cl~airr.

Z L - 5 0 OHMS

F

matchcd load conditions, there is no output from the VSWIL detector. The control amplifier is saturated, ant1 the gain-controlled rf amplifier operates a t maximnn~ gain. The po\ver amplifier, thercforc, is opcr- atcd a t maximum power output. If a mismatch occurs, a negative volt-

age from the VSIVR bridge brings the control amplifier out of saturation, which, in turn, reduces the gain in the gain-controlled rf amplifier. Gain is reduced because the base of the rf amplifier becomes ,nore negative with respect to the emitter, and because the unsaturated control


/ ~i~~~~ system ~ ~ ~ , i ~ l ~ ~ ~ f o r low distortion, and emitter ballast resistance for stability and de- . ~

generation. In high-powei amplifiers, transistor junctions experience

ur wide excursions in temperature and a means must be provided to sense the collector-junction temperature so tha t a n external circuit can be used to provide bias compensation t o prevent a n excessive shift in operating point and to avoid catastrophic device failure a s a result of thermal runaway.

~dvantages of - SSB Transmission ,"', &- - c'-

, .- .I i'!

c?-- '. 7 Fir. ?l5-Loncl-rrrisr~ra/cIt profecriorl Single-sideband communication sys- I\:

circlrit. tems have many advantages over AM :: amplifier has a degenerative effect and F M systems. In on the rf amplifier. With the reduc- reliability of transmission as well tion in the gain of the gain-controlled a s power conservation a r e of prime rf amplifier, the drive to the power concern, SSB transmitters a r e usual- amplifier is decreased to safe levels. ly e m p l o y e g T h e main advantages Once the load mismatch is removed, of SSB operation include reduced the sys te~n returns instantaneously power consumption for effective to normal operating fonditioqs. transmission, reduced channel width

i . , , . . , . , ... , . ,,;%,., . , ,,- f . ? . >..A> ,? t o permit more transmitters to be

' SING LE-s I DEBAN D ! , / c n , operated within a ! ? TRANSMITTERS f ' :"""' range, and improved

ratio. The increase in comnlunication In a

traWc, especially in the hf and vhf modulated AM transmitter, two- i ranges, necessitates more effective thirds of the total power delivered use of the frequency spectrum so by the power amplifier i s a t the car- that more channels can he assigned rier frequency, and contributes noth- to a given spectrum. I t has been ing to the transmission of intelli- shown that one of the more eficient gence. The remaining third of the methods of communication is through total radiated power is distributed the use of single-sideband (SSB) equally between the two sidebands. techniques. In the past, the power- Because both sidebands a re identical a~nplifier stages of an SSB trans- in intelligence content, the transmis- mitter invariably employed tubes be- sion of one sideband would be suf- cause of the lack of suitable high- ficient. In AM, therefore, only one- frequency power transistors. Recent sixth of the total r f power is fully transistor developments, however, utilized. In an SSB system, no power have made i t feasible and practical is transmitted in the suppressed side- to design and construct all-solid- band, and power in the carrier is state single-sideband equipment for greatly reduced or eliminated; a s a both portable and vehicular appli- result, the dc power requirement is cations. substantially reduced. In other

Unlilte most co~n~nercially avail- fo r the same dc input power, t h

:1nd shoultl have a flat beta curve AM transmitter.

able rf power transistors, which a re peak useful output power of a n SSB normally designed pri~narily for transmitter, in which the carrier i s class C oper:ltion, an SSl3 transistor completely suppressed i s theoretical- is designed for linear applications ly six times t h a t of a conventional


Another advantage of SSB trans- tude under pealc power condition, the mission is that elimination of one average powcr of one tone of a two- sideband reduces the channel width tone signal is one-fourth the single- rcquired for transmission to 011c-1i:llf . frequency power. For two tones, con- that required for Ahl transmission. / versely, the PEP rating of a single- Theoretically, therefore, two SSB- sideband system is two times the transniitters can he operated within average power rating. a frequency spectrum tha t is normally required for one ARI trans- ~n i t t e r . - bf = I TO 2 KHZ

111 a single-sideband system, the _I

signal-to-noise power ratio is eight a r t inirs as great a s that of a fully a

nlodulated double-sideband system l-Jl.- f~ I 2 for the same peak power. FREQUENCY

I Linearity Test For__a~i-a~~~liff ier to he linear, a

,? relationship must exist such tha t the ou t l~u t voltage is directly proportional to the input voltage for all s i ~ n a l amplitudes. Because a single- frequency signal in a perfectly linear single-sidcband system remains unchanged a t all points in the sign:ll path, the signal cannot be distin- guished from a cw signal-or from a n unmodulated carrier of an A M transmitter. To measure the linearity of an amplifier, i t is necessn- r y to use a signal that varies in amplitude. In the method cotilmon- ly used to measure nonlinear distortion, two sine-wave voltages of different frequencies are applied to the anil)lifier input simultaneously, and thc s u ~ n , difference, and various conlhination frequencies that are produced by nonlinearities of the amplifier a re observed. A frequency difference of 1 to 2 ltIIz is used widely for this purpose.,'A typ- icla two-tone signal without distortion. as displayed on a spectrum analyzer, is shown in Fig. Zl(i.)The resultant signal envelope varies'corl- tinunusly between zeio and niaxi- muln a t an audio-frequency rnte. li'l~llen the signals are in ph;lse, tlie peak of the two-frequency envelope is limited by the voltage and current ratings of the transistor to tlie smile power rating a s that for the single-frequency case. Because the amplitude of each two-tone frequcncy is equal to one-half thc cw ampli-

Fig. 216-Freqrcerrcy spectrctru for a Iypicnl tw.o-torrc si,q~rol ~virhorrt disrorriorr.

Intermodulation Distortion Nonlinearities in an amplifier

generate intermodulation (IM) dis- ., tortion. The important I M products are those close to thc desired output frequency, which occur within the pass band and cannot be filtered out by normal tuned circuits. If f t and fl are the two desired output signals, third-order IRI products take the form of 2fl - f, and 2f2 - f,. The matching third-order terms are 2fl + f2 and 2f, 4- f,, but these m a t c h i n ~ terms correspond to frequencies near the third h:umonic output of the amplifier and are ~ r e a t l y attenuated by tuned circuits. I t is important to note that only odd-order distortion products appear near the fundamental frequency. The frequency spectrum shown in Fig. 217 illustrates the frequency relationship of some tlistortion

FUNDAMENTAL FREQUENCIES

THIRD-ORDER DISTORTION

FREOUENCY

Fig. 217-Frcqricrrcy spcctrrrur slron~irtg 111e freqrrerrcy rclotiorrslri~t o f .corrre di.rtor-- tioft prodrrcfs to two rc.rr si.q~rols f , N I I ~ fn.

/ RF Power Amplification and Generation

products to the test signals f , and f,. All such products are either in the difference-frequency region or in the harmonic regions of thc original frequencies. Tuned circuits or filters following the nonlinear elements can effectively remove all products generated by the even- order conlponents of curvature. Therefore, the second-order component tha t produces the second harmonic does not produce any distortion in a narrow-band SSB linear amplifier. This factor explains why class AB and class B rf amplifiers can be used a s linear amplifiers in SSB equipment even through the collector-current pulses contain large amounts of second-harmonic current. In a wideband linear application, however, i t is possible fo r harmonics of the operating frequency to occur within the pass band of the output circuit. Biasing the output transistor further into class AB can greatly reduce the undesired harmonics. Operation of two transistors in the push-pull configuration can also result in can- cellation of even harmnics in the output.

20

30

400 20 40 60 80 100 PEAK ENVELOPE POWER OUTPUT-W

Fiq. 218-Tj,picol irrterrrtodrrlorion di.rtor- 1 iior~ ill RCA-4 /75 nurrsixtor nt voriorrs orttprrt polrver levels.

The signal-to-distortion ratio (in dB) is the ratio of the amplitude of one test frequency to the amplitude of the strongest distortion product. A signal-to-distortio~l specification of -30 dB means tha t no distortion product will exceed this value for a two-tone signal level

up to the PEP rating of the amplifier. A typical presentation of IM distortion for a 40675 transistor a t various output-power levels is shown in Fig. 218.

Transistor Requirements Most high-frequency power tran-

sistors a re designed for class C operation. Forward biasing of such devices fo r class AB operation places them in a region where second breakdown may occur. The susceptibility of a transistor to second breakdown is frequency-dependent. Experimental results indicate tha t the higher the frequency response_ : of a transisby, the-more se-verg-the second-breakdown limitation be-' comes. F o r an rf power transistor, the second-breakdown energy level a t high voltage (greater than 20 volts) becomes a small fraction of its rated maximum power dissipation. This behavior is one of the reasons that vacuum tubes have traditionally been used in single- sideband applications.

A power transistor designed especially for use a s a linear amplifier is required to perform satisfactorily when forward-biased for class AB operation, a s well a s to exhibit the desired high-frequency response. The ability of the transistor to with- s t a n q o n d breakdown is improved y subdividing the emitter into many small sites and resis- tively ballasting the individual sites. The RCA 2N5070 and 40675 transistors a re designed specifically for linear-amplifier service in SSB applications. Current-limiting resistors are placed in series with each emitter site between the metalizing and the emitter-to-base junction.

Bias Control Operation of the transistor in a

class AB amplifier to improve linearity requires the use of a positive base voltage for a n n-p-n silicon transistor. The magnitude of the positive voltage must be large enough to bias the transistor to a

152 RCA Tran isistor, Tliyristor, & Diode Manual

point slightly beyond the tl~rcshold of collector-current conductiori. The class AD l ~ i a s condition nir~st be ~naintained over a wide tcn~licratme i,alife to prevcnt an increase in itll- ing current to the level a t which the transistor can bc dcstroycd a s n result of tlicrnial runn\v;l;\- :111d to niininiize distortion tha t results

. from a shift in the quiescent point. I t is pa~~ticular ly dillicult to main-

tain the bins current of a transistor high-powcr class AB mnlrlifier a t a constant level. A s tlie drive in- crcaseq, the dissipation increases and tlie junction temperature ~ i s c s . If the conventional biasing technique is employed (an ac-bypassed eniitter resistor and a constant rolt- age supply to tlie base), the vnry- ing emitter current tha t results from the varyine drive chnnecs the voltage drop across the emitter rcsistor and causes the I~ ias to shift with drive. If a constant-current base-bias supply is used, the drive po\ver is rectified and tlie bias point is changcd.

The prol)leni of maintaing n st:ihle rluic,sccnt current is caused I)y a rcduction in thc V,,,; of the transistor when the temperature rises. The I~asc-to-emitter voltage dcn~cnses a t a rate of approximately 2 millivolts per "C lnise in ten~per:iture. Unless this condition is colnpcnsntcd for (i.e., bias voltage made to vary ac- cording to the V I ~ E decre:ise), the tr:lnsistor is destroyed by the tllernlal ebects.

Bias-point control fo r the 40675 SSB transistor i s accomplishetl 11y use of a diode placed next to tlie transistor pellet in the same pnrlr- age. The cathode of the diotle is connected internally to the emitter lead. The anode of the diode is connected to a fourth tern~inal, a s shown in Fig. 219. The diode is fern-ard-1,iased between 1 to 5 niilli- amperes to provide a formartl-volt- axe drop that is temperaturc-sensitive. At such a low cur~<ent , the diode operates in the low-conduct- nnce region where it does not provide the stiff voltage necessary for the transistor bias. In this case. the

Fi,?. 219-Pnckrrfe ortrlirre for tile RCA - 40675 SSR trartsirror slrotc'irr,y irrterrr~rl- package diode rrserf jor rrorrsistor bias-

poirtr co~rrrol. t

diode acts merely a s a thermonieter; an external amplifier must he used for current amplification. Compcn- sation is achieved because the diode has approximately the same temperature coeficient fo r its forward- voltage drop a s does tlie base- enlitter junction of the transistor. Good tracking is obtained by n~ounting tlic diode and transistor pellets in the same case in very close prox- iinity to minimize any thcrnmal time lag. Tenlperature coeflicicnt depends, to a large extent, upon the opcrat- ing current. If the diode current can be adjusted so that it is approximately equal to tlie base current, good compensation can be achieved. The bloclc diagram of a current nmplifier that uses a low- contluctance diode is shown in Fig. 220.

The schematic diagram of the cur- - rent (bias-control) anlplificr is shown in Fig. 221. The current nnl- plifier eniploys a dc differential amplifier. The output voltage is the bias source for the power transistor. The use of a differential amplifier makes the entire amplifier relatively insensitive to temperature variations. Two additional s tages a re used for current aniplification with ncga- tive fectlback for stability.


I I I

BIAS RFC I CONTROL - I

AMPLIFIER I I -

LOW-CONDUCTANCE -I-- COMPENSATING DIODE

Fig. 220-Block diagrarrr of 30-MHz amplifier that uses a low-cot~ductartce diode for retIlpera1ure cor?rpertsalion.

+28 V A

TO TRANSISTOR BASE

TO ANODE OF DIODE

I- - Fig. 221-Birrs-cortrr-ol stages for lirrear 30-MHz ar~rplificr with lerriperarrtre-corrtpensafir~g

circrlir.

Transistor collector-bias current pensation occurs when diode current can be adjusted by varying the is greater than the base current. Fig. potentiometer connected in series 222(a) shows collector quiescent with the temperature-compensat- current, intially biased a t 10 milli- ing diode. The diode current es- amperes, a s a function of case tem- tablished by R,,,, . determines the perature. With compensation, the degree of compensation. Overcom- transistor is thermally stable even

154 RCA Transistor, Thyristor, & Diode Mariual

fo r case t r ~ n p c ~ . : ~ t r ~ r c a s h i ~ l i :IS

150°C. \\Tithout co~iipcnsntion, Iio\v- ever, the t rans is tor tends toward t l ier~iinl ruri:r\v:iy a t a cnsc ternl)c5~.:~- tu re of approximately 75°C.

d

$ I00 rz LT

2 s o w

60

.I= g 40 0 I- 20 Z W U

- '20 40 60 80 100 120 140 160 3 CASE TEMPERATURE-'C 0

( 0 )

- -

CASE TEMPERATURE -OC I- u a (b) 5

Fig. 222-Prrjor~rrorrcc cl~nrncterislics /or. rlrc 30-AlHi otr~pli/ier: ( a ) co l l~~c tor crrrrcrrt or o frrrrcliorr o f cose lerrrperatrrre 11.illi crfld ~~~i t l ror t t rorrpernrrrre corrrpc~rrsatiori; ( h ) orrtprrt pou,cr arid irrter~rrodrrlotio~~ distor-

rinrl ns o jrrrrclior~ 01 cose rerrrpernrrtrc.

Because both input and output a r e isolated through rf choltes, the external circuit provides conlpensa- tion ~ v i t h o u t degrading the rf pcr- formance of the p o x e r amplifier. F ig . 222(b) shows t h a t no apprc- c i a l~ le decrease in ou tpu t power nor much incrcase in the third-order IRT distortion occurs with increasing case temperature u p t o TI = 130°C. The slight decrease in distortion,

t o g c t l ~ c r with a dccrcnsc in collcc- t o r eliicicncy, can be a t t r i l~u ted to a ~ i s e in rf sa tura t ion voltage and a drcreasc in t rans is tor l x t a : ~ t high temperature.

Despite the ex t r a circuit nccdcd to achieve t empe~ .a tu re stal)ilization, the approach provides a practical solution f o r acliievcment of reliablc opcration of a class AB anll)lifier ovcr a wide temperature range. The use of a small diode a s a tcrnpcra- ture-sensing element offers the following advantages:

( a ) Diode and t rans is tor pellets need not be niatched f o r forward- voltage drop.

(11) Transis tor quiescent cu r rcn t can be either overcompensated or undercompensated agains t changes in temperature by variation of the diode current.

(c) A diode idling current a s low a s 1 to 5 milliamperes can I)c used.

( d ) Current of less than 50 milli- anlperes a t 28 volts is needed to opei.ate the external compensating circuit.

Typical Linear Amplifier The common-emitter configuration

should be used f o r the powey amplifier because of i t s stability and high power gain. Tuning is less critical, and the amplifier is less sensitive to variations in parameters among transistors. The class AB mode is used to obtain low inter~nodula t ion distortion. Nei thcr resistive loading nor neutralization is used to ini- prove l inearity because of the result ing drastic reduction in power ga in ; fur thermore , neutralization is difficult f o r l a rge signals because parameters such a s outl)ut capacitance and output and input impctl- ances va ry nonlinearly over the l imits of signal swing.

In low-power linear amplifiers, the use of temperature-conll)ensal- i ng circuits is sometimes not ncces- s a ry provided t h a t the t rans is tor output power is less than 50 per cent of i t s n l ax in~um cm power ra t- ing. The RCA-2N5070 t rans is tor i s


uscful in sucli app1ir:~tion. This t ransistor is specified f o r SSB applications without temperature compensation a s follows:

Frequency = 30 MHz P,, (PEP) a t 28 V = 25 W Power Gain = 13 dB (min.) Collector Efficiency =

40 r/o (min.)

Fig. 223 shows a 2-to-30-MHz wideband linear amplifier t h a t uses o ther types of RCA r f transistors. A t 5 w a t t s (PEP) output, I M distortion products a r e more than 40 dB below one tone of a two-tone signal. Power gain is g rea te r t han 40 dB.

Fig. 224 shows a 150-watt 2-to- 30-MHz push-pull amplifier t h a t

Fig. 223-2-to-30-MHz lirrear power atnplifier.

RCA

50-OHM

BIAS v c ~ ~ 2 8 V

Fig. 224-2-to-30-AfHz, 130-watt (PEP) push-pull littear arriplifier.

' 156 RCA Transistor, Thyristor, & Diode Manual

uses a pair of 40675 transistors. sets minimum requirements on radio ! Typical performance curves fo r this performance which a re based on

amplifier a re shown in Fig. 225. the maximuln authorized altitudes

- for the plane, whcthcr paying pas- a scngers a re carried, and on the RU-

IL w FREQUENCY -MHz

thorization for instrument flying. The FAA gives a desirable TSO cer- tification to radio equipment tha t satisfies thcir standards of air- worthiness.

The FCC checlcs aircraft-radio transmitter designs for interference and other electrical characteristics ( a s i t does all transmitters). Addi- tional requirements a re specified for radios intended for use by scheduled airlines by a corporation supported by the airlines themselves.

Fig. 226 shows a broadband ali~plifier t h a t can supply 15 watts - of

Fig. 225-Tjpicol pcrfornmrrcc crrrves fo r carrier power fo r aircraft trans- ntrrpli{icr show~r irr f ig. 224. mitters,

AIRCRAFT RADIO The aircraft radios discussed in

I this ser:tion a r e of the type used for communication between the pilot and t l ~ e airport tower. The transmitter opcrates in an AM mode on specific chnrincls between 118 and 1313 nIHz. Radios of this type a re rrgulatecl by both the FCC and the FAA (Federal Aeronautics Adniin- istration). The FCC assigns frequencies t o airports and places some

a requirements on the transmitters, particularly a s regards spurious radiation and interference. The FAA

J

VHF AND UHF MILITARY RADIO

Military radios, which operate i n the vhf and uhf ranges, vary greatly in requirements. Telemetering devices may operate with a s little output a s 0.25 watt, while communication systems may require outputs of 50 watts and more. Dlodulation mny be AM, FRI, PM (pulsc modulation), o r 1JCR.I (pulse-code niodulation). Equipment may be dcsigned for fixed, mobile, airborne, or even space applications. Althougll the circuits described in this section apply

Fi& 226-Anrpli1rcdr-r1to~Itrl~1ed broodbrtrrrl r~r~rplifier fo r 118-to-136-AfHz opcrctrio~l. I

R F Power Amplification and Generation 157

only to specific military applications, they a r e representative of the general design techniques used in all military vhf and uhf radio equipment.

Sonobuoy Transmitters A sonobuoy is a floating sub-

marine-detecting device t h a t incor- porates a n underwater sound detector (hydrophone). The audio signals 'eceived a r e converted to a frequency-modulated rf signal which is transmitted to patrolling aircraft o r surface vessels. The buoy is battery-operated and is designed to have a very limited active life.

Typical requirements fo r the rf- transnlitter section of the sonobuoy a re a s follows:

Frequency = 165 MHz Supply Voltage = 8 to 15 volts CW Output = 0.25 to 1.5 watts Over-all Eficiency = 50 per cent Harmonic Output = 40 dB down

from carrier

Figure 227 shows the circuit configuration of a n experimental sono- I)uoy transmitter designed to pro- dltce a power output of 2 watts a t l(i0 MHz. Only three stages, including the crystal-controlled oscillator

section, a re required. Efficiency is greater than 50 per cent (overall) with a battery supply of 12 to 15 volts.

The 2N3866 or 2N4427 transistor can be used in a class A oscillator- quadrupler circuit which is capable of delivering 40 milliwatts of r f power at 80 MHz. Narrow-band frequency modulation is accomplished by "pulling" of the crystal oscillator. The crystal is operated in i t s fundamental mode at 20 MHz. The oscillator is broadly tuned to 20 MHz in the emitter circuit and is sharply tuned to 80 MHz in the collector circuit. The supply voltage to the oscillator section is regulated a t 12 volts by means of a Zener diode. Spectrum-analyzer tests indicate tha t this stage is highly stable even though rather high operating levels a r e used.

The oscillator-quadrupler section is followed by a 2N3553 class C doubler stage. This stage delivers a power output of 250 milliwatts a t 160 MHz from a 12- to 15-volt supply. The over-all output of the sonobuoy can be adjusted by varying the emitter resistance of this stage.

The final power output is developed by a n RCA-2N2711 transistor which operates a s a straight-through class C amplifier at 160 MHz. A p i

OSCILLATOR-QUADRUPLER DOUBLER

Fig. 227-2-1t~otI (rJ power otctprcf) sottobrroy frar~srrriffer.

network matches this output to the 50-ohm line. The spurious output (measured directly a t the output port) is more than 35 dB down from the carrier. This suppression is achieved l,y means of series resonant t r a p circuits between stages and the use of the pi network in the output.

hlany sonobuoy systems require power outputs in the range of only 0.25 to 0.5 watt, preferably with a supply voltage of 8 to 12 volts. The 2N4427 transistor is suitable for use a s the doubler and also the final output device in such low-power applications. Fig. 228 shows a diagram of a n output stage which uses the 2N4427 a s a straight-through 175- MHz class C amplifier. This circuit can deliver output power of more than 500 milliwatts with a supply voltage of 10 volts and a drive power of GO milliwatts.


Sonobuoy circuits, in general, must be reliable, simple, and low in cost. The three-stage transmitter circuit shown in Fig. 227 is intended to be representative of the general design techniques used in these systems. IIowever, four-stage sonobuoy transmitter systems a r e also in common use a t the present time. Typically, a four-stage arrangement consists of a n oscillator-tripler stage, a second tripler stage, a buffer stage. and a final amplifier stage. Most present-day sonobuoy applications require CW power output between 0.25 and 0.5 watt.


Air-Rescue Beacon 1 The air-rescue beacon is intended

to aid rescue teams in locating air- plane crew members forced down

I on land or a t sea. The beacons a re amplitude-modulated or continuous- " tone line-of-sight transmitters. They a r e battery-operated and . small enough to be included in survival gear.

Typical requirements fo r rescue beacons a r e a s follows:

Frequency = 243 MHz (fixed) Power Output = 300 milliwatts

I (carrier)

Efficiency = greater than 50 per cent

Supply Voltage = 6 to 12 volts Modulation = AM, up t o '100

per cent

F i g . 228--05-\cart 175-MHz sot~ohltoy 1-1 ~ O I I ' E ~ o t t fp l~ t stage.

I

F o r the lower power-output requirement a t low supply volt-

' ages, the oscillator-quadrupler stage should use lower-power transistors such a s the 2N1491 or 2N914. Only 10 to 15 milliwatts of fourth harmonic power is required in this case. The bias-network resistors (Rr and Ra) should be adjusted for reliable oscillator starting conditions a t these

I lower supply voltages.

The 2N4427 transistor is especially suited for this service. A general circuit for the driver and output stages is shown in Fig. 229. Collec- tor modulation, a s well a s some driver modulation, is used to achieve good down-modulation of the final amplifier. Conventional transformer- Coupled modulation is used; however, a separate power supply and resistor network in the driver circuit a re provided to adjust the modulation level of this stage independ- ently of the output stage.

The rf-amplifier design is conventional; pi- and T-matching networks a re used; simpler circuits (e.g.,

AUOIO INPUT

POUT= 300 mW (CARRIER)

Fig. 229-Driver and output stage for a 243-MHz beacon fransrt~iifer.

device-resonated tapped coils), however, could be used. The T-matching network a t the driver input is used to match the amplifier to a 50-ohm source for test purposes. A 10-to-20- milliwatt input signal is needed to develop a 300-to-400-milliwatt carrier output level.

Broadband Power Amalifier R F power transistors a re often

used in broadband amplifier circuits for comniercial and military applications. Transistor transmitters are superior to tube transmittem a i t h respect to broadband capability, reliability, size, and weight. The aircraf t communication bands of 116 to 152 MHz (discusscd in previous section) and 225 to 400 MIIz a r e of interest for both military and com- mcrciill al)plications. Another area of interest is ECRI (electronic counter-measures equipment) applications. Transistors suitable for 1)roadband analications must be ca-

power output within the entire frequency range of interest and constant gain within the pass band. The bandwith of a transistor power amplifier i s limited by the following three factors: (1) intrinsic transistor structure, (2) transistor parasitics, and (3) external circuits such as input and output circuits.

Transistor Structure+-The parameters which determine the bandwidth of a transistor structure are the emitter-to-collector transit time, the collector depletion-layer capacitance, and the base-spreading resistance. The emitter-to-collector transit time, which represents the sum of the emitter capacitance charging delay, the base transit time, and the collector depletion-layer transit time, affects the over-all time of response to an input signal. The emitter-to- collector transit time i s inversely proportional to the gain-bandwidth product f~ of the transistor. A high f~ is essential for broadband oper-

pable of pro;iding both the required ation; in addition, a constant fT with


current level is required for large- of Fig. 230(a) from f l to fr.. External signal operation. The ratio of the feedback is then applied to control f~ to the product of the base-spread- the input drive and flatten the power ing resistance and the collector de- output over a broad frequency band. pletion-layer capacitance (rl,C,.) c o w rises the gain function of a transistor.

Under conjugate-inatclied input and output conditions, the power gain, which is equal to f.r/8;f2r~oC,, falls off a t a rate of 6 dB per octave. In a pow-el- amplifier, the power gnin is usually decieased by less than G dB per octave, a s shown in I'ig. 230(a), hecause the load resistance HI, presented to the collector is not equal to the output resistance of the transistor but is dictated by the required power output and the collector voltage swiug. The curve in Fig. 230(a) indicates tha t one approach to achieving n broadl)and transistor amplifier is to optimize the matching a t the higher end of the freclucncy band and to introduce mis~natcli in the input or output, o r both, a t the lower end of the band so that a constant power output i s obtained from ft to f,; this latter approach is shown in Fia. 230(b). The power output that can be obtained with a transistor broadband a~nplifier is comparable lo that measured a t the high end of the band in a narrowband amplifier; efliciency and power gain a re slightly lower than in a narrowband anipli- ficr because the load and source impedance cannot be ideally matchcd to the transistor over a broad frequency band. The disadvantage of this approach to producing a broadband t r a n ~ i s t o r amplifier is the ye- sultant lelntivrly high input VSWR a t the low rnd of the band.

A more sophisticatcd approach to achieving bro:tdband l~erforniance is to considcr the transistor structurc. the transistor parasitic elements, and the external circuits ac part of the over-all Iinnd-l~ass s t ruc t~u c, in wliich the input and output c irr l~i ts are couplcd togethcr by thc transis- toi. fccdl~nrk rapncit:~ncc. T1.k I -qm- bined structure ~.cprod~lccs the lio\vcr-outpnt or ] ) o n ~ ~ . - g a i n c111.v~

FREQUENCY

Fig. 230-40) Or~lprrt polcwr as a f~rrirtiorr of jreqrrer~cy iri art nrrrplijicr arirh corr- jtr,qa/e-rrtatchcd irrprrt artd orrcprrt corrdi- tiorrs; (6) a rrtetlrod of correctirr.~ rlir de-

crease irl power gain-shosvr irt (a).

Parasitic Limitation-Evcry discrete transistor contains parasitic elements which inipose fur ther limitations on bandwidth. The most critical parasitics arc the emitter- lcad inductance L, and the base inductance L I , . Thcsc parasitic in-

'ductances range from 0.1 to 3 nanohcnries in conime~cially available rf power transistors. In the simple equivalent c i r c ~ ~ i t of a common-emtter transisto1 input circuit a t high frequency show~l in Fig. 231, the i ~ ~ d u c t a n r e LI, , represents the st1111 or the base palasitic inductance and the rcfccted emitter 1)al.nsitic inductance; I t ! , , is the dynn~nic illput


resistance. The real par t of the impedance, R,,, is inversely proportional to the collector area and the power-output capability of the device; i.e., the higher the power output, the lower the value of Ron. A low ratio of the reactance of LI. to R,, is important as the first step in broadband amplifier design. Un- less the reactance of LI. is appreciably lower than the input resistance Rot,, the reactance must be tuned out and thus the bandwidth limited.

collector load must be maintained to provide the necessary voltage and current swings. In addition, the input matching network must be capable of transforming the low input impedance of the transistor to a relatively high source impedance.

Suitable output circuits fo r broadband amplifiers includes constant-K low-pass filters, Chebyshev filters (both transmission-line and lumped- constant types), baluns, and tapered lines. Fig. 232(a) shows a conventional constant-K low-pass filter. The in7 put impedance Zll is substantially constant a t frequencies below the cutoff frequency U. = (LrCr)'". A con- s tan t collector load resistance can be obtained if the shunt arm (1-1)

0 - d of Cn is split into two capacitances, a s shown in Fie. 232(b). P a r t of the - . .

F ; . ~ - E ~ ~ ~ ~ I ~ I I I i l r t i c l i o capacitance represents the output rf polver ~rarwistor. capacitance of the transistor, Co;

the other par t has a value which External Circuits-For a broad- makes the total capacitance equal

hand amplifier circuit to deliver con- t o CK. Further improvement of s tant power output over the fre- bandwidth can be obtained by the quency range of interest, a proper cascading of more sections.

Fi~r . 232-A corn~eritiorral corutar~t-K lowpass lilfer (a), a rrrerhod of obrair~ing a constarit collector lood resisrarrce (b). a short-step rtricrostrip irlrpedance rransformer (c), a lumped

eqrtivalenr Cliebysl~rv impedance Irarisformer (d).

RF Power Amplification and Generation 162 RCA Transistor, Thyristor, & Diode Manual

Fig. 232(c) shows a short-step pass filter. If the cutoff frequency niicrostrip impedance transformer ,, = 1/(L,,C)Ih is high compared to which consists of short lengths of . the frequency of interest (f. in Fig. J relatively high-impedance t rans~nis- 230), the total conlhilled input im- I sion line alternating with short pedance of the transistor input and I lengths of relatively low-impedance the capacitance C is approximately

Fig. 234-225-ro-400-MHz broadband artrplifier rrsirrg ZN5919.

transmission line. The sections of R I , , / ( l - ,",,') and is constant if transn~ission line a re all of the same (,"/,,.')<< 1.

I length Af16). A constant load The remaining step in broadband resistance can be maintained across transistor power amplifier design is the collector-emitter terminals over the design of a network to provide a wide frequency band if the circuit the necessary impedance transfor- is designed to include a Chebyshev nlation over the entire frequency transmission characteristic. Fig. band. Circuits suitable fo r the input 232(d) shows a lumped-element include multisection constant-K fil- Chel)ysliev impedance transformer ters, Chebyshev filters, and tapered which consists of a ladder network lines. A more sop}~isticated approach of series inductances and shunt ca- to obtaining a broadband transfor- pacitances. Transmission-line a s mation in the input is to t reat the well a s strip-line baluns with difler- parasitic inductance L I n of Fig. 233 ent stepdoJr.n ratios (4 to 1, 9 to 1, a s part of the transformation net- and 16 t~ 1) can also he used in the work. For example, LI,, can be con- ,, output to provide the broadl~and im- sitlered a s one a r m of the Chehyshev pedance transformation. low-pass filter of Fig. 232(d). For

Olle dificulty encountered in a given bandpass characteristic, the broadband transistor-power-allIp]i- number of sections increases with . fier design involves the attainment the value of LI.. Again, therefore, of the desired bandwitlth in an input low package parasitic inductance is circuit which provides the required i n l ~ o r t a n t - impedance transformation from the

a n RCA 2N5919 transistor in con- MICROWAVE junction with a Chebyshev input and POWER AMPLIFIERS

I

butput. Fig. 235 shows typical per- fonnance curves for this circuit. With a n input of 4 watts, the circuit is capable of a minimum power output of 15 watts with a variation of 1.5 dB from 225 to 400 MHz; the collector efficiency is greater than 70 per cent.

extremely low input impedance of a transistor to a relatively high

I source inipcdance. The design of the Xin' ~ L i n

I i

input circuit dcpends on the ap- pinc Xin/Rin I proach chosen: optimization of the 1 match a t the high end only, or the use of transistor parasitic elcmcnts

(a)

a s par t of a low-pass structurc. A sunple \vay of optimizing the match a t the high end is to introduce a capacitance between the base and the emitter terminals of the tran- X ~ ~ . X ~ ~ [ I + I / ( Q ~ ~ I ~ ]

sistor to tune out the reactive part of the parallel equivalent input in?- ( b l

pedance of the transistor. The net- I I

works in Fig. 233 show that the Re [yBE] l/Rin(l +pin2) I

lower the inductance L,, or QI,,, ' (w/w012 -1 the less frequency-sensitive is the equivalent parallel resistance R,,.

i xin [ I ~ I / ( O ~ ~ ) ~ ] 1

The networks shown provide a first I step-up transformation for the real (cl

part of the input impedance of the Fig. 233-Ne~!rorks der~rorrsrrnri~rr: rlre I

transistor. when a is con- eflccf OI i)rducfnrrce Lln. Qln 011 c(/rrib)a- 1 nected to the network of Fig. 233(a), lerrr parallel resismt~ce R,,,.

the circuit has the same fonn as a Fig. 234 shows a 225-to-400- half-section of a constant-K low- MHz broadband amplifier that uses 1

at COLLECTOR SUPPLY VOLTAGE (Vccb 2 8 V

- 5 0 2 5 >

O W C 3 Q

Z- INPUT, VSWR

8 2.1 * - 1.1

225 250 275 300 325 350 375 400 FREQUENCY ( f )-MHz

The power-output and frequency capabilities of rf power transistors have been increased many-fold during recent years so that the frequency spectrum over which these devices can provide useful power output now extends well into the microwave regjon.

In comparisons of transistor per- formances. eain and efficiencv. a s - - - ~. . - - ,

well a s power output and frequency, a r e important considerations. The use of more than one low-gain t ransistor to obtain the same gain a s one high-gain transistor results in reduced collector efficiency. For example, Fig. 236 illustrates the use of two transistors which have the same power output, but different gain and collector efficiency. The high-gain unit shown in Fig. 236(a) is capable of delivering an output of 10 watts a t 1 GHz with a gain of 10 dB and a collector efficiency of 50 Der cent. The low-gain unit shown in Fig. 236(b) is also capable

Fig. 235-Typicnl hrondbarld perforr,rorrce of 10 watts output a t 1 GHZ, but of rlre 225-ro-400-MHz arrrplifier circrrit with a gain of only 5 dB and a col-

sho~vtr ir~ Fig. 234. lector efficiency of only 30 per cent.

RCA Transistor, Thyristor, & Diode Manual / RF Power Amplification and Generation

As shown in Fig. 236, two low-gain transistors are required to provide t,hc samc performance a s one I i i ~ h - gain, high-efliciency unit. Besides using an additional transistor, the system of Fig. 236(b) requires twice a s much dc power a s that of Fig. "(;(a); the additional 5 dB of gain required to match the high-gain transistor can I)e achieved only a t the expense of 24 watts of dc power.

20 WATTS PIN' t Po =

l WATT I0 WATTS

TRANSISTOR I '&--F}FI pdcl ' Pdc2'

10.5 WATTS 33.5 WATTS

I0 WATTS WATTS

l GHz l GHz

TOTAL Po= PIN (PGI.PG2) TOTAL Po

TOTAL COLLECTOR EFFICIENCY = pdcl +pdc2

Fi.p. 23(j-Corrrporisor~ o f orre o~rrl f ~ v o - rmrrsitor s).sterrrs ivitll tlre strrrre power orrt~1rrr hrtr d i f l r ror~ ~nirr orrd collecror

eficicr~cies.

From the practical point of view, the system of Fig. 236(b) is more coml>lex, and the higher dissipation of the output transistor is undcsir- able.

Thc 2N510R transistor can I I C used in the comnion-emitter amplificr nlode a t 1,-band frequencies. A typical circuit configulation capable of operation in the 1-to-1.5-GHz range is sIio\vn in Fig. 237. This circuit can provide an output power of 1

wat t a t 1 GHz with a 28-volt power supply. The transistor enlitter is directly grounded to the ground plane of the strip-line circuit board. The input circuit consists of capacitors CI and C2 and the parasitic lead inductance of the 2N5108 transistor.

50 n OUTPUT

Fig. 237- I-GHz power a~rr/)li/irr ~rsirr!: 2 N5 108 ~rrtrzsis~or.

The output circuit uses a capacitive- ly loaded 50-ohm section of strip- line which is resonant a t the operating frequency. The amplifier power gain is in the order of 6 dB; collector efficiency is about 35 per cent.

The RCA-2N5921 coaxial transistor is designed for operation a t high L-band or low S-band frequencies. Fig. 238(a) shows a coaxial-line amplifier circuit which can provide 6 watts of output power a t 2 GHz with a 28-volt power supply. In this circuit, the coaxial transistor is placed in series with the center conductors of the coaxial lines, and the base is properly grounded to sep- a ra te the input and output cavities. The input line L,, in conjunction with capacitance Cl and CL', trans- forms the complex input impedance to 50 ohms of real resistance.

The transistor output load impedance required for a 6-watt output

50 tl

RFC -4- -

NOTE : +"CC RFC IS 3 TURNS N0.32 WIRE, 1/16" ID X 3/16" LONG

la)

COAXIAL OUTPUT

COAXIAL INPUT 'ORON CONNECTOR i SLEEVE

MATERIAL: CENTER CONDUCTOR-COPPER OUTER CONDUCTOR FOR INPUT

* AND OUTPUT-BRASS CONHEX 50-045-0000 SEALECTRO CORE.

OR EOUIV.

12

m 40 10 - In I- 5 " I- z 5 O 4 LL e 2

1 1.2 1.4 1.6 1.8 2 2.2 24 FREQUENCY (t)-GHz

Fig. 238-.4 coaxial-line arrlplifier circrrfl that can provide 6 worts o f ortrprir power at 2 G H z w,ith a 28-volt supply: ( 0 ) circrtit diograr~r; (h ) the Irard~oare reqrtired for rhe circltir in (a); (c) rf power orttprtt as a jlrrrction of freqrterlcy fo r the 2N5921

Irarrsistor.

is 2.5 -I- j2.4 ohms a t 2 GBz; the combination of a 7.8-ohm line L2 (1-inch long) and capacitors C7 and C,, provides the transformation from 50 ohms to this value.

The hardware required in the circuit of Fig. 238(a) is shown in Fig. 238(b). A heat sink is provided by pressing the flange of the transistor to the outside conductor of the cavities. Additional heat flow is obtained through the use of a boron nitride cylinder which makes direct contact between the coaxial-line conductors over the entire length of the cavity. This arrangement improves heat conduction and thus is more suitable for high-power microwave. transistors. In addition, the boron nitride, which has electrical and thermal properties comparable to aluminum oxide, is readily machineable and nontoxic. As a result of the use of the boron-nitride cylinder, coaxial-line lengths a r e substantially reduced.

When operated a t 28 volts, the circuit of Fig. 238(a) can deliver cw power output of 6 watts a t a gain of 7 dB; collector efficiency is greater than 45 per cent. Because of the excellent input and output circuit isolation (within the 2N5921 transistor a s well a s in this coaxial circuit design), the common-base circuit configuration shown in Fig. 238 is extremely stable. Fig. 238(c) shows the power output a s a function of frequency of a 2N5921 transistor a t 28 volts.

It has been established that a well-designed coaxial transistor package (such a s the 2N5921) generally outperforms other transistor packages (including strip-line packages) a t microwave frequencies. This performance can be related to the low values of the parasitic elements and the excellent isolation between the input and output circuits which is possible in the coaxial configuration. Coaxial transistors can also be used in microstrip o r strip-line amplifier circuits which have thermal and electrical performance equal t o tha t of the coaxial-line circuits.


Fig. 239(a) shows the circuit collector stud to the metal block; mounting arrangement of the 2N5921 the block also serves a s both a heat coaxial transistor. The transistor is sink and a ground. The diameters mounted vertically in a llole through of the holes through the metal block a metal block. The cross-sectional and the cylinder of beryllium oxide view of the metal block can also (or boron nitride) a re determined by be seen in Fig. 239(a). The bottom the desired characteristic impedance side of the metal block is counter- of the short coaxial-line section bored so that the base flange of the which is formed by this mounting transistor can be placed flush with technique. The beryllium oxide and

I the metal 1)lock. The hole throug-11 boron nitride have excellent heat

I the metal block has a soniewhat conductivity and low electrical losses larger diameter than that of the and thus provide satisfactory heat ceramic portion of the transistor dissipation from the coaxial transis- which separates the base flange and tor without adversely affecting the the collector stud. A cylinder of be- rf performance. ryllium oxide or boron nitride is The arrangement shown in Fig. ! press-fit between the transistor and 239(a) is suitable for use in micro- the metal block to provide an addi- strip, strip-line, and lumped-element tional heat-conducting path from the circuits. The output circuit can be

constructed on the top portion of the metal bloclc and the input cir-

COLLECTOR cuit on the bottom portion. This a r - a TERMINAL rangement provides excellent isob-

I tion between the input and ,output

I circuits. For example, Fig. 239(b) shows the construction of the micro-

! I

strip-line circuit. The output circuit is constructed of standard microstrip line mounted on the top surface of the metal block. The input circuit is constructed of another microstrip line placed directly over

I MITTER the bottom surface of the metal

block. A strip-line circuit can be ( a ) formed by placing. another s t r ip of

INE dielectric material and round plane I

ET) above the conductor strips of Fig.

In the microstrip amplifier circuit shown in Fig. 240, a 2N5921 transistor is mounted in a 0.350-inch-ID hole in a 0.210-inch-thick aluminum block. The base flange is mounted flush to one surface of this block. The collector section, however, is mounted through the hole in the

INPUT(EMITTER) LINE 'block; a boron-nitride sleeve in the (COPPER SHEET) hole serves a s an additional heat

I b) sink for the transistor. I i

The input and output lines a r e thin (5-mil) copper strips that a r e taped down on 5-mil Dupont H-Film,*

I Fi,r. 239-fn) Circrtir rtrorrrtri~~g nvvorrpe- r l rcsr l r of rlrr RCA-?N5921 coo.~iol trnrr- sisror, nrrd fhl n ~rlicroslrip-lirre circrrit * Registered trademark, Dupont

r~lnkirr~ rtse of the arrc~riga~rrrrrr irr ((1). DeNemours & Co.


which serves a s the dielectric medium of the nlicrostrip circuit. The circuits a r e fixed-tuned a t about 2 GHz. The ceramic capacitors CI and C. (used for dc isolation a t the input and output ports) a re slightly inductive a t 2 GHz. The electrical performance of the circuit is equal to tha t of the coaxial-cavity circuit shown in Fig. 238.

RFc wRFc Z SHORT SECTION OF TRANSMISSION LINE

FORMED BY COLLECTOR STUD SURROUNDING METAL BAR (CHASSISI

Fig. 241(a) shows the configuration for a 2-GHz alnplifier tha t uses the same layout a s t h a t shown in Fig. 239. The metal block is aluminum. The input and output circuits are constructed on 1132-inch Teflon* fiberglass board which i s mounted atop the aluminum so t h a t the input and output lines a r e on opposite sides of the aluminum block.

When operated a t 28 volts with a typical 2N5470 transistor in the circuit, the 2-GHz amplifier can deliver a power output of 1.2 watts

with a gain of 6 dB. The collector efficiency is 43 per cent, and the 3-dB bandwidth i s 12 per cent. The performance of this microstrip-line amplifier is equivalent to that of a cavity o r coaxial-line amplifier circuit.

A similar 1.5-GHz amplifier is shown in Fig. 241(b). The output circuit of this amplifier is constructed on 1132-inch Teflon fiberglass board which is mounted on one surface of a n aluminum block. The input line is constructed on the opposite side of the aluminum block; the block serves a s the ground plane of the line. The input line is formed by mounting a 5-mil copper sheet over a 5-mil dielectric sheet (DuPont H-film) which is placed directly over

- ~

* Kegistercd trademark. Dupont DeNemours & Co.

Fig. 241- (a) A 2-GHz, arrd Ib) a 1.5-GHz striplit~e at~rplifier usirrg t l ~ e type 2N5470

rransistor.

168 RCA Trail

thc nlunlinum bloclr surface. This amplifier circuit, when operated a t 28 volts with a typical 2N5470 transistor included, can providc output power of 1.5 watts with a gain of 8.5 d B and a collector efficiency of 50 per ccnt.

MICROWAVE POWER GENERATION

Microwave power can he gene]-ated by operation of a power tlansistor a s a fundalnental-frequency oscillator o r a s an amplifier incorporated with a low-po~ver, crystal-controllecl multiplier chain. Both modes of operation a re important in n~icrowave applications. Fundan~erital-frequency oscillators are now widely used in local oscillators and sonde oscillators, and for backward-wave oscillator (BWO) replacement.

Fundamental-Frequency Oscillators

Transistors capable of power amplification are also suitable for pow- e r oscillation. The most important par t of every oscillator is a n element of amplificntion. I t is then necessary only to provide a path that fecds back a par t of the power output to the input in the proper phase and a source of dc powcr. The rnaxi- nium frcqucncy of oscillation, which is related to f,,, in a small-signal transistor, is usually difficult to dc- fine in a niicromave power transistor becaose of the added parasitic ele- mcnts. The circuit-design for an oscillator circuit is similar to that discussed previously fo r amplifier circuits.

Fig. 242 shows Colpitts,. Rartley, 2nd Clapp transistor oscillators suitable for use in nlicrowave applications. The inductances and the ca- pncitnnces of the oscillntor shown in Fig. 242(a) can sometimes be considered a s the parasitic clcments of the pacltage. Such parasitic elements can he used to form a transistor oscillator capable of operation a t microvave frequencies provided the f rcquency of oscillation can be controlled. Although the

lsistor, Thyristor, & Diode Manual

transistor configuration is not too .tvell defined in these oscillator circuits, the device can be grounded in high-frequency operation a t the collector, the base, or the emitter without affecting its performance.

l o COLPITTS ( b ) HARTLEY

L C ) CLAPP

Fig. 242-Basic transisfor oscillaror circuits: (a) Colpirts, (6) Ifarrley, attd

fc) Clapp.

L-Band Oscillators-Fig. 243 sho~vs the circuit configuration of a 1.68- CIIz fundamental-frequency oscillator which uses the 2N5108 transistor. This transistor is pacltaged in sr TO- 30 case, and its collector is grounclcd to the ground plane of a 1116-inch Te- flon-fiberglass microstripline board. Power output is talten from the Imse through a 0.75-inch section of 50- ohm microstripline and the capacitor network composed of C, and C.. Power output greater than 0.3 wat t can be obtained a t 1.08 GHz with the 2N5108 transistor. Transistor

-efficiency is 20 per cent a t a supply voltage of 25 volts.

The basic oscillator circuit shown in Fig. 243 is useful over thc range of 1 to 2 G H z with only slight modi- fications in the length of the transmission line L,. F o r example, a n increase of line length to 0.80 inch optimizes the circuit for operation a t 1.6 GHz. Output power of 400


TYPE power output of this circuit is typically 0.3 watt; the efficiency is in the order of 1 G pe r cent. The collector is grounded and power output is taken from the base circuit. All - leads in the circuit must be kept a s short a s possible for highest fre-

7'~ quency response. Capacitor C, forms a part of the feedback loop of the

F 9 circuit, which is basically a Hartley arrangement because L, and the par- astic inductances of C, make up a tapped inductor in the feedback loop. Capacitor Cn is used for tuning while capacitor Cs is used for maintaining output match with tuning.

"cc -"cc + k c

Fig. 243-1.68-GHz j~mdanrental-jreq~certcy . c4 oscillafiotz usit~g a 2NSIO8 rratrrisror.

R I 4 - R 3 w ( I - T

milliwatts (with a 24-volt supply) c31pQ - =

can be expected a t this frequency. < 1

In another interesting modification of the 0.80-inch line, operation is optimized at 1.25 GHz when capacitor Ce is moved to the dotted position. This modification results in a n improved output transformation network wliich can develop better than 800 milliwatts of output power a t 1.25 GHz with the 24-volt supply.

S-Rand Oscillators-Although the 2N5470 coaxial transistor is designed for stable operation in the common- base amplifier mode a t 2.3 GHz, i t Fig. 244-~ Z-GHr Irtr,lped-constarrr 0s- can also deliver a Power output of cillaror using a 2N5470 Irarrsistor. 0.3 watt a t 2.3 GHz a s an oscillator. In this device, the very low values Fig. 245 shows another oscillator of the parasitic elements a re used circuit, a Colpitts type, in which the to simplify circuit requirements; fo r 2N5470 transistor can be used over example, lumped-constant, S-band the range of 1.8 to 2.2 GHz. The circuits can be designed around this base of the transistor is directly unit. However, because of the low grounded to the ground plane on the feedback capacitances of the unit, strip-line board; collector heat i s exte

r

nal feedback loops a r e needed conducted to this board through a for sustained oscillation a t S-band bery]liunl oxide insulating washer. frequencies. Feedback is provided by the phase-

Fig. 244 shows a sinlple lumped- resonant loop composed of L and constant circuit using the 2N5470 C,. The output line makes use of transistor. The circuit is tunable standard microstrip-line techniques: over the range of 1.8 to 2.3 GHz. L2 provides the reactance needed t o At 2 GHz with a 24-volt supply, the tune out the output capacitance; Lt,

170 RCA Transistor, Thyristor, & Diode Manual 1

n quarter-wave t ransforn~cr , trans- forms the real collector load impcd- ance to about 50 ohms. This circuit can also produce about 0.3 watt output a t 2 GIlz with a 24-volt supply.

son

Fig. 24.7-?&Hz ffrioostrip-lirrc o.~cillrtor ttsi~rg n 2N5470 tra~rsistor.

Transistor Frequency- Multiplier Circuits

nccause the output-current waveform of power transistors can be made to contain both fundamental ant1 harmonic frequency components, power output can be obtained a t a desired harmonic frequency by use of a special type of output circuit coupled to the collector of the transistor. Transistors can be connected in either the common-base or the common-emitter configuration f o r frequency n~ultiplication.

The d e s i ~ n of transistor frequency-multiplier circuits consists of selection of a suitable transistor and design of filtering and matching networks for optimunl circuit performance. The transistor must be capable of power and gain a t the funtlanien- tal frequency and capable of con- verting power from the fundan~ental to a harmonic frequency. At a given input power level, the output power a t a dcsil-ed harmonic frequency is equal to the product of the power

gain of the transistor a t the drive frequency and the conversion eficiency of the frequency-multiplier I circuit. Conversion gain can be obtained only when the power gain of the transistor at the fundamental frequency is larger than the conversion loss of the circuit.

Various types of instabilities can occur in transistor frequency-

I multiplier circuits, including low- frequency resonances, parametric oscillations, hysteresis, and high-frequency resonances. Low-frequency resonances occur because the gain of the transistor is very high a t low frequency compared to t h a t a t the operating frequency. "Hysteresis" refers to discontinuous mode jumps in output power when the input power or frequency is increased or decreased. A tuned circuit used in the output coupling network has a .* different resonant frequency under strong drive than under weaker driving conditions. I t has been found experimentally t h a t hysteresis effect can be minimized, and sometimes eliminated, by use of the common- emitter configuration. I

Perhaps the most troubleson~e in- I stability in transistor frequency- multiplier circuits is high-frequency resonance. Such instability shows up in the form of oscillations at a frequency very close to the output frequency when the input drive power is removed. This effect suggests tha t the transistor under this condition behaves a s a locked oscillator a t the I

fundamental frequency. Common- emitter circuits have been found to

i be less critical f o r high-f requency oscillations than common-base circuits. High-frequency resonance is also strongly related to the input drive frequency, and can be elimina ted if the input frequency is kept below a certain value. The input frequency a t which stable operation can be obtained depends on the method used to ground the enlitter of the transistor, and can be increased by use of the shortest possible path from the emitter to groiund.


Varactor diodes a r e also used to provide frequency multiplication. Fig. 102 and associated text given previously in the section on Other Solid-State Diodes define the requirements f o r this type of application.

400-To-800-MHz Doubler-Fig. 246 shows the complete circuit diagram of a 400-to-800-MHz doubler that uses the 2N4012 transistor. This circuit uses lumped-element input and

Fig. 247 shows the power output a t 800 MHz a s a function of the power input a t 400 MHz for the doubler circuit, which uses a typical 2N4012 operated a t a collector supply voltage of 28 volts. The curve is nearly linear a t a power output level between 0.9 and 2.7 watts. The power output is 3.3 watts a t 800 MHz for a n input drive of 1 watt a t 400 MHz, and rises to 3.9 watts a s the input drive increases to 1.7 watts.

idler circuits-and a coaxial-cavity output circuit. The transistor is placed inside the cavity with its emitter properly grounded to the 3.8

chassis. A pi section (C,, Cn, L,, L2, and Cr) is used in the input to match the impedances, a t 400 MHz, of the 3.4

driving source and the base-emitter P junction of the transistor. L? and C3 1 3.0

provide the necessary ground return 5 fo r the nonlinear capacitance of the transistor. L and C, form the idler $2.6

loop for the collector a t 400 MHz. The output circuit consists of an ge, open-ended 1 '/a-inch-square coaxial a

cavity. A lumped capacitance C is added in series with a 1%-inch hollow- 1.8 center conductor of the cavity near the open end to provide adjustment for the electrical length. Power output a t 800 MHz is obtained by direct coupling from a point near the 1.0,

0.4 0.8 1.2 1.6 2.0 shorted end of the cavity. POWER INPUT-W

Fig. 247-Outpltt power n ~ t d collector ef - ficiency ns n fltrrctiotr o f irrprtl power for

INPUT the 400-to-800-MHz frequerrcy dortbler. 400 MHz t20 V

The collector efficiency, which is defined a s the ratio of the rf power output to the dc power input a t a supply voltage of 28 volts, is also shown in Pig. 24'7. The efficiency is 43 per cent measured a t a n input power of 1 watt. The 3-dB bandwidth of this circuit measured a t power output of 3.3 watts is 2.5 per cent. The fundamental-frequency component measured at a power-

OUTPUT 800 MHz

output level of 3.3 watts is 22 dB down from the output carrier. High- er attenuations of spurious com-

Fig. 246-400-to-800-MHz c o ~ r z ~ ~ r o ~ t - e ~ t ~ i t - ponents can be achieved if more filter trarrsistor freqrrerrcy niultiplier. tering sections a r e used.

172 RCA Transistor, Thyristor, & Diode Manual I RF Power Amplification and Generation 173 I

The variation of power output with collector supply voltage a t an input drive level of 1 wat t is shown in Fig. 248. This curve is obtained \vith the circuit tuned a t 28 volts. The curves of Figs. 247 and 248 indicate that the transistor amplifier- multiplier circuit is capable of amplitude modulation.

tOLLECTOR SUPPLY VOLTAGE-V

3.4

3.0

3 1 2.6 I- 3 P

Fi 22 0

P W

g 1.8 a

14

1.0

Fig. 248-Po~vrr orrtprrf as a jltrrcriorr 01 srrpplv ~'olrn,ce jor tllc 400-to-SOU-MHz

freqltolcy dotrbler.

367-To- 1100-MHz Triplcr-The 367-to-1100-MIIz tripler shown in Fig. 249 is essentially the same a s

10 14 18 22 26 30

INPUT 367 MHz

9

r

INPUT POWER= I W

OUTPUT I .I GHZ

TYPE

/

.

Fig. 249-367-h4Hz-10-1 .I-GHz C O ~ I I I I O ~ I - e~~ritter Ira~rsislor Ireqrte~~c)? rr-ipler.

-

ZN4012

/

the doubler shown in Fig. 246 except tha t an additional idler loop, (L,, C,,) is added in shunt with the collector of the transistor. This idler loop is resonant with the transistor junction capacitance a t the second harmonic frequency (734 MHz) of the input drive.

Fig. 250 shows the power output of the tripler a t 1.1 GHz a s a function of the power input a t 367 MHz. This circuit also uses a typical 2N4012 transistor operated a t a collector supply voltage of 28 volts. The solid-line curve shows the pow- e r output obtained when the circuit is retuned a t each power-input level. The dashed-line curve shows the power output obtained with the circuit tuned a t the 2.9-watt output level. A power output of 2.9 watts a t 1.1 GHz is obtained with drive of 1 watt a t 367 MHz. The 3-tlB bandwidth measured a t this power level is 2.3 per cent. The spurious- frequency components measured a t the output a re a s follows: -22 dB a t 340 MHz, -30 dB a t 680 MIIz, and -35 dB a t 1360 MHz.

The variation of power output with collector supply voltage a t an input drive level of I watt is shown in Fig. 251. The variation of collector eficiency is also shown. These curves were obtained with the circuit tuned a t 28 volts.

A 367-MHz amplifier tha t used the same circuit configuration and components a s those of the tripler circuit shown in Fig. 249 was constructed to compare the performance between amplifier and tripler. The conversion efficiency for a large number of tripler units was then measured. The conversion efficiency of the tripler is defined a s the l.lGHz power obtained from the tripler divided by the 367-MHz power obtained from the amplifier a t the same power-input level ( 1 watt) . The eficiency varies between 60 to 75 per cent, and has an average value of 65 per cent; this performance is comparable to that of a good varactor multiplier in this frequency range.

POWER INPUT-W

deliver a power output of 0.5 wat t a t 1.5 GHz with a n input drive of 0.25 wat t a t 500 MHz.

150-To-450-MHz Tripler Circuit- Fig. 252 illustrates the use of the 2N4012 transistor in a 150-to-450- MHz frequency tripler. The input coupling network is designed to match the driving generator to the base-to-emitter circuit of the transistor. The network formed by CI and L? provides a ground return for harmonic output current a t 450 MHz. The idler network in the collector circuit (L, L,, and C,) is designed to circulate fundamental and second-harmonic components of current through the voltage-variable collector-to-base capacitance, CI,~.

Fig. 250-Power orttprtt as a jrtrictiorr of power input /or the 367-MHz-to-1.1-GHz

freqrcer7cy rripler.

COLLECTOR VOLTAGE-V

Fig. 251-Pow>cr orrfprrt as a jrorctiorr o f collcclor sltpp1.v ~ 0 l l l r ~ e for 11ie 367-MHz-

to-1.1-Glfz jreqlterrcy tripler.

A similar tripler circuit that uses a selected 2N3866 and tha t is operated iron1 500 MHz to 1.5 GHz can

fi* fo'

450 MHz

Fig. 252-150-to-450-MHz conirr~orr-emit- rer lrar~sistor frequerlcy tripler.

The network formed by CG1 CO, C I I

L,, and Ln provides the required collector loading for 450-MHz power output. Fig. 253 shows the 450-MHz power output of the tripler as a function of the 150-MHz power input. F o r driving power of one watt, power output of 2.8 watts is ob-

I 174 RCA Transistor, Thyristor, & Diode Manual !

tained a t 450 MHz. The rejection of the 2.8-watt, 450-MHz level. The va- I I

fundamental, second, and fourth har- riation of power output with supply i monics was measured as 30 dB bclow voltage is shown in Fig. 254. i

RF POWER INPUT AT 150 MHz-W

Fi.p. 253-Po~-er ortrprrt as a jrtrrcriorr of por.er itrpril for rlrc 150-to-450-Mff: jrc-

qrtefrcy Iriplcr.

- - - -

DC COLLECTOR-TO-EMITTER VOLTS-V

Fig. 254-Power orctp~it ns o jltrrcrio~l oj collccror strpply voltage jor tlre 15O-to-450-

M H z freqrtorcy friplcr.

TV Deflection

I

F o r reproduction of a transmitted number scanning lines. The field repe- picture in a television receiver, the tition rate is thus 60 per second, and

face of a cathode-ray tube is scanned the vertical scanning rate is 60 Hz. with an electron beam while the in- (For color systems, the vertical

! I

tensity of the beam is varied to con- scanning rate is 59.94 Hz.) I trol the enlitted light a t the phosphor The geometry of the standard odd-

screen. The scanning is synchronized line interlaced, scanning pattern is with a scanned image a t the TV trans- illustrated in Fig. 255. The scanning mitter, and the black-through-white beam starts a t the upper left corner picture areas of the scanned image of the frame a t point A, and b e e p s

! are converted into an electrical sig- across the frame with uniform ve- nal that controls the intensity of the locity to cover all the picture ele-

! electron beam in the picture tube a t ments in one horizontal line. At the ,;the receiver. ,lend of each trace, the beam is rapidly

.,,..&..-, ..7 -. , JPY 2q,t,.-./9 returned to the left side of the frame, I c "SCANNING Y UN D A M E N T ~ as shown by the dashed line, to begin ; -+fl --. the next horizontal line. The horizon- . ~. I i.f ' The scanning procedure used in tal lines slope downward in the di-

the United States employs hori- rection of scanning because the " ,; ",? zontal linear deflecting signal simultane- 1. k:'; ,, linejnterlaced produces a vertical scanning

' " .'scanning pattern for motion, which is very slow compared I /:Lb"' tems includes a total of 525 with the horizontal scanning speed.

taAs~anning lines in a The slope of the horizontal line trace i J,c..pfram.~:having an aspect ratio of 4 from left to right is greater than the : 3 3 h e frames are repeated a t a slope of the retrace from right to left ! of 30 per second, with two fields because the shorter time of the re-

i laced in each frame. The first trace does not allow as much time each frame consists of all odd-number for vertical deflection of the beam.

j scanning lines, and the second field Thus, the beam is continuously and in each frame consists of all even- slowly deflected downward as i t scans

I Fig. 255-The odd-litre blrerlnced scarlnirlg procedure.


the horizontal lines, and i ts position is successively lower a s the horizontal scanning proceeds.

At the bottom of the field, the vcr- tical retrace begins, and the beam is brought back to the top of the frame to begin the second or even-number field. The vertical '!flyback" t i ~ n e is very fas t compared to the trace, but is slow conlpared to the horizontal scanning speed; therefore, some horizontal lines are produced during the vertical flyback.

All odd-number fields begin a t point A in Fig. 255 and are the same All even-number fields begin a t point C and a re the same. Because the beginning of the even-field scanning a t C is on the same horizontal level a s A, with a separation of one-half line, and the slope of all lines is the same, the even-number lines in the even fields fall exactly between the odd- number lines in the odd field.

SYNC

In addition to picture information, the composite video signal from the video detector of a television receiver contains timing pulses to assure that the picture is produced on the faceplate of the picture tube a t the right instant and in the right location. These pulses, which a r e called sync pulses, control the horizontal and vertical scanning generators of the receiver.

Fig. 256 shows a portion of the detected video signal. When the picture is hright, the amplitude of the signal is low. Successively deeper grays are represented by higher amplitudes

until, a t the "blanking level" shown in the diagram, the amplitude represents a complete absence of light. This "black level" i s held constant a t a value equal to 75 per cent of the maximum amplitude of the signal during transmission. The remaining 25 per cent of the signal amplitude is used for synchronization information. Portions of the signal in this region (above the black level) cannot produce light.

In the transmission of a television picture, the camera becomes inactive a t the conclusion of each horizontal line and no picture information is transmitted while the scanning beam is retracinfi to the beginning of the next line. The scanning beam of the receiver is maintained a t the black level during this retrace interval by means of the blanking pulse shown in Fig. 256. Immediately af ter the beginning of the blanking period, the signal amplitude rises further above the black level to provide a horizontal-synchronization pulse tha t initiates the action of the horizontal scanning generator. When the bot- ton1 line of the picture is reached, a s in~i la r vertical-synchronization pulse initiates the action of the vertical scanning generator t o move the scanning spot back to the top of the pattern.

The sync pulses in the con~posite video signal a re separated from the picture information in a sync-separator stage. a s shown in F i ~ s . 257 and 258. This stage is biased sufficiently beyond cutoff so that current flows and a n output signal is produced only a t the peak positive swing of the input signal. In the

MAXIMUM LEVEL

BLACK LEVEL OR w BLANKING LEVEL a

PICTURE _I a

ANFORMATION 25&&& - -

U J -- MAXIMUM WHITE

LEVEL 0 1

Fi,?. 256-Dct~.cfe[l video sigtlal. I

TV Deflection

diode circuit of Fig. 257, negative bias fo r the diode is developed by R and C a s a result of the flow of diode current on the positive extreme of signal input. The bias automatically adjusts itself so

Fig. 257-Diode ~)~nc-separafor circrtit.

tha t the peak positive swing of the input signal drives the anode of the diode positive and allows the flow of current only for the sync pulse. In the circuit shown in Fig. 258, the base-emitter junction of the transistor functions in the same manner a s the diode in Fig. 257, but in addition the .pulses a r e amplified.

Fig. 258-Transistor syr~c-separator circuit.

After the synchronizing signals are separated from the composite video signal, i t is necessary t o filter out the horizontal and vertical sync signals so tha t each can be applied to its respective deflection generator. This filtering is accomplished by RC circuits designed to filter out all but the desired synchronizing signals. Although the horizontal, vertical, and equalizing pulses are all rectangular pulses of the same amplitude, they differ in frequency and pulse width. a s shown in Fig. 259. The horizontal

sync pulses have a repetition rate of 15,750 per second (one for each horizontal line) and a pulse width of 5.1 microseconds. (For color system, the repetition ra te of the horizontal sync pulses is 15,734 per second.) The equalizing pulses have a width approximately half the horizontal pulse width, and a repetition rate of 31,500 per second; they occur a t half-line intervals, with six pulses immediately preceding and six following the vertical synchronizing pulse. The vertical pulse i s repeated a t a rate of 60 per second (one for each field). and has a width of approximately 190 microseconds. The serrations in the vertical ~ u l s e occur a t half-line intervals, diGding the complete pulse into six individual pulses tha t provide horizontal synchronization during the vertical retrace. (Although the picture is blanked out during the vertical retrace time, i t is necessary to keep the horizontal scanning generator synchronized.)

All the pulses described above a r e produced a t the transmitter by the synchronizing-pulse generator; their waveshapes and spacings a re held .within very close tolerances to provide the required synchronization of receiver and transmitter scanning.

The horizontal sync signals a re separated from the total sync in a differentiating circuit t h a t has a short time constant compared to the width of the horizontal pulses. When the total sync signal is applied to the differentiating circuit shown in Fig. 260, the capacitor charges completely very soon a f te r the leading edge of each pulse, and remains charged for a period of time equal to practically the entire pulse width. When the applied voltage is removed a t the time corresponding to t h e trailing edge of each pulse, the capacitor discharges completely within a very short time. As a result, a positive peak of voltage is obtained for each leading edge and a negative peak f o r the trailing edge of every pulse. One polarity i s produced by the charging current f o r the leading edge of the applied pulse, and the

178 RCA Transistor, Thyristor, & Diode Manual I TV Deflection 179 ,

HORIZ. EQUALIZING PULSES PULSES 6 3 . 5 p 3 , 0.5H7

VERTICAL PULSE LEADING I90.5pS EDGE

' - 3 H - l \

\TRAILING EDGE

5.1 ps F ~ K . 259-I4'avefor111 of TV sy~rchrorrizi~~x prtlsrs (H = Irorizorrtal Iitrr period of 1115,750

sc'corrr1.s. or 63.5 as).

opposite polarity is obtained from the discharge current corresponding to the trailing edge of the pulse.

As mentioned above, the serrations in the vertical pulse a r e inserted to provide the differentiated output needed to synchronize the horizontal scanning generator during the time of vertical synchronization. During the vertical blanking period, many more voltage peaks a re available than a re necessary f o r horizontal synchronization (only one pulse is used for each horizontal line period). The check marks above the differentiated output in Fig. 260 indicate the voltage peaks used to synchronize the horizontal deflection generator fo r one field. Because the sync sys- ten1 is made sensitive only to positive pulscs occurring a t approximately the right horizontal timing, the negative sync pulses and alternate differentiated positive pulses produced by the equalizing pulses and the ser- rated vertical information have no

effect on horizontal timing. I t can be seen tha t although the total sync signal (including vertical synchronizing information) is applied to the circuit of Fig. 260, only horizontal synchronization infornlation appears a t the output.

The vertical sync signal is separated from the total sync in a n integrating circuit which has a time constant t h a t i s long compared with the duration of the 5-microsecond" horizontal pulses, but short compared with the 190-microsecond .vertical pulse width. Fig. 261 shows the general circuit configuration used, together with the input and output signals fo r both odd and even fields. The period between horizontal pulses, when no voltage is applied to the RC circuit, is so much longer than the horizontal pulse width tha t the capacitor has time to discharge almost down to zero. When the vertical pulse is applied, however, the integrated voltage across the capacitor builds

HORIZ. EQUALlZlNO VERTICAL EQUALIZING HORIZ. PULSES PULSES PULSE PULSES PULSES

'SYNC I N P U T 0 s E ' 2 ~ D l F F E R E N T I A T E D -OHMS - OUTPUT

ODD FIELDS

nnmn n INPUT

EVEN FIELDS OUTPUT --- ------- -- ----- ----- -- - -

\ - - - - - - - - - - - - - -- - - - - - Fie. 261-Sepamtiort of lwrtical sytrc sig- trals fror~r the total syrrc for odd and even fields ~viirk 110 eqltalizing pltlses. (Dashed Iirre iridicates triggerir~g level for vertical

scarrtring getrerator.)

up to the value required f o r triggering the vertical scanning generator. This integrated voltage across the cnpacitor reaches i ts maximum amplitude a t the end of the vertical pulse, and then declines practically to zero, producing a pulse of the tri- angular wave shape shown f o r the complete vertical synchronizing pulse. Although the total sync signal (including horizontal information) is applied to the circuit of Fig. 258, therefore, only vertical synchronization information appears at the output.

The vertical synchronizing pulses a r e repeated in the total sync signal a t the field frequency of 60 per second (59.94 per second in color sys- tctns). Therefore, the integrated output voltage across the capacitor of the RC circuit of Fig. 261 can be coupled to the vertical scanning generator to provide vertical synchronization. The six equalizing pulses imniecliately preceding and following the vertical pulse improve the accuracy of the vertical synchronization f o r better interlacing. The equalizing pulses tha t precede the vertical pulses make the average value of applied voltage more nearly the same f o r even and odd fields, so tha t the integrated voltage across the capacitor adjusts to practically equal values fo r the two fields before

the vertical pulse begins. The equalizing pulses that follow the vertical pulse minimize any difference in the trailing edge of the vertical synchronizing signal for even and odd fields.

VERTICAL DEFLECTION

The vertical-deflection circuit in a television receiver is essentially a class A audio amplifier with a complex load line, severe low-frequency requirements (much lower than 60 H z ) , and a need f o r controlled linearity. The equivalent. low-frequency response for a 10-per-cent deviation from linearity is 1 Hz. A simple circuit configuration is shown in Fig. 262.

The required performance can be obtained in a vertical-deflection circuit in any of three ways. The amplifier may be designed t o provide a flat response down to 1 Hz. This design, however, requires a n extremely large output transformer and im- mense capacitors. Another arrangement is to design the amplifier for fairly good low-f requency response and predistort the generated signal.

a The third method i s to provide ex- t r a gain so tha t feedback techniques can be used to provide linearity. I f loop feedback of 20 or 30 dB is used,, transistor gain variations and nonlinearities become fairly insignificant. The feedback automatically provides the necessary "predistor- tion" to correct low-frequency limi-

Fig. 262-Sir~rple wrtical-deflecfiot~ circuit. tations. In addition, the coupling of miscellaneous signals (such a s power- supply hum or horizontal-deflection signals) in the amplifying loop i s suppressed.


Fig. 263 shows a vertical-deflection system tha t employs bipolar and MOS transistors. A positive pulse fed back from the output circuit triggers the oscillator Q1. The high input impedance of the hIOS transistor Q2, used a s a predriver, permits the use of relatively large resistors and small capacitors in the gnte-No.1 circuit. Negative sync is injected a t gate No. 2. Only 4 to 5 volts of sync a t the integrator input provides exceptionally good inter- lace.

The thermal compensating stage, Q., provides thermal tracking during warmup and also prevents thermal runaway. The peak current of the output stage, Q,, is monitored hy connection of the base of Qs to the emitter side of the emitter resistor of Ql. The output voltage developed at the collector of QG is pl.oporlional to the peak current of the vertical output stage and is fed back to gate No.1 of the predriver Q.. by means of the bias-linearity control. If some condition exists which causes the peak current of

the output s tage to increase, the thermal-compensating transistor Q:. conducts more heavily and causes a reduction in the average voltage a t i ts collector. This decreasing voltage changes the bias of the predriver Q2. Because the predriver, driver, and output stages a r e all direct-coupled, the changes in the peak current of the output stage are coupled back to the base of the output stage in such a polarity a s to adjust the dc operating conditions of the output stage to compensate for any change in peak current. .

There a re two linearity potentiometers in the circuit. The first is a bias potentiometer which sets the bias on the predriver and, in turn, on the output unit so tha t the output unit commences scan from cut* off. The second potentiometer is located in the integrating circuit, which shapes a sawtooth waveform taken from the output and feeds i t back to gate No.1 of the prcdriver to provide the required parabolic correction for good linearity.

9 IJEG. SYNC. t21 V0-y-1 &[- 1 T.

TO HORIZONTAL

CIRCUIT

BOARD

= DRIVER

+21 v BIAS LIN.

Fi.q. 263-Vr1ti~~crl-rl~~,Rt~ctio1t cirrrit for color TV receiver.

! TV Deflection

The parabolic sawtooth voltage required for convergence is obtained from the collector of the output transistor Q.. This sawtooth voltage is coupled to the base of the convergence amplifier Qn and then applied to the convergence board.

For vertical blanking, the negative retrace pulse from the second-

1 ary of the vertical output transformer is amplified and inverted by a blanking transistor, and i s then applied to the cathodes of the picture tube.

HORIZONTAL DEFLECTION In the horizontal-deflection stages

of a television receiver, a current t h a t varies linearly with time and has a

I sufficient ' peak-to-peak amplitude I must be passed through the horizon-

tal-deflection-yoke winding t o develop a magnetic field adequate to deflect the electron beam of the television picture tube. After the beam is deflected completely across the face of the picture tube, i t must be returned very quickly to i t s starting point. (As explained previously, the beam is extinguished during this retrace by the blanking pulse incorporated in the composite video signal, o r in some cases hy additional external blanking derived from the horizontal-deflection system.)

Basic Circuit Requirements The simplest form of a deflection

circuit is shown in Fig; 264(a). I n this circuit, the yoke impedance L is assumed to be a perfect inductor. When the switch is closed, the yoke current s ta r t s from zero and increases linearly. A t any time t, the current i is equal t o EtIL, where E is the applied voltage. When the switch is opened a t a later time t,, the current instantly drops from a value of E t r / L to zero.

Although the basic circuit shown in Fig. 264(a) crudely approaches the requirements fo r deflection, i t presents some obvious problems and limitations. The voltage across the switch becon~es extremely high, the-

oretically approaching infinity. In addition, if very little of the total time is spent a t zero current, the circuit would require a tremendous

(a) SIMPLE DEFLECTION CIRCUIT

(b) ADDITION OF CAPACITOR

(c) YOKE CURRENT [top) AND SWITCH VOLTAGE (bottom) FOR CIRCUIT (b)

(d l YOKE CURRENT (top) AND SWITCH VOLTAGE [bottom) FOR SWITCH CLOSED AT tr

[el ADDITION OF DAMPER DIODE

Fig. 264-Development o f horizonfal-de- ffccfion circ~cit.

RCA Transistor, Thyristor, & Diode Manual I TV Deflection

atnount of dc power. Further~i ior t~, the operation of the switch would be rather critical with regard to both i ts opening and i ts closing. Finally, because tlie deflection field would be phased in only one direction, the beam would have to be centered a t the extrc~ile left of the screen for zero yoke current.

If a capacitor is placed across the switch, a s shown in Fig. 2'72(11), the yoke current still increases linearly when the switch is closcd a t time t = 0. Ilowever, when the switch is opened a t time t = t , , a tuned circuit is formed 1)g the parallel coml>ination of L and C. The resulting y3lre currents and switch voltages are then as shown in Fig. 264(c). The current is a t a niaxin~um when tlie voltage equals zero, and the voltage is a t a maximum when the current equals zero. If i t is assunled that there a re no losses, the ringing frequency f,., is equal to 1 / ( 2 ~ d E ) .

If the switch is closed again a t any time the capacitor voltage is not equal to zero. an infinite switch current

wavefornls f o r this new condition. If the switch is again opened a t

t,, closed a t tc, and so on, the desired sweep results, the peak switch voltage is finite, and the average supply current is zero. The deflection system is then lossless and efficient and, because the average yoke current is zero, beam decentering is avoided. The only faul t of the circuit of Fig. 264(b) is the critical tinling of the switch, particularly a t time t = t,. However, if the switch is shunted by a damper diode, a s shown in Fig. 264(e), the diode acts a s a closed switch a s soon a s the capacitor voltage reverses slightly. The switch may then be closed a t any time between t, and t?. Transistor Horizontal-Deflection

Circuits In horizontal-deflection circui ts ,

the switch can be a transistor, a s shown in Fig. 265. Although the transistor is forward-biased prior to t. (shown in Fig. 264), i t is not an effective switch for the reverse collector current; therefore, the

flows a s a result of the ,-aDacitive dis- damper diode carries most of this IIowever, if thi switch is cume"t. High voltage is generated

by use of the step-up transformer closed a t the precise moment tz t h a t in parallel with the This the capacitor voltage equals zero, the step-up transformer is so capacitor current effortlessly trans- that its leakage inductance, dis- fcrs to the switch, and a new transient tributed caDacitance. and outDut condition results. Fig. 264(d) shows stray capac;tance cdmplement \he the yoke-current and switch-voltage yoke inductance and retrace tuning

FLYE~ACK A

TRANSFORMER PICTURE-TUBE HIGH-VOLTAGE LANODE [ULTOR)

RECTIFIER CAPACITANCE -

f. CAPACITOR ~z~~~~~~~

capacitance in such a manner that the peak voltage across the primary winding is reduced and the peak voltage across the secondary winding is increased, a s compared to the values that would be obtained in a perfect transformer. This technique, which is referred to a s "third-harmonic tuning", yields a voltage ratio of secondary-to-primary peak voltage of ap- proxin~ately 1.7 times the value expected in a perfect transformer.

To provide linearity correction for wide-angle television picture tubes, i t is necessary to retard the sweep rate a t the beginning and end of scan. Therefore, a suitable capacitor C2 is placed in series with the yoke, a s shown in Fig. 265, so tha t the direct current required to replenish circuit losses is fed through the flyback- transformer primary. A parabolic waveform is then developed across C, (called the S-shaping capacitor) so that the trace voltage across the yoke is less a t the ends of the sweep than in the middle of the sweep. (This

capacitor actually provides a series resonant circuit tuned to approximately 5 kHz so tha t a n S-shaped current portion of a sine wave results.) I t is desirable to place the S-shaping capacitor and the yoke between the collector and the emitter of the transistor so t h a t the yoke current does not have to flow through the power supply.

The highest anticipated peak voltage across the transistor in Fig. 265 is a function of the dc voltage obtained a t high ac line voltage and a t the lowest horizontal-oscillator frequency. ( A t these conditions, of course, the receiver is out of sync.) The tolerance on the inductors and capacitors alters the trace time only slightly and usually may be ignored if a 10-per-cent tolerance is used for the tuning capacitor.

Fig. 266 shows a schematic of a transistor horizontal-deflection cir- cut for a color TV receiver. The horizontal output transistor, Q,, is a high-voltage silicon transistor.

Fig. 266-florizorrrol-dcfiectio~l circrril and Irigli-voltage and low-voltage power sirpplies.


The normal collector-emitter pulse voltage across Q , includes a n ample safety factor t h a t allows for any increased pulse t h a t may result from out-of-sync operation, line surges, and other abnormal conditions.

A unique feature of the horizontal-deflection circuit is the low- voltage supply of approximately 23 volts tha t is derived from it. This features makes it possible to eliminate the power transformer in the power supply. The low-voltage power is used to operate all but the high- voltage receiver stages, such a s the video-output stage, the audio-output stage, and the horizontal oscillator and driver. The vertical oscillator is supplied from the same point which supplies the horizontal output in such a way t h a t the actual voltage is a function of beam current; this connection compensates for the tendency for picture height to change with brightness settings.

The transistor deflection circuit achieves commercially acceptable high-voltage regulation without the use of the high-voltage shunt regulator used with tube-type deflection circuits. With a flyback transformer of normal design and a low-voltage power supply with about 3-per-cent regulation, high-voltage regulation from zero beam to full load of 750 microan~peres is about 3 ltilovolts and is accompanied by a considerable increase in p i c t ~ ~ r e width. Improvement of this behavior with brightness changes is achieved by utilizing the accompanying changes of direct current to the deflection circuit in two ways. First, the a i r gap of the transformer is reduced to permit core saturation to decrease the system inductance :IS the high-voltage load is increased. When this method is used, regulation is improved to about half tha t of the normal transformers with no circuit instabilities, but picture-width change is still greater than desired. Second, series resistance is added t o the B supply to decrease power input a t full load

and thereby reduce the change in picture width ( a t some sacrifice in high-voltage regulation). The net result of both changes is a regulation of about 2.8 ltilovolts fo r the high voltage, with very little variation in picture size.

A secondary benefit of the inherently good regulation of the transistor deflection system is a reduction in the size of the flyback transformer. The size reduction is accomplished .by a reduction in the area of the "window" in the flyback core. A reduction in the size of the high- voltage cage required to maintain adequate isolation of the high-volt- a g e winding from ground is possible because of the snlaller flyback transformer.

The transformer-coupled driver stage takes advantage of the high- voltage capability and switching speed of the horizontal driver t ransistor which is designed ,primarily f o r video-output use. A sine-wave stabilized multivibrator type of horizontal oscillator is used. This type of oscillator is especially useful in experimental work with deflection systems because i t permits on-time and off-time periods to be easily varied.

The afc phase detector operates on the principle of pulse-width variation of combined sync and reference pulses. I11 the circuit shown in Fig. 266, timing infornlation is related to the leading edges of the sync pulses, and the retrace process is initiated prior t o the leading edge of the sync pulse; performance of the circuit is very satisfactory.

SCR Horizontal-Deflection Circuit

A highly reliable horizontal-deflection system tha t uses silicon controlled rectifiers (SCR's) has been developed f o r use in color television receivers. This system, shown in Fig. 267, illustrates a new approach to horizontal-circuit design t h a t represents a complete departure from the approaches currently used in com-

TV Deflection

I mercial television receivers. The

i switching action required t o generate the scan current in the horizontal- yoke windings and the high-voltage pulse used to derive the dc operating voltages fo r the picture tube is controlled by two SCR's tha t a r e used in conjunction with associated fast- recovery diodes to form bipolar switches.

The SCR's used to control the trace current and to provide the

TO VERTICAL

TO HEIGHT

VERTICAL CIRCUIT

commutating action to initiate trace- retrace switching exhibit high voltage- and current-handling capabilities together with the excellent switching characteristics required for reliable operation in deflection- system applications. The switching diodes, (trace and commutating diodes), provide f a s t recovery times, high reverse-voltage blocking capabilities, and low turn-on voltage drops. These features and the fact

Fig. 267SCR horizontal-deflection circuit.


that, with the exception of one non- critical triggering pulse, all control voltages, timing, and control polarities a r e supplicd by passive elements within the system (rather than by external drive sources) contribute substantially to the excellent reliability of the SCR deflection system.

Fig. 267 shows the circuit configuration of the conlplete horizontal-deflection system. The system operates directly from a conventional, unrcgulatcd dc power supply of +I55 volts, and proviclcs full- screen deflection a t angles up to 90 degrees a t full bean1 current. The current and voltage waveforms required for horizontal deflection and for generation of the high voltage are derived essentially from LC rcsonarit circuits, As a result, f as t and abrupt switching transients which would impose strains on the solid-state device a r e advoided.

A regulator stage is included in the SCR horizontal-deflection circuit to maintain the scan and the high voltage within acceptable lilnits with variations in the ac line voltage or picture-tube beam current. The system also contains circuits that provide full protection against the effects of arcs in the picture tube or the high-voltage rectifier, and linear i ty and pincushion correction circuits.

The SCR horizontal-deflection system enlploys two bidirectional switches, each of which consists of an SCR and a diode in an inverse parallel connection. Fig. 268 shows a simplified schematic of the basic deflection circuit. SCRT and diode Dr a r e used to control the current in the yoke winding L, during the trace interval; SCRr and diode Dr provide the comniutating action required for retrace.

A t the beginning of the trace interval, the trace-switch diode conducts the yoke current established during previous circuit action. The trnce-switch diode conducts a linearly decreasing current until the yoke current reaches zero to produce the first half of the scan current. Be- fore the zero-yoke-current point is

reached, the trace-switch SCIL is made ready to conduct by application of a positive pulse to its gate electrode. When the yoke current crosses the zero point from negative t o positive, the current transfers from the trace-switch diode to the trace-switch SCR. Capacitor C,. then begins t o discharge through the trace-switch SCR to supply current to yoke winding L, during the second half of the trace interval. The voltage across capacitor C, remains essentially constant throughout the trace-retrace cycle. This constant voltage results in a linearly rising current through the yoke winding to complete the trace period.

COMMUTATING S ~ ~ ~ ~ $ HIGH- SWITCH VOLTAGE

- - - - ~ i < 2 6 8 z ~ o s i c circrtir for gorc'rafiort of 111e def7ectiot1-crtrretrf tc*aveforttr it1 rlrr

hori;ottral-yoke tvitrditrg.

J u s t prior t o the end of trace, the commutnting-switch SCR is gated on by the horizontal oscillator. Capaci- to r Ce then discharges a pulse of current through inductor LN and the trace and commutating SCR's. This current pulse, referred to a s the commutating pulse, increases until i t exceeds the yoke current and thereby causes the trace diode DT to turn on. The conduction of diode DT reverse-biases the trace SCR for sufficient time to allow i t to turn off. When the commutating pulse declines to a value less than the yoke current, diode DT opens, and the energy in the yoke winding produces a current that charges the retrace

TV Deflection

capacitors CR and CA during the first half of retrace. This current then rings back into the yoke winding during the second half of retrace. The circuit fo r the ringing oscillation during the second half of retrace is completed through the commutating-switch diode and allows suflicient time for the commutating-switch SCR to turn off. When the yoke current reaches i ts peak negative value, the trace- switch diode begins to conduct t o s ta r t the trace interval.

During the time the commutating switch is closed, the input inductor LCC is connected across the B+ supply, and energy is stored in this inductor. This stored energy charges the retrace capacitors CR and CA to replenish the energy loss in the circuit.

Fig. 269 shows the current and voltage waveforms applied to the trace and commutating switches a s a result of the circuit actions described in the preceding paragraphs.

The SCR horizontal-deflection system offers a number of distinct advantages over the conventional types of systems currently used in commercial television receivers. The following list outlines some of the more significant circuit features of the SCR deflection system and points out the advantage derived from each of them:

1. Critical voltage and current waveforms, and timing cycles a r e determined by passive components in response to the action of two SCR-diode switches. The stability of the system, therefore, is determined primarily by the passive components. When the passive components a r e properly adjusted, the system exhibits highly predictable performance characteristics and exceptional operational dependability.

2. The only input drive signal required for the SCR deflection system is a low-power pulse which has no stringent accuracy specification in relation to either amplitude or time duration. The deflection system, therefore, can be driven di-

rectly from a pulse developed by the horizontal oscillator.

3. This deflection system is unique in that, although i t operates from

I COMMUTATINO-SWITCH VOLTAGE

(DIODE AND SCR)

0 COMMUTATING- SWITCH

I DIODE CURRENT

1

I 1 I COMMUTATING-SWITCH SCR I M GATE SIGNAL I

1 -- I

TRACE-SWITCH VOLTAGE (DIODE AND SCR) I

I DIODE CURRENT

Fig. 269-Volrage and cttrrerrt wavefortns applied to the switching SCR's and diodes

in the horizontal-deflecriot~ system.

a conventional B+ supply of +I55 volts, the flyback.pulse is less than 500 volts. This level of voltage stress is substantially less than t h a t in conventional line-operated systems, and this factor contributes to improved reliability of t h e switching devices.

4. Regulation in the SCR deflection system is accomplished by control of the energy stored by a reactive element. This technique

188 RCA Transistor, Thyristor, & Diode ManuaI }

avoids the use of resistive-load rcgu- zero current level through the re- lating elements required by many verse recovery of high-voltage p-n I other types of systems and, there- junctions in the deflection diodes. fore, makes possible higher over-all The diode junctions are not limited ; svstem efliciency and reduces input- in volt-ampere switching capabili- power requirements. ties for either normal o r abnormal

5. All switching occurs a t the conditions in the circuit. Power Switching and Control

RANSISTORS have already is- tablished themselves in switching

applications in radar, television, telemetering, pulse code communications, and computing equipment. More re-

; cently, triacs, discs, and SCR's have been used in these applications and in arc-lamp ballasting circuits, automobile ignition systems, and heat,

J 1 light, and motor controls. This section describes the circuits used in these applications and discusses special consideration required for their operation.

NONSINUSOIDAL OSCILLATORS

Oscillator circuits which produce nonsinusoidal output waveforms use a regenerative circuit in conjunction with resistance-capacitance (RC) or resistance-inductance (RL) con~ponents to produce a switching action. The charge and discharge times of t h e reactive elements (which a re directly proportional t o R x C or LIR) a r e used to produce sawtooth, square, o r pulse output waveforms.

The switching action in a nonsinusoidal oscillator occurs when a n externally applied signal causes a n instantaneous change in the operating s tate of the circuit; when this instantaneous change occurs the circui t ' is said t o be triggered. Trig- gered circuits may be astable, monostable, or bistable.

Astable triggered circuits have no stable state; they operate in the active linear region and produce relaxation-type oscillations. A monostable circuit has one stable s tate

in either of the stable regions (cutoff or saturation) ; a n external pulse "triggers" the transistor to the other stable region, but the circuit then switches back to its original stable s tate a f te r a period of time determined by the time constants of t h e circuit elements. A bistable (flip-flop) circuit h a s a stable s tate in each of the two stable regions. The transistor is triggered from one stable s tate to the other by a n external pulse, and a second trigger pulse is required to switch the circuit back to its original stable state.

The multivibrator circuit shown in Fig. 270 is . a n example of a monostable circuit. The bias network holds transistor Q8 in saturation and transistor Q, a t cutoff during the quiescent or steady-state period. When an input signal is applied through the coupling capacitor C1, however, transistor QI begins to conduct. The de- crea_sing col lector v o l t a ~ C of-, (coupled tp the-base of Q, through capacitor G) ( causes the -, base current

0-l w CI

Fig. 270-Mor~ostable r~~ultivibrator.


and collector current of Qz to de- .-.- crease. The i n c r F Z S i ~ ~ I ~ t ~ f . ~ age of Q, (coupled to the base of Q, through resistor R,) then increases the forward base current of QI. This regeneration rapidly drives transistor QI into saturation and transistor Q2 into cutoff. The base of transistor Q? a t this point is a t a negative potential almost equal to the magnitude of the battery voltage V,,.

Capac i to r Ct t h e n d i scharges through resistor R2 and the low saturation resistance of transistor Q1. As the base potential of Qz becomes slightly positive, transistor Qz again conducts. The decreasing collector INPUT A

potential of Qr is coupled to the base of QI and transistor QI is driven into CZ cutoff, transistor Q2 becomes r i g . 271-Eccles-Jorclrr,r rype bisrable saturated. This stable condition is rrrul~ivibrotor. maintained until another pulse triggers the circuit. The duration of the output pulse is primarily determined by the tinie constant of capacitor Cx and resistor R? during discharge. In other words, the oscillating frequency of the multivibrator is determined by the values of resistance and capacitance in the circuit.

The Eccles-Jordan type multivi- \)rator circuit shown in Fig. 279 is a n example of a bistable circuit. The resistive and bias values of this circuit are chosen so tha t the initial application of dc power causes one transistor to be cut off and the other t o

J circuit to i ts original state. (Collector triggering can be accomplished in a similar manner.) The capacitors CJ and C, a r e used t o speed up the regenerative switching action. The out- pu t of the circuit is a unit step voltage when one trigger is applied, o r a square wave when continuous pulsing of the input i s used.

A blocking oscillator is a form of nonsinusoidal oscillator which conducts for a short period of time and is cut off (blocked) fo r a much longer period. A basic circuit fo r this type of oscillator is shown in Fig. 272.

be driven into saturation. Because of the feedback arrangement, each transistor is held in i ts original s ta te by the condition of the other. The application of a positive trigger pulse to the base of the O F F transistor or a C

negative pulse to the base of the ON transistor switches the conducting s tate of the circuit. The new condition is then maintained until a second pulse triggers the circuit back t o the original condition. kc=-

In Fig. 271, two separate inputs a r e shown. A trigger pulse a t input A will change the s tate of the circuit. input An B or input an input of the of same opposite polarity polar- a t !f3+c ~ i ~ . 2 7 2 - ~ ~ ~ ; ~ rircrri, b/orkiim

ity a t input A will then return the oscillaror.

Power Switching and Control 191

Regenerative feedback through the tickler-coil winding 1-2 of transformer TI and capacitor C causes current through the transistor to rise rapidly until saturation is reached. The transistor is then cut off until C discharges through resistor R. The output waveform is a pulse, the width of which is primarily determined by winding 1-2. The time between pulses (resting or blocking time) is determined by the time constant of capacitor C and resistor R.

SWITCHING REGULATORS

Fig. 273 shows the basic configuration of a switching type of transistor voltage regulator. In this circuit, the pass transistor is connected in series with the' load and is pulse-duration nlodulated by the signal supplied from the pulse generator o r multivibrator. The ON time of' the multivibrator is controlled by a dc comparison between a reference voltage and the output. The pulsed output from the series transistor is integrated by the low- pass filter. When the transistor is conducting, current is delivered to the load from the input source. In the O F F condition, the diode conducts and the energy stored in the reactive elements supplies current to the load. If the output voltage tends to decrease below the reference voltage, the duration of the ON-time pulse increases. The pass

transistor then conducts for a longer period of time so tha t the output voltage increases to the desired level. If the output voltage tends to rise above the reference voltage, the duration of the ON-time pulse decreases. The shorter conduction period of the pass transistor then results in a compensating decrease in output voltage.

When a step-down regulator is required (e.g., 100 volts down ' to 28 volts), the efficiency of a switching regulator i s considerably higher than t h a t of a conventional series regulator. If very precise regulation is required, the switching regulator can be used a s a pre-regulator followed by a conventional regulator circuit: this confimration ootimizes the advantages of both Gpes of - regulators. Over-all efficiency for such a combination circuit is typically about 80 to 85 per cent, a s compared to values of 25 to 30 per cent f o r a conventional series-type step-down regulator. In addition, total power dissipation is reduced from several hundreds of wat t s to less than 50 watts.

Fig. 274 shows a switching regulator included in the design of a mercury-arc-lamp ballasting system. DC potential is applied to the Vln terminals so that the transistor switch QI (par t of the switching regulator) is slightly forward-biased by a small current through Ra (approximately 3 milliamperes). Through positive feedback, Ql is immediately saturated by L2, which also powers the control circuit. Cur-

Fig. 273-Basic diagrant of switching regulator.

I RCA Transistor, ~hyr istor , & Diode Manual Power Switching and Control I

VOLTAGE

D I O D E

Fig. 2 7 4 - S ~ ~ ~ i f c h i r r ~ - r ~ ~ g r i I ~ 1 1 0 r tlr~si,vrr for sol i t l -~fofc~ rrrercltry-arc-larrrp 6iillasfirrg.

rent rises a t a linear rate until the voltage across R, causes the control circuit to shunt tlie base-emitter junction of 4,. Q , is shut off and held off by L2 until the current t l~rough L, is zero. The inductive kickback voltage is clamped by the communtating diode and, therefore, is the saliie a s tlie output voltage on C2. L, charges C. to a voltage proportional to Vor.~. During the next cycle, the control circuit samples a conlbination of the voltage on C3 and the current in RI.. The output waveshapes for the circuit a re shown in Fig. 275; performance data a r e sllown in Fig. 276.

The unique feature of this circuit

is that only the high-current switching element Q, must meet the breakdown-voltage requirement imposed by the high input voltage; with this one exception, all of the control- circuit transistors a re of the IOW- voltage, low-dissipation type. The circuit is able to withstand operation under short-circuit conditions.

DC INPUT VOLTAGE TO FILTER CAPACITOR CI

Fig. 276-Perfor rrrcrrrce crrrvcs /or circrrir o f Fig. 274.

A 175-watt switching-regulator ballast circuit utilizing the approach just described is shown in Fig. 277. For three-phase operation, no CI filter element is necessary provided that the dc input voltage to the switching regulator never drops below 200 volts. An input voltage drop below this level would extinguish the bulb.

Switching-regulator techniques are also utilized in motor-control systems. A servo motor control is shown in Fig. 278.

Switching-mode servo controls af- ford a n efficient means for amplification of directional information. As a n alternative to the use of cascaded linear stages to drive a class B push-pull output stage, this switching mode of control allows the active elements of the amplifier to operate in either saturation or cutoff. Because a relatively small length of time is spent in the active region of the devices, where power dissipation is high, the average power dissipation is lower. The efficiency oE the over-all system, therefore, is t--igher. Switching servos are used in stable platforms for guidance and

Fig. 277-175-wolf switching-regulafor ballast.

navigational systems, control of memory access devices in computer and data-processing systems, and other applications in which efficiency is a prime factor.

An ever-expanding application f o r switching systems is in the ac motor-control field. Sometimes this

application is necessary because the standby power is dc. More generally, however, high-speed inverters or switching circuits a re used because the higher-frequency motors are more efficient and weigh less than their lower-frequency counter- parts.

MOTOR VOLTAGE 3F +40 V

Fig. 278-P1clsc-widfh-r11odulufed servo-rr~ofor-driven orrfpuf sfage. !


CONVERTERS AND INVERTERS

In many applications, the apti- mum value of voltage is not available from the primary power source. In such instances, dc-to-dc converters o r dc-to-ac inverters may be used, with o r without regulation, to provide the optimum voltage for a

given circuit design. An inverter is a power-conversion

device used to transform dc power to ac power. If the ac output is rectified and filtered to provide dc again, the over-all circuit is referred to a s a converter. The purpose of the converter is then to change the magnitude of the available dc voltage.

+ SATURABLE CORE ->

(a1

\SL-] " w SATURABLE CORE

REVEaS CORE SATURATION DIODE (U ) CONDUCTION 7'

LOADED

NO LOAD

R I N G I N G

I

(C) Fir. 279-Sirriplc corlverter circrri~s that nray be rtssed t o replace vibralor-type corrver/ers irz arrror~rohile radios: fa ) corrverter circrtit tllat uses separate otrtprrt arrd feedback trarrs- forrrrers; (h) converier circrrir in w,hicll the feedback wirtdirrgs are irrcluded 0 1 1 the orrt-

put trarrsforr~rer; ( c ) typical voltage atrd cltrrerrr waveJorn~s.


Transistor Converters and Inverters

Iqg. 279 shows two simple converter circuits which can be used in place of the conventional vibrator- type converter in automobile radios. The switching drive to the two transistors is supplied by a separate, small, saturable transformer in the circuit of Fig. 279(a), and by a n

/ additional center-tapped drive winding on a single saturable trans-

! former in Fig. 279(b). The characteristic hysteresis loop of the auto-transformer used in the circuit of Fig. 279(b) is shown in Fig. 280. Transformer parameters such a s frequency, number of turns,

B

Fig. 280-Chnmcteristic hysteresis loop of artlo-trarrsforrrrer rtsed in circrtit o f Fig.

279(b)

and size and type of core material a re determined by the operating requirements f o r the circuit. Once the transformer has been established, a change in supply voltage results in a change in the operating frequency.

Switching is accomplished a s a result of the saturation of the transformer. When the slope of the hysteresis loop shown in Fig. 288 is small, the magnetizing inductance is small and the magnetizing current increases rapidly. This situation exists a s the loop i s traversed in a counter-clockwise manner from point 1 to point 2. From point 2 to point 3, the magnetizing current increases

very slowly because the magnetizing inductance is high. A t point 3, the core i s in saturation, and the magnetizing current again increases rapidly. A s the current continues t o increase (between points 3 and 4), the ON transistor comes out of saturation. When point 4 has been reached, the voltages across the primary windings of the transformer have dropped to zero, and the battery voltage is applied across the collector-to-emitter terminals of each transistor. The magnetizing current then begins to decay, and voltages of opposite polarity a re induced across the transformer. A t point 5, the magnetizing current has been reduced to zero, the second transistor is in saturation, and the first transistor has twice the battery voltage across i ts emitter-to- collector junction. This sequence of events is repeated during each half- cycle of the operation of t h e circuit, except f o r a reversal of polarity.

The approximate load line of the converter circuit of Fig. 279(b) is shown in Fig. 281. Many of the important transistor ratings can be

I

Vcc *vcc COLLECTOR-TO-EMITTER VOLTAGE

Fig. 281-Approxi~~late load Ible for con- verler circrtit show11 in Fig. 28716).

determined from this curve. F o r example, the collector-to-emitter sustaining voltage under reverse-bias conditions, VCRV(SUS), i s given by

where VCC is the collector-supply

RCA Trans

voltage and AVcc is the magnitude of the supply variations o r "spikes." The second-breakdown voltage limit ES/D for the transistor i s given by

Esln 2 34 (~1n)'Ll where p is the common-emitter forward-current transfer ratio, I n i s the base current, and L1 is the total series inductance of the transformer and the load reflected to the input. As mentioned previously, the collector-to-emitter saturation voltage Vca(sat) of the transistor should be low.

Fig. 282 shows the basic circuit configuration for a ringing-cholre dc-to-dc converter. In this converter,

Fig. 282-Rnsic circrrit cori/igrtr.ariorr o j rr ringirrg-choke dc-to-dc corrr.crrer.

n blocking oscillator (chopper circuit) is transformer-coupled to a half-wave-rectifier type of output circuit. The rectifier converts the pulsating oscillator output into a fixed-value dc output voltage.

When the oscillator transistor Q, conducts (as a result of either a forward bias or external drive), energy is transferred to the collector inductance presented by the primary winding of t ransfor~ner TI. The voltage induced across the transformer feedback minding connected to the transistor base through resistor Rn increases the conduction of Q1 until the transistor is driven illto saturation. The rectifier diode CRI in series with the secondary winding of transformer TI is oriented so that no power is delivered to

istor, Thyristor, & Diode Manual

the load circuit during this portion of the oscillator cycle.

With transistor QI in saturation, the collector current through the primary inductance of transformer TI rises linearly with time (-dildt = EIL) until the base drive supplied by the transformer feedback winding can no longer maintain QI in saturation. As the current through QL decreases from the saturation level, the voltage induced into the feedpack winding decreases, and transistor Q1 is rapidly driven beyond cutoff. The energy stored in the collector inductance (primary of transformer TI) is relased by the collapsing magnetic field and couplcd by the secondary winding of transformer TI, through rectifier diode CR,, to the load resistance Rt. and filter capacitor CI. The filter capa; citor stores the energy it reccivcs from the collector inductance. Whc~l no current is supplied to the load circuit from the oscillator (i.e., during conduction of transistor Q,), capacitor C1 supplies current to the load resistance Rr. to maintain the output voltage a t a relatively con- s tant value. The switching action of rectifier diode CRl prevents any decrease of the energy stored by capacitor Cl because of the negative pulse coupled from the oscilla- to r during the periods that transistor Q, conducts.

The operating efficiency of the ringing-choke converter i s low, and the circuit, therefore, is used primarily in low-power applications. In addition, because power is delivered to the output circuit for only a small fraction of the oscillator cycle (i.e., when Ql is not conducting), the circuit has a relatively high ril)l,le factor which substantially increases output filtering requirements. This converter, however, provides tlefi- nite advantages to the system designer in terms of design siniplicity and compactness.

In a converter, the c h a n ~ e in frequency of operation with supply voltage is not usually important hecause the output voltage is rectified and

1 Power Switching and Control

filtered. In a n inverter circuit, however, the frequency may be very important and is generally controlled by adjustment of the supply voltage. Typically, the dc supply voltage is controlled by means of a voltage regulator inserted ahead of the converter to stabilize the input voltage and a power amplifier following the converter t o isolate the converter from the effects of a varying load.

Inverters may be used to drive any equipment which requires an ac supply, such a s motors, ac radios, television receivers, or fluorescent lighting. In addition, a n inverter can be used to drive electromechanical transducers in ultrasonic equipment, such as ultrasonic cleaners and sonar detection devices.

Fig. 283 shows a block diagram of a typical inverter circuit. The output frequency is directly depend- en t on the induced voltage of the

VOLTAGE POWER REGULATOR SUPPLY

I I FEEDBACK I I

When t h e inverter is used t o provide dc-to-dc conversion; the square- wave voltage is usually applied to a full-wave bridge rectifier and filter.

Fig. 284 shows the configuration of the pudh-pull switching converter. The single saturable transformer controls circuit switching and provides the desired voltage transformation for the square-wave output delivered to the bridge rectifier. The rectifier and filter convert the square-wave voltage in a smooth, fixed-amplitude dc output voltage.

When the voltage VCC is applied to the converter circuit, current tends to flow through both switching transistors QI and Q,. It is very unlikely, however, tha t a perfect balance can be achieved between corresponding active and passive components of the two transistor sections; therefore, the initial flow of current through one of the transistors is slightly larger than t h a t through the other transistor. I f transistor Q, is assumed to conduct more heavily initially, the rise in current through i ts collector inductance causes a voltage to be induced in the feedback windings of transformer TI which supply the base drive to transistors Q1 and Q2. The base-drive voltages a re in

I i the proper polarity to increase the CONVERTER POWER current through Ql and to decrease

AMPLIFIER the current thruogh Q1. A s a result - of regenerative action, the conduction of Q, is r a ~ i d l y increased. and

i 1 Qa is dr>ven to cutoff.. Fig. 283-Block diagram o f typical , inverter circuit. The increased current through

Q1 causes the core of the collector converter transformer. The feedback shown samples this induced voltage and adjusts the output of the voltage

I regulator to maintain a constant induced voltage i n the converter and thus a constant output frequency. If a regulated output voltage i s not required, the second voltage regulator i s omitted.

The push-pull switch in^ inverter is probably the most widely used type of power-conversion circuit. F o r inverter applications, the circuit

I provides a square-wave ac output.

inductance t o saturate. The inductance no longer impedes the rise in current, and t'he transistor current increases sharply into the saturation region. For this condition, the magnetic field about the collector inductance is constant, and no voltage is induced in the feedback windings of transformer TI. With the cutoff base voltage removed, current is allowed to flow through transistor Qa. The increase in current through the collector inductance of this transistor causes voltages to


be induced in the feedback nlindings in the polarity that increases the current through Q.. and dccrrases the current through QI. This effect is aided by the collapsing magnetic field about the collector inductance of Q, thnt results from the decrease in currcnt through this transistor. Tlie fecdl~nck voltages produced by this col1al)sing field quicltl y drive Q , beyond cutoff and further increase the condurtion of Qz until the core of thc collcctor inductance for this transistor saturates to initiate a new cycle of operation. The square wave of voltage produced by the switcliing action of transistors Q1 and Q, is coupled hy transfo1,lner TI to the bridge rectifier and filter, ~vhich develop a snlootli, constant- amplitude dc voltage across tlie load resistance RT.. The small ripple produced by tlie square ~ v a v e greatly simplifies filter requirements.

Fir. 284-Basic circrrit corrfiprtr.nrio~r o j a sirtglc-trarxforrttcr prcsh-prrll slc~ifchirrg COr7-

verter.

I'ush-pull transformer-coupled converter .~ \%-it11 full-wave rectification provide power to the load continuously and are, therefore, well suited for low-impedance, high-power applications. Although not a s economical a s the ringing-choke design, thc push-pull configuration provides hinher efRcicncy and inlproved regulation.

In higl*l)ower driven invcrters, i t is not uncommon to use a Darling- ton connection to increase the current gain. However, this configura-

tion increases the VCR saturation of the output and does not permit a fast turn-off. The boosted 1)arliug- ton inverter shown in Fig. 285 uses two small additional transformer . windings (N:, and N,) and eliminates both problems.

Fig. 285-Boosted Darlitc.~forr irtverlcr wifh frrrrr-on dri1.c.

The polarity of Nn and N I is shown for Q, ON and Qs OFF. N,

.and N u a r e wound on core No. 1 which could be a motor o r other magnetic structure. The voltage developed across NI allo~vs QI to saturate fully while the voltage across NI allows Qa to have a reverse bias applied, thus helping the device to turn off. The diodes provide a path fo r reverse bias when the transistor turns off and hloclts voltage while the transistor is on; thus, they allow the driver transistors to control the output units.

Three-phase bridge invcrters f o r induction motors are usilally used to convert dc, GO-Hz, or 400-Hz input to a much higher frequency, pos- sibly a s high a s 10 kHz. Increasing frequency reduces the motor size and increases the horscpower-to- weight ratio, desirable features in military, aviation, and portable industrial power-tool markets. Fig. 286 shows a typical three-phase bridge circuit with base driving signals and transformer primary currents.


Tr is not allowed to saturate; therefore, the peak collector current through the transistor is determined

A principally by the value of the load impedance.

Because no two transistors a re perfectly matched, one of the transistors in the inverter circuit con-

i ducts more rapidly than the other i B when the power is turned on. This

transistor, Q? f o r example, tends toward saturation and causes positive voltages to appear a t the dotted ends of the transformers. Thus, there is a n effective positive feed-

C back tha t causes Q1 to switch off

l and Q, t o switch on. The voltage from the collector of QI t o the collector of Qa is then positive and equal to twice the collector supply voltage VCC. The voltage Vnm across

DC SUPPLY the feedback resistor RI\I is essen- (a) tially the product of the resistance

Rlb and the base current referred to

I the primary of T.. The voltage across TI is equal to 2 VCC - V~fb.

A t the beginning of the next half- cycle, the voltage across RII, in-

i creases very slowly with the slowly increasing magnetizing current through Tn. When TI reaches i ts saturation flux density, the magnetizing current increases very

1

(bl " ~ f b --4

O" TO 120' 120' TO 240° 240' TO 360° 1

Fig. 286-Three-plrasc bridge itrvcrter: (a) circtrif cortfigrtrcrliorr; (b) hnsc driving sig- rmls; ( c ) trarlsjorrrrer pririlary current

switchitlg.

Fig. 287 shows the schematic diagram of n two-transistor, two-transformer inverter circuit. A saturable base-drive transformer Ta controls the inverter switching operation. A linearly operating output trans- forlner TI transfers the output power to the load. The output transformer

Fig. 287-Two-trar1sistor/two-tra1tsformer ir~vertcr.

rapidly and causes a rapid increase in VlIrl,. A s a result, the voltage across Tn decerases rapidly and Q, comes out of saturation. The collector voltage of Qz then rises, and

200 RCA Transistor, Thyristor, & Diode Manual Power Switching and Control 201

~,cgcncr:ltivc action c:iuses Q , and Q.. to reverse states. As tlicse provesacs arc rcl)catcd drll,ing succeeding half-cycles, oscillations a re sustained.

SCR Inverters

SCR inverters offer an ellicient and ccononiical mcthod for conversion of direct current to allernating current. In the design of an SCR inverter, the fact tha t tlie SCIZ is 1)asically a "latching" device must be considcrcd. Anode current can be initiated a t any titlie by application of a signal of the proper polarity to the gate. However, thc gate loses control a s soon a s conduction begins, and current continues to flow, regardless of any gate signal which nlay be applied, a s long a s the anode remains positive. Special commutating circuitry is required t o turn off tlie SCR a t the proper time. A basic conlmutation circuit is shown in Fig. 288(a).

TVhen conduction is initiated 1)y application of a positive pulse to the gate, the voltage acroqs the SCR decreases rapidly a s current increases through it because of the voltage drop across the inductor L. The capacitor C charges th1.oug11 the resistor It i r ~ the polarity indicated. If the switch S is then closcd, tlie capacitor will be connected across the SCR in such a polarity that the anode of the SCR is suddenly driven negative. Conduction of the SCR then ceases as soon as the charge stored in the device has bcen removed 1)y the reverse recovery current.

The tinic ~.equired for the SCR to recover its for\vartl l~locliinp cap- nbility, a s shown in Fit. 288(b), limits the maxin~um operating frequency of the inverter. If the SCR 11as not recovered its bloclii~>g capability by the time the anode swings positive, continuous conduction results, and no ac power is generated. Sprcial fast-turn-off SCR's, which permit operation a t frequencies up to 25 ICIIZ, are currently available.

FORWARD- T R I G G E R E D ON B~~~~~

SWITCH

f O R W A R D C U R R E N T

SCR

I 'REVERS E R E C O V E R Y C U R R E N T

( b l

Fig. 288-C011tt11rc/ariott o j utl SCR: (u) bnsic cor~rtttrrtnriot~ circrrit; ( 6 ) ~olrage arrd

cfirrcrtl ivai~cforttt.~.

Fig. 289(a) shows the basic configuration for an inverter circuit. An ac output can he generated l)y alternately closing and opening switches S, and SL.. A more practical nlethod of producing an ac output is to replace switches S , and S, with SCR's, a s shown in Fig. 289(b). Capacitor C is used, a s previously described, to commutate SCRl and SCR, alternately.

Inverter circuits may use other methods of comnlutation. For example, auxiliary SCR's may be used to produce a negative co~nmutating pulse across the inverter SCR a t the proper time, o r a saturable reactor may be used in series with a capacitor to produce a c o m m u t a t i n ~ pulsc a t the proper time.

DC POWER

SUPPLY

s 2 - - AC (a1 OUTPUT

SUPPLY

AC OUTPUT

0 GATE S I G N A L

(b l

Fig. 289-111verfer circuits: (a) basic cott- figrrruliot~; (b) SCR Dlverter.

Fig. 290 shows a typical high- frequency SCR switching inverter; Fig. 291 shows the waveshapes across the SCR and the output of the transformer. For resistive loads, this inverter is capable of delivering 500 watts of output power a t an operating frequency of 8 kHz, and is provided with regulation from a no-load condition to full load. With proper output derating, this circuit can also accom~uodate inductive and capacitive loads. Under a capacitive load, the power dissipation of the SCR's is increased; under an inductive load, the turn-off time is decreased.

The inverter can be operated a t any optional frequency up to 8 kHz provided that a suitable output transformer is used and the timing capacitors are changed in the gate- trigger-pulse generator. A change in operating frequency, however, does not require any change in the comniutating components C1 and L1.

The operation of the SCR inverter is very similar to that of the two- transistor push-pull inverter except that external gate-trigger signals are required to initiate the SCR switching action.

Fig. 290 shows the two thyristors SCR, and SCR? connected to the output transformer TI. These thyristors a re alternately triggered into conduction by gate-trigger pulse generator to produce an alternating current in the primary of the power transformer. Fig. 291 shows typical operating wave forms f o r the SCR 1

inverter. The thyristors are commutated

by capacitor C,, which is connected between the anodes of SCRI and SCR,. The flow of current through the circuit can be traced more easily if i t is assumed that initially SCR, is conducting and SCR2 is cut off and t h a t the common cathode connection of the SCR's i s the reference point. For this condition, the voltage a t the anode of SCR? is twice the voltage of the dc power supply, i.e., 2E,,. The load current flows from the dc power supply through one-half the primary winding of transformer TI, inductor L2, SSRl, and inductor L,. When the firing current is applied to the gate of SCR,, this SCR turns on and conducts.

During the "ON" period of SCR?, the capacitor Cl begins to discharge through L., SCR-, SCRI, and Lz. In- ductors Ls and L function to limit the rate of rise of the discharge current d i /d t so that the associated stresses a re maintained within the capability of the device during the turn-on of the SCR. The effect of this control is to decrease the turn- on dissipation, which becomes a significant portion of the total device dissipation a t high repetition rates.

The discharge current through SCR, flows in a reverse direction, and af ter the carriers a r e swept out (and recombined) the SCRI switch opens (i.e., SCRl switches to the "OFF" state). At this time, the voltage across the capacitor C1, which


circuit adjusts the pulse width to maintain a fixed-voltage output current. Tiraveforms a re shown for a lightly loaded and a heavily loaded case.

AUTOMOBILE IGNITION SYSTEM

Fig. 293 shows a simple ignition system tha t uses an n-p-n transistor; performance curves fo r the circuit

Fig. 293-Solid-state acttorrrohilc i.yrririorr syslent.

are shown in Fia. 204. The advantages of this circuit include less maintenance of points and spark plugs, better perfornlance a t high engine speeds, and easier engine starting.

3 0 2 I

2 0

5 9 z 10 0 - Z w -

0 1000 2 0 0 0 3 0 0 0 4 0 0 0 5 0 0 0 ENGINE SPEED- r/n.ln

Fig. 294-Igniriorr voltage as a frtc~ctior~ of crrgitle speed.

CHARGING IMPEDANCE I-, CHARGING SCR

PULSE MODULATORS

Silicon controlled rectifiers a r e often used in pulse circuits in which the ratio of peak to average current is large. Typical applications include radar pulse modulators, inverters, and switching regulators. The limiting parameter in such applications often is the time required for forward current t o spread over the whole area of the junction. Losscs in the SCR a r e high, and a re concentrated in a small region until the entire junction area is in conduction. This concentraton produces undesirable high temperatures.

A typical SCR pulse modulator circuit is shown in Fig. 295; basic waveforms for the circuit a re shown in Fig. 296. The capacitors of the energy-storage network a re charged by the dc supply. The SCR

SUPPLY L SATURABLE REACTOR

r ( 0 E L A Y ~ I TO 2 ~ 5 1

I

Fig. 295-13nsic prtlsr ~r~odrtlntor circrtit.


is triggered by pulses from the gate- trigger generator No.1, and the energy-storage network discharges through an inductance and the load (transformer). Fig. 296 shows t h a t the discharge of the storage network is oscillatory; the half-sine- wave shape is characteristic of a single LCsection energy-storage network.

F o r turn-off, the load is "mismatched" to the discharge-circuit impedance so tha t a negative voltage is developed on the capacitor a t the end of the pulse.

A s a n example, the rise-time portion of turn-on i s defined a s the time interval between the 10-per- cent and 90-per-cent points on the current wave shape when the SCR is triggered on in a circuit that has rated forward voltage and sufficient

s ta te forward voltage of only 1 or 2 volts under such conditions. An interval many times greater than the turn-on time may be required before the forward voltage drop reduces to the steady-state level.

AC Power Controls

Thyristors have been widely ac- cepted in power-control applications in industrial systems where .high- performance requirements justify the economics of the application. Historically, in the commercial high-volume market, economic considerations have precluded the use of the thyristor. However, with the development of several families of thyristors by RCA designed specifically f o r mass-production economy and rated f o r 120- and 240-volt line operation, the use of these devices in controls for many types of small electric motors, incandescent lighting, and electric heating elements has been made economically feasible. The controls can be designed to provide good performance, maximum efficiency, and high reliability in compact packaging arrangements.

Basic Requirements

The simplest form of half-wave power control is shown in r i g . 297. This circuit provides a simple,

i F = 3 0 0 A non-regulating half-wave power control t h a t begins a t the 90-degree conduction (peak-voltage) point and

7~

Fi.r. Z ~ G - T I ~ ~ I I - O ~ I reqrtirerricr~ts for a prtlse-rnoditla~or SCR.

resistance to limit the current t o 8c+ rated values. F o r a 600-volt device, MIN

the end of the turn-on interval occurs (a1 MIN (b)

when the forward voltage drop Fifi. 297-Degree of cor~trol over condrrc- across the SCR is 60 volts. This riorz arrgles when ac resistive network is value contrasts with the steady- riscd ro trigxer (a) SCR's and (bJ triacs.


may be adjusted to within a few degrees of full conduction (180- degree half-cycle).

The half-tvave proportional control sl~own in Fig. 298 is a non-regulating circuit ~vhose function depends upon an RC delay network for ate phase- lag control. This circuit is better than simple resistance firing circuits because the phase-shifting characteristics of the RC networlc permit the firing of the SCR heyond the peak of the impressed voltage, resulting in small conduction angles. On the positive half-cycle of the applied voltape, capacitor C is charged through the networlc R, and R,,. When the voltage across capacitor C exceeds the gate- firing voltage of the SCR, the SCR is turned on; during the remaining portion of the half-cycle, a c power is agplicd to the load.

Fig. 298-SCR half-ivrrrr propor./io~rctl power corrlrol circrtil.

The delay in firing the SCR depends upon the time-constant network (R,, RI,, C) which produces a gate-firing v o l t a ~ e tha t is shifted in 1)liase with respect to the supply v o l t a ~ e . The amount of phase shift is adjusted by R,,. With maximum resistance in the circuit, the RC time constant is longest. This condition results in a large phase shift with a correspondingly smal l concluction annle. With minimum resistance, the phase shift is small, and essentially the full line voltage is applied to the load.

The control circuit uses the breakdown voltiige of a diac a s a threshold setting for fir in^ the SCR. The diac is specifically designed for handling the high-current pulses required to trigger SCR's. When the voltage across capacitor C

reaches the breakdown voltage of the diac, i t fires and C discharges through thc diac to i ts maintaining voltage. A t this point, the diac again reverts to its high-impedance state. The discharge of the capaci- to r from breakdown to maintaining voltage of the diac provides a current pulse of sufficient magnitude to fire the SCR. Once the SCR has fired, the voltage across the phase- shift network reduces to the forward voltage drop of the SCR for the remainder of the half-cycle.

Two SCR's are usually required to provide full-wave power control. Because of the bidirectional switch- in^ characteristics of triacs, however, only one of these tlevices i s needed to provide the same type of control. Fig. 299 shows three full- wave power controls tha t cnlploy thvristors.

In circuits of this type, a rapidly rising off-state voltage can .occur across the thyristor when the device changes from a conducting s tate to a blocking s tatc (commutates) . The influence of this dvldt stress on the operation of the power- control element is described below. Consideration is given only to those circuit applications that utilize a triac a s the main power-control element.

The dvldt stress in a circuit with a resistive load (such a s those just described) can be illustrated by consideration of a circuit with a G- ampere load tha t has a power factor close to unity. The load resistance in this circuit is 20 ohms for a source voltage of 120 volts. If the total circuit inductance is assumed to be 500 microhenries and the total triac and s tray capacitance is 500 picofarads, the circuit factor for the conduct in^ s tate is 0.99996, lagging. Thus, the load current laas the line voltage 1)y the small phase delay of approximately 25 microseconds. At the time tha t the triac commutates current, the line voltajie is 1.G volts. A t this time, a transient damped oscillation occurs a s a result of the interaction of the triac


junction capacitance and the circuit inductance. For the circuit parameter values given (R = 20 ohms, L = 500 n~icrohenries, and C = 500 picofarads), the frequency of oscillation is 3.2 x 10' Hz. Calculation of the maximum dvld t stress across the triac yields a value of 1.97 volts per microsecond. The voltage a t the time of commutation i s then 1.6 volts, and the maximum commutating dv ld t becomes 3.15 volts per micrc- second.

ClRCUlT

TRIAC

TRIGGER SUPPLY CIRCUIT

Fig. 299-Full-wave rhyrisror niotor corr/rol circrri/s rcsing (a) bridge rectifier and a sitigle SCR; (b) irrverse parallel SCR's;

(c) a rriac.

Thus, it can be seen that a definite dv ld t stress is imposed on the triac even when the load is primarily resistive. Because all resistive circuit configurations have some small inductance associated with them, a

commutating dvldt stress is produced in all resistive circuits. Fig. 300 shows a commutating dvldt waveshape f o r a resistive load of 6 amperes in a 120-volt triac control circuit.

Fig. 300-Triac principal voltage during cornn~uratior~ o f a resistive load.

The use of triacs fo r full-wave ac power control results in either fixed or adjustable power to the load. Fixed load power is achieved by use of the triac a s a static on-off switch which applies effectively all of the available line voltage to the load, o r by use of the triac in a fixed-phase firing mode which applies only the desired portion of the line voltage to the load. The latter method of operation i s but one point of a n infinite number of available points whicn can be attained by variable- phase firing operation.

Fig. 301- shows the current and voltage waveshapes produced when a triac is used to control a c power to a highly inductive load f o r on-off triac operation; Fig. 302 illustrates the waveshapes for phase-control operation. Because the load is highly inductive (wL>>R), the load current lags the line voltage by some phase angle e. When the current through the triac (i.e., the load current) goes to zero (commutates), the triac turns off. In static control operation, the triac is immediately turned on by continuous application, or re-application, of the ga te triggering signal; thus, this signal causes the triac to continue conducting f o r the desired number of successive half-cycles.

208 RCA Transistor, Thyristor, & Diode Manual Power Switching and Control 209

As shown in Fig. 301, a t time t,, the gate is opened and the triac continues to conduct for the remainder of tha t half-cycle of load current. A t the end of the half- cycle, commutation occurs and the triac is subjected to an off-state blocking voltage which has a polarity opposite t o the conducted current and a magnitude equal to the value of line voltage a t tha t instant. Be- cause the triac goes from a conducting state to a blocking s tate in a very short period of time, the rate of rise of off-state voltage is very rapid. This rapidly rising off-state voltage produces a dvldt across the main power terminals of the triac and can result in the triac going into conduction if the triac is incapable of \vithstanding the dvldt.

Fix. 301-Pri~tcipal voltage arrd crtrrcvrt for smric-sn3irch rriac operatior1 with an

bmdrcrive load.

Fig. 302 shows the waveshnpes produced for phase-control operation \vith an inductive load. The oscillations which are present on the pcalts of the voltage waveform are the result of interaction of the circuit inductance and capacitance. For this type of operation, the stress caused by the comnlutating dvldt is produced each time the current crosses the zero-axis and, therefore, occurs a t a frequency cqunl to twice the line-voltage frequency. If thc triac is incapable of sustaining the dv ld t which is produced, i t goes

PRINCIPAL I CURRENT I

PRINCIPAL I

I dv/dl I 1

1 1 9

Fig. 302-Prir~cipnl voltage arid crtrrerlt for phase-confro1 triac operariori witl~ art

irrductive load.

into a conducting s tate and remains in continuous conduction, supplying current to the load. This malfunc- tion is illustrated in Fig. 303.

Fig. 303-Principal voltage arrrl CrlrreIrt s h o w i ~ ~ g r~tal/ri~rctior~ of triac as a resrrlt o f corrrrr~urafiri,q dv/dr prod~tceci by arr

iridrtctive load.

Fig. 304(a) shows the circuit diagram of a series connection of voltage source, triac, and load. An equivalent circuit for this series connection is shown in Fig. 304(b). When the triac is in conduction, the triac

junction capacitance is shunted by a low-value, nonlinear resistance which minimizes the effect of triac capacitance. However, when the triac

nm

INDUCTIVE LOAD .-

TRIAC POWER

SOURCE

CIRCUIT

INDUCTIVE

yTl I b l

Fig. 304-(a) Series-circuit corlrlection of triac, Lldrtcfive load, and ac power source;

and ( b ) equivalent circuit.

goes out of conduction, the resistive component becomes very large and the equivalent triac shunting capacitance becomes significant. Because the circuit is basically a series RLC circuit, the voltage waveshape and the ra te of rise of voltage across the triac a t commutation a r e determined by the magnitude of source voltage and the circuit inductance,

, capacitance, and resistance. Thus the rising off-state voltage across

1 the triac can be a n overdamped, critically damped, o r underdamped oscillation.

The increased complexity of aircraft control systems, and the need for greater reliability than electromechanical switching can offer, has led to the use of solid-state power switching in aircraft. Because 400- Hz power is used almost universally in aircraft systems, triacs employed for power switching and control in such systems must have a substantially higher commutating dvldt capability than a re those employed similarly in 60-Hz systems. (The increase in commutating dvldt stresses

on triacs with increases in frequency was explained previously in the section on Thyristors.) RCA of- fe rs a n extensive line of triacs rated f o r 400-Hz applications.

Areas of application f o r 400-Hz triacs on aircraf t include:

1. Heater controls f o r food-warm- ing ovens and for windshield de- frosters.

2. Lighting controls f o r iristru- ment panels and cabin illumination.

3. Motor controls. 4. Solenoid controls. 5. Power supply switches Fig. 305 shows a low-current triac

in use in a simple, common, proportional-control application; the circuit consists of a single RC time constant and a threshold device. The trigger diac is used a s a threshold device to remove the dependence of the trigger circuit on

Fig. 305-Simple control circuit using a single time constant.

variations in g a t e t r igger characteristics. The circuit can provide sufficient control f o r many applications, such a s heaters and motor-speed and switching controls. Because of its simplicity, t h e circuit can be packaged in confined areas where space is a t a premium. Electrically, it dis- plays a hysteresis effect and initially turns on f o r resistive loads with a conduction angle which may be too large; however, i t provides maximum power output a t the full "on" position of the control potentiometer.

The hysteresis effect produced by a single-time-constant circuit can be reduced by addition of a resistor (Rs) in series with the trigger diac, a s shown by the dotted lines in Fig. 305. The series resistor reduces t h e capacitor discharge time and thus


providcs reduced time lag bccnl~se Fig. 308 shows a circuit in which of the diac turn-on-characteristics. a n SCR controls the triggering and

The circuit shown in Fig. 306 uses operation of a triac in a n integral- a double-time-constant control to cycle control circuit which is radio- improve on the performance of the frequency-interference free. A basic sinrle-time-constant control circuit. SCR gate-trigger o r gate-control hi$ circuit minimizes the hystcresis effect and allows the triac to turn on a t small conduction angles. The circuit has the a d v a n t a ~ e s of low hysteresis, bidirectional operation a t

Fig. 306-Corztrol circrrit rrsbrg a dortble tinle col~slatll.

small conduction angles, and continuous control up to the maximum conduction angle. In addition, the fixed resistor Rf can be replaced by a trimmer potentiometer for minimum control a t low conduction angles.

The circuit shown in F ~ R . 307 uses a neon bulb a s a t l~ res l~o ld device rather than the solid-statc diac. This circuit has the advantnges. of low hysteresis, bidirectional operation a t small conduction angles, and

Fig. 307-Cotrtrol circrtit rtsitrg o tleotr- brrlb rlrresl~old device.

continuous control up to the maximum conduction angle. 13ecause the neon-bulb threshold voltage is higher than that of a solid-stage diac, however, full 3GO-dcgree control may not bc achieved.

f t I,-/ CONTROL -

Fig. 308-Integral-cycle co~rrrol circ~ril.

circuit can be represented by a voltage source and a series resistance, a s shown in Fig. 309. The series resistance should include both the ex-

CIRCUIT RESISTANCE 2 L I

Fig. 309-Eqrtivaletrt gate trigger circrril.

tcrnal circuit resistance and the internal generator resistance. With this type of equivalent circuit, the conventional load-line approach to ga te trigger-circuit design can be used. With pulse-type triggering, i t is assumed initially t h a t the time required to trigger all SCR's of the same type is known, and that the maximum allowable ga te trigger pulse widths for specific peak gate power inputs a re to be determined. The magnitude of gate trigger current required to turn on an SCR of a given type can be determined from the turn-on characteristics shown in the section on Thyristors.

Power Switching and Control

The triae in Fig. 308 is not triggered as long as the SCR is on. When the SCR is turned off by renloval of the gate signal and application of a negative anode potential, the triac is triggered on a t the beginning of the next half-cycle. When the triac conducts, the capacitor charges up to the peak supply voltage and re- tains i ts charge to trigger the triac on in the next half-cycle. When the triac conducts in the reverse direction, the negative charge on the capacitor is held to a low value so tha t i t does not trigger the triac when the supply voltage reverses. If the SCR is still off, the triac repeats its conduction angle. I f the SCR is conducting, the triac does not trigger on, but remains off until the SCR is again turned off. This circuit provides the unique function of integral- cycle switching, i-e., once the triac is triggered on, i t completes one full cycle before turning off. This type of switching eliminates dc components present with half-wave control. The circuit also provides synchronous switching, i.e., the triac turns on a t the beginning of the cycle and does not generate RFI.

Light Dimmers

A simple, inexpensive light- dimmer circuit can be constructed wit11 a diac, a triac, and a n RC cllarge-control network. I t is important to remember tha t a triac in this type of circuit dissipates power a t the rate of about one wat t per ampere. Therefore, some means of removing heat must be provided to keep the device within i ts safe operating-temperature range. On a small light-control circuit such a s one built into a lamp socket, the lead-in \\,ire serves a s a n effective heat sink. Attachment of the triac case directly to one of the lead-in wires provides sufficient heat dissipation for operating currents up to 2 amperes (rms) . On mall-mounted controls operating up to G amperes, the combination of faceplate and wallbox serves a s an effective heat sink. F o r higher-power

controls, however, the ordinary faceplate and wallbox do not provide sufficient heat-sink area. In this case, additional area may be obtained by use of a finned face plate tha t has a cover plate which stands out from the wall so a i r can circulate freely over the fins.

On wall-mounted controls, it is also important tha t the triac be electrically isolated from the face .plate, but a t the same time be in good thermal contact with it. Although the thermal conductivity of most electrical insulators is relatively low when compared with metals, a low-thermal-resistance, electrically isolated bond of triac to faceplate can be obtained if the thickness of the insulator is minimized and the area for heat transfer through the insulator is maximized. Suitable insulating materials a re fiberglass tape, ceramic sheet, mica, and polyimide film. Fig. 310 shows two

MOUNTING ELECTRICAL INSULATION TAB

\ SOLDER /

ELECTRICAL INSULATION

Fig. 310-Exatnples o f isolated rno~tnling o f triacs.

exan~ples of isolated mounting for triacs in a TO-5 package and the new plastic package. Electrical insulating tape is first placed over the inside of the faceplate. The triac

21 2 RCA Transistor, Thyristor, & Diode Manua l

is then ~iiounted to the insulated faceplate by use of epoxy-resin cement.

Because the light output of an incandescent lanip dcpcnds upon thc voltage impressed upon the lamp filament, changes in the lamp voltage vary the brightness of the l a~np . When ac source voltages are used, a triac can he used in series with an incandescent lanip to vary the voltage t o the lamp by changing its conduction angle; i.e., the portion of each half-cycle of ac line v o l t a ~ e in which the triac condurts to provide voltage to thc lanq) filament. The triac, therefore, is very attractive a s a switching element in light-dimming applications.

To s\vitcli incandescent-lamp loads reliably, a triac must be able to withstand the inrush current of the lainp load. The inrush current is a result of the difference hetween the cold and hot resista~ice of the tungsten filament. The cold resistance of the tungsten filament is much lower than the hot resistance. The resulting inrush current is approximately 12 times the normal operating current of the lamp.

The simplest circuit t h a t can be used for liglit-dimming applications is shown in Fig. 311. This circuit uses a diac in series with the gate of a triac to minimize the variations in

Fig. 3I1-Sirr,qle-tirr1e-co11.rta11t liglrt- ditiirr~er circr~it.

gatc trigger characteristics. I11 applications where space is a t a pl.emium, the RCA-40431 or RCA-40432, \ ~ h i c h combines the functions of I)otli triac and diac, may be used. Cliangcs in the resistance in series wit11 thc ca-

pacitor change the conduction angle of the triac. Because of i ts simplicity, this circuit can be packaged in confined areas where space is a t a premium.

The capacitor in the circuit of Fig. 311 is charged through the control potentiometer and the series resistance. The series resistance is used to protect the potentiometer l)y limiting the capacitor charging current when the control potentiomctcr is a t i ts minimum resistance setting. This resistor may be eliminated if the potentiometer can withstand the peak charging current until the triac turns on. The diac conducts when the voltage on the capacitor reaches i ts breakover voltage. The capacitor then discharges through the diac to produce a current pulse of sum- cient amplitude and width to trigger the triac. Because the triac can be triggered with either polarity of gate signal, the same oper a t' ]on occurs on the opposite lialf-cycle of the applied voltage. The triac, therefore, is triggered and conducts on each half-cycle of the input supply voltage.

The interaction of the RC network and the trigger diode results in a hysteresis effect when the triac is initially triggered a t small conduction angles. The hysteresis effect is characterized by a difference in the control potentiometer setting when the triac is first triggered and when the circuit turns off. Fig. 312 shows the interaction between the RC net- worlr and the diac to produce the hysteresis effect. The capacitor voltage and the ac line voltage are shown a s solid lines. As the resistance in tlie circuit is decreased from its maximum value, the capacitor voltage reaches a value which fires the diac. This point is designated A on the capacitor-voltage waveshape. When the diac fires, tlie capacitor discharges and triggers the triac a t an initial conduction angle 8,. During the foi.mii~g of the gate trigger pulse, the capacitor voltage drops suddenly. The charge on the capacitor is sinaller than


when the diac did not conduct. As a result of the different voltage conditions on the capacitor, the breakover voltage of the diac is reached earlier in the next half-cycle. This point is labeled B on the capacitor- voltage waveform. The conduction angle O? corresponding to point B is greater than 8,. All succeeding conduction angles a r e equal to e2 in magnitude. When the circuit resistance is increased by a change

LlNE VOLTAGE

Fig. 3I2-M/avrfor111s sho~virtg irtteractiorz of coritrol t~etnlork atid trigger diode.

in the potentiometer setting, the triac is still triggered, but a t a snlaller conduction angle. Eventu- ally, the resistance in series with the capacitance becomes so great that the voltage on the capacitor does not reach the breakover voltage of the diac. The circuit then turns off and does not turn on until the circuit resistance is again reduced to allow the diac to be fired. The hysteresis effect makes the voltage load appear much greater than would normally be expected when the circuit is initially turned on.

The double-time-constant circuit in Fig. 313 improves on the perform-

V Fig. 313-Dorrble-tirrte-cowant light-

di~rrrrrer circiiit.

ance of the single-time-constant control circuit. This circuit uses a n additional RC network to extend the phase angle so that the triac can be triggered a t small conduction angles. The additional RC network also minimizes the hysteresis effect. Fig. 314 shows the voltage waveforms for the ac supply and the trigger capacitor of the circuit of Fig. 313. Be- cause of the voltage drop across R3, the input capacitor Cs charges to a higher voltage than the trigger capacitor CJ. When the voltage on CJ reaches the breakover voltage of the diac, i t conducts and causes the capacitor to discharge and produce the gate-current pulse to trigger the triac. After the diac turns off, the charge on C z is partially restored by the charge from the input capacitor C2. The partial restora- tion of charge on Cs results in better circuit performance with a minimum of hysteresis.

LlNE VOLTAGE TRIGGER-

CAPACITOR VOLTAGE

Fig. 3 l k V o l t a g e waveforms of double- time-constant cor~trol circuit.

Fig. 315 shows a lamp-dimmer circuit in which the use of a n RCA-CA3059 integrated-circuit zero- voltage switch in conjunction with a 400-Hz triac results in minimum RFI. (The CA3059 is described briefly in the section on Heater Con- trols. A detailed description of this integrated circuit is given in the manual on RCA Linear Integrated Circuits, Technical Series IC-42, in RCA Application Notes ICAN- 4158 and ICAN-6268, or in the Tech- nical Bulletin on the CA3059, File No. 397.


1

Fig. 315--Circrcit diagrarri for 400-Hz zero-voltage-switched lar~tp dintntcr.

Lamp dimming is a simple triac application t h a t demonstrates an advantage of 400-Hz power over 60 Hz. Fig. 316 shows the adjustment

Fig. 316-IVaveforrrts for 60-Hz phase- cot~rrolled lalrrp dirnrr~cr.

of lamp intensity by phase control of the 60-Hz line voltage. Because RFI is generated by the step functions of power each half cycle, extensive filtering is required. Fig. 31'7 shows a means of controlling power to the lamp by the zero-voltage- switching technique. Use of 400-Hz power makes possible the elimination of complete o r half cycles within a period (typically 17.5 milliseconds) without noticeable flicker. Fourteen different levels of lamp intensity can be obtained in this manner. In the circuit shown in Fig. 315, a line- synced ramp is set up with the desired period and applied to terminal

400-HZ LINE VOLTAGE

I I -1 I LINE- I I1

SYNCED -: / / ! + / 1 1 RAMP I I ! I / (1 -

I I I I \y I

Fig. 317-W'avejorma /or 400-Hz zero-voltage-switched lump dit~lr~tcr.


No. 9 of the differential amplifier within the CA3059. The other side of the differential amplifier (terminal No. 13) uses a variable reference level, set by the potentiometer R3. A change of the potentiometer set- t ing changes the lamp intensity.

In 400-Hz applications, i t may be necessary to widen and shift t h e CA3059 output pulse (which is typically 12 microseconds wide and centered on zero voltage crossing) to assure tha t sufficient latching current is available. The resistor Ra (terminal No. 12 t o common) and the capacitor C. (terminal No. 5 t o common) a r e used f o r this adjustment.

Heat Controls There a r e three general categories

of solid-state control circuits f o r electric heating elements: on-off control, phase control, and proportional control using integral-cycle synchronous switching. Phase-control circuits such a s those used for light dimming a re very effective and efficient f o r electric heat control except f o r the problem of radio-frequency interference (RFI) . In higher-power applications, the RFI is of such magnitude tha t suppression circuits to minimize the interference become quite bulky and expensive.

On-off controls have only two levels of power input t o the load. The heating coils a r e either ener- gized to full power or a re a t zero power, Because of thermal time constants, on-off controls produce a cyclic action which alternates between thermal overshoots and undershoots with poor resolution.

This disadvantage is overcome and RFI is minimized by use of the concept of integral-cycle proportional control with synchronous switching. In this system, a time base is selected, and the on-time of the triac is varied within the time base. The ratio of the on-to-off time of the triac within this time interval depends upon the power required to the heating elements t o maintain the

desired temperature. Fig. 318 shows the on-off ratio of t h e triac. Within the time period, the on-time varies by a n integral number of cycles from full ON to a single cycle of input voltage.

TRIAC OFF

-7RIAC ON-

TRIAC ON

LOW HEAT

7 ' p-TRlAC O F F 1

~ I M E BASE--^ HlGH HEAT

Fig. 318-Triac duty cycle.

One method of achieving integral- cycle proportional control is to use a fixed-frequency sawtooth generator signal which is summed with a dc control signal. The sawtooth generator establishes the period or time base of the system. The dc control signal is obtained from the output of the temperature-sensing network. The principle is illustrated in Fig. 319. As the sawtooth voltage increases, a level is reached which turns on power to the heating elements. A s the temperature a t the sensor changes, the dc level shifts accordingly and changes the length of time that the power is applied t o the heating elements within the established time.

When the demand for heat is high, the dc control signal is high and high power is supplied continuously

TRIGGER T LEVEL n r

CO-KTROL LOW HEAT DEMAND HlGH HEAT DEMAND

L O A D I A n A n ~ n V V v VOLTAGE

92LS-3081)

Fig. 319-Proportional-controller wave- sltapes.

RCA Transistor, Thyristor, & Diode ~ a n u a l

to the heating elements. When the denland for heat is completely satis- fied, the dc control signal is low and low power is supplied to the heating elements. Usually a system using this principle operates continuously somewhere between full ON ant1 full O F F to satisfy the demand for heat.

The RCA-CA3059 integrated-circuit zero-voltage switch is intended primarily a s a trigger circuit for the control of thyristors and is particularly suited for use in thyristor temperature-control applications. This multistage circuit employs a diode limiter, a threshold detector, a differential amplifier, and a Darlington output driver to provide the basic switching action. The dc supply voltage for these stages is supplied by an internal zener-diode-regulated power supply tha t has suficient current capability to drive external circuit elements, such a s transistors

and other integrated circuits. The trigger pulse developed by this circuit can be applied directly to the gate of an SCR or a triac. A built- in fail-safe circuit inhibits the application of these pulses to the thyristor gate circuit in the event tha t the external sensor fo r the integrated-circuit switch should be inadvertently opened or shorted. The CA3059 may be employed a s either an on-off type of controller o r a proportional controller, depending upon the degree of temperature regulation required.

Fig. 320 shows a functional block diagram of the CA3059 integrated- circuit zero-voltage switch. Any triac tha t is driven directly from the output terminal of this circuit should be characterized for operation in the I ( + ) o r I I I ( + ) triggering modes, i.e., with positive gate current (current flows into the gate for

1 +NTC=NEGATIVE TEMPERATURE COEFFICIENT

Note: Detailed descriptive informntion and the complete circuit dingrnm for the CA3059 nre given in the RCA 1,inenr Inteprated Circuits hlnnual, Technical Series IC-42. or in RCA r\pplirntinn Notex ICAN-4158 and ICAN-6268 and the RCA Technical Bulletin on the CA3059. File No. 397.


both polarities of the applied ac voltage). The circuit operates directly from a 60-Hz ac line voltage of 120 or 240 volts.

The limiter stage of the CA3059 clips the incoming ac line voltage to approximately plus and minus 8 volts. This signal is then applied to the zero-voltage-crossing detector, which generates a n output pulse during each passage of the line voltage through zero. The limiter output is also applied to a rectifying diode and an external capacitor tha t comprise the dc power supply. The power supply provides approximately 6 volts a s the dc supply to the other stages of the CA3059. The onloff sensing amplifier is basically a differential comparator. The triac gating circuit contains a driver f o r direct triac triggering. The gating circuit is enabled when all the inputs a re a t a high voltage, i.e., the line voltage must be approximately zero volts, the sensing-amplifier output must be "high," the external voltage to terminal 1 must be a logi- cal "1," and the output of the fail- safe circuit must be "high."

Fig. 321 shows the position and width of the pulses supplied to the gate of a thyristor with respect to the incoming ac line voltage. The CA3059 can supply sufficient ga te voltage and current to trigger most RCA thyristors a t ambient temperatures of 25°C. However, under worst- case conditions (i.e., a t ambient- temperature extremes and maximum tr igaer requirements), selection of the higher-current thyristors may be necessary for particular applications.

As shown in Fig. 320, when terminal 13 is connected to terminal 14, the fail-safe circuit of the CA3059 is operable. If the sensor should then be accidentally opened or shorted, power is removed from the load (i.e., the triac is turned off). The internal fail-safe circuit functions properly, however, only when the ratio of the sensor impedance a t 26"C, if a thermistor is the sensor, to the im-

Fig. 321-Timing relationship between the output pltlses of the CA3059 and the ac

line voltage.

On-Off Temperature Controller- Fig. 322 shows a triac and a CA3059 used in a n on-off temperature-controller configuration. The triac is turned on at zero voltage whenever the voltage Vs exceeds the reference voltage V,. The transfer characteristic of this system, shown in Fig. 323, indicates significant thermal overshoots and undershoots, a well- known characteristic of such a system. The differential or hysteresis of this system, however, can be further increased, if desired, by the addition of positive feedback.

Proportional Temperature Con- troller-For precise temperature- control applications, the proportional-control technique with synchronous switching is employed. The transfer curve for this type of controller is shown in Fig. 324. In this case, the duty cycle of the power supplied to the load is varied with the demand for heat required and the thermal time constant (inertia) of the system. For example, when the t,emperature set t ing is increased in a n "on-off" type of controller, full power (100 per cent duty cycle) i s supplied to the system. This effect results in significant temperature excursions because there is no an t ic i~a tory cir-

) pedance of the potentiometer R. is cuit to reduce the power gradually Fig. 320-F~rtlctional block diagram o f the integrated-circuit zero-voltnge switch. less than 4 to 1. before the actual set temaerature is

120 V A C 60 Hz

T


- Fig. 322-CA3059 018-off te~nperature controller.

Power Switching and Control 21 9

TEMPERATURE SETTING OVER

/ SHOOT DIFFERENTIAL

UNDER SHOOT

v TIME

Fig. 323-Tra~!sfer chnractcristics o f nrr on-off te~nperature-cor~lrol syste111.

TEMPERATURE

,-<DI:F~,~~NCTIAL

Before such a system is imple: mented, a time base is chosen so t h a t the on-time of the triac is varied within this time base. The ratio of the on-to-off time of the triac within this time interval depends on the thermal time constant of the system and the selected temperature setting. Fig. 325 illustrates the principle of proportional control. For this operation, power i s supplied to the load until the ramp voltage reaches a value greater than the dc control signal supplied to the opposite side of the differential amplifier. The triac then remains off f o r the remainder of the time-base period. As a result, power is "pro- ~or t ioned" to the load in a direct relation to the heat demanded by the system.

a

RAMP S I G N A L

- t-

TIME

I I F ~ R . 324-Trar1,~/er c /~nrnctrr;~t ics of a

proport ior~a~ tenlpcratrrre-cor~trol syster~l. O N ~ ~ 5 " . ~ ~ 5 0 ° ~ 0 ~ ' 7 5 0 ~ ~ j POWER POWER POWER OUTPUT OUTPUT OUTPUT

achieved. However, in a propor- POWER tional control technique, less power is s u ~ n l i e d to the load (reduced dutv cycl;)*as the error signal is reduced

d L I

TIME-- (sensed temperature approaches the Fig. 325-Prb~ciplts o f proporrio~rol set temperature). corztrol.

AC TO PIN 2

O V + = + G V I N

TO PIN 9 O OUTPUT

120 VAC 60 Hz

TO PIN 7

COMMON COMMON

CONNECTIONS RCA CA3059 REFER TO

Fig. 326-Ra118p generator.

For this application, a simple ramp generator can be realized with a minimum number of active and passive con~ponents. Exceptional ramp linearity is not necessary for proportional operation because of the nonlinearity of the thermal system and the closed-loop type of control. In the circuit shown in Fig. 326, ramp voltage is generated when the capacitor C? charges through resistors R, and Rn. The time base of the ramp is determined by resistors R1 and R2, capacitor C,, and the breakover voltage of the IN5411 diac. When the voltage across CI reaches approximately 32 volts, the

diac switches and turns on the 2N3241A transistor. The capacitor Ca then discharges through the collector-to-emitter junction of the transistor. This discharge time is the retrace or flyback time of the ramp. The circuit shown can generate ramp times ranging from 0.3 to 2.0 seconds through adjustment of R?. For precise temperature regula- tions, the time base of the ramp should be shorter than the thermal time constant of the system, but long with respect to the period of the 60-Hz line voltage. Fig. 327 shows a triac and a CA3059 connected for the proportional mode.

120 V A C

Hz

V - Fig. 327-CA3059 proporti011al tentperature controller.

220 RCA Transistor, Thyristor, & Diode 'Manual

Integral-Cycle Tcniperature Con- troller (No half-cycling)-If a temperature controller \vhicli i s corn- pletely devoid oT half-cycling and hysteresis is required, then the circuit shown in Fig. 328 may be used This type of circuit is essential for applications in which half-cycling and the resultant dc component coultl cause overheating of a power transformer on the utility lines.

In the circuit shown in Fig. 327, the sensor is connected between

trolled is low, tlie resistance of thc thermistor is high and an output signal a t terminal 4 of zero volts is obtained. The SCR, therefore, is turned off. The triac is then t r ic- gered directly from the line on positive cycles of the ac voltage. When the triac is triggered and supplies power to the load RI., capacitor C is charged to the peak of the input voltage. When the ac line swings negative, capacitor C discharges through the triac gate to trigger the

At-

- * FOR PROPORTIONAL OPERATION OPEN TERMINALS 10.11, AND 13, AND CONNECT POSITIVE RAMP VOLTAGE TO TERMINAL 13

Fig. 328-CA3059 itltegral-cycle tetuperatlcre controller in which half-cycling enecr is e l i t~~i t~a ted .

terminals 7 and 9 of the CA3059. This arrangement is required because of the phase reversal introduced by the SCR. TJTith this configuration, terminal 12 is connected to terminal 7 for operation of the CA3059 in the dc mode (ho~vever, the load is switched a t zero voltage). Because the position of the sensor has been changed for this configuration, the internal fail-safe circuit cannot be used (terminals 13 and 14 are not connectetf).

In the integral-cycle controller, wlien the temperature being con-

triac on the negative half-cycle. The diode-resistor-capacitor "slaving network" triggers the triac on negative half-cycles of the ac input voltage af ter i t is triggered on the positive half-cycle to provide only integral cycles of ac power to the load.

When the temperature being controlled reaches the desired value, a s determined by the thermistor, then a positive voltage level appears a t terminal 4 of the CA3059. The SCR then s ta r t s to conduct a t the beginning of the positive input cycle to shunt the trigger current away


fro111 the gate of tlie triac. The triac motors and perform switching, o r is then turned off. The cycle re- any other desired operating con- peats when the SCR is again turned dition tha t can be obtained by a by a reversal of the polarity of the switchinp action. Because most - applied voltage. motors a re line-operated, the triac

The circuit shown in Fig. 329 i s can be used as a direct replacement similar to the configuration in Fig. 328 except that the fail-safe circuit for electromechanical switches. A incorpor&.ed in the CA3059 can be very simple triac static switch f o r

-\ TYPE

ZN3241A

- *FOR PROPORTlONAL OPERATION OPEN TERMINALS I0,II;AND 13,

AND CONNECT POSITIVE RAMP VOLTAGE TO TERMINAL 13

Fig. 329-CA3059 irrtcgral-cycle fe~nperature controller that features fail-safe operation and no half-cyclitlg eflect.

used. In this latter circuit, the NTC control of a c motors is shown in sensor is connected between termi- Fig. 330. The low-current switch nals 7 and 13, and a transistor in- verts the signal output a t terminal 4 to nullify the phase reversal introduced by the SCR. The internal power supply of the CA3059 supplies bias current to the transistor. ~&IET~& CURRENT The circuit shown in Fig. 329 can z40vAc readily be converted to a true pro- ~ O H Z

portional integral-cycle tempera- J

ture controller sinlply by connection Fig. 33O-Sit?lple triac static switch.

of a positive-going ramp voltage to terminal 9 (with terminals 10 and controlling the gate trigger current 11 open). can be any type of transducer, such

a s a pressure switch, a thermal Motor Controls switch,- a photocell, or a magnetic

reed relay. This s i m ~ l e type of cir- Triacs and SCR's can be used cuit allows the motor to be switched

very effectively to apply power to directly from the transducer switch


without any intcmmedintc power switch or relay.

Triacs can also be used to change the operating characteristics of niotors to obtain many different speed and torque curves.

For dc control, the circuit of Fig. 331 can be used. By use of the tfc triggering modes, the tri:lc can be directly triggered from transistor

I I

Fir. 331-AC trinc sa~ifch corrrrol froi~r [fc ilrpllf.

circuits by either a pulse or continuous signal. A transistor series- switching regulator al)proacll can also be used t o control the armature current of a dc motor, a s shown in Fig. 332. Usually the transistor is full on or full off and the duration of the pulse (or the duty cycle) dctcrmines the motor spced. I t s typical high-power application is in the drive illotors of electric ve- hicles o r submarines.

Fig. 332-DC ~rrofor aritrarrrrcj co~r~rol .

I

L~

R A

I

Many fractional-horsel,ower motors a re series-wound "universal" motors, so named because of their ability to

operate directly from either ac or dc power sources. Fig. 333 is a schematic of this type of motor operated from an ac supply. Because most domestic applications today require 60-Hz power, universal motors a r e usually designed to have optimum performance characteristics a t this frcqucncy. Most univers:~l nlotors run faster a t a given dc voltage than a t the same 60-1Iz ac voltage.

The field winding of a universal motor, whether distributed or lumped (salient pole), is in series with the armature and external circuit, a s shown in Fig. 333. The current

> AMPUFIEU

INPUT FROM F E E D CONTROL AND SENSING > A

COMHUTATING DIODE

(IF REWlREDl

+

I- MOTOR TERMINALS 1

4,

EXTERNAL CONTROL

r BATfERY

OR ENERGY SOURCE

Fig. 333-Series-\vorr~ld rririvrrsal rjtoror.

-

through the field winding produces a magnetic field which cuts across the armature conductors. The :~ction of this field in opposition to the field set up by the a r ~ n a t u r e current sull- jects the individual conductors to a lateral thrust ~vhich results in armature rotation.

AC operation of a universal motor is possit)le because of the nature of its electrical connections. As the ac source voltage reverses every half- cycle, the magnetic field produced by the field winding reverses i ts direction sin~ultaneously. Because the armature windings a re in series s i t h the field windings through the brushes and commutating scgnienls, the current through the armature winding also reverses. Because both the magnetic field and a r ~ n a t u r e current are reversed, the direction of the lateral thrust on the armature windings remains constant. Typical performance characteristic ckuves for a universal motor a r e sho\vn in Fig. 334.


STALL NO LOAD

Fig. 3 3 k T y p i c a l performance curves for a ldtliversal n~otor.

One of the simplest and most em- cient means of varying the impressed voltage to a load on a n ac power system is by control of the conduction angle of a thyristor placed in series with the load. Typi- cal curves showing the variation of motor speed with conduction angle fo r both half-wave and full-wave impressed motor voltages a r e illustrated in Fig. 335.

CONDUCTION ANGLE-DEGREES

F k . 335-Typicul perfonnonce cltrves for a rtrliversal i ~ ~ o t o r with phase-angle control.

Half-Wave Control-There a r e many good circuits available fo r half-wave control of universal motors. The circuits a r e divided into two classes: regulating and non- regulating. Regulation in this in- stance implies load sensing and compensation of the system to prevent changes in motor speed.

The half-wave proportional control circuit shown in Fig. 336 is a non-regulating circuit tha t depends upon a n RC delay network for gate

phase-lag control. This circuit is better than simple resistance firing circuits because the phase-shifting characteristics of the RC network permit the firing of the SCR beyond the peak of the impressed voltage, resulting in small conduction angles and very slow speed.

UNIVERSAL MOTOR

I A I

Fig. 336-Half-wave motor control with no regulafio~t.

Fig. 337 shows a fundamental circuit of direct-coupled SCR control with voltage feedback. This circuit is highly effective f o r speed control of universal motors. The circuit makes use of the counter emf induced in t h e rotating armature because of the residual magnetism in the motor on the half-cycle when the SCR is blocking.

The counter emf is a function of speed and, therefore, can be used a s a n indication of speed changes a s mechanical load varies. The gate- firing circuit i s a resistance network consisting of R, and Rz. During the positive half-cycle of the source voltage, a fraction of the voltage is developed a t the center-tap of the potentiometer and is compared with the counter emf developed in the rotating armature of the motor. When the bias developed a t the ga te of the SCR from the potentiometer exceeds the counter emf of the motor, the SCR fires. AC power is then applied t o the motor f o r the


FI RI SCRl

C R 2

CRI

SUPPLY VOLTAGE

MOTOR VOLTAGE

remaining portion of the positive half-cycle. Speed control is accomplished hy adjustment of potentiometer R,. If the SCR is fired early in the cycle, the motor operates a t high speed because essentially the full rated line voltage is applied to the motor. If the SCR is fired later in the cycle, the average value of voltage applied to the motor is reduced, and a corresponding reduction in motor speed occurs. On the negative half-cycle, the SCR blocks voltage to the motor. The voltage applied to the gate of the SCR is a sine wave because it is derived from the sine-wave line voltage. The minimum conduction angle occurs a t the peak of the sine wave and is restricted to 90 degrees. Increasing conduction angles occur when the gate bias to the SCR is increased to allow firing a t voltage values which a re less than the peak value.

At no load and low speed, sltip cycling operation occurs. This type of operation results in erratic motor speeds. Because no counter EMF is induced in the armature when the motor is standing still, the SCR will

fire a t low bias-potention~eter settings and causes the motor to accel- erate to a point a t which the counter emf induced in the rotating armature exceeds the gate firing bias of the SCR and prevents the SCR from firing. The SCR is not able to fire again until the speed of the motor has reduced, a s a result of friction losses, to a value a t which the induced voltage in the rotating armature is less than the gate bias. A t this time the SCR fires again. Bc- cause the motor deceleration occurs over a number of cycles, there is no voltage applied to the motor; hence, the tern1 skip-cycling.

When a load is applied to the motor, the motor speed decreases and thus reduces the counter emf induced in the rotating armature. With a reduced counter emf, the SCR fires, earlier in the cycle and provides increased motor torque to the load. Fig. 337 also shows variations of conduction angle with changes in counter emf. The counter emf appears a s a constant voltage a t the motor terminals when the SCR is blocking.

Half-Wave Rlotor Control Limita- tions-If a universal motor is operated a t low speed under a hcavy mechanical load, i t may stall and cause heavy current flow through the SCR. For this reason, low-speed heavy- load conditions should be allowed to exist for only a few seconds to prevent possible circuit damage. In any case, fuse ratings should be carefully determined and observed.

Nameplate data for some universal motors are given in developed horsepower to the load. This mechanical designation can be converted into its electrical current equivalent through the following procedure.

Internal motor losses are taken into consideration by assigning a figure of merit. This figure, 0.5, represents motor operation a t 50-per- cent efficiency, and indicates that the power input to the motor is twice the power delivered to the load. With this figure of merit and the input voltage V.,, the rnls input


current to the motor can be calculated a s follows:

rms current = mechanical horse~ower x 746

0.5 V., For an input voltage of 120 volts, the rms input current becomes rms current = horsepower x 12.4 For an input voltage of 240 volts, the rms input current becomes rms current = horsepower x 6.2 The motor-control circuits de-

scribed above should not be used with universal motors that have calculated mms current exceeding the values given. The circuits will ac- comrnodate universal motors with ratings up to % horsepower a t 120 volts input and up to 1 % horsepower a t 240 volts input.

Full-Wave Universal and Induction Motor Controls-Fig. 338 shows a single-time-constant full-wave triac circuit which can be used a s a satisfactory proportional speed control for universal motors and with cer-

IZOVAC 0 R

24 OVAC * 60Ht

T Fig . 338-I~rd~tctiotl nrotor corltrol.

tain types of induction motors, such a s shaded pole or permanent split- capacitor motors when the load is fixed. No regulation is provided with this circuit. This type of circuit is best suited to applications which require speed control in the medium to full-power range. I t is specifically useful in applications such a s fans or blower-motor controls, where a small change in motor speed produces a large change in a i r velocity. Caution must be exercised if this type of circuit is used with induction nlotors because the motor may stall suddenly if the speed of the motor i s reduced below the drop-out speed for the

specific operating condition determined by the conduction angle of the triac. Because the single- time-constant circuit cannot provide speed control of an induction motor load from maximum power to full OFF, but only down to some fraction of the full-power speed, the effects of hysteresis described previously a r e not present. Speed ratios a s high a s 3:l can be obtained from the single-time-constant circuit used with certain types of illduction motors. Care must be taken to avoid continuous low-speed operation of induction motors in which sleeve bearings are used as improper lubrication will result.

Because motors a re basically inductive loads and because the triac turns off when the current reduces to zero, the phase difference between the applied voltage and the device current causes the triac to turn off when the source voltage is a t a value other than zero. When the triac turns off, the instantaneous value of input voltage is applied directly to the main terminals of the triac. This commutating voltage may have a rate of rise which can re- trigger the triac. The commutating dv /d t can be limited to the capability of the triac by use of a n RC network across the device, a s shown in Fig. 338. The current and voltage waveshapes fo r the circuit a r e shown in Fig. 339 to illustrate the principle of commutating dvldt.

In applications in which the hysteresis effect can be tolerated o r which require speed control primarily in the medium to full-power range, a single-time-constant circuit such a s tha t shown in Fig. 338 for induction motors can also be used for universal motors. However, i t is usually desirable to extend the range of speed control from full- power ON to very low conduction angles. The double-time-constant circuit shown in Fig. 340 provides the delay necessary to trigger the triac a t very low conduction angles with a minimum of hysteresis, and also provides practically full power

RCA Transistor, Thyristor, & Diode ManuaI

~ z ; f m L " j VOLTAGE 1

Fig. 339-IYavrdmpes o f cor~~rnrttatir~g dvldr characteristics.

to the load a t the minimum-resistance position of the control potentiometer. When this type of control circuit is used, a n infinite range of motor speeds can be obtained from very low to full-power speeds.

Fig. 34O-Dortble-tir1re-co11s~at1t r~rotor corttrol.

lleversing Rlotor Control-In many illdustrial applications, i t is necessary to reverse the direction of a motor, either manually o r by means of an auxiliary circuit. Fig. 341 shows a circuit which uses two triacs to provide this type of reversing motor control fo r a split-phase capacitance motor. The .reversing switch can be either a manual

switch or a n electronic switch used with some type of sensor to reverse the direction of the motor. A resistance is added in series with the capacitor to limit capacitor discharge current to a safe value whenever both triacs a r e conducting simultaneously. If triac No.1 is turned on while triac No.2 is on, a loop current resulting from capacitor discharge will occur and may damage the triacs.

The circuit operates a s follows: when triac No.1 is in the off state, motor direction is controlled by triac No.2; when triac No.2 reverts to the off s ta te and triac No.1 turns on, the motor direction i s reversed.

The triac motor-reversing circuit can be extended to electronic garage- door systems which use the principle fo r garage-door directioq.

MOTOR

RESISTANCE 120 OR vAck ::.2;k 24ovac L - - - - - - - 60 Hr

TRlAC T R l A C No.1

0 DIRECTION 1 CONTRCX I I

-0 DIRECTION CONTROL

Fig. 341-Reversirrg rrrotor cor~trol.

control. The system contains a transmitter and a receiver and provides remote control of door opening and closiag. The Idock diagram in Fig. 342 shows the functions required for a complete solid-state system. When the garage door is closed, the gate drive to the DOWN triac is disabled by the lower-limit closure and the ga te drive to the U P triac is inactive because of the s tate of the flip-flop. If the transmitter is momentarily keyed, the


receiver activates the time-delay monostable multivibrator so that i t then changes the flip-flop s tate and provides continuous gate drive to the U P triac. The door then continues to travel in the U P direction until the upper-limit switch closure dis- ables gate drive to the U P triac. A second keying of the transmitter provides the DOWN triac with gate drive and causes the door to travel in the DOWN direction until the gate drive is disabled by the lower

limit closure. The time in which the monostable multivibrator is active should override normal transmitter keying for the purpose of eliminat- ing erroneous firing. A feature of this system is that, during travel, transmitter keying provides motor reversing independent of the upper- o r lower-limit closures. Additional features, such a s obstacle clearance, manual control, o r time delay for overhead garage lights can be included very economically.

RECEIVER REVERSING

FLIP- FLOP

TRlAC

Fig. 342-Block diagram for remore- cotltrol solid-stare garage-door system.

DC Power Supplies

C power supplies convert the out- D put of a prime source, such a s a generator, to a form useful to the circuit to be powered. The supply of power usually requires rectification to change ac to dc, filtering to smooth out the ac ripple in the output of the rectifier circuit, and regulation to assure a constant output from the power supply in spite of variations in the input voltage and output load.

RECTIFICATION

The niost suitable type of rectifier circuit for a particular application depends on the dc voltage and current requirements, the amount of rectifier "ripple" (undesired fluctua- tion in the dc output caused by a n ac component) that can be tolerated in the circuit, and the type of ac power available. Figs. 343 through 349 show seven basic rectifier configurations. These illustrations include the output-voltage waveforms for the various circuits and the current waveforms for each individual rectifier in the circuits. Filtering of the output of the rectifier circuits is discussed later in this section. Ideally, the voltage waveform should be a s flat a s possible (i.e., approaching almost pure dc). A flat curve indicates a peak-to-average voltage ratio of one.

The single-phase half-wavc circuit shown in Fig. 343 delivers only one pulse of current for each cycle of ac input voltage. A s shown by the current waveform, the single rectifier conducts the entire current flow. This type of circuit contains

OUTPUT RECTIFIER VOLTAGE CURRENT

Fig. 343--Si~rg(e-~hase half-wove circliit.

a very high percentage of output ripple.

Fig. 344 shows a single-phase full- wave circuit t h a t operates from a

OUTPUT VOLTAGE

RECTIFIER CURRENT

Fifl. 344-Sirtgle-plrase fir//-wave circtiil 1tlit11 cetller-rapped power rratlsformer.

DC Power Supplies 229

center-tapped high-voltage transformer winding. This circuit has a lower peak-to-average voltage ratio than the circuit of Fig. 343 and about 65 per cent less ripple. Only 50 per cent of the total current flows through each rectifier. This type of circuit is widely used in television receivers and large audio amplifiers.

The single-phase full-wave bridge circuit shown in Fig. 345 uses four

OUTPUT RECTIFIER VOLTAGE CURRENT

can be used to supply twice as much output voltage a s the circuit of Fig. 344 for the same transformer voltage, o r to expose the individual rectifiers to only half a s much peak reverse voltage for the same output voltage. Only 50 per cent of the total current flows through each rectifier. This type of circuit is popular in amateur transmitter use.

The three-phase circuits shown in Figs. 346 through 349 are usually found in heavy industrial equipment such a s high-power transmitters. The three-phase Y half-wave circuit shown in Fig. 346 uses three rectifiers. This circuit has considerably less ripple than the circuits discussed above. In addition, only one- third of the total output current flows through each rectifier.

Fig. 347 shows a three-phase full-wave bridge circuit which uses six rectifiers. This circuit delivers twice a s much voltage output a s the circuit of Fig. 346 fo r the same transformer conditions. In addition, this circuit, a s well a s those shown in Figs. 348 and 349, has a n extremely small percentage of ripple.

In the six-phase "star" circuit shown in Fig. 348, which also uses

Fig, 345-SjnR(e-phase frtll-wave six rectifiers, the least amount of wirhour certrer-tapped power trartsfornrer the total output current (one-sixth)

(i.e., bridge-rectifier circuit). flows through each output rectifier. rectifiers, and does not require the The three-phase double-Y and inter- use of a transformer center-tap. I t phase transformer circuit shown in

WTPUT VOLTAGE

RECTIFIER CURRENT

Fig. 346-Three-phase "Y" half-wave circ~cit.


OUTPUT VOLTAGE

RECTIFIER CURRENT

Fig. 347-Three-plrrrse "Y" fitll-wave circrrir.

Fig. 349 uses six half-wave rcctifiers in parallel. This arrangement delivers six current pulses per cycle and twice a s ~ n u c h output current a s the circuit sho\vn in Fig. 346.

Table I V lists voltage and current ratios for the circuits shown in Figs. 3.13 through 349 for resistive or inductive loads. These ratios apply for sinusoidal ac input voltages. I t is

generally recommended t h a t inductive loads rather than resistive loads be used for filtering of rectifier current, except fo r the circuit of Fig: 343. Current ratios given for inductive loads apply only when a filter choke is used between the output of the rectifier and any capacitor in the filter circuit. Values shown do not take into consideration voltage drops

RECTIFIER CURRENT

OUTPUT VOLTAGE -

0 L--I ~ - 4

-.

Fig . 348-Six-plrasc "star" circrtit.

DC Power Supplies

n

Fig. 349-Three-phase "double-Y"

which occur in the power transformer, the silicon rectifiers, o r the filter components under load conditions. When a particular rectifier type has been selected for use in a specific circuit, Table IV can be used t o determine the parameters and characteristics of the circuit.

In Table IV, all ratios a r e shown a s functions of either the average output voltage En, or the average dc output current I.,, both of which are expressed a s unity f o r each circuit. In practical applications, the magnitudes of these average values will, of course, vary f o r the different circuit configurations.

FILTERING

Filter circuits a re used t o smooth out the ac ripple in the output of a rectifier circuit. Filters consist of txvo basic types, inductive "choke" input and capacitive input. Combina- tions and variations of these types are often used; some typical filter circuits a rc shown in Fig. 350.

The simplest of these filtering circuits is the capacitive input type. This type of filtering is most often uscd in low-current circuits in which

OUTPUT VOLTAGE

TTTWv\

and interphase-transformer circuit.

a fairly large amount of ripple can be tolerated. Such circuits a re usually single-phase, half-wave or full- wave. In this type of filter, t h e capacitor charges up to approximately the peak of the input voltage on each half-cycle tha t a rectifier conducts. The current into the load is then supplied from the capacitor rather than from the power supply until tlie point in the next half- cycle when the input voltage again equals the voltage across the capacitor. A rectifier circuit tha t uses a smoothing capacitor and the voltages involved a r e shown in Fig. 351. The input and output voltage wave- fors f o r this circuit a r e shown in Fig. 352.

Higher average dc output voltages and currents can be obtained from this type of circuit by the use of larger capacitors. A larger capacitor also tends to reduce the ripple. However, .care must be taken t h a t the capacitor is not so large t h a t excessive peak and rms currents cause overheating of t h e rectifier.

The next simplest filter is the inductive input filter. This filter performs the same function as a capacitive input filter in t h a t i t smooths


reducing the dissipation and current requirements in the high-voltage device Q1.

Ema, I m o x I mar ' P ~ ~ m o x ' 4

R I = €IN ( m a r )

IOUT (mar)

In the circuit of Fig-. 354, the niaximum power dissipated in Q, or QI. is approxinlately one-fourth of the power that would be dissip:ited in a conventional series-pass stage. The I~alance of the power is dissipated in resistor R,.

In many high-current applications including sc~, ies regulators, a L)nrl- ington configuration is utilized to improve the current gain, as shown in Fig. 855. A serious lin1it:ition of this method, however, is the high ponrer dissipated in the pass element because this dcvice cannot reach saturation.

0 ~ F E (TOTAL) = ~ F E ~ + ~ F E ~ + ~ F E ~ ~ F E ~

VCEZ = 'CE +VBE

Fi,y. .?55-l)c1rIi1rylr111 c < > ~ ! / ; , y r i r i r / i ~ ~ ~ r .

A typic:il auto~nol~i le volt:lgc- r r ~ u l a t o r cirruit for an auto with a 12-colt system is shown in Fig. 356. T~.ansistor (1: presents a varii111le re-

sistance in series with the field. If the battery is fully charged and the electrical loading is small (e.g., only from the ignition circuit), the 10-volt zener diode breaks down, turning (II on and Q2 off (i-e., high resistance). The consequent reduction in field current reduces the armature voltage E, so tha t the

ELECTRICAL SYSTEM

battery supplies the load current. If the battery requires c h a r ~ i n g , o r if the electrical load is heavy, then the lower terminal voltage is not suficient to break down the zener. For this condition, QI is off and Q.: is on full (i-e., driven into saturation). As a result, ficld current is h i ~ h , the armature voltage is high, and the alternator supplies current to the load and also charges the battery. Under normal operation, the transistor may be fully on, fully off, or some\vliere in between (i.e., on 11ut in the active region rather than in saturation). The actual transistor operating conditions depend on hattery condition and electrical load.

Shunt regulator circuits a re not a s rficient a s series regulator circuits for most applications, but they have the advantage of greater siniplicity. In the shunt voltage regulator cirruit shown in Fig. 357, the ciirrcnt through the shunt element consisting of transistors Q, and QI varies with c.hnnges in the load current o r the input voltage. This current variation

DC Power Supplies 235

is reflected across the resistance R, in series with the load so tha t the output voltage VO is maintained nearly constant.

Fig. 357-Typical shunt-regulator circrtit.

A third type of regulator, the switching regulator, was discussed previously in the section on Power Switching and Control. This type of voltage regulator is recommended for dc power-supply applications that require high efficiency, but only moderate regulation and noise im- munity.

SCR Regulated Power Supply Fig. 358 shows the circuit con-

figuration for a regulated dc power supply t h a t uses an SCR a s a series

pass element. This type of circuit is designed to provide approximately 125 volts, regulated to 2 3 per cent for both line and load. Ripple is less than 0.5 per cent rms.

The power supply is basically a half-wave phase-controlled rectifier. The capacitor C1 between the cathode and gate of the SCR charges up during half of each cycle and is discharged by the firing of the SCR. The firing angle of the SCR is ad- vanced o r retarded by the charging current flowing into the capacitor C,. Some of the current which would normally charge this capacitor i s shunted by the collector of the control transistor Q,. A s the current in the control transistor increases, current is shunted around the ,capacitor, through the ballast lamp I,, so tha t the capacitor charging time is increased. As a result, the firing angle of the SCR is retarded, and a lower output voltage results.

The controlling voltage on t h e control transistor is derived from both the dc output and from the line voltage in such a manner a s to provide load and line regulation respectively. The voltage-dependent

SCR TYPE

2N3228

BALLAST LAMP

V O R 1

Q I

- Fig. 358-SCR regulated power supply.


resistor (VDR?) in the base circuit of the control transistor decreases resistance for an increase in line voltage and thus increases hase current (and collector current) a s line voltage is increased. In addition, the lamp I , exhibits an increase in resistance with increasing line voltage, and, thus, tends to retard the firing

angle of the SCR. Changes in dc output voltage t h a t result from variations in load current a re fed back to the base of the control transistor by a voltage divider a t the input to the filter in the proper polarity to adjust collector current in a direction to compensate for changes in dc output voltage.

Testing and Mounting

T his section covers the testing and installation suggestions which a r e

generally applicable to all types of solid-state devices. Careful ob- servance of these suggestions will help experimenters and technicians to obtain the best results from solid- s tate devices and circuits.

TESTING The ability to determine the con-

dition of solid-state devices is a n important requisite f o r servicemen, experimenters, and others who a r e required to operate and maintain electrical equipment tha t employs such devices. Although thorough, comprehensive evaluations of solid- s tate devices are hindered by the limited amount of commercially available test equipment, simple techniques and circuits can be readily devised to provide golno-go type sf indications or t o measure significant characteristics of the devices. The following paragraphs outline various test methods, indicate some of the available test equipment, and describe simple test circuits tha t may be constructed f o r use in the test and evaluation of different types of solid-state devices.

, . ,' ,, - .

. .:' ' L ~ i p o l a r Transistors / Fig. 359 shows a golno-go tes t

circuit for bipolar transistors. The connections shown are for an n-p-n transistor. When the base resistor is connected to the negative terminal of the battery, the lamp should go out. For p-n-p transistors, the same results should be obtained with the battery polarities reversed.

n-p-n TRANSISTOR

Fig. 359-''Go/?1o-g0" test circuit for bipolar transistors.

A quick check of bipolar transistors can also be made prior to their installation in a circuit by resistance measurement with a conventional ohmmeter. The resistance between any two electrodes should be very high (more than 10,000 ohms) in one direction and considerably lower in the other direction (100 ohms or less between emitter and base or collector and base; about 1000 ohms between emitter and collector). It is very important to limit the voltage applied by the ohmmeter in such tests (particularly between emitter and base) so t h a t the breakdown voltages of the transistor will not be exceeded; otherwise, the transistor may be damaged by excessive currents.

In addition to the test to determine open or shorted elements described above, any comprehensive evaluation of bipolar transistors must include measurements of the two most important transistor characteristics, beta and leakage. Com- mercial transistor testers a r e avail-


able to perform these nieasurements.; Because there is no efficient substi- tutc way to cvaluntc thcse charnc- tcristics, a transistor tcster is :I mortliwhile instrunlent for use in the servicing of equipnlents t h a t employ bipolar transistors.

The beta, or conimon-emitter forward-current transfer ratio (li,,) , of a bipolar transistor expresses tlie gain char;lcteristics of the device. This characteristic can be dctermincd I)y use of nc or dc test voltages.

Collector-to-base leakage ( l , .~ , , , ) , measured with the emittcr open, is tlie critical leakage of both germanium and silicon transistors. How- ever. these two basic transistor tvnes

Transistor Tester Requirenients- The value of a transistor tcster depends on its dcsic~i and how i t is used. For accurate ineasure~iients of a wide range of transistor types, the tester must incorporate several specific design features. The more inl- portant considerations a re a s follows:

1. The capability to measure beta a t the collector-current level .best suited to the transistor type or its application. This capability should extend to the handling of devices ranging from small-signal rf transistors that have nominal collector currents of a few milliamperes to high-power types that have ratings . .

can display wide differences in their UI' One ampere' leakage values and in levels of ac- 2. The facility to provide beta

readings with a n accuracy of &5C/o ceptability. both in and out of circuit. ( I t should

A transistor tester should nieasure I,, however, that beta leakage directly in milliamperes or is directly affected bv the collector microamperes. current.)

IN CIRCUIT ZERO ADJUST

0-F CIRCUIT

RANGE

TKFxJeJ S ~ H

COAASE SIC REAR

R4" I

Fig. 360-Circrrir diagra~rr for RCA W T - 5 0 l A rmrisistor tesler.

Testing and Mounting 239

3. An adjustment which permits leakage currents to be "bucked out" before the beta measurement is made; otherwise, the beta reading may be upset by the leakage current. In the case of high-leakage germanium power transistors, the resultant beta reading may be significantly inaccurate. This rule applies to both in-circuit and out-of- circuit tests.

4. Means for calibrating the beta test fo r each transistor tested.

5. A facility for reading leakage current directly in values a s low a s one microampere.

The considerations listed above define the primary requirements of a good transistor tester. Other features a re desirable, of course, to make the tester completely reliable and easy to use.

Transistor Tester-A11 of the necessary and desirable features have been included in the RCA WT-501A Transistor Tester, a measurement instrument t h a t combines service speed and simplicity with laboratory-measurement qualities. Fig. 360 shows the overall schematic and Fig. 361 shows a photograph of the WT-501A transistor tester. This tester is designed to measure transistor collector-to-base leakage (Icnv), collector-to-emitter leakage (ICE,,), and dc beta. Collector current (Ic) is continuously adjustable up to 1 ampere in four ranges. The WT-501A can also be used for in- circuit beta tests of a transistor.

A 100-microampere meter movement is used in the measuring circuits f o r the various test functions. Precision resistors a re used t o assure accurate test results.

An N-P-NIP-N-P switch provides the proper bias polarity to the transistor. Two dual potentiometers provide coarse and fine adjustment of collector current (CAL) and in- circuit zero.

The instrument has two internal 1.5-volt 'ID"-size batteries. One battery i s used in n-p-n tests and the other is used in p-n-p tests. The batteries a r e also used during in-circuit

tes ts to provide voltage in reverse polarity to cancel the effect of circuit leakage.

Fig. 361- RCA W T - 5 O I A transistor tester.

Beta-measuring circuit: A simplified diagram of the dc-beta test circuit is shown in Fig. 362. Resistors Rb and R, serve both to establish

Fig. 3 6 2 S i m p l i f i e d beta-rneas~rrin~ circuit for 0-lo-100-~nilliampere range.

the collector current, and t o shunt the meter to the required sensitivity. Values for Rb and R, are a s follows:

240 RCA Transistor, Thyristor, & Diode Manual Testing and Mounting 241

R a ~ g e R b Rc 1 ~ i i A 1000 ohms 110 ohms

10 mA 110 ohms 10 ,011nis 100 niA 10 ohms 1 ohm

1 A 1 ohm 0.1 ohm

When tlie range switch is set to the CAL function, the meter is in the collector circuit. Collector current is deterniined by the value of the collcctor resistor fo r the particular range, and by the setting of the CAL control.

In the BETA function, the meter is switched to the base circuit. DC beta is defined a s the ratio of the steady-state collector current to the 1)asc current. Because the collcctor current is established a t a known value by the CAL adjustment, the hnse-current meter rentling can I)e interpreted in terms of dc beta for tlie transistor.

Irnc) measuring circuit: Ivrro is the culTrent flow, o r leakage, from the collector to the base with the emitter open. As shown in Fig. 363 1.5 volts is applied to the collector ant1 base of the transistoy, and the

n- p- n TRANSISTOR U N D E R TEST

Fig. 363-Sirrrplified II-IIO lest circ~rit.

meter is connected in the collector circuit. Collector-to-base leakage is indicated directly in n~icroamperes.

IIT:,, n~casuring circuit: Irl:o represents the lealtage from collector to cinitter, with the base open. F ~ E . 3fi1 sliows ;I simplified t l i agra~i~ of the Ic.,:,, test circuit. A volt;ige of 1.5 volts is applied to the transistor, ant1 the meter is connectcd in the collector circuit. The resistor slinnting the meter rrduccs the meter sensitivity to 10 milliamperes.

hIeasuremcnt of Icv:,, is normally

n- p-n

UNDER TEST

Fig. 364-Sirrrplifird I,,,:,, rest circrtit lor I-rrrillin~rrpo.e rartge.

made on the CAL position of the 1- milliampere range. If Irl:,, exceeds 1 milliampere, however, the range switch can be set to the 10-milliampere or 100-milliampere range a s necessary. Collector-to-cmittcr leakage is indicated in millia~i~peres, depending on the current range tha t is used.

In-circuit beta test: The test circuit used to measure in-circuit current gain is similar to t h a t used for out-of-circuit beta measurement. As shown in Fig. 365, the IN- CIRCUIT ZERO ADJUST control applies a voltage of reverse polarity

CAL

IN-CIRCUIT ZERO A D J

L-----_1

IN-CIRCUIT TRANSISTOR UNDER TEST

Fig. 365-Sir~rldifieri irr-circrrit /)etrr lest circrtif /or. O-to-100-1~1illiu111[~~~re rutr.re.

to the collector metering circuit. This voltage compensates f o r the collector-to-emitter leakage through the components in the circuit under test, and permits the meter to be set to zero.

The CAL adjustment and the metering circuit a re the same a s for out-of-circuit measurement.

The resistance of the measuring circuit is low in value so t h a t no significant loading effect occurs from the circuit being tested.

MOS Transistors

In the servicing of electrical equipment tha t employs MOS transistors, i t is readily determined that the test techniques required to measure the characteristics of these devices are not the same a s those used for bipolar transistors. An entirely new set of techniques, aimed specifically a t the unique properties of MOS transistors, is required. Simple golno-go types of tes t circuits, however, may still be used for detection of open or shorted devices.

The test circuit shown in Fig. 366 can be used to test n-channel depletion or p-channel enhancement MOS transistors for opens or shorts. The substrate and source of the device being tested should be connected to terminal No. 1, the gate should be connectcd to terminal 2, and the drain should be connected to terminal No. 3. If the MOS transistor is a dual-gate type, the gates a r e tested separately. For n-channel depletion types, if the lamp lights when the switch is open and does not light when the switch i s closed, the transistor is good. If the lamp lights with the switch in either position, the transistor is shorted. If tlie lamp remains off with the switch in either position, the transistor is open. For p-channel enhancement types, the reverse indications a re obtained.

In the section of this Manual on 310s Field-Effect Transistors, the susceptibility of these devices to

OHMS m I+ 6V f -

2

b OHMS

A

. l ' I - 6 . 4 9 LAMP 2v

60 mA

Fig. 366-"Go/no-go" test circuit for MOS.lrartsis/ors.

possible damage from the discharge of electrostatic charges was pointed out. Integral gate-protection systems used in certain types of dual- gate devices a re very effective in guarding against the effects of electrostatic charges. The following special precautions, however, a re neces- sa ry in handling MOS-transistors which do not contain integral-gate protection systems:

1. Prior to assembly into a circuit, all leads should be kept shorted together by either (a) use of metal shorting springs attached to the device by the vendor, a s shown in Fig. 367, or (b ) use of conductive foam such a s "ECCOSORB LS26" or equivalent. (ECCOSORB is a Trade Mark of Emerson & Cuming, Inc.). Note: Poly- styrene insulating "SNOW" can acquire high static charges and should not be used.

2. When devices a r e removed by hand from their carriers, the hand being used should be a t ground potential. Personnel handling MOS transistors during testing should ground themselves, preferably a t the hand or wrist.

3. Tips of soldering irons should be grounded.


4. Devices should never be inserted into or renloved from circuits with power on.

Fig. 367-Illrtsrrnrion shon,s slrortiri~ spririg for RCA hfOS field-eflecr tmir.rbtors illat do nor coritain the iriregral gnte protection. (Spring sl~ortld iior be reitroved rrrrtil after

the device is soldered iiito circuit.)

Silicon Rectifiers

In general, silicon rectifiers and most other types of solid-state diodes can be adequately tested by resistance nleasurements with a conventional ohmmetcr . (For procedures used in the testing of tunnel diodes, refer to RCA Tunnel Diodes, Technical hlnnual TD-30.) Resistance measurements a re talten in both the forward and reverse directions. The ratio of the "reverse" resistance reading to the "for\vardP' resistance reading should be greater than 10 to 1. For the forward- direction measurement, i t is important to assure tha t the forwnrd- voltage rating of the rectifier is greater than the voltage applied by the ohmmeter (the battery voltage of a conventional ohmmeter is 1.5 volts); otherwise, the rectifier may be damaged by excessive current. The front-to-back ratio of rectifiers can also be checked a t various current levels with the RCA WT-501A T

r

ansistor Tester described in the pnragraph on testing of Bipolar 'rransistors.

There a re a number of easily constructed golno-go types of test circuits tha t may be used to detect open o r shorted rectifiers. Several of these test circuits a re shown in the following paragraphs.

Fig. 368 shows a simple "go/no- go" test circuit fo r silicon rectifiers operating a t 120 volts. With the connection shown, the lamp operates at half-power. When the switch i s closed, the lamp should brighten if the diode under test is good. If there is no change in brightness when the switch is closed, the lamp was burn- ing a t full power with the switch open; in this case, the diode is shorted. If the lamp is out with the switch open but lights when the switch is closed, the diode is open.

Fig. 368-"Go/no-go" rest circuit for high-voltage silicor~ rcctificrs.

Fig. 369 shows a "go/no-go" tester fo r all silicon rectifiers in this Manual tha t operate a t low voltages

SILICON

R E C T ~ F ~ E R T Fig. 369-"Go/rio-go" test circrtit for low-voltage silicori rectifiers exclttdirig

types 134A nrld IN270.


except the 1N34A and 1N270. The test circuit fo r these two types i s shown in Fig. 370.

With a diode connected a s shown in Fig. 3G9 and with the polarity of the battery a s shown, the lamp should light; when the polarity of the battery is reversed, the lamp should not l i ~ h t . If the lamp lights regardless of the polarity of the battery, the diode is shorted; if the lamp does not light with either polarity, the diode is open.

When the anode of a 1N34A or IN270 diode is connected to terminal No. 1 in Fig. 370, the lamp should light if the diode is good; when the anode is connected to terminal No. 3 the light should go off. If the light remains lit regardless of the connection, the diode is shorted; if the light is off regardless of the connection, the diode is open.

OHMS P? TEST,

TYPE

NO 49 LAMP ZV

60 mA

Fi.r. 370-"Go/no-go'' lest circrtit for silicori rectifier types 1N34A and lN270.

SCR's and Triacs Similar tes t procedures and cir-

cuits may be used for testing SCR's and triacs. The triac, however, should b? tested f o r operation in all four .operating modes. F o r convenience of illustration, the test circuits described show only SCR's. Triacs tested in these circuits should be connected in one direction and then reversed f o r each test. I n addition,

the triacs should be tested for both negative and positive gate signals for each direction in which they a r e connected.

Fig. 371 shows a golno-go type of test circuit t h a t can be used to test thyristors t h a t operate directly from the line voltage. When the

Fig. 371-Simple test circrtit for SCR's.

switch is closed, a current of approximately 20 milliamperes flows through the 25-watt lamp, the 5600- ohm resistor, and the switch; this amount of current is not enough to light the lamp. When the switch is opened, the light should brighten t o approximately half maximum brightness. Under these conditions, the SCR should be triggered into operation (shunting the 5600-ohm resistor) on each positive half-cycle of input by the 20-milliampere current flowing in the gate-cathode circuit. If the lamp lights to full brightness, the SCR is shorted. If the lamp does not brighten regardless of the position of the switch, the SCR is open.

Fig. 372 shows a simple, inexpensive test circuit t h a t may be used t o evaluate the OFF-state voltage capabilities of thyristors,

Fig. 372-Test circuit used to determine dc forward- and reverse-voltage-blocking capabilities and leakage currerzt of thy-

ristors.

and for reverse-hloclting (SCR's) and leakage tests. Resistor R1 and capacitor Cl a re included in the test circuit to limit the rate of rise of applied voltage to the thyristor under test. Resistor R? limits the discharge of capacitor Cl through the thyristor in the event tha t the thyristor is turned on during the test. Resistor Rn provides a discharge path for capacitor C1.

F~E. 373 shows a simple tcst circuit that may be used t o determine the holding and latching currents of thyristors. For the holding- current tests, the value of potentiometer Ill is adjusted to apl)roxi- mately 50 ohms, and the spring- loaded push-button switch PB1 is momentarily depressed to turn on the thyristor. The value of R1 is

RCA Transistor, Thyristor, & Diode Manu-al

OHMS


Fi.0. 373-Test circlrit rrserl to dcterrnirie holding a~td latchir~g atrrcrrts of thpristors.

then gradually increased to the point a t which the tllyristor turns off.

F o r the latching-current test, the value of potentiometer R1 is initially adjusted so t h a t the main-terminal current is less than the holding level. The value of R, is then dc- creased, a s push-button switch PBr is alternately depressed and re- leased, until the thyristor latches on.

Fig. 374(a) shows a simple test circuit tha t may be used to determine the dvldt capability of a thyristor. The curves in Fig. 374(b) define the critical values fo r linear and exponential rates of increase in reapplied forward OFF-state voltage for a n SCR. The critical value for the exponential ra te of rise of for~vard voltage is the rating given

Fig. 374-Test circrrit nrrtl ivaveforr~rs rr.~cd to determine dv/dr capability of a thy-

ristor.

in the manufacturer's tes t specffi- cations. This rat ing i s determined from the following equation:

rated value of dv thyristor voltage (Vno) -- d t - RC time constant x 0.632

Fig. 375 shows a sinlple tcst circuit used to determine turn-on times of thyristors. The value of resistor R1 is chosen so t h a t the rated value

GATE TRIGGER PULSE

Fig. 375-Test circrrit artd i r~a~v for~r~s ctsrd to clctenribre trrrrr-oft rirne of thyristors.

of current flows through the thyristor. Turn-on time is specified by the thyristor manufacturer a t the rated blocliing voltage. I t is defined (for resistive loads) a s the time interval between 10 per cefit of the gate voltage and the period required for the current to rise to 90 per cent of its maximum value.

Fig. 376 sho\rVs a simple test circuit used to measure turn-off time. The circuit subjects the thyristor to current and voltage waveforms similar to those encountered in most typical applications. In the circuit diagram, SCRl is the device under test. Initially, both SCR's a re in the OFF-state; push-button switch SW1 is momentarily closed to s ta r t the test. This action turns on SCRl and load current flows through this SCR and resistor R?. Capacitor C1 charges through resistor R3 to the voltage developed across R2. If the second push-button switch SWs is then closed, SCR, is turned on.

FACTORY TESTED ~ " F B RATED

1 is2 CLOSED

5, CLOSED

Fig. 376-Test circrtit and voltage wave- forrrrs tcsed to deterrrti~~e turn-off times

o f tlryristors.

SCRl is then reverse-biased by the voltage across capacitor C,. The discharge of this capacitor causes a short pulse of reverse current to flow through SCRl until this device recovers i ts reverse-blocking capability. A t some time tl, the

anode-to-cathode voltage of SCRl passes through zero and s tar ts to build up in a forward direction a t a rate dependent upon the time constant of C1 and R2. The peak value of the reverse current during the recovery period can be controlled by adjustment of potentiometer R& If the turn-off time of SCRl is less than the time tl, the device will turn off. The turn-off interval tl can be measured by observation of the anode-to-cathode voltage of SCRl with a high-speed oscilloscope. A typical waveform is shown in Fig. 376.

The gate voltage and current required to switch a thyristor to i t s low-impedance s tate a t maximum rated forward anode current can be determined from the circuit shown in Fig. 377. The value of

2 - Fig. 377-Test circuit used to determine

gate-trigger-pulse reqttirements of thyristors.

resistor R2 is chosen so t h a t maximum anode current, a s specified in the manufacturer's current rating, flows when the device latches into i t s low-impedance state. The value of resistor R1 is gradually decreased until the device under test i s switched from i t s high- impedance state'. t o i ts low-impedance state. The values of ga te current and ga te voltage immediately prior to switching a re the gate voltage and current required to trigger the thyristor.

HEAT-SINK REQUIREMENTS

All solid-state devices a re temperature-sensitive, some to a greater degree than others. A s a result, the device temperature o r power dissipation must be kept below the maximum specified rat ing


either by limiting the input power requirements to maintain a limited power dissipation or by providing some external ~ n e a n s of re111oving the excess heat generated during normal operation. Generally, low- power semiconductor devices have sufficient mass and heat-dissipation area to conduct away the detrimental heat energy formed a t their semiconductor junctions. For higher- power devices, such a s power trxn- sistors, thyristors, and silicon rectifiers, however, a heat sink must be used.

Under steady-state conditions, the inasi~num dissipation c:~pability of a solid-state device that has a heat sinlc attached depends on the sun1 of (a ) the series thermal resistances from the semiconductor junction to the ambient, (b) the lnaxi~num junction temperature, and (c) the ambient temepratnre a t which the device is operated. The total ther~na l resistance of the device fro111 junction to ambient €)I-&

can be expressed a.s follows:

where (.t.l-r is the thermal resistance from the semiconductoi~ junction to the case of the device, ( 3 ~ - s is the thermal resistance between the device case and the surface of the

mon practice to use the chassis of the unit as a heat sink. In any case, the heat-dissipation capability of the heat sink is based on i ts thermal resistance Ha..+. The thermal-resistance value of the heat sink should be small enough t o obtain a power- dissipation capability, a s expressed in the above equation, t h a t exceeds the power-dissipation rating of the semiconductor device. For high- power devices, the interface thermal resistance 8 ~ - s between the semiconductor case and the surface of the heat sink can be maintained a t a low value (1 to 2°C per watt) by use of epoxy glue o r silicone grease.

Fig. 378 shows a useful nomograph for obtaining the physical dimensions of a heat sink a s a function of i ts thermal resistance. The data in this nomograph pertain to a heat sink tha t cools by convection and radiation and tha t is of natural bright finish of copper or aluminum. The heat-sink area is selected from the left-hand column and a line is drawn horizo~ltally froni this point. The value of thermal iesistance OS.., is read directly from the graph, depending on the type and thickness of the heat-sink material and the mounting position of the heat sink, either horizontal or vertical, with respect to t h e mounting board.

TRANSISTOR MOUNTING heat sink, and ns-\ is the t h e ~ m a l The collector, base, and resistance of the heat sink (from terminals of transistors can be con- its surface to the ainbient air).

The maximum dissipation capa- nected to associated circuit elements I,ility of a solid-state device by means of socltets, clips, or solder Prl(max) with a heat sinlc at- connections t o the leads or pins. If tached is given by connections a r e soldered close to the

lead or pin seals, care must be talten

\vhcre TJ(max) is the nlaxi~nuni junction tcniperature obtaincd from the manufacturer's data and T(anib) is the nmljient tenlpcrature.

Discrete heat sinks are sold coln- mercially in various size, shapes, colors, ant1 materials. It is also com-

to conduct excessive heat away from the seals, otherwise the heat of the soldering operation may craclc the glass seals and damage the transistor. When dip soldering is employed in the assembly of printed circuits using transistors, the temperature of the solder should be limited t o about 225 t o 250°C for a maximum immersion period of 10 seconds. Furthermore, the leads


Fig. 378-T1rcr111al rcsi.rta~rce as u firtrc~ior~ of heal-sink dirr~errsions (Nonlograph re- pririred fro111 ELECTRONlC DESIGN, A~cg. 16 , 1961).

should not be dip-soldered too close to the transistor case. Under no circu~nstances should the mounting flange of a transistor be soldered to a heat sink because the heat of the soldering operation may perma- nently damage the transistor.

Metal-Package Types In some transistors, the collector

electrode is connected internally to the metal case to improve heat-dissipation capabilities. More efficient cooling of the collector junction in these transistors can be accomplished by connection of the case to a heat sink. Direct connection of the case

to a metal surface is practical only when a grounded-collector circuit is used. .For other configurations, the collector is electrically isolated from the chassis or heat sink by means of a n insulator tha t has good thermal conductivity. Suggested mounting arrangement for RCA transistors supplied in hermetically sealed metal packages a re shown in detail in the section on Mounting Hardware.

For small general-purpose tran- 'sistors, such a s the 2N2102, which use a JEDEC TO-5 package, a good thermal method of isolating the collector from a metal chassis o r printed circuit board is by means of a beryllium oxide washer. The use


of a zinc-oxide-filled silicone compound between the washer and the chassis, together with a ~notlerate amount of pressure froni thc top of the transistor, helps to improve thermal dissipation. An alternate method is the use of a fin-type heat sink. Fir. 379 illustrates both types of n~ounting. Fin-type heat sinks a r e especially suitable when transistors a r e mounted in Teflon sockets which

SILICONE GREASE 8, 0 WASHER

I I I CHASSIS

FIN-TYPE HEAT SINK

Fig. 379-S~i,qgcstcd rnor~r~tir~g nrrallge- mert1.c for traruirtors bl JEDEC TO-5

package.

provide no thermal conduction t o the chassis o r printed circuit board.

For power transistors xvhich use a JEDEC TO-3 package, such a s the 2N3055, i t is recommended t h a t a 0.002-inch mica insulator o r a n ano- dized aluminum insulator having high thermal conductivity be used between the transistor base and the hcnt sink o r chassis. The insulator sliould extend beyond the mount in^ clamp, a s shown in Fig. 380. I t should be drilled or punched to provide both the two mount in^ holes and the clerancc holes for the elnit- ter and base pins. Burrs should be removed from both the insulatol. and the holes ill the chassis so tha t the insulatinls Inger will not be destroyed during mounting. It is also recommended tha t a n insulating washer

be used between the mounting screws and the chassis, as shown in Fig. 380, to prevent n short circuit between them.

For large power transistors, such as the 2N2876, which use a double- ended stud package, connection to the chassis o r heat sink should be made a t the flat surface of the transistor perpendicular t o t h e threaded stud. A large mating surface should be provided to avoid hot spots and high thermal drop. The hole fo r the stud should be only a s large a s neces- sa ry for clearance, and should contain no burrs or r i d ~ e s on its perim- eter. A s mentioned above, the use of a silicon grease between the heat sink and the transistor improves thermal contact. The transistor can be screwed directly into the heat sink or can be fastened by means,of a nut. In either ease, care must be taken to avoid the application of too much torque lest the transistor semiconductor junction be damaged. Al- though tho studs a r e made of relatively soft copper t o provide 1iip.h thermal conductivity, the threads should not be relied upon to provide a mating surface. The actual heat transfer must take place on the

u CLAMP

I

I HEAT SINK

> I ; +NYLON INSULATING WASHER - METAL WASHER

&I

F ~ E . 380-S11,qgcsted nlorolfirrg nrmtt,qcrrlcrrl for rrar~sistors ill JEDEC 7'0-3 packnge.


underside of the hexagonal par t of the package.

The use of a n external resistance in the emitter or colleclor circuit of a transistor is a n effective deterrent to damage which might be caused by thermal runaway. The minimum value of this resistance for low-level stages may be obtained from the following equation:

where E is the dc collector supply voltage in volts, Po is the product of the collector-to-emitter voltage and the collector current a t the desired operating point in watts, and &-A is the thermal resistance of the transistor and heat sink in degrees cen- tigrade per watt (8,-c + ~ C - S + 0 s - A )

Plastic-Package Types

RCA transistors a r e also available in two basic types of molded-silicone- plastic packages, which a r e supplied in a wide range of power-dissipation ratings and a variety of package configurations to assure flexibility of application. These types include the

RCA Versawatt packages for medium-power applications and the RCA high-power plastic packages. Each basic type offers several different package options, and the user can select the configuration best suited to his particular application.

Fig. 381 shows the options currently available f o r RCA Versawatt packages. The JEDEC Type TO- 220AB in-line-lead version, shown in Fig. 381(a), represents the basic style. This package features leads t h a t can be formed to meet a variety of specific mounting requirements. Fig. 381(b) shows a modification of the basic type tha t allows a Versa- watt package to be mounted on a printed-circuit board with a 0.100- inch grid spacing and a minimum lead spacing of 0.200 inch. Fig. 381(c) shows a JEDEC Type TO- 220AA version of the Versawatt package. The dimensions of this type of transistor package a r e such tha t i t can replace the JEDEC TO- 66 transistor package in a commercial socket o r printed-circuit board without retooling. The TO-220AA Versawatt package is also supplied with a n integral heat sink.

The RCA molded-plastic high- power packages a r e also supplied in several configurations, a s shown

Fi,q. 381-RCA Versawatt tmrrsislor packages: (a) JEDEC No. TO-220AB in-litre-lead vrrsiot~; (I>) corrfigrtmtior~ designed for rr~o~otfirrg of t printed-circrtit hoard.$; ( c ) JEDEC No. TO-220AA version, wlriclr tlray be ltsed as a replacrnrenf for JEDEC No . TO-66

metal packages.


Fig. 382-RCA hi,qh-power plartic frarrsisfor packa~es: (a) JEDEC N o . TO-219AB vcrsiorr, which r c p r ~ ~ ~ c r ~ t s the basic corrfi~rtrafion; (b) JEDEC No. TO-219AA versiori, which ttray he used as f l rc~plflce~~rrrrt for JEDEC TO-3 nrefal packages; (c ) cot~fig~tratiorz

drsigtied lor rirortrifirlg or1 printed-circuit boards.

in Fig. 382. The JEDEC Type TO- 219AB, shown in Fig. 382(a), is the basic high-power plastic package. Fig. 382(b) shows a JEDEC Type TO-219AA version of the high-po\vcr plastic package. With the addition of an NR193B top clamp, the TO- 219AA package can be used a s a direct replacement for the hermetically sealed JEDEC TO-3 package. The RCA high-power plastic package i s also available with a n attached header-case lcad, a s shown in Fig. 382(c). This three-lead package is designed for mounting on a printed-circuit board.

Itecommended mounting arrangements and sugcested hardware for the Versawatt transistors a re sIlo\vn in the section on blot lnt in~ llard- ware. The rectangular masher (NR231A) used in the mounting of these devices i s designed to minimize distortion of the mounting f l n n ~ e when the transistor is fastencd to a heat sink. Excessive distortion of the flange could cause damage to the transistor. The washer is particularly importxnt when the size of the mounting hole exceeds 0.140 inch (6-32 clearance). Larger holes a re needcd to acconllnodate insulating bushings; however, the holes should

not be larger than necessary to provide hardware clearance and, in any case, should not exceed a diameter of 0.250 inch. Flange distortion is also possible if excessive torque is used during mounting. A maximum torque of 8 inch-pounds is specified. Care should be exercised to assure tha t the tool used to drive the mounting screw never comes in contact with the plastic body during the driving operation. Such contact can result in damage to the plastic body and internal device connections. An excellent method of avoiding this problem is to use a spacer o r combination spacer-isolating bushing which raises the screw head or nut above the top surface of the plastic body. The material used for such a spacer o r spacer-isolating bushing should, of course, be carefully sc- lected to avoid "cold flow" and consequent reduction in mounting force. Suggested materials fo r these bushings a re diallphthalate, fibcrglass- filled nylon, or fiberglass-filled poly- carbonate. Unfilled nylon should be avoided.

Modification of the flange can also result in flange distortion and should not be attempted. Thc transistor should not be soldered to the heat


sink by use of lead-tin solder because the heat required with this type of solder will cause the junction temperature of the transistor t o become excessive.

The TO-220AA plastic transistor can he mounted in commercially available TO-GG sockets, rsuch a s UID Electronics Corp. Socket No. PTS-4 or equivalent. For testing purposes, the TO-220AB in-line package can be mounted in a Jetron Socket No. CD74-104 o r equivalent.

The recommended hardware and mount in^ arrangements f o r RCA high-power molded-plastic transistors a re also shown in the section on Mounting Hardware. These types can he mounted directly in a socket such a s the Industrial Hardware Corporation No. LST-1702-1 (or equivalent) o r they can be mounted in a standard TO-3 socket with the NR193B clamp. The precautions given for the Versawatt packages should also be followed in the mounting of the high-power molded-plastic packages.

The maximum allowable power dissipation in a solid-state device is limited by i ts junction temperature. An important factor to assure tha t the junction temperature remains below the specified maximum value is the ability of the associated thermal circuit to conduct heat away from the device.

When a solid-state device is operated in free air, without a heat sink, the steady-state thermal circuit is defined by the junction-to-free-air thermal resistance given in the published data on the device. Thermal considerations require that there be a free flow of a i r around the device and tha t the power dissipation be maintained below that which would cause the junction temperature to rise above the maximum rating. When the device is mounted on a heat sink, however, care must be taken to assure tha t all portions of the thermal circuit a re considered.

Operation of the transistor with heat-sink temperatures of 100°C or greater results in some shrinkage of

the insulating bushing normally used to mount power transistors. The degradation of contact thermal resistance is usually less than 25 per cent if a good thermal compound is used. (A more detailed discus- sion of thermal resistance can be found in the RCA Power Circuits Manual, Technical Series SP-51.)

During the mounting of RCA molded-plastic solid-state power devices, the following special precautions should be taken to assure efficient heat t ransfer from case to heat sink:

1. Mounting torque should be between 4 and 8 inch-pounds.

2. The mounting holes should be kept a s small a s possible:

'

3. Holes should be drilled o r punched clean with no burrs or ridges, and chamfered t o a maximum radius of 0.010 inch.

4. The mounting surface should be flat within 0.002 inchiinch.

5. Thermal grease (Dow Corning 340 or equivalent) should always be used (on both sides of the insulating washer if one is employed).

6. Thin insulating washers should be used (thickness of factory- supplied mica washers ranges from 2 to 4 mils).

7. A lock washer o r torque washer should be used, together with materials t h a t have sufficient creep strength t o prevent d e g radation of heat-sink efficiency during life.

A wide variety of solvents is available f o r degreasing and flux removal. The usual practice is to sub- merge components in a solvent bath for a specified time. From a reliabil- i ty standpoint, however, it i s extremely important tha t the solvent, together with other chemicals in the solder-cleaning system (such a s flux and solder covers), not adversely affect the life of the component. This consideration applies to all non- hermetic and molded-plastic components.

I t is, of course, impractical t o evaluate the effect on long-term tran-

RCA Trausistor, Thyristor, & Diode Manual

sistor life of all c l e a n i ~ ~ g solvents, whicl~ a re marketed under a variety of brand names with numerous ad- tliLives. Chlorinated solvcllts, g:.;leo- line, and other hydrocarbons cause the inner encapsulant to swcll and damage the transistor. Alcohols a re accepta1)le solvents and a re rccom- lnended for flux ren~oval whenever possil~le. Several examples of suitable alcohols a re listed below:

1. mcthanol 2. ethanol 3. isopropanol 4. hlcnds of the above When considerations such a s sol-

vent flammability a re of concern, se- lccted Freon-alcol~ol blends are usable when exposure is limitcd. Solvcnts such a s those listcd below should be safe when used for normal flux removal operations, but rare shoultl 11e talten to assure their suit- ability i l l tllc cleaning procedure:

1. Freon TE 2. Freon TE-35 3. Freon TP-35 (Freon P C )

Thcse solvents may be uscd foi- a maximum of 4 hours a t 25°C or for a n~axinluni of 1 hour a t 50°C.

Care must also be used in the selection of fluxes in the soldering of leads. Rosin or activated-rosin fluxes a re recomn-rendcd; organic fluxes a re not.

THYRISTOR MOUNTING

For most efficient heat sinks, inti- mate contact sliould exist I~etwecn the heat sinlc and a t least one-half of the package base. The thyristor p n c l t a ~ c can be mounted on the heat sink mechanically, with ~ l u e or epoxy adhesive, or by soldering. The JEDEC TO-48, TO-GG, and stud- rnountcd pacltagcs are mounted mechanically. In thesc cases, silicone grease shoultl be used hetwccn thc device and the heat sink to eliminate surface voids, prevent insulation build-un due to oxidation. and

exist a t the interface. To mi~linlize this interface resistance, an adhesive material with low thermal resistance, such a s IIysol*' Eposy I'atch Material No. GC or Wakefield" Delta Bond No. 152, or their equivalent, should be used.

Fig. 383 shows the special press-fit package used for some SCR's and triacs. Press-fit mounting depends upon an interference fit between the

thyristor case and the heat sinlc. As the thyristor is forced into the heat- sink hole, metal from the heat sink flows into the knurl voids of the thyristor case. The resulting close contact between the heat sink and thyristor case assures low thermal resistances.

A recommended mounting mcthod, shown in Fig. 384, shows press-fit knurl and heat-sink hole dimensions. If these dimensions are maintained, a "worst-case" condition of 0.0085 inch interference fit will allow press- fit insertion below the maximunl allowable insertion force of 800 pounds. A slight chamfer in the heat-sink hole will help center and guide the press-fit package properly into the heat sink. The insertion tool should be a hollow shaft having an inner diameter of 0.380 -c- 0.010 inch and a n outer diameter of 0.500 inch. These dimensions provide sufficient clearance for the leads and assure that no direct force is applied to the glass seal of the thyristor.

help conduct heat across the * P,oducrs of I-lyson Corporation, Olcan, face. Although glue or el)oV ad- New York, and Wakcficld Engineering. hesivc provides good bonding, a Inc.. Wakefield. Mazsachusclls. rcsncc- significant amount of resistance may tively.


The press-fit package is not restricted to a single mounting arrangement; direct soldering and the use of epoxy adhesives have been successfully employed. The press-fit case is tin-plated to facilitate direct soldering to the heat sink. A 60-40 solder should be used, and heat should only be applied long enough to allow the solder to flow freely.

800 LB. MAX.

the heat sink is preferable because i t is most efficient. Not only is the bond permanent, but the thermal resistance BC-B from the thyristor case to the heat sink is easily kept below 1 ° C per wat t under normal soldering conditions. Oven or hot- plate batch-soldering techniques a re recommended because of their low cost. The use of a self-jigging arrangement of the thyristor and the heat sink and a 60-40 solder pre- form is recommended. If each unit is soldered individually with a flame or electric soldering iron, the heat source should be held on the heat sink and the solder on the unit. Heat should be applied only long enough to permit solder to flow freely. Be- cause RCA thyristors a re tin-plated, maximum solder wetting is easily obtainable without thyristor overheating.

The special high-conductivity leads on the two-lead TO-5 package permit operation of the thyristor

Fig. 384-Sugges!ed n~ororting arrange,,lent at current levels that be 'On- for press-fit types. sidered excessive for a n ordinary

TO-5 package. The special leads can For the JEDEC TO-5, TO-8, and be bent into almost a n s confirrura-

low-profile packages, shown in. Fig. tion to fit any monting requirement; 385, soldering of the thyristor to however, they a r e not intended to

Fig. 385-JEDEC TO-5, TO-8, and low-profile packages.


tillce repeated bending and unl)end- ing. In particular, repeated bending a t t l ~ c glass should be avoidcd. The leads a rc not especially brittle a t this point, but the glass has a sharp edgc which produces a n excessively small radius of curvature in a bend made a t the glass. Repeated bending with a small radius of curvature a t a fixed point will cause fatigue and !)real(age in alnlost any material. For this reason, right-angle bends should be made a t least 0.020 inch from the glass. This practice will avoid sharp bends and maintain suficient electrlical isolation between lead connections and header. A safe bend can be assured if the lead is gripped with pliers close to the glass seal and then bent the requisite amount with the fingers, a s shown in Fig. 386. When the leads of a number of devices a re to

is given, together with approximate dimensions. The thyristors in the illustrations a re soldered t o the heat sink; if epoxy is used, a n additional thermal resistance OC-s of 1 to 2 ° C per wat t must be added to the thermal-resistance values sho\vn. The junction-to-case thermal-resistance value f o r the particular thyristor bcing used should be added to the values shown to obtain the over-all junction-to-air thermal resistance of each configuration. In the designs

INSULATION HEAT S I N K

I Fig. 386-Afcrhod of belrdi~rp lcnd.~ or1 @s-A=180C/W

~h).ristor packarc.

bc bent into a particular configuration, i t may be advantageous to use a lead-bending fixture to assure t h a t all leads a re bent to the same shape and in the correct place the first time, so t h a t there is no need for repeated bending.

RCA thyristors are also available in plastic packages. The inforniation ~ i v e n pl.eviously on the mounting

8 ~ - ~ ' 3 o ' C / w and handling of plastic-pacltage transistors is, in general, applicable to plastic-package thyristors a s well.

Typical Heat-Sink Configurations

Fig. 387 shows some typical heat- sink conf gurations tha t can be used with RCA tlbyristors in a TO-5 pacli- age. The thermal-resistance 0 s - A for Fig. 387-Typicnl frcot-si~rk confirtratiorrs each of the easily fabricated sinks for ~tsc witlr TO-5 pockage.


shown, electrical insulation of the heat sink from the chassis or equipment housing may be required.

Chassis-Mounted Heat Sinks

In many applications, i t is desirable and practical to use the chassis or equipment housing a s the heat sink. In such cases, the thyristor must be electrically insulated from the heat sink, but must still permit heat generated by the device to be efficiently transferred t o the chassis o r housing. This heat transfer can be achieved by use of the heat- spreader mounting method. In this method, the thyristor is attached to a metal bracket (heat spreader) which is attached to, but electrically insulated from, the chassis. The heatrsink configurations shown in Fig. 387 can serve a s heat spreaders, a s well a s the special clip shown in Fig. 388. (Triacs soldered to this heat spreader a re available from RCA a s type numbers 40638 and 40639; SCR's on this spreader a r e available a s type numbers 40656 and 40657.)

Electrical insulation may consist of material such a s alumina ceramic, polyimide film or tape, fiberglass tape, o r epoxy. The metal bracket itself has a low thermal resistance, and spreads the heat out over a larger area than could the thyristor case alone. The larger area in contact with the electrical insulation allows heat to transfer from bracket to chassis through the insulation with relatively low thermal resistance. Typical heat sinks, such a s those shown in Fig. 387, provide a much lower thermal resistance when used a s heat spreaders than when used a s heat sinks.

Heat-spreader dimensions can be varied over a wide range t o suit particular applications. F o r example, area o r diameter c.ul be increased, or shape changed, a s long a s the heat-tmnsfer area in contact with the electrical insulation is sufficient. An area of 0.2 square inch or more is usually desirable. The exact

thermal resistance of any heat spreader d e ~ e n d s on the heat- transfer area, type of metal used, type of insulation used, and whether the thyristor is fastened t o the heat

Of (CASE TO HEAT SINK1 3 TO 6. C/W MOUNTING

rELECTRICAL INSULATION TAB

SOLDER /

SELF-JIGGING TAB EPOXY

CCI F- IICCINC

Fig. 388-Self-jigging heat spreader.

spreader . with solder o r epoxy. Soldered construction yields a thermal resistance about 1 ° C per wat t less than t h a t obtained with epoxy. Alumina o r polyimide insulation provides a thermal resistance about 1 to 2 ° C per wat t less than tha t obtained with thermosetting fiberglass- tape insulation. The heat spreader can be made of any material with suitable thermal conductivity, such a s copper, brass, o r aluminum. Solderable plating for aluminum is commercially available.

RECTIFIER MOUNTING

The maximum forward-current ratings f o r RCA silicon rectifiers apply specifically for operation in free a i r (natural convection cooling). The average (dc) forward-current and the peak recurrent forward- current capabilities of these rectifiers a r e substantially higher than those shown in the maximum ratings when the rectifiers a re attached to heat sinks.

Rectifiers used for low-power applications normally do not require an external heat sink to dissipate the heat generated a t their p-n junctions. Most rectifiers in this category a r e packaged in the same


Fix. 389-Variorru paclinnr tlr,sktrc /or RCA silicorr rc~cti/ier.s. Z

sniall case used for tlie JEDlCC TO- 1 package. For medium-currcnt (1- to 2-ampcre) higli-voltagc applications, tlie rectifier is pacltagcd in a flange-case, axial-leal JEDEC DO-1 case. For higher-current applications, the DO-4 and DO-5 packages are used. These paclcagc configurations a re shown in Fig. 380.

Fig. 390 shows two suggcstcil nicthods for attaching the flange- case, axial-lead pacltage to a heat sinlr. The flange of the rectifier lnay also LC soldered directly to the heat sink, provided the flange temperature during soldering does not cx- cccd 253°C for a miximum pcriod of 10 seconds. Permanent damage to the rectifier may result if these limits a re exceeded.

The flcxiblc lends of some RCA rectifiers nre usually soldcred to the circuit elements. I t is desirable in all installations to provide some slaclr o r an expansion elbow in each lcad to prevent excessive tcrision on the lends. JIanual soldering should bc performetl carefully and quickly to avoid daninge to tllc rcc- ficr by cscessire heating. To minimize hcnting the rectifier junction durinq ~nanua l soldering, it is dc- si+nl>le t o grip the flcxiblc lcad 1 ~ -

i ~ ~ g soldcred between the case and Ihc soldering point with a pair of pliers.

RECTIFIER

ALUMINUM PLATE

\

SILICONE GREASE

RECTIFIER I n ,s%zzT SILIEONE IJ HEAT'SINK GREASE

Fig. 390--Sltgxc~t~(i rrrrthods jor attach- irtg recti/ier typrs IN2858A tl~rough

IN2864A to lrrnt sink.

I

' Testing and Mounting

When dip soldering is used in the assembly of printed circuits, the temperature of the solder should not exceed 255°C for a maximum immersion period of 10 seconds. The leads should not be dip-soldered beyond points, "A" and "B" indicated in Fig. 391.

POINT A k POINT B I

F ~ R . 391-Dingram s l ~ o ~ v i n ~ nreas hej7011d whiclt dip-soldering sltovld ,lot extend.

Fig. 392 shows the suggested mounting of the higher-current-type DO-4 and DO-5 packages. Mounting components of the type shown are furnished with each rectifier. With these mounting components, the in-

crease in thermal resistance ~ C - S

from the rectifier case to the heat- sink surface is approximately 3°C per watt.

Fig. 392-4lrggested nrortrzting arrange- rnenis for 0 0 - 4 and DO-5 packages.

RCAGLOBAL

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