Polar Transmitter

2276 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 44, NO. 9, SEPTEMBER 2009

Design of Highly Efficient Wideband RF PolarTransmitters Using the Envelope-Tracking Technique

Jerry Lopez, Student Member, IEEE, Yan Li, Student Member, IEEE, Jeremy D. Popp,Donald Y. C. Lie, Senior Member, IEEE, Chia-Chang Chuang, Kevin Chen, Stanley Wu, Tzu-Yin Yang, and

Gin-Kou Ma, Member, IEEE

Abstract—This paper discusses the design issues of highly effi-cient and monolithic wideband RF polar transmitters, especiallythe ones that use the envelope-tracking (ET) technique. Besidesfirst reviewing the current state-of-the-art polar transmittersin the literature, three focus topics will be discussed: 1) thesystem-on-a-chip (SoC) design considerations of the monolithicpolar transmitter using ET versus EER (envelope elimination andrestoration); 2) the design of highly efficient envelope amplifiercapable of achieving the high efficiency, current, bandwidth, accu-racy and noise specifications required for wideband signals; and3) the design of high-efficiency monolithic Si-based class E poweramplifiers (PAs) suitable for ET-based RF polar transmitters. Adesign prototype of a polar transmitter using ET and a monolithicSiGe PA that passed the stringent low-band EDGE (EnhancedData rates for GSM Evolution) transmit mask with 45% overalltransmitter system efficiency will be given; the simulated data ofthe entire polar transmitter system is also compared against themeasurement. Further investigations on how to solve the technicalchallenges to successfully implement linear and high-efficiencyET-based polar transmitter for broadband wireless applicationssuch as WiBro/WiMAX are also discussed.

Index Terms—DC-DC converter, envelope amplifier, envelope-elimination-and-restoration (EER), envelope-tracking (ET),monolithic RF SiGe power amplifier (PA), polar transmitters,power-added efficiency (PAE), radio-frequency (RF), switch-modePA, system-on-a-chip (SoC), wideband transmitter.

I. INTRODUCTION

I T IS critical to maximize the power-added efficiency (PAE)while minimizing the number of off-chip components for

any radio-frequency (RF) transmitter (TX) system, especiallyfor portable and space-borne wireless applications. Both thepeak and average PAE of a RF transmitter can heavily impactthe size of battery and heat dissipation, therefore playing thedominate roles on the final product’s form factor miniaturiza-tion, reliability, yield, and cost. The PAE of a RF TX systemcan be significantly enhanced by using a highly efficient non-linear RF power amplifier (PA), especially if novel linearizationtechniques (such as digital predistortion) can be applied on thesaturated PA [1]–[3]. Nonlinear switch-mode or saturated PAs

Manuscript received January 09, 2009; revised March 18, 2009. Current ver-sion published August 26, 2009.

J. Lopez, Y. Li, D. Y. C. Lie, and C.-C. Chuang are with the Departmentof Electrical and Computer Engineering, Texas Tech University, Lubbock, TX79409 USA (e-mail: [email protected]).

J. D. Popp is with the Boeing Company, Seattle, WA, USA.K. Chen, S. Wu, T.-Y. Yang, and G.-K. Ma are with the SoC Technology

Center, Industrial Technology Research Institute (ITRI), Hsinchu, Taiwan.Digital Object Identifier 10.1109/JSSC.2009.2022669

are more efficient than linear PAs, and they may be easier formonolithic integration on silicon since the driver stages do nothave to be linear. These nonlinear PAs are also less noisy andless sensitive to shifts in the operating point that might be causedby process-voltage-temperature (PVT) variations [3]. A very at-tractive TX architecture for PAE enhancement utilizes the polarmodulation architectures and nonlinear PAs, where the base-band signal is modulated in the amplitude/phase domain ratherthan the in-phase/quadrature (i.e., I/Q) domain. Polar modu-lation can operate in either a closed-loop or open-loop mode,and the resulting TX system is typically called as “polar trans-mitter” [4]–[6]. When one restricts the polar operation to thesignal modulator only but not extending it to the high-powerPA, the transmitter is called a “small-signal polar transmitter”or a “polar lite transmitter” [6]–[9]. In this case, the amplitudemodulation (AM) signal at the output of the I/Q modulator canbe read off from an AM detector or directly generated digitallyat the baseband and then fed into the voltage control input of avariable gain amplifier (VGA). The VGA will recreate the am-plitude modulation by varying the signal level to the input ofa linear PA. Therefore for the small-signal polar operation, theAM and phase modulated (PM) signals are recombined at theVGA. If, however, the AM and PM signals are recombined atthe high-power PA (often off-chip), the transmitter is called asa “large-signal polar transmitter” or a “direct polar transmitter”[4], [6], [9]–[11]. When a polar transmitter applies closed-loopfeedback control of both AM and PM portions of the signal fromthe high-power PA output, this closed-loop transmitter is calleda “large-signal closed-loop polar transmitter” or simply a “polarloop transmitter” [6], [11]. Strictly speaking, a polar loop TXsystem can use either large-signal or small-signal polar modu-lation and can have one or two feedback paths for AM and/orPM signals.

In general, the advantages of a large-signal polar transmitterinclude the improved PA efficiency, reduced wideband outputnoise floor (leading to elimination of bulky off-chip filters), andreduced sensitivity to PA oscillation with varying output loadimpedance over those of a linear PA system. Large-signal polartransmitters have recently demonstrated impressive resultsusing the 57-year-old Envelope-Elimination-and-Restoration(EER; i.e., Kahn’s technique) where the output power is directlymodulated by the drain/collector voltage of the highly efficientnonlinear PA (i.e., “plate modulation”) [1], [9]. In the past,polar transmitters were mostly used for high-power base stationapplications to effectively reduce heat dissipation; however,they have recently become very successful for wireless handsetTX design in volume production due to their significant better

0018-9200/$26.00 © 2009 IEEE

LOPEZ et al.: DESIGN OF HIGHLY EFFICIENT WIDEBAND RF POLAR TRANSMITTERS USING THE ENVELOPE-TRACKING TECHNIQUE 2277

Fig. 1. Block diagram of a typical direct-conversion transmitter using I/Q mod-ulation. Note a SAW filter is required near the final stage of the transmitter [10].

efficiency and lower cost [4]–[12]. One latest developmentis to apply the Envelope-Tracking (ET) technique to imple-ment monolithic large-signal polar transmitters for wirelessapplications, as excellent system efficiency and linearity hasbeen demonstrated, which shall be the focus of this paper[13]–[18]. We would like to begin our discussions of ET-basedpolar transmitters by briefly reviewing some state-of-the-artmonolithic RF polar transmitters for handset applications thatused EER-based polar TX architectures. Our ET-based polarTX research reported here builds upon the foundation of theseEER-based polar transmitters with important modifications andimprovement, as will be described and explained next.

II. REVIEW OF SOME STATE-OF-THE-ART POLAR

TRANSMITTERS IN THE LITERATURE

Fig. 1 shows a block diagram of a typical traditional di-rect-conversion transmitter using I/Q modulation. The directquadrature conversion has the advantages of supporting highdata rates and it is very flexible with respect to different mod-ulation formats, including non-constant envelope modulationmethods such as QPSK or QAM [19]. In this case, a RF-VGAis typically required to amplify the modulated RF signal tothe desired output power. A RF SAW filter or a duplex filteris always required to suppress the TX noise floor in the re-ceiver (RX) band to filter out noise several MHz away fromthe channel frequency to meet the strict ETSI requirement forhandsets. For a quad-band handset, since one SAW filter isneeded for each band in the TX path, it seriously adds cost andsize to the final product. Also, the traditional I/Q transmitterrequires the use of a linear PA, which suffers from lower PAEthan that of a saturated PA.

One major benefit for using the polar modulation approachfor handset operation is the elimination of the TX SAW filter.Depending on where the SAW filter is placed (before or afterthe high-power PA), it may not directly lower the PAE of the PA.However, the non-negligible insertion loss, cost and bulkiness ofthe off-chip SAW filters will most likely increase the total powerconsumption, form factors and the Bill of Materials (BOM) ofthe final products. Today’s polar transmitters successfully re-moved these SAW filters with low TX noise in the RX band. Forexample, over 100 million units of highly integrated large-signalpolar transmitters have been shipped for GSM/EDGE handsets[20]. This open-loop polar transmitter is a two-chip solution thatincludes a transceiver and a TX module, while the latter includesan integrated PA, switch, filter, DC-DC converter, voltage ref-erence and control circuitry, etc. The transceiver front-end in-tegrates VCOs inside a fractional-N synthesizer based digital

GMSK modulator with associated loop filters, system oscillator,DSP based digital channel filters, auxiliary DACs and other cir-cuits on-chip and with no IF and TX SAW filter [20].

One of the main challenges of implementing a polar trans-mitter for EDGE/GSM handset is the tradeoff between the RFspectrum and noise. The bandwidth of the circuits in the AMand PM paths for a polar modulator is much wider than thatof the composite signal due to the nonlinear I/Q to polar trans-formation [13]–[15]. For example, in the case of EDGE modu-lated signals, system simulations show that the phase and ampli-tude bandwidths need to be close to 3 MHz in order to meet theRF spectrum and EVM specs [7]. For EDGE exhibiting typicaloutput power of 27 dBm, the output noise must be lower than

144 dBc/Hz at 10 MHz and 156 dBc/Hz at 20 MHz offset(low band) to meet the RX band noise specs (for GSM mode,

150 dBc/Hz at 10 MHz and 162 dBc/Hz at 20 MHzoffset). To achieve low noise at these frequency offsets, narrowbandwidths are required and tight control of the phase and en-velope bandwidths are usually needed using calibrations priorto each burst.

As shown in Fig. 2(a) and (b), open-loop polar transmitterscan use feed-forward pre-distortion to linearize the AM-AM andAM-PM distortion in the PA. This enables elimination of powerdetectors, couplers, feedback circuits and many other functionsrequired to support feedback loops. Power consumption is lowerin open loop systems due to reduced complexity and less inser-tion loss after the PA. Fig. 2(b) shows the TX data from thebaseband is split into its AM and PM components. The PMcomponents are pre-distorted to compensate for the PLL loopfilter roll-off and are then combined with the channel selectionword of the fractional-N synthesizer, which provides the phasemodulation of the 8-PSK (Phase Shift Keying) signal for EDGEmodulation. The AM components are scaled according to thePA ramping control signal applied to the PA controller to mod-ulate the saturated PA output directly, which PA control blockoffers a highly linear amplitude transfer function between theinput control signal voltage and the output RF voltage. Carriersuppression is excellent as there is no upconversion in the TXsystem. This complete GSM/GPRS/EDGE radio system solu-tion achieves higher functionality at lower cost for cellular hand-sets than one would obtain using the traditional I/Q transmitterapproach.

Another commercially successful GSM/EDGE handsetlarge-signal polar transmitter with a saturated GSM type PAadopts the large-signal transmit polar loop architecture withseparate feedback control of the amplitude and the phaseof the PA output signal and it also meets all the GSM typeapproval requirements for both EDGE and GMSK in quadband as shown in Fig. 2(c) [6], [11]. The polar loop enablesthe radio to transmit both constant and non-constant envelopesignals through the same TX path to minimize the numberof external components as no pre-PA filters are required fornoise filtering. The T/R front end module includes separateGSM850/EGSM900 and DCS1800/PCS1900 PA blocks, a PAcontrol block, impedance-matching, an integrated coupler, aPHEMT switch and a diplexer for excellent EVM (Error VectorMagnitude) and phase error performance up to 6:1 VSWRwithout external isolator. In general, systems with feedback


Fig. 2. The block diagrams of: (a) the TX module for a large-signal polar transmitter operated in open loop for GSM/EDGE transmitter. The PA controller blockcan include a DC-DC converter and a low dropout regulator (LDO), etc.; (b) an open-loop large-signal polar transmitter (PM path shown in shaded blocks; adaptedfrom [10], [20]); and (c) a large-signal polar loop transmitter after Sowlati et. al. [6].

have increased complexity, and somewhat higher power con-sumption but can do well with VSWR changes. Other workusing polar transmitter for handsets utilizes small-signalpolar operation. In one case, the VGA is critical to both the

noise and output spectrum [7]; in the other, the discrete-timesampled system of all-digital PLL (ADPLL) handles thewideband PM path and the AM circuits are fully digital aswell [8].


Fig. 3. A simplified block diagram of a RF polar transmitter system using the ET technique, where �� and � �� representing the inputs to the EnvelopeAmplifier and the RF-PA, respectively.

Polar modulation schemes can be used for efficiency en-hancement only if the amplitude modulator is of very highefficiency operation. One drawback of polar transmitter, how-ever, is that the modulators will require higher bandwidth dueto I/Q to polar nonlinear transformation. Disadvantages alsoinclude a need to fully characterize the PA in compressedoperation mode across the output dynamic range, control oftime alignment of the AM and PM signal component paths,correction (generally through pre-distortion) of AM-AM andAM-PM distortion effects, etc. With these in mind, we areready to discuss the system design considerations next forimplementing polar transmitters using ET.

III. SYSTEM DESIGN CONSIDERATIONS OF RF POLAR

TRANSMITTER USING ET/EER

In practice, there are some serious technical hurdles to over-come to realize a RF polar transmitter, especially for portableapplications. For example, polar TX system is known to be sen-sitive to timing mismatch between the AM and PM paths [24].The group delay of the two signal paths must be matched to min-imize PA distortion, which is difficult to control across all PVTcorners [2]. The other major obstacle to overcome is the largerbandwidth required for the circuits in the polar TX system. Tobegin, the I-Q to polar transformation in the baseband is non-linear, which inevitably expands the bandwidth of both AM andPM output signals [21], [22]. Depending on the specific mod-ulation scheme and system specs, for EER-based polar TX theconstant-amplitude phase signal path may need to have roughlyten times larger bandwidth than the modulation input signal topass the TX transmission mask requirement and/or the Error-Vector-Magnitude (EVM) specs [15], [22], [23]. These issueshave been painfully resolved in the cellular industry by applyingcareful calibrations and predistortions to the PAs and driver am-plifiers, albeit with compromised transmitter efficiency as thehigher circuit bandwidth required for EER implementation willinevitably consume more power, offsetting the PAE improve-ment one can gain from using saturated PAs. We will, therefore,present in this paper that the major issues mentioned above canbe significantly relieved by using an ET-based polar TX archi-tecture (also known as hybrid-EER or H-EER architecture), asshown in Fig. 3 [15]–[17].

Compared to EER-based large-signal polar TX system, itis found that an ET-based polar TX system has the followingbenefits:

1) Higher gain at low output power. This is because the PA is“nearly saturated” but not always fully saturated as in thecase of EER;

2) Lower sensitivity to timing mismatch between the RFversus amplitude paths than EER [13], [17];

3) Lower bandwidth requirement for the envelope amplifierthan that in the case of EER [15]. This can be critical asthe efficiency of the envelope amplifier can be the limitingfactor for an ET/EER system;

4) Relaxed bandwidth requirement for the circuits used in theRF path versus that in the case of EER. Since this ET-basedpolar TX architecture uses the RF modulation signal as theinput to the saturated PA (instead of the PM signal), thePA needs to cover only the modulation signal bandwidth,making ET more suitable for broadband wireless applica-tions than EER [23]. The high bandwidth RF limiter re-quired for EER can also be power hungry, while it is notneeded for ET;

5) ET will have less RF feed-through signal that can appear asdistortion in the TX output. Since the drive signal is hard-limited for the case of EER, it has sidebands that can causeintermodulation distortion (IMD) by the large gate-drain orbase-collector capacitance in the final RF power device tocouple to the output to cause EVM issues [27]; ET is betterin this.

For reasons listed above, this ET architecture can be very at-tractive for implementing low power portable RF transmitterwith excellent PAE [14], [23]. In this section, we will focus ourdiscussions on the system timing mismatch analysis between theRF versus amplitude paths to compare an ET-based large-signalpolar system against an EER-based system. To analyze the in-fluence of timing mismatch distortion for both ET and EERsystem, we would first consider that the excitation is the tra-ditional two-tone, described by its complex envelope:

(1)

where is the baseband modulation frequency and isa square-wave with the same period as the modulation frequencyand an amplitude level of 1 [24]. If the PA is assumed ideal, theresulting output signal from the EER system will be a delayedversion of the input signal. Therefore in that case,

(2)


Fig. 4. Simulation of a two-tone RF signal input to an EER system with time-mismatch between amplitude and RF phase (� � �� MHz, time delay � �� ns).

Fig. 4 shows the simulation of EER delayed amplitude signal,phase (RF) signal and distorted output signal for a two-toneinput with a timing mismatch of 3 ns for an ideal PA, indicatingthis small timing misalignment produces significant errors foran EER system.

Next, we will use a linearized modified Cann’s model to an-alyze the influence of timing mismatch distortion for an ETsystem. The RF output amplitude signal of an ET system usingthe modified Cann’s model on a class E SiGe PA can be ex-pressed as [25]:

(3)

where is the dynamic collector voltage and it is a lineartime delayed function of the input envelope signal, i.e.

(4)

We assume an ideal pre-distortion is introduced to the PAsuch that it can eliminate any intermodulation and harmonicproducts generated by the nonlinearity in (3), the only distortionleft in the ET system will be caused by timing mismatch of theamplitude delay versus that of the RF signal path. As a resultof the assumed ideal pre-distortion, approaches infinity, andthe denominator of (3) becomes a “hard limiter” (i.e., linearizedCann’s model). Fig. 5 shows the ET system block diagram withlinearized Cann’s model. To only focus on the distortion causedby timing mismatch, we neglect all DC components and assumethe saturation value is always bigger than amplitude and can getthe delayed amplitude as [13]:

(5)

where

(6)

(7)

Fig. 5. ET-based polar transmit system block diagram with the linearizedCann’s model.

Fig. 6. Simulated time-mismatch distortion of an ET-based polar system be-tween amplitue and RF signal versus an EER system �� V� � �� ns�� MHz�.

We have used this modified Cann’s model with coefficientsextracted from our monolithic SiGe PA measurement data andplotted the simulated distortion in Fig. 6 for an ET-based polarTX system with a two-tone input signal. One can see clearly thatET is considerably less sensitive to timing misalignment thanEER. This is because the “ ” term is greater than one for thecase of ET and that decreases the timing mismatch , while thevalue of for an EER system is always unity as expressed by(2). Please note that is dependent of the PA bias voltagein our modified Cann’s model, which indicates that careful de-termination of optimum can achieve best timing alignmentfor ET [13].

Besides investigating the timing mismatch issue mathemati-cally, detailed system simulations of the overall ET/EER-basedpolar TX system (including digital DSP blocks and RF/analogcircuits) were performed in Agilent’s ADS environment forboth EDGE and WiBro applications. The system simulation


Fig. 7. System simulation block diagram that enables RF/analog and digital circuits co-design for the overall ET/EER-based polar TX system.

schematic consists of baseband waveform generator, modula-tors, envelope amplifier and our SiGe class E PA (SPICE circuitmodels based on measurement data), etc., as shown in Fig. 7(c).These simulation schematics enable powerful co-designingof the RF/analog and digital circuits for the optimal systemperformance. Note that to focus on the comparisons of ETversus EER, no pre-distortion was applied in the simulationresults presented in this paper.

Two principal sources of nonlinear distortion for ET/EERsystems are the finite envelope bandwidth and the differentialdelay between amplitude and RF/phase signal paths, as pointedout by Raab [26]. To compare ET and EER on the effects oftiming mismatch for EDGE signals, we used a delay line in-stead of realistic envelope amplifier in simulation to mimic thestatic timing mismatch between amplitude and RF/phase paths,as shown in Fig. 7(a). Fig. 8 presents the simulated TX outputspectrums for ET and EER with different static delays. As can beseen here, the ET topology is less sensitive to timing mismatchthan EER for a difference of 1/128 symbol time ( 25 ns). How-ever, when a timing mismatch of 1/64 symbol time ( 50 ns) isapplied, both ET and EER polar transmitters fail the EDGE TXspectral mask with this switch-mode PA. It is, however, impor-tant to notice that the EER output spectrum has a higher spec-tral growth than that of ET at the first knee of the EDGE TXmask (200 kHz to 400 kHz offset from the center frequency;i.e., from 880.6 MHz to 880.8 MHz in Fig. 8) and also with ahigher noise floor, making it more prominent to EDGE TX maskfailure. These results of complete RF/Analog/Digital systemsimulations corroborate with the mathematical derivations thatwe showed earlier. An ET-based polar system, therefore, doeshave higher resilience against static timing mismatches betweenthe amplitude and the RF/phase paths.

Aside from the static delay mismatch, finite bandwidth ofthe envelope amplifier can also cause non-negligible group de-

Fig. 8. Simulated EDGE TX output spectrums using (A) ET and (B) EER fromADS system simulations on polar transmitters with different static timing mis-matches between the amplitude path versus the RF/phase path. Note a realisticclass E SiGe PA SPICE model based on measured results was used in the systemsimulations.


Fig. 9. Simulated EDGE ET-based polar TX spectrums with a 600 kHz first-order envelope filter at the amplitude path, with and without the filter having thesame group delay mismatch at the RF path. The PA is modeled in SPICE usingmeasured class E SiGe PA data.

lays, which will further hamper the ability to combine phase andamplitude signals correctly in a polar TX system, resulting inout-of-band spectral growth [21], [27]. To investigate the effectsof differential group delays in an ET/EER system, a first-orderButterworth low-pass filter (LPF) was used in system simula-tion to model the finite bandwidth of the envelope amplifier[Fig. 7(a)], which mainly accounts for the group delay in theamplitude path. Fig. 9 shows the simulated output spectrum ofan ET-based polar TX system for EDGE, when the envelopesignal is filtered by this first-order LPF with a bandwidth of600 kHz. Once an additional filter with the same group delayresponse is added into the RF path as well [see Fig. 7(b)], theresults are plotted in Fig. 9 and they indicate that the TX outputspectral distortions can be reduced. These findings suggest thatin practice, a timing alignment algorithm should compensateboth static delay and group delay in an ET/EER based polar TXsystem. Predistortion techniques can further reduce the groupdelay mismatch to meet the EDGE TX spectrum mask, whichis outside the scope of discussions for this paper [21].

IV. DESIGN OF ENVELOPE AMPLIFIERS FOR ET-BASED

POLAR TRANSMITTERS

The design of high-efficiency envelope amplifier is criticalto the overall system efficiency for any large-signal polar TXsystem using either EER or ET techniques. This is because thetotal TX system efficiency is determined by the product of theenvelope amplifier efficiency and the PAE of the RF PA [16].While the design principle of ET-based large-signal polar trans-mitter has been known for many years, it has not been com-mercialized until recently, partly because of the difficulty in de-signing this envelope amplifier capable of tracking the rapidlyvarying envelope signal with high accuracy, wide bandwidth,large current, low noise, and yet still with excellent efficiency[28]. As an example, the envelope amplifier (i.e., sometimes justa DC/DC converter) in a GSM/EDGE large-signal polar handsettransmitter must be able to handle the non-constant envelopesof EDGE transmit bursts. Since the 8-PSK signal envelope haslarger than 17 dB peak-to-minimum ratio (i.e., PMR), currents

Fig. 10. Normalized envelope spectrums from simulations from (A) EDGEand (B) WiBro input signals. Over 99% of the envelope power resides within200 kHz for EDGE, and within 8 MHz for WiBro.

of the envelope amplifier will vary with the envelope signal, re-sulting in the output voltage to have overshoots and undershoots.During an EDGE burst, currents of the envelope amplifier canramp up and down from a few 100 mA to 2 A, causing signifi-cant load transient ripples and distortions of the amplitude signalthat can degrade system EVM [29].

Another difficulty in designing this circuit block for large-signal polar transmitter is that the switching frequency of a tradi-tional envelope amplifier (i.e., a DC-DC converter) is required tobe at least several times of the signal bandwidth, therefore chal-lenging to realize high efficiency for a broadband system. Forexample, to achieve low EVM for the WLAN 802.11a systemthat uses OFDM modulation with 20 MHz envelope bandwidth,the required switching frequency of a traditional DC-DC con-verter for an EER-based large-signal polar modulator will needto be 60–100 MHz to achieve acceptable EVM [15]. Thishigh switching frequency will introduce significant switchingloss for the envelope amplifier and therefore degrade the effi-ciency of the wideband TX system. Fig. 10 shows our simu-lated data for an EDGE and a WiBro system, respectively, whichindicates that over 99% of the envelope power resides within200 kHz bandwidth for EDGE while within 8 MHz for WiBro.The exact peak-to-average-ratio (PAR) has some dependency onthe number of carriers for an OFDM modulated system [30];the WiBro signal we used has 1024 and a PAR 10–12 dB.

To meet the stringent requirements for high efficiency, band-width, and slew rate with low ripples and large current handlingcapability, many groups have reported different approaches forthe envelope amplifier design, including buck converters usingpulse-width modulation (PWM) and delta–sigma modulation


Fig. 11. A split-band linear-assisted switch-mode envelope amplifier designused in this work [13]–[15], [37].

[31]–[33], multiphase converters [34], cascade of buck andboost converters [35], linear-assisted switch-mode converters[36], [37], etc. We show here a promising linear-assistedswitch-mode converter using split-band design in Fig. 11,which was recently detailed by Wang et al. [14], [15], [37]. Theenvelope amplifier consists of a wideband linear stage using anop-amp (PAE 30%) and a narrowband bulk converter witha large PMOS as the switcher stage (PAE 80%–90%). Thewideband high PAE envelope amplifier uses hysteretic currentfeedback control to realize the smooth power split between theswitcher stage and the linear stage. This split-band design al-leviates the switching requirements of the bulk converter sincethe fast transients of the envelope signal will be taken care of bythe fast op-amp, while the bulk converter will handle DC andthe slow moving transients. The feedback gain of the op-ampin our design is 2 with feedback resistors of 1000 ohm each.The 3 dB bandwidth of the op-amp is 190 MHz; hysteresisvalue of the comparator is 7 mV. The equivalent load resistorthat mimics the PA collector is obtained from measurement(roughly from 33–47 ohms), as it varies depending on theoperating regions of the switch-mode PA.

The envelope amplifier circuit in Fig. 11 has three differentmodes of operation [15]:

1) Linear Operation for Small-Signal Envelope (i.e., Small-signal operation): this is when the average slew rate ofswitcher current is much larger than the average slew rateof load current. In this case, the buck converter can fullysupport the load current; i.e., the switcher stage can provideboth DC and AC components of envelope signal.

2) Large-signal operation: this is when the average slew rateof the switcher current is much smaller than the averageslew rate of load current. The switcher stage can only pro-vide the DC component of envelope, while the AC com-ponent will be provided by the linear stage. The averageswitching frequency of the buck converter is the same asthe signal frequency which the current sensor can detect.

3) Matched slew-rate point: this is when the average slew rateof switcher current is equal to the average slew rate of theload current.

Fig. 12. The input, output and switching waveforms �� of the envelopeamplifier with EDGE envelope signal (a) and WiBro envelope signal (b) withADS SPICE simulation. EDGE average output envelope � �� V, WiBro av-erage output envelope � ��V; switching inductor values: 56 �H (EDGE) and15 �H (WiBro); � � �� V (EDGE) and 5.5 V (WiBro).

Fig. 12 plots the input, output and switching waveforms of theenvelope amplifier with both EDGE and WiBro envelope signalsfrom SPICE simulations in ADS. Notice the differences in thetime-scale and supply voltage for EDGE versus WiBro designs.This difference is primarily due to the much higher bandwidthand PAR of WiBro signals compared to those of EDGE, there-fore dictating a higher supply voltage to prevent clipping forWiBro. Fig. 13 is a picture of our discrete board design of thelinear-assisted envelope amplifier as shown in Fig. 11. Fig. 14shows the measured input, output, and switching voltage at thedrain of PMOS of this envelope amplifier with the EDGE enve-lope signal as input.

For both SPICE simulation and lab measurement of the split-band envelope amplifier design, we swept the inductance valueswith EDGE input signal (Fig. 15). The simulation results matchquite well against the measurement data, validating our ampli-fier design methodology. The efficiency of the envelope am-plifier for EDGE, however, is significantly higher than that ofWiBro (i.e., 60%–65% versus 45%–50%). The lower efficiencyof the WiBro envelope amplifier is expected, as the WiBro en-velope signal has a considerably higher PAR and signal band-width. This in turn demands a higher supply voltage than EDGE(i.e., 5.5 V versus 3.6 V) to balance the efficiency versus signalfidelity. Therefore, the overall power consumption of the WiBroenvelope amplifier is significantly higher than that for the EDGEapplication. We also found that the efficiency of this envelope


Fig. 13. A picture of our discrete board design of the linear-assisted envelopeamplifier as shown in Fig. 11.

Fig. 14. Measured input (top), output (middle) and switching waveforms(bottom) of the envelope amplifier (shown in Fig. 11) with an EDGE envelopeinput signal. EDGE average output envelope � 2.1 V, � � 3.8 V.

amplifier becomes insensitive to the inductance values once it ishigher than 33 H for EDGE, while only 8 H for WiBro.

Fig. 16 plots the measured efficiency versus the supplyvoltage at several load resistances for the split-band envelopeamplifiers designed for both EDGE and WiBro envelope sig-nals. An interesting trend can be seen from Fig. 16 that asthe supply voltage decreases, the efficiency always increases.This suggests that if clipping can be avoided by some smartde-cresting algorithms or novel circuitry, one can successfullylower the operating supply voltage of the envelope amplifier toconsiderably increase its efficiency [14], [38]

Note that the switching stage does generate noise. Thisnoise will make the voltage across bigger than thehysteresis value, which will false trigger the switcher. In orderto avoid this from happening, the value of should bevery small. However, switching noise will be present at output

Fig. 15. (a) Simulated and measured efficiency of envelope amplifier withEDGE envelope signal versus different inductance values at different loadresistances; � � �� V. (b) Measured efficiency of envelope amplifier withWiBro-like envelope signal; � � �� V.

and selecting an op-amp with large bandwidth can help toreduce it (see Fig 14; our measured “Env. Output” waveformappears to be rather clean). Therefore, careful system-levelmeasurements must be conducted to verify if the split-banddesign can meet the very tough spec of TX-induced switchingnoise in the RX band for handset operations. However, dueto the reported much more relaxed spec on TX noise in theRX band for WiMAX-like broadband wireless applications,we believe that the adoption of linear-assisted switch-modeenvelope amplifiers in the polar TX SoC implementation shouldbe justified and can be potentially integrated with commercialchipsets for WiMAX-like modulations.

V. DESIGN OF MONOLITHIC SATURATED PA FOR ET-BASED

POLAR TRANSMITTERS

A traditional class AB PA can offer good PAE at the peakRF output power, and with careful design this high PAE can beachieved over a wideband. However, with high PAR RF inputsignals, most of the time the PA output is well below its peak


Fig. 16. (a) Measured efficiency versus supply voltage � of the split-bandenvelope amplifier with EDGE envelope signal (inductor � 56 �H); and(b) WiBro envelope signal (inductor � 15 �H). The measurement was done atseveral different load resistances as well.

output power and therefore mostly operating in the low PAE re-gions. For high PAR input signals, the PA also has to operate ina “back-off” mode to maintain good linearity, therefore furtherdegrading its efficiency. The average PAE for class AB PA withhigh PAR signals, therefore, is quite poor and this is a major bar-rier for battery size reduction. It is possible, however, to achievea significant efficiency improvement using monolithic switch-mode PA with envelope tracking [16], [17]. For base station ap-plications, this ET technique has proven to offer considerablereduction in heat dissipation for 3G, WiMAX and DVB trans-mitters [28], [39], [40]. Specifically, in ET-based large-signalpolar transmitter, the voltage supplied to the final stage of theRF PA is changed dynamically, synchronized with the RF signalpassing through the device to ensure that the PA remains satu-rated and efficient.

As an example, it is well-known that the EDGE waveformoccupies 200 kHz TX channels (within 880–910 MHz) with amoderate PAR of 3.3 dB and a PMR of 17 dB but it has strin-gent TX spectral mask specs as GSM at 54 dBc (400 kHz)and 60 dBc (600 kHz), and a worst case rms EVM of 9%

Fig. 17. Schematic of a single-stage monolithic class E SiGe PA at 900 MHz.

[27]. Typically, the EDGE linearity requirement is achieved byusing traditional current-mode PA typologies (class AB) and op-erating the amplifier several dB “back-off” from its point,which inevitably degrades its PAE significantly. Switch-modePAs (i.e., class D/E/S) can provide the highest possible PAE byoperating the devices as switches to minimize the overlappingof current and voltage waveforms. A class E PA is easier forintegration compared with a class F PA, and it is arguably themost efficient class of PA; if the optimal switching conditionsthat minimize power dissipation in the device are achieved, itsPAE can be theoretically 100% [41]–[43]. In reality, non-ideal-ities such as finite switching speed, switch resistive loss, pas-sive component loss, device breakdown and voltage rail limita-tions, etc. have kept the PAE of the best Si-based class E PAsbelow 70% at RF frequencies of 2 GHz and above [44]–[51].These measured low PAE values strongly suggest that it is ex-tremely challenging, if not impossible, to meet all of the optimalclass E switching conditions to achieve ideal I-V waveformsfor Si-based monolithic PAs at the high-GHz range. The broad-band nature of low-Q on-chip lumped components and theirlow self-resonant frequency values definitely limit their use inharmonic controlled applications above a few gigahertz. There-fore, a fully integrated Si-based high-efficient class E PA wouldlikely be in a “quasi-class E” or “near-class E” mode in actualoperations. For example, Negra and Bächtold have recently re-ported a lumped-element load-coupling circuit design methodfor class E approximation to provide improved second harmonictermination and simultaneous fundamental load transformationon Si [52]. They used the IBM 6HP technology for the mono-lithic SiGe PA design and achieved an impressive PAE 51%at 5 GHz for the sub-optimal class E SiGe PA. We have usedsomewhat similar design methodology and achieve 66% PAEat 900 MHz and 62% at 2.4 GHz for our monolithic SiGe PAs[17], [53]–[56].

A simplified schematic for our monolithic one-stage sub-op-timal class E PA for 900 MHz operation is shown in Fig. 17.We purposely left the RF choke (RFC) inductors off-chip be-cause of the available low Q and large size on-chip inductorsat 900 MHz. SPICE simulations using IBM SiGe7HP designkit with Cadence Spectre RF were performed on high-break-down SiGe HBTs for monolithic quasi-class E PA design to ob-tain best PAE targeted for 900 and 2300/2400 MHz wirelessapplications [53]–[55]. This technology offers typical high-


Fig. 18. Die pictures of monolithic class E SiGe PAs designs for (a) 900 MHz(1.1 � 1.7 mm with pads), and (b) 2.3 GHz (1.3 � 1.5 mm with pads).

breakdown devices with 25/57 GHz,4.2 V, 12.5 V. Our fabricated SiGe PA dies werebonded onto PC boards by bondwires for testing, as shown inFig. 18.

Fig. 19 shows the measured and simulated single-tone testdata on our 900 MHz single-stage class E SiGe PA at differentbase bias voltages and supply voltage . The simulatedoutput power is within 1 dB of the measured results, and theyboth increase with higher base bias voltage. The measuredPAE values are quite high at 10 dBm where the PA issaturated (i.e., 61%–64% at 3.6 V), while no externalI/O matching was used and the only off-chip inductor is thechoke inductor. Fig. 19(b) shows that the PAE of our 900 MHzSiGe PA continues to increase with higher supply voltage ,where a peak PAE of 63% is achieved at 3–3.5 V.Fig. 20 also presents the data of PAE and versus ofour 900 MHz class E SiGe PA.

Fig. 21 plots the measured single-tone test data on our2.3 GHz single-stage class E SiGe PA at different base bias

and supply voltages . Similar to our 900 MHz PAdesign, no external I/O matching was used. The best PAE inmeasurement is achieved at of 0.68–0.72 V [Fig. 21(a)],while the measured PAE values are high at 2.3 GHz [i.e.,63% at 2.5 V, Fig. 21(b)]. The lower supply voltageprovides higher PAE in the case of our 2.3 GHz SiGe PA, whileits PAE improves with higher input power at 0.65 V(Fig. 22).

Fig. 19. Single-tone measurement of 900 MHz SiGe class E PA: (a) �versus � (� � 10 dBm, � � 3.6 V); and (b) � and PAE versus� (� � 10.9 dBm, � � 0.65 V). All matching on-chip except for thechoke inductor placed off-chip.

Fig. 20. Single-tone measurement of PAE versus � for our 900 MHz mono-lithic SiGe class E PA �� 0.65 V�. All matching on-chip except for thechoke inductor placed off-chip.

VI. MEASURED AND SIMULATED SYSTEM RESULTS FOR

ET-BASED POLAR TRANSMITTERS

Switch-mode PAs are highly efficient but intrinsically non-linear so they need to be somehow linearized for non-constant


Fig. 21. Single-tone measurement of 2.3 GHz SiGe class E PA: (a) PAE versus� (� � 5 dBm) and (b)� , Gain and PAE versus� (� � 0.65 V,� � 7 dBm).

Fig. 22. Single-tone measurement data of our 2.3 GHz class E SiGe PA�� 0.65 V�.

envelope modulated systems. Classical ET techniques havebeen applied mainly to current-mode PAs [13]–[15], howeverrecent work indicates that ET can be extended to class E PAsand provide better linearization than EER without even the needfor pre-distortion for low PAR signals such as EDGE [16], [17],[54]. We will, therefore, present in this section the measured

Fig. 23. (a) Simulated AM-AM and (b) measured AM-AM performance of alow-band EDGE modulated ET-based polar TX system with our switch-modeSiGe PA �� 0.6 V� � � 12 dBm� � � 19.9 dBm� � �2.25 V�. No PA predistortion algorithms applied.

versus simulation results from a complete ET-based polar TXsystem using our SiGe class E PA described in Section V,and the split-band envelope amplifier detailed in Section IV.The measurements are carried out as shown in Fig. 7 with thePA output downconverted for AM-AM, AM-PM and EVManalysis.

As we have described earlier, both the static and group de-lays mismatches will produce system distortions for an ET/EERpolar TX, often caused by the finite bandwidth of the envelopeamplifier circuitry. Since a saturated nonlinear class E PA isused in our system to recombine both phase and amplitude sig-nals, this PA will introduce major nonlinearities in the systemthat can be described in terms of the AM-AM and AM-PMdistortions [21], [39]. In our experimental setup, we adoptedan open-loop polar TX architecture where the output of thePA is downconverted and sampled for analysis only [39], [54].AM-AM distortion is measured by the presence of unwantedamplitude signals at the output due to variation of the input am-plitude to the system. In our case, we measure the input ampli-tude to the PA and compare that with its output amplitude (after


Fig. 24. (a) Simulated AM-AM and (b) measured AM-AM performance ofa low-band EDGE modulated EER-based polar TX system with our switch-mode SiGe PA �� 0.6 V� � � 12 dBm� � � 19.7 dBm� � �2.25 V�. No PA predistortion algorithms applied.

downconversion to baseband). AM-PM conversion of an ampli-fier is the measurement of unwanted phase changes at the outputcaused by the variation of input amplitudes to the system. In ourcase, we assess the AM-PM conversion by measuring the ampli-tude input to the PA and compare it with the output phase signalof the PA. The measured and simulated AM-AM and AM-PMcharacterizations of both the ET and EER polar TX systems arepresented in Figs. 23–26 for direct comparisons.

Fig. 23 shows the AM-AM performance for our modulatedET-base polar TX system. Both simulated and measured datapresents great fidelity across the entire dynamic range, even forvery low input amplitude levels. This data indicates that our non-linear class E PA used in an ET-based TX should be able toachieve the necessary amplitude fidelity for EDGE modulatedsystems. Furthermore, our SiGe PA is also used for an EER-based TX polar system with low-band EDGE modulation for aclear comparison versus the ET architecture. Fig. 24(a) and (b)show the simulated and measured AM-AM performance forour EER system, respectively. For an EDGE-modulated EER

Fig. 25. (a) Simulated and (b) measured AM-PM performance of theET-based polar TX system with our switch-mode SiGe PA (� � 0.6 V,� � 12 dBm, � � 2.25 V; � � 19.9 dBm, low-band EDGEmodulation). No PA predistortion algorithms applied.

system, a slight increase of nonlinear behavior at high ampli-tudes is observed (for normalized input of 0.6 to 1). Further-more, a more pronounced distortion is clearly present at smallinput amplitudes, especially when it is close to zero. This unde-sired behavior can be seen in the simulation results and then cor-roborated in the measured data. This is an unacceptable leakagebehavior not present in the ET system, probably caused by theRF signal feedthrough from the input as a feed-forward cur-rent [27]. Further pre-distortion would be needed to alleviatethis unwanted AM-AM distortion for EER. Note in our EERsystem using the switch-mode PA, this AM-AM behavior shownin Fig. 24 is the best one obtained as we swept the biasfor optimal PA linearity. Comparing Fig. 23 with Fig. 24, theET-based system presents a clear immunity in term of AM-AMdistortion, thus making it a better choice for implementing polarTX SoC using our switch-mode PAs.

In addition to AM-AM measurement and simulations,phase behavior is of great interest when dealing with EDGEmodulated signals. Fig. 25 presents the AM-PM character-istics of an EDGE-modulated ET-based polar TX system.Fig. 25(a) and (b) show the simulated and measured cases,respectively, exhibiting good agreement between the two.Please note that initial phase values at zero input amplitude


Fig. 26. (a) Simulated and (b) measured AM-PM performance of the EER-based polar TX system with our switch-mode SiGe PA (� � 0.6 V, � �

12 dBm, � � 2.25 V; � � 19.7 dBm, low-band EDGE modulation).No PA predistortion algorithms applied.

are arbitrary. Also, the cluster-like outputs in Fig. 25(b) arenot replicated on the simulated results in Fig. 25(a) since thesimulated values are done by means of a fixed set of SPICEmodels (we were not using the Monte Carlos method), whilethe measurement is done by means of a set of statisticallyaveraged measurements. The relative phase settlement of ETsystem is within 5 at 0.3 (normalized input amplitude), butsimulated results [Fig. 25(a)] exhibit a slightly faster phasedistortion settling than the measurement data [Fig. 25(b)].Likewise, an EER-based polar TX system was also tested underlow-band EDGE modulation for AM-PM characterization. Thesimulated and measured results of the EER system are shownin Fig. 26(a) and (b), respectively. A phase settling of 5 canbe seen between the normalized input amplitudes of 0.3 and 0.4for both the simulated and the measured case. Therefore, theAM-PM conversion of the EER system is comparable to that ofthe ET system. In short, aside from the similar behavior foundbetween the two systems, the significantly better performancein AM-AM behaviors highly suggest the ET architecture as thesystem of choice for polar TX SoC design.

To see how this ET-architecture can provide higher PAE ata given output power, Fig. 27 shows the single-tone simulationof our switch-mode SiGe PA at different collector voltage for

Fig. 27. Simulation data of the PAE of our switch-mode SiGe PA at 900 MHzobtained by sweeping input power at different collector voltages of the PA (rangeof � � 0.7 to 3.9 V; single-tone input, � � 0.65 V).

the 900 MHz case. Fig. 27 indicates that optimal system PAEcan be selected at a required by varying the PA collectorvoltage , where the simulated PAE of the PA is mostlyabove 50%. The measurement results of versus PAE at dif-ferent collector voltage are not available, however the measuredPAE at 3.3 V is 5%–10% higher than the simulated dataaccording to our previous publication [16], [17]. Additionally,these PAE versus curves can be tabulated to be potentiallyused in real time for an ET-based polar system to reach max-imum system PAE [15], [57]. An ET-based large-signal polartransmitter with a saturated PA can operate with good PAE atall envelope signal levels, therefore significantly improved theaverage efficiency for high PAR wideband envelope signals.

This ET-based polar TX system with the split-band enve-lope amplifier and switch-mode SiGe PA has exhibited goodAM-AM and AM-PM behaviors, and is therefore expected toprovide low values of EVM and spectral re-growth at the TXsystem output. This is indeed the case, as we will show the mea-sured system testing results next. The measurement data of thecomplete low-band EDGE ET-based polar TX system are pre-sented in Figs. 28–31, where our split-band envelope amplifierand switch-mode class E PA are used. The supply voltages of theop-amp and the switcher are 3.8 V and 3.4 V, respectively. Thisis because if supply voltages of the op-amp and the switcherare both 3.4 V, we will see a degradation of the system linearityand EVM as certain clipping at the op-amp output is observedin both simulation and measurement. Fig. 28(a) shows the TXsystem output power versus the base bias voltage of our classE SiGe PA, where and system PAE reaches 20.5 dBm and41%, respectively. Fig. 28(b) plots the versus curvefor the polar TX system. All of the data shown in Fig. 28 passedboth the EDGE low-band TX mask and EVM specs.

In our system measurement, higher envelope amplifier effi-ciency can be achieved by slightly overdriving the envelope am-plifier, resulting some clipping on the output voltage [16]. How-ever, excessive clipping will degrade EVM and fail the EDGETX spectral mask requirements. Fig. 29 shows the measuredoutput power, PAE and EVM of the ET-based polar TX system.As can be seen from Fig. 29, when the rms PA collector voltage


Fig. 28. System measurement data of the entire low-band EDGE ET-basedpolar TX: (a) � versus the bias � of the switch-mode SiGe PA (� �10 dBm); (b) � versus � (PA � � 0.6 V). For the measurement setupof the split-band envelope amplifier, the supply voltage of the op-amp is 3.8 V,while that of the PMOS switcher is 3.4 V. All of the data shown passed boththe EDGE low-band TX mask and EVM specs. No PA predistortion algorithmsapplied.

Fig. 29. System measurement data of the entire low-band EDGE ET-basedpolar TX (� � 0.6 V, � of op-amp � 3.8 V; � of PMOS switcher �3.4 V, � � 12 dBm). No PA predistortion algorithms applied.

(i.e., output of the envelope amplifier) is at 2.6 V, the ET systemhas a high EVM of 8% that barely passed the ETSI spec; how-ever, the system noticeably failed to meet the EDGE TX trans-mission mask (not shown).

Fig. 30 plots the simulated and measured spectrums of ourEDGE ET-based polar TX system. One can see the ADS sim-ulation of the entire polar TX system matches quite well with

Fig. 30. (a) Measured and (b) simulated output spectrum for our ET-basedentire polar TX system versus EDGE mask; � � 12 dBm, � � 0.6 V,PA collector voltage �rms� � 2.47 V. Overall measured system efficiency �44.4%. No PA predistortion algorithms applied.

Fig. 31. Measured large-signal polar TX system output spectrum before andafter ET/EER linearization versus the EDGE TX mask using our split-band en-velope amplifier and class E monolithic SiGe PA �� 12 dBm� � �3 V� � � 0.6 V�. No PA predistortion algorithms applied.

the measurement data, which demonstrates the usefulness andthe powerful predictive capability of our system design method-ology that includes the co-design and optimization of all RF,analog and digital circuits used in the polar TX system. Fig. 31shows the experimental results of our highly efficient large-signal polar transmitter using both ET and EER architectureswith the low-band EDGE modulated input signals. One can seethat ET easily outperforms EER in terms of TX output distor-tion. Table I summarizes the performance of the ET and EERPA systems for comparison. On can clearly see that ET outper-forms EER for implementing polar TX systems, with superior


TABLE ICOMPARISON OF ET VERSUS EER SYSTEM USING OUR SIGE CLASS E PA WITH

LOW-BAND EDGE MODULATED SIGNALS �� V� � � �� dBm

Note: The required EVM is 9% by EDGE specs.Overall System PAE� (RF modulated output power- RF input power)/overallDC power.Overall System CE � RF modulated output power/overall DC power.No PA predistortion algorithms were applied in ET or EER.

Fig. 32. Simulated ET-based TX polar system output spectrums using our split-band envelope amplifier and switch-mode class E SiGe PA versus the 802.16eTX mask (� � 18.5 dBm, PA � � 3.6 V, � � 0.73 V). The supplyvoltage of the split-band envelope amplifier is 5.5 V. No PA predistortion algo-rithms applied.

EVM performance, improved PAE, and significantly lower dis-tortion in the TX output spectrum (note no predistortion is ap-plied in the systems). We also compared the distortion caused bytiming misalignment between the amplitude and RF paths, andinvestigated experimentally their dependence on the PA biasingpoint. Note that according to our bias-dependent Cann’s modeldiscussed in Section III, the best timing alignment for ET is pre-dicted to be at a base bias of 0.55–0.6 V for the SiGe PA,which is again consistent with our measurement findings.

In addition to investigating and proving the usability of ourdesign under an EDGE ET-based polar TX system, we expectthat our class E SiGe PA can perform well in OFDM mod-ulation systems with high PAR for WiMAX/WiBro applica-tions. Fig. 32 shows the simulated system output spectrum ofour ET-based polar TX system for WiBro application. Whendealing with broadband signals such as WiMAX/WiBro, theirinherent wideband envelope signals will require the envelopeamplifier to have a much wider bandwidth to meet the linearityrequirement. Such high requirement makes these systems moresusceptible not only to static but also group delays. To fur-ther explore these effects when dealing with broadband wirelesssystems, we made use of either an ideal op-amp or a realisticsplit-band envelope amplifier to illuminate/isolate the distortion

caused by the switching modulator. As seen from Fig. 32, whenno additional delay is applied in the RF path, the ET TX outputspectrum using our linear-assisted switch-mode envelope am-plifier shows 3–8 dB higher spectral distortion than the caseusing an ideal op-amp, resulting in the output spectrum slightlyfailing the stringent 802.16e mask (say, at above 2.365 GHz).This failure is mainly due to the group delay of the realistic enve-lope amplifier caused by high bandwidth requirement, togetherwith the envelope output clipping and possible switching noise.To compensate the group delay caused by the envelope ampli-fier, we inserted a 3 ns static delay line in the RF path [as shownin Fig. 7(b)], which reduced the spectral growth by 2–3 dB andthe TX output then successfully passed the 802.16e TX mask(see Fig. 32). Since we have demonstrated excellent agreementbetween measured versus simulated results for our ET-basedpolar TX system using EDGE signals, we expect somewhat sim-ilar agreement between the measured versus simulated data willbe seen as well for WiBro input signals. However, PA predistor-tion and PAR de-cresting algorithms may be necessary in theWiBro polar TX system to further reduce the distortions andthe high supply voltage of the envelope amplifier (now at 5.5 V)to improve the overall TX system efficiency and its usefulnessfor mobile wireless applications. Test bench characterization iscurrently in development of WiBro polar TX system to assesthe agreement between the simulation data presented here tothe upcoming bench results. The robustness of the switch-modePA (i.e., mismatched under several VSWR conditions) needsto be carefully examined in measurement as well. The esti-mated system PAE of the WiBro/WiMAX ET-based polar TXsystem using our class E PA and split-band envelope amplifiercan reach 30%, showing a great promise for future highly ef-ficient broadband wireless TX SoC design.

VII. CONCLUSION

We discussed the design issues of highly efficient and mono-lithic wideband RF polar transmitters, especially the ones thatuse the envelope-tracking (ET) technique. Some state-of-the-artpolar transmitters are reviewed, and the SoC design considera-tions especially for highly efficient envelope amplifiers and PAsfor ET-based RF large-signal polar transmitters are also dis-cussed. Very good agreement has been found between the sim-ulation and measurement results on our entire polar TX systemfor low-band EDGE input, including AM-AM, AM-PM, TXoutput spectrum, etc. This powerful TX system simulation plat-form enables the co-design of the RF, analog, and digital cir-cuits to achieve the optimal system performance, applicable forbroadband wireless applications such as WiBro/WiMAX ap-plications as well. We believe, therefore, that ET-based large-signal polar transmitter architecture is very attractive for re-alizing highly efficient wideband monolithic transmitters formobile wireless communications.

ACKNOWLEDGMENT

The authors are indebted to Mr. D. Kimball, Prof. L. E.Larson, and Prof. P. Asbeck (all from UCSD) for their guid-ance on the ET system measurement. The authors also deeplyappreciate the help from Mr. D. Meng (at TTU), and thankIBM for IC fabrication.


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Jerry Lopez (S’09) received the B.S. and M.E.degrees at the University of California at San Diego(UCSD) in 2001 and 2005, respectively. He iscurrently working towards the Ph.D. degree in theDepartment of Electrical and Computer Engineering,Texas Tech University, Lubbock, TX.

Previously, he worked for Northrop GrummanCorp. in research and development involving thedesign of highly efficient power amplifiers and polarmodulators including custom RFICs design. Hehas also worked at the US Navy where he was a

member of the RFIC design group. He has been the recipient of the NorthropGrumman Fellowship and NSF Educational Fellowship. His research interestsinclude the design of analog/RF integrated circuits such as highly integratedhighly efficient CMOS PA/transmitter and circuits and systems that are highlynonlinear.

Yan Li (S’09) received the B.S.E.E. degree fromSouthwest Jiaotong University, China, in 2007. Heis currently working towards the Ph.D. degree in theDepartment of Electrical and Computer Engineering,Texas Tech University, Lubbock, TX. He has beenthe recipient of the AT&T Chancellor’s EndowedFellowship from Texas Tech University.

His current research is on the design of highly effi-cient and linear wideband polar transmitters with in-tegrated power amplifiers for WiMAX-like wirelessapplications.

Jeremy D. Popp received the B.S.E.E. degree fromPortland State University and the M.Eng. degree fromthe University of California at San Diego.

He is currently the Mixed Signal ASIC De-sign Leader at the Boeing Company’s Solid StateElectronics Development group and manages thedevelopment of high-speed PLLs and SerDes prod-ucts for space-based applications. Previously, hewas a Senior Member of Technical Staff and LeadCircuit Designer at Orora Design Technologies,where he led Orora’s PLL IP development in DSM

CMOS. He also worked as a program technical leader for the US Navy wherehe successfully lead several high profile defense electronic system design andadvanced technology programs. He has twelve technical publications, holdstwo patents, and has received several awards for his exceptional contributions.

Donald Y. C. Lie (S’86–M’87–SM’00) receivedthe M.S. and Ph.D. degrees in electrical engineering(minor in applied physics) from the CaliforniaInstitute of Technology, Pasadena, CA, in 1990 and1995, respectively.

He has held technical and managerial positions atcompanies such as Rockwell International, Silicon-Wave/RFMD, IBM, Microtune Inc., SYS Technolo-gies, and Dynamic Research Corporation (DRC). Heis currently the Keh-Shew Lu Regents Chair Asso-ciate Professor (tenured) in the Department of Elec-

trical and Computer Engineering, Texas Tech University, Lubbock, TX. He isalso an Adjunct Associate Professor, Department of Surgery, Texas Tech Uni-versity Health Sciences Center (TTUHSC). He is instrumental in bringing inmultimillion dollars research funding and has also designed real-world com-mercial communication products sold internationally. He has been a VisitingLecturer to the ECE Department, University of California, San Diego (UCSD)since 2002 where he taught upper-division and graduate-level classes and af-filiated with UCSD’s Center of Wireless Communications and co-supervisedPh.D. students. He has authored and co-authored over 75 peer-reviewed tech-nical papers and book chapters and holds several US patents.

Dr. Lie has been serving on the Executive and RF Design committee ofthe IEEE Bipolar/BICMOS Circuits and Technology Meeting (BCTM), theTechnical Program co-chair for IEEE SiRF, and also serving on various IEEEVLSI-DAT, SoCC and DCAS committees. He has received numerous awardsfrom DRC, IBM, Rockwell, and has given many invited talks and shortcourses at IEEE conferences and workshops. Dr. Lie and his students havewon several Best Graduate Student Paper Awards in international conferencesand also various prestigious scholarships. He was a Rotary InternationalScholar 1989–1990, and awarded with internships at Motorola Inc. sponsoredby SRC (Semiconductor Research Corporation) at 1993–1994, and also atthe NASA Jet Propulsion Laboratory (JPL), 1992–1993. Dr. Lie has servedas the Area Editor-in-Chief for the International Journal on Wireless andOptical Communications and also as reviewer for various IEEE journals. Heis working on a book on SiGe BiCMOS RFIC design with the CambridgeUniversity Press and articles on the history of modern science versus ChristianFaith and Chinese Culture. His professional research interests are low-powerRF/analog integrated circuits and system-on-a-chip (SoC) design and test, andinterdisciplinary research on medical electronics, biosensors and biosignalprocessing.

Chia-Chang Chuang received the B.S. degree inelectrical engineering from the National YunlinUniversity of Science and Technology in 2002. He iscurrently working towards the M.S.E.E. degree in theDepartment of Electrical and Computer Engineeringat Texas Tech University.

From 2005 to 2007, he worked as a Power De-sign Engineer with Universal Microelectronics Ltd.,Taichung, Taiwan, where he designed high-frequencyswitching power supplies. He holds one U.S. patenton a solar energy pulse charge device.


Kevin Chen received the M.S. and Ph.D. degrees inelectrical and electronic engineering from Univer-sity of Bristol, UK, in 2006. His doctoral researchconcerned with the high-efficiency power amplifierdesign for wideband applications.

In June 2006, he joined the Industrial Tech-nology Research Institute (ITRI), a top researchcenter in Taiwan, where he was involved in thedigital RF receiver design. He is currently leadinga research team towards the development ofhigh-efficiency and high-linearity polar transmit-

ters for WiMAX application.

Stanley Wu was born in Hsinchu, Taiwan, on Jan-uary 8, 1969. He received the M.S. degree in elec-trical engineering from the National Taiwan Univer-sity (NTU) in 1996.

Since 2001 he has been a design engineer withthe SoC Technology Center (STC)/Industrial Tech-nology Research Institute (ITRI), involved in RF andanalog integrated circuit design.

Tzu-Yin Yang was born in Taichung, Taiwan, in1973. He received the B.S. degree in electronicsengineering from National Chiao-Tung University,Taiwan, in 1995, and the M.S. degree in electricalengineering from National Taiwan University,Taipei, in 1997.

After graduation, he worked for the ElectronicResearch and Service Organization (ERSO), Indus-trial Technology Research Institute (ITRI), Hsinchu,Taiwan, on RF integrated circuit design. Since 2001,he has been with SoC Technology Center (STC),

ITRI, Taiwan. His research interests are integrated circuits and systems forwireless communications.

Gin-Kou Ma (S’85–M’90) received the Ph.D. de-gree in electrical engineering from the University ofFlorida, Gainesville, in 1989.

He was with The Athena Group Inc. USA, during1985–1989, where he was responsible for theMONARCH-DSP CAD tool project. From 1990to 1998, he was working as an R&D Manager ofHigh-speed Broad-band Information Networks:Communication Technologies and Services for theComputer Communication Research Laboratories(CCL) of Industrial Technology Research Institute

(ITRI), Taiwan, R.O.C. During 1998 to 2000, he was the R&D Manager ofAdvanced Microelectronics System Technologies for Electronics Research andService Organization (ERSO) of ITRI. During 2000 to 2006, he was the SeniorLeader Researcher of Advanced RFIC and AMS Circuit and DSP Designs,and worked on CMOS SoCs including low-power TDS-CDMA, WCDMA,WLAN, Bluetooth RF transceiver, DVB-T/H tuner, 3.1–10.6 GHz UWB RFIC,various low-power high-bandwidth high-resolution ADC, reconfigurable radioprocessor, 10 Gbps optical transceiver, low-power high-performance PAC-DSPfor multimedia CODEC, advanced hearing aid system, as well as others. Heis currently the Deputy General Director of SoC Technology Center (STC),ITRI, and is responsible for broadband wireless (WiMAX/4G) communicationsystem and chip designs and portable better-life CPE projects.

Dr. Ma received the 1997 National Outstanding Information Engineer Award,MOEA, Taiwan, the 2002 Outstanding Engineer Award, IEC, Taiwan, for hisachievements and contributions to multimedia, broadband and wireless com-munications, and the 2003 Outstanding MOEA Project and 2006 Best MOEAProject Awards. He has published over 90 conference and journal papers andhas twelve US patents. He is interested in research in broadband wireless com-munication and digital signal processing technologies. He was the secretary ofIEEE Taipei Section from 2005 through 2006.

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