Offline LED applications · 2019-08-02 · 1 Off-line LED applications From PFC basics to constant...
Transcript of Offline LED applications · 2019-08-02 · 1 Off-line LED applications From PFC basics to constant...
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Off-line LED applications
From PFC basics to constant current LED drive
CompelFest 2013
EU Design Services
Roberto Scibilia
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AGENDA
Review of Power Factor
EU Specs and Energy Star®
Limits for lighting equipments: EN61000-3-2 Class C
Selecting the right topology:
Buck in average current mode
Buck + Charge pump Boost
Buck + Voltage Feed Forward
Continuous, Discontinuous and Transition Mode
Boost
Flyback
Closing the loop on the output current
DC current sensing
Transformer on secondary side
Peak current stabilizing
Low and Mid-Power LED driver Portfolio
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Review of Power Factor
Power Factor is the Ratio of Real Power (Watts) to Apparent
Power (RMS Volt-Ampere product)
Power Factor has two components –
Displacement Factor (DispF)
Distortion Factor (DF)
Power Factor PF is the product of DF and DispF
)(Power Apparent
Power Real
VA
(W)PF
2
1
1
1
THDI
IDF
rms
cosDispF
cos1
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THDPF
Vin
Iin
Current Lags Voltage by
(Displacement Factor)
Vin
Iin
Current has high harmonic
content (THD)
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Review of Power Factor
Reactive currents, either capacitive or inductive, including reactive harmonics result in
circulating currents and associated I2R losses in the power transmission system but do not
develop power in the load.
Loads presenting the AC line with high power factor minimize unnecessary power losses
in the transmission system.
Loads presenting the AC line with low current THD minimize losses and interference with
adjacent loads.
Ideal resistive loads have a power factor of 1.0 and generate no harmonics (THD = 0).
Legacy incandescent lamps are nearly ideal resistive loads. (temperature coefficient of
filament causes some distortion,.
AC
Line Voltage
Line Current
Volt/
Amp/
AC
Line Voltage
Line Current
Volt/
Amp/
PF = 0.90
THD = 43.5%
PF = 1.0
THD = 0
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E U Specs and Energy Star ®
Classes
EN-61000-3-2
sets harmonic content for any power supply sold in EU
4 classes
D ~ personal computers and TVs
C ~ lighting equipment
B ~ portable tools
A ~ everything else
Energy Star®
Power supplies with greater than or equal to 100-W input power
must have a true power factor of 0.9 or greater at 100% of rated
load when tested at 115 V, 60 Hz
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EN61000-3-2 and Energy Star ®
Limits
Power Factor not important!
Only harmonic currents are limited, up to 39th harmonic, nominal
230 VAC
Limits depend on Class: Absolute limit in amps (Class A, B)
Percentage of fundamental (Class C lighting)
Amps/watt up to absolute max limit (Class D 75 W ≤ Pin ≤ 600 W)
Energy Star®
Power factor drives the limit here
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Lighting limits: EN61000-3-2 Class C
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Lighting limits: EN61000-3-2 Class C (P<25W)
For class C equipment with an input power smaller or
equal than 25W either:
1. The limits of table 3 (column two) apply
2. Or the third harmonic current shall not exceed 86% and
the fifth harmonic current shall not exceed 61% of the
fundamental current (for further details refer to the
standard).
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Fixing the maximum limits (P>25W)
1st, 3rd, 5th, 7th, 9th Harmonic (7th is 180 deg. shifted)
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Fixing the maximum limits
Approximation with a simple waveform:
A trapezoidal line
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Fixing the maximum limits
The trapezoidal line fulfill the class C limit since it’s
an approximation of the built waveform
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Is it enough also for Energy Star® ?
The THD of the waveform generated from the
EN61000-3-2 limits is ~ 11.5%
Consequently the PF, assuming there is no
displacement factor, is 94.7%
This value is well higher than the 0.9 dictated from
the Energy Star® regulations
BUT: how do we get this waveform?
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Selecting the right Topology: Buck
An average current mode Buck converter might do the job, but
the conduction angle is not enough to fulfill Class C
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Selecting the right Topology: Buck
Peak current mode, sensing on Mosfet’s current
Constant output current: you get the “smiling” input current
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Selecting the right Topology: Buck
Constant output current and voltage = Constant Power
The “smiling current” has even worse PF (~0.55) and bad THD Real Measurement
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Solution with Feed Forward injection
Absorbed current almost sinusoidal (PF>0.85)
Low cost, no feedback loop
Output current slightly dependant on input AC voltage (±20% on Vin translates into ± 10% on Iout)
This dependence can be reduced by injecting a DC bias ∩ Vin
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CCM, TM, DCM…what’s the difference?
IPEAK
IAVERAGE
(C) DRM
(b) DCM
IPEAK
IAVERAGE
IAVERAGE
(a) CCM
Some very important
differences:
Ripple Current
Drives filtering requirements, ac
losses in magnetics
Peak current
Drives semiconductor stress,
losses, peak flux in magnetics
Frequency
CRM is variable frequency
Helps on EMI, but enters quickly
the 150KHz lower limit
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CCM Loss Analysis
In CCM diode experiences large reverse recovery
current
PFC boost is the worst case for diode
MOSFET losses are increased
Ripple currents are small
low ac losses
low peak current stress
RMS currents are lower
conduction losses are lower
t (a) CCM
t (b) CRM
Diode Current
Switch Current
Diode Current Switch Current
I L_ pk_ccm
IL_valley_ccm
C
V OUT L
Q1
I L I D
I Q
I L_ pk_crm
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TM and DCM Loss Analysis
Peak current in TM is 2 x CCM (even higher in DCM)
High ac losses
High peak current stress on Mosfet and Diode
TM and DCM benefit: the diode stars conducting
always when the energy in the inductor is zero: no
reverse recovery issue
Reduced risk of MOSFETs failure due to shoot through by
reverse recovery of the Diode
RMS currents are higher than CCM
conduction losses are higher
Worst case in DCM
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TM Current Loop
TM current loop employs hysteretic type control
Lower boundary is zero
Upper boundary is set by multiplier
No “loop” to design
Simply chose sense resistor based on large signal considerations
Need to sense when zero current crossing occurs
Signal taken from already existing auxiliary winding
Small inductance of TM is traded-off for increased
filter size
Use low losses cores and Litz wire, when possible ($$$)
Variable frequency operation can help
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CCM: Closing the loop
The inner current loop corrects the Power Factor
The outer voltage loop regulates the output voltage
The input voltage feed-forward speeds-up the voltage regulation
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Input Average Current: why don’t we get PF=1?
The current loop tries to
stabilize a constant
output voltage (or
current)
Since we have 100Hz
ripple on output cap, the
error signal (Verr) will
have also that ripple
This error signal will add
distortion on the
absorbed current while
trying to regulate “inside”
it
The consequence is a
reduced Power Factor
and higher THD
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Boost Topology: TM mode
t
(a) CCM
t
(b) CRM
Diode Current
Switch Current
Diode Current
Switch Current
I L_ pk_ccm
IL_valley_ccm
C
V OUT
L
Q1
I L I D
I Q
I L_ pk_crm
Ripple current in CCM is
“small”
Peak current in CRM is 2x
CCM
If TON is constant, the peak
input current is proportional to
the sinusoidal input voltage
The cycle-average input
current is half the triangular
switching waveform area
resulting in a sinusoidal input
current
L
Vin*|sin(ωt)| ILCRM_PK
*TON
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Flyback Topology: TM mode
If TON is constant, the peak input current is proportional to the sinusoidal input voltage
The cycle-average input current is the average of the switch current, which is NOT sinusoidal because TOFF depends on ILCRM_PK
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Duty Cycle and Frequency vs. Time
The output power is then calculated by integration of the transferred energy cycle by cycle
POUT0
x1
2LP IPKprim x( )
2 FS x( )
d
0 0.314 0.628 0.942 1.257 1.571 1.885 2.199 2.513 2.827 3.14250
90
130
170
210
250
0.2
0.36
0.52
0.68
0.84
1
212.869
78.646
F.S x( )
1000
1
0.369
Duty_Cycle x( )
0 x
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Input Average Current: why don’t we get PF=1?
The average input current is NOT a half of the peak inductor current but it is averaged with duty-cycle
Iinavg x( )1
2IPKprimx( )
TON x( )
TON x( ) TOFF x( )
0.435
0
Iinavgvminx( )
0 x
IinavgvminIinavgvmin
0 0.5241.0471.5712.0942.6183.1420
0.083
0.167
0.25
0.333
0.417
0.5
0.435
0
I.inavg x( )
0 x
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Typical application LED Driving
The loop is closed to the output current, so VSENSE pin needs only a bias voltage
Primary aux. winding used for zero current switching
Secondary aux. winding for biasing and sensing the reflected input voltage (Triac dimming)
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UCC28810 Transition Mode Controller
Suitable for Boost, Flyback, Sepic, Buck, as PFC controller and constant output current generator as Buck controller.
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Closing the loop on the output current:
Current sense resistor + op-amp + TL431
R5 senses the output current
U1-B is a differential amplifier
U1-A has a TL431 (pin 3) + op-
amp inside
U3 transfers the error signal to
the EAOUT pin of UCC28810
D9, Q3 provide the Bias
voltage for U1: note the dots
on transformer; the forward
voltage is rectified, which is
independent of the output
voltage (variable from 40V to
120V)
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Closing the loop on the output current:
Current sense transformer + TL431
C6 + Llk of T1
define the
resonant
frequency (LLC)
T3 senses the
output current
R11 contains the
information of
the output
current
U3 + U4 close the
loop and create
the error voltage,
which modulates
the switching
frequency
U2, D5 work as
OVP
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Closing the loop on the output current: Stabilizing
the peak current on the output inductor (TM)
R1 senses the switch current (same peak value of the inductor)
The value is compared inside UCC28811 with a fixed reference
An hysteretic mode allows the converter to work into Transition Mode (Bang-Bang modulation)
The average output current is half of the stabilized peak
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Closing the loop on the output current: Stabilizing
the peak current on the output inductor (CCM)
The peak current through R4 is compared to the internal maximum limit
CCM and duty-cycle > 50% need slope compensation (R5)
C4, D3, D4, supply the Bias
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Flyback Loop Compensation: Power Stage Gain
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Loop Compensation: Power Stage Gain
Calculate the power stage gain at 95Vdc input and full
load of a 80Vout@2A, Flyback converter
Transformer primary inductance Lp=70uH and turns
ratio 1:1
The input power is always:
If the efficiency is 89%, Pin = 180W
VINPK = 120.2V, the 95Vdc is reached at 0.91 radians
From the switching frequency graph we pick the
equivalent frequency at 0.91 rad.: ~ 60KHz
“Flying” back to the TL431, the total
gain is: Gps = 30.5dB
Pin1
2Lp Ipk
2 Fsw
VIN t( ) VINPKsin t( )VIN t( ) 95VDC
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Loop Compensation: Closing the loop
Choose a crossover frequency << 100/120Hz
Place the zero to gain enough phase margin
18Hz, 17.9dB
4.23Hz
18Hz, -17.9dB 160Hz
7.24Hz
-30
-25
-20
-15
-10
-5
0
5
10
15
20
25
30
35
1 10 100 1000 10000
Frequency (Hz)
Gain
(d
B)
Power Stage Compensation Total Open Loop
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Loop Compensation: Closing the loop
ΦEA = -180º TL431 inverting input (error
amplifier)
ΦINT = -90º due to the integration of the
same amplifier
θZ = Tang-1(Fc/Fz) Fc= crossover
frequency, Fz = zero freq.
θP = Tang-1(Fp/Fc) - 90º Fp = pole frequency
The phase margin without compensation is:
Mφ = 360+ ΦEA + ΦINT + θP = 13.2º …..not enough!
We start choosing a pole at 160Hz (we loose here 6.4º)
We need then a zero that has a phase lead, at least:
θZ = 75º - (13.2º - 6.4º) = 68.2º, where 75º is our ideal
phase margin
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Loop Compensation: Real Measurement
The real measurement has only a slightly higher
phase margin than calculated (85º instead of 75º)
The crossover frequency and the -1 slope match the
measurement
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Test results on a 42W isolated LED string driver
89.5
90.0
90.5
91.0
91.5
92.0
92.5
40.0 50.0 60.0 70.0 80.0 90.0 100.0 110.0 120.0
Output Voltage (V)
Eff
icie
nc
y (
%)
180Vdc 230Vdc 265Vdc
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Driven by a constant current power supply with
ballast resistors
Unbalanced channel current
Poor efficiency
Driven by a constant voltage power supply with
constant current linear LED drivers
Balanced channel current, but
Poor efficiency
TRADITIONAL LINEAR DRIVING
METHOD
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DRIVING LED BY A CONSTANT
CURRENT POWER SUPPLY
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VLED6
VLED5 VLED4
VLED3
VLED2
VLED1
Constant
Current
Power Supply
Largest current through the
lowest LED voltage channel
Lowest current through the
highest LED voltage channel
Unbalance current through each channel
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DRIVING LED BY LINEAR CONSTANT
CURRENT DRIVERS
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VREG6
VLED6
VLED5
VREG5
VLED4
VREG4
VLED3
VREG3
VLED2
VREG2
VLED1
VREG1
Fix Rail Voltage
Constant
Voltage
Power Supply
Linear
constant
current
driver
Significant
power
dissipation
across
linear
drivers
=>Poor
efficiency
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LM3466 is a linear LED driver which acts like an intelligent ballast
resistor. Each IC communicates with other IC’s to equalize the current in
each channel which derives from a constant current power supply, i.e.,
divides the current equally. Thus it is very easy to construct a high
power lighting fixture by combining an off-the-shelf constant current
power supply and LM3466’s.
DEVICE HIGHLIGHT
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Works with a constant current power supply
Equalizes the current of every active LED string automatically
Maintains constant output power if some strings open (inactive), and the
current of remaining active LED strings will be equalized automatically
No communication to/from the constant current power supply is required
Operating with minimum voltage overhead to maximize power efficiency
(up to 99%)
Wide operating voltage from 6-70V, drives up to 20 LEDs per string
Up to 1.5A driving current
Protections : Thermal shutdown, fault status output
Linear circuitry does not deteriorate EMI
Package: PSOP8
Target application : Street lamp, tunnel lamp, parking lot lamp,
panel lamp
LM3466 OVERVIEW
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LM3466 BUILDING BLOCK
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TYPICAL APPLICATION
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Lighting Power Products – Combined Portfolio
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AC-DC LED Driver Controller
< 50W PAR 38, Industrial
Fixtures, Area Lights
• Dimming, dissipative
• PF >80 %
• THD < 40%
• Eff > 65%
Perf
orm
an
ce
Output Power
< 25W
A19, PAR
20/38
< 10W
A19, PAR 20,
downlights
< 5W
GU10, E14
Candles
TPS92210
LM3445
LM3444
TPS92070 (Q3/11) Market Base
Requirements
LM3450
TPS92001
TPS92010
UCC28810
Integrated Dimmer Detection
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TPS92070 LED lighting Driver controller
System Block Diagram (Isolated driver)
Near Lossless Dimmer Triggering
TRIAC Dimmer detect
Dimmer Angle detect
Valley Fill
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LM3445
TRIAC Dimmable Offline LED Driver
Features Benefits TRIAC Dimming Decoder for LED Dimming Integrated TRIAC Detection Reduces
Component Count and Solution Size
Master/Slave Operation Single TRIAC Controls Multiple Strings with
Consistent Dimming Performance
Application Voltage Range (80-277Vac) Supports Residential and Commercial LED
Lighting Applications
Controls LED Currents of Greater than 1A
Adjustable Switching Frequency
Adaptive, Programmable Off-Time Control
Thermal Shutdown, UVLO, Current Limit
Applications Dimmable Residential LED Lighting Drivers:
A19 (E26/27, E14), PAR30/38, GU10
Lighting Applications: Light Bulb
Replacement, Wall Sconces, Wall Washers,
Architectural and Display Lighting,
Commercial Troffers and Downlights
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LM3444 AC/DC Offline LED Driver
Features Benefits Application Voltage Range (80-277Vac) Supports Residential and Commercial
LED Lighting Applications
Controls LED Currents of Greater than
1A
Supports a Wide Variety of LED
Configurations
Adjustable Switching Frequency
Adaptive, Programmable Off-Time
Control
Thermal Shutdown, UVLO, Current Limit
Applications Non-Dimming Residential LED Lighting
Drivers: A19 (E26/27, E14), PAR30/38,
GU10
Lighting Applications: Light Bulb
Replacement, Wall Sconces, Wall
Washers, Architectural and Display
Lighting, Commercial Troffers and
Downlights
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TPS92210
PFC Offline LED Lighting Driver Controller
Features Benefits Flexible Operating Modes: Peak Primary
Current, Constant On-Time, or both
Constant On-Time implements Single
Stage Power Factor Correction (PFC)
Cascoded MOSFET Configuration Fast start up; Line Surge Ruggedness
Better Than Internal HV FET
Works with TRIAC Dimmers Continuous Exponential Dimming
Transformer Zero Energy Detection High Efficiency, Low EMI
Discontinuous Conduction or Transition
Mode Operation
No Reverse Recovery Loss in Output
Rectifier
Advanced Over-Current Protection and
Integrated Over-voltage Protection
Protects Driver Against Fault Conditions
Applications Residential LED Lighting Drivers: A19
(E26/27, E14), PAR30/38, GU10
Lighting Applications: Light Bulb
Replacement, Sconces, Wall Washers,
Architectural and Display Lighting,
Commercial Troffers and Downlights TOOLS
•TPS92210EVM-647 (110V)
•TPS92210EVM-613 (230V)
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TPS92010 High Efficiency Offline LED Lighting
Driver Controller
Features Benefits High Efficiency LED Lighting Current
Quasi resonant and low power modes
87% Achievable Efficiency – Higher than
Standard Flyback Topologies
High Performance TRIAC dimming with
application circuit
Less than 400mA Standby Power Allows
Efficient Deep Dimming
Programmable Overvoltage Protection 20% More Efficient Dimming Compared with
Other Methods
Internal Over-temperature Protection Safely Shuts Down Driver if Open or Over
Temperature is Condition is Present
TrueDrive Gate Drive 1A sink, 0.75A Source Lower Switching Losses Reduces System
Cost
Current Limit Protection Cycle-by-cycle Power Limit
Primary Side Over-current Hiccup
Restart Mode
Protects Driver from Fault / Abnormal
Conditions
Applications Residential LED Lighting Drivers
Lighting Applications: Wall Sconces, Pathway,
Overhead Lighting, wall washing and display
lighting TOOLS
•TPS92210EVM-592 (110V)
•TPS92210EVM-631 (230V)
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TPS92001/2General Purpose LED Lighting Driver
Features Benefits Ideal for Single Stage LED Driver
Designs
Power Factor >0.7
Isolated and Non-Isolated Topologies Supports Wide Configuration of LED
Loads
TRIAC Dimmable Application Circuit
with Low External Component Count
Low Cost Deep Dimming Solution with
Small Form Factor
Convenient 5V Reference Power for MCU or Linear Circuits
Two Under-Voltage Lockout Options
(10V or 15V)
Protection from Abnormal Operating
Conditions
Integrated Gate Drive: 0.4A Source /
0.8A Sink
External Gate Drivers Not Required
Applications Residential LED Lighting Drivers: A19
(E26/27, E14), PAR30/38, GU10
Lighting Applications: Light Bulb
Replacement, Wall Sconces, Wall
Washers, Architectural and Display
Lighting, Commercial Troffers and
Downlights
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Single Stage Flyback AC-DC Controller:
Primary side flyback LED current regulation
Doesn’t require opto-coupler or secondary side circuitry
Adaptive ON-time control with inherent PFC
Critical-Conduction-Mode (CrM) with Zero-Current-Detect (ZCD) for valley
switching
Reduces EMI filter design complexity
LED current setting with external sense resistor
No loop compensation required
Gate driver with slew rate control
Eases EMI filter design
Output voltage protection (OVP) through ZCD
VCC Under Voltage Lock Out (UVLO)
Thermal Shutdown
SOIC-10 package
TPS92310 Features Description
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Typical schematic (COT)
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Typical schematic (PCM)
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What is difference between COT and
PCM operation.
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Constant ON time
>> High power factor
Peak Current Mode (Pull MODE1 pin to GND)
>>Low output ripple current
IAC
VAC
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Design consideration
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Current sense
Resistor
OVP
Resistor
Snubber
Fly wheel Shottky diode
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Design example and component
selection
1. Snubber circuits: use 600V ultra fast diode.
2. Flywheel diode : use the 100V…200V 2A schottky.
3. ZCD / OVP resistor : Suggest set the Normal voltage is ~3.5V . Current is 1mA.
4. Current sense: IOUT = N*IREF/ RCS
N is turn ratio of transformer (primary : aux ): (3.8:1)
REF = 0.14
0.35 mA = 3.8 x 0.14 /1.5Ω. (*PS: 0.14 is Internal reference.)
5. Power MOSFET :
600V or upper. 2A e.g. 3N60E
6. TDLY resistor selection: 1. First step : 1/4 (TDLY )= 1/(2 Pi x SQRT(Lm x C )) . Assume Cds= 200p, L = 5mH.
2. Calculation : TDLY = 2/(pi*SQRT(Lm x C) =636ns = 8K.
3. Based on the MOSFET drain to source waveforms. Fine tune RDLY.
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Design example and component
selection
7. Pull up resistor (R4):
The current through resistor R4 must less than 1mA. Otherwise OVP can’t
restart. (can not pull low by device itself)
8. Input capacitor (C2):
C2 can’t too small, We assume the VIN is constant within a switching
cycle. If value of C2 too small. The input current can’t estimate the output
current correctly.
For 220VAC the base is 0.15uF > C2> 0.1uF, The C2 too large will impact
the power factor, too small will impact the current regulation.
9. Output capacitor (C3):
C3 can smooth the IOUT current more smooth. Bigger C3 value can
reduce the output current, but will impact the start up time. 430uF or
330uF are typical.
10. Diode (D7,D6): Diode D7 can limit the ZCD pin > -0.3V, it can avoid a negative current
through the IC device. D6 is a zener diode for protection the VCC voltage.
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Design transformer consideration:
Transformer:- Use a better couple transformer.
Consider the transformer layout. e.g. use the interleaved
windings.
Interleaved winding has low eddy current loss.
Full cover the bobbin length.
Switch-node pin should be connect the transformer inside for reduce
the EMI.
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TPS92310 line regulation (ILED
=
350mA)
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0,250
0,275
0,300
0,325
0,350
0,375
0,400
0,425
0,450
90 95 100 105 110 115 120 125 130 135 140
Iou
t (A
)
AC Input (V)
Iout vs AC Input Voltage LIM_HI LIM_LO
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TPS92310 I_LED vs Temperature
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-3%
0%
3%
-30 -15 0 15 30 45 60 75 90
I_L
ED
(%
)
Temperature (°C)
I_LED vs Temperature
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Input voltage and current
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VAC
IAC
PF = 0.91, Efficiency = 83%
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Start up operation waveforms
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VLED
IOUT
VIN
VCC
Test condition: - Apply VDC = 0 to 250VDC, measure the output LED voltage and current.
CH1 :- Controller input Voltage VCC
CH2:- VLED voltage
CH3 :- VDC Input DC Supply Voltage,
CH4 :- ILED output current.
Test result: - Normal, In upper start up case, LED light up time is 0.6sec. In typical operation, (220VAC ) start up time within 1.2
sec by current silicon.
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Typical operation waveforms
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V_Lx
I_TX
Test condition :- Apply 250VDC to light bulb supply pin, measure from drain to source voltage and
Transformer primary side current.
CH1 : NC,
CH2 : NC,
CH3: Drain to source voltage at external main MOSFET,
CH4 : Main switch current,
Test result :- Normal. The convertor can achieve CRM switching and zero current switch. Thus, design
EMI and EMC filter are easy.