Novel RF MEMS Devices for W-Band Beam-Steering Front-Ends516789/FULLTEXT01.pdf · Novel RF MEMS...

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Novel RF MEMS Devices for W-Band Beam-Steering Front-Ends Nutapong Somjit MICROSYSTEM TECHNOLOGY LABORATORY SCHOOL OF ELECTRICAL ENGINEERING ROYAL INSTITUTE OF TECHNOLOGY ISBN 978-91-7501-296-4 ISSN 1653-5146 TRITA-EE 2012 : 011 Submitted to the School of Electrical Engineering KTH - Royal Institute of Technology, Stockholm, Sweden in partial fulfillment of the requirements for the degree of Doctor of Philosophy Stockholm 2012

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Novel RF MEMS Devices for W-Band Beam-Steering

Front-Ends

Nutapong Somjit

MICROSYSTEM TECHNOLOGY LABORATORY SCHOOL OF ELECTRICAL ENGINEERING

ROYAL INSTITUTE OF TECHNOLOGY

ISBN 978-91-7501-296-4 ISSN 1653-5146

TRITA-EE 2012 : 011

Submitted to the School of Electrical Engineering KTH - Royal Institute of Technology, Stockholm, Sweden

in partial fulfillment of the requirements for the degree of Doctor of Philosophy Stockholm 2012

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The front cover shows a SEM image of the first prototype of the fabricated three-stage dielectric-block phase shifter composed of 15°, 30°, and 45° stage (left). The different phase-shifts of the silicon block are achieved by tailor-making the block with different etch-hole sizes. The right picture is a SEM image of silicon grass (or black silicon) observed during the 890-µm deep-reactive ion-etch (DRIE) step of the though-silicon holes for the semiconductor-substrate-integrated helical antenna fabrication. The aspect ratio between the etch-depth and the opening area is 22.25. Copyright © 2012 by Nutapong Somjit All rights reserved for the summary part of this thesis, including all pictures and figures, unless otherwise indicated. No part of this publication may be reproduced or transmitted in any form or by any means, without prior permission in writing from the copyright holder. The copyrights for the appended journal papers belong to the publishing houses of the journals concerned. The copyrights for the appended manuscripts belong to their authors. Printed by Universitetsservice US-AB, Stockholm 2012. Thesis for the degree of Doctor of Philosophy at the Royal Institute of Technology, Stockholm, Sweden, 2012.

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Abstract iii

Novel RF MEMS Devices for W-Band Beam-Steering Front-Ends

Abstract This thesis presents novel millimeter-wave microelectromechanical-systems (MEMS) components for W-band reconfigurable beam-steering front-ends. The proposed MEMS components are novel monocrystalline-silicon dielectric-block phase shifters, and substrate-integrated three-dimensional (3D) micromachined helical antennas designed for the nominal frequency of 75 GHz. The novel monocrystalline-silicon dielectric-block phase shifters are comprised of multi-stages of a tailor-made monocrystalline-silicon block suspended on top of a 3D micromachined coplanar-waveguide transmission line. The relative phase-shift is obtained by vertically pulling the suspended monocrystalline-silicon block down with an electrostatic actuator, resulting in a phase difference between the up and downstate of the silicon block. The phase-shifter prototypes were successfully implemented on a high-resistivity silicon substrate using standard cleanroom fabrication processes. The RF and non-linearity measurements indicate that this novel phase-shifter design has an excellent figure of merit that offers the best RF performance reported to date in terms of loss/bit at the nominal frequency, and maximum return and insertion loss over the whole W-band, as compared to other state-of-the-art MEMS phase shifters. Moreover, this novel design offers high power handling capability and superior mechanical stability compared to the conventional MEMS phase-shifter designs, since no thin moving metallic membranes are employed in the MEMS structures. This feature allows MEMS phase-shifter technology to be utilized in high-power applications. Furthermore, the return loss of the dielectric-block phase shifter can be minimized by appropriately varying the individual distance between each phase-shifting stage. This thesis also investigates 3D micromachined substrate-integrated W-band helical antennas. In contrast to conventional on-chip antenna designs that only utilize the surface of the wafer, the novel helical radiator is fully embedded into the substrate, thereby utilizing the whole volume of the wafer and resulting in a compact high-gain antenna design. The performance of the antenna is substantially enhanced by properly etching the substrate, tailor making the antenna core, and by modifying size and geometry of the substrate-integrated ground plane. A linear line antenna array is composed of eight radiating elements and is demonstrated by simulations. Each antenna is connected to the input port through a multi-stage 3-dB power divider. The input and output of the single-stage 3-dB power divider is well matched to the 50-Ω impedance by four-section Chebyshev transformers. The simulation results indicate that the novel helical antenna arrays offer a narrow radiation beam with an excellent radiation gain that result in high-resolution scan angles on the azimuth plane. The proposed helical antenna structures can be fabricated by employing standard cleanroom micromachining processes. Nutapong Somjit, [email protected] Microsystem Technology Laboratory, School of Electrical Engineering KTH Royal Institute of Technology, SE-100 44 Stockholm, Sweden

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Abstract iv

Novel RF MEMS Devices for W-Band Beam-Steering Front-Ends

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Contents v

Novel RF MEMS Devices for W-Band Beam-Steering Front-Ends

Contents Abstract iii List of Publications vii 1. Introduction

1.1 General introduction 1 1.2 Structure of this thesis 2

2. RF MEMS beam-steering front-end 2.1 RF MEMS background 3 2.2 Electronic beam-steering front-end 3 2.3 RF MEMS components for beam-steering front-ends 4

2.3.1 Some selected MEMS Phase shifters 4 2.3.1.1 MEMS TTD phase shifters 4 2.3.1.2 DMTL phase shifters 6

2.3.2 Micromachined antennas 7

3. Novel W-band phase shifters 3.1 Novel phase-shifter concept and design 11 3.1.1 Single-stage phase shifters 11 3.1.2 Multi-stage phase shifters 14

3.1.2.1 Linear-coded phase shifters 14 3.1.2.2 Binary-coded phase shifters 14

3.1.3 MEMS actuator design 15 3.2 Prototype fabrication 17 3.3 RF measurements 18 3.3.1 Single-stage phase shifters 18 3.3.2 Linear-coded phase shifters 19 3.3.3 Binary-coded phase shifters 20 3.4 MEMS characterization and reliability 23 3.4.1 Actuator characteristics 23 3.4.2 Life-cycle and reliability test 24 3.5 Non-linearity analysis 25 3.5.1 Power modulation 25 3.5.2 Intermodulation products 26 3.6 Thermal property and power handling analysis 28 3.6.1 Design comparisons of conventional MEMS phase shifters 28 3.6.2 Measurement setup and results at 3 GHz 30 3.6.3 Extension of results to 75 GHz 33 3.6.4 Comparisons of power handling 36 3.7 Design optimization for minimizing signal reflection 37 3.7.1 Two-stage phase shifters 38 3.7.2 Multi-stage phase shifters 39

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3.7.3 Simulation results 41 3.7.4 Measurement results 43

4. Substrate-integrated helical antenna 4.1 Substrate-embedded square helical antennas 47 4.2 Influence of substrate 49 4.3 Influence of dielectric core 52 4.4 Ground plane variations 55 4.5 Antenna arrays 57 4.6 Feed network and matching 59 4.6.1 Four-stage Chebyshev matching transformer 59 4.6.2 3-dB power divider 60

5. Conclusions 63

References 65

Acknowledgement 79 Summary of the appended papers 81 Paper reprints 83

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List of Publications vii

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List of Publications The thesis is based on the following papers in reviewed international journals:

1. "Binary-Coded 4.25-Bit W-Band Monocrystalline-Silicon MEMS Multi-Stage Dielectric-Block Phase Shifters," N. Somjit, G. Stemme, J. Oberhammer. IEEE Transactions on Microwave Theory and Techniques, vol. 57, issue 11, pp. 2834-2840, Nov. 2009.

2. "Deep-Reactive Ion-Etched Wafer-Scale-Transferred All-Silicon Dielectric-Blocks Millimeter-Wave MEMS Phase Shifters," N. Somjit, G. Stemme, J. Oberhammer. IEEE/ASME Journal of Microelectromechanical Systems, vol. 19, issue 1, pp. 120-128, Feb. 2010.

3. "Power-Handling Analysis of High-Power W-band All-Silicon MEMS Phase Shifters," N. Somjit, G. Stemme, J. Oberhammer. IEEE Transactions on Electron Devices, vol.58, no.5, pp.1548-1555, May 2011.

4. "Microwave MEMS Devices Designed for Process Robustness and Operational Reliability," M. Sterner, N. Somjit, U. Shah, S. Dudorov, D. Chicherin, A. Räisänen, J. Oberhammer. EuMA International Journal of Microwave and Wireless Technologies, vol. 3, special issue 5 (RF MEMS), pp. 547-563, Oct. 2011.

5. "Design Optimization of Millimeter-Wave MEMS Dielectric-Block Phase Shifters," N. Somjit, J. Oberhammer. Submitted for journal publication.

6. "Three-Dimensional Micromachined Silicon-Substrate Integrated

Millimeter-Wave Helical Antennas," N. Somjit, J. Oberhammer. Submitted for journal publication.

7. "RF Characterization of Gold-Coated Nickel-Wire Through Silicon Vias Fabricated by Magnetic Assembly," N. Somjit, A. Fischer, S. Bleiker, U. Shah, G. Stemme, J. Oberhammer, F. Niklaus. Manuscript for journal publication.

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The contribution of Nutapong Somjit to the publications: 1. Major part of design, all fabrication, all experiments, and major part of

writing. 2. Major part of design, all fabrication, all experiments, and major part of

writing. 3. Major part of design, all fabrication, all experiments, and major part of

writing. 4. Part of design, part of fabrication, part of experiments, and part of writing. 5. Major part of design, all fabrication, all experiments, and major part of

writing. 6. Major part of design, and major part of writing. 7. Major part of RF design, all RF experiments, and major part of writing.

The work has been presented at these reviewed international conferences: 8. "Novel RF MEMS Mechanically Tunable Dielectric Phase Shifter,"

N. Somjit, G. Stemme, J. Oberhammer. Proceedings of IEEE International Conference on Infrared, Millimeter, and Terahertz Waves, Pasadena, CA, USA, Sept. 2008, oral presentation.

9. "Novel Concept of Microwave MEMS Reconfigurable 7×45º Multi-Stage Dielectric-Block Phase Shifters," N. Somjit, G. Stemme, J. Oberhammer. Proceedings of IEEE International Conference on Micro Electro Mechanical Systems, Sorrento, Italy, Jan. 2009, oral presentation.

10. (*) "Phase Error and Nonlinearity Investigation of Millimeter-Wave MEMS 7-Stage Dielectric-Block Phase Shifter," N. Somjit, G. Stemme, J. Oberhammer. Proceedings of IEEE European Microwave Conference/IEEE European Microwave Integrated Circuit Conference, Rome, Italy, Sept. 2009, oral presentation.

11. "Comparison of Thermal Characteristics of Novel RF MEMS Dielectric-

Block Phase Shifter to Conventional MEMS Phase-Shifter Concepts," N. Somjit, G. Stemme, J. Oberhammer. Proceedings of IEEE European Microwave Conference, Paris, France, Sept. 2010, oral presentation.

12. "High Aspect Ratio TSVs Fabricated by Magnetic Self-Assembly of Gold-Coated Nickel Wires," A. Fischer, S. Bleiker, N. Somjit, T. Haraldsson, N. Roxhed, G. Stemme, F. Niklaus. IEEE Electronic Components and Technology Conference, San Diego, CA, USA, May 2012, oral presentation by A. Fischer.

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List of Publications ix

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13. "Minimum-Return-Loss Optimization for Multi-Stage Millimeter-Wave MEMS Phase-Shifter Concepts," N. Somjit, G. Stemme, J. Oberhammer. IEEE European Microwave Conference, Amsterdam, The Netherlands, Oct. 2012 (Submitted)

14. (**) "Semiconductor-Substrate Integrated 3D-Micromachined W-band Helical Antennas," N. Somjit, J. Oberhammer. IEEE International Symposium on Antennas and Propagation & USNC/URSI National Radio Science Meeting, Chicago, IL, USA, July 2012, poster presentation.

The work has also been presented at the following international workshops: 15. "Microwave MEMS Activities at the Royal Institute of Technology,"

J. Oberhammer, N. Somjit, M. Sterner, F. Saharil, S. Braun, G. Stemme. GigaHertz Symposium, Göteborg, Sweden, Mar. 2008, oral presentation by J. Oberhammer.

16. "Reconfigurable Three-Dimensional Micromachined Dielectric-Loaded CPW Phase Shifter," N. Somjit, G. Stemme, J. Oberhammer. International Symposium on RF MEMS and RF Microsystems, Heraklion, Greece, Jun. 2008, oral presentation.

17. “Monocrystalline-Silicon Microwave MEMS Devices: Multi-stable Switches, W-band Phase Shifters, and MEMS Tuneable Frequency-selective Surfaces,” J. Oberhammer, M. Sterner, N. Somjit, G. Stemme. NATO-Advanced Research Workshop: Advanced Materials and Technologies for Micro/Nano Devices, Sensors and Actuators, St. Petersburg, Russia, Jun. 2009, oral presentation by J. Oberhammer.

18. "Ultra-Wideband Low-loss All-Silicon MEMS Phase Shifters for High-Performance Electronic Beam-Steering Applications," N. Somjit, G. Stemme, J. Oberhammer. GigaHertz Symposium, Lund, Sweden, Mar. 2010, oral presentation.

19. "Microwave MEMS Activities at KTH- Royal Institute of Technology" J. Oberhammer, N. Somjit, M. Sterner, Z. Baghchehsaraei, U. Shah, F. Saharil, S. Braun, G. Stemme. Micronano System Workshop, Stockholm, Sweden, May 2010, oral presentation by J. Oberhammer.

20. "Novel Low-loss 4.25-Bit All-Silicon Millimeter-Wave MEMS Phase Shifters for High-Performance W-Band Applications," N. Somjit, G. Stemme, J. Oberhammer. Micronano System Workshop, Stockholm, Sweden, May 2010, poster presentation.

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21. "Investigation of Power Handling Capability of Conventional and Novel All-Silicon RF MEMS Phase Shifters," N. Somjit, G. Stemme, J. Oberhammer. International Symposium on RF MEMS and RF Microsystems, Otranto, Italy, Jun. 2010, oral presentation.

22. "Thermal Characteristics and Power-Handling Capability of High-Power Millimeter-Wave MEMS Dielectric-Block Phase Shifters," N. Somjit, G. Stemme, J. Oberhammer. GigaHertz Symposium, Stockholm, Sweden, Mar. 2012, oral presentation.

The work has been published in book chapters:

23. "Reconfigurable Three-Dimensional Micromachined Dielectric-Loaded CPW Phase Shifter," N. Somjit, G. Stemme, J. Oberhammer. Series in Micro and Nanoengineering vol. 15: New Developments in Micro Electro Mechanical Systems for Radio Frequency and Millimeter Wave Applications, Publishing House of the Romanian Academy, pp. 163-167, 2008.

24. "Monocrystalline-Silicon Microwave MEMS Devices: Multi-stable

Switches, W-band Phase Shifters, and MEMS Tuneable Frequency-selective Surfaces," J. Oberhammer, M. Sterner, N. Somjit. Springer NATO series: Advanced Materials and Technologies for Micro/Nano Devices, Sensors and Actuators, Springer, 2010.

25. "RF MEMS for Automotive and Radar Applications," J. Oberhammer, N. Somjit, U. Shah, Z. Baghchehsaraei. MEMS for Automotive and Radar Applications, Woodhead Publishing Ltd., 2012.

Other international reviewed conference papers, by Nutapong Somjit, not included in the thesis:

26. "Numerical Optimization and Design of a 3-GHz Chopper/Prebuncher System for the S-DALINAC," N. Somjit, R. Eichhorn, H.-D. Gräf, C. Heßler, Y. Poltoratska, A. Richter, W.F.O. Müller, T. Weiland. European Particle Accelerator Conference, Edinburgh, UK, Jun. 2006, poster presentation.

27. "Status of Polarized Electron Gun at the S-DALINAC," G. Iancu, M. Brunken, J. Enders, H.-D. Gräf, C. Heßler, Y. Poltoratska, M. Roth, K. Aulenbacher, N. Somjit, W. Ackermann, W.F.O. Müller, B. Steiner, T. Weiland. European Particle Accelerator Conference, Edinburgh, UK, Jun. 2006.

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28. "Longitudinal Beam Dynamic Simulation of S-DALINAC Polarized Injector," B. Steiner, W.F.O. Müller, N. Somjit, T. Weiland, R. Eichhorn, J. Enders, C. Heßler, A. Richter, M. Roth. Linear Accelerator Conference, Tennessee, USA, Aug. 2006.

Part of this work has been awarded the following international prizes:

1. (*) Best Paper Award (European Microwave Integrated Circuit Prize).

European Microwave Week 2009, Rome, Italy.

2. (**) Honorable Mention in Student Paper Competition IEEE AP-S International Symposium on Antennas and Propagation 2012, Chicago, IL, USA.

3. IEEE Graduate Fellowship 2010. IEEE Microwave Theory and Techniques Society.

4. IEEE Graduate Fellowship 2011. IEEE Microwave Theory and Techniques Society.

5. IEEE Doctoral Research Award 2012. IEEE Antennas and Propagation Society.

The work is published in the Education Column of the IEEE Antennas and Propagation Magazine as the final report of the IEEE Antennas and Propagation Society Doctoral Research Award. "Novel Millimeter-Wave MEMS Components for High-Performance Reconfigurable Beam Steering Front-Ends," N. Somjit, J. Oberhammer. IEEE Antennas and Propagation Magazine (To appear).

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List of Publications xii

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1. Introduction 1

Novel RF MEMS Devices for W-Band Beam-Steering Front-Ends

1. Introduction 1.1 General introduction Microelectromechanical systems (MEMS) are micro-scale devices that integrate electrical and movable mechanical structures. MEMS devices employ principles of physics and chemistry such as force, movement, temperature, and electromagnetic reaction to sense any changes in the system. The electronic section processes the information from the sensing unit by converting the information into an electrical signal and responding to that change. These kinds of MEMS devices are categorized as "sensors". Additionally, MEMS devices can be employed as "actuators" by converting an electrical signal into force, movement, or heat. Examples of MEMS devices used as actuators include micromotors, microswitches, micropumps, and microvalves. The fabrication processes of MEMS devices are developed based on a well-established semiconductor-fabrication technology called microfabrication that is generally composed of four major techniques: bulk micromachining, surface micromachining, bonding, and high-aspect-ratio micromachining. Interest in MEMS technology has substantially increased due to increasing demand for miniaturized devices. Moreover, MEMS components can be fabricated in a batch as in semiconductor integrated-circuit production, which results in a low cost-per-device unit. Additionally, it is possible to integrate MEMS devices and semiconductor integrated circuits on a single chip, which results in high-performance miniaturized devices with low power consumption that can be efficiently integrated or embedded in a system. These features make MEMS technology very attractive for research and industry. Radio-Frequency Microelectromechanical Systems (RF MEMS) present a new kind of miniaturized RF device that involves mechanical, electrical, and RF-signal interactions. RF MEMS devices offer nearly ideal characteristics such as extremely low signal losses, ideally zero-power consumption, and high linearity, as compared to their semiconductor electronic counterparts. Moreover, RF MEMS devices are usually fabricated employing low-temperature micromachining processes, and, thus, are compatible with post-processing semiconductor circuit integration such as CMOS, GaAs, and SiGe. Therefore, RF MEMS components can substitute the conventional costly and large off-chip discrete elements by enabling integrated solutions that can be batch-fabricated. RF MEMS is rapidly growing due to its impressive figure of merit and a promising commercial potential. Today, research and development in RF MEMS are emphasizing towards system integration, reliability, and designing of devices with novel configuration to meet commercial requirements such as RF MEMS switches, tunable filters, low-loss phase shifter, and reconfigurable substrate-integrated antennas. This thesis focuses on designing, fabricating, and characterizing novel RF MEMS monocrystalline-silicon dielectric-block phase shifters, and investigating three-dimensional (3D) substrate-integrated micromachined helical antennas for W-band beam-steering front-ends. The thesis introduces a novel design strategy for avoiding the utilization of thin moving metallic membranes as MEMS structures in dielectric-block phase shifters. The novel design concept offers solutions for many critical issues related to the design of conventional MEMS

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1. Introduction 2

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phase shifters such as reliability, power-handling capability, and performance. Additionally, this thesis discusses an optimization strategy for minimizing the return loss of the dielectric-block phase shifters. The second part of this thesis investigates novel substrate-integrated helical antennas. In contrast to the conventional on-chip antennas that utilize only the surface of the substrate, this 3D micromachined helical radiator is fully embedded into the wafer, which results in compact high-gain antenna structures. This section also discusses the design aspects of substrate-integrated helical antennas in terms of enhancing their radiation performance e.g., maximum gain, front-to-back ratio (F/B), and half-power beamwidth (HPBW). This thesis also discusses the design constraints, performance, and feed networks of the antenna array before proposing its fabrication process. 1.2 Structure of this thesis This thesis is subdivided into five chapters. Chapter 1 presents an introduction to MEMS technology with a short description of RF MEMS while Chapter 2 provides more detail on RF MEMS technology as well as a short introduction to electronic beam-steering front-ends. Furthermore, some RF MEMS components that enable high-performance beam-steering front-ends, are shown. These interesting MEMS components include various concepts and designs of the state-of-the-art MEMS phase shifter, as well as some selected micromachined antenna structures. Chapter 3 presents novel W-band monocrystalline-silicon dielectric-block phase shifters. The design strategy, micro-fabrication of the phase-shifter prototype, and phase-shifter optimization are also discussed in this chapter. Moreover, an extensive characterization of the proposed devices, and measurement results are reported, including RF performance, MEMS and life-cycle evaluation, non-linearity, and power-handling capability. Chapter 4 investigates 3D micromachined substrate-integrated helical antennas. This chapter studies extensively the design considerations and the influence of the surrounding substrate, dielectric core, and ground plane, on antenna performance. Moreover, a linear line antenna array composed of the proposed helical antennas is discussed. Additionally, the feed-network design is presented based on a 3-dB power divider and a multi-section Chebyshev matching transformer. Chapter 6 summarizes all the work presented in this thesis. Summary of the appended papers gives a brief overview of the reviewed journal publications which are related to this thesis and are reprinted in Paper reprints section.

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2. RF MEMS Beam-Steering Front-Ends 3

Novel RF MEMS Devices for W-Band Beam-Steering Front-Ends

2. RF MEMS beam-steering front-ends 2.1 RF MEMS background Over the past several decades, there has been rapid growth in the number and functionality of components and devices in communications and wireless systems. This drastic advancement is an effect of the tremendous achievement of the semiconductor and electronics industries that enable large amounts of information and data to be processed in a quick and inexpensive way. The need for tinier and more versatile devices in, e.g., mobile phones, automotive radar transceivers, GPS receivers, and remote sensors, is enabling intensive integration and the development of novel technologies to achieve the technological requirements. RF MEMS technologies including tunable capacitors [1]-[10], micromachined inductors [11]-[19], resonators [20]-[30], switches [31]-[42], and tunable filters [43]-[54] are emerging as enabling technologies that can fulfill the requirements for the next generation of components and devices. Although the technology is still maturing, MEMS is attractive for RF and wireless systems. Miniaturization and high integration, nearly-zero power consumption, high linearity, and capability for batch fabrication are among many reasons why MEMS is an important technology for these systems. The required performance such as low loss and high quality factor, of discrete passive components has yet to be matched by integrated circuits. The high demand for these off-chip passive components is restricting the further miniaturization of some systems. RF MEMS are better suited for integration, and therefore, allow for significant size reduction and decreased power consumption in devices. The availability of components and devices with improved performance allows for higher flexibility in designing systems. Moreover, producing devices with batch fabrication drastically decreases the cost-per-unit when replacing these components. This is an important consideration for future devices and systems that require high-performance, but low-cost components. Due to their superior performance, RF MEMS components can be used in high-performance electronic beam-steering front-ends, and wireless communication systems. They can also replace many conventional passive components such as phase shifters [55]-[66], antennas [67]-[78], and power divider [79]. 2.2 Electronic beam-steering front-ends Electronic beam steering is a key issue in phased arrays and is widely used in military and commercial applications such as radar systems and wireless communications. Many studies have been carried out on electronic beam steering. The most popular method is to use phase shifters in combination with an antenna array [80]-[81], as shown in Figure 1.1. Phase shifters are critical components because they are the general devices used in phased arrays. This is especially true in transceiver modules because phase shifters can produce phase shifts between the elements of an antenna array, and can steer the beam to the desired direction. However, the cost of a phase shifter accounts for nearly half of the cost of an entire electronically scanned phased array. In addition, most of the popular

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2. RF MEMS Beam-Steering Front-Ends 4

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semiconductor-based phase shifters have high losses, especially at the millimeter-wave frequencies.

Figure 1.1 Drawing of the basic principle of a beam-steering front-end based on a phased-array antenna. 2.3 RF MEMS components for beam-steering front-ends RF MEMS devices and circuits have attracted interest in applications such as electronic beam-steering systems, particularly in the millimeter-wave frequency band, due to their near-ideal signal performance and compatibility with semiconductor fabrication technology. The following sections introduce the state-of-the-art device architectures and describe the most commonly engaged RF MEMS components and circuits such as phase shifters and on-chip antennas in electronic beam-steering front-ends. 2.3.1 Some selected RF MEMS phase shifter Phase shifters are widely employed in electronic beam-steering systems that are based on phased antenna arrays. MEMS technology offers much lower insertion loss, higher linearity over a large bandwidth, and lower power consumption compared to solid-state technology. Ferrite-based phase shifters have good performance, but cannot be integrated easily and are more expensive to fabricate than MEMS technology. The following section discusses two conventional types of widely-discussed millimeter-wave MEMS phase shifters: (a) MEMS-switched true-time delay-line (TTD) phase shifter networks [82]-[84]; and (b) distributed MEMS transmission-line (DMTL) phase shifters [84]-[85]. 2.3.1.1 MEMS-switched TTD phase shifters MEMS-switched TTD phase shifters consist of various phase-shifting sections in a cascade arrangement. MEMS switches are employed in different lines to switch the line length of the signal, which results in different phase shifts. Since RF MEMS switches are used to switch between different paths, this kind of phase shifter inherits all the advantages of RF MEMS switches, which results in excellent performance. However, they are not suitable for millimeter wave

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frequencies, including W-band, because the multiple switches and necessary lengths of the transmission line degrade performance. Figure 1.2 shows an image of a one-bit Ka-band MEMS-switched TTD phase shifters based on single-pole double-throw (SPDT) switches [82]. Stehle et al. [83] describe a phase shifter that consists of a 45° loaded-line phase shifter element, a 90° and a 180° switched-line phase shifter element that results in a 3-bit RF MEMS phase shifter. For the complete device, the return loss is better than -12 dB and the insertion loss is -5.7 dB at 76.5 GHz. The design equations for the MEMS TTD phase shifter are comprised of impedance-matched slow-wave unit cells with the optimization goal of maximizing the figure-of-merit. B. Lakshminarayanan el al. present Δφ/dB [84], and to verify the equations, 1-bit phase shifters are implemented by a cascading number of unit cells that corresponded to various maximum design frequencies. The phase shifter, with a maximum frequency of 110 GHz, showed an insertion loss of approximately -2.65 dB, measured Δφ/dB of 150°/dB, and a return loss below -19 dB.

(a)

(b)

Figure 1.2. Photomicrograph of (a) MEMS SPDT switch, and (b) a one-bit Ka-band MEMS TTD phase shifters [82].

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(a)

(b)

Figure 1.3. A low-loss distributed 2-bit W-band MEMS DMTL phase shifter: (a) single cell with its corresponding profile, (b) entire 2-bit DMTL phase shifter composed of 24 switches [86]. 2.3.1.2 DMTL phase shifters The DMRL phase shifter is based on periodically loading the transmission line with capacitive MEMS bridges to vary the line capacitance. Consequently, the propagation coefficient of the line and thus the signal phase between the input and the output of the phase shifter are altered. The transmission line impedance is changed by varying the capacitance of the line which results in changing the propagation coefficient. This effect imposes another limitation on the maximum usable capacitance ratio, in addition to the ones required by the switch design parameters, according to the acceptable mismatch. The capacitance ratio is limited to approximately 1.3-1.6 [87]. DMTL phase shifters generally have excellent performance in the millimeter-wave regime compared to the TTD phase shifters. However, they have the same disadvantages that must be taken into consideration. Capacitive MEMS bridges that are incorporated into DMTL phase shifters and MEMS switches that are utilized in MEMS TTD phase shifters are composed of thin metal bridges that

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cannot handle large induced current densities at high RF power because of limited heat conductivity to the substrate due to their suspension above the substrate. This results in reliability issues due to buckling (plastic deformation) or even melting of the thin metal layer. Additionally, thin gold bridges, as employed in both types of MEMS phase shifters, are subject to drastic loss of their elastic behavior even at slightly elevated temperatures (around 80°C), which results in decreased reliability [88]. It is also possible to load the transmission line digitally by employing fixed capacitances, which are switched to load the line. This alternative method provides the possibility of increasing the capacitance ratio, which is more suitable in the millimeter-wave regime [87]. Barker and Rebeiz [85] discussed the design and optimization of DMTL phase shifters for U-band and W-band with analog tuning capability of the bits on a quartz substrate. The W-band DMTL phase shifter consisted of 48-bridge DMTL elements with pull-down voltage of just over 26 V and a corresponding capacitance ratio of 1.15. The measured phase shift per decibel loss is 70°/dB from 75 to 110 GHz. The average measured insertion loss is -2.5 dB and the return loss is -11 dB at 94 GHz. Hung, Dussopt, and Rebeiz [86] demonstrated a low-loss distributed 2-bit W-band MEMS phase shifter on a glass substrate, as shown in Figure 1.3. Each unit cell of this phase shifter consists of a MEMS bridge and the sum of two MAM capacitors, which are fabricated using the cross-over between the MEMS bridge and the CPW ground plane. Since the simulated phase shift of the unit cell is about 1.2° at 80 GHz, a 90° section with eight switches and a 180° degree section with 16 switches were cascaded. Phase shifts of 0°, 89.3°, 180.1°, and 272° are measured at 81 GHz, which are close to the designed frequency and phase shifts. The return loss is better than -11dB, the average insertion loss is -2.2 dB, and the phase error is equal to ±2°. The worst insertion loss is measured at -2.9 dB at 94 GHz.

Figure 1.4. Micro-mechanically driven antenna platform: (left) schematic view of the proposed RF antenna; (right) design layout [89]. 2.3.2 Micromachined antennas Kim et al have demonstrated a single antenna platform steered by an external magnetic field where the monolithic microwave integrated circuit (MMIC) and

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capacitors are vertically integrated [89]. The electrical steering scheme using phase shifters is already mature and shows a fast scan speed. However, it requires a number of phase shifters and power amplifiers to obtain a large scan angle, which results in bigger and more expensive systems. A mechanical steering antenna can radiate a beam directly as it has a constant RF gain at any scan angle. The absence of multiple phase shifters and amplifiers even at large scan angles means the mechanical beam steering antenna is smaller and more efficient in RF capability. The required deflection of the antenna in [89] is over 1.3 mm. Figures 1.4 presents a schematic of the RF antenna and steering method. The antenna consists of two pairs of mechanical springs made of BCB with a thickness of 40 μm. Two metal–insulator–metal (MIM) capacitors are implemented by conventional micromachining and are vertically flip-chip bonded to the suspended antenna substrate. Currents with the same magnitude and the same direction then flow through a pair of adjacent coils under the movable antenna substrate and generate the driving magnetic force. When the current flows through the two pairs of coils, the magnetic field is exerted to the nickel beneath the silicon substrate, and the plate rotates. The average tilting angles are 5.4°, 8.2°, 13.4°, and 18.3°, respectively, when the applied currents are 200, 300, 400, and 500 mA in the H-plane. The average tilting angles are 4.7°, 6.8°, 12.1°, and 17.7° in the E-plane. Figure 1.5 depicts the measured radiation beam patterns and tilting angles. Figure 1.5 presents the shifts in the the beam radiation patterns originating from the rotation angles of −14 °, 0°, and +18° in the H-plane and the beam patterns at −18°, −12°, 0°, +12°, and +16° in the E-plane.

Figure 1.5. Radiation beam pattern of the mechanically tilted RF antenna actuated by magnetic field: (left) H-plane; (right) E-plane [89]. In [90], a planar antenna structure is dynamically reconfigured to steer the radiation beam or change the shape of the beam using electrically controlled microactuators. Figure 1.6 shows the concept and the cross section of the reconfigurable Vee-antenna, respectively. Microactuators move the antenna arms of the Vee-antenna by pulling or pushing them. One end of the antenna arm is held by a fixed rotation hinge that is locked on the substrate, which allows the arm to rotate with the hinge as the center of circle. Both antenna arms were rotated by 30° and 45° in the same direction, while the Vee-angle was kept at 75°. Figure 1.7 shows that the main beams shifted

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by 30° and 48°. It also shows that the first null shifted from 35° to 15° for the 30°-steering and from 35° to 0° for the 45°-steering.

Figure 1.6. Concept of micro-mechanically reconfigurable Vee-antenna [90].

Figure 1.7. E-plane beam-steering patterns for a 17.5 GHz MEMS reconfigurable Vee-antenna [90].

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3. Novel W-band phase shifters This chapter introduces the novel monocrystalline-silicon dielectric-block MEMS phase shifters. First, the concept and design consideration of single-stage phase shifters are discussed extensively. This phase-shifter concept is based on multiple steps of deep-reactive ion-etching (DRIE) of monocrystalline-silicon blocks that are transfer-bonded on a high-resistivity silicon (HRS) substrate containing a 3D micromachined coplanar wave-guide (CPW). This chapter also describes two types of multi-stage dielectric-block phase shifter with different phase-shift configurations. In addition, the prototype fabrication of the dielectric-block phase shifters is presented. The prototypes are evaluated extensively for their RF figure-of-merit and MEMS characteristics. Additionally, the various types of device characterization such as reliability testing, non-linearity measurement, thermal properties, and power-handling evaluation, are conducted. Finally, the chapter discusses the different optimization strategies for significantly minimizing the return loss of the multi-stage phase shifter. 3.1 Novel phase-shifter concept and design 3.1.1 Single-stage phase shifter Figure 3.1. Schematically functional drawing of a single-stage dielectric-block phase shifter with actuation contact points for electrostatic actuation and its cross-sections, explaining the operation of the phase shifter. The phase-shift values are given for the length of the single-stage block only. Figure 3.1 presents the operational concept of a single-stage dielectric-block MEMS phase shifter consisting of a high-resistivity monocrystalline-silicon block anchored above a 3D micromachined CPW transmission line. The relative phase shift (∆φ) is achieved by vertically pulling the monocrystalline-silicon block down above the CPW by electrostatic actuation. This results in a different propagation speeds of the electromagnetic signal. The relative phase shift of the proposed devices depends on the vertical displacement of the monocrystalline-silicon block that is suspended above the CPW transmission line. In addition, the 50-µm deep-

actuation voltage

monocrytalline-silicon block

Au CPW 60/20 µm signal line/gap with 50-µm deep slots

100-nm thick Si3N4 distance keepers

serpentine flexures

anchor HRSS

up state

down state

380 µm 5 µm

50 µm

35 µm

φ = -135°

φ = -180°

∆φ = -180°

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etched slots along the gaps of the CPW in the HRSS are introduced to decrease substrate loss and increase the sensitivity of the propagation constant to displacement changes of the dielectric block. The length of the dielectric block is half the wavelength at the nominal frequency of 75 GHz to minimize the electromagnetic signal reflection from both edges of the dielectric block, which cancels out for an electrical length of the block of 180°. For a digital-type operation (upstate or pulled-down), an optimum operation point of high phase-shift sensitivity with an appropriate displacements that is realizable with MEMS electrostatic actuators is 5 µm for the displacement between the up and downstate. The relative phase shift (∆ φ) in radians per meter of the digital-type operation of the single-stage phase-shifter design is calculated to be:

∆𝜑 = 2𝜋𝑓𝑍0𝜀𝑟,𝑒𝑓𝑓

𝑐 1𝑍𝑢− 1

𝑍𝑑, (3.1)

where f is the design frequency (Hz), Z0 is the unloaded characteristic impedance of the high-impedance 3D CPW (Ω), εr,eff is the effective dielectric constant of the unloaded transmission line, c is the speed of light in free space (m/s), and Zu and Zd are the loaded impedances in the up and in the down state (Ω), respectively. Table 3.1. Simulated and calculated circuit parameters of the single-stage 45º dielectric-block phase shifter. design freq.

(GHz) Z0

ε (Ω)*; r,eff

λ;

g/2

Z

(µm)

uε (Ω); r,eff

λ;

u/2

Z

(µm)

dε (Ω); r,eff

λ;

d/2

Δφ

(µm) HFSS

(º)

Δφ Eq.(3.1)

(º)

75 54.3; 3.4; 1080

47.6; 4.5; 940

37.3; 7.2; 743

-44.2 -39.8

Note: * without loading the CPW transmission line with the monocrystalline-silicon block. From Ansys HFSS, the simulated characteristic impedance, Z0, of the unloaded line is 54.38 Ω, resulting in a loaded transmission line with the dielectric block in the upstate that is close to 50 Ω. Although the unloaded transmission-line impedance cannot be matched for the up and down state, the impedance mismatch is very low (see Table 3.1). This is proven by the measurement results presented in the following sections. Since the guided half wavelength in the upstate (940 µm) is different from the downstate (743 µm), the nominal length of the dielectric block is chosen as a compromise (760 µm) between these two values. The nominal length is closer to the downstate half-wavelength because the return loss in the downstate is larger than in the upstate; therefore, it is more important to minimize the return loss in the downstate. Table 3.1 presents the simulated and calculated circuit parameters of the single-stage dielectric-block phase shifter. Figure 3.2 presents the simulation results of the relationship between the relative phase shift and the displacement of the dielectric block in analog tuning mode. For the digital-type operation, an optimum operation point for high phase-shift sensitivity and decent deflection is 5 µm which is suitable for displacements that are typically achieved by MEMS electrostatic actuators. The relative phase shift

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of a 760-µm long block, at a relative displacement of 5 µm above the CPW, results in a relative phase shift of 45°. Figure 3.2. Simulation results of a phase shift corresponding to the displacement of the dielectric block. The optimum operation point for high phase-shift sensitivity and decent deflection is 5 µm. Square patterns are etched periodically into the thick monocrystalline-silicon block, and the ratio of the etched volume to the unetched volume allows for artificial tuning of the macroscopically effective dielectric constant of each individual block. Figure 3.3 shows the effective dielectric constant and the characteristic impedance of the transmission line loaded by the dielectric block as a function of the etch-hole size. Three different embodiments with a relative phase shift of 45°, 30°, and 15° for the nominal frequency of 75 GHz are used as building blocks for the multi-stage phase shifters. The chosen lengths of the single-stage 30° and 15° phase-shifters are 800 µm and 850 µm, respectively. Figure 3.3. Effective dielectric constant and the characteristic impedance of the transmission line loaded by the dielectric block as a function of the etch-hole size (left). The SEM image shows different phase-shifter designs with different etch-hole sizes. The etching factors are 3.4% for the 45°, 20% for the 30°, and 50% for the 15° phase-shifter stage.

downstate (0.1 µm)

upstate (5 µm)

operational

displacement

etched 3.4% (45°)

etched 20% (30°)

etched 50% (15°)

15° 30° 45°

760 µm 800 µm 850 µm

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All MEMS and mechanically moving structures of the dielectric-block phase-shifter concept are fabricated out of a single monocrystalline-silicon block. This method offers the best available mechanical characteristics in MEMS due to its excellent strength and highly elastic properties, which ensure creep-free actuation. This presents an inherent reliability advantage, when compared to conventional MEMS phase shifter concepts that utilize metallic bridges, either for MEMS switches in TTD designs or for MEMS switched capacitors in DMTL designs. These designs are subject to stress-related warping and plastic deformation which limits their life spans, especially at slightly elevated operation temperatures. 3.1.2 Multi-stage phase shifters Multi-stage phase shifters, consisting of a series of single-stage phase shifters, are required to achieve higher phase shifts. This section discuss two examples of a multi-stage phase shifter: linear-coded devices and binary-coded phase-shift devices. These phase-shifter designs offer a range of phase shifts, and phase-shift resolutions. 3.1.2.1 Linear-coded phase shifters A multi-stage linear-coded phase shifter with 45° phase-shift steps can be achieved by placing seven single 45° stages in series. Figure 3.4 shows a drawing of a seven-stage 45° phase shifter with full 360° phase-shift capability with a 45°-step phase resolution. The distance between each stage of the multi-stage linear-coded phase shifter is 10 µm, which minimizes the total length of the phase shifter, and ultimately, minimizes the insertion loss. The total length of the multi-stage linear-coded 7×45° phase shifters is 5.4 mm. Figure 3.4. Schematic drawing of a multi-stage linear-coded phase shifter with total phase-shift of 360° (left), and its phase resolution of 45° (right). 3.1.2.2 Binary-coded phase shifters The multi-stage binary-coded phase-shifter designs are composed of two phase-shifter stages of 15° and 30°, and five stages of 45° phase shifters, with a total phase-shift resolution of 270° in 15° steps, i.e. 19 different phase-shift states. Thus, this phase-shifter configuration has a phase-shift resolution of more than

distance between 2 stages: 10 µm

stage 1

stage 7

phase resolution

totel length 5.4 mm

90° 45°

135°

180°

215° 270°

315°

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four bits, which are compared to linear-coded 7×45° phase shifters with a total phase-shift of 360° in 45° steps, i.e., eight different phase states (3 bits). However, the total length of the multi-stage binary-coded 15°+30°+5×45° phase shifters is 5.51 mm, which is slightly longer than the multi-stage 7×45° linear-coded phase shifters. 3.1.3 MEMS actuator design The monocrystalline-silicon block is suspended above the 3D micromachined CPW by four silicon serpentine flexures that provide excellent mechanical properties in terms of strength and elasticity for high reliability. The DC actuation voltage of the dielectric-block phase shifter is applied between the RF ground of the CPW transmission line and the silicon surface at the anchor (see Figure 3.1). The mechanical spring structures [91]-[93] were designed by a finite-element-method (FEM) based simulation known as COMSOL Multiphysics. Figure 3.5. Electrostatic model with COMSOL Multiphysics of the actuation principle in the upstate position when HR monocrystalline silicon and the bonding polymer are considered perfect dielectric materials, even for DC. In the high-frequency regime, the monocrystalline HR silicon behaves as an excellent dielectric material with relatively low signal losses. However, if the monocrystalline silicon were considered a perfect dielectric material for the DC actuation, the actuation of the dielectric-block phase shifter would not work properly. Figure 3.5 explains this actuation characteristic by showing the 2D simulation results of the electrostatic actuation, which represents the potential distribution of the dielectric-block phase shifter over the cross-section, while applying a DC voltage (Vac). In the figure, all dielectric materials are considered perfect dielectric materials, i.e., without taking the leak current that flows through

Vac

Vac

anchor dielectric block shared grounds

between DC and RF

50-µm deep etched slots

shared ground between DC and RF

anchor bonding polymer dielectric

block

effective actuation area

serpentine flexure

30 V

20

10

0

5

15

25

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the bonding polymer into account. The potential drastically drops inside the silicon block, and creates only a very small potential difference between the electrodes in the actuation area (approximately 0.07 V), which is not sufficient to pull the monocrystalline-silicon block down. However, the silicon block has a finite resistivity of 4 kΩ∙cm, which - in connection with the leak current flowing through the bonding polymer underneath the anchor with much higher resistivity as compared to the silicon layer - is largely responsible for the dropping actuation voltage over the bonding polymer. As a result, the block has nearly the same potential everywhere and creates a potential difference between the effective electrodes, i.e., the bottom part of the silicon block and the ground of the CPW. Figure 3.6 shows the simulation results of the stationary flow-field of the potential distribution over the cross section of the dielectric-block phase shifter. By applying an actuation voltage of 30 V at the contact area between the anchor and the ground, the potential difference between the effective electrodes in this basic model is 27.46 V. Figure 3.6. Electrostatic actuation principle simulated with a stationary flow-field model, i.e., taking the leak current that flows through the bonding polymer between the anchor and the ground of the CPW transmission line into account. Moreover, a 3D electromechanical model in COMSOL Multiphysics simulated the pull-in characteristics. For 5-µm displacement, the simulation results indicate that pull-in occurs at 27.5 V and 6 V for phase shifters with three and five-meander serpentine flexure, respectively. The calculated effective spring constants are 36.6 N/m for the three-stage serpentine flexure and 3.5 N/m for the five-stage flexure.

Vac

Vac

anchor dielectric

block shared grounds between DC and RF

50-µm deep etched slots

anchor bonding polymer

serpentine flexure

dielectric block

effective actuation area

shared ground between DC and RF

30 V

20

10

0

25

15

5

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3.2 Prototype fabrication A high-resistivity (> 4 kΩ∙cm) silicon wafer with a loss tangent of 0.0006 (±50% error) measured at 75 GHz, and a measured dielectric constant of 11.9 is used to fabricate the dielectric-block phase shifters. Figure 3.7 schematically depicts the process flow. The 1-µm-thick sputtered-gold CPW is patterned by wet etching on a 500-nm SiO2 layer with a 100-nm thick adhesive Ti layer on top of the high resistivity silicon substrate. To prevent a DC short circuit between the CPW and the dielectric block during pull-in, an array of 100-nm thick Si3N4 distance keepers with an area of 5×5-µm2 are fabricated on top of the transmission line (see Figure 3.1). The fill-factor of the Si3N4 distance keeper array to as compared to a complete film is 1.59%, which is intended to reduce charging problems in the Si3N4 layer. The slots of the CPW are further etched by DRIE into the silicon substrate to create 50-µm deep trenches. Figure 3.7. Cross-sectional fabrication process flow after the major process steps.

(a) depositing gold CPW (20/60/20 µm) on a 525-µm thick HR silicon substrate covered with 500-nm SiO2.

(b) DRIE process and depositing of a Si3N4 distance keeper array.

(c) adhesive polymer bonding with an SOI wafer. The polymer is 5 µm thick

(d) removing SOI handle wafer and patterning of the movable block in ICP machine with DRIE process.

(e) patterning serpentine flexures with another DRIE step

(f) release movable structures with O2-plasma.

1-µm thick Au CPW

thermal SiO2: 500 nm

HR Si substrate

100-nm thickSi3N4 distant keepers

50-µm DRIEed slot

HR Si substrate

HRSS SOI wafer

35-µm thick device layer bonding polymer: 5µm

DRIE etch-holes

monocrystalline-silion block

serpentine flexures remaining bonding polymer

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A high-resistivity silicon-on-insulator (SOI) wafer is bonded face-to-face to the previous processed wafer by adhesive polymer bonding with mr-I 9250xp thermal nanoimprint lithography photoresist [95], spun on the device layer of the SOI wafer to provide a flat surface. The 5-µm-thick photoresist sacrificial bonding layer is obtained by spinning the photoresist twice with a 2.5-µm standard thickness. Hard baking of the first photoresist layer at 120°C for one minute between the spinning steps is required to create a stable photoresist layer for the next spinning step. The SOI handle wafer is then removed by plasma etching in an inductively coupled-plasma (ICP) machine. The buried-oxide layer of the SOI wafer is used as an etch-stop layer and subsequently as oxide masks for further processing steps. The periodical release-etch holes in the monocrystalline-silicon block are patterned by DRI-etching the 35-µm device layer of the SOI wafer. The second DRIE step is performed to shape the 4-µm-thick serpentine flexures and to define the outlines of the silicon dielectric block. Finally, the patterned structure is released by etching the photoresist sacrificial bonding layer in the O2 plasma through the etch-holes in the device layer. Since the sacrificial bonding layer is only partly underetched during the 15-minute O2-plasma etch process with 1000-W RF power, the polymer layer below the anchors mostly remains. Therefore, the block is suspended above the transmission line at a distance that corresponds to the thickness of the polymer layer. Figure 3.8 shows a picture of a prototype device that consists of a fabricated 30° and 15° phase shifter compared to a 45° stage, and a three-meander flexure. Figure 3.9 presents microscope pictures of a full 7×45º linear-coded phase shifter, and a 15°+30°+5×45° binary-coded phase shifter. Figure 3.8. SEM pictures of fabricated 30º and 15º phase shifter stages with different etch-hole size patterns, compared to a 45º stage (left); and a three-stage serpentine flexure (right). 3.3 RF measurements 3.3.1 Single-stage phase shifters In contrast to conventional MEMS phase shifter designs where the actuation voltage must be applied to the signal line and the CPW ground behaves as a common DC and RF ground, the DC actuation voltage for the dielectric-block phase shifter is applied to the floating block via the anchor. Therefore, no DC signal on the signal line, thus eliminating the necessity of DC and RF decoupling circuits. Moreover, no DC charging/loading effects of the transmission line that

45° stage 5×5 µm2

30° stage 17×17 µm2

15° stage 38×38 µm2

three-stage serpentine flexure

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might degrade the RF performance are observed. Furthermore, seven separate DC probes are employed in order for the measurement to actuate each phase-shift stage independently.

(a) linear-coded 7×45° phase shifters

(b) binary-coded 15°+30°+5×45° phase shifters Figure 3.9. Optical microscope pictures of the fabricated linear-coded 7×45° dielectric-block phase shifter (top), binary-coded 15°+30°+5×45° phase shifter (bottom), and their phase resolutions. Fig. 3.10(a) plots the RF measurement results for the single-stage phase shifters for the 45°, 30°, and 15° phase-shifts. At the nominal frequency of 75 GHz, a single-stage 45° performs better than -1.7 dB insertion loss and -20 dB return loss. Over the whole bandwidth up to 110 GHz, the return and insertion loss of a single 45° stage is better than -15 dB and -1.7 dB, respectively. The measurements also show that the return loss of the single 30° and 15° phase shifter is better than -10 dB from 10-100 GHz in both the up and downstate position, with the maximum insertion loss of -1.7 dB. At the design frequency of 75 GHz the return loss is better than -25 dB for the 30° phase-shifter stage and better than -12 dB for the 15° phase shifter with an insertion loss less than -0.9 dB. Figure 3.10(b) shows the relative phase shift of all fabricated single-stage phase shifters versus frequency from 1-110 GHz. 3.3.2 Linear-coded phase shifters Figure 3.11 plots the RF measurement results for the linear-coded 7×45° phase shifters. For the seven-stage linear-coded phase shifter, the return loss is better than -12 dB while the insertion loss is less than -4.1 dB for all combinations of all eight phase-shift steps at the nominal frequency of 75 GHz. Over the whole bandwidth up to 110 GHz, the 7×45°linear-coded phase shifter has a return loss that is better than -12 dB with an insertion loss that is better than -5.1 dB. Therefore, despite the fact that HRS is a substrate material with inferior microwave properties compared to glass and quartz, the novel 7×45° phase shifter has a phase shift performance of 74.1°/dB at 75 GHz and performs a phase shift

phase resolution:15°

phase resolution:45° 5.4 mm

5.51 mm

45° 45° 45° 45° 45° 45° 45°

45° 45° 45° 45° 45° 30° 15°

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per length of 592.1°/cm and at 110 GHz its phase shift per loss is 97.1°/dB, and its phase shift per length is 916.7°/cm.

(a) measured return and insertion losses

(b) measured phase shifts

Figure 3.10. Measured RF results: (a) upstate and downstate figure-of-merit of 30° and 15° phase shifter as compared to 45°; (b) comparison of different measured phase shifts.

Figure 3.11. RF measurements of the multi-stage 7×45° linear-coded dielectric-block phase shifter: return and insertion loss for actuating multiple stages (left), phase of S21 for different configurations (right). Note that both pictures use the same color scale.

3.2.3 Binary-coded phase shifters Figure 3.12 shows the measured insertion and return loss of the binary-coded 15°+30°+5×45° phase shifter for one to all seven stages actuated. Its phase shift is measured from 1 to 110 GHz. At the design frequency of 75 GHz, maximum return and insertion loss are -17 dB and -3.5 dB, respectively, which correspond to a loss of -0.82 dB/bit, and a phase shift efficiency of 71.1°/dB and 490.02°/cm.

upstate downstate

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The phase shifter is exceptionally broadband over the whole W-band with a maximum return loss better than -12 dB and a maximum insertion loss of less than -4 dB (98.3°/dB; 715.6°/cm at 110 GHz). Figure 3.12. Measured return and insertion loss, and measured phase shift for all actuation states of multi-stage binary-coded 15°+30°+5×45° phase shifter. Figure 3.13 plots the phase shifter efficiency (phase shift/length and phase shift/loss) for the frequency range from 75-110 GHz. Table 3.2 compares the figure of merit between the linear-coded 7×45° phase shifters and the binary-coded 15°+30°+5×45° phase shifter. Table 3.3 compares the novel binary-coded 15°+30°+5×45° phase-shifter designs to HEMT, MEMS switched delay line, and distributed MEMS transmission line phase shifters [83], [96]-[99]. The binary-coded MEMS phase shifter presented in this thesis has better insertion loss per bit and better return loss than any phase shifter previously reported, both for their respective nominal frequency and for the whole W-band. In designs [96] and [98], three-bit and one-bit phase shifters have better phase shift per loss at their nominal frequencies, respectively, but perform worse than the presented 4.25-bit phase shifter over the rest of the W-band. This performance data of the first prototype designs demonstrates the advantage of the new MEMS phase shifter concept over other MEMS and GaAs-switch phase shifters. This is true despite the fact that the present technology utilizes silicon both as a substrate and for the dielectric block and as a substrate, in contrast to the other high-performance phase shifters that are fabricated using low-loss glass or GaAs substrates. Figure 3.13. Measured performance of the 15°+30°+5×45° phase shifter: measured phase shift per length and phase shift per loss for the frequency range from 75-110 GHz.

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Table 3.2. Comparisons of design data and measurement results between multi-stage linear-coded 7×45° and binary-coded 15°+30°+5×45°.

linear coded 7×45°

binary-coded 15°+30°+5×45°

design data stage numbers

7

7

phase-shift combinations 19 8 max. Δφ at 75 GHz (°) 315 270 phase resolution at 75 GHz (°) single-stage length (µm) - 45° stage - 30° stage - 15° stage total length (mm) measured key performance max. IL 75-110 GHz (dB) max. RL 75-110 GHz (dB)

45

760 - -

5.4

-5.1 -12

15

760 800 850 5.51

-4 -12

Δφ/loss at 75 GHz (°/dB) 74.1 71.05 Δφ/length at 75 GHz (°/cm) Δφ/loss at 110 GHz (°/dB) Δφ/length at 110 GHz (°/cm)

592.1 97.1

916.7

490.02 98.3

715.6

Table 3.3. Comparisons of several state-of-the-art W-band phase shifters.

reference [97] [96] [83] [98] [99] This work

binary coded

linear coded

substrate GaAs glass Si quartz quartz Si Si nominal frequency fn (GHz)

94 78 76.5 90 80 75 75

number of bits 4 3 3 1 2 4.25 3 configuration possibilities 16 8 8 2 4 19 8

max. Δϕ of fn (˚) 337.5 315 315 180 282 270 315 max. IL of fn (dB) >-8 >-3.2 >-5.8 >-2.5 >-6.1 >-3.5 >-4.5 max. IL/bit of fn (dB/bit) -2 -1.07 -1.93 -2.5 -3.05 -0.82 -1.5

max. Δϕ/loss of fn (˚/dB) 56.25 95.75 55.26 85.71 70.5 71.05 74.1

max. Δϕ/lossa (˚/dB) - 83.3 - 32.85 - 98.3 97.1

max. RL of fn (dB) <-15 <-12 <-12 <-12 <-9.5 <-17 <-12

max. IL at W (dB) >-10 >-6 >-8 >-6 >-7 >-4.1 >-5.1

max. RL at W (dB) <-8 <-9 <-7.5 <-3 <-9 <-12 <-12

Note: at W: over the whole W-band 75-110 GHz. a at 110 GHz

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3.4 MEMS characterization and reliability 3.4.1 Actuator characteristics The pull-in behavior and the surface leveling of the electrostatic actuator are characterized by a white-light interferometric profilometer. Figure 3.14 plots the actuation curves compared to the simulation results by COMSOL Multiphysics for a three-stage and a five-stage meander flexure. The measured pull-in voltage of the three-stage serpentine flexure is 29.9 V and 7.1 V for the five-stage meander flexure, which matches the simulated values of 27.5 V and 6 V, respectively. With the measured pull-in voltages and the calculated effective spring constants of the whole structure are 36.67 N/m and 3.53 N/m for the 3-stage and 5-stage meander flexures, respectively. Figure 3.14. Measured and simulated DC-actuation characteristics of three-stage serpentine flexures. For the five-stage meander flexure, only simulation results are given because the flexure is too soft and, therefore, too unstable to provide reproducible pull-in results. The displacement is measured by a white-light interferometer. Figure 3.15. 3D surface scan of the height profile from a white-light interferometric profilometer (left) and a local surface roughness profile of the dielectric block (right). The monocrystalline-silicon block also provides excellent surface roughness and leveling relative to the bottom wafer. The measured local root-mean square (RMS) roughness of the top surface of the dielectric block is smaller

3-stage serpentine flexure, k= 36.67 N/m

5-stage serpentine flexure k= 3.53 N/m

-1.3 µm 40 µm 20 10 30

-118 nm -32 nm 0 -30 -60 -90

monocrystalline-silicon block

serpentine flexure

anchor

DRIEed hole

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than the measurement resolution of the optical profilometer of 3 nm RMS (see Fig. 3.15). Figure 3.16 shows the leveling profile of the movable silicon block which is measured by referring to the flat CPW surface on the bottom wafer (x-y plane). The measured data shows that the leveling of the silicon block along the x-z plane is as good as 62.5 nm over the block width of 380 µm and 15.4 nm along the y-z plane over the block length of 760 µm.

(a)

(b) Figure 3.16. Cross-section image of the dielectric-block phase shifter with x-z and y-z cut plane (a); (b) leveling profile and cross section of x-z cut plane (h1-h2=62.5 nm), and leveling profile and cross section of y-z cut plane (h1-h2=15.4 nm) over the block width of 380 µm and the block length of 760 µm. 3.4.2 Life-cycle and reliability test The susceptibility to dielectric charging by the actuation voltage was investigated by measuring the pull-in voltage of 100 subsequent cycles with unipolar DC actuation voltage. Neither stiction nor any tendency to change the pull-in voltage was observed (see Fig. 3.17), which proves the low susceptibility of the Si3N4 bumps to dielectric charging, compared to completely filled dielectric layers [36]. The mean value of this specific actuation sequence is 29.57 V with a standard deviation of 0.32 V.

y-z plane

x-z plane

reference plane (x-y) (CPW)

reference

h2 h1

x-z plane y-z plane

dielectric block

anchor

reference

h2 h1

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Life-cycle tests evaluated the mechanical reliability of the dielectric-block phase shifter in a nonhermetic environment by applying a 50-V unipolar rectangular-pulse actuation voltage of 10 kHz with a duty-cycle of 50%. All the tested phase shifters with three-meander 36.67 N/m mechanical springs (pull-in voltage 29.9 V) survived 1 billion cycles after which the tests were discontinued. A single device with much-weaker five-stage 3.53-N/m meander springs (pull-in voltage of 7.1 V) failed at 110 million cycles by stiction. Figure 3.17. Pull-in voltage measurements of subsequent 100 cycles with 50-V unipolar actuation, showing no evidence of dielectric charging. Furthermore, the acceleration sensitivity of the dielectric block to the external mechanical shock was calculated by:

𝜑

𝑎= 𝑚

𝑘∙ ∆𝜑

𝑥 (3.2)

where φ/a is the acceleration sensitivity (°∙s2∙m-1), m is the mass of the dielectric block (kg), k is the effective spring constant (N/m), and Δφ is the phase shift (°) at the displacement x (m). From Equation (3.2), the acceleration sensitivity of the 45° phase shifter block at 75 GHz in the up-state position was calculated to a relative phase shift of 0.0056°∙s2∙m-1. The down-state is even less sensitive to the external acceleration because the block is electrostatically clamped, which drastically increases its effective spring constant. The parameter, Δφ/x, was extracted from Figure 3.2 by reading out the slope of the plotted curve between the phase shift and displacement. 3.5 Non-linearity analysis 3.5.1 Power modulation Despite the fact that the mechanical resonance frequencies of the device are a magnitude of at least 104 lower than the electrical signals in the GHz range, the effective value of the signal line voltage creates a force that acts on the moveable parts in any RF MEMS device. Therefore, although the component is passive and inherently linear, a single-tone input signal also results in a single tone output signal, and variations in the voltage (power) level of the electrical signal result in variations in the upstate position of the dielectric block. This causes a nonlinear

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phase modulation that occurs in every MEMS phase shifter. The degree of the nonlinearity depends on the stiffness of the mechanical suspension of the moving parts [100]-[104]. The phase shift in S21, which depends on the input power level, is measured for the presented devices using a power amplifier, filters for filtering the nonlinearities of the amplifier, two 50-Ω high-power impedances and two circulators at the ports to protect the signal generator and the power amplifier from the reflected mismatched power. Figure 3.18 shows the measured deviation in the phase shift of a 45° single stage plotted over the input power level and measured at 3 GHz and up to 40 dBm (the phase shift is extrapolated to the nominal frequency of 75 GHz). Figure 3.18. Measured phase shift and phase error in the dependence on input RF power. The simulated mechanical resonant frequency is 63.1 kHz for the 45° phase shifter block at 75 GHz in the upstate position. The downstate is even less sensitive to the external acceleration because the block is electrostatically clamped, which drastically increases its effective spring constant. Therefore, a power modulation (amplitude modulation) with a modulation frequency, fm, of the RF signal, fc, results in an additional phase modulation with the frequency, fm, as displayed in Figure 3.19 for a single 45° stage, if the frequency, fm, of the amplitude modulation is below the mechanical resonance frequency, f0, of the device, which is in the order of a few tens of kHz. Thus, the output power spectrum will include phase-modulation sidebands at fc ± n∙fm beside the amplitude modulation sidebands, as shown in the figure for different power modulation parameters. For a power modulation between 10 to 20 dBm, the sidebands created by the phase modulation are below -11.71 dBc, and for a power modulation between 10 to 40 dBm, the sidebands rise to -6.57 dBc. 3.5.2 Intermodulation products Similar to the phase modulation discussed in previous subsection that results from an amplitude modulation of a single-tone signal, a dual-tone, unmodulated signal may cause phase modulation if the Δf = f1-f2 is smaller than the mechanical cut-off frequency, fm, of the device. This is caused by the fact that the envelope of the total power of the two steady-state signals is modulated with Δf, and results in a

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phase modulation, much like the subsection of the upstate position of the device discussed in the previous subsection.

(a) input signal modulation

(dBm)

carrier and modulating signal frequency

spectral components (single-sideband power spectral density: SSB PSD)

fc (GHz) fm (kHz)

fc+2fm (dBc)

fc+3fm (dBc)

fc+4fm (dBc)

fc+5fm (dBc)

10-12 dBm 75 10 -84.1 -149.3 -218.1 -265.7 10-20 dBm 75 10 -61.4 -117.1 -176.4 -238.1 10-30 dBm 75 10 -50.1 -98.5 -150.4 -204.7 10-40 dBm 75 10 -42.4 -84.2 -129.4 -177.2

(b) Figure 3.19. Responses of a single 45° stage to a power-modulated signal: (a) schematic illustration of phase modulation caused by a power modulated signal, if the modulation frequency, fm, is smaller than the mechanical cut-off frequency, f0; (b) spectral components of the input and output signals, calculated from the measured power dependent phase shift (Figure 3.18) for modulating power as listed in the table. The nonlinearity characteristics of the 45° stage can be calculated from the capacitance modulation derived from the measurements (see Figure 3.18) and by using the models introduced in [100]-[104]. Figures 3.20 plots the third intermodulation level and intercept point of the third order (IIP3) with different RF input power from the transmitted signal, which shows an IIP3 of 49.27 to 47.97 dBm for total input power levels of 10 to 30 dBm. The nonlinearity

V V

t t

envelope (fm)

Δφ

AM AM+PM input signal

output signal f0

f Δφ

P

PSD PSD fc-fm fc+fm fc-fm fc+fm

fc+2fm fc-2fm

fm f0 fc fm f0 fc

AM AM+PM

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decreases with greater mechanical stiffness of the structure with the additional parameter of the block distance from the transmission line in the upstate position (see Figure 3.20). The effective spring constant of 36.67 N/m and the initial gap of 5 µm in the presented design are a good compromise between an acceptable low actuation voltage of 30 V, which is characterized by white-light interferometric profilometer, and good RF performance in terms of nonlinearity and phase shift per unit length. Figure 3.20. Nonlinearity characteristic of a single 45°-stage phase shifter with two-tone signal at 75 GHz + 10 kHz (Δf): third IM level and IIP3 with different RF input power (left); IIP3 for different initial distances of the block for varied mechanical spring constant for the total signal power of 40 dBm (right). 3.6 Thermal properties and power-handling analysis 3.6.1 Design comparison of conventional MEMS phase shifters Figure 3.12 presents the schematically functional drawings and concept comparisons of the two conventional MEMS phase-shifter concepts and the novel unconventional concept. The two conventional concepts employ air-suspended thin metallic bridges, either for capacitive switches in a time-delay network, as in the TTD concept in Figure 3.21(a) or for capacitive loading of the CPW as in the DMTL concept in Figure 3.21(b). The thin metallic bridges employed in these conventional designs are bottlenecks in terms of power handling. The bridges must be relatively thin for acceptable actuation voltages, which results in high current densities at elevated power levels, especially at higher frequencies. These high power densities, in connection with the poor thermal conductivity of the freestanding bridges to the substrate heat sink, result in a large temperature rise in the bridges [88], [105]. The electrical resistance of the gold bridges also increases with temperature and frequency, which results in a much higher temperature rise, particularly when the gold bridges are in the downstate. Gold is the most common material for fabrication of bridges, which is subject to degradation even at slightly elevated temperatures; the mechanical elasticity of the material degrades, which results in altered actuation behavior and drastically limits the life time of the devices [106]-[111]. At even higher temperatures, significant thermo-mechanical deformations of the gold bridges can occur, resulting in serious reliability issues including permanent plastic deformation, thermal-strain-induced buckling, creeping, or even melting. Furthermore, most of the MEMS phase shifters are fabricated on quartz or glass to minimize substrate losses, but offer poor thermal properties. For example, fused quartz has approximately 111 times lower thermal conductivity than silicon

design

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[104]. Therefore, the fused-quartz substrate is a bad heat sink for the thin movable gold bridges. This also limits the power handling capability of the conventional phase-shifter designs significantly.

(a) conventional concept: TTD phase shifters

(b) conventional concept: DMTL phase shifters

(c) novel concept (this work): dielectric-block phase shifters Figure 3.21. Operational concept comparison between (a) switched TTD network; (b) DMTL phase shifter, and (c) dielectric-block phase shifter (this work). The novel all-silicon phase-shifter concept developed in this thesis and shown in Figure 3.21(c), employs vertically moving dielectric blocks for tuning the effective dielectric constant of the transmission line. Therefore, the wave propagation speed creates a linear phase shift. Metallic losses are only caused in the t-line itself, as the moving dielectric blocks are only subject to minor dielectric losses. All mechanically moving parts, i.e., the dielectric block, the suspending springs and the anchors, are fabricated out of a single monocrystalline-silicon

downstate

upstate

CPW Si3N4 layer

2-bit TTD phase shifter [98]

2-bit DMTL phase shifter [112]

upstate

downstate

thin air-suspebded gold

bridge

thin air-suspebded gold

bridge

Si3N4 layer CPW

upstate

downstate

4.25-bit phase shifter

monocrystalline-silicon block

Si3N4 bumps

anchor serpentine flexure

CPW with etched slots

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block, which offers the best-possible mechanical reliability in microsystem technology. The micromachined CPW is fabricated on a low-loss HRSS, which also is an excellent heat sink for the power dissipated in the transmission line. Because this concept eliminates the metallic bridges and the all-silicon design, the dielectric-block phase shifter is more suitable for high-power and high-frequency applications since the power handling is only limited by the transmission line and not by any MEMS moving parts. Moreover, this concept offers better RF performance over the whole W-band compared to all previously published conventional MEMS phase shifters on any substrate, including quartz and glass [113]. This novel concept also offers excellent mechanical properties with high reliability as discussed in the previous sections [114]. Figure 3.22. High-power measurement setup with infrared microscope and large-signal network analyzer with input power up to 43 dBm (20 W) at 3 GHz. The dielectric-block phase shifter is placed on a temperature-controlled chuck with a constant temperature of 65°C. 3.6.2 Measurement setup and results at 3 GHz Figure 3.22 presents the block diagram of the high-power measurement setup. The setup is composed of a signal generator connected to a 3-GHz tunable solid-state power amplifier with a maximum tuning range of up to 46 dBm (40W). A circulator is used to protect the power amplifier from the reflected signal, so the reflected power is dissipated at a 50-Ω termination that is connected to the other port of the circulator. Four directional couplers are required to detect the transmitted and reflected signals at the input (a1 and b1) and output (a2 and b2) port of the phase shifter. These parameters are used to control and stabilize the gain of the power amplifier as it reaches the required RF power level at the input port of the device under test (DUT) by the large-signal network analyzer. Although DC blockage during actuation is not required because the DC probes were launched on the spring anchor and CPW ground, rather than on the signal line and ground as in the conventional designs, the bias-tee circuit is used to protect the measurement setup from all possible DC leakage. The dielectric-block phase shifter is then placed on a temperature-controlled chuck to stabilize temperature of the DUT at 65°C, which protects the measurement from the unwanted external heat source. Additionally, another 50-Ω load is required to dissipate the power at the end terminal of the measurement setup. The feed-back automatic gain-controlled setup presented in this thesis is more reliable than the commonly used setup [115] where the input RF power at the terminal of the

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device can drastically affects the required power due to power losses at the circulator, bias-tee circuit, t-line, and RF probes.

(a) single 45° stage: upstate (left), and downstate (right)

(b) single 30° stage (left), and 15° stage (right) in the upstate

(c) t-line without block (single-stage) Figure 3.23. Top-view infrared-microscope pictures of the single 45°, 30°, and 15° stage phase shifters (the phase-shift values of 45°, 30° and 15° are offered at 75 GHz) with 40 dBm (10 W) signal power at 3 GHz. The color scale is for all plots. Figure 3.25 presents the measured and simulated cross section plots (y-yʹ) of the temperature rise. Figure 3.23 shows the top-view infrared-microscope pictures of the single 45°, 30°, and 15° stage phase shifters with the input power of 40 dBm (10 W) at 3 GHz. The measurement results along the y-yʹ line show that the maximum temperature rise of the single 45° stage in the upstate position is 10.56°C and 12.19°C for the downstate position. The hottest spots with input signal power of

CPW probe

y-yʹ

serpentine flexure

anchor

760 µm

y-yʹ

y-yʹ y-yʹ

dielectric block

760 µm

y-yʹ

DC probe tip

800 µm

800 µm

850 µm

36°C

0

24

12

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37 dBm (5 W) and 43 dBm (20 W) in the upstate are 6.37°C and 27.28°C, respectively. For the single 30° and 15° stage, the peak temperature increases at 40 dBm in the upstate position are 11.75°C and 11.67°C, respectively, while the hottest temperature rise of the unloaded CPW without the dielectric block is only 10.77°C. These hotspots locate on the signal line of the gold CPW, which are determined by the skin effect and input power of the signal line.

(a) all up state (total length = 5.4 mm)

(b) fifth stage was actuated Figure 3.24. Top-view infrared-microscope pictures of the 7×45° multi-stage phase shifters with 40 dBm (10 W) signal power at 3 GHz: (a) all stages are in the upstate position; (b) the fifth stage is in the down state. All pictures have the same color scale. Figure 3.25 presents the cross-section plots (y-yʹ) of the temperature rise. Figure 3.25. The cross-section plots (y-yʹ) of the temperature rise: single 45° stage phase shifter compared to the t-line from Fig. 3.23 (left); and multi-stage phase shifter in Fig. 3.24 (right). The high-power measurement is performed at 3 GHz and 40 dBm (10 W). Figure 3.24 presents the thermal measurements of the 7×45° multi-stage linear-coded phase shifter. For all silicon blocks in the upstate position, the peak temperature rise along the y-yʹ line is 11.06°C and 10.63°C when the fifth-stage silicon block is actuated, i.e., pulled down. The hottest spots occur on the signal line of the CPW and are determined by the skin effect and power level of the input RF signal. As shown in Figure 3.24(b), the actuated silicon block performs as an

CPW probe

stage 1 stage 2 stage 3 stage 4 stage 5 stage 6 stage 7

stage 1 stage 2 stage 3 stage 4 stage 5 stage 6 stage 7

y-yʹ

y-yʹ

pulled-down

y-yʹ y-yʹ

36°C

24

12

0

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Novel RF MEMS Devices for W-Band Beam-Steering Front-Ends

additional heat sink for the devices, which results in an approximate 3°C lower temperature rise at the corners of the silicon block compared to when it is in the upstate position. Figure 3.25(a) and (b) present the y-yʹ cross-section plots of the single 45° stage, both in the upstate and the downstate position, compared to the unloaded t-line and the 7×45° linear-coded phase shifters, respectively. 3.6.3 Extension of results to 75 GHz RF/electro-thermal simulations (CST Studio Suite) mapped the measurement results at 3 GHz with a signal power of 40 dBm (10 W), see Figure 3.23, and the simulation parameters were calibrated from microwave and thermal measurement results. Figure 3.26 shows a simulated temperature rise of the cross-section y-y´ of the phase shifter with the thermal contour plot. In the upstate position, the temperature rise of the phase shifter is 9°C and is distributed uniformly across the suspended silicon block. In the down state, the thermal resistance between the movable silicon block and the structures underneath (t-line and substrate) decreases, which results in much better heat flow properties. Therefore, heat can flow from the heat source (signal line) through the middle of the movable silicon block and flows to the heat sink (substrate) where the temperature is slightly lower. Thus, the distribution of the rising temperature changes from 12°C in the middle of the silicon block to 8°C at the edges. Figure 3.26. Simulated results at 3 GHz and 40-dBm power of single 45° stage with contour plots of the temperature-rise distribution of cross-sections of the dielectric-block phase shifters; upstate (left) and downstate (right) position. Figure 3.27. Demonstrations of heat transfer characteristics of the dielectric-block phase shifter under infrared microscope measurement; upstate (left) and downstate (right) position.

12°C 9°C

6.2°C 3.4°C

9°C

6.2°C

3.4°C monocrystalline-

silicon block

CPW with deep etched grooves

anchor

upstate downstate

12°C

11°C 8°C 5°C

6.5°C

4.5°C

monocrystalline-silicon block

anchor

upstate downstate

heat from t-line heat from block heat from block heat from t-line

HRS substrate HRS substrate

anchor anchor

gold gold SiO2 SiO2

bonding polymer

bonding polymer

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Novel RF MEMS Devices for W-Band Beam-Steering Front-Ends

To correlate the thermal simulations with the infrared measurements, it is necessary to consider that silicon has good transparency at the infrared wavelength. However, due to the lattice vibration, impurity, free carrier absorption, plasma resonance and surface roughness, the transparency in silicon can decrease by up to 50% [116]. Therefore, the movable silicon blocks of the phase shifters are partially transparent during the thermal power measurements with an infrared microscope camera. Therefore, the temperatures measured by the infrared camera in Figure 3.23 and 3.24 consist of two infrared-radiation components: (1) emission of the heated transmission line due to metallic losses radiated through the silicon block, and (2) the actual temperature radiation of the air-suspended silicon block (see Figure 3.27). The thermal results were mapped and the two components for the silicon block and the CPW were extracted from the simulation. The superposition of these two components agrees well with the measurements. Figure 3.28 plots the measured temperature rise of the single 45° stage phase shifter, both in the up and down state, and compare them to the simulated results from CST Studio Suite. Figure 3.28. Temperature-rise distribution comparisons (at 3 GHz and 40-dBm power) between simulated and measured results of the cross section y-yʹ of the single 45°-stage dielectric-block phase shifters; upstate (left) and downstate (right) position. The simulated maximum temperature rise was calculated by the maximum value between the temperature rise of the t-line and the dielectric block.

Figure 3.29. Comparison of the peak temperature rises of the single 45° stage dielectric-block phase shifter with the power sweep from 37 to 43 dBm (5 to 20 W) at 3 and 75 GHz.

upstate downstate

y-yʹ y-yʹ

simulation simulation measurement measurement

t-line (simulated)

t-line (simulated)

dielectric block (simulated)

dielectric block (simulated)

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Since no power sources of 10W at 75 GHz were available, the temperature rise at 40 dBm (10 W) signal power was measured up to 3 GHz, and these measurement results were used as calibration points to map the RF/thermal simulations, with CST Studio Suite, from 3 up to 75 GHz. In this way, frequency dependent thermal effects such as increased metallic loss and skin effects are taken into account. Figure 3.29 shows the comparison of the measured and simulated peak temperature rise of the single 45° stage dielectric-block phase shifter at 3 GHz with power sweep from 37 dBm to 43 dBm (5 W to 20 W). The extended results of the maximum temperature rise at 75 GHz are also plotted.

(a) novel concept: dielectric-block phase shifters on silicon

(b) conventional concept: TTD phase shifters on quartz [98]

(c) conventional concept: TTD phase shifters on silicon

(continued)

0 5 10 15 20 25 30°C (peak temperature rise)

0 108 216 325 433 542 650°C (peak temperature rise)

0 55 110 165 220 275 330°C (peak temperature rise)

downstate upstate

dielectric block

serpentine flexures

anchor

CPW direction of propagation

direction of propagation

direction of propagation

direction of propagation

upstate downstate

CPW

MEMS switch

upstate downstate

direction of propagation

direction of propagation

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3. Novel W-Band Phase Shifters 36

Novel RF MEMS Devices for W-Band Beam-Steering Front-Ends

(continued)

(d) conventional concept: DMTL phase shifters on quartz [112]

(e) conventional concept: DMTL phase shifters on silicon Figure 3.30. Simulated temperature rise at 75 GHz and 40 dBm of (a) a single-stage 45° dielectric-block phase-shifter on high-resistivity Si substrate, (b) single-switch of TTD phase shifter on quartz, and (c) on HRSS, (d) a 45° stage of a DMTL phase shifter on quartz, and (e) on silicon substrate, respectively. The thermal simulations were conducted at 75 GHz with 40 dBm (10 W) signal power with the same chip areas for all simulations. Note that the color scales are different for the different subfigures. 3.6.4 Comparisons of power handling A comparative study on the thermal heating induced by RF power was carried out for the novel MEMS phase-shifter concept, by comparing it to the two conventional MEMS phase-shifter concepts (DMTL, and TTD). Figure 3.30 compares the simulated temperature rise of single MEMS phase-shifter modules of the three concepts at 75 GHz and 40 dBm signal power, for both discrete states (up and downstate positions). In an effort to reproduce the RF performance of the conventional designs, the geometry and material parameters of the TTD switch and the DMTL W-band phase shifters for the RF and electro-thermal simulation models are taken from [98] and [112], respectively. Figure 3.31 plots the simulated temperature increase of the hottest spots for the different designs at 75 GHz, over signal powers from 20 to 40 dBm (10 W). For the novel dielectric-block phase shifter (45° stage), the maximum

0 50 100 150 200 250 300°C (peak temperature rise)

0 20 40 60 80 100 120°C (peak temperature rise)

upstate downstate

direction of propagation

direction of propagation CPW

bridge 1

bridge 2

bridge 3

bridge 4

upstate downstate

direction of propagation

direction of propagation CPW

bridge 1 bridge 2 bridge 3 bridge 4

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temperature rise is only 30°C, whereas the maximum temperature rises for the TTD switch and DMTL phase shifter (45°) are as high as 650°C and 300°C, respectively, which is by a factor of more than 20 and 10 times more than the novel dielectric-block phase shifter, respectively. Even if a silicon substrate, offering a better heat sink, would be used for the conventional phase shifters, which, however, would drastically degrade their RF performance by 2.5-4 dB in the W-band, the temperature increase would still be 330°C and 120°C for the switched-TTD and the DMTL designs, respectively (11 and 4 times higher than the novel design, respectively). Figure 3.31. Simulated peak-temperature rises of the three MEMS phase-shifter concepts, at a signal power from 0.1 to10 W (20-40 dBm) at 75 GHz. Furthermore, it must be noted that the conventional MEMS phase shifter concepts are more susceptible to small temperature increases because they are made of soft metals whose mechanical elasticity degrades at increased temperatures, which affects actuation reproducibility and reliability. For gold, a very common material for fabricating the bridges of the conventional phase shifters, peak temperatures must be kept below an absolute temperature of 80°C in order to avoid degrading the gold bridges [106]-[108]. Thereore, the MEMS TTD and DMTL designs can only be operated up to 31 dBm (1.5 W), and 34 dBm (2.5 W) on quartz substrates, or 33.8 dBm (2.4 W) and 38.39 dBm (6.9 W) on silicon substrates, which gives much worse RF performance compared to the novel concept. Aside from the fact that the novel dielectric-block phase shifters' peak temperature rise is only 30°C even at 40 dBm at 75 GHz, all mechanically active parts are fabricated from monocrystalline-silicon, which maintains its mechanical properties well up to 900°C [117]. Thus, the thermal power handling of the novel device concept is only limited by the t-line rather than by the MEMS parts. Table 3.4 compares the overall performance of the different concepts. 3.7 Design optimization for minimizing signal reflection Although this dielectric-block phase shifter already offers an excellent overall RF performance, the return loss is initially only designed for the best performance for a single-stage phase shifter. However, the average return loss of all these states can further be optimized in multi-stage devices where different numbers of blocks are in the up and downstate.

conventional concepts

this work

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3. Novel W-Band Phase Shifters 38

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Table 3.4. Comparative study of power-handling capability of three MEMS phase shifter concepts. dielectric block

(novel) DMTL

(conventional) TTD

(conventional)

commonly-used substrates Si* glass/quartz/ (Si**)

glass/quartz/ GaAs/(Si**)

structural material Si* metals (Au) metals (Au) substrate thermal conductivity very good Poor very poor

dissipated power in moving elements extremely low very high extremely high

temperature rise at high signal power very low very high extremely high

overall high-power handling capability excellent very bad at high

frequency extremely bad

peak temperature rise (40 dBm at 75 GHz) 30°C 300°C 650°C

power handling at 75 GHz > 40 dBm (> 10 W)

34 dBm (2.5 W)

31 dBm (1.5 W)

limitation of power handling t-line metallic bridge*** MEMS

switches*** plastic deformation temperature of the MEMS part

very high very low very low

RF losses for the whole W-band low low/moderate high

MEMS reliability at high power very good Moderate low-moderate

linearity very good very good very good design complexity at high frequency low Low moderate

Notes: *high-resistivity silicon. **Si improves thermal behavior, but drastically decreases RF performance. ***power handling capability is lower due to plastic deformation of the gold bridges

(around 80°C). 3.7.1 Two-stage phase shifters For a single-stage dielectric-block phase shifter, the length of the dielectric block is chosen to minimize the reflected signal from both sides of the block, while the width and thickness of the block are already optimized to achieve the maximum relative phase shift between the upstate and the downstate of the block. Therefore, the only parameter for optimizing a multi-stage device that consists of a sequence of dielectric blocks that can be individually actuated to achieve the different phase states, is the distance between the stages. Figure 3.32 plots the simulation results of the return and insertion loss of the two-stage dielectric-block phase shifter at the design frequency of 75 GHz as a function of the distance between the two stages. The data is presented for the three basic operation cases: 1) both dielectric blocks in the upstate position, 2) both

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3. Novel W-Band Phase Shifters 39

Novel RF MEMS Devices for W-Band Beam-Steering Front-Ends

blocks in the downstate position, or 3) one block in the upstate and one block in the downstate position. A minimum return loss of -41.3 dB is obtained for both blocks in the down-state at a distance, L, of 290 µm. For both blocks in the up-state, the minimum return loss of -37.9 dB is achieved at a distance of 750 µm. With one block in the up-state and the other block in the down-state, a distance of 190 µm offers the best return loss of -25.4 dB. It is important to consider that the distance between the two stages should be as short as possible because the insertion loss, which does not have any explicit minima, increases linearly with the total length of the device and thus the distance between the blocks. No specific distance optimizes the phase-shift state of a two-stage device. Therefore, it can be assumed that the optimization of a phase shifter with more than two stages is even more complicated and an equidistance optimization strategy (i.e., constant distance between the blocks of a multi-stage device) may not be favored. Figure 3.32. Simulation results of the return and insertion loss of the three different phase-shift combinations at the nominal frequency of 75 GHz (combination1: both blocks in the upstate; combination 2: both blocks in the downstate and combination 3: one block in the upstate and one block in the downstate position) as a function of the distance (L) between the two-stage dielectric-block phase shifter. 3.7.2 Multi-stage phase shifters The previous sections (see Figure 3.36(a)) demonstrate that different phase-shift states have different optimum frequencies for the return loss. Therefore, for the design frequency with given bandwidths around it, the average return loss of all phase-shift combinations is the criteria for the optimization. The return-loss optimization for the multi-stage dielectric-block phase shifter is achieved by varying the distances between each phase-shifter stage. The overall device length and the distances between the blocks must be kept at a minimum to keep the insertion loss acceptably low. By taking into account the specific sequence for actuating the blocks. It might be beneficial to employ non-equidistant lengths between each stage by taking the specific actuating sequence into account, and also a non-symmetrical device, as the actuation sequence for achieving the different phase-shift values is non-symmetrical.

stage 1

L

two-stage phase shifter

stage 2

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Figure 3.33. Seven different multi-stage dielectric-block phase-shifter designs with different strategies for choosing the distances between each stage (L1-L6). Figure 3.33 presents seven different multi-stage phase-shifter designs whose distances between the seven phase-shifter blocks (L1-L6) are varied according to the different optimization strategies. Table 3.5 lists the individual distances and the total device length. The distance of the design 1 (in this thesis) is kept at a minimum of 10 µm to minimize the total device length (5380 µm) and, ultimately, the insertion loss. The distances in design 2 are 420 µm for all sections, which is a quarter of the wavelength for the unloaded t-line and results in a total length of 7840 µm. The distances of all the sections in design 3 are chosen from Figure 3.32. The distances follow the strategy of minimizing the return loss for all dielectric-blocks in the down-state position and have a total length of 7060 µm. Design 4 has a total length of 9940 µm, and the distance between each stage is 770 µm following the strategy of Figure 3.32, where the return loss is minimized for all blocks in the up-state position. In designs 5 and 7, non-equidistant strategies are employed, and the distances are chosen to increase exponentially to result in a non-equidistant phase-shifter design with a total length of 5950 µm and 18970 µm, respectively. To compare the performance of such a non-equidistant strategy to an equidistant device of the same insertion loss design 6 is an equidistant design with a length of 5950 µm that results in a distance of 105 µm between the stages.

design 1 (Fig. 3.9(b)): Ln+1=10 µm, where n=0...5. shortest length; equidistant.

design 2: Ln+1=420 µm, where n=0...5. equidistant; λ/4 distance.

design 3: Ln+1=290 µm, where n=0...5. equidistant; distance minimized for two blocks down Fig. 3.32.

design 4: Ln+1=770 µm, where n=0...5. equidistant; distance minimized for two blocks up Fig. 3.32.

design 5 (best performance): Ln+1=10×2n µm, where n=0...5.

design 6: Ln+1=105 µm, where n=0...5. equidistant; same length as design 5.

design 7: Ln+1=10×4n µm, where n=0...5. non-equidistant; longest length.

5380 µm 7840 µm

7060 µm 9940 µm

5950 µm 5950 µm

18970 µm

L1 L1

L1 L1

L1 L1

L1

L2 L2

L2 L2

L2 L2

L2

L3 L3

L3 L3

L3 L3

L3

L4 L4

L4 L4

L4 L4

L4

L5 L5

L5 L5

L5 L5

L5

L6 L6

L6 L6

L6 L6

L6

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Table 3.5. Distances between each phase-shifter stage of seven different optimization strategies (Figure 3.33).

design distance between each phase-shifter stage total

length (µm)

L1 (µm)

L2 (µm)

L3 (µm)

L4 (µm)

L5 (µm)

L6 (µm)

1 10 10 10 10 10 10 5380 2 420 420 420 420 420 420 7840 3 290 290 290 290 290 290 7060 4 770 770 770 770 770 770 9940 5 10 20 40 80 160 320 5950 6 105 105 105 105 105 105 5950 7 10 40 160 640 2560 10240 18970

3.7.3 Simulation results Figure 3.34 plots the CST-Studio-Suite-2011 simulated return and insertion loss of designs 1, 2, 3, 5, 6, and 7 for all phase-shift states at their nominal frequencies, 500-MHz and 1-GHz bandwidth around that frequency. The dashed lines connect the average return and insertion loss of all phase-shifter states. The simulation results from design 4 are not plotted since it is obvious that the total length of this design results in an unacceptably high insertion loss. Table 3.6 shows the simulated results of the average return and insertion loss at their nominal frequency, including the bandwidth of 500 MHz and 1 GHz of the designs presented in Figure 3.34. It can be seen that the non-equidistant design 5 offers the lowest average return loss for all considered bandwidths, while still maintaining a good average insertion-loss performance that is second only to design 1 with minimum distances between the blocks. At the nominal frequency, design 5, as compared to design 1 (published), gives a return-loss improvement of 11.8 dB and, 9.57 dB and 8.06 dB for 500-MHz and 1 GHz bandwidth, respectively. The average insertion loss of design 5 is slightly higher than the previously-published work (design 1) by 0.7 dB. Design 2 also has an improved average return loss of at least 1.95 dB, over design 1, but it suffers a much worse insertion loss increase by at least 2.86 dB. Design 3, which actually follows the optimization strategy identified in Figure 3.32 for a two-block device, results in worse insertion loss and an even worse average return loss. Although the total length of design 6 (equidistant) is the same as design 5 (best design, non-equidistant) and results in the same insertion loss, the average return loss of the equidistant design 6 is much worse than the return loss of design 5 (best performance, non-equidistant). The second non-equidistant design concept (design 7) has a degraded insertion loss because of its long length, and has a much worse return loss.

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Figure 3.34. Simulated return and insertion loss of all phase-shift combinations (0-360°) for the seven dielectric-block phase-shifter designs. The plots display the simulated performance at their nominal frequencies, within 500-MHz, and 1-GHz bandwidth. The dashed lines connect the average return and insertion loss of all phase-shift states. Table 3.6. Simulated return and insertion loss. design 1 2 3 5 6 7 nominal frequency (GHz) 76.25 75 75 75 74 75.9 average RL at nominal frequency (dB)

-19.2 -25.9 -20.5 -31.0 -25 -20.8

average RL at 500-MHz bandwidth (dB)

-19.3 -21.7 -17.3 -28.8 -24.1 -18.3

average RL at 1-GHz bandwidth (dB)

-17.8 -19.8 -15.2 -25.9 -22.7 -16.2

average IL at nominal frequency (dB)

-3.6 -6.49 -6.06 -4.35 -4.41 -7.65

average IL at 500-MHz bandwidth (dB)

-3.6 -6.51 -6.04 -4.35 -4.39 -7.78

average IL at 1-GHz bandwidth (dB)

-3.65 -6.51 -6.03 -4.35 -4.39 -7.9

design 1 design 2 design 3

design 6 design 7 design 5 (best design)

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Figure 3.35. Microscope and SEM image of the best-performance minimum-return-loss multi-stage dielectric-block MEMS phase shifter (design 5) with a total length of 5.95 mm. These simulation results show that the optimization of the multi-stage dielectric-block phase shifter with non-equidistant, exponential distribution of the distance between each phase-shifter stage, as implemented in design 5, is the best of the investigated optimization strategies. However, design 7 shows that the overall length must be short. The fact that the best performance is delivered by an asymmetrical, non-equidistant device is attributed to the specific actuation sequence of how this seven-stage phase-shifter operates. If the actuation sequence was mirrored, i.e. start from block 7 to block 1 instead of block 1 to block 7, the return loss in design 5 would be worse by approximately -10 dB. 3.7.4 Measurement results Designs 1, 5, and 7 were fabricated and measured to verify the simulation results. Figure 3.35 shows a photomicrograph and an SEM image of the best-performance device (design 5, non-equidistant). Figure 3.36(a)-(c) present the measured return and insertion loss for the whole W-band of designs 1, 5, and 7 for all phase-shift combinations, and compares them to the simulation data by showing a very good agreement between the simulations and measurements. It can be seen that this phase-shifter concept (i.e., loading the line by vertically moving dielectric blocks) has a very good broadband performance, even if the return loss has distinct minima. Figure 3.37 shows the plots of the measured return and insertion loss for all phase-shift combinations at the nominal frequencies, and within 500-MHz and 1-GHz bandwidth; an analogy of Figure 3.34. For the best-performance design (design 5), the measurement results show the average return loss at the nominal frequency of 75 GHz of -26.2 dB, with the average insertion loss of -5.12 dB. The average return and insertion loss at 500-MHz bandwidth are -21.1 dB and -5.11 dB, while offering the average return loss of -18.108 dB and the average insertion loss of -5.15 dB at the bandwidth of 1 GHz. Design 1 has an average return loss of -19.2 dB and an average insertion loss of -4.20 dB at the nominal frequency of

stage 3

stage 4

stage 5

stage 1

stage 2

stage 3

stage 4

stage 5

stage 6

stage 7

L1 10µm

L2 20µm

L3 40µm

L4 80µm

L5 160µm

L6 320µm

total length 5950 µm

direction of propagation

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76.25 GHz. The average return and insertion loss of design 2 at 500-MHz and 1 GHz bandwidth are -17.6 dB and -4.23 dB, and -17.5 dB and -4.24 dB, respectively. For design 7, an average return and insertion loss of -19.9 dB and -8.81 dB are observable at the nominal frequency of 75.9 GHz. The average return and insertion loss at 500-MHz bandwidth are -18.3 dB and -8.89 dB, with an average return and insertion loss of -17.3 dB and -8.87 dB at 1-GHz bandwidth.

(a) design 1

(b) design 5 (best performance)

(c) design 7 Figure 3.36. Plots of measured and simulated return and insertion loss of the dielectric-block phase shifters; (a) design 1 (minimum distance, equidistant, (b) design 5 (best performance, non-equidistant), and (c) design 7 (non-equidistant). From the measurement results, design 5 (best-performance design) shows the best average return loss with an improvement of 6.97 dB, as compared to the previous work (design 1), at the nominal frequency. The average return loss of

measured

measured

measured

simulated

simulated

simulated

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design 5 at 500-MHz and 1-GHz bandwidth are 3.46 dB and 0.57 dB better than design 1. The average insertion loss of design 5 is only 0.92 dB higher than design 1 at the nominal frequency, with 0.87 dB and 0.91 dB higher at 500-MHz and 1-GHz bandwidth, respectively. Compared to design 7, design 5 offers a better average return loss of 6.24 dB at the nominal frequency, while the average return-loss improvements are 2.78 dB and 0.72 dB at 500-MHz and 1-GHz bandwidth. Moreover, the average insertion loss of design 7 is 3.68 dB higher than design 5 at the nominal frequency, and 3.77 dB and 3.72 dB higher at 500-MHz and 1-GHz bandwidth. Table 3.7 summarizes the measurement and simulation results in analogy to Table 3.6. Therefore, both the measurement and simulation results agree that design 5 offers the lowest return loss, compared to the other investigated dielectric-block phase-shifter designs, compromised to the minimum distance design (design 1) only by the insertion loss, which is still the smallest of all investigated alternatives. Figure 3.37. Summary of measured return and insertion loss of the dielectric-block phase-shifters design 1, 5 and 7 at their nominal frequencies, 500-MHz and 1-GHz bandwidth. The dashed lines connect the average return and insertion loss of the phase shifter for all phase-shift combinations. Table 3.7. Measured average return and insertion loss. design 1 5 7 nominal frequency (GHz) 76.2 75 75.9 average RL at nom. freq. (dB) -19.2 -26.2 -19.9 average RL at 500-MHz BW (dB) -17.5 -21.1 -18.3 average RL at 1-GHz BW (dB) -17.6 -18.1 -17.3 average IL at nom. freq. (dB) -4.20 -5.12 -8.81 average IL at 500-MHz BW (dB) -4.24 -5.11 -8.89 average IL at 1-GHz BW (dB) -4.23 -5.15 -8.87

design 1 design 5 (best design) design 7

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4. Substrate-integrated helical antennas Axial mode helical antennas offer many desirable advantages over other antennas such as high gain, wide operational bandwidth and circular polarization [118]-[122]. With these attractive characteristics, helical antennas play an important role in various wireless applications and remote sensing such as radar systems, mobile networks, radio-astronomy, satellite networks and fixed communication links [123]-[129]. New helical-antenna designs and their enhanced characteristics have been intensively investigated such as multifilar helical antennas [130]-[134], helical-fed dielectric-rod antennas [135]-[138], helicone antennas [139]- [141] and square helical antennas [142]-[144]. Furthermore, a design study has recently been published that investigates the feasibility of a folded helical antenna that can be integrated in a printed circuit broad, but no radiation patterns are reported [145]. Future wireless and radar technologies are moving towards higher frequencies, therefore; the dimensions of microwave systems and antennas become more attractive and feasible for integration into a semiconductor wafer substrate, often even together with integrated circuits. There is only one report in the literatures that demonstrates on-chip helical antennas to date based on a wire-boding based semi-helical antenna for 5.2 GHz that is fabricated on the surface of a high-resistivity silicon semiconductor substrate [146]. However, this structure resulted in poor radiation performance, since most of the electromagnetic energy is confined inside the substrate. To the knowledge of the author, it has never been reported on an on-chip helical antenna above 10 GHz, or even on integrating such an antenna inside a semiconductor-grade substrate. This chapter investigates a novel concept of a 3D micromachined axial-mode dielectric-core helical antenna designed for 75 GHz that is integrated into a seminconductor-grade HRS wafer. In contrast to the conventional on-chip antenna design that typically utilizes only the surface of the substrate, the proposed antenna concept combines bulk and surface micromachining processes to utilize the whole volume of the wafer for the antenna body. This novel concept results in a chip-area efficient, high-gain 3D helical radiator. The maximum gain of this antenna is enhanced by patterning a tailor-made silicon core as well as modifying the ground structure of the antenna. This chapter also investigates the influence of silicon substrate and dielectric core etching, the matching transition between the helical structure and the CPW, and the size and geometry of the ground structure on the performance of the antenna. Furthermore, a line array of eight elements is demonstrated. 4.1 Substrate-embedded square helical antennas Square helical antennas loaded with a dielectric core are well studied and have demonstrated their excellent performance, while offering a circular polarization [142]-[144], as compared to conventional circular-cross-section helical antenna. For W-band wavelengths, the size and cross-section of the square helical structure are feasible to be implemented on a standard semiconductor wafer that can be achieved by 3D micromachining, which is based on fabrication processes for microelectromechanical systems (MEMS) that utilize standard semiconductor manufacturing facilities. These facilities offer the advantages of high level of

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integration, high product uniformity, large-volume manufacturability, and potential low cost. Figure 5.1 represents a perspective model of the proposed antenna concept, for an eight-turn substrate-embedded axially radiating square helical antenna that is completely embedded inside the volume of a HRSS with a resistivity that is greater than 4 kΩ∙cm, a dielectric constant of 11.9 and a loss tangent of 0.0006 (±50% error; measured at 75 GHz) [88]. All of the simulation and gain reports includes these dielectric losses, as well as the metallic losses of the gold film of finite conductivity and thickness, which are significant for these types of helical antennas in the W-band. Figure 4.1. 3D model of an eight-turn substrate-embedded square-cross-section helical antenna fully embedded in a HRSS. The antenna is designed for the nominal frequency of 75 GHz. Part of the substrate inside the core of the antenna is removed to eliminate the imaginary part of the input impedance. Note that the substrate in the model is drawn with slight transparency for visualization improvement. The geometry of the eight-turn 75-GHz square helical antenna in Figure 4.1 was analytically determined [118]-[122] and optimized by Ansys HFSS. For this design study, the fabrication feasibility is taken into account throughout the antenna design and optimization process in order to manufacture the antenna with standard micromachining processes. The antenna is composed of a gold square-cross-section helical that is fully embedded in an 870-µm-thick HRSS. A 2-µm-thick gold pattern on top of the front and back of the wafer are connected by 40×40-µm2 metalized through-wafer holes, which can be implemented by a standard DRIE process, and then coating the sidewalls with a gold film of 500 nm thickness. Each side of the cross-section of the antenna is 870-µm long, which corresponds to a quarter of the electrical wavelength, and the separation between each turn is 1020 µm in order to provide axial radiation. The substrate-integrated helical antennas proposed in this paper are fed via a transmission line on the surface of the wafer which allows for compatibility with planar circuits on the RF substrate. Insert I of Figure 4.1 shows that the 50-Ω

total length 8160 µm

I

I

II

II

z

y x

2-µm-thick gold

8-mm-long ground plane

870-µm-thick HRS wafer

direction of propagation

50-Ω CPW feed

105-Ω CPW λ/4 matching transition

substrate removed

874 µm

870 µm

1020 µm

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transmission line (signal line/gap : 65/40 µm) feeds the antenna via a 105-Ω λ/4 CPW matching transition (signal line/gap : 3/50 µm). To eliminate the reactance part of the input impedance of the antenna [135]-[138], a part of the silicon wafer inside the core must be completely removed by deep-etching the silicon wafer (insert II in Fig. 4.1) at the beginning of the helical structure. This process results in a pure resistive input impedance of 220 Ω. An 8-mm-long, laterally expanding ground plane is designed for the helical radiator by vertically embedding a gold layer in the substrate that is perpendicular to the wafer plane. This vertical ground plane can be implemented in the same fabrication process as the metalized through-wafer holes. 4.2 Influence of substrate Simulations of the reflection coefficient and a radiation characteristic of the basic square helical antenna design were performed (see Fig. 4.1). However, such an antenna design results in a poor signal reflection S11 of only -3.77 dB at the nominal frequency, because the electromagnetic (EM) wave propagating inside the core of the helical structure is reflected back to the feed point. This occurs because a huge impedance mismatch exists at the boundary interface between the air and silicon inside the core. This antenna also results in poor radiation performance and radiation efficiency, with a maximum gain of only -6.5 dB for the forward radiation. The reason for the poor performance is that for silicon with a high dielectric constant, even though it results in a compact design, EM fields and signal energy are mostly confined inside the substrate, and only a small fraction of EM power radiates as intended from the helical structure. To reduce these effects, the silicon substrate that surrounds the antenna and silicon core should be partly removed to decrease the effective dielectric constant of the helical structure in order to increase the radiation efficiency. Compared to the forward direction, the back radiation of the EM field on the y-z plane is considerably high because the ground plane size in the vertical direction is not sufficiently large for functioning as a good reflector. Different possibilities for improving the design are analyzed in this chapter. To increase the radiation efficiency of the design introduced in Figure 4.1, the effective dielectric constant of the antenna structure must be reduced. This can be achieved by partially removing the substrate surrounding the helical antenna and patterning the silicon core. From a fabrication feasibility point of view, this can be implemented by micromachining through a DRIE step, that is similar to the through-wafer connections. Moreover, the back radiation characteristic of the antenna can be drastically reduced by extending the ground plane in the vertical direction. Figure 4.2 presents a 3D model of the modified substrate-integrated square-cross-section helical antenna with the silicon substrate partially removed and an extended vertical ground plane.

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Figure 4.2. 3D model of a substrate-integrated square-cross-section helical antenna with silicon substrate partially removed and extended vertical ground plane. The antenna is designed for the nominal frequency of 75 GHz.

Figure 4.3. Simulated reflection coefficients from 60-100 GHz of air-core substrate-integrated square cross-section helical antenna with different silicon-to-air etch-ratios of the substrate. To investigate the influence of the silicon substrate surrounding the antenna structure on the radiation characteristics, the helical antenna is modeled for different etch-perforations (silicon-to-air etch-ratios) of the substrate, with the air core illustrated in Figure 4.2. Figures 4.3 and 4.4 show the simulated reflection coefficients and the maximum-gain performance of an air-core helical antenna with different silicon-to-air etch-ratios of the HRSS. The basic antenna design without any perforations (Si 0% removed) results in a reflection coefficient that is worse than -5 dB for the whole band from 60-110 GHz (see Figure 4.3). Furthermore, the simulated gain

z

y x

8000 µm

4.87 mm total length 8160 µm

etched-patterns

square helix

silicon core

direction of propagation

antenna feed

air core

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pattern (Figure 4.4) shows that this antenna design radiates very poorly since the silicon substrate surrounding the helical structure has a high dielectric constant, which results in most of the EM energy being confined outside the helix.

maximum gain (dB)

(a) x-z plane (phi=0°)

(b) y-z plane (phi=90°) Figure 4.4. Simulated maximum gain patters of air-core substrate-integrated square cross-section helical antennas and their nominal frequencies for different silicon-to-air substrate etch-ratios for the frequency found to have best radiation performance (in parentheses).

Si 0% removed (75 GHz) Si 50% removed (71.5 GHz) Si 90% removed (82.5 GHz) Si 100% removed (83.5 GHz)

Si 0% removed (75 GHz) Si 50% removed (71.5 GHz) Si 90% removed (82.5 GHz) Si 100% removed (83.5 GHz)

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For an antenna with an etch-ratio of 50%, S11 is -26.2 dB at the best radiation performance frequency of 71.5 GHz and the maximum gain is 6.7 dB at the F/B ratio of 10.8 dB. However, the antenna still has a very wide HPBW in both the x-z and y-z planes, as well as a very narrow operational bandwidth of 500 MHz. When removing the silicon substrate to 90%, the helical radiator achieves a reflection coefficient of better than -16.7 dB at 82.5 GHz and the operational bandwidth is greatly improved to 15 GHz. This antenna design has a maximum gain of 10 dB with an F/B ratio of 15.7 dB and a HPBW of 40° and 48° on the x-z and y-z plane, respectively. For an antenna with an etch-ratio of 100% (no substrate surrounding the antenna), the helical radiator has a reflection coefficient of better than -19.8 dB at the best-performance frequency of 83.75 GHz. The antenna offers a slightly lower maximum gain of 9 dB, compared to the previous case, with an F/B ratio of 16.7 dB. The antenna has a HPBW of 44° on the x-z plane and 56° on the y-z plane. Therefore, the etched-pattern with a 90% substrate removal is the best strategy for improving the performance of the antenna. 4.3 Influence of dielectric core The influence of the substrate perforations in the air core has been investigated. The HFSS model is composed of a helical structure, a feed network with a matching transition, and a silicon core with a varying silicon-to-air ratio without any surrounding substrate.

Figure 4.5. Simulated reflection coefficients from 60-100 GHz of air-core substrate-integrated square cross-section helical antenna with different silicon-to-air etch-ratio of the core. Figure 4.5 shows the simulated S11 of the helical antenna with a different etch-ratio. The antenna with a full silicon core (0% removed) has a sharp resonance in the return loss of -41.6 dB at 88 GHz, while the design with 25%-etched core has such a return loss resonance of -48.4 dB at 92.25 GHz. For the antenna design with 50% etch-pattern, a return loss of -34.7 dB is achieved at 99.5 GHz. Moreover, the return loss of -25.3 dB (at 88 GHz) and -19.8 dB (at 83.75

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GHz) are achieved for the helical antennas with 75% and 100%-etched core, respectively. However, the best directivities of these antenna designs are achieved around 75 GHz, the nominal frequency for the given antenna geometry, and not at the best-return-loss frequencies. Therefore, the radiation figure-of-merits of these antenna designs are discussed and compared for 75 GHz (Figure 4.6), where the reflection coefficients of these designs are between -7 and -10 dB.

maximum gain (dB)

(a) x-z plane (phi=0°)

(b) y-z plane (phi=90°)

Figure 4.6. Simulated radiation characteristics of the core substrate-integrated square cross-section helical antenna at the nominal frequency of 75 GHz with different percentages of the substrate of the core removed.

Si 0% removed Si 50% removed Si 90% removed Si 100% removed

Si 0% removed Si 50% removed Si 90% removed Si 100% removed

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Figure 4.6 plots the simulated radiation characteristics of these helical-antenna designs with different etch-ratios of the silicon core. The maximum gain of the antenna structure with a full solid core is only 2.2 dB for a front radiation and an F/B-ratio is 7.5 dB at 75 GHz and a HPBW of more than 172° in both planes. For the design with a 25% etched-core, the antenna has a 6.8-dB gain, an F/B ratio of 17.5 dB, and a HPBW of 64° in both planes. The antenna design with a 50% etched core offers a maximum gain of 11.3 dB, an F/B ratio of better than 20 dB, and a HPBW smaller than 40° for both radiation planes. The designs with an etched core of 75% and 100% achieve maximum gains of 8.2 and 8.4 dB, respectively. These designs have an F/B ratio better than 12.2 dB with a HPBW of better than 60°. Therefore, the best performance is achieved when 50% of the silicon core of the helical antenna is removed.

(a) design 1

(b) design 2 (c) design 3 Figure 4.7. 3D model of optimized substrate-integrated helical antennas with substrate and core removed by 90% and 50%, respectively.: (a) design 1 with extended ground plane (six times wafer thickness or 1.5λ); (b) design 2 with ground plane width corresponding to the wafer thickness (λ/4); (c) design 3 with extended ground plane in planar-cup shape.

z

y x

8000 µm

4870 µm

total length 10.2 mm

ground plane

air core mechanical

support

square helix

tailore-made silicon core

direction of propagation

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4.4 Ground plane variations The optimum substrate-integrated helical antenna design with the surrounding substrate and silicon core are removed by approximately 90% and 50%, respectively, as shown in Figure 4.7(a). The 90% etching of the surrounding substrate is modeled by etch holes with a pitch of 435 µm, which is λ/8 and thus small enough for the perforated substrate to be perceived as a near-ideal homogeneous material of a tailor-made permittivity of close to two for the etched area of 90%. With this pitch and etched area, the remaining silicon bridges are 22 µm wide, which requires a depth-to-width etching aspect ratio of 38:1. This is clearly below the capability of state-of-the-art silicon deep etching processes of 50:1 in commercially available equipment. From a mechanical point of view, the 90% etched-area-perforations result in a stability decrease by a factor of 10 compared to the initial 870-µm-thick wafer; however, the total resulting stability corresponds to the stability of an unetched wafer with a thickness of 410 µm. Therefore, the 90% perforated 870-µm-thick substrate has still higher mechanical stability than a 300 µm thick substrate, which is the minimum standard size of 100 mm diameter wafers. Figure 4.7 shows that the two ends of the helix antenna have been further optimized: close to the ground plane, the silicon core is removed completely for impedance matching purpose, and for the tip section of the antenna, both the helix and the surrounding supporting substrate are removed to further increase the directivity. The design 1 (Figure 4.7(a)) comprises an extended ground plane that is approximately six times thicker than the helix core (1.5λ). For a helix antenna to be completely integrated in the substrate, the ground height should preferably be as thick as the helix thickness which is equal to the wafer thickness, which is approximately λ/4, and is investigated in design 2 (Figure 4.7(b)). Furthermore, this section analyzes the influence of a planar-cup shaped ground plane on the directivity, as seen in design 3 (Figure 4.7(c)). Fig. 4.8 plots the simulated reflection coefficients of the substrate-embedded helical antenna designs in Fig. 4.7. All designs have a return loss better than -22 dB at their best frequency which is around the design frequency and all designs offer a return loss below -15 dB, a bandwidth of at least 6 GHz.

Figure 4.8. Simulated reflection coefficients from 60-100 GHz of the square cross-section helical antenna designs of Figure 4.7. The antennas are designed for 75 GHz .

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maximum gain (dB)

(a) x-z plane (phi=0°)

(b) y-z plane (phi=90°)

Figure 4.9. Simulated radiation characteristics of the square cross-section helical antenna designs in Figure 4.7: (a) x-z plane; (b) y-z plane. Figures 4.9(a) and (b) show the radiation patterns (maximum gain) of these three antenna designs, in the x-z and y-z planes, respectively. The maximum gain of the design 1 is 12.2 dB, while the gain of the design 2, which limits the ground plane to a single-wafer thickness, shows a worse performance by 3 dB. Design 3, with extended and planar-cup in-plane shaped grounds, achieves a 13.2 dB gain. Furthermore, for the sidelobe levels in the x-z plane, design 3 shows the best performance, but design 3 is second to design 1 and about equal to design 2 in terms of the sidelobe levels in the y-z plane. The half-power beam-widths of designs 1 and 2 are approximately equal and around 40° both for the x-z and y-z planes, while the planar-cup shaped ground of design 3 shows a drastic

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improvement to 28° for the y-z plane. The planar-cup shaped ground also performs best in the front-to-back firing ratio, with 23.6 dB compared to 20.0 and 9.6 dB in designs 1 and 2, respectively. Table 4.1 summarizes the key performance data. Table 4.1. Summary of the key performance of design 1, 2, and 3 of Figure 4.7. design design 1: extended

ground plane (six times wafer

thickness)

design 2: single-wafer ground plane (single-

wafer thickness)

design 3: extended ground plane,

planar-cup shaped

max. gain (dB) 12.2 9.3 13.2

side-lobe level x-z plane (dB)

-10.8 -9.6 -14.7

side-lobe level y-z plane (dB)

-12.82 -9.6 -9.77

HPBW x-z plane (°) 40 40 40 HPBW y-z plane (°) 40 44 28 front-to-back ratio (dB) 19.95 9.6 23.55

4.5 Antenna arrays Placing a planar radiator into a wafer substrate allows for creation of line arrays of antennas in the wafer plane, as shown in Fig. 4.10. This drastically increases the directivity of the radiation in the wafer plane and allows for electronic beam-steering, i.e., a complete substrate-integrated beam-steering system could be embedded on a single semiconductor substrate.

Figure 4.10. Substrate-integrated linear helical antenna array in wafer plane, modeling of line array of eight antennas with spacing of 0.5λ (2 mm) at the best-radiation performance frequency of 74.5 GHz (design frequency is 75 GHz). Figures 4.11(a) and 10(b) plots the radiation patterns of different line antenna array configurations based on the single antenna in design 1. The parameters and radiation performance are summarized in Table II. For a spacing of 0.5λ, a number of antennas of four elements achieves maximum gain of 18.2

antenna 1 antenna 2

antenna 3 antenna 4

antenna 5 antenna 6

antenna 7 antenna 8

total length: 22 mm.

0.5λ (2 mm) direction of propagation

z

y x

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dB with sidelobe levels of 12.2 and 20 dB for y-z and x-z planes, respectively. A larger number of antennas, as demonstrated for 8 and 16 line elements, results in a lower half-power beamwidth due to the increasing total aperture, but also creates a higher sidelobe since the half-power beamwidth of a single antenna is about 40°. A line array with 16 elements of λ/2 spacing, is 38 mm wide and achieves a half-power beam width in the y-z plane of 6° and a gain of 24.2 dB.

maximum gain (dB)

(a) x-z plane (phi=0°)

(b) y-z plane (phi=90°)

Figure 4.11. Substrate-integrated linear helical antenna array in wafer plane: (a) and (b) simulated x-z, and y-z radiation patterns of antenna gain for line arrays of 4, 8, and 16 elements, with spacing of 0.5λ (2 mm) at the best-radiation performance frequency of 74.5 GHz (design frequency is 75 GHz).

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Table 4.2. Summary of the linear array performance composed of N-elements helical antennas from Figure 4.7(a). N* 4 8 16 spacing 0.5λ 0.5λ 0.5λ max. gain (dB) 18.2 21.2 24.2 side-lobe level x-z plane (dB) -12.3 -12.3 -12.2 side-lobe level y-z plane (dB) -21.2 -17 -14 HPBW x-z plane (°) 38 38 38 HPBW y-z plane (°) 24 12 6 front-to-back ratio (dB) 20 20.1 20.2 total length (mm) 14 22 38 Note: *N is the number of elements 4.6 Feed network and matching 4.6.1 Four-stage Chebyshev matching transformer A Chebyshev or equal-ripple multi-section matching transformer provides a very large operational bandwidths at a given number of transmission line sections compared to the other matching techniques such as the λ/4-transformer and the binomial multi-section matching network. The increased operational bandwidth of the Chebyshev matching transformer comes at the cost of an increased ripple over the passband of the matching network. However, the maximum allowable reflection coefficient for the design of the Chebyshev transformer may be designated. The Chebyshev transformer exploits the characteristics of the Chebyshev polynomials. Using the properties of the Chebyshev polynomials, it is possible to design matching networks with a reflection coefficient at or below a prescribed level over a wide bandwidth.

matching section

calculated matching impedance (Ω)

calculated geometries of CPW signal-line

width (µm)

gap (µm)

length (µm)

1 92 9 53 300 2 78 17 50 300 3 63 32 45 300 4 54 50 40 300

Figure 4.17. Four-section Chebyshev transformer network for matching a 50-Ω antenna to a transmission line with a characteristic impedance of 100 Ω at 75 GHz. The calculated geometries of the CPW transmission lines for such a matching network are included in the table.

l1=λ/4 l2=λ/4

l3=λ/4 l4=λ/4

t-line Z0=100 Ω Z1=92 Ω Z2=78 Ω Z3=63 Ω Z4=54 Ω

antenna Za=50 Ω

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Figure 4.17 shows the calculated impedances required for a four-section Chebyshev transformer that matches a 50-Ω antenna to a 100-Ω transmission line at the nominal frequency of 75 GHz. If the magnitude of the reflection coefficient (|Γ|) is smaller than 0.2 (-14 dB), the calculated matching impedances, Z1 to Z4, are 92 Ω, 78 Ω, 63 Ω, and 54 Ω, respectively (Figure 4.17). Theoretical calculations demonstrate that the individual length of each matching section is 300 µm, which is equal to one quarter of the wavelength of the design frequency. This four-section Chebyshev matching network achieves an operational bandwidth of 109.5 GHz with a signal reflection better than -14 dB from 20.2 GHz to 129.7 GHz. Furthermore, the maximum ripple of this matching-network design is calculated to be -33.7 dB (|Γ| = 0.02). Figure 4.17 also presents the calculated geometries of the matching transmission-line sections with CPW technology for integrating the matching network with the antenna structures discussed in the previous section. 4.6.2 3-dB power divider A 3-dB T-junction power divider is utilized in the proposed beam-steering front-end, which provides two equal in-phase EM powers from a signal generator to two output ports that can be connected to the antennas or the input port of the other stage of the power divider. To increase the operational bandwidth, the multi-stage Chebyshev matching transformer is integrated with the 3-dB T-junction power-divider design.

Figure 4.18. Schematic drawing of the 3-dB T-junction power divider integrated with four-section Chebyshev matching transformers implemented on a HRS substrate. The power divider is designed for the nominal frequency of 75 GHz. The distance between port 2 and 3 (between two antennas) is 0.5λ (2 mm). Figure 4.17 shows the calculated geometries of the CPW of each matching section. Figure 4.18 shows the schematic model of the ultra-wideband 3-dB T-junction power divider integrated with four-section Chebyshev matching networks at the nominal frequency of 75 GHz. The EM power from a microwave source is fed to the network at port 1 and equally divided to ports 2 and 3. The design performance can be improved by employing metal air bridges at the T and L-discontinuities. Moreover, the operational bandwidth of the 3-dB power divider is greatly enhanced by integrating the four-section Chebyshev transformers along the signal paths from port 1 to ports 2 and 3, respectively.

input port 1 (50 Ω)

output port 2 (50 Ω)

output port 3 (50 Ω)

100-Ω lines

50-Ω line 50-Ω input impedance

air bridge

92 Ω

78 Ω

63 Ω

54 Ω

0.5λ (2 mm)

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Figure 4.19. Simulated S-parameters from 50 to 110 GHz of the 3-dB T-junction power divider integrated with four-section Chebyshev matching transformers implemented on a HRS substrate in Figure 4.18. The power divider is designed for the nominal frequency of 75 GHz. The simulated S-parameters from 50-110 GHz of the proposed 3-dB power splitter with four-section Chebyshev matching transformers are shown in Figure 4.19. From the simulation results, the 3-dB power splitter achieves a signal reflection at port 1 (S11) of better than -31.4 dB at the nominal frequency of 75 GHz, while offering S11 lower than -25 dB from 50-110 GHz. Signal reflections at ports 2 and 3 (S22 and S33) are -8.5 dB at 75 GHz, and better than -7.5 dB for 50 to 110 GHz. Moreover, the signal isolation between ports 2 and 3 (S23 and S32) is -7.7 dB at the design frequency, and lower than -7.3 dB for 50-110-GHz band. Finally, the signal transmissions between ports 1, 2 and 3 (S21, S12, and S13) are -3.7 dB at 75 GHz. The additional loss of approximately 0.7 dB of the 3-dB power divider comes from the transmission-line length and the air bridges. For an eight-element linear antenna array, multiple stages of the 3-dB power divider can be placed in a series to compose a one-input eight-output (nine port) network. However, the transmission-line lengths of each stage must be considered to fulfill the requirements of the matching network and the antenna array design.

S11

S12,S13, S21 and S31

S22 and S33 S23 and S32

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4. Substrate-Integrated Helical Antennas 62

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6. Conclusions 63

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6. Conclusions The first part of this thesis presented a novel single and multistage all-silicon W-band MEMS phase-shifter concept. The concept was based on multiple-step deep-reactive-ion-etched monocrystalline-silicon dielectric blocks that were transfer bonded to an RF substrate containing a 3D micromachined CPW transmission line. The concept features the following design elements for superior RF performance and improved reliability over conventional MEMS phase shifters, i.e. DMTL phase shifters and MEMS TTD networks, all based on moving thin metallic bridges.

• The dielectric-block phase shifters are fabricated out of high-resistivity silicon wafers that offer very low signal losses and the best RF performance to date in terms of return loss and maximum insertion loss per bit, either at the nominal frequency or at the whole W-band.

• All mechanical parts, including the moving block, mechanical springs, and

support anchors, are etched out of the same, continuous monocrystalline silicon bulk layer for the best mechanical reliability, in contrast to the thin metallic bridges that are prone to plastic deformation. The life-cycle test reveals that this novel phase shifter can be operated continuously for more one billion cycles without observing any degradation of actuation characteristics.

• The dielectric constant of the moving block can be tailor-made in a wide

range by varying the etch hole size, which results in different relative phase shifts and increases the phase shift resolution.

• Since dielectric blocks are used for tuning impedance of the transmission

line, the power handling is only limited by the transmission line, in contrast to conventional phase shifters where the currents induced in the thin bridges result in substantial device heating and thus power limitations.

• No detectable dielectric charging as no closed dielectric isolation layer is

used.

• Reduced substrate losses and substrate charging by deep-etched grooves in the transmission line.

Moreover, this thesis studied the novel concept of W-band substrate-integrated axial helical antennas, where the square helices are proposed to be 3D micromachined inside a high-resistivity silicon semiconductor substrate. Optimum designs were identified in terms of feasibility for microfabrication and radiation performance. Furthermore, antenna feeding and matching were successfully optimized from a feeding coplanar waveguide. This study found that the surrounding substrate and the core of the antenna must be partially etched in order

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6. Conclusions 64

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to provide high radiation performance without compromising mechanical stability. Furthermore, it was found that the radiation performance could be enhanced if the ground layer width goes beyond the thickness of a single wafer. Finally, substrate-integrated line antenna arrays were investigated, for higher gain in the wafer plane. In conclusion, substrate-integrated, in-plane radiating axial helical antennas were found to be feasible for W-band applications, and could be realized by state-of-the-art standard micromachining processes in the high-resistivity silicon substrate.

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Acknowledgements 79

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Acknowledgements My study period at Microsystem Technology Laboratory (MST), KTH-Royal Institute of Technology, is one of the most valuable times in my academic and personal life. There are so many people here who I really appreciate their help and support. I would like to thank Prof. Joachim Oberhammer and Prof. Göran Stemme for giving me this good chance in conducting fruitful researches, and in studying toward my PhD under their excellent supervision. I can still remember the moment when I came for the job interview here, and the first time in my life that I saw a real fully equipped cleanroom which inspired me to explore the world of microsystem technology. During my time at MST, I have learned a lot from people at MST about how to conduct a top-class research and how to write an excellent research paper, especially, from my main supervisor, Prof. Joachim Oberhammer. I, of course, should not forget the funding agencies who fully funded me for all these years under NORDITE-SARFA project which is supported by VINNOVA (Sweden), TEKES (Finland), and RCN (Norway). Moreover, our partners, both academic and industrial, in this project always give me nice suggestions and supports during the time that we work together. Also, many thanks to the IEEE Microwave Theory and Techniques Society, the IEEE Antennas and Propagation Society, and the European Microwave Association for giving me special funding of approximately 130,000 SEK for my personal expenses through their awards and fellowship programs. Furthermore, I would like to thank the following people for their technical support, discussion, and valuable time in helping me out with my research and experiment. I would not be able to finish my research work, and study without them. Firstly, Prof. Antti Räisäinen (Aalto University), and Dr. Tauno Vähä-Heikkilä (VTT-Technical Research Center of Finland) for supporting me in RF measurement with their network analyzer. Secondly, Prof. Jahn Grahn, and Prof. Kristoffer Andersson (Chalmers University of Technology) for assisting me in using their large-signal network analyzer, and a lot of expensive microwave equipments for measuring thermal property, power handling capability, and non-linearity of my devices. Also, µFab network (MC2-Chalmers, and Ångström-Uppsala) in using their modern cleanroom facilities. Moreover, I would like to thank Prof. Gabriel Rebeiz from the University of California, San Diego, for very fruitful discussions, and for giving me very useful advices in RF MEMS. Finally, with my great appreciation for their valuable time, I would like to thank the opponent, Prof. Pierre Blondy (Université de Limoges), and the thesis committee: Prof. Mikael Östling (Integrated Devices and Circuits Laboratory, KTH), Prof. Staffan Rudner (FOI-Swedish Defence Research Agency), and Prof. Vassen Vassilev (Microwave Electronics Laboratory, Chalmers University of Technology). I also would like to thank the professors and the senior researchers at MST. A very big thank to Prof. Wouter van der Wijngaart, who always gives me nice advices about academic and personal life, and suggesting me when I have problems. Also, Prof. Hans Sohlström for guiding me in teaching various courses

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at MST and reviewing my thesis. Not to be forgotten, Prof. Frank Niklaus and Prof. Niclas Roxhed in suggesting me many fruitful ideas regarding the cleanroom process. Moreover, many thanks to my colleagues and ex-colleagues: Erika, Fredrik C., Kjell, Kristinn, Tommy, Sergey, Mikael A., Zargham, Simon, Valentin, Andreas, Fredrik F., Hithesh, Henrik, Staffan, Mikael K., Martin, Gaspard, Farizah, Niklas, Stephan, Umer, Mikael S., Fritzi, Thomas, Adit, Björn, Stefan B. Biggest special thanks to Adit who was helping me everything when I moved to Sweden. Also, a special thank to Mikael S., Cecilia, and Stefan B. in teaching me a lot of thing in the cleanroom when I was working there for the first time. In addition, I would like to thank my family and friends who always understand and support me either in good or bad time. Thank you very much!

Nutapong Somjit KTH Royal Institute of Technology

Stockholm, Sweden May 2012.

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Summary of Appended Papers 81

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Summary of appended papers Detailed summary Paper 1: Binary-Coded 4.25-Bit W-Band Monocrystalline-Silicon MEMS Multi-Stage Dielectric-Block Phase Shifters. This paper introduces a novel ultra-broadband digital-type multiple-stage binary-coded microwave MEMS phase shifter concept with best loss/bit at the nominal frequency and best maximum return and insertion loss performance over the whole W-band applications. The relative phase shift is controlled by vertically moving a dielectric, monocrystalline silicon block above the 3D micromachined coplanar waveguide. Periodically etched patterns in the dielectric block with three different sizes are introduced to tune the effective dielectric constant artificially, resulting in a relative phase shift of 45°, 30°, and 15° for single stages at 75 GHz. Paper 2: Deep-Reactive Ion-Etched Wafer-Scale-Transferred All-Silicon Dielectric-Blocks Millimeter-Wave MEMS Phase Shifters. This paper reports the design, fabrication, and characterization of a novel multi-stage microwave MEMS phase-shifter, based on multiple-step deep-reactive ion-etched monocrystalline-silicon dielectric blocks that are transfer-bonded to an RF substrate containing a 3D micromachined coplanar waveguide. The measurement results show an excellent mechanical and RF performance. Paper 3: Power-Handling Analysis of High-Power W-band All-Silicon MEMS Phase Shifters. The thermal characteristics and power handling capability analysis of a novel MEMS phase shifter concept are evaluated and compared to two conventional TTD and DMTL phase shifter concepts that utilize an air-suspended thin gold membrane. The novel concept was found to be the first MEMS phase shifter concept whose thermal power handling capability is not limited by the MEMS structures, but only by the transmission line itself and by the heat sink capabilities of the substrate. Paper 4: Microwave MEMS Devices Designed for Process Robustness and Operational Reliability. This paper provides an overview of novel MEMS phase shifters concepts, tunable microwave surfaces, reconfigurable leaky-wave antennas, multi-stable switches, and tunable capacitors, featuring innovative design concepts for overcoming performance bottlenecks in reliability and robustness to process variations of conventional RF MEMS devices. Device design, fabrication, and measurements are discussed in the context of comparing the design concepts with conventional MEMS device designs.

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Paper 5: Design Optimization of Millimeter-Wave MEMS Dielectric-Block Phase Shifters. This paper investigated optimization methods for achieving the best RF performance of a multi-stage millimeter-wave MEMS dielectric-block phase-shifter, with the distance between the stages as the main optimization parameter. The paper addresses the general problem of parameter optimization for multi-state transmission-line based RF MEMS devices. The optimization strategies were benchmarked for seven different optimization strategies by simulating and was confirmed by measurements of fabricated devices. It was found that a design with an asymmetric, non-equidistant distance distribution resulted in highest return loss performance improvement, while only slightly worsening the insertion loss. Paper 6: Three-Dimensional Micromachined Silicon-Substrate Integrated Millimeter-Wave Helical Antennas. This paper examined the concept of W-band substrate-integrated axial helical antennas, where the square helices are proposed to be 3D micromachined inside a high-resistivity silicon semiconductor substrate. Optimum designs, in terms of feasibility for microfabrication and radiation performance, were identified. Antenna feeding and matching, from a feeding coplanar waveguide, was successfully optimized. Finally, substrate-integrated line antenna arrays were investigated, for higher gain in the wafer plane. In conclusion, substrate-integrated, in-plane radiating axial helical antennas were found feasible for W-band applications, and could be realized by state-of-the-art standard micromaching processes in high-resistivity silicon substrate. Paper 7: RF Characterization of Gold-Coated Nickel-Wire Through Silicon Vias by Magnetic Self-Assembly.

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Paper Reprints 83

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Paper reprints

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Paper Reprints 84

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