NON-IMAGE GLIDESLOPE INVESTIGATIONThe analysis and design of a broadside glideslope antenna array...

Report No. FAA-ID-71 .. 48

NON-IMAGE GLIDESLOPE INVESTIGATION

Robert C. Voges Cllarles Liu

• RAFIC LIBR.ARY .~.

.267f: , .

February 1916

Documllnt is available to the public through the National Technical Information Service.

Springfield. Virginia 22151

Prepared for f· .

U. S. DEPARTMENT OF TRANSPORTATION

FEDERAL AVIATION ADMINISTRATION Systems Researcll & Development Service

Wasllington, D. C. 20591

>...

NOTICE

This document is disseminated under the. sponsorship of the Department of Transportation in the interest of information exchange. The United State Government assumes no li;lbility for its contents or use thereof.

!i· ~

Technical Report Documentation Page

3. Recipient's Catalog No.1. R_rtNo. 2. Go".""""" Ac-;ession No.

FAA-RD-76-48

t-,,.....,=""":------o-=-:---:----------'-----------------j-;c--;=--.---:::--~-----..--..... --..--...--- -.--... Title on" Su1>title 5. Report Data

February 1976 ' 6. Pe,fonning Orgonizotion Cod-;

N('I1-'m~{ge Glideslope Investigation

t-:=--:-"'--:-:-------------------------------1I. Performing Orgonizotion Report 'N-;:----7. Author' .)

l{<)l)o,'ll c:. Voges and Charles Liu

l'eX;ls Instruments Incorporated 13:,()() North Central Expressway P.O. Box 6015 Dallas, Texas 75222

12. Sponserino Aoency N_e _d Address

Department of Transportation Federal Aviation Administration

14. Spons~~i"g-A;;~;yCod-;--·-- .~--_._~---,--Systems Research & Development Service Washington. D.C. 20591 ARD-74J

r-:-:------~------------------------'----_._--..-.--..----~.- ..---- 15. Suppl__tary Notes

1'6. Al>o'rClct

The analysis and design of a broadside glides lope antenna array and its shielded loop elements are presented. TIlis non-image array is intended to replace the glideslope image array in regions where the latter does not perform well, such as in unlevel terrain or over tidal water. The synthesis of the array is presented with an analysis of its required height. The tower design and the design of a corner reOector behind the loops are given. The major effort was the design of the shielded loop, a single-turn loop of &tripline construction. Its impedance is matched with a series stub and a broadbanding circuit integral in the stripline feed. An integral monitor design is also described. The sensitivity of the loop impedance to rain and ice is investigated. as well as the performance of various radome approaches. An integral radome and a conventional type are discussed. Finally, a tluee-element array with a small distribution network is described and its performance presented. Full-sized distribution and monitor networks are discussed.

The feasibility of using the loop element in existing image arrays is demonstrated, and a detailed design of a coaxial construction with a single-rod reflector is given.

17. Key Wo,d.

Glideslope Non-imaging Loop element Shielded loop

18. Distributi_ Stot....ent

Document is available to the public through the National Technical Information Service, Springfield, Virginia 22151

19. Security CI...iI. (of "'i. r_t) 20. Socurity CI....if. (of thio p ..ge) 21. No. 01 P"lIes 22.. Price

UNCLASSIFIED UNCLASSIFIED 102

Fa.. DOT F 1700.7 (8-72) R......ducti_ 0' co....I...... pogo ...,tho.i"ee!

METRIC CONVERSION FACTORS

Approximate Conversions to Metric Measures '" -....

Approximate Conversions from Metric Maasures

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Symbel When Yeu Knew Multiply by Te Find Symbol - ;;

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LENGTH ~

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'" TEMPERATURE IUBCI) -

TEMPERATURE (exact) ... °c Celsius 9/5 (then Fahrenheit of

of Fahrenheit 5/9 lafter Celsius °c -- .. temperature add 32) tempet'ature

temperature subtracting temperature - - of 32l .. of 32 98.6 212

·1 in = 2.54Iexactlvl. For other exact conversions. and more detailed tables. see NBS Misc. Publ. 286. Units of Weights and Measures. Price $2.25. SO Catalog No. C1J.10:286.

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TABLE OF CONTENTS

Section Title Page

I INTRODUCTION · 1

II ARRAY DESIGN · 3

A. Aperture Synthesis · 3 B. Theoretical Array Performace .6

1. Error Analysis (Includes Mutual Coupling) .6 2. Array Focus . . . . . . . . 10 3. Site Grade Requirements 10

C. Distribution Network and Monitor 11 1. Network Design 12 2. Performance.... 18 3. Misalignment Monitor 26

D. Tower Design ..... 26 1. Tower Analysis 26 2. Tower Support and Tilt Adjustment 27 3. Tower/Array A-ssembly and Erection 30

III SHIELDED LOOP ARRAY ELEMENT DESIGN 37

A. Loop Theory 37 1. Small Loop 38 2. Shielded Loop 40"

3. Mutual Coupling 43 B.. Loop/Reflector Experimental Investigation 45

1. Loop Size . ; . 45 2. Reflector . , . 46 3. Loop Excitation 50 4. Summary ... 51

C. Loop Impedance Matching 52 1. Reflector' Effects 52 2. Series Stub. . . . . 54 3. Broadbanding 54

D. Investigation of Environmental Effects 59 1. Environmental Test Results 60 2. Performance Requirements 61

E. Loop Final Design and Performance 61 1. Element Design 66 2. Performance . . . . . 67

IV IMAGE LOOP INVESTIGATION 77

A. Single-Frequency Coaxial Loop 77 1. Experimental Investigation 77 2. Design and Performance 80

B. Stripline Loop . . . . . . 80 1. Design and Performance 80 2. Environmental Tests 91

C. Conclusions and Recommendations 93

iii

V FABRICATION PLAN 95

A. Reflector 95 B. Support Structure 95 C. Safety Climbing Equipment 98 D. Loop and Monitor . 100 E. Distribution Network .100

VI CONCLUSIONS .101

UST OF ILLUSTRATIONS

Figure Title Page

Glideslope Tower Offset From Runway Centerline Versus Tower Height .4

2 Number of Array Elements Versus Aperture Length for Various Spacings .4

3 Carrier Plus Sideband Radiation Pattern .7 4 Sideband Only Radiation Pattern .8 5 Array Excitation .9 6 Course Crossover Linearity for Constant Range and

Variable Elevation Angle 10 7 Carrier Plus Sideband Pattern With 2 Degrees and

0.2 dB rms Error 11 8 Sideband Only Pattern With 2 Degrees and

0.2 dB rms Error 12 9 Glideslope Location for 3-Degree Glideslope Angle and

55-Foot Threshold Crossing Height 13 10 COl for Nonfocused Array 14 11 COl for Focused Array 14 12 Array Focus 15 13 Monitor Design Alternatives 16 14 Monitor Probe in line Versus Antenna 16 15 Distribution Networks for Upper Half Array 19 16 Stripline Components 20 17 Phase Response of Schiffman Phase Shifter 21 18 Integral Monitor Performance 22 19 Distribution Network for TWA 23 20 Recombination Network for TWA 24 21 Phase Response of Model Phase Shifter 25 22 Directional Coupler Response 25 23 Non-Image Glideslope Antenna Array '. 28 24 Support Tower Base 29 25 Tower Modifications, Side View 30 26 Tower Modifications, Top View 31 27 Tilt Adjustment 32 28 Antenna Array, Top View 33 29 Tower Assembly, Side View 34 30 Tower Erection 35

iv

31 Electrical Design of Shielded Loop 37 32 Theoretical Loop Impedance 40 33 Evolution of the Loop 41

. 34 Balanced and Unbalanced Loops 43

37 E-Plane Patterns of a Loop in Right Angle Reflectors

72 Stripline Loop Antenna Circuit Including Matching Stubs

35 Parameter Definitions for Mutual Coupling 44 36 E-P1ane Patterns of Different Sized Loops 46

of Different Widths 47 38 Impedance of Loop in Reflector With Different Spacings 48 39 E-Plane Patterns of Loop in Different Reflectors~· 49 40 E-P1ane Patterns of Balanced and Unbalanced Fed Loops 50 41 Effect of Gap Width on Loop Impedance 51 42 Full-Sized Loop and Reflector 53 43 Sample of Scaled Loops 54

.' 44 Loop Impedance Matching Method 55 45 Shielded Loop Matching Techniques 56 46 Broadbanding With Transformer 57 47 Design Approach to Loop Broadbanding 58 48 Impedance of Broadbanded Loop 59 49 Impedance of Ice-Covered Loop . 60 50 Loop With Simulated Ice .. 62 51 Ice-Covered Radome of Loop Antenna 63 52 E-Plane Pattern of Loop and Iced Radome 64 53 Loop With Integrated Radome .. 65 54 Impedance of Ice-Loaded Integrated Radomes 66 55 Mechanical Design of Shielded Loop 67 56 Electrical Design of Stripline 68 57 Three-Element Loop Array 69 58 E·Plane Pattern of Loop in Array Environment 70 59 H~P1ane Pattern of Loop in Array Environment 71 60 E-Plane Pattern for Three-Element Array 72 61 H-Plane Pattern for Three-Element Array 73 62 Measured Mutual Coupling of Loops . 74 63 Mutual Coupling of Dipoles Over Ground Plane 75 64 Experimental Model of Image Loop 78 65 Image Loop With Three·Element Reflector 78 66 E-Plane Patterns for One- and Three-Reflector Elements 79 67 Design Detail of Image Loop 81 68 E·P1ane Pattern of Delivered Image Loop With Pole 82 69 H-P1ane Pattern of Delivered Image Loop 83 70 Impedance of Image Loop Element 84 71 One of Loops Delivered 85

and Integral Monitor 86 73 E-Plane Pattern for Stripline Loop Antenna 87 74 H-Plane Pattern for Stripline Loop Antenna 88 75 Impedance of Loop Under Normal Test Conditions 89 76 Electrical Performance of Integral Monitor 90 77 Cross Section of Antenna, Radome, and Support 91 78 Antenna and Supporting Brackets 92 79 Stripline Loop Antenna as Delivered 92 80 Stripline Loop Antenna With ~-Inch Thick Simulated Ice 93

v

The shielded loop radiating element was also considered for use in an image array, Le., null reference, sideband reference, or capture effect glideslope system. The objectives for the image loop design were low mass and low wind resistance so that the support tower could be simple and frangible. These requirements necessitated the consideration of a small reflector to minimize the element frontal area and mass. The image element design resulted in an element similar to the non-image loop but with a single rod reflector. The final image loop designed and delivered to the FAA had a single reflector and was fabricated with provisions for attachment to a circular pole.

In the course of this investigation, it has been shown that this shielded loop array possesses important characteristics. First, the mutual coupling was expected to be low and was found to be a factor of 2 lower for this loop design than for dipoles over a ground plane. Also, this shielded loop design was used to reduce the stray capacitance of environmental effects lower than might otherwise be expected. In addition, the stripline approach to element fabrication is known for its reliability and is expected to be low cost in volume production.

The mechanical design making use of spaced thin rods reduces the wind loading on the tower. A solid panel with dipole elements would produce about 30 percent more torque at the tower base. This fact is expected. to reduce the tower cost by about 40 percent. Field service was "

also considered in this design. The networks will not be embedded in molded foam and are not ';'

required to be placed on the tower. It would be possible, by adding more cable, to place the networks in the shelter. All itroubleshooting could then be done in one convenient location on the ground rather than over :the length of the tower. Even with the shortest cables, the circuits are in two accessible places Oin the tower.

The following sections of this report discuss in detail the array design, array element design, and image array element design accomplished during this investigation. In addition, a fabrication plan is included to describe the fabrication of a non-image glideslope antenna array.

2

SECTION II

ARRAY DESIGN

The objective of this program was to investigate the design of a broadside array of sufficient length to minimize the influence of the ground plane, Le., an antenna which does not rely upon the ground image to achieve the required glide path guidance. This antenna array must give Category II level guidance for sites where the ground height is variable (tidal waters) or where the terrain in front of the antenna is irregular (valleys, hills, etc.). To achieve this objective, the carrier plus sideband (CSB) and sideband only (SBO) patterns of the basic antenna (free space) must have rapid rolloff at the horizon and low elevation sidelobe levels below the horizon. The pattern rolloff at the ground is achieved with a sufficiently large aperture, whereas the sidelobe level can be synthesized with a particular amplitude and phase distribution.

, As with any physical problem in optimization, constraints must be considered. For instance,

the aperture length is bounded by the airport environment and the minimum sidelobe levels are constrained by whether the aperture distribution is physically realizable. The relationship for determining the offset of the array from the runway centerline is illustrated in Figure I. For sites where the ground elevation decreases away from the runway, the array may be placed closer to the runway. The physical realization of the synthesized aperture distribution must also be considered through an error analysis. In addition, the number of array elements or array spacing must be considered. For the aperture lengths between 18 and 23 wavelengths, the relation between the number of elements, array spacing, and aperture height are depicted in Figure 2. The following sections contain a discussion of Texas Instruments approach to the synthesis problem and the results.

A. APERTURE SYNTHESIS

Two techniques have been used to synthesize the array aperture. The Schellkunoff polynomial or associated polynomial l was found to give rapid results on the computer, but fine-tuning the resulting distribution· was tedious. A second technique, using a constrained optimization method developed at Texas Instruments, was used to fine-tune the distribution.

The Schellkunoff polynomial method is a technique where the zeros of an array radiation pattern are described in terms of the array element excitations: The polynomial associated with an N-element linear array may be described as

N -1

(I)f(2) = L gn zn

n=:O

where the array phase and amplitude excitation are given by gn and the complex variable Z is given by kod sin () for which ko is the free space wave number and d is the array spacing. Equation (I) may also be expressed as the product

(2)

'R. Collin and F. Zucker, Antenna Theory, Part 1, McGraw-Hill Book Company, (New York, 1909), p. 158.

3

100

t-G.... \.LI :c 0: \.LI:= ~ \.LI D

3 (/) \.LI C-.J

80

60

40

20

./

.//

f---7 V1---- ---- ---

7/ V

1-00___ 1----I

// 23A

V -

18A

Cl

o 200 300 400 500 600 700 800

Gl-IDESl-OPE OFFSET FROM RUNWAY (FEET)185979

Figure 1. Glideslope Tower Offset From Runway Centerline Versus Tower Height

80

70

~ \.LI \.LI lL 60 'OJ

:r l-t!)

·z \.LI 50.J \.LI 0: ::> I0: 40\.LI Doe(

30

20 16 20 24 28 32 36 40

NUMBER OF ARRAY ELEMENTS185980

Figure 2. Number of Array Elements Versus Aperture Length for Various Spacings

4

where Z through ZN are the N zeros of the associated polynomial. Defining the N desired zeros and multiplying out Equation (2) gives a power series for which the coefficients are the element excitations described by Equation (1). Equation (2) can be easily solved and the array excitation extracted with a computer. Included with the computer technique to solve Equation (2) was a routine developed to define, analytically, the zero locations so that prescribed sidelobe levels could be maintained near broadside.

The polynomial method of array synthesis proved efficient to use on a computer, but the array pattern could not be completely constrained. Hence, a computer optimization technique was used to fine-tune the aperture distribution while constraining the important parameters. The optimizing procedure used was a penalty function Il?-ethod based on a relatively new viewpoint regarding the significance of Lagrange multipliers in nonlinear programming. In the Generalized Lagrange Multiplier (GLM) method, a value is assigned to the multiplier d~fined to be associated with each constraint. A Lagrangian penalty function is formed by adjoining the constraints to the objective function with the corresponding multipliers, and the Lagrangian function is optimized over the set of decision variables. In general,. the solution obtained is neither feasible nor optimal in the constrained mathematical program; but it is an optimal solution to a mathematical program for which the constraints are satisfied identically (at the obtained levels). Solution of the desired problem generally requires a convergent algorithm for modifying the multiplier values, iteratively, until the constraint functions (evaluated at the solution) are satisfied at the desired levels. The global optimization strategy employed in the synthesis procedure treats the question of local optima as a hypothesis testing problem; the procedure developed is based on the "sequential-testing" concept of Wald. 2

Subsequently, the glideslope antenna synthesis problem was cast as an efficient computer program with the optimization routine GLAMOR (for a Generalized Lagrange Multiplier Optimization Routine) using the techniques described above. The GLAMOR optimization program has been extremely successful in synthesizing shaped, doubly and singly curved reflectors with multiple feeds.

Generally, the pattern characteristics required to give Category II performance for most sites have been defined by Westinghouse, both analytically and experimentally, at LaGuardia and Lynchburg, Virginia.3 The conclusion derived from the Westinghouse report was that the sideband only radiation pattern is of greater importance than the carrier plus sideband radiation pattern in that reflections have a greater influence upon the SBO null than the CSB peak. The measurements by Westinghouse at LaGuardia and Lynchburg, Virginia, for a tidal site and a severe upgrade site, respectively, have shown that the SBO peak sidelobes below the horizon to -12 degrees or lower should be less than -35 dB from the peak SBO level, and the rolloff of the SBO pattern at the horizon should be 10 dB or more below the level at the SBO lower course edge. These characteristics were used in the synthesis as guidelines for usable pattern levels.

Since the SBO pattern was considered more critical than the CSB pattern, the initial synthesis· was directed toward 20-, 21-, and 22-wavelength (A) apertures to generate the SBO pattern. The SBO pattern is, in general, a difference pattern with a null in the glide path

"A..Wald, sequentiill Analysis, John Wiley & Sons (New York,1947)..

3 R A. Moore, et ai., "Analysis of Instrument Landing System Glide Slope Broadside Antennas," Report No. FAA-RD-72-139, (August 1972).

5

direction. The array was synthesized with a broadside null so that a planar guidance null could be achieved as. opposed to a conical null for a scanned array. In practice, the array will be tilted mechanically to give the required glide path angle. For analysis purposes, the glide path was assumed to be 3 degrees; hence, the horizon would be -3 degrees from broadside. The objective of the synthesis was to lower the radiation levels between -3 and -15 degrees or more from broadside using the Schellkunoff polynomial. The SBO array distribution was synthesized for 20, 21, and 22A apertures and the 21A aperture was chosen as the optimum. The 20A aperture seemed adequate, but more margin in sidelobe level was considered necessary. The added performance given by a 22A aperture did not seem worthwhile for the added height and resulting added offset from the runway centerline. The CSB pattern was synthesized for the 21 A aperture with the same requirements as the SBO pattern except that the CSB pattern was a sum type pattern with peak radiation at broadside. The peak radiation did not occur precisely at broadside in that a small amount of scan was necessary to minimize radiation at -3 degrees.

The results of the synthesis are illustrated in Figures 3 and 4 for the CSB and SBO patterns, respectively. Note that the sidelobe levels Qelow the broadside direction are at least 48 dB below the pattern peaks out to -30 degrees. The pattern values normalized to the lower path edge are given in Table 1. The glide path angle is 3 degrees (same as 0 degrees in Figures 3 and 4), and odegrees is the horizon. Note also that the level at the horizon is -15 dB, which is less than the desired -10 dB, thus allowing a greater margin for an up-slope site.

In addition to the aperture length, the number of array elements was determined. Figure 2 shows the relation between aperture length, number of array elements, and element spacing. The number of array elements was chosen on the basis of minimum number of elements to maintain flexibility for array synthesis. In addition, the number of elements was constrained to be even. The number of array elements was chosen to be 26 with an approximate spacing of 0.8 wavelength. Figure 5 illustrates the excitations for the array in amplitude and phase for both CSB and SBO. Note that no rapid variations in either amplitude or phase are evident, making physical realization practical. The crossover linearity is illustrated in Figure 6 for a 3-degree glide path using the excitation given by Figure 5.

B. THEORETICAL ARRAY PERFORMANCE

The following is a discussion of the performance of the non-image glideslope antenna for the synthesized patterns discussed in the previous section. Factors influencing either the physical realizability of the synthesized patterns or the array performance may be fabrication errors, array

.focusing, mutual coupling, and site grading.

1. Error Analysis (Includes Mutual Coupling)

As is usually the case for array antennas for which very low sidelobes are synthesized, I.e., less than -30 dB, phase and amplitude errors introduced in fabrication and assembly are very important. The non-image array is not immune to such considerations. Figures 7 and 8 illustrate patterns computed with 2 degrees and 0.2 dB 1-0 errors on the array distribution for the CSB and SBO arrays, respectively. For manufacturing purposes, this error corresponds to a maximum (3-0) error of 6 degrees and 0.6 dB. The patterns given in Figures 7 and 8 are considered adequate down to -12 degrees below broadside, but care in distribution network fabrication and assembly is required.

6

o

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0( " '" ~ -30 ~ J

f\'"II:

f\ f\

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1\ f-0+-50 --+-'-+-40

I -30

t -20 -10" o 10 20 30 40

185981 ELEVATION ANGLE (DEGREES)

Figure 3. Carrier Plus Sideband Radiation Pattern

Another source of error which will influence the array perfonnance is mutual coupling. The shielded loop radiator was chosen for its low mutual coupling. To -demonstrate the synthesized pattemunder mutual coupling conditions, a computer program developed by Texas Instruments was ·used. The program gives the far-field pattern of an array of dipoles including mutual coupling. Dipoles were used in the program because the coupling is well known and well documented and the loop elements developed under this program were shown to have lower mutual coupling than the dipole. Hence, the dipole coupling was considered a worst case. The results for a 26-element dipole array spaced 0.808;\ were a 2-dB increase in peak sidelobe level in the 0- to -12-degree region and a negligible change in the level at the horizon. These results are satisfactory, and the loop array should perform even better,'

7

o

1\ f\

-10

-20

'iii' e 111 c ::l !:: ..J Do ~ «

-30111 > !( ..J 111 II::

f\

-40

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- ... 40


Figure 4. Sideband Only Radiation Pattern

TABLE 1. ARRAY PATTERN VALUES (normalized to lower path edge)

Max S.L.L. 3.7° 2.3° 0° _1.5° _3° _6° _9° _12° oto _120

SBO(dB) +7.0 0 -15.7 -27.9 -24.0 -28.0 -35.4 -42 -24.0

CSB(dB) +6.3 0 -29.5 -48.5 -42.1 -70.6 -50.1 -55.6 ~0.8

8

CSB1 .0

0.8

..... ~ 0.6

.......

o o 2 4 6 8 10 12 14 18 20 22 24 26

ELEMENT NUMBER

360

180 ..... III 1&1 1&1 It: Cl 1&1S. 0

1&1 III <:r a.

-180

-360 L..-_.....l..__-'--_---L__....L.._---l__....L-__L...-_-'-__.l------L.-_-'--_.....l.._----'

o 2 4 6 8 10 12 14 16 18 20 22 24 26

ELEMENT NUMBER 185983

Figure 5. Array Excitation

9

50

150 ,...------------...

3-DEGREE GL.1DE PATH

...... < 100 :l 'oj

z 2 I

~ Q Z

~ 0 1------------------'\---------------- I< ~ Q -50 \1/til It: :J

8 -100

-150 o 1 .0 2.0 5.0 6.0


Figure 6. Course Crossover Linearity for Constant Range and Variable Elevation Angle

2. Array Focus

The non-image array is defocused at threshold because of its size and location. The distance to threshold, as shown in Figure 9, is less than 700 feet which is within the array near field; therefore, the array is defocused. This defocusing manifests itself in a strong fly-down signal near threshold as illustrated in Figure 10.. This phenomenon can be compensated for by focusing the array transversely, Le., in a direction normal to the runway centerline. A constant angle glide path will be achieved, as illustrated in Figure 11, if the· array elements are placed on an arc described by a constant length with the center above the runway centerline and at the array phase center height, as shown in Figure 12. If a fly-up command is desired, the focus can be exaggerated by decreasing the focus radius. This transverse focus does not influence the remainder of the glide path (Figure 12) for longer ranges. The focus can be implemented by using a piecewise approximation to the arc curvature; Le., groups of four elements, each one tilted to approximate the arc. Over the four-element subarray, the error introduced in the arc approximation is less than 0.1 inch, and the maximum offset at the top and bottom of the array is approximately 9.6 inches for a 600-foot runway offset.

3. Site Grade Requirements

The array patterns shown in Figures 3 and 4 require a smooth ground plane extent of less than 200 feet. This smooth ground plane will be required in a semicircle about the broadside direction of the array with emphasis on the sector from broadside to the direction normal to the

10

o

ELEVATION ANGLE (DEGREES)185985

Figure 7. Carrier Plus Sideband Pattern With 2 Degrees and 0.2 dB nos Error

403010o-10-20-30-40

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20

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~ -20 m e "I 0 :::l t .J Il. ~ < 1JJ > -30

~ .J 1JJ a:

-10

runway centerline (as illustrated in Figure 9) for a typical location where the tower base is at the same elevation as the runway.

c. DISTRIBUTION NETWORK AND MONITOR

The distribution network is an RF circuit, located on or near the antenna tower, which divides the two transmitter signals, CSB and SBO, into the appropriate 26 components,one for each element. The recombination network does the opposite, processing the 26 signals from the inte8ral monitors into the desired monitor signals.

11

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~ ~

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1\

f\

f\ 1&1 II:

f\

1\ -40 A f\~N'1~ rJ

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-so -40 -30 -20 -10 o 10 20 30 40


Figure 8. Sideband Only Pattern With 2 Degrees and 0.2 dB nns Error

1. Network Design

The approach is to use stripline for the antenna elements and integral monitors and to use coaxial cables to connect the processing networks. These networks will be divided into upper and lower halves and will be contained in two boxes mounted about ¥.J and % the distance up the array (23 feet and 54 feet from the ground)~

In one box, there will be three distribution circuit boards and three recombination circuits. Feeding the antenna will be one power dividing board each for CSB and SBO as well as a summation board consisting of 13 separate hybrid couplers to add the two signals for each element. A duplicate board will be needed to evenly sum the 13 monitor signals for input to the

12

o 500

THRESHOLD

500

23 A GLIDESLOPE

1000

X IMAGE GLIDESLOPE

RUNWAY

185987

Figure 9. Glideslope Location for 3-Degree Glideslope Angle and 55-Foot Threshold Crossing Height

13-way recombination circuits discussed below. All circuits except the CSB and SBO networks can be duplicated in the other box. To combine the boxes, there will be five hybrids, two for feeding and three for the monitor signals.

a. Integral Monitor

The monitor will be a 20-dB stripline edge coupler incorporated in the loop stem. It will be 'A/4 long for high directivity and positioned on the generator side of all matching components to avoid standing waves. This coupler is directional, and the loop will have a connector for each direction as shown in Figure 13(A). The forward direction will normally be monitored for course

13

150

FLY UP 100

so

..... 0(

3 o Q o

-50

3.0-DEGREE GLIDE; PATH. NON-FOCUSED ARRAY

2.64°

FLY -100 DOWN

185988

150

~~Y 1 0.0

50

o Q o

-50

FLY -100 DOWN

o Q 5.0 10.0 15.0 20.0 .J o X

RANGE FROM GPIP (FEET X 1000)

Ul III 0:

~ Figure 10. COl for Nonfocused Array

3-'DEGREE GLIDE PATH

~ ..::...:..::::..__F_O_C_U_S_E_D_A_R_R_A_Y__ - 2,640f"---

L 3.00 __---

-150 L.........I...------JL.....--------L......,..O----......-----......------'--------' 0

Q 10, 15.0 20.0 25.0 30,

.J o RANGE FROM GPIP (FEET X 1000)

X Ul

185989 III 0: X I-

Figure 11. COl for Focused Array

14

UNFOCUSED ARRAY

+-ARRAY PHASE CENTER

/ ..../"

--------~

~ --

185990

Figure 12. Array Focus

width and boresight in lieu of a far-field monitor, while the reflection port will be monitored for element fault detection.

Two alternative designs have been investigated and their merits considered. A probe antenna in the loop, as shown in Figure 13(C), is the most obvious approach. Another possibility would be a probe in the line leading to the antenna which would monitor voltage like a slotted line probe. This would also be the signal formed by summing the two ports of a bidirectional coupler with a Wilkenson Tee. Figure 13(B) is such a line probe.

An important performance parameter would be the sensitivity of these signals to various deviations which might occur in antenna characteristics. For instance, consider Zs and Zp in Figure 14, a circuit representation of the antenna. To the right of the "feedpoint" reference plane is the antenna with impedance, Za' the undesired shunt impedance, Zp, and the series impedance, Zs' which is nominally the matching stub. Normally, Zp = 00 and Zs = Zo - Za' in which case the voltage at the reference is 1. Then:

1 1=-· =1Z n

o

v = eikd = V (3)n

15

(A) TWO-PORT MONITOR

t (B) LINEVQLTAGE PROBE

t (c) LOOP CURR ENT PROBE

185991

Figure 13. Monitor Design Alternatives

EQUIVALENT CIRCUIT OF ANTENNA

I- d---" Zo LINE

REFERENCEV

NORMAL OPERATING CONDITIONS ARE Zp = 00

185992 Z + Z .. Z SaO

Figure 14. Monitor Probe in Une Versus Antenna

16

FEED POINT

The sensitivities will be defined as the derivatives of relative V and I as a function of relative impedance. The relative parameters are:

VV =

f V n

I 1=

f I n

(4)

Table 2 shows that the sensitivity to error in Zs or Za is the same for either probe with the exception of phase. However, when Yp = l/Zp varies from zero, the antenna probe is more sensitive than the line probe. Fault detection, or· response to catastrophic failure, is similar for both cases except that the line probe signal can be doubled as well as nulled depending on the choice of d, the probe position.

TABLE 2. RELATIVE CHANGE IN I AND V FOR SMALL CHANGES IN ZsAND Zp

Line Probe Antenna Probe

al _f = -1/2aZSf

av Z __f =_~ e-j2kdZp finite aVpf 2

2jkdLimit vf

= 1 + e

(Derivatives evaluated for normal operating conditions)

The design approach for the antenna probe case would be to etch a line along the inside edge of the loop between the stem and the series stub, which would couple with the radiating currents on the loop. This is probably the second best method, although both these alternatives have the advantage of requiring only one monitor cable per element. The choice of design was basically for reliability in detecting faults and could readily be changed on the basis of a cost tradeoff. The higher cost of the proposed design would be for 26 more cables with. their connectors, two more stripline circuit boards, and perhaps an increase in alann circuitry.

17

b. Distribution Network

The distribution network synthesis was done with emphasis on keeping the lines short because, at this frequency, size is an important constraint. The results are shown in Figure 15. The three stripline boards indicated will each be about 30 by 30 by 0.5 inches. Branchline couplers and Schiffman phase shifters were also chosen with this constraint in mind. These components are more amenable to hardware adjustments than alternatives, which is an important feature given the tight tolerances on array excitation. The components sketched in Figure 16 are both adjusted by changing the impedance of the lines, which can be done by inserting a thin piece of dielectric between the laminations. Twenty-four such couplers will be used for each mode, CSB and SBO, as well as 28 hybrid rings and 26 phase shifters. These components have been fabricated and tested at Texas Instruments for existing instrument landing systems. Phase adjustments have been made on the Schiffman devices by trimming the line back and by adding dielettric tape. Figure 17 indicates how varying the coupling varies the slope of the phase curve at the design frequency. The slope corresponds to a compensating line length response but is offset by as much as 90 degrees. The branchline coupler is also designed so that a piece of tape .,

,can be placed over. the branch lines (wide) while leaving the main lines unaffected.

c. Recombination Network

The recombination network for fault detection will be contained on one board and will look similar to the SBO network. It was synthesized so that each element is equally weighted for the sum of CSB and SBO and fault detection sensitivity will be equivalent for all elements.

The recombination for the coarse monitors is simple since uniform signal addition is desired. An "N-way" power divider is appropriate for the CSB monitoring function. The sensitivity monitor is required to simulate the far field off boresight which requires a linear phase progression along the elements. Thus, this monitor is also an N-way divider with the addition of delay lines to simulate the different distances from the elements to the required point in space. These networks will be contained on two boards, one with the hybrids and one with the two N-way dividers.

2. Perforlllance

The performance of the edge coupler monitor has been measured on a fabricated loop and is illustrated in Figure 18. This 17. 5-dB edge coupler can be seen to have very high directivity since the forward monitor port is unaffected by the return, signaL Directivity was measured at about 35 dB, but it may be better since the reverse signal in the main line was close to -35 dB.

Texas Instruments has had considerable experience designing and fabricating stripline networks. For instance, the ILS localizer, which has 14 traveling wave elements, requires very similar networks built by Texas Instruments. The schematic of the distribution network shown in Figure 19 indicates that both edge couplers and branch-line couplers and included. Similarly, one of the monitor networks which contains the N-way power dividers is shown in Figure 20. Over 60 of these circuits are now in service.

18

CSB BOARD r BOARD SBO BOARD

CSB IN SBO IN NETWORK FOR LOWER HALF

IS SYMMETRICAL.

B = BRANCH LINE COUPLER

H =HYBRID

P =PHASE SHIFTER

N = ELEMENT TERMINAL

185993

Figure 1s. Distribution Networks for Upper Half Array

19

BRANCH LINE COUPLER

A/4

•. PHASE SH I FTER

185994

Figure 16. Stripline Components

VSWRs are typically limited to 1.15 and tolerances to ±0.3 dB and ±3 degrees from beginning of distribution network to beginning of element cables; These error limits are applied to two circuit boards and their interconnecting cables much like the non-image system will be. Since frequency and tolerance requirements are different, it cannot be reliably determined whether Texas Instruments networks can be made to meet the specifications over the whole band without actually building a complete feed system and testing it. If riot, the circuits would have to be tuned to their operating frequency.

20

15

10 ,COUPLING LEVEL

3 4· 5 6 7 8 9 10

1 FREQUENCY (GHZ) 185995

Figure 17. Phase Response of Schiffman Phase Shifter

In order to discuss the acceptability of measured data, one must have an idea of performance requirements. Beginning with the pattern analysis r~sults that a random error with a standard deviation, a, less than 0.2 dB and 2 degrees is required, about a third of the error is allocated to each, stripline board, cable, and element. Noting that at a given frequency these three sources of error are independent, they are combined by adding their variances, a2 • This means that the error through the circuit board can have a a6 as high as aT/vl3 where subscript T stands for total. This ac = (0.12 dB, 1.2 degrees) corresponds to the TWA network specification, so the common usage that tolerance is equal to ±3a provides a circuit board tolerance of ±0.36 dB and ±3.6 degrees that is higher than for the TWA network.

Some components fabricated by Texas Instruments were tested for the non-image project. Figures 21 and 22 show the phase response of a Schiffman phase shifter designed for 3 GHz. Notice that it has been adjusted for precisely 45 degrees, that it is a very broadband device, and that its "random error" is less than ±l degree. Also appearing is the amplitude measurement of a hybrid type coupler. There is a wide band which has amplitude variations less than ±O.l dB. This brings up the question of component tolerances. Referring to Figure 15, a maximum of six stripline components are in series in the network. Again using independence:

(5)

then· an = 0.12/vl6 = 0.05 dB, which makes the tolerances ±0.15 dB and 1.5 degrees for each component. These error requirements appear to be reasonable based on this cursory analysis, but testing of the completed hardware is ultimately required.

21

FREQUENCY (MHZ)

226 228 230 232 234 236 238

...... lD a 0: III ~

~

BAND .1 I I I I I I I I I I I

REFLECTION MONITOR I + 20 DB

FORWARD MONITOR

~

185996

Figure 18. Integral Monitor Perfonnance

22

CSB ~;>:Jir----;:::===~2~I~D~B==~=L ~

16 DB

21DB

~ 51 ';'

'- ~ J3 1 W

--+------------J-2-5--4~

.-------------1 51 ':' '----4J4 lW

..~2 1850 lOW

~-------....'--.L---------J-2-l;5~~ 50 ':' 2W

'----4.15

'------+---------------~J7

.----------------- __~J8

3.5 DB

3 ~J-,-9---../--'------------------4J9 15 DB50 ..... ..",.-

25W~ ~

19 DBC ..""..

SBO ~ ......2 ~ ,

1./

30

\..

~20 '1,./:}

l~i/1 ~.

./ .....5 :J21

50 c::: W ~ ,4 :,

0

"

6 J22 :SO 25W ' ~ 13D~1

.\.. .....7 :J23

50 lOW ~

185997

Figure 19. Distribution Network for TWA

23

" J27/.~ 51 ':' lWI ,

Jl0/

\..' J28/.~

50 ':'I , 25W J 1 1 / , JI2/ , J..1 3

~ J14

~ JI57 \..

J167

-~----------------------

W67P7 ELEMENT 1 LEFT







J7

-7>

J6

-7>

J5

-7>

J4

-7>

J3

-7>

J2

-7>

J1

-7>

R53

~7A J21 r-----'. I

<~I R54 50 - I 47 - IW1/4W L _____ .J,U I

R55 47 1/4W

R56 47 1/4W

R57 47 ,/4W

.f"4,'='

J20 r - - - -""1 <~I

50 I 1W I- 1- ____ ...1

~ J19 r - - -- -,

Il..t<~I R58 .:. II 50 '=' II1 W471/4W L ___ ...J

R59 47 ~ 1/4W

JI8 r - - - --, .. <~I,V IR60 _ R18_1- I50

47 - I I W 1/4W L. ____ J

R61 47 ~ 1/4W

J17 r - -- -.., <~I

R62 ,l..t I .,7 I50 -47 .:. I 1 W· I 1/4W L ___ -.J

R63 47 ~ 1/4W

JI6 r----..,

rU<~I R64 I R 1 6 '=' I 47 .:. I ~~ I '/4W L ____J

R65 47 ~ 1/4W

J15 r - - - -1

<~I I R15,=, I

R66 50 IrU I47 - 1W1/4W L ____ -J

I

R2 R33 ~~~ RX~31

R32 R35

R4 R2 ..... -~ /R16

R7/:

.R5 ='" RIR3

R18 16

R19 16

.

,

J34 W4 P34

~

J35 W3P 35

J R17 16

J29 W1P29

/

ON COUR SE OUTPUT

185998

Figure 20. Recombination Network for. TWA

24

40e 1--------....;.------------------------------a:: ClJ III. a.....

OL

2.0

-..L

2.5

- __....

3.0

--l-,-....

3.S

.........

4.0

.. FREQUENCY (GHZ)

185999

Figure 21. Phase Response of Model Phase Shifter

5 Q. Z-

,.. lD8

o

-,...5

.. -

~ III > § IIIa:: a:: III ~

2

-15

1 .2

"'~9 L.-__~i--___J,

1 • 1 1 .5

._&._-_--'

186000 FREQUENCY (GH~)

Figure 22. Directional Coupler Response

2S

3. Misalignment Monitor

. A misalignment detector will be required to detect tilt of the array tower in any direction away from the correct tilt. An alarm output is generated if the tilted condition remains for great~r than about 2 minutes. This monitor will be located at the top of the tower to measure any possible deflections caused by bending of the tower as well as tilt.

Texas Instruments has a suitable misalignment detector in production and in the field so that redesign and replanning fabrication are not necessary. The transducer is an electrolytic tube with a bubble and curvature much like a carpenter's level. It has three electrodes extending into the resistive electrolyte with an appropriate bridge circuit to sense any movement of the electrolyte caused by tower tilt. There is a transducer for each direction, and the processing circuit combines their outputs so that a circular tilt limit pattern is effected.· The circuit also produces the time lag before switching the alarm relay. A thermostatically controlled heating resistor is required to maintain a constant temperature for the transducer and circuits.

D. TOWER DESIGN

1. Tower Analysis

A finite element analysis of the antenna tower assembly was performed using MRI/STARDYNE. STARDYNE is a static and dynamic structural analysis system of computer programs which analyze linear elastic structural models. STARDYNE has been in use since 1968 and has been used extensively at Texas Instruments to predict the structural integrity of antenna arrays with excellent results.

A preliminary worst case finite dement model was generated to determine the required support tower size and the required truss size and configuration. A 75-foot support tower with a 66-foot corner reflector was modeled. The model consisted of 275-nodal points connected by 739 uniaxial pipe elements with tension-compression, torsion, and bending capabilities. The element has 6 degrees of freedom at each node: X, Y, and Z translation and rotation. The material properties for the tower. elements were: modulus of elasticity (E) = 29 X 106

pounds/inch2 , Poisson's ratio (v) = 0.292, and mass density (U) = 7.32 X 10-4 (pound-second2 )

inch4 • The structural properties of the corner reflector facing were neglected for this model. The loading conditions considered for the antenna tower assembly were

Static load caused by member weight (RF cable runs and the RF distribution unit)

Static load caused by an ice accumulation of 0.5-inch radial thickness of clear ice

Aerodynamic loads caused by 100-mph wind with ice accumulation.

In the iced condition, the corner reflector was treated as a solid panel. These loading conditions were analyzed to determine their cumulative effects, resulting in the initial selection of a triangular support tower with a cross-section size of 42.2 inches on leg centers.

After the array design was finalized with a 26-element array and 0.808-wavelength spacing, the finite element model of the antenna tower assembly was revised. The final model simulated a 7Q:-foot support tower with a 62.2~footcorner reflector. The corner reflector was changed from a solid panel in the iced condition to a thin rod reflector consisting of 0.25-inch aluminum tubing

26

spaced on 1.91-inch centers. This produced a 29.7-percent reduction in the frontal area of the iced comer reflector and resulted in a significant decrease in the aerodynamic load as shown.

Solid Reflector

Co P y2 (2.0)(0.075)(146.67)2P = ---'-- = = 50.11 pounds/foot2 (6)

2g 2(32.2)

F = PA = (50.11)(156.97) = 7,865.9 pounds (force) (7)

Thin-Tube Reflector '::'

CD P y2 (1.2)(0.075)(146.67)2P= = -------- = 29.79 pounds/fooe (8)

2g 2(32.2)

F = PA = (29.79)(156.97) =4,676.1 pounds (force) (9)

This permitted the support tower cross-section size to be reduced from 42.2 inches on leg centers to 36.0 inches. A static analysis of the final configuration of the antenna tower assembly was performed with the following results:

Pressure load equivalent to 100-mph wind on an iced reflector

Maximum translation 0.87 inch

Maximum rotation 0.07 degree

Maximum principal stress 14,700 psi

Safety factor for this loading 3.1 *

2. Tower Support and Tilt Adjustment

The support tower will be supported by a single truss structure and stabilized with two guy cables as shown in Figure 23. Both the support tower and the truss structure designs have been based on commercially available sectional triangular .towers. The support tower has a cross-section size of 36.0 inches on leg centers, and the truss structure has a cross-section size of 24.3 inches on leg centers. The base of the support tower will be modified to .include a pinned joint as shown in Figure 24. The third section of the support tower and the upper end of the truss structure will also be modified to include a pinned joint as shown in Figures 25 and 26. The modifications will simplify field assembly and permit the antenna assembly to be tilted. A manually adjusted worm gear actuator will be added to the base of the truss structure to change the ant~nna tilt (glide angle) over a minimum range of 2 to 4 degrees. This allows the antenna tilt to be changed by an individual at ground level by manually cranking the actuator to the new setting. This adjustable linkage at the base of the single truss is shown in Figure 27.

*Factor of safety is defined as yield stress/maximum stress.

27

CLEARANCE LIGHT

MISALIGNMENT DETECTOR

SUPPORT TOWER THIN TUBE CORNER REFLECTOR

SINGLE ELEMENT RADOME

SINGLE TRUSS STRUCTURE

186001

Figure 23. Non-Image Glideslope Antenna Array

28

c

39.00

186002

PINNED JOINT

CONCRETE FOOTING

GROUND LEVEL

Figure 24. Support Tower Base

29

-----------------

SUPPORT TOWER

PINNED JOINT

SINGL.E TRUSS STRUCTURE

Figure 25. Tower Modifications, Side View

186003

3. Tower/Array Assembly and Erection

The main support tower will be assembled on its side on the ground. The tower sections will be bolted together using the vendor-supplied hardware. Then the safety climbing equipment will be installed. The reflector section will be clamped to the tower cross braces on the front face of the tower in a slight arc. The radius of this arc will equal the distance at the site between the antenna array and the runway centerline. The cable trough will then be mounted to the back side of the tower face in line with the center of the reflector. The stripline loops and the radomes will be installed in the reflector support brackets. The stems of the loops will extend into the cable trough to protect the connectors and cables from inclement weather as shown in Figure 28. The two distribution units also mount to the back side of the tower face (Figure 29).

30

186004

Figure 26. Tower Modifications, Top View

To minImIZe cable lengths, one unit is mounted 24 feet from the tower base, and the second distribution unit is mounted 55 feet from the tower base. After the distribution units are mounted, the monitor and signal cables are installed. The antenna array is then ready to be tested.

Once the testing has been completed, the monitor and signal cables must be removed. The support tower sections are separated and the hardware saved for use at the installation site. It is not necessary to dismantle the array sections from the tower sections for shipping. The four tower sections will be numbered, crated, and shipped to the installation site. At the installation site, the tower sections will be reassembled and the monitor and signal cables reinstalled. Then the obstruction lights and the misalignment detector will be ·added and wired. The base of the tower is then pinned to the concrete footing, and the two guy cables are attached to the tower. While the antenna array is still on its back on the ground, an impedance check of the array will be made to verify that it is operational. The antenna will then be erected with a single large crane as shown in Figure 30. The support tower will be erected with the crane's main hoist cable. Then the single truss structure will be raised with the crane's jib cable. and the upper end of the truss structure attached to the support tower. The lower end of the truss structure and the two guy cables will be fastened to their respective concrete footings. Once the guy cables have been tightened, the crane can be disconnected and the antenna tilt adjusted.

31

186005

t \

'\

24-INCH SUPPORT TOWER

10-TON WORM GEAR ACTUATOR

GROUND LEVEL

CONCRETE FOOTING

Figure 27. Tnt Adjustment

32

CORNER REFLECTOR

. STRIPLINg LOOP

SU PPORT TOWER

MONITOR CABLes SI(;NAL CABLES

186006

Figure 28. Antenna Array, Top View

33

70 FT

55 FT

24 FT

186007

DISTRIBUTION UNIT

DISTRIBUTION UNIT

Figure 29. Tower Assembly, Side View

34

JIB CABLE

186008 ...1-_---' --.1.....

Figure 30. Tower Erection

35/36

SECTION In

SHIELDED WOP ARRAY ELEMENT DESIGN

This section includes the details of the shielded loop investigation and design with its integral monitor and comer reflector. Also included are the performance tests of individual elements and an engineering model array of three elements.

A. LOOP THEORY

The shielded loop antenna is a conducting loop with a feedpoint that is a gap excited by a line running inside the element as shown in Figure 31. This element type was chosen for two basic reasons:

The fields are small on its axis (reduced mutual coupling).

Performance is insensitive to stray capacitances (environmentally hardened).

Specifically, any small loop will do the former, whereas the latter is accomplished by "shielding." The theory behind these two qualities will be examined here in some detail since it is essential to an understanding of the 'design approach.

UNBAL.ANCED GENERATORTRANSMISSIONI

I L.INE

~ .

~---- ... -------% FEED

POINT (GAP)

186009

Figure 31. Electrical Design of Shielded Loop

37

1" Small Loop

Loop antennas have been extensively analyzed in published literature,4 especially single-tum circular loops which are discussed in this section.

The radiation electric field E for a circular loop of radius b fed with a 6-generator at l/J =° is given in terms of the vector potential A:

I: I cos f) n At) = -21TG - exp [jn (l/J + 1T12)] In(kb sin f)) -.- kb

~ sm f)

Aq, = -j21TG I: ~ exp [jn (l/J + 1T/2)] Jln (kb sin f))

where

-jVeJ.Lo b e-jkr

G=--41T2 ~o r

and

k = the propagation constant 21Tlx

Ve = the generator voltage

Jn = Bessel functions

~ = coefficients calculated by King, et aI.4

f) is measured from the loop axis.

For small loops, the first two terms are adequate to determine the far field. For kb < I, approximate

. J 1 (u) ~ u/2

J~ (u) =J (u) - J .cu)/u ~ Y2o

Then for n =0,

W 1TGW E = - - =-- kb sin f)

80 k B80 a o

4R.E. Collin and F.J. Zucker, Antenna Theory Book, Part 1, McGraw-Hill Book Company (New York, 1969) p. 458.

38

and n = 1,

_ W Bel ::: -j21TGW cos e sin ¢ k al

W -j21TGWEC/>l = -k B4>1 = cos ¢

al

where

kb 8b j a ~ - (~n - ~2) -- (kb)4o 1T a 6

-18b jal ~ - (~n - -2) -- (kb)2

kb1T a 3

The term b is the loop radius and a is the cross-sectional radius of the loop wire.

For convenience, this "zero" mode will be called the loop mode because it is the field caused by a uniform loop of current. The equi-amplitude surfaces are toroidal in shape and independent of the azimuthal angle, ¢. The next mode will be called the dipole mode because it is caused by equal currents at ¢ =0 and 1T, which are in the same direction [i.e., on one side of the loop, the current is in the -+V>(+y) direction and on the other side the current is in the -</>(+y) direction]. For very small loops with circumferences less than O.IA(kb < 0.1), only the loop mode is significant, but for the size of interest here (kb ~ 0.7), both modes are present.

The above expressions indicate that for small kb, EI lEo a: kb at a given angle. This means that the contribution of the dipole mode to the total radiation increases with loop size. Furthermore, the fields on the loop axis are caused entirely by this dipole mode, and they increase like l/y'al *al a: kb. This is true even for kb as large as the region of interest:

8b ~n - -2 ~ 3

a

so that

1 .,. ---- ~kb·

kb4

+9

Therefore, a design guideline has been to keep the loop small to avoid axis fields, which are closely 'related to coupling between elements.

The limiting factor in reducing loop size is, of course, efficiency. That is, if the, reactive component of the impedance is very high with respect to the real part (radiation resistance), the loop will be lossy because high currents will be required in the' loop. The theoretical impedance has also been extensively published5 and is shown in Figure 32 as a function of loop size

~K .. Iizilka. et al.; "Self and Mutual Admittances of Two Identical Circular Loop Antennas in a Conducting Medium and in Air:' IEEE Trans.• Vol. AP 14. No.4. p. 440.

39

Figure 32. Theoretical Loop Impedance

(frequency). Clearly, if the theory is accurate, loops of kb < 0.6 will have very low efficiency; in fact, kb ~ 0.5 looks much like an open circuit. Loops smaller than 0.5A in circumference have a very low radiation resistance; therefore, the design was limited to the 0.6 to 0.75 region where matching at the feedpoint requires a large series inductance. This theoretical impedance was computed for a thinner loop wire than the one used. The thickness (nt . '" I0) and shunt capacitance at the gap were found to affect impedance significantly.

2. Shielded Loop

"Shielded" refers to static electricity as might be encountered on an outboard airborne loop. For such de voltages, the whole loop appears to be at ground potential except at the gap which is perpendicular to this voltage gradient. Thus, no static charges can affect reception

40

(A) ELEMENTARY LOOP

+

LINE

(8) PRACTICAL UNl3ALANCED ....OOP

(c) BALAHCE:O LOOP

186011

Figure 33. Evolution of the Loop

41

(REF.ERENCE PLANES)

(A) LOOPS 30 INCHES APART IN REFLECTOR

_---"'T"-------4I.....----"T""----!y1 2 t---.....,r-----...---.,

I

o (B) EQUIVALENT CIRCUIT FOR COUPLED LOOPS

I

I I J

Y1 Y 2I I I

I

0 AND 0 (C) REFERENCE FOR MEASUREMENT

186013

Figure 35. Parameter Definitions for Mutual Coupling

44

S12 can then be expressed in terms of Y11 = Ys' Y22 = Ys' and Y12 = Ym by solving these simultaneous equations for Y1 and Y2 :

Y~ :::

Substituting into the expression for S12 and letting Y be small, m

where G denotes the real part. Note that S12 is not being measured at these reference planes, but since the matching networks are lossless, the new reference of Figure 35(C) is readily accounted for. Inserting lizuka's admittance parameters for loops of kb = 0.75 with 0.81 A spacing into this formula yields 16 dB, which was also measured in the absence of the reflector.

B. EXPERIMENTAL INVESTIGATION

Extensive experimentation was accomplished to determine the best loop and reflector design. Principal variables were loop size, reflector configuration, and feedpoint configuration, with performance criteria on radiation patterns and impedance.

1. Loop Size

As indicated in the theoretical discussion, the impedance has a strong dependence on loop size with larger loops being easier to match. Loops were tested at many frequencies, hence at many sizes of loop circumference in wavelengths, kb; but effort was concentrated on the values kb = 0.58 and 0.75. It was found that with the same matching device, a VSWR lower than 1.5 could be achieved with the larger loop, but the smaller loop had a VSWR greater than 2.3. These measurements are at the band edges, ±1 percent, when the center frequency is precisely matched.

A comparison ,was also made of the radiation patterns produced by the two loop sizes in front of a comer reflector. Figure 36 shows the horizontal (E-plane) patterns for a spacing of 8 inches from reflector vertex to the nearest point in the loop centerline. 'The difference in patterns because of loop size is negligible. However, mutual coupling was found to be higher with the larger loop as expected, e.g., about 2.5 dB without the reflector.

45

.'10' o' ,0

180

10'.'

110'

CIRCUMFERENCE

-0.75X ---0.58X

186016

Figure 36. E-Plane Patterns of Different Sized Loops

2. Reflector

These performance characteristics were also investigated as a function of reflector width, vertex angle, and spacing from the loop. In particular, patterns were measured for various reflector widths and vertex angles, but the impedance was not appreciably affected by these parameters. For angles varying from 90 to 180 degrees (flat), the best patterns were obtained from the 90-degree vertex reflector. Also, the best reflector size was found to be 20 inches on a side, although this dimension was not as critical. Figure 37 shows comparable patterns from Wand 20-inch reflector widths respectively. These cases have a vertex-to-loop spacing of 6 inches. As this distance increases, the patterns improve for the 10-inch case, but deteriorate for the 20-inch case. This is especially true of the front-to-back ratio which is only 2 dB better for the former case at a spacing of 12 inches.

46

WIDTH OF EACH SIDE 1511· 10· o·

o

o ISO

_ 20 INCHES 10 INCHES

186015

Figure 37. E-Plane Patterns of a Loop in Right Angle Reflectors of Different Widths

This reflector spacing affects the impedance, as well as the patterns, as shown in Figure 38. The data were measured on a scale model which was matched at about 1.4 GHz with a series

,,: stub. On the full-sized antennas, as well as on the models, decreasing the spacing was shown to increase the effective shunt capacitance at the feedpoint. This phenomenon is further discussl;\d in the section on impedance. Finally, a smaller loop-to-reflector spacing was found to decrease mutual coupling between elements, e.g., extension from 6 to 8 inches increased coupling about 0.5 dB.

47

, I

Figure 38. Impedance of Loop in Reflector With Different Spacings

Besides pattern and impedance perfonnance, the wind loading on the tower must be considered in the electrical and mechanical design. In the interest of reducing wind loading, another reflector was fabricated with O.25-inch aluminum rods held in horizontal position by I-inch tubing around the edges of the reflector. Measurements were made on this reflector for vertical spacings of I inch and 2 inches with very little difference. There was some difference between the performance of this thin rod reflector and the solid reflector as indicated in Figure 39. The rod reflector generally had better performance than the solid one in that the azimuth beamwidth generally decreased from 79 to 73 degrees, and the front-to-back ratio improved by about 3 dB.

48

186017

Figure 39. E-Plane Patterns of Loop in Different Reflectors

In the absence of diagonal members in the reflector, the cross polarization in the principal planes is theoretically zero. Therefore, cross polarization measurements are basically an indication of loop-reflector alignment and antenna polarization alignment, which have little to do with intrinsic. antenna properties. However, with nominal alignment effort, levels were typically measured at a maximum of -25 dB, and occasionally, as low as ~30 dB. It was found that diagonal braces in the plane of the reflector produced cross polarization. Therefore, these diagonal braces to the reflector edges will be extended back to the tower legs rather than the reflector vertex. The impact on mechanical design is negligible.

49

Ii r !

186018

Figure 40. E-Plane Patterns of Balanced and Unbalanced Fed Loops

3. Loop EXcitation

There is more than one way of feeding the loop which can be described as "shielded." Specifically, it can be balanced or unbalanced, and the gap width can vary. A typical pattern for the unbalanced feed case is shown in Figure 40 compared to the balanced case of the same configuration. The differences are quite insignificant. Similarly, no difference in impedance was found between the balanced and unbalanced cases except for the important consideration that the balanced feed requires a balun. These results are discussed more extensively in the section on loop theory.

50

..

..

O.lt 0.13

186019 "'0 Lt"O

GAP WIDTH ---0.100 --0.125

Figure 41. Effect of Gap Width on Loop Inpedance

The gap Width was not found to affect patterns, but it does affect impedance. As expected, Widening the .gap decreases the shunt capacitance across the feedpoint, much like extending the loop out from the reflector. Figure 41 shows the change in the impedance of a stub matched coax loop caused by a gap increase from 0.100 to 0.125 inch. The reference plane for these plots is more than a wavelength from the feedpoint.

4. Summary

Table 3-1 contains a summary of the pattern characteristics. It should be noted that the H-plane pattern is not too significant since it is dependent on the reflector height, which was only 30 inches in this case. Measurements with a 90-inch reflector showed a narrower H-plane pattern and a slightly better front-to-back ratio.

51

TABLE 3. PATTERN CHARACTERISTICS

BW3 Value at 90 Degrees Minimum Back (degrees) (dB) (dB) (dB)

E-Plane 73 -11.4 -30 -16

H-Plane 101 -8.6 -19

Other conclusions resulting from this investigation are:

Loop size has little impact on patterns.

Increased loop size improves impedance but increases coupling.

There is no difference between balanced and unbalanced feeds.

Pattern changes over the band are insignificant.

E-planewidth is minimized for reflector spacing between 6 and 8 inches.

Spacings less than 5 inches cause pattern degradation.

Spacings less than 10 inches affect impedance.

Directivity increases with reflector width.

Figure 42 shows one piece of the full-sized hardware built for this investigation. This reflector is of perforated metal which, electricallY, acts like a solid surface. A few of the various models tested are illustrated in Figure 43. On the left is a scale model of the large reflector; in the center a narrow reflector corresponding to 10 inches full-sized; and on the right is a noncircular loop in a 120-degree angle reflector.

C. LOOP IMPEDANCE MATCHING

The. theoretical impedance of Figure 32 was essentially verified by experiment with a thin loop. However, Texas Instruments shielded loops are significantly thicker (nt = 9-10) than that of the theoretical calculation. Increased thickness basically shifts the frequency along the same locus so that, at a given frequency, the capacitive reactance decreases. With this starting point, the inipedance matching approach is described below.

1. Reflector Effects

The effect of the narrow gap and reflector proximity as discussed earlier can be evaluated by adding shunt capacitance as shown in Figure 44. It can be seen that addition of shunt capacitance is almost equivalent to rotating the impedance curve clockwise on the Smith Chart, which reduces the radiation resistance and capacitive reactance. This means that by narrowing the gap and reducing the loop-reflector spacing, both of which introduce shunt capacitance, the radiation resistance can be brought down to 50 ohms (from 90 ohms as in Figure 32 while· reducing the reactance).

52

186020 (47-1611)

Figure 41. Full-Sized Loop and Reflector

53

186021 (47-1612)

Figure 43. Sample of Scaled Loops

2. Series Stub

After appropriately adjusting the shunt capacitance, the mismatch is almost entirely due to capacitive reactance, which means inductance must be added in series. Several ways which have been investigated are illustrated in Figure 45. The bazooka balun in A was used to in trod uce inductance by adjusting its short. The open stem of C is inductive if less than 11./4 long, but has the disadvantage of leaving currents on the stem, i.e., leaving it sensitive to environmental effects.

The series stub was chosen as the most straightforward implementation of the required series inductance. The new impedance is found by rotating about the circles of constant resistance. Note that, since the impedance locus for this frequency band nearly coincides with the real 50-ohm circle, a change in this inductance can be used to change the match frequency. This is accomplished by changing the position of the short along the loop, i.e., making the stub shorter for a match at a higher frequency. This method has achieved VSWR measurements of nearly 1.4 over the band, depending on loop thickness.

3. Broadbanding

Broadbanding the antenna by detuning it and introducing a length of line with a transformer has been investigated. Detuning would involve changing the shunt capacitance by widening the gap or moving the reflector to make the impedance lie on about the 70-ohm circle. Then, a few wavelengths of 50-ohm transmission line would broaden the band because it looks

54

(A) ORJGINAL Z (B) Y INCLUDING REFLECTOR'

186022 (C) Z WITH R:::FLECTOR (D) Z WITH SERIES STUB

Figure 44. Loop Impedance Matching Method

shorter at low frequencies and longer at high frequencies. Figure 46 illustrates the steps. Finally, a quarter-wave transformer of about 59-ohm line would rematch the system. The major drawbacks of this method are that the transformer would be inconveniently far from the antenna and the monitoring coupler would be affected by standing waves if placed near the antenna.

More recently, a resonant circuit has been investigated with superior results. It is basically an inductance and capacitance in parallel with the feedpoint, but on the generator side of the series stub as shown in Figure 47. Hardware limitations require the circuit to be Ag /2 or Ag away from the actual feedpoint, but this has little effect for the narrow band being considered.

55

(A) BAZOOKA BALUN

BALUN

(B) SERIES INDUCTOR

(C) SHORTED STEM LINE

186023

(D) SERIES STUB

Figure 45. Shielded Loop Matching Techniques

Intuitively, this circuit is designed to have no effect at the center frequency and to have. inductive susceptance at low frequencies where the antenna is capacitive and vice versa at high frequencies. As shown in Figure 47, Ym = Yb + Yl' To design this circuit, two conditions determine Land c:

where W is center frequency. o

where We is edge frequency.

56

(A) MATCHED LOOP (p) DETUN!tD WITH CAPACITANCE.

(C) LOOKING THROUGH nA/z L.U'&: (D) WITH )./4 TRANSFORMER

186024

'Figure 46. Broaclbanding With Tran\$former

This circuit was implemented with stripline in the stem of a stripline loop by putting an open stub and a short stub on' opposite sides of the transmission line. Thus, the .conditions become:

(5)

(6)

where

p = w/c 20 , 25 are open and short stub lengths

BI = Im(Yl)'

57

l66025

Va ANTENNA

Cr = REFLECTOR EFFECT

L s = SERIES STUB

= ADMITTANCE LOOKING INTO REFERENCE PL.ANE l

= BROADBANDING CIRCUIT ADMITTANCE

= ANTENNA ADMITTANCE LOOKING FROM 50...oliM LINE

Figure 47. Design Approach to Loop Broadbanding

Finally, after approximating Equation (6) with

-B=

1

one has:

Qo + Qs = >../4

and

58

__ ~.," ... "".,_",.,." "',Or,, .",,· ".

·186026

'1'0 ~I'O

• LCO

Figure 48. Impedance of Broadband Loop

. These are 'tl!e . stripline .broadbanding design equations. Implementation and testing yielded the impedance plot of Figure 48.

D. . INVESTIGATION OF ENVIRONMENTAL EFFECTS

In order to minimize environmental effects, specifically those of ice and rain, on the performance of the IQop element; many experimental measurements were taken. Rain was simulated by a spray of water which could be varied from a light mist to a heavy stream. Ice was simulated by a methyl hydroxystearate wax (Paracin I) which has a dielectric constant of about 3 at· the glideslope frequency. This is reasonably close to ice, which would have approximately

59

Figure 49. Impedance of Ice-Covered Loop

€ = 3.3 at this frequency. Similar investigations have been dorie elsewhere with this wax with good results.' ,.

1. Environmental Test Results

A small radome over the gap with a honeycomb spacer was first fabricated and tested. It greatly improved the performance in rain, but the performance in the iced condition was unsatisfactory. One-half-inch radial simulated ice had the impedance chart shown in Figure 49. The resonance was shifted down in frequency so that the VSWR at the upper band. edge was

'R.W. Wesley and P.G. Accardi, "Effects of Simulated Icing on Radome and Antenna Radiation Patterns," Proc. of Eleventh Symp. on Electromagnetic Windows, August 1974.

60

about 4.3. Furthermore, as the ice was removed symmetrically, little by little, beginning at the gap, the impedance did not recover until nearly all the ice was removed. The iced loop is shown in Figure 50 with icicles. The icicles were removed and found to have little effect.

Second, a conventional type radome of prolate spheroidal shape, as shown in Figure 51, was then iced and tested with very good results. The maximum impedance of a coax loop was degraded only ~lightly' from 1.6 to 1.7 when the Paracin I was applied. The radiation pattern beamwidth was broadened only slightly as seen in Figure 52.

The third approach was to integrate the radome with the loop element. Two basic types of such "integral" ladomes were investigated, one with honeycomb and one with dielectric. The purpose of the former is merely to space the radome and, therefore, the ice from the loop surface, whereas the latter was designed to preload the antenna with material similar to ice so that addition of ice would be a small perturbation of existing loading. Figure 53 shows the integrated radome configuration. Impedance plots of these two loops are in Figure 54. It was found that the spacing between radome and radiating element is critical. The further they are separated, the less are the effects of environment. The polypropylene case had a VSWR of 1.35 before icing which, as shown, was degraded to 2.3, whereas the honeycomb case was degraded from 1.15 to about 3.1. Thus, the preloading approach was moderately effective.

. 2. Performance Requirements

It must now be considered what VSWR is necessary for adequate pattern performance. As discussed in Section II, of the total signal error [aT = (0.2 dB, 2 degrees)], about one-third of ' the variance is allowed for element mismatch; Le., ae = (0.12 dB, 1.2 degrees) for independent errors. This means the tolerance is about ±3a = ±(0.36 dB,. 3.6 degrees). But a reflected signal e cannot increase the radiated power which means there cannot be a positive error in decibels. Therefore, the power error limits can be translated to 0 dB, -0.72 dB, where -0.72 dB represents a VSWR of 2.3. Recall that this limit is for random error, and if the loops were all iced evenly, the error limits could be even wider. Thus, the integrated radome approach has marginal performance and its viability cannot be resolved without pattern measurement of an actual array.

In summary, it has been shown that unavoidable effects are caused by ice being too close to the loop element. This is fundamental and would occur for ice on or close to a dipole as well. It also has been shown that, while an integral radome has possibilities, a small cylindrical or spheroidal radome can definitely. solve the problem. Furthermore, there are other considerations which would lead to the choice of a nonintegral radome. For instance, birds could sit or even nest on an integral radome, most likely in an asymmetrical manner. A bird on the cylindrical radome would be far enough away not to affect impedance. Finally, the cost of the integral radome. approach is expected to be higher and the difficulty of tuning much increased. The difference in wind loading has been calculated to be negligible.

E. FINAL LOOP DESIGN AND PERFORMANCE

. Two basic designs of the shielded loop have been investigated, the coax construction and the stripline construction. The coax approach has been chosen for the image loop design because of its simplicity and low cost in the absence of monitor and broadbanding requirements.

61

186028 (47-1708)

Figure 50. Loop With Simulated Ice

62

186029 .(47-1782)

Figure 5 . Ice-Covered Radome of Loop Antenna

63

186030

Figure 52. E-Plane Pattern of Loop and Iced Radome

However, the stripline version, which has identical pattern performance, has several advantages for the non-image array application. The integral monitor is inside the stripline loop requiring no connectors, and likewise, the broadbanding circuit is much easier to implement. In addition, the • construction tolerances are easier to maintain with stripline so there will be essentially no standing waves between these components. Finally, the matching stub is more readily adjusted in the stripline case. The only significant disadvantage to this approach is that it will either cost more per loop (in 'production quantity) or be lossy if inexpensive dielectric is used.

64

~ ~

LAYER OF HONEYCOMB

0\ VI

COPPER SIDE

ETCHED ON THIS SIDE ONLY

, SINGLE-SIDED COPPER CLAD DIELECTRIC SUBSTRATE

DOUBLE-SIDED COPPER CLAD DIELECTRIC SUBSTRATE

Figure 53. Loop With Integrated Radome

186031

Figure 54. Impedance of Ice-Loaded Integrated Radomes

1. Element Design

The mechanical design is shown in Figure 55. The loop-to-reflector spacing mainly affects impedance but is not critical. The loop could be made thinner with plated laminate edges if desired. The stem is reinforced with a ground plane plate, but; in order to save material and weight, the loop is not reinforced.

The electrical design is shown in Figure 56. All line impedances are 50 ohms. The monitor coupler is -20 dB 'and has a type N connector on each end. The feedpoint is a gap in the ground plane but not in the dielectric.

66

t 2 11 DIAMETER RADOME

8.511

186033

Figure 55. Mechanical Design of Shielded Loop

2. Performance

To determine the performance of the basic loop and reflector design, a three-element array was built as shown in Figure 57. The loops are all identical to the single-element hardware but the reflector is 90 inches long and made of sheet metal. Patterns were measured on a single element in the presence of two other loaded elements to determine the effects of coupling on the pattern. The E- and H-plane patterns are shown in Figures 58 and 59, respectively. As would be e~pected because of the long reflector and the coupling, the H-plane pattern has narrowed . considerably. The front-to-back ratio has also improved over that of the small solid reflector. Patterns were measured with the three elements fed with equal amplitude and phase with the

67

BROADBANDING STUBS

SHORT

_ FAULT MONITOR

_.- MONlTORlNPUT

186034

Figure 56. Electrical Design of Stripline .

results shown in Figures 60 and 61. Here the E-plane beamwidth has decreased somewhat and the H-plane is beginning to look like the well-known sin xix pattern. As the whole array is included, it is expected that the E-plane will change very little from this pattern and the H-plane will, of course, .become very narrow.

Essentially, no change was found in the loop impedance· because of· the presence of the other loops, which means that the loops may be tuned separately if desired. Mutual coupling was measured and found· to be low compared to dipoles over a ground plane. Figure 62 shows the transmission loss between the three combinations of two loops in the three-elemerit . array.

68

Figure 57. Three-Element Loop Array

69

186036

Figure 58. E-Plane Pattern of Loop in Array Environment

Between adjacent loops, the isolation was 22 dB and between end elements, it was about 26 dB. Resonant dipoles, placed A/4 above a ground plane, have been investigated at Texas Instruments with the result shown in Figure 63. The analytical curve follows the actual measured coupling quite closely. The 0.808A spacing will produce dipole coupling of -19 dB, which is 3dB higher than the loop case.

Again the question of performance requirements must be addressed. Mutual coupling is one of the more harmless antenna problems in the sense that it can usually be compensated for. For instance, if the isolation is not high enough, the distribution network should not be designed for

70

1fII· 10· 10· 350·

;;lto;r-~~~"""'"J1J110• 170° 110

0

186037

Figure 59. H-Plane Pattern of Loop in Array Enviromnent

the . desired excitations, but rather to produce the desired excitations in the presence of other elements. Furthermore, if the array element pattern is different from the single-element pattern, these desired excitations can be synthesized for the coupled element case. Thus, all the coupling effec;ts can be cancelled by an appropriate distribution network design. However, since design simplicity is desirable, a concerted effort has been made to keep the loop coupling low. It is believed that the coupling is low enough that no special compensation is required in the distribution network.

71

186038

Figure 60. E-Plane Pattern for Three-Element Array

72

Figure 61.H-Plane Pattern for Three-Element Array

73

FREQUENCY (MHZ)

328 330 332 334 336 o

1 I I. BAND .. \

I I I I

-10 I I"" ID I I8

l!).

: I-..I zIL I I:l

8 -20 I :ADJACENT LOOPS

_-l--------------------~-------~---I I

I -----·------ --- .~ ......... --_..-.1 I

I OUTER LOOPS -30

186040

Figure 62. Measured Mutual Coupling of Loops

74

DISTANCE BETWEEN DRIVEN EL-EMENT (A)

o 0.5 1 .0 1.5

-10

~ ...I 0::J o U

-20

-30 186042

Figure 63. Mutual Coupling of Dipoles Over Ground Plane

75/76

SECTION IV

IMAGE LOOP INVESTIGATION

The feasibility of using the shielded loop element developed under this contract as the radiating element in the. image glideslope arrays has been demonstrated. This type of element is desirable because its low mass makes it suitable for mounting on a frangible mast. Furthermore, in two add-on contracts, the following items were fabricated, tested, and delivered:

Two engineering prototype, single-frequency, coaxial loop antennas

Two production-type prototype, whole glideslope frequency band, stripline loop antennas with integral monitor and radome.

The following sections provide a·description of these antennas.

A. SINGLE-FREQUENCY COAXIAL LOOP

1. Experimental Investigation

Since a small loop has low directivity, a simple reflector has been added as shown in Figure 64. The following parameters were varied for optimum performance:

Reflector angle

Loop-reflector spacing

Number of elements

Reflector element spacing

Reflector length.

It was again found that an angle of 90 degrees at the reflector vertex was optimum for directivity, whereas the spacing from the loop to the vertex gave the best performance at 6.5 inches. Reflectors of three elements (Figure 65) were tried, as well as reflectors of one element, producing a negligible increase in front-to-back ratio. Figure 66 compares the reflectors. The effect because of a change in spacing between reflector rods from 2 inches to I inch was insignificant. However, the reflector length was critical, with 10 inches being best for both the one- and the three-element reflectors. The reflector l~mgthwas adjusted by means of a telescoping rod as shown in Figure 65. This type of rod led to the discovery that a stepped reflector of part 0.5 inch and part 0.25-inch rod was better than a solid rod of either size.

Since the single rod reflector performs as well as the three-element reflector, it was chosen with the loop as the radiating element for the image array. The impedance of this element is different from the planar corner reflector because the rod reflector tends to resonate. Basically, the rod reflector makes the system more narrow-banded with band edges at a VSWR of about 2. The broadbanding circuit discussed above can be applied to this element as well as the non-image loop.

77

166042 (47-1673)

Figure 64. Experimental Model of Image Loop

166043 (47-1709)

Figure 65. Image Loop With Three-Element Reflector

78

186044

Figure 66. E-Plane Patterns for One- and Three-Reflector Elements

The shielded loops of stripline construction were also investigated with the single rod reflector. The performance was similar, and this approach facilitates addition of the broadbanding circuit. However, this approach was not pursued because a long lead time was required for fabrication, and the short delivery time could not have been met. Furthermore, the broadband match is not necessary since it will be tested only at one frequency: Thus, the coaxial loop with a series-matching stub at the feedpoint was the recommended design.

Environmental effects were investigated by simulating rain and ice on the loop as was done with the noh-image element. Results were similar.

79

2. Design and Performance

The shielded loop is a piece of 50-ohm coaxial cable bent into a circle and connected with silver-loaded epoxy to make an electrical loop. The outer conductor is cut to form a 0.2-inch gap directly opposite the stem. A short is then introduced in the line in the side of the loop opposite the feedline.

The gap is covered with a block of closed-cell urethane foam sealed with a glass-filled epoxy coating to minimize the effect of rain. The mounting bracket allows adjustment in 3 degrees of freedom:

Loop-to-reflector spacing

Extension away from the mast

Height on the mast.

The mounting bracket is aluminum and supports two aluminum reflector rods as shown in Figure 67. The E- and H-plane pattern performance of these image elements is shown in Figures 68 and 69 for maximum offset from the support mast. The E-plane beamwidth is about 84 degrees, and the front-to-back ratio is 10 dB. Note that the vertical pattern is not desired to be narrow since it must illuminate the ground to produce the image.

Figure 70 shows the impedance (referenced to the antenna connector) of one of the delivered elements which is matched at 333.8 MHz. Figure 71 is a photograph of the element.

B. STRIPLINE LOOP (WHOLE GLIDESLOPE FREQUENCY BAND)

1. Design and Performance

a. Antenna Element Design

In order to cover the whole glideslope frequency band, the stripline loop approach (as described in Subsection III.e.3) was. taken. This approach provided a highly repeatable and accurate fabrication method. During the development phase, the stripline approach also provided convenient access to the open- and short"circuit impedance matching stubs for tuning purposes as shown in Figure 72. This would have been more difficult with a wholly coaxial configuration.

The dielectric material used for the stripline antenna is a glass-filled, cross-linked polystyrene made by Atlantic Laminates. This material has excellent, very low loss, electrical properties and is commonly used in stripline circuits at much higher microwave frequencies. Key factors used in the selection of this material were its moderate cost and low electrical loss properties.

b. Monitor Configumtion

The antenna/monitor configuration shown in Figure 56 has an overall length of 30 1/8 inches. It is possible to reduce the length of the antenna stem by moving the fault monitor coupler to the loop side of the tuning stubs as has been done for the delivered hardware. This arrangement (Figure 72) has resulted in a shortening of the antenna stem by 5 inches, which is superior from a mechanical point of view.

80

186045

'~G . r~ ~ ...II

-I

Figure 67. Design Detail of Image Loop

81

186046

Figure 68. E-Plane Pattern of Delivered Image Loop With Pole

82

t 86047

Figure 69. H-Plane Pattern of Delivered Image Loop

83

Figure 70. Impedance of Image Loop Element

84

..... o Q)c: o

85

STRIPLINE LOOP ANTENNA ELEMENT

DIRECTIONAL COUPLER FOR INTEGRAL MONITOR

MONITOR OUTPUT

" .... ANTENNA INPUT

MATCHING STUBS

196712

Figure 72. Stripline Loop Antenna Circuit Including Matching Stubs and Integral Monitor

Because of the existing glideslope system, only one input is available for the monitor. The monitor input port normally provides energy coupled in the forward direction for the course width and boresight in lieu of a far-field monitor. The reflected signal caused by element failure is added out of phase to the forward coupled signal for element fault detection.

It was initially intended that the stripline antenna would be a bonded (hermetically sealed) assembly. During the development phase, it was determined that conventional bonding techniques caused an unpredictable detuning phenomenon in the antenna. Available time and funding did not permit a thorough investigation of the cause and cure of this situation. In view of this, the three antennas deHvered (two production prototypes and one engineering model) employed O.090-inch thick aluminum clamping plates secured by 2·56 stainless steel screws. The edges of this assembly were sealed with an epoxy sealant. Appropriate steps were taken to treat the interfacing surfaces of the copper-clad ground plane and the aluminum clamping plates to eliminate the possibility of corrosion.

The electrical performance of the production-type stripline loop antennas proved satisfactory. Figures 73 and 74 show typical E- and H·plane patterns. Figure 75 shows typical antenna impedance. The coupling and fault detection characteristics of the monitor are as· indicated in Figure 76.

86

c

Figure 73. E-Plane Pattern for Stripline Loop Antenna

87

Figure 74. H-Plane Pattern for Stripline Loop Antenna

88

Figure 75. Impedance of Loop Under Normal Test Conditions

89

...,0

\ REFERENCE POWER LEVEL AT ANTENNA INPUT PORT

-10

" ID / COUPLED FORWARD POWER LEVEL AT MONITOR PORT UNDER NORMAL TEST CONDITIONS

"C '-'

\0 0

.J W > w .J -20

o 328.6 MHZ

0 332.0 MHZ

335.4 MHZ

o a: W ~

~

-30 .... I POWER LEVEL AT MONITOR PORT DUE TO -.... ANTENNA IS SHORTED .... ...... ....

'G. ...

COMBINED FORWARD AND REFLECTED SIGNAL WHEN ,----,.Z'

,. '335. 4 MHZ

328.6 MHZ ......... ...-- 332.0 MHZ

- - - - - - - - - - e - - - - ;' "" "" - ---'" 196716

Figure 76. Electrical Performance of Integral Monitor

c; Radome

A cross section of the protective covering and base support for the antenna is illustrated in Figure 77. The antenna rests on a ~-inch thick layer of low-density foam supported by a fiberglass· channel as shown. The purpose of the foam is to remove the antenna from the proximity effects of the fiberglass channel. The upper hemispherical portion of the radome has a large radius to minimize the detuning effects of ice on the radome. If the radome were a lower profile, the detuning effects of ice would not be acceptable.

d. Supporting Structure

The antenna and radome are supported by the hardware bracket arrangement shown in Figure 78. The bracket is designed to prevent the possibility of the antenna rotating on the mast. Figure 79 shows the antenna in its completely assembled form.

2. Environmental Tests

The antenna was tested with a Y2-inch thick coating of simulated ice as shown in Figure 80. The antenna impedance with the Y2-inch thick ice coating is as shown in Figure 81, indicating that the VSWR is well within the maximum specified requirement of 2.0: 1. In addition, the production-type prototype was tested in the environmental laboratory under the temperature and humidity service conditions called out in Enyironment III of FAA-G-21 OO/lb. When these tests were completed, the antenna was found to have experienced no detrimental effects.

HEMISPHERICAL RADOME

FOAM SPACER 1/4-INCH TH ICK ANTENNA

196717

Figure 77. Cross Section of Antenna, Radome, and Support

91

196718

Figure 78. Antenna and Supporting Brackets

196719 (65-47) II

Figure 79. Stripline Loop Antenna as Delivered

92

Figure 80. Stripline Loop Antenna With Y2-lnch Thick Simulated Ice

The wind loading te~:l was conducted by loading the antenna with a static load of sand in accordance with Lnvironment III of FAA-G-2100jlb. No apparent damage from the wind loading tests was observed. Thus, the antenna was determined to have passed the environmental test req uiremen ts.

C. CONCLUSIONS AND RECOMMENDAnONS

As discussed in Subsection IV.B.I, the initial intentions were to bond the stripline an tenna assembly. This, however, met with difficulties introduced by the detuning effects of the bonding process. Appropriate bonding techniques are very desirable for mass production and will require additional investigation prior to implementation.

The stripline loop antenna that is the final product of this contract covers the whole glideslope frequency band and includes an integral fault-detection monitor. This is a definite improvement over the earlier coaxial single-frequency loop antenna with no integral monitor. The broadband stripline loop antenna with integral monitor is a cost-effective antenna suitable for large-quantity production and can also be used for the non-image broadside array application.

93

Figure 81. Impedance of Stripline Loop Antenna With and Without Simulated Ice Loading

..

94

SECTION V

FABRICATlON PLAN

A. REFLECTOR

The reflector consists of six sections 113.22 inches long and a half-section 56.61 inches long. Each section is a welded assembly of aluminum tubing. The horizontal tubes forming the reflector screen are 0.25 inch in diameter and are spaced on 1.91-inch centers. Supporting these tubes are two 1.00-inch diameter tubes that form the outside edges of the comer reflector. Another 1.00-inch tube supports the horizontal tubes in the middle of the reflector. Directly above and below the element a.50-inch diameter tubes are used in place of the a.25-inch diameter tubes for increased support. In construction, the welding is performed using a fixture which accurately holds the 90-degree angle between the reflector halves. A frontal view,of a reflector section is shown in Figure 82.

Before the completed reflector section is removed from the welding fixture, two welded subassemblies of 1.0o-inch diameter and 0.5O-inch diameter aluminum tubing are welded to the back of the reflector section. The back structure is added while the reflector is still on the fixture to prevent any change in the angle between the reflector halves. Front and side views of the reflector back structure are shown in Figure 83. After the back structure has been attached, the reflector assembly can be removed from the fixture and the loop antenna support brackets added. A strip~ine antenna loop mounted in a reflector assembly is shown in Figure 84.

B. SUPPORT STRUCTURE

A vendor-supplied support tower has been selected that will meet or exceed the requirements of Specification FAA-G-2100/1 and operate under the ambient conditions of Environment III (paragraph 1-3.2.23). The support tower selected for the array is an equilateral triangle with tubular steel legs and welded cross bracing. The cross-section size is 36.0 inches on leg centers. The leg sections are 2.5-inch-diameter schedule 40 pipe and the cross bracing is 0.75-inch-diameter solid rod. The tower is available in lo-foot and 20-foot sections. A 20-foot section is shown in Figure 85. The tower sections will be modified to include a tilt adjustment as discussed in Section II. Additional horizontal cross braces will be added to one side of the tower to provide a mounting surface for the antenna array sections. The existing tower cross bracing is

, all diagonal and does not provide adequate attachment points for the antenna array sections. The added braces will also be used to support the cable trough that will extend the length of the tower. The trough will be made out of sheet metal' and will have a resealable access cover. The monitor and signal cables will be routed to the two distribution units through this trough which will have built-in cable supports.

The single truss structure and the two guy cables are commercially available from tower manufacturers. The truss structure is an equilateral triangle with tubular steel legs and welded cross bracing. The cross-section size is 24.3 inches on leg centers. The leg sections are 2 O-inchdiameter 0.180 inch steel tube and the cross bracing is 0.625-inch-diameter solid rod. The tower is available in lo-foot and 20-foot sections. The tower sections will be modified as discussed in Section II. The worm gear actuator is a standard stock unit that is readily available from several commercial sources.

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112.965

1-------....; 28.284 --:-0.12

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1=-== -I==='~

F- -- -

p====- V 0. 50-INCH

'l===:===-= TUBING

V

113.22

-y ~--= ~=====~~_.

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/I

t====---= <l=======J

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Figure 82. Reflector Section Frontal View

96

59 SPACES AT 1.914 = TOL NON ACCUM

.'

186050

THIS BECOMES THE MIDDLE TUBE OF THE CORNER REFLECTOR

V I I ----

~ 1.00

~ TUBING

k---

1 I

;

.22

1 II

if

lS7.44 28.72

'~r-

I If I

L,9.00 3.83 4 PLACES

/"" 0.50 TUBING

86. 16

113

..

1 1 • 61

l186051

Figure 83. Reflector Back Structure

97

RADOME

REFLECTOR BACK STRUCTURE

STRIPLINE LOOP

186052

Figure 84. Reflector Assembly

C. SAFETY CLIMBING EQUIPMENT

The safety climbing equipment shown in Figure 86 will be supplied by the tower vendor. This equipment limits the maximum possible length of fall that may be sustained by a user with a safety climbing rail that is notched every 6 inches to provide a positive stop in which the climbing sleeve locking pawl engages. A sectional view of the locking mechanism is shown in Figure 87. This safety climbing equipment meets all requirements of Interim Federal Specification RR-S-OOI301 dated June 1967.

98

3.75

III N

20 FEET

"'--1-

STEEL ROO 1"----+---- 0.75 THICK SOLIO

2.50 SCHEOULE 40 STEEL PIPE

186053

Figure 85. Support Tower Section

99

SAFETY RAIL

,I I I I I

LOCKING PAWLI I I I I CLIMBING SJ-EEVE

I I I 1

186056186054

Figure 86. Safety Climbing Equipment Figure 87. Sectional View of Locking Mechanism

D. LOOP AND MONITOR

The loop and monitor fabrication is straightforward. The artwork has been done for a particular dielectric material and could easily be modified for material suitable for production, such as Poly CR. The laminate is then etched according to standard procedure and cut in the form of the loop (Poly CR is readily machinable). The two dielectric slabs will be bonded since nuts and bolts involve too much labor. They will then be plated where shorts are required on stubs, and the stem supports and connectors will be installed. Finally, the radome will be added as shown in Figure 53. The radome will consist of low-density foam molded around the loop and then sealed with an appropriate coating. An alternate method would consist of the separate radome shown in Figure 84. It is constructed in two pieces and bonded together. The sections are laminated fiberglass formed on a jig. The back of the radome is slotted to allow the radome to be removed without removing the stripline loop. To remove the radome, the fasteners are removed and the radome rotated 90 degrees to clear the loop.

E. DISTRIBUTION NETWORK

Double-clad dielectric laminate will be etched with the appropriate circuit and clamped between two 1/8-inch aluminum plates for rigidity. The connectors will be installed and measurements made of VSWR and the amplitude and phase of transmission. The plates will then be disassembled and pieces of thin dielectric tape placed over those couplers and phase shifters which are not within specification. Type N connectors will be used to connect cables which exit the distribution box and miniature connectors on those ports which connect to other boards in the same box.

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SECTION VI

CONCLUSION

The objectives of designing a broadside array with minImUm ground-plane influence and . designing an array element using a shielded loop have been accomplished. The theoretical

performance predicted for the non-image array will allow Category II glideslope operation at sites previously allowing only Category I operations.

The array design using the Schellkunoff polynomial was proven successful for the synthesis of low sidelobe levels below the horizon. With the computer program developed by Texas Instruments, other broadside arrays can be readily synthesized for various pattern constraints and array sizes. The radiation patterns synthesized for this program have rapid rolloff at the horizon and low sidelobes below the horizon sufficient to allow the use of the broadside array at difficult sites. To maintain these low sidelobes, care must be taken in fabrication of the array to minimize errors. The error levels required to maintain satisfactory radiation patterns are achievable with some adjustment of the distribution network. Furthermore, effects of mutual coupling have been shown to be much lower than for dipoles.

The monitor philosophy adopted was to use integral monitors with monitor outputs for the course, course width, and fault indication. The monitor will be a directional coupler in each antenna to monitor both forward and reverse power levels. The fault monitor was considered necessary so that changes in the lower sidelobe levels because of antenna faults could be reliably monitored. A full size array must be built before a definite decision can be made as to the number of elements to be monitored.

A shielded loop antenna element was designed and tested. The loop was designed with enough bandwidth to cover the glideslope frequency range using stub tuners in stripline. The monitor directional coupler was also included in the stripline portion of the antenna, thus eliminating two additional coaxial. connectors. In addition, environmental protection was considered for the loop. Isolating the loop a short distance from the ice was found to be successful in reducing ice influence upon the loop impedance. To determine the effects of ice on the array radiation pattern, a full size array must be built and tested. The use of a small radome around the loop was also investigated with excellent results.

The tower design was completed and a commercial tower was selected with safety climbing equipment. In addition, the array reflector was designed using an open lattice to minimize wind loading. This approach is estimated to save 40 percent on tower costs because a small standard size is used.

The distribution system has been designed for easy accessibility. It will not be sealed in foam or ·distributed over the tower, but the stripline networks will be contained in two boxes on the tower. With longer element cables, the networks might even be placed at ground level for field service convenience.

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The next step in the design of the broadside array is to build a full size array with distribution networks. The purpose of the second phase is to determine the sensitivity of the radiation pattern to fabrication errors and the presence of ice. The broadside array would be assembled and tested at the Texas Instruments Antenna Range under controlled conditions to characterize the array performance.

iThe use of a shielded loop in an image array was found to be feasible, and two engineering prototypes were fabricated and delivered. Further work is required in the image array loop to J. r add a monitor and to environmentally harden the antenna. Finally, study is required to reduce I

the azimuth beamwidth for sites where multipath signals are a problem.

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NON-IMAGE GLIDESLOPE INVESTIGATIONThe analysis and design of a broadside glideslope antenna array...

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