Njhx410e - Link Budgets
Transcript of Njhx410e - Link Budgets
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www. masoncom. com
Mason Communica tions Ltd 2001
WCDMA Radio Planning Course4 Network Design4.1 Link Budgets
Mason Communications Training: WCDMA Radio Planning Course
Module 4: Network Design
Section 4.1: Link Budgets
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4.1.2 Mason Communica tions Ltd 2001
Where are We Now?Introduction
UMTSOverview
AccessTechnologies
WCDMAIntroduction
ModelArchitecture
UMTSStandards
Mobile RadioChannel
NarrowbandChannel
WidebandChannel
Local MeanSignal
Path Loss
Diversity
DesignElements
Basic Radio
Principles
Antennas andFeeders
Interference
MatchedFilters and
Rake Receivers
WCDMAPhysical Layer
NetworkDesign
OperatorsDesign Guides
The PlanningProcess
Polygons
Site Placement
AntennaPlacement
FrequencyPlanning
ForwardCapacity
Planning
CourseOverview
ConventionalOptimisation
3GOptimisation
Radio ResourceManagement
Optimisation
CourseWash Up
LinkBudgets
Where are We Now?
The Course Map shows which section we are now on.
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What is in This Section?
Introduction
Classical 2G Link Budgets
UMTS Link Budget How It Differs
Summary
UMTS Uplink Link Budget
UMTS Downlink Link Budget
UMTS Link Budget Analysis
NetworkDesign
OperatorsDesign Guides
The PlanningProcess
Site Placement
AntennaPlacement
FrequencyPlanning
ForwardCapacity
Planning
Polygons
LinkBudgets
What is in This Section?
The Radio Planning section will concentrate upon the current industry and academic approaches in
analysing the UMTS or WCDMA Link Budget. The aim of this section is to demonstrate the issues,
and limitations in the UMTS Link Budget, when compared to conventional Link Budget analysis forGSM or TDMA. As a result of these limitations the need for more sophisticated approaches is
introduced. The use of Static and Dynamic simulation techniques for Link Budget and Detailed Radio
Planning, using RF Planning Tools (such as Aircoms Asset), are presented in the next Section, The
Radio Planning Process.
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Why is this Section Important to You?
Very Important that the UMTS Link Budget is understood
The UMTS Link Budget will be used in dimensioning anetwork or an area of a network
The Link Budget is the precursor to using a Network PlanningTool
The Network Planning Tool uses the Link Budget and isextremely dynamic in UMTS
Why is this Section Important to You?
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How Will You Learn?
Discussion
Worked Examples Exercises
Demonstrations
The differences betweenGSM and UMTS LinkBudgets
That the confidence ofthe UMTS Link Budget islimited when we considerall the UMTS parameternetwork interactions
The need for Simulationsto predict UMTSnetwork performance
How Will You Learn?
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Where Are We Now?
Introduction
Classical 2G Link Budgets
UMTS Link Budget How It Differs
Summary
NetworkDesign
OperatorsDesign Guides
The PlanningProcess
Site Placement
AntennaPlacement
FrequencyPlanning
ForwardCapacity
Planning
Polygons
LinkBudgets
UMTS Uplink Link Budget
UMTS Downlink Link Budget
UMTS Link Budget Analysis
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Approaches to Radio Planning
Generally there have been two key approaches to Radio Planning (i.eGSM Planning, or more precisely TDMA Planning):
Network Dimensioning using Link Budget Analysis
Network Planning using Radio Planning Tools
UMTS Presents New Challenges for the Radio Planner and RadioPlanning Process
Conventional Approaches are limited when considering UMTS RadioPlanning
There are classically two approaches to Radio Planning.
Network Dimensioning using Spreadsheet based Link Budget analysis has been used to provide a
basic understanding of Cell Range for GSM and TDMA networks. Business Plans for country wide
networks often use the Spreadsheet Link Budget to estimate the quantity of Base Stations required toprovide Coverage and Capacity, and the sensitivities behind the Link Budget and the overall site count
estimate.
Network Planning using Radio Planning Tools offer a detailed estimate of specific coverage and
service levels over specific areas for GSM and TDMA networks. Planning tools use Terrain and
Buildings information of the specific area and aim to estimate coverage levels at a resolution at the
order of the resolution of the Terrain data (e.g. 25m resolution). This allows collections of Base
Station locations, heights, and antenna configurations to be optimised or engineered to best meet the
coverage and capacity expectations for the area.
GSM or TDMA based Radio Planning is relatively straightforward. The Link Budget and RadioCoverage Levels are predictable, since all of the parameters which make up the GSM or TDMA
Radio Link are not dependant upon one another. In UMTS or WCDMA, there are many parameters of
the Radio Link which are inter-dependent upon one another. The classic WCDMA parameter
dependency is that of Cell Breathing, where the Cell Loading, or number of users per cell increases
the Interference Levels within that cell, which in turn reduce the potential range. These
interdependencies present new challenges to the Radio Planning process and approaches to UMTS
Planning.
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The Link Budget is used in dimensioning exercises, and tounderstand basic coverage principles.
Link Budget equations are normally developed using Spreadsheetformulas.
There are many different ways of presenting the same Link Budget.
A Link Budget should generally have 3 sections:
Tx Parameters Element
Rx Parameters Element
Propagation Parameters Element
The Link Budget
The Link Budget is one of the fundamental tools a Radio Planner should be familiar with. Radio Link
budgets are used in dimensioning networks, and more importantly can give valuable insight to how a
radio link might behave, in terms of Coverage Range, and Capacity.
There are many ways of presenting a Link Budget. Vendors, Operators, Standards bodies,consultancies often present the link budget in different ways. There is no universally accepted
approach to presenting a link budget. However, a Link Budget should be made up of three elements;
Transmission, Receiver, and Propagation Parameters. We shall use a certain approach, which largely
reflects the Link Budgets shown in [1].
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Link Budgets track Power levels along a transmission pathfrom Transmitter Output Port to Receiver Input Port
Two Distinct Links in a Duplex Communications Channel:
Mobile (transmit) to Base Station (receive) - uplink
Base Station (transmit) to Mobile (receive) - downlink
Tx Rx
Rx Tx
UPLINK
DOWNLINK
The Link Budget
Link Budgets strictly track Power Levels along a Transmission Path or Link from a Transmitters
Output Port to the Receiver Input Port. They can also be re-arranged in many ways, but normally Link
Budgets are re-arranged in a manner which provides information, such as the Maximum Acceptable
Propagation Loss, given all other Parameters. This Propagation Loss can then be used to estimate
Range, and hence estimate Coverage.
There are always two Link Budgets in a Duplex Communications Channel. A Simplex Channel (such
as TV Broadcast has one Link Budget). We often refer to these links as Uplink and Downlink. In the
US (and technically correct), theses are the Reverse and Forward Channels.
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Duplex Communication Link having two links are ideallybalanced in terms of their link budgets
Uplink: Low Power Mobile Transmission to HighSensitivity Base Receiver
Downlink: High Power Base Station to Low SensitivityMobile Receiver
Tx Rx
Rx Tx
UPLINK
DOWNLINK
The Link Budget
Having a Power Balanced Link is ideal in terms of Range, or Coverage. However, the Uplink and
Downlink power budgets are not reciprocal, as the Mobile Stations are limited in terms of the Power
they can Transmit, and the Mobile Stations must be made for a Mass Market and hence the Receiver
Sensitivity they can afford, can not be highly engineered. This presents a potential link imbalance,
which is counted by having a more highly engineered Base Station receiver, and hence Receiver
Sensitivity, and a higher power can be transmitted from a Base Station. Generally speaking the Uplinkoften presents the limiting case.
In GSM or TDMA planning, there are a number of parameters at the RF engineers disposal to
improve the overall Link Budgets, or to attempt to balance the Link Budgets. These parameters are
discussed later in this section, but include the use of Base Station Receive Antenna Diversity, and the
use of LNAs at the Base Station Receiver.
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Where Are We Now?
Introduction
Classical 2G Link Budgets
UMTS Link Budget How It Differs
Summary
NetworkDesign
OperatorsDesign Guides
The PlanningProcess
Site Placement
AntennaPlacement
FrequencyPlanning
ForwardCapacity
Planning
Polygons
LinkBudgets
UMTS Uplink Link Budget
UMTS Downlink Link Budget
UMTS Link Budget Analysis
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The 2G Link Budget
BTS
+30dBm
-104dBm
Tx Parameters Rx ParametersEnvironment Parameters
Tx
Power
Antenna
Gain
Body
Losses
Penetration
Losses
Path
Loss
LogNormal
FadeMargin
Antenna
Gain
Diversity
Gain
Feeder
Losses
Rx
Power
Tx
Power
RxPower
Rx
Sensitivity
RxSensitivity
The three parametric elements which make up the Link Budget are shown; Transmission,
Environment, and Receiver Parameters.
The Link Budget shows the Power Received at the Base Station Receiver (Uplink) given the all the
Link Parameters. If the Log Normal Fade Margin is set to zero, then the Rx Power calculated is theAverage Rx Power received for all locations (at the same distance, and in the same Environment of
course). In other words the actual Rx power can vary above (50%) and below (50%) this Average Rx
Power, due to location variability. The shaded/blurred elements represent the Locations Variability in
a certain environment.
The Log Normal Fade Margin is added such that the Rx Power represents the Rx Power Received for
a certain percentage of locations. This might be 90% of locations. That is Rx Power represents the
minimum Rx Power expected for 90% of locations variability.
The Link Budget aims to ensure that the Rx Power (for a certain %locations, given the LN Fade
Margin), does not fall below the Minimum Rx Sensitivity, and the slide indicates the MaximumAllowable Propagation Path Loss acceptable to maintain the Radio Link.
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The 2G Link Budget
BTS
+30dBm
-90dBm
Tx Parameters Rx ParametersEnvironment Parameters
Tx
Power
Antenna
Gain
Body
Losses
Penetration
Losses
Path
Loss
LogNormal
FadeMargin
Antenna
Gain
Diversity
Gain
Feeder
Losses
Rx
Power
Tx
Power
Rx
Power
RxSensitivity
Rx
Sensitivity
This slide simply shows the same Power Link Budget as before, but with a smaller Path Loss. The Rx
Power in this slide represents the Rx Power exceeded for 90% of locations (assuming that a Log
Normal Fade Margin for 90% Locations is added). In this example this Rx Power (90% Locations) is
well above the minimum Rx Sensitivity Threshold.
What would happen in reality is that the Mobile Tx Power would reduce in a GSM system such that
the Rx Power (90% locations) is closer to the Rx Sensitivity Threshold, thereby conserving Battery
Power, reducing unnecessary interference (to other cells), and still maintaining the Link Budget.
If one was to imagine an animation of the above slide, the Path Loss would vary, as the distance
varied, between MS and BS. Also the instantaneous Rx power level due to location variability would
dance around (obeying a Log Normal Probability distribution the shaded red parts) which would
result in Tx Power Variations, such that the Rx Power was => Rx Sensitivity Threshold.
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The 2G Link Budget
The Rx Power can be calculated given Path Loss and the Link BudgetParameters
The Max Path Loss can be calculated given the Minimum RxSensitivity and Link Parameters
TxPower
AntennaGain
BodyLosses
PenetrationLosses
PathLoss
Log
NormalFade
Margin
AntennaGain
DiversityGain
FeederLosses+ - - - - + + -
RxPower =
Path
Loss
Tx
Power
Antenna
Gain
Body
Losses
Penetration
Losses
Log
Normal
FadeMargin
Antenna
Gain
Diversity
Gain
Feeder
Losses+- - +-=Rx
Power - + +-
MaxPath
Loss
Tx
Power
Antenna
Gain
Body
Losses
Penetration
Losses
LogNormal
FadeMargin
Antenna
Gain
Diversity
Gain
Feeder
Losses+- - +-=Rx
Sensitivity- + +-
The Link Budget can be re-arranged such that the Maximum Allowable Path Loss can be calculated.
The first equation is that shown before in the previous examples.
The second equation is the first re-arranged to represent Path Loss.
The Third equation is the second equation, but using Rx Sensitivity = Rx Power (for 90% locations) todetermine the Maximum Path Loss.
This common re-arrangement is used to find out Maximum Path Loss, and hence Cell Range.
The Rx Sensitivity for TDMA and GSM Systems is a function of:
Noise Power Bandwidth
Receiver Noise Figure
SNR Margin above Noise+Noise Figure such that a Minimum acceptable decoded BER for
a Service (such as Coded Speech) is maintained
The SNR Margin varies for different Services, Speeds, and Environments (Typical Urban, Hilly, Bad
Urban, etc.).
Environments give rise to different Multipath Models, and hence Fading Dynamics.
The Bit/Frame Interleaving mechanisms used in GSM assist the Error Correction algorithms (by
randomising errors) in the Multipath Channel.
The Worst Case speed, Environment, and acceptable BER is usually taken resulting in the Worst Case
SNR. GSM Specifications stipulate that the Base Station Receiver should have a minimum Rx
Sensitivity of 104dBm.
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2G Uplink
BTS
Tx Power Rx Sensitivity
(TU50 or
RA130)
Mobile Tx Power = 30dBm (1 Watt) variable due to manufacturer variances
Mobile Antenna Gain = 0 dBi variable due to polarisation and near-field effects
Body Loss = 4 dB variable due to orientation, body position wrt BTSPenetration Loss = 5 dB dependent upon location (i.e building or direction of car)
LN Fade Margin = 5 dB based upon % location probability of coverage over cell area
Path Loss = ? dB The maximum path loss can be calculated to achieve % loc prob.
Base Station Antenna Gain = 18 dBi variable due to (azimuthal) antenna pattern
Base Station Diversity Gain = 4 dB variable due to extent of multipath de-correlation
Base Station Feeder Loss = 2 dB dependent upon quality and size of feeder, and length
Base Station Splitter Loss = 2 dB dependent upon how the signal is shared between multiple TRXs
Int. Degradation margin = 3 dB with interference limited design
Rx Sensitivity = -104 dBm Function of:
(residual BER = 0.2%) noise floor in 200kHz (-120 dBm)
receiver noise figure (8 dB)
Eb/No in fading environment (8 dB)
Path Loss = 30+0+(-4)+(-5)+(-5)+18+4+(-2)+(-2)+(-3)-(-104) = 135dB
Tx
Power
Antenna
Gain
Body
Losses
Penetration
Losses
Path
Loss
LogNormal
Fade
Margin
Antenna
Gain
Diversity
Gain
Feeder
Losses
In this Uplink Link Budget Example, real figures are used for a GSM/TDMA scenario.
The Slide also illustrates the variability of each Link Budget parameter.
Generally speaking parameters such as:
Tx Power
Rx Sensitivity
Antenna Gains
Feeder Losses
are very deterministic, and vary very little we dont worry too much about these.
Parameters such as:
Penetration Loss
Body Loss
Fade Margin
Diversity Gain
can be quite variable quantities, and are usually quoted as statistical limits. These can vary depending
upon specific MS-BS Orientation, diversity schemes used, etc. Penetration Loss can vary enormously
for buildings, ranging from a few dB to 25dB. This variability is seen between buildings, and the
locations variability within the buildings. Ideally the penetration margin should represent the
penetration losses experienced for a certain % of locations within all buildings, e.g. 95% rather thanan average penetration loss.
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2G Downlink
BTS
Tx power Rx Sensitivity(TU50 or
RA130)
Base Station Tx Power = 40dBm (10 Watts) variable due to manufacturer variances
Base Station Combiner Loss = 4 dB dependent upon how the TRXs are shared with common antennas
Base Station Feeder Loss = 2 dB dependent upon quality and size of feeder, and lengthBase Station Antenna Gain = 18 dBi variable due to (azimuthal) antenna pattern
Path Loss = ? dB The maximum path loss can be calculated to achieve % loc prob.
LN Fade Margin = 5 dB based upon % location probability of coverage over cell area
Penetration Loss = 5 dB dependent upon location (i.e building or direction of car)
Body Loss = 4 dB variable due to orientation, body position wrt BTS
Mobile Antenna Gain = 0 dBi variable due to polarisation and near-field effects
Int. degredation margin = 3 dB with interference limited design
Rx Sensitivity = -102 dBm Function of:(residual BER = 0.2%) noise floor in 200kHz (-120 dBm)
receiver noise figure (10 dB)
Eb/No in fading environment (8 dB)
Path Loss = 40+(-6)+18+(-5)+(-5)+(-4)+0+(-3)-(-102) = 137dB
Tx
Power
Antenna
Gain
Feeder/
CombinerLosses
Penetration
Losses
Path
Loss
LogNormal
Fade
Margin
Antenna
Gain
Body
Losses
The example shown above applies to the previous example, but for the Downlink channel.
The same variability and issues apply as discussed in the previous slide.
There is no receive diversity on the downlink for GSM/TDMA.
The Downlink can suffer a slightly greater Path Loss than the uplink channel in this example.
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2G Link Budget Calculation
Given Link Parameters, a maximum path loss can be calculated, in ourUplink example this is:
Loss = 30+0+(-4)+(-5)+(-5)+18+4+(-2)+(-2)+(-3)-(-104) = 135dB
Given environment and appropriate path loss model a maximumdistance, hence cell size can be calculated to a % locations probability
For example, assume a countryside environment with in-carpenetration and maximum path loss of 135dB, then using a stochasticmodel, such as Hata, we can work out the cell radius
Maximum Path Loss or Propagation Loss can be translated to a Maximum Cell Radius or Range when
applied to a suitable Empirical loss equation, such as a Hata Model.
A more exact range can be computed for all locations within a cells area if we were to use a Radio
Planning Tool. The Radio Planning tool computes the specific path loss for every point to the BaseStation. All points-BS links which fall below a the Maximum acceptable Path Loss means that
Coverage Service is available.
If a very accurate terrain/buildings model is used with the planning tool then the Locations Variability
can be resolved. In this case we would not need to consider the Log-Normal Fade Margin in the
Maximum Path Loss.
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2G Link Budget Calculation
100
105
110
115
120
125
130
135
140
145
150
0 2 4 6 8 10 12 14 16
Hb
= 50 m
Hm
= 1.5 m
f = 900MHz
Path Distance (km)
Pa
thLos
s(dB)
HATA Model Rural (Quasi-Open)Environment
For a 135dB Path Loss, a Cell radius of ~ 9km can be
achieved with 90% Area Locations Probability, given
system parameters in example for in-car penetration in a
rural environment
Applying the previous GSM Maximum Path Loss to a Hata Empirical Equation, for Quasi-Open Rural
Environment, a Base Station height of 50m, and a MS height of 1.5m at 900MHz, we could achieve
about 9km. This means that service is available to 9km which maintains that the Rx Power is above
the minimum Rx Sensitivity Threshold for 90% Locations at the cell edge, or at 9km.
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2G Link Budget Sensitivity
We shall examine the effect of an additional 2dB margin ona link budget
With previous example a path loss of 135dB corresponded to
a cell radius of 9km Using same path loss model and system parameters, except
that the uplink now has an extra 2dB of link margin,therefore making a maximum path loss of 137dB. What cellradius can be achieved now?
If we were to gain an extra 2dB on the Uplink Maximum Path Loss, such that it now balances with the
Downlink Maximum Path Loss, we could examine the potential new range, using Hata. This
improvement might be achieved through improved engineering of the Uplink, by the inclusion of
LNAs and/or low loss feeder at the Base Station.
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2G Link Budget Sensitivity
We can now achieve a cell radius of ~ 10.5km
increase in cell area from 9km2 to 10.5km2
increase in cell area from 254km2 to 346km2
36% Area Increase !
100
105
110
115
120
125
130
135
140
145
150
0 2 4 6 8 10 12 14 16
Hb
= 50 m
Hm
= 1.5 m
f = 900MHz
Path Distance (km)
Pa
thLos
s(dB)
HATA Model Rural (Quasi-Open)
Environment2dB
A 2dB improvement on the Uplink can give a new range of 10.5km, which means an improvement of
36% in service area.
This is quite surprising, and demonstrates the sensitivity of the Link Budget with respect to estimation
of quantities of sites. This is only 2dB. When we consider that parameters such as Penetration Margin,Body Losses, and diversity Gain have variability over many dBs we can begin to understand the
difficulty, or the limitations in the confidence a Spreadsheet Link Budget approach to site count
estimation gives.
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2G Link Budget Sensitivity
When considering network build an extra couple of dBs in alink margin can have significant effect on site numbers andultimately cost of network!
Rural (Quasi-Open) Suburban Urban Urban Indoor
Cell Area Coverage increase due to extra 2dB Link margin
The key message for ensuring that a network gets the most out of the Link Budget, is to engineer the
Links such that extra Path Loss can be tolerated. This can result in fewer sites for coverage.
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2G Link Budget Sensitivity
URBAN
0dB + 2dB
SUBURBAN
0dB + 2dB
RURAL
0dB + 2dB
Cell
Radius (km2)
Effective Cell
Area (km2)
Area to be
Covered (km2)
No. of CellSites
Relative Cost
per Cell Site (k)
Network Build
Costs(M)
0.84 0.95 1.35 1.53 6.74 7.62
1.84 2.34 4.73 6.12 118.2 153
1,500 1,500 6,000 6,000 100,000 100,000
815 641 1268 980 846 653
300 324 300 324 200 208
245 208 380 318 169 136
168M lower Network Build Costs for +2dB link margin (1997)
This slide shows a simple demonstration of how the Link Budget can improve Network Costs. The
example applies to GSM, and is purely for illustrative purposes only. The same argument would apply
to UMTS, in that extra coverage can be achieved through Link Budget improvement. Alternatively, in
UMTS this Link Budget improvement can be traded for extra capacity. This will be demonstrated
later in this section. Also see p167 in Reference [1].
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2G Link Budget Optimisation
The need to balance uplink and downlink budgets
Uplink budget usually the limiting case
Uplink can benefit from additional budget over downlinkthrough:
Minimise Feeder Losses
Base Station Receive Diversity Gain
Higher Base Station Receive Sensitivity
Use of Head Amplification and LNAs
We have shown that Link Budget improvement can be used to improve the Network Build
requirement. Generally speaking, the Uplink is the limiting case for GSM/TDMA systems. In UMTS
this is also generally the case, but the Link Imbalance can vary much more due to cell loading,
Asymmetric loading, and the level of WCDMA Interference from other cells and Mobiles connected
to other cells. We shall show this later in this section.
The Uplink can be improved through various methods. Four methods are discussed in the following
slides, and relate to Uplink Link Budget improvement through:
The use of low loss feeder at the Base Station
The use of Base Station Receive Diversity Reception
The use of better Base Station sensitivities (better Noise Figures)
The use of Low Noise Amplifiers (LNAs) at the Base Station Receiver
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2G Link Budget Optimisation - Feeder Losses
Feeder Losses can be minimised through use of:
Short Feeder Runs
Minimum number of connectors
Use of high quality and thick feeder cable
FEEDER LOSS AT 960 MHz
7.51 dB/100m 4.2 dB/100m 3.1 dB/100m
11 2 "dia"dia 7 8"dia 1 4
The slide shows examples of Feeder loss at 960MHz, I.e for GSM. The feeder losses at
1900/2100MHz would be even greater, and more reason to choose a High Quality, Thick Feeder
Cable. Such cable could only be used on Tower or Macro type base stations, and probably not on
Micro Base Stations. Micro Base Stations can incorporate the Transceiver(s) directly behind an
Antenna, rather than Antenna-Feeder-Transceiver arrangement. In this case, of course the Feeder run
losses are negligible.
Connectors inherently introduce losses, through two mechanisms
Contact losses the fact that conductivity across two surfaces is not as good as
continuous cable
Reflection losses impedance mismatches result in power reflections and hence
reduced power transfer
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2G Link Budget Optimisation Diversity Gain
Diversity Combining on Uplink path can give extra link margin
Relies on sufficiently separated receive antennas at Base Stationsuch that the transmission channels from mobile to each antenna arede-correlated in terms of fast fading
Fast Fading Signal at
Antenna 1
Fast Fading Signal atAntenna 2
Diversity CombinedSignal
D
Base Station Receive Diversity is often called Micro Diversity, since the difference in signals is due to
microscopic fading, or fast fading phase differences, which occur over a number of wavelengths.
Diversity attempts to increase the median signal strength at the receiver, by the reducing the
probability of deep fades. Section x.x.x discusses the theory, practice, and benefits offered by
Diversity in much more detail.
Diversity Gain is not really an Active Gain product, but is the difference between:
Power Level at Receiver for a Certain BER with Diversity
Power level at Receiver for same BER without Diversity
The BER is usually the minimum acceptable BER for a certain Multipath Channel (e.g. TU50). The
diversity gain represents the equivalent dB improvement had there not been any Diversity.
Diversity Gain will therefore vary depending upon BER, Multipath Channel, and Separation ofreceive antennas.
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2G Link Budget Optimisation BS Specification
GSM Specification was put togetherin 1980s
Radio Receiver Technology can now
offer Base Station ReceiverSensitivities below -110dBm
Nortel claim -117dBm
Equipment maturity and supplierissues must be considered
GSM SpecificationRx Sensitivity = -104dBm
Rx Sensitivity = -108dBm
1991
2001
GSM Vendors have recognised the importance of increasing the Link Budget. As a result there has
been huge investment in attempting to better the fundamental Receiver Sensitivity. This improvement
is by improving the Noise Figure of the receivers.
When GSM Specifications were first released the Base Station Receiver Sensitivity (for a certainBER, and Multipath channel) was 104dBm. Vendors have steadily improved on this, at almost 1dB a
year, and now below 110dBm is common.
There is a fundamental limit to this improvement. Assuming a Noise Figure of 0dB, and a Gaussian
Channel, then the fundamental Receiver Sensitivity might be
Rx Sensitivity = 10Log10(kTB) + Noise Figure (dB) + Eb/No (BER=10-3)
Rx Sensitivity = 10Log10(1.38x10-23 x 290 x 200x106) + 0dB + 5dB
= -121dBm + 0dB + 5dB
= -116dBm
Where Eb/No = 5dB represents the decoded Bit Error Rate for a Gaussian Channel (estimate).
When Nortel claim 117dBm, they are claiming the fundamental figure, in Gaussian Channel, and
possibly lower Noise Temperature.
This value is academic since it assumes no additional Noise introduced by the Receiver, and it
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2G Link Budget Optimisation MHA/LNAs
Receiver amplification provided at antenna
Receiver front-end stage is situated directly at antenna
Effect is though whole receiver at antenna therefore removingany feeder losses
Improvement is in overall receive system Noise Figure
Feeder
AntennaHead
BS Rx Feeder
AntennaHead +
LNA
BS Rx
If we remove the Feeder losses by situating the Base Station receiver directly at the Antenna, we
improve the link as discussed earlier, e.g. the micro base station receiver. However, it can be
impractical to do this for Macro or Tower Base Stations, as we would have to mount all the equipment
at say 20m on a head frame, This presents some challenges for maintenance, access, and reliability.
A Receiver is usually made up of various Amplification stages, as discussed in section x.x.x. If we
move some of the amplification stages, or at least the first stage to the antenna, we can achieve a
similar improvement in the link by essentially removing the feeder losses, whilst maintaining a
practical deployment solution.
Depending upon the Active Gain(s) of the first amplification stages, we afford improvement in the
link budget. This is shown through the Cascade Equation.
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2G Link Budget Optimisation MHA/LNAs
NF1 NF2 NFn
G1 G2 G3
NF NFNF
G
NF
G G
NF
G G GSYS
n
n= +
+
+ +
1
2
1
3
1 2 1 2 1
1 1 1...
...
Cascade Equation
NFSYS
- Overall Noise Figure of Receiving System
NFx
- Noise Figure for each Receiver stage/elementG
x- Gain of each Receiver stage/element
NF and G input as Linear quantities, not logarithmic (dBs)
The slide illustrates a number of amplification stages as might be expected in a Receiver. The Overall
Noise Figure (in linear terms) for a Receiver (NFSYS)is shown in the slide, and is a composite sum of
Gains and individual Noise Figures of the stages as shown. We can represent the Feeder in this
diagram as having less than unity Gain, and a Noise Figure equal to its attenuation. So a 3dB Feeder
Loss would have a Gain of 0.5, and a NF of 2.
The next two slides present an example of calculating the overall Noise Figure assuming no head
amplification and with head amplification, to demonstrate the improvement in the overall Noise
Figure and hence the link budget.
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2G Link Budget Optimisation MHA/LNAs
NF1=3dBNF1 = 2
NF2=6dBNF2=4
G1=-3dB
G1=0.5
NF NFNF
G
NF
G GSYS = +
+
1
2
1
3
1 2
1 1
Attenuator
Amplifier
NF3=6dBNF3=4
G2=10dB
G2=10
G3=10dB
G3=10
NFS YS = +
+
24 1
0 5
4 1
0510. . .
NFSYS = + + = =2 6 0 6 8 6. . 9 . 3 4 d B
The example shows a conventional Receiver Deployment, with Antenna-Feeder-Receiver
arrangement. In this example the Receiver has an effective Noise Figure of 9.34dB.
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2G Link Budget Optimisation MHA/LNAs
NF2=3dBNF2 = 2
NF1=6dBNF1=4
G2=-3dB
G2=0.5
NF NFNF
G
NF
G GSYS = +
+
1
2
1
3
1 2
1 1
Attenuator
Amplifier
NF3=6dBNF3=4
G1=10dB
G1=10
G3=10dB
G3=10
NFS YS = +
+
42 1
10
4 1
0510. .
NFSYS = + + = =4 0 1 0 6 4 7. . . 6 . 7 2 d B
2.6dB increase in Link
Margin over feederthen amplifier
cascade
The example shows a Receiver Deployment with Head Amplification, with Antenna-LNA-Feeder-rest
of Receiver arrangement. In this example the Receiver has an effective Noise Figure of 6.72dB, a
2.6dB increase in Link Budget.
If we moved all the receiver, i.e. all of the amplification stages to the antenna, an improvement of 3dB(=Feeder Losses) would be seen. The example above demonstrates also that the first amplification
stage has the largest influence on overall Noise Figure. Because of this, the first amplifier stage is
often the best engineered, in terms of reducing its Noise Figure, and the use of Low Noise Amplifiers
(LNAs) are typically used.
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Where Are We Now?
Introduction
Classical 2G Link Budgets
UMTS Link Budget How It Differs
Summary
NetworkDesign
OperatorsDesign Guides
The PlanningProcess
Site Placement
AntennaPlacement
FrequencyPlanning
ForwardCapacity
Planning
Polygons
LinkBudgets
UMTS Uplink Link Budget
UMTS Downlink Link Budget
UMTS Link Budget Analysis
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UMTS Radio Planning - Differences
In TDMA, the Noise (or Interference)Level with which a Mobile or Base Stationmust operate remains essentially Constant.
In TDMA, there is no real concept of avariable Processing Gain.
In WCDMA, the Interference Level variesdue to Loading of the Cell, which in turnaffects Maximum Path Loss, and hencecoverage.
In WCDMA, there are many Services, whichhave different Datarates, which give rise todifferent Processing Gains.
There are key differences between WCDMA and TDMA/FDMA link budgets
When we move from our GSM or TDMA Link Budget to UMTS we need to consider a number of
new, and variable Link Budget Parameters. The slide shows the key differences in GSM and UMTS
Link Budget Parameters.
In GSM/TDMA systems the design is Interference Limited, that is Cell Frequencies, and Time Slotsare re-used, such that a predictable, and reasonably steady state of Interference is present. The idea of
introducing an Interference Margin in the GSM Link budget (shown earlier) represents the fact that
the wanted signal in the Link Budget competes against Noise and Interference. In WCDMA or UMTS
however, the Interference levels vary much more widely since the same downlink Carrier Frequency
is re-used in every cell and the same uplink frequency by every mobile. As a result the Interference
levels in an Uplink Link Budget, for example will vary with the number of active mobiles, both in the
home cell (Intracell interference) and the number of active mobiles and their positions in other cells
(Intercell interference).
In WCDMA/UMTS many different services can be supported which demand/consume different Data
Rates, Latency, and Throughput. These result in different DS-CDMA Processing Gains, as well as
adding different Interference Levels to the Cell. These different datarates give rise to different link
budgets, and hence range. In GSM/TDMA, there is essentially a limited subset of services, i.e. EFR
Voice, Data (14.4kbps), and some GPRS data rates where error correction coding is traded for extra
datarate capacity at the expense of Receiver Sensitivity.
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UMTS Radio Planning - Differences
In TDMA, the Service is normally alwaysvoice, which dictates a certain Eb/No, andhence Rx Sensitivity, based upon a worstcase Environment, and minimum acceptable
BER is a Constant (e.g. 104dBm for GSM).
In TDMA, there is Hard Handover whichhas no influence on the Radio Link Budget.
In TDMA, there is simple slow PowerControl, which preserves Battery Life, andensures that the MS-BS AverageInterference power is kept in order.
Rx Sensitivity is a function of Eb/No, whichis dependent upon actual service type,datarate, speed, Multipath environment,diversity schemes and RAKE Receiver
Algorithms
In WCDMA, soft handover is possible, whichgives rise to Macro-Diversity Gains againstLog Normal Fading
WCDMA requires that all MS Powersreceived at the BS are equal. To achieve thisWCDMA employs fast power control tocounter Rayleigh fading. A Fast PowerControl Margin (or Headroom is needed forMobiles at the Cell Edge)
There are key differences between WCDMA and TDMA/FDMA link budgets
In GSM/TDMA a certain Bit Energy to Noise Power Density (Eb/No) is required for a certain BER
for say EFR Voice in a certain Multipath channel, which leads to a certain reference Receiver
Sensitivity. Eb/No varies with Service, and Multipath Channel. This might be a decoded EFR voice
stream at 10-3 BER, in Bad Urban 50km/h. Normally in GSM 104dBm is used as the reference
sensitivity. In UMTS Eb/No also varies as above, but is also variable with datarate, data service (10 -6
might be needed for data), specific RAKE Receiver Scheme (no. of RAKE Fingers), and effect of FastFading Power Control, which also varies with speed. This leads to a much wider spectrum of Eb/No
values relating to different environments, and services.
In GSM/TDMA there is only Hard Handover which has no influence on the Link Budget. In
UMTS/WCDMA soft handover is possible, which can afford a Macro Diversity Gain against Log
Normal Fading, over the non-soft handover case. Macro diversity reduces the influence of fading
(good for reducing stress on Error Correction/Interleaving), and allows a higher median received
signal.
Fast Fading Power Control is available in UMTS for both Uplink and Downlink. This allows the
Eb/No to be effectively reduced in a Multipath environment. However, at the cell edge the Mobile is
Power limited, and will not be able to fully negate deep fades in the channel. As a result the Eb/No at
the cell edge deteriorates. In order for the Link Budget to be consistent a Fast Fading Margin (or
Power Control Headroom) is added to represent this limitation.
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Where Are We Now?
Introduction
Classical 2G Link Budgets
UMTS Link Budget How It Differs
Summary
NetworkDesign
OperatorsDesign Guides
The PlanningProcess
Site Placement
AntennaPlacement
FrequencyPlanning
ForwardCapacity
Planning
Polygons
LinkBudgets
UMTS Uplink Link Budget
UMTS Downlink Link Budget
UMTS Link Budget Analysis
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UMTS Uplink Link Budget
BTS
Tx Parameters Rx ParametersEnvironment Parameters
Tx
Power
Antenna
Gain
Body
Losses
Penetration
Losses
Path
Loss
LogNormal
FadeMargin
Antenna
Gain
Diversity
Gain
Feeder
Losses
Rx
Power
Tx Parameters Rx ParametersEnvironment Parameters
TxPower
AntennaGain
BodyLosses
PenetrationLosses
PathLoss
Log
NormalFade
Margin
AntennaGain
DiversityGain
FeederLosses
Processing
Gain
InterCell
Int.
IntraCell
Int.
Eb/No
Target
Rx
Sensitivity
RxSensitivity
RxPower
BTS
LogNormal
FadeMargin
Soft
HandoverGain
Fast FadeMargin
If we refer back to our depiction of the Link Budget, the additional UMTS Parameters can be thought
of as influencing the Rx Sensitivity as shown.
The actual Rx Power remains the same. This can not change of course.
1. The variable Intercell and Intracell Interference quantities add to the Noise Power and limit the RxSensitivity.
2. The Processing Gain (Service Dependant) influences the equivalent Rx Sensitivity.
3. The Eb/No (Dependant upon Service, Datarate, Speed, and Multipath Channel) will influence the
equivalent Rx Sensitivity.
4. The Soft Handover Gain will reduce the Log Normal Fade Margin needed, or can be thought of in
the Link Budget as Log-Normal Fade Margin (without Soft Handover) + Soft Handover Gain. Since
the Link Budget is often used to find the Maximum Range, this must be considered (assuming a
continuum of cells).
5. The Fast Fading Margin represents the limit in Power Control for Mobiles at the Cell Edge. It
represents the deterioration in Eb/No due to not being able to adequately follow the fast fading
because of Power Limiting. Since the Link Budget is often used to find the Maximum Range, this
must be considered. The Fast Fading Margin can be considered as reducing the effective Rx
Sensitivity, or included as part of the overall Fade Margin (i.e. with Log Normal Fade Margin, and
Soft Handover Gain).
The Link Budget becomes dynamic, changing every time:
A Mobile user moves within the cell (the Interference to other cells will change)
A Mobile user in another cell moves (the Interference to the home cell will change)
A Mobile user becomes admitted/handed-off/removed to/from the home cell
A Mobile user becomes admitted/handed-off/removed to/from other cells
A Mobile user changes datarate (say for VBR Service Type)
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UMTS Uplink Link Budget
+21dBm
-110dBm
Tx
Power
Rx
Power
Rx
Sensitivity
Tx Parameters Rx ParametersEnvironment Parameters
Tx
Power
Antenna
Gain
Body
Losses
Penetration
Losses
Path
Loss
LogNormal
Fade
Margin
Antenna
Gain
Diversity
Gain
Feeder
Losses
ProcessingGain
InterCellInt.
IntraCellInt.
Eb/NoTarget
Rx
Sensitivity
Rx
PowerBTS
Log
NormalFade
Margin
SoftHandover
Gain
Fast FadeMargin
This slide simply shows the same Power Link Budget as before, but with a smaller Path Loss. The Rx
Power in this slide represents the Rx Power exceeded for 90% of locations (assuming that a Log
Normal Fade Margin for 90% Locations is added). In this example this Rx Power (90% Locations) is
well above the minimum Rx Sensitivity Threshold.
What would happen in reality is that the Mobile Tx Power would reduce in a GSM system such that
the Rx Power (90% locations) is closer to the Rx Sensitivity Threshold, thereby conserving Battery
Power, reducing unnecessary interference (to other cells), and still maintaining the Link Budget.
If one was to imagine an animation of the above slide, the Path Loss would vary, as the distance
varied, between MS and BS. Also the instantaneous Rx power level due to location variability would
dance around (obeying a Log Normal Probability distribution the shaded red parts) which would
result in Tx Power Variations, such that the Rx Power was => Rx Sensitivity Threshold.
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UMTS Uplink Link Budget
MaxPathLoss
TxPower
AntennaGain
BodyLosses
PenetrationLosses
LogNormalFadeMargin
AntennaGain
DiversityGain
FeederLosses+- - +-=
RxSensitivity- + +-
ProcessingGain
InterCellInt.
IntraCellInt.
Eb/NoTarget
ThermalNoisePower
NoiseFigure+ + + - +
LogNormalFade
Margin
SoftHandover
Gain-
Eb/No
Target
Fast
FadeMargin-
Similar to the GSM/TDMA Link Budget, the UMTS Link Budget can be re-arranging such that the
Maximum Path Loss can be calculated. We have to pay particular attention to the variable UMTS
parameters if we are to estimate Maximum Path Loss.
The key variables are:
Intracell Interference
Intercell Interference
Processing Gain
Eb/No
Fast Fading Margin.
We shall briefly look at each of these UMTS specific Link Budget variables.
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UMTS Uplink Link Budget InterCell Interference
MaxPathLoss
TxPower
AntennaGain
BodyLosses
PenetrationLosses
LogNormalFadeMargin
AntennaGain
DiversityGain
FeederLosses+- - +-=
RxSensitivity- + +-
ProcessingGain
InterCellInt.
IntraCellInt.
Eb/NoTarget
ThermalNoisePower
NoiseFigure+ + + - +
LogNormalFadeMargin
SoftHandover
Gain-
Eb/NoTarget
FastFadeMargin
-
Pj.PLjIntercell Interference =j = mobiles in other cells, PL = Path Loss, P = Power
The Intercell Interference experienced is the Sum of the received powers (at the Cell Base Station)
from all Mobiles in all other cells. This is dependant upon many factors, which include:
Position (and hence distance of Mobile to Cell) and Powers of other Mobiles in other cells
Quantity and service rates of other Mobiles in other cellsCell Antenna Downtilt
Base Station Cell Sectorisation
Macro or Micro cell (if micro then there may not be any Intercell Interference)
It will be shown that the Intercell Interference has a significant effect on the usable available Capacity
in the cell, dictated by what is called the Pole Capacity. It will be shown for example that a Micro
Cell with one Transceiver can have twice as much capacity as a Macro Cell with one Transceiver. A
Micro cell has no, or little Intercell Interference as they are normally deployed in isolation, or within
confined spaces, allowing isolation from adjacent cell Interference. The Macro Cell will normally be
in a sea of Interference from its neighbouring cells.
If we use downtilt, cell sectorisation, or careful site positioning we can minimise the Intercell
Interference experienced in the Macro Cell case. For example we would not want the situation where
high traffic demand is along the boundary of two cells (e.g. Football Ground). In this case there would
be many mobiles on high power communicating with say Cell1, this would also present a very high
Intercell Interference to Cell2 since the mobiles on Cell1 are at there nearest point to Cell2.
This prompts the question of should we use 4-sector sites or 3-sector sites?. A 4-sector site will
offer more capacity per Base Station, and possibly more other-cell isolation, but most existing sites
are geared up for 3-sector deployments and head frames.Discussion Point.
We shall look at Intercell Interference later and demonstrate that we experience at the Home Cell an
effective Noise Rise per Interfering Subscriber in other cells. This Noise Rise influences the
Intercell Interference, but not the Intracell Interference when we consider a design.
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UMTS Uplink Link Budget InterCell Interference
MaxPathLoss
TxPower
AntennaGain
BodyLosses
PenetrationLosses
LogNormalFadeMargin
AntennaGain
DiversityGain
FeederLosses+- - +-=
RxSensitivity- + +-
ProcessingGain
InterCellInt.
IntraCellInt.
Eb/NoTarget
ThermalNoisePower
NoiseFigure+ + + - +
LogNormalFadeMargin
SoftHandover
Gain-
Eb/NoTarget
FastFadeMargin
-
Pj.PLjIntracell Interference =j = mobiles in own cell, PL = Path Loss, P = Power
The Intracell Interference experienced is the the Sum of the received powers at the Base Station cell
from all mobiles within the cell. As the number of mobiles increases, or the capacity loading on the
cell increases, the Intracell Interference increases. From this statement the rate of increase would
appeargradual(first order) in nature, but as more mobiles are added each mobile has to increase its
power to overcome the increased noise rise at the Base Station cell, which in turns adds more
Interference. This produces asecond orderrate of increase, such that a theoretical infinite Interferenceis reached this is termed the Pole Capacity.
It is normal to impose a hard limit on the number of mobiles, or more precisely a hard limit on the cell
capacity, to avoid Intracell Interference rising above a certain level. This allows the range and
capacity of the cell to become more deterministic. A Cell Load of 50% means 50% of Pole Capacity,
results in an Intracell Interference of 3dB. A Cell Load of 75%, results in an Intracell Interference of
6dB. This Interference reduces Link Budget margin, and Path Loss, and hence potential range.
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UMTS Uplink Link Budget Processing Gain
A
B
A
B
Mobile 1
Mobile 2
Rx 1
Rx 2
In the UMTS Link Budget Processing Gain represents the effective improvement in power from a
wanted Signal carrying information (which has been produced by multiplying the information with a
Pseudo-Noise Scrambling code running at 3.84Mcps) to the resulting signal power of the signal
produced through decoding or decorrelation (i.e. multiplying again by the same Scrambling
sequence).
The decoding or correlation process produces a narrowband Baseband signal at the datarate of the
DPDCH Channel, from the Wideband original signal. What is happening is a process of trading
Bandwidth (Wide in the original signal, to low in the decoded signal) for Power (Low in the original
signal, High in the decoded Signal). There exists a Power-Bandwidth Conservation (rather like
conservation of Momentum in Physics), and the Processing Gain is always equal to {Chip
Rate/Information Rate}, where the Chip Rate > Information or Data Rate.
Processing Gain will vary depending upon Information Bandwidth (Service Datarate), For UMTS the
following Processing Gains are available:
Strictly speaking the WCDMA Processing Gain is equal to (Chip Rate/Channel datarate) and not
(Chip Rate/User Information Rate), since it is the DPDCH Physical Channel which receives WCDMAspreading. In fact we use the (Chip Rate/User Information Rate) to loosely define overall Processing
Gain since there is effective gain from Channel Coding, and Interleaving. Different services may
have different coding, interleaving, etc. and therefore their Processing Gains may be different for the
Service Information Rate Chip Rate Linear Log
8kbps (Voice) 3.84Mcps =3840/8 480 26.8dB12.2kbps (Voice) 3.84Mcps =3840/12.2 314 25.0dB
64kbps (LCD Data) 3.84Mcps =3840/64 60 17.8dB144kbps (LCD Data) 3.84Mcps =3840/144 26.7 14.2dB
384kbps (LCD Data) 3.84Mcps =3840/384 10 10.0dB
2Mbps(LCD Data) 3.84Mcps =3840/2000 1.92 2.8dB
Processing Gain in Rx
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UMTS Uplink Link Budget Eb/No
W = 3.84MHz
Eb
No
R x Eb
R x No
R bps 1 Hz
Linear Scaled Graphs
Frequency Domain
Noise Power = PSD x Bandwidth = 10-13 Watts
Wanted Signal Power = PSD x Bandwidth = 10 -14 Watts
Despread
Signal
Power
=
10 -14 W
Despread
Noise
Power
=
1.7.10-15 W
PowerSpectraldensity(W/Hz)=Energy(J)
RF SNR = 0.1 = -10dBDespread SNR =
RF SNR x W/R =
0.1 x 3.84/0.64 =
6 = 7.8dB
Eb/No = 7.8dB
Energy
(in 1Hz)
A Bit of Theory First - Eb/No is one of those terms which can be confusing! In UMTS we
use Ec/No and Eb/No. They are related to one another as Eb/No = Ec/No x Spreading
Factor.
The Energy in a User Information Bit (Eb) comes from the summing or Integration of theEnergies in every chip (Ec) during a bit duration through the de-spreading process in the
receiver. If we use the same Scrambling code in the receivers de-spreading process as used
in the spreading process in transmission we achieve voltage Integration on the received
signal. Noise power on the other hand is de-correlated, and in fact averages, when we sum
the chips. The voltage of the wanted signal is integrated and the noise component averages
which produces integration of Energy. Theprobability that a bit will be received in Error is
a function of the Energy in the Bit (Joules) and the Average Noise Energy (or Noise Power
Spectral Density in W/Hz = Joules). We have aprobability since actual Noise Power (even
after integration) can vary around an average value, which follows Gaussian statistics in
general. The greater the ratio of Bit Energy to Noise Energy (Eb/No) the lowerprobability that the Bit will be received in Error.
The bits in this case are the User Information Bits. In UMTS these bits undergo channel
coding with Error correction coding schemes, and as a result the Eb/No for the channel
(coded) bits can be lower than the Eb/No for the User Information bits at the expense of
reduced datarate, since channel coding adds redundancy. Generally speaking for low User
Bit error probabilities, channel coding can offer a better User Bit error probability for the
same user datarate as the uncoded channel case.
Therefore if we have a slow bit rate we have many chips per bit and hence can achieve ahigher Eb than for a higher user datarate. Hence we can state that Eb = Ec x Spreading
Factor. Alternatively we can achieve the same Eb for different user bitrates by varying Ec.
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UMTS Uplink Link Budget Eb/No
Eb/No is a parameter to definethe Energy per User InformationBit divided by the Noise PowerSpectral Density.
There are Waterfall curves to
characterise the trade-offbetween Eb/No against Bit ErrorRate (BER) for differentModulation Schemes.
SNR against BER (a more tangiblequantity) can be derived fromSNR = Eb/No x Bit Rate/NoiseBandwidth.
Eb/No is a notional quantity itcan not be directly measured.
Eb/No is theoretically
independent of datarate
Probability of Bit Error (or BER) for QPSK Modulation and
Coherent Detection at Receiver
1.00E-17
1.00E-16
1.00E-15
1.00E-14
1.00E-13
1.00E-12
1.00E-11
1.00E-10
1.00E-09
1.00E-08
1.00E-07
1.00E-06
1.00E-05
1.00E-04
1.00E-031.00E-02
1.00E-01
1.00E+00
-5 0 5 10 15 20
Eb/No
Pb(BER
Eb/No is defined as the (Energy per User Information Bit) divided by (Noise Power Spectral Density),
required to yield a specified bit error probability. Different user services require different BER or Bit
Error Probabilities. Voice may require 10-3 BER, MPEG2 Video 10-4, and FTP Data Transfer 10-6.
Ec/No is the same for a single Chip.
In order to send more information (bits) without increasing the Bandwidth, Baseband informationstreams are split into groups of say 2,3, or 4 bits and these groups are sent as different modulation
states over the radio channel. Each group of bits is a Symbol. Different modulationstates are different
Amplitude and/or Phases of the RF Signal, which can be differentiated at the Receiver. QPSK can
represent 4 changes in phase, 64QAM can represent 64 changes in Phase and Amplitude. The rate of
change of this Modulation state, represents the Symbol Rate, which dictates the RF Bandwidth
Occupancy. E.g. an RF Signal changing Phase and/or Amplitude at 3.84Million times per second will
occupy about 3.84MHz. In UMTS we dont modulate User Information bits in this way but modulate
Chips. A number of chips of course represents a User Bit depending upon the Spreading Factor used.
If we consider a QPSK modulation with state changes (or Symbol Rate) at 1MSymbols/sec. As each
Modulation State can represent two bits (or one complex bit), we effectively have a channel bitratethroughput of 2Mbps. Why dont we send more chips for every Modulation State, and get a higher
user bit rate throughput? Well we could, we would need say 8-PSK modulation to increase our
information throughput twofold. We now have less differentiatingspace between phase states, and
there is an increased risk in decoding the wrong phase state at the receiver (I.e. increase in BER). In
order to combat this we need to increase the S/N ratio. This is the fundamental trade-off of
Information Rate against S/N (related through Shannons Theorem). We would need a whole family
of graphs to represent Bit Error Rate against SNR for all bitrates. To avoid this we use the notional
parameter of Eb/No, where we assign a notional Energy for each bit, although bits are physically
transferred in blocks of 2 as a complex bit.
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The Eb/No Waterfall curvesshown previously assumes aGaussian Radio Channel. Thatis Perturbations of the Signal(due to Noise or Fading) follow
Gaussian Statistics. We can generate these
Waterfall Curves fromSimulation (Eye Diagrams, andConstellation Diagrams).
Diagram shows theConstellation Diagram forQPSK Signal with SNR=11dB(Eb/No = 8dB). This gives us aBER of about 0.0002 (1 BitError in 5000).
Demonstration
Signal
Voltage
Noise
Voltage
Example of non-WCDMA QPSK Modulated Waveform Constellation.
SNR = 11dBEb/No = 8dB
2 bits/symbol
This is not a UMTS/WCDMA Waveform
Eb/No are therefore strictly independent of Datarate, and hence there are Waterfall Curves to
Characterise the BER against Eb/No used to represent the performance at all information datarates.
We can convert between Eb/No and the more familiar SNR (S/N) for UMTS through the following
equation:
Where;
B = Bandwidth (Hz). I.e. 3.84MHz
W = Chip Rate (cps). I.e. 3.84Mcps
R = User Information Datarate (bps), e.g. 8kbps, 12.2kbps, 64kbps, 144kbps, etc.
To help illustrate why and how errors occur in a digital modulated channel we can use the ideas of a
Constellation Diagram. The Constellation diagram is a useful tool to represent the symbol decisionstates for say in QPSK Modulation. Each symbol representing bits of 00, 01, 10, and 11. The distance
from the Origin represents the Signal Strength, and the Angle represents the Phase. Alternatively the
Amplitude and Phase are shown as I and Q on the Constellation Diagram, as shown above. The
addition of Noise or Channel Fading will add another vector (Noise Power) which has Random
(Rectangular Distribution) Phase and Random (Gaussian Distributed) Amplitude.
Given enough Noise Power the a QPSK Modulated Symbol may end up nearer another QPSK state,
and be incorrectly decoded, that is we get a Symbol Error or Bit Error(s).
BR
NE
NS
o
b =RW
NE
NE
o
c
o
b =
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Diagram shows the TemporalEye Diagram for QPSK Signalwith SNR=11dB (Eb/No = 8dB).This gives us a BER of about0.0002 (1 Bit Error in 5000).
The Four Colours correspondto the 4 phase states asshown in the ConstellationDiagram.
-1.50
-1.00
-0.50
0.00
0.50
1.00
1.50
0
0.4
0.8
1.2
1.6
2
2.4
2.8
3.2
3.6
4
4.4
4.8
5.2
5.6
6
6.4
Likewise the Eye Diagram can be used to help illustrate the concept of Symbol Errors, or
Bit Errors for a Signal in Noise.
The above images represent the symbol decision boundaries for a QPSK modulated signal
perturbed by Noise which obeys Gaussian Statistics. This means that the received signal issimply the Original Transmitted QPSK signal is added to Noise (Voltage Terms). This
represents the case when we have a signal at a receiver and the power of the signal is only a
few dBs higher than the Thermal Noise Power the signal is competing with. The above
diagrams illustrate the case when we have a SNR of 8dB. Given the statistics of Noise there
exists a (albeit a low) probability that the Noise Voltage will be high, which will perturb the
QPSK signal such that a symbol is incorrectly decoded and a but error(s) results. The above
constellation represents about 11dB SNR and corresponds to a BER of 0.0002.
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Rayleigh Fading Radio Channelhas significant impact uponBER performance.
Since Fast Fading goesthrough deep fades which arein fact Phase Reversals thiscan flip a Symbols State.
Fast Fading alwaysexperiences Phase Reversals,regardless of SNR, or Eb/No.This results in the signalalways experiencing someerrors, which are irreducible.
Probability of Bit Error (or BER) for QPSK Modulation and
Coherent Detection at Receiver
1.00E-17
1.00E-16
1.00E-15
1.00E-14
1.00E-13
1.00E-12
1.00E-11
1.00E-10
1.00E-09
1.00E-08
1.00E-07
1.00E-06
1.00E-05
1.00E-04
1.00E-031.00E-02
1.00E-01
1.00E+00
-5 0 5 10 15 20
Eb/No
Pb(BER
Rayleigh Channel: Severely degraded BER
Gaussian Channel: Normal Eb/No vs. BER
Not all Transmission Channels are Gaussian. Cables, Waveguides, and Satellite-Earth
Radio links are very much Gaussian and the Waterfall Curves for BER against Eb/No
apply. However, in the mobile communications world we have a Rayleigh or Fast Fading
Transmission Channel. We dont just have perturbations in Signal Strength around an
Average Signal Power, but also rapid Phase reversals of the composite signal due to theMulti-path channel.
To help imagine the mechanism of the Fast Fading channel on the QPSK Constellation
diagram we wouldnt see small clouds of decoded constellation points around 4 phase
states. We would see the four clouds but also lots of decoded constellation points around
the origin, as the instantaneous signal has undergone a deep fade, and hence phase reversal.
As datarate slows down we can imagine a Symbol may be at one Phase State (say 45o) on
the constellation diagram, then a deep fade occurs and appears at 215o, before the next
information symbol comes along! The decoder wouldnt know what the Symbol should be.Therefore one would have a family of curves to represent different datarates. Eb/No is then
no longer independent of datarate, and we could now use SNR.
As a Fast Fading signal always encounters phase reversal (transitions through or near the
origin), increasing the SNR (or Eb/No) will not significantly improve BER. It can be shown
that for a Fast Fading Signal, and certain datarate there exists an Irreducible BER figure.
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Eb/No Waterfall Curves canbe improved by introducingError Correction Schemes atthe expense of reducedthroughput.
The effect of adding ErrorCorrection (part of theChannel Coding) is tointroduce a Knee in theWaterfall Curve such thatbeyond a certain Eb/NoErrors can not be correctedand the information collapses.
Probability of Bit Error (or BER) for QPSK Modulation and
Coherent Detection at Receiver
1.00E-17
1.00E-16
1.00E-15
1.00E-14
1.00E-13
1.00E-12
1.00E-11
1.00E-10
1.00E-09
1.00E-08
1.00E-07
1.00E-06
1.00E-05
1.00E-04
1.00E-031.00E-02
1.00E-01
1.00E+00
-5 0 5 10 15 20
Eb/No
Pb(BER
Rayleigh ChannelBER Without
Error Correction
Gaussian ChannelBER Without
Error Correction
Gaussian ChannelBER With
Error Correction
Rayleigh ChannelBER With
Error Correction
In most communication systems the Information is normally protected with various Error
Correction schemes. The introduction of Error Correction Coding will in effect add
redundancy of information, and reduce the overall throughput given a constant bit rate
channel. However, the Error correction schemes will be able to correct a certain percentage
of randomly occurring errors.
The raw Channel still encounters errors (as per the usual Waterfall Curves), but the
Information or decoded channel will appear errorless due to error correction and
information restoration.
If we add Error Correction coding the decoded channel Waterfall curve becomes more of
a two stage curve: the lower part representing complete recovery of information for Bit
Errors, up to a certain Channel Bit Error Rate, and the upper portion representing rapid
deterioration of information since there are too many errors to try and correct.
In the Fast Fading Channel we saw the Waterfall curve for Eb/No against BER reach a
point where there is Irreducible Bit Error rate performance. As long as these errors are
randomised and can be handled by Error Correction Routines then we can achieve a similar
decoded channel Waterfall curve as above.
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Typical Uplink and Downlink Eb/NoValues are shown left.
Eb/No varies with:
Up/Down Link
Datarate Channel Type (and Speed)
QoS for Service
Fast Fading Power ControlLimits
Micro-diversity schemes(such as Spatially separatedantennas)
Eb/No values are determinedthrough experiment orsimulations.
Mobility Pedestrian Mobility Pedestrian Mobility Pedestrian
8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4
LCD144 1.7 0.5 1.7 0.5 2.2 0.5
LCD384 2.0 0.7 2.7 1.4 3.0 2.2UDD64 2.0 0.7 2.7 1.4 3.0 1.2
UDD144 0.9 0.7 0.9 0.7 2.2 1.5UDD384 0.9 -0.4 0.9 -0.4 1.6 -0.2
Base Station
Eb/No
Urban Suburban Rural
Source: Nortel Networks
Mobility Pedestrian Mobility Pedestrian Mobility Pedestrian
8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4
LCD144 1.7 0.5 1.7 0.5 2.2 0.5
LCD384 1.7 0.4 1.7 0.4 2.2 1.2UDD64 1.7 0.4 1.7 0.4 2.2 0.7
UDD144 0.9 0.7 0.9 0.7 2.1 1.5UDD384 0.6 -0.7 0.6 -0.7 1.2 -0.5
Rural
Mobile Eb/No
Urban Suburban
Eb/No values are shown above for different Datarates, Services, and Speeds for the Uplink
and Downlink.
Eb/No values can only be realistically derived from simulations or trials A theoretical
approach would be too complex. In the UMTS system the Receivers are constantly takingmeasurements of BER, and adjusting the Target Eb/No such that the Service Quality (or a
certain minimum BER) is maintained. One can imagine the Target Eb/No varying as the
mobile terminal movement speeds up/down, encounters interference from another cell(s),
or changes datarate (for a variable bit rate service).
A lower Eb/No can be achieved when the mobile can effectively compensate for the Fast
Fading Radio Channel (shown in the next few slides). However, the Eb/No Target will also
ramp up as the Maximum Power is reached on the Mobile Terminal. As the Mobile reaches
Full Power, it can not effectively compensate for Fast Fading. This results in the need for a
higher Eb/No at the Cell Edge.
LCD = Low Constrained Delay data (low latency, high QoS, such as Voice, ISDN, or
Video streaming type services)
UDD = Unconstrained Delay Data (variable latency, variable QoS, such as FTP, Web
Access, email, and other non time critical services)
LCD and UDD are terms used to generally describe the Service Container. All Services can
be mapped to UDD and LCD together with QoS Targets, BER, FER, mi nimum bandwidth,
maximum bandwidth, latency, throughput, etc. Different QoS, BER, and FERs can bedesigned through use of coding, interleaving, etc.
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Generally Speaking Eb/No ishigher for:
Delay intolerant services
Lower Datarate Services
Higher Mobile Speeds
Higher Power DelaySpread Environment
Circuit Switched (LCD)services over the samedatarate PacketSwitched (UDD) services
The Uplink (only forPacket services)
Mobility Pedestrian Mobility Pedestrian Mobility Pedestrian
8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4
LCD144 1.7 0.5 1.7 0.5 2.2 0.5
LCD384 2.0 0.7 2.7 1.4 3.0 2.2UDD64 2.0 0.7 2.7 1.4 3.0 1.2
UDD144 0.9 0.7 0.9 0.7 2.2 1.5UDD384 0.9 -0.4 0.9 -0.4 1.6 -0.2
Base Station
Eb/No
Urban Suburban Rural
Source: Nortel Networks
Mobility Pedestrian Mobility Pedestrian Mobility Pedestrian
8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4
LCD144 1.7 0.5 1.7 0.5 2.2 0.5
LCD384 1.7 0.4 1.7 0.4 2.2 1.2UDD64 1.7 0.4 1.7 0.4 2.2 0.7
UDD144 0.9 0.7 0.9 0.7 2.1 1.5UDD384 0.6 -0.7 0.6 -0.7 1.2 -0.5
Rural
Mobile Eb/No
Urban Suburban
The same Eb/No values are shown above for different Datarates, Services, and Speeds for
the Uplink and Downlink. Why do we get variability in Eb/No values. The general rules are
shown in the slide and are discussed in the next few slides.
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Delay Intolerant Servicesrequire a higher Eb/No than asimilar more delay tolerantQoS service. Assuming allother factors are the same.
Not many service examplesbut could be Voice comparedto Voice Messaging Service.
Not shown on table but theabove applies for 64kbpsbeing more delay tolerant than8kbps Voice (assuming theyalso had the same QoS BER)
Mobility Pedestrian Mobility Pedestrian Mobility Pedestrian
8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4
LCD144 1.7 0.5 1.7 0.5 2.2 0.5
LCD384 2.0 0.7 2.7 1.4 3.0 2.2UDD64 2.0 0.7 2.7 1.4 3.0 1.2
UDD144 0.9 0.7 0.9 0.7 2.2 1.5UDD384 0.9 -0.4 0.9 -0.4 1.6 -0.2
Base Station
Eb/No
Urban Suburban Rural
Probability of Bit Error (or BER) for QPSK Modulation andCoherent Detection at Receiver
1.00E-17
1.00E-16
1.00E-151.00E-14
1.00E-13
1.00E-12
1.00E-11
1.00E-10
1.00E-09
1.00E-08
1.00E-07
1.00E-06
1.00E-05
1.00E-04
1.00E-03
1.00E-02
1.00E-01
1.00E+00
-5 0 5 10 15 20
Eb/No
BER With
Error Correction and80ms Interleaving
BER WithError Correction and
20ms Interleaving
Eb/No = 5dB
BER = 10 -3Eb/No = 3.5dB
BER = 10 -3
Delay intolerant services such as Conversational Voice require a higher Eb/No over a
more delay tolerant, similar datarate and QoS target service. This is because services such
as Conversational Voice can only receive a 20ms interleaving depth of data during
Physical Layer, Transport Sub-Layer processing. In fact Conversational Voice always
uses a 20ms interleaving depth. Higher interleaving depths mean greater randomisation ofbit errors and hence better Error Coding performance at the expense of greater processing
delays. A Service such as a non-conversational Voice Messaging service, or Audio
Streaming Radio Service could in principle use a deeper interleaving depth and benefit
from a lower Eb/No.
The Eb/No table does not show this but the values ringed could in principle represent a
Voice service and a 64kbps service, where the 64kbps service has the same QoS BER
target, and use a deeper interleaving depth. A 64kbps LCD service could be used to
transport an Audio Streaming service. The lower graph illustrates the Eb/No vs. BER
performance for two services, one Conversational Voice, and the other Non-Conversational Voice where the latter uses a deeper interleaving depth. The graph is
purely illustrative and the curves are not based upon any real simulations or measurements.
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Lower Datarate Servicesgenerally require a higherEb/No than a similar delaytolerance, and QoS service.Assuming all other factors arethe same.
Service examples include sayFTP using different datarates.
384kbps LCD requires higherEb/No than 144kbps. DuringTransport Formattingpuncturing is used to ratematch the 384kbps service,whereas repetition is used for
144kbps service.
Mobility Pedestrian Mobility Pedestrian Mobility Pedestrian
8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4
LCD144 1.7 0.5 1.7 0.5 2.2 0.5
LCD384 2.0 0.7 2.7 1.4 3.0 2.2UDD64 2.0 0.7 2.7 1.4 3.0 1.2
UDD144 0.9 0.7 0.9 0.7 2.2 1.5UDD384 0.9 -0.4 0.9 -0.4 1.6 -0.2
Base Station
Eb/No
Urban Suburban Rural
Probability of Bit Error (or BER) for QPSK Modulation andCoherent Detection at Receiver
1.00E-17
1.00E-16
1.00E-151.00E-14
1.00E-13
1.00E-12
1.00E-11
1.00E-10
1.00E-09
1.00E-08
1.00E-07
1.00E-06
1.00E-05
1.00E-04
1.00E-03
1.00E-02
1.00E-01
1.00E+00
-5 0 5 10 15 20
Eb/No
Pb(BER
BER with codinggain due to 64kbps
information rate
Eb/No = 3dB
BER = 10 -4
Eb/No = 2dB
BER = 10 -4
BER with greater coding
gain due to 144kbpsinformation rate
Lower datarate services such as 64kbps LCD require a higher Eb/No than similar QoS
higher datarate services (such as 144kbps LCD). This is because higher datarate services
carry more bits per Transmission Time Interval (TTI) during Physical Layer, Transport
Sub-Layer processing and as such the Error Coding can perform better when applied on a
greater bit length. There can be differences, e.g. the 384kbps uplink LCD service in theabove table needs a slightly higher Eb/No than 144kbps uplink LCD service. This is
because the Reference Physical Channels for the above tables correspond to the
Reference Transport Formats detailed in Module 3, section 3.5 (WCDMA Physical
Layer, and in 3GPP TS25.101) where puncturing is applied to the 384kbps channel during
rate matching whereas repetition is applied to the 144kbps uplink channel.
The Eb/No shows this with typical values ringed and could represent two similar services at
different datarates. The lower graph illustrates the Eb/No vs. BER performance for two
services, one FTP at 64kbps, and the other FTP at 144kbps where the latter benefits
from more efficient error correction perfromance. The graph is purely illustrative and thecurves are not based upon any real simulations or measurements.
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Services at higher mobilespeeds require a higher Eb/Nothan a the same service at alower speed.
Service examples include sayany two services at twodifferent mobile speeds.
Fast Power Control providessome equalisation of a FadingChannel seen at the BaseStation. A fading channel canbe equalised when at lowmobile speeds. Lessequalisation occurs at higher
speeds.
Mobility Pedestrian Mobility Pedestrian Mobility Pedestrian
8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4
LCD144 1.7 0.5 1.7 0.5 2.2 0.5
LCD384 2.0 0.7 2.7 1.4 3.0 2.2UDD64 2.0 0.7 2.7 1.4 3.0 1.2
UDD144 0.9 0.7 0.9 0.7 2.2 1.5UDD384 0.9 -0.4 0.9 -0.4 1.6 -0.2
Base Station
Eb/No
Urban Suburban Rural
Probability of Bit Error (or BER) for QPSK Modulation andCoherent Detection at Receiver
1.00E-17
1.00E-16
1.00E-151.00E-14
1.00E-13
1.00E-12
1.00E-11
1.00E-10
1.00E-09
1.00E-08
1.00E-07
1.00E-06
1.00E-05
1.00E-04
1.00E-03
1.00E-02
1.00E-01
1.00E+00
-5 0 5 10 15 20
Eb/No
Pb(BER
Eb/No = 6dB
BER = 10 -3Eb/No = 5dB
BER = 10 -3
Voice at 3km/h
Voice at 50km/h
Mobiles at low speeds can effectively combat the fast fading channel through fast power
control. As a result the channel appears Gaussian at the receiver and Equalisation (at least
in terms of power fluctuation) is achieved. A mobile travelling at a higher speed will not be
able to combat the fading channel as efficiently and the channel appears more Rayleigh at
the Base Station receiver. This is discussed in more detail a few slides ahead.
The Eb/No shows this with typical values ringed and could represent two similar services at
different mobile speeds. The lower graph illustrates the Eb/No vs. BER performance for
two voice services, one with a mobile travelling at 3km/h and the other with a mobile
travelling at 50km/h where the latter benefits from more efficient Fast Power Control and
equalisation of the fading channel. The graph is purely illustrative and the curves are not
based upon any real simulations or measurements.
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Services in Higher Power DelaySpread Environments require ahigher Eb/No than a the sameservice in a less dispersivechannel.
Service examples include say anytwo services in two differentMultipath environments.
In a less dispersive channel lesschip energy is spread in time.Although the Rake Receiverrecovers energy for each chipspread in time, any chip energyspread over over chip periodsintroduces inefficiencies.
Mobility Pedestrian Mobility Pedestrian Mobility Pedestrian
8k Voice 4.4 3.3 4.4 3.3 5.0 3.7LCD64 2.7 1.1 3.2 1.1 2.9 2.4
LCD144 1.7 0.5 1.7 0.5 2.2 0.5
LCD384 2.0 0.7 2.7 1.4 3.0 2.2UDD64 2.0 0.7 2.7 1.4 3.0 1.2
UDD144 0.9 0.7 0.9 0.7 2.2 1.5UDD384 0.9 -0.4 0.9 -0.4 1.6 -0.2
Base Station
Eb/No
Urban Suburban Rural
Probability of Bit Error (or BER) for QPSK Modulation andCoherent Detection at Receiver
1.00E-17
1.00E-16
1.00E-151.00E-14
1.00E-13
1.00E-12
1.00E-11
1.00E-10
1.00E-09
1.00E-08
1.00E-07
1.00E-06
1.00E-05
1.00E-04
1.00E-03
1.00E-02
1.00E-01
1.00E+00
-5 0 5 10 15 20
Eb/No
Pb(BER
Eb/No = 6dB
BER = 10 -3Eb/No = 5dB
BER = 10 -3
Voice in Sub-urban
Voice in Rural
The greater the time dispersion of the channel the more energy per chip is spread over other
chip periods (the Wideband Channel). Using a Rake receiver with a number of fingers
aligned at delayed received energy peaks attempts to recover the total energy per chip.
However, not all energy will be recovered due to the finite number of Rake Receiver
fingers. Also, any single finger and hence delayed chip sequence will in principle have alow cross-correlation with other users chip sequences, but not zero cross correlation, from
the properties of the Scrambling codes used by each user in the Uplink. As a result chip
sequence energy recovery in a Rake Receiver finger will never be perfect. If all the energy
of each chip of a users chip sequence falls within one chip period we do not need to use
Rake Receiver fingers, and as a result better receiver performance can be achieved or
alternatively seen as an improvement in Eb/No. The actual values of Eb/No in different
Multipath environments will be a function of the Rake Receiver performance and in
particular the number of Rake Receiver fingers. The Rake Receiver and number of fingers
is not specified by the 3GPP specifications and is left to the vendor to engineer. Refer also
to section 3.2 (Rake Receiver and Matched Filters) of the Course notes for further
information.
The Eb/No table shows this with typical environments leading to different dispersive
characters ringed and could represent two similar services in different environments. The
lower graph illustrates the Eb/No vs. BER performance for two voice services, one in a
Rural environment and the other in a