Low-Power High-Voltage Power Modulator for Motor Insulation Testing

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IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 44, NO. 4, JULY/AUGUST 2008 1059

Low-Power High-Voltage Power Modulatorfor Motor Insulation Testing

Yuseph Montasser, Associate Member, IEEE , Mostafa I. Marei, andShesha H. Jayaram, Fellow, IEEE

Abstract—Variable-speed pulsewidth-modulated (PWM) drivesallow for precise speed control of induction motors, as well asa high power factor and fast response characteristics, comparedwith nonelectronic speed controllers. However, due to the highswitching frequencies and the high dV/dt, there are increaseddielectric stresses in the insulation system of the motor, leading topremature failure, in high power and medium- and high-voltagemotors. Studying the degradation mechanism of these insulationsystems on an actual motor is both extremely costly and im-practical. In addition, to replicate the aging process, the samewaveform that the motor is subjected to should be applied to the

test samples. As a result, a low-power two-level high-voltage PWMinverter has been built to replicate the voltage waveforms for agingprocesses. This generator allows for testing the insulation systemsconsidering a real PWM waveform in which both the fast pulsesand the fundamental low frequency are included. The results showthat the effects of PWM waveforms cannot be entirely replicatedby a unipolar pulse generator.

Index Terms—High voltage, insulation testing, inverter-feddrive motors, power electronics, pulsewidth modulated (PWM).

I. INTRODUCTION

T HE INTRODUCTION of the first medium-voltage (MV)drive in 1983 [1] ushered in a new era in the operation

and control of MV induction motors. These new drives quicklybegan to supersede devices such as gearboxes and eddy-current

clutches. In addition, the rapid development of technology

and manufacturing processes in the electronics industry hasallowed for the development of semiconductor products, such

as switches, with ever increasing current and voltage ratings.

Currently, solid-state-based MV drives with operating voltagesof 2.3, 3.3, 4.16, 6, 7.2, and 13.8 kV are available on the

market. As such, it is possible to increase the operating voltage,

and, hence, the power ratings of these drives while keepingthe current at low levels, in order to keep the physical size

Paper MSDAD-07-68, presented at the 2006 EAS/IEEE/IEJ/SFE Joint

Conference on Electrostatics, Berkeley, CA, June 20–23, and approved forpublication in the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS bythe Electrostatic Processes Committee of the IEEE Industry ApplicationsSociety. Manuscript submitted for review September 1, 2006 and releasedfor publication December 3, 2007. Published July 23, 2008 (projected). Thiswork was supported in part by the Natural Sciences and Engineering ResearchCouncil (NSERC) of Canada.

Y. Montasser and S. H. Jayaram are with the Department of Electricaland Computer Engineering, University of Waterloo, Waterloo, ON N2L 3G1,Canada (e-mail: [email protected]).

M. I. Marei was with the University of Waterloo, Waterloo, ON N2L 3G1Canada. He is now with the Electrical Power and Machines Department,Faculty of Engineering, Ain Shams University, Cairo 11517, Egypt (e-mail:[email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TIA.2008.926234

of the components reasonably small with good thermal perfor-

mance [2]. Moreover, the newer solid-state drives are much lessprone to equipment breakdown [1] and offer significant energy

savings over their mechanical alternatives for variable torque

applications [1], [3].Despite many advantages, the application of solid-state

variable-speed pulsewidth-modulated (PWM) drives for use

with induction motors has created concerns with regard to thenegative impact that these drives have on the insulation system

of the motor [4], [5]. This is mainly due to the fact that theinsulation systems are not designed to cope with the impulse

like voltages produced by the PWM voltage source converters

(VSCs), as they have been mainly designed to operate at powerfrequency (50/60 Hz). Most of the problems that occur due to

the use of these drives result from the repetitive steep front

pulses (high dV/dt) and added harmonic content of the outputwaveforms. These problems include large overshoots at the

motor terminals [6], increased motor heating [7] which may

accelerate the thermal degradation of the insulation, and bearingcurrents [8]. The common causes of insulation degradations due

to such high frequency fast transients are as follows.

1) An increased growth rate in the material’s microcavitiesdue to the local electromechanical energy storage and

electrical fatigue with high frequency components.

2) Dielectric heating that usually occurs because of high fre-quency components. Hot spots developed on the surface

of the ground-wall insulation system, specifically in thestress grading (SG) region, can accelerate the degradation

process.

3) Space charge injection/accumulation due to steep wave-fronts, high dV/dt, and high frequency components. The

presence of such components leads to a delay (out of step)

in the polarization of some of the dipoles, and the chargesmight not disappear with polarity reversal. Space charge

can cause field perturbation between two consecutiveturns, which can lead to premature failure.

4) Partial discharge (PD) activity due to overshoots in the

voltage waveforms and space charge fields. The PD ac-

tivity can gradually destroy the SG coatings and, eventu-ally, the ground-wall insulation. Furthermore, the voltage

distribution along the SG coating is frequency (or dV/dt)dependent because of the capacitive coupling of the SG

coating with the high-voltage conductor. The voltage at

every point along the SG coating depends on dV/dt, asthis potential is defined by the ratio between the longitu-

dinal impedance of the SG coating and the impedance of

the main insulation.

0093-9994/$25.00 © 2008 IEEE



1060 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 44, NO. 4, JULY/AUGUST 2008

To suppress the problems associated with the drives, po-

tential solutions include the use of an inverter duty motor or

a filter between the motor and the converter [9]. Often, the

inverter duty motor is a standard motor with improved cooling

and added ground-wall and turn-to-turn insulation [10], [11].

This solution does not completely eliminate the aging and

degradation problems found in the insulation system. It simplylengthens the aging process before any serious problems or

failures develop. Further increasing the insulation level is not

a good solution because it reduces the thermal performance

of the motor. Instead, motor manufacturers should focus on

developing insulation systems that are much more resilient to

the operating conditions produced by VSCs.

The use of filters may be appropriate for a variety of situ-

ations; however, they are not always suitable. For example, in

propulsion applications for ships, the use of a filter negates the

weight savings obtained through the use of a PWM-VSC [12].

Filters can also be unappealing due to the fact that the resonant

frequency of the filter can potentially limit the fundamental

frequency of the output, limiting the applications of the drive

that can be used in [13].

An additional concern to motor manufacturers is the avail-

ability of switches with higher voltage ratings, which allow

drive manufacturers to achieve the same operating voltage

levels with a reduced number of stages in their designs. This

trend would reduce the complexity, as well as the number of

components required in the drive, but it may create added insu-

lation problems for motor manufacturers [14]. This is because

by reducing the number of levels utilized in the design will

simply increase the electrical stress that these drives place upon

the motor insulation.

Multilevel inverters are preferred from an insulation stand-point, as they produce much more motor friendly waveforms

compared with a standard two-level inverter. The increased

number of levels reduces the overall dV/dt in the output voltage

waveform, which occurs at the motor terminals; this, in turn,

reduces the stress on the insulation. In addition, multilevel con-

verters allow the elimination of the use of output transformers

for large induction motor applications [15].

Although significant work has been done with respect to

drive design, the effects of PWM inverters on MV insulation

systems [16]–[19] have not been investigated in depth. There

has only been a small focus on developing improved magnet

wire coatings [20], [21] to resist corona activities. Whereasthe insulation problems discussed previously can be found in

both low voltage and MV motors, the solutions developed to

suppress these problems in low voltage random-wound motors

cannot be directly applied to their MV counterparts because of

their differing constructions.

As stated previously, there are few published works on

insulation studies in the MV class. Currently, the majority

of the research in this area utilizes unipolar square waves,

exponentially decaying pulses, or high frequency ac as test

waveforms in analyzing insulation performance. Such voltage

waveforms do not expose the test sample to the full effects

that the converter output produces, namely, the fundamental

component, the high dV/dt, and the large harmonic content.As a result, a comparison between the aging effects under the

conventional test waveforms and an actual PWM test waveform

would prove to be very beneficial. To address this need, the

objective of this paper is to design a device that is capable of

producing a high-voltage bipolar PWM output suitable for use

in insulation testing. This paper showcases the laboratory built

low-power high-voltage inverter used to replicate the output

waveforms of an MV drive, as well as the studies conductedon the performance of the insulation in MV motor coils under

these waveforms. Whereas this device is based on a single-

phase inverter, it has been called a modulator because, by

swapping the firmware in the controller, the device can produce

both unipolar and bipolar pulse waveforms. The unipolar pulse

waveforms were, in fact, used in the initial testing of the device.

As a result, in this paper, the terms pulse modulator and low-

power high-voltage inverter are used interchangeably.

II. DESIGN OF THE BIPOLAR PULSE GENERATOR

As previously discussed, the voltage sources currently usedin motor insulation studies are either exponential (unipolar)

pulse or high frequency ac waveforms. A high-voltage PWM

signal can be generated by using two potential methods; a low

voltage signal could be generated from a standard inverter,

and the magnitude can be amplified by using a high-voltage

transformer. The other method involves the use of switching

devices with suitably high-voltage ratings to build an inverter

to generate the high-voltage signal.

The use of a high-voltage transformer is not feasible for a

number of reasons. Due to the magnetic limitations of the trans-

former core, the duty cycle of the input pulse must be limited

so as to avoid core saturation. This limitation on the duty cycle

means that the long duty cycles used in generating parts of the

PWM signal will not be possible. As the transformer method

not being feasible, the second method which requires the use of

high-voltage gated switching devices has been implemented in

this paper.

A. Basic Configuration

The basic topology of the pulse modulator, as shown in

Fig. 1, is based on a single-phase inverter. Because the leakage

current through the test object (insulation sample) is very small

during normal testing, a resistor is connected at the output ter-minals to allow the semiconductor switches to operate properly.

In addition, this resistor will discharge the capacitance of the

test object so that, when the switches in the inverter commutate,

the voltage across the test object will drop to zero during that

commutation time. The resistor in Fig. 1 is a variable resistor

because its value is selected, depending on the capacitance of

the test object as well as the external dc-link voltage applied to

the inverter.

The two-level inverter topology was selected, as it offers the

simplest control algorithm, as well as easy construction. As

discussed earlier, almost all VSCs use a multilevel inverter in

their designs. Whereas a two-level inverter will not be able to

produce the exact output of an MV-PWM VSC, it will replicatethe most important aspects of an MV VSC’s waveform: the fast



MONTASSER et al.: LOW-POWER HIGH-VOLTAGE POWER MODULATOR FOR MOTOR INSULATION TESTING 1061

Fig. 1. Single-phase inverter topology utilized in the bipolar pulse generator.

rise time (dV/dt), the presence of a significant fundamental

component, and the large harmonic content. It should be noted

that, for the same dc-link value, a two-level inverter will pro-

duce a higher dV/dt than a multilevel inverter. Nonetheless,

the waveforms used in this paper have the dV/dt in the range

that reflects the drive output very closely. The operating voltage

range for this power supply falls within that of the MV drives

which are between 1 and 13.8 kV. Based on the aforementioned

values, an output waveform with a magnitude of 14 kVpp witha dc-link voltage of ∼7 kV is selected as reference.

B. Component Selection

The selection of components for the proposed power modu-

lator design requires a balance between higher voltage and low

current ratings. This is due to the fact that the pulse modulator is

required to supply very low currents for insulation testing. Due

to the limitation of the two-level inverter, a switching device

with a voltage rating of 8 kV is required, which is based on

the 7-kV dc-link voltage. Currently, the only available active

switching devices on the market, which have these voltageratings, are thyratrons, thyristors, and gate turnoffs (GTOs).

Thyratrons and GTOs have very low repetition rate and will not

be able to operate at the required switching frequency. Thyris-

tors are naturally commutated, and the pulsewidth cannot be

easily controlled. As a result, a series chain of either insulated-

gate bipolar transistors (IGBTs) or MOSFETs can be utilized.

MOSFETs that are available with voltage ratings of 1200 V

come with turn-on/off times on the order of tens of nano-

seconds, but these switching times are much faster than the

operations of those switches found in an MV drive. Alterna-

tively, discrete IGBT units are available on the market with

voltage ratings of up to 1700 V and current ratings of up to

75 A. Based on these data and the need to minimize the numberof switches in series, so as to limit the size of the modulator,

Fig. 2. Measured voltage across each switch in a series chain of two switches

with resistive snubbers installed.

IGBT switches with voltage and current ratings of 1700 V and

16 A, respectively, were selected for use.

C. Snubbers

Under ideal circumstances, a series chain of switches with no

voltage sharing scheme may be acceptable, but in reality, device

variation and other factors must be considered. Hence, some

form of voltage sharing must be utilized to keep the switches

operating within their safe operating area. The most common

voltage sharing technique is the use of parallel resistors forsteady state voltage sharing. For this design, 2-MΩ shunt re-

sistors were selected by basing on the 50-µA leakage current

specified on the datasheet of the switch [22].

The initial operation of the chain of switches demonstrated

that this snubber was enough to reliably operate the series chain.

Later measurement with a differential probe demonstrated that,

whereas the steady state voltage was indeed equalized, the

transient voltage across each switch was markedly different.

This is shown in Fig. 2, which shows the voltage across each

switch in a series chain consisting of two switches. The figure

also indicates that the top switch is stressed more than the

bottom switch.To correct this problem, snubber capacitors were placed

in parallel with the switches [23]. The size of the required

capacitors was roughly determined from the collector-emitter

capacitance [22] and by testing a variety of capacitors. Based

on these criteria, 1-nF capacitors were selected for use. This

solution equalized the transient voltage but created another

problem; with the capacitors installed, the turn-off time became

much longer, distorting the pulse shape. The effect is shown

in Fig. 3, which compares the pulse shapes with and without

a snubber capacitor installed. The bend that is apparent in the

falling edge of the pulse is due to a slight delay in the gate

signal of the bottom switch, as compared with the gate signals

of the other switches. This delay was due to imperfectionsin the driver circuit, and it was later eliminated through the




Fig. 3. Comparing the turn-off time of the pulse with and without 1-nFcapacitive snubbers installed.

use of a printed circuit board with identical gate circuits forall switches.

For testing insulation systems where the loads are predomi-

nantly capacitive, the added capacitance of the snubber further

increases the turn-off time of the pulse. If the turn-off time of

the output becomes too long, this will limit the dV/dt of the

device, making it difficult to properly age the samples, as well

as influencing other applications, this device can be used for

[24] and [25]. After much testing with different snubber values

coupled with different capacitive loads connected at the output,

it was determined that it was best not to use the capacitive

snubber.

D. Controller

The pulse modulator is controlled by a PIC16F73 micro-

controller programmed with a custom firmware that can be

used to generate both a bipolar square wave and a sinusoidal

PWM waveform. The PWM output is generated by using the

built-in PWM module, as well as a lookup table stored in

the microcontroller’s Flash memory. The user can vary the

fundamental frequency by controlling the step size between the

data points of the readout of the lookup table. Fig. 4 shows

the output PWM waveform generated by the custom built

supply with a fundamental frequency of 60 Hz, a switchingfrequency of 1.25 kHz, and an amplitude of 6.2 kV peak to

peak. The superimposed sine wave is a symbolic represen-

tation of the fundamental, but the actual Fourier spectra of

the modulator output PWM voltage waveform are presented

later in Fig. 8 under the section “Output waveforms” and

analyzed.

One of the settings in the firmware, which had to be carefully

considered, was the insertion of a dead time in the output. The

dead time is the period between the conducting switches of

the inverter being turned off and the opposite switches being

turned on. This dead time is shown in Fig. 5, which highlights

a zoomed-in section of the voltage output of the modulator

in its bipolar mode. In the firmware, the dead time was setat 8 µs. The length of the dead time is particularly important

Fig. 4. Sinusoidal PWM output produced by the laboratory-built power

modulator. This waveform was captured by using two high-voltage probes,which were then subtracted by using the math function to obtain the actualwaveform.

Fig. 5. Zoomed-in output of the bipolar pulse modulator, showing the deadtime which was programmed into the firmware of the modulator.

with capacitive loads because, as discussed previously, the turn-

off time of the pulse increases with the capacitance of the

load. Turn-off times on the order of 2, 5, and even 10 µs

were observed, depending on the capacitance of the test object,

during testing with unipolar pulses.

E. Drive Circuit

One of the most important areas of the modulator is the

drive circuit. If the drive circuits are poorly designed, then

the switching performance of the IGBTs will be poor. The

gate circuit must be able to supply enough peak current so

as to turn on the switch by fully charging the input capac-

itance and putting the switch into its low impedance op-

erating state. In addition, it must simultaneously maintain

electrical isolation between the gate of the switch and thecontroller.




Fig. 6. Switching waveforms during turn-on: (Ch 1) V CE, (Ch 2) gate voltageV GE, and (Ch 4) gate current I G. These waveforms were measured at thebottom switch connected in a series chain of switches.

Because the device discussed in this paper depends on the

use and operation of series chains of switches, the electrical

isolation and synchronization of the gate signals are particularly

crucial. The synchronization of the gate signals will be achieved

through the construction of identical gate circuits for all the

switches.

To maintain the electrical isolation and generate the required

floating gate signals, several methods were investigated. One

technique discussed in the literature is the passive triggering

of series MOSFETs through capacitive coupling [26]. This

technique offers the advantages of low component count, as

well as few space requirements. However, the results presentedfor series IGBTs are not very promising [26], and therefore, this

method was not implemented.

The use of gate drive transformers was also investigated.

Like the previous method, gate drive transformers offer the

advantage of being compact. This method could not be im-

plemented due to the fact that there are no available units

which can operate at the switching frequency of the modulator,

which is between 1 and 3 kHz. Almost all gate drive units are

manufactured to operate at switching frequencies of 10 kHz and

above.

As a result of the limitations of the previous two methods,

optocouplers were selected for use. One downside of usingoptocouplers is that they each require an isolated power supply.

This means that a series chain of switches will require quite a

bit of space for the drive circuitry. The optocoupler, which was

selected, had an adequate insulating voltage but was only able

to supply a very limited current. As a result, additional circuitry

was required to supply adequate current to the gate terminal of

the switch, ensuring proper turn-on. This was shown in Fig. 6,

which shows the gate current, collector-emitter voltage, and

gate-emitter voltage waveforms of a selected IGBT during its

turn-on. During the turn-on of the switch, the recorded peak

current is around 50 mA. The optocoupler selected for use

in this design has a peak output current of 16 mA; hence, it

would increase the turn-on time for the switch because its inputcapacitance would be charged much slower, as otherwise, the

Fig. 7. Low-power high-voltage inverter which was built in the laboratory forinsulation testing prior to being packaged in a metal enclosure.

Fig. 8. Fourier spectra of the PWM waveform, which was recorded in Fig. 4.

optocoupler may be damaged from attempting to supply more

than its rated current. This is because the gate of the switch is

directly connected to the optocoupler.

III. RESULTS

A. Output Waveform

The pulse modulator that was designed and built in labora-

tory is shown in Fig. 7. After the initial testing of the modulator,

it was mounted inside a grounded metal enclosure to minimizethe electromagnetic interference emitted by the device. This

pulse modulator has been operated safely and stably at output

voltages of up to 6.2 kVpp, Fig. 4. In theory, this modulator

should be able to operate with a dc-link voltage of 6.8 kV,

which would produce an output voltage of 13.6 kVpp. Due to

packaging and safety issues with operating the device at such

high voltages, this level could not be safely reached.

The Fourier spectra of the modulator output PWM voltage

waveform were calculated, and they are shown in Fig. 8. The

plot in Fig. 8 shows the significant fundamental component

at 60 Hz. In addition, the peak at a switching frequency of

1.25 kHz is apparent, along with the peaks which occur at multi-

ples of the switching frequency. The output waveform producedby the modulator does, in fact, replicate all the features of a test




Fig. 9. Recorded (top) line-to-ground voltage and (bottom) current applied toa 4.16-kV coil, with all its turns shorted together, connected to the output of the modulator. It should be noted that the output of the modulator is floating;thus, measuring the line-to-ground output voltage only gives half the output

waveform.

voltage waveform that a practical drive output voltage would

have on the insulation system. The following sections briefly

cover the tests which were conducted, using the modulator.

B. Full 4.16-kV Coil Test

One of the initial ideas behind the modulator was to build

a test bench that is capable of aging full coils. This would

provide invaluable data on the degradation process that occurs

in MV inverter-fed induction motors. To determine whether this

would be possible with the current design, a 4.16-kV coil was

connected to the output. One connection was made to the steelplates attached to the coil, which would simulate the stator core,

and the other connection to the shorted windings in the coil.

The system was then energized, and the dc-link voltage was

slowly increased. As the voltage increased, large currents were

drawn by the test object, as shown in Fig. 9. Due to the low-

power rating of the dc source connected to the modulator, the

dc-link voltage was limited to 1.2 kVpp. In Fig. 9, the current

waveform shows that the capacitance of the coil is the dominant

component, which is apparent from the fact that there is only

current drawn when there is a transition in the voltage. When

the voltage is constant, such as during a longer pulse, the current

through the load becomes zero.

C. MV Ground-Wall Insulation Tests

The major insulation problem for large machines, operating

above 3.3 kV, is most likely in the SG material on the end

winding; however, it is essential to understand the performance

of the ground-wall insulation, when enhanced PD activity and

dielectric heating are present. Therefore, test samples with

specialized SG coatings, which can withstand the aging effects,

are used to determine the breakdown voltage of the ground-

wall insulation under different types of voltage waveforms. The

description of the laboratory that developed small stator bars

is shown in Fig. 10. The small copper bars are wound withground-wall mica tapes, as shown in Fig. 10(a), and prepared as

Fig. 10. Prepared bar samples that were used to represent a stator winding.

(a) Before any conductive paint of SG material was applied. (b) Completedbar sample with all coatings applied and all the relevant components labeled.(c) Experimental setup used to age the bar samples.

per manufacturer guidelines. The bar samples were first vacuum

dried for approximately 6–8 h. Following the vacuum cycle,

the specimens were flooded with a low viscosity resin. The wet

stator bars were removed from the tank and placed in an oven

to cure the resin at 120 C–150 C for 2–3 h. The tapes used for

vacuum pressure impregnation are mica paper tapes reinforced

with thin glass cloth on one side. They are made with 10% to

25% of the binder resin needed to fully saturate the insulation,

which is sufficient to bond the layers of the tape together during

overlapping. The binder resin is of the same resin family as thefinal impregnating resin. Silicon carbide loaded coatings in the

form of paints are applied to the stator bar specimens, along

with SG, to limit erosion from PDs, particularly due to the edge

effects. A completed sample and the experimental setup used in

aging are shown in Fig. 10(b) and (c), respectively.

After the samples had been aged under pulse application for

a set period of time, their dc breakdown strength was measured.

Three samples are used in each test condition. In all cases, the

tests were initiated with virgin samples. In this research paper,

the breakdown value is measured to establish the residual life

of the sample instead of measuring the time to failure. The

reported values of the breakdown voltages are average of threemeasurements. The deviation in the data was observed to be

in the range of 0.41–0.55 kVp in all cases. Based on the small

scatter in the data and also due to the limited number of data

points, no statistical analysis was applied.

All of the breakdowns occurred in the aged samples due to

the puncture in the mica tape. The results were then compared

with the results of other aging tests, using different test wave-

forms. The comparison in Fig. 11 shows that samples aged

under a PWM waveform had a significantly lower breakdown

voltage as compared with samples aged by using unipolar

square wave and unipolar exponentially decaying waveforms.

The mechanism behind the rupturing of ground-wall in-

sulation was different under pulse and ac conditions. In thecase of samples subjected to unipolar pulse, and PWM-VSC




Fig. 11. Recorded breakdown strength of ground-wall samples aged undervarious test waveforms.

waveforms, the degradation is observed between two mica

layers due to PD activity, which propagates in axial direction

with the formation of tree channel, thus leading to insulation

breakdown. However, the degradation process under ac wave-

forms started in defects or voids if present in the ground-wall

insulation, then causing a local breakdown due to increased

PD activity. Furthermore, under PWM conditions, the electric

field concentrates right at the simulated slot exit [17], which

can accelerate the aging and result in an eventual failure of

the affected area. This aspect coupled with the large amount

of heating produced in the insulation by the fundamental com-

ponent could explain the lower breakdown voltage (decreasedresidual life).

D. SG System Test

SG coatings are applied to the coil ends of electrical ma-

chines in order to prevent PD damage that can occur with high

concentrations of the electric field [27]. The SG system utilizes

materials that have field dependent conductivities. Hence, as

the field concentration increases, the conductivity increases,

causing the electric field to spread out more. Current SG

systems have been designed to operate under power frequency

conditions (50/60 Hz). As a result, their performance rapidlydegrades under pulse conditions, which occurs when the motor

is fed by a PWM-VSC. This problem has been verified through

simulation and experimental work in [17].

To correct the aforementioned problem, a new SG coating

was developed, which would be able to operate under inverter-

fed conditions. The new SG coating was tested by using a

circular geometry because it produces an electric field in the

sample that is very similar to that found at the slot exit, and it

is easy to control the thickness of each layer when making the

sample. To determine whether any electrical stress was being

generated from the applied waveform, a thermal camera was

used to observe any hot spots. Under PWM, two hot regions

(rings) are observed on the SG coating, as shown in Fig. 12, dueto the SG coating grading the high-frequency components to the

Fig. 12. Testing the new SG system with an actual sinusoidal PWM waveformto show that the system is able to grade both the 60-Hz ac and pulse electricfields.

inside interface and the 60-Hz component to the outer interface.

The verification of whether this design would actually work was

only possible by using the laboratory-built power modulator.

Thus, it is recommended that, in voltage endurance tests, a real

PWM voltage source should be used; this way, both coatings

will be under stress and, at the same time, making it possible

for a correct evaluation of the system.

IV. CONCLUSION

This paper has attempted to show the reasoning behind

developing a high-voltage low-power modulator for testing

insulation systems. Some of the important considerations in the

design of this pulser, such as the topology, the controller, and

the switches used, have been reviewed and discussed.

The modulator that was constructed has been tested with avariety of applications, and the results from these tests have

demonstrated a potential need for this type of device. For

example, the effectiveness of a new SG system under a real

PWM waveform was demonstrated.

Whereas this modulator was designed with insulation testing

specifically in mind, it could very easily be used in other pulse

power applications. This is particularly true for areas where

the used loads are much more conductive, as in liquid food

sterilization using pulsed electric fields. A bipolar voltage will

enhance microbial inactivation due to sudden polarity reversal.

This paper also shows that the idea has proven to be feasible

and achievable with regular off-the-shelf components; hence,the next step is to scale up the device so that it can be used

for reliably aging full coils. This scaling up will entail, using

a higher power HV-DC source, switches with higher current

ratings, as well as improved packaging and shielding. The

availability of such power supply could provide valuable data

and information on the degradation mechanisms which occur

within inverter-fed motors.

ACKNOWLEDGMENT

The authors would like to thank F. Espino-Cortes and

S. Ul-Haq for providing the insulation samples which were

tested by using this low-power high-voltage pulse generator, aswell as their valuable input and suggestions.




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Yuseph Montasser (S’06–A’06) received the B.Sc.degree in electrical engineering from the Universityof Alberta, Edmonton, AB, Canada, in 2004. He iscurrently working toward the M.A.Sc. degree in theDepartment of Electrical and Computer Engineering,University of Waterloo, Waterloo, ON, Canada.

Mostafa I. Marei was born in Alexandria, Egypt, onJune 17, 1975. He received the B.Sc. and M.Sc. de-grees in electrical engineering from Ain Shams Uni-versity, Cairo, Egypt, in 1997 and 2000, respectively,and the Ph.D. degree in electrical engineering fromthe University of Waterloo, Waterloo, ON, Canada,in 2004.

He is currently an Assistant Professor with theDepartment of Electric Power and Machines, AinShams University. His research interests includepower electronics, hybrid electric vehicles, custom

power supplies, artificial intelligence applications in power systems, digital-control-based microcontrollers and digital signal processors, power quality, anddistributed generation.

Shesha H. Jayaram (M’87–SM’97–F’08) receivedthe B.A.Sc. degree in electrical engineering fromBangalore University, Bangalore, India, in 1980, theM.A.Sc. degree in high-voltage engineering fromthe Indian Institute of Science, Bangalore, in 1983,and the Ph.D. degree in electrical engineering fromthe University of Waterloo, Waterloo, ON, Canada,in 1990.

She is a Professor with the Department of Elec-trical and Computer Engineering, University of Waterloo, and an Adjunct Professor with the Univer-

sity of Western Ontario, London, ON. Her research interests are developingdiagnostics to analyze insulating materials, industrial applications of high-voltage engineering, and applied electrostatics.

Prof. Jayaram has been an active member of the IEEE Dielectric andElectrical Insulation Society and the Electrostatic Processes Committee (EPC)of the IEEE Industry Applications Society. In both, she has contributed as aBoard Member, Chair of EPC during 1998–1999, Session Organizer/Chair,and a member of the Paper Review Process Committee. She is a RegisteredProfessional Engineer in the Province of Ontario.

Low-Power High-Voltage Power Modulator for Motor Insulation Testing

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