Linear Thecnology Rewue

OCTOBER 1994 VOLUME IV NUMBER 3

The LT1372 500kHzSwitching Regulator

by Bob Essaff

IN THIS ISSUE . . .

COVER ARTICLEThe LT1372 500kHzSwitching Regulator ....... 1Bob Essaff

Editor's Page ................... 2Richard Markell

LTC in the News............. 2

DESIGN FEATURESPCMCIA Card SocketPower Management ......... 3Doug La Porte

An Ultra-Selective BandpassFilter with Adjustable Gain........................................ 6Philip Karantzalis

New LT1182/1183 DualCCFL/LCD ContrastSwitching Regulator ....... 8Anthony Bonte

The LT1366 Family:Precision, Rail-to-RailBipolar Amplifiers ......... 13William Jett and Sean Gold

LTC1325 BatteryManagement System OffersUnparalleled Flexibility..17Anthony Ng, Teo Yang Long, and Robert Reay

DESIGN INFORMATIONSimple Thermal Analysis —A Real Cool Subject for LTCRegulators..................... 23Alan Rich

DESIGN IDEAS................................. 25–29(complete list on page 25)

New Device Cameos ....... 30

Design Tools .................. 32

Sales Offices ................. 32

IntroductionThe LT1372 is a 500kHz, constant-

frequency, current-mode bipolarswitching regulator with an on-chippower switch. The LT1372 is similarto the popular LT1172 but offershigher efficiency, and, due to the in-creased switching frequency, usessmaller external components, whichsave on board space, weight, andcost. An entire DC-to-DC converterbased on the LT1372 consumes only0.5 square inches. Packaged in an 8-lead SOIC or DIP, it operates over a2.5V to 35V input supply range andits current-limited switch can carryup to 1.5A with 0.45Ω on resistance.

New design techniques increasethe LT1372’s flexibility while main-taining ease of use. Switching is easilysynchronized to an external logic-level source. A logic low on theshutdown pin reduces supply cur-rent to 15µA. Unique error-amplifier

circuitry can regulate positive or nega-tive output voltages while maintainingsimple frequency-compensation tech-niques. Nonlinear error-amplifiertransconductance reduces outputovershoot on start-up or overload re-covery. Oscillator frequency shiftingprotects external components duringoverload conditions.

The LT1372 uses a new, high-speedprocess that is tailored to high-fre-quency switching regulators. Thoughslow when compared with state-of-the-art digital technologies, thisprocess is a compromise betweenspeed and breakdown voltage, whichis important for switching regulatordesign. Smaller device geometries al-low for faster circuits at loweroperating currents, increased circuitcomplexity, and lower parasitic resis-tance for a given die area.

continued on page 20

Figure 1. LT1372 Block diagram

–

+ –

+

–

+

1372_1.eps

1.24V REF

GND SENSE GND

0.1Ω

SWITCH

5:1 FREQUENCY SHIFT

SHUTDOWN DELAY & RESET

500kHz OSC

SYNC

SD/SYNC

NFB

FB

50k

LOW DROPOUT 2.3V REG

ANTI-SAT & DRIVE BOOST

SW

NFB100k

EA IA

AV ≈ 6VC

VIN

LOGIC DRIVER

COMP

and LTC are registered trademarks and LT is a trademark of Linear Technology Corporation.

LINEAR TECHNOLOGY LINEAR TECHNOLOGY LINEAR TECHNOLOGY

2 Linear Technology Magazine • October 1994

EDITOR'S PAGE

On Growing and GrowthAnother feature article examines the

LTC1325 battery management systemIC. This part offers the designer a com-plete solution to the problem of charginga wide variety of batteries, includingNiMH, NiCad, lead-acid, and otherbattery chemistries. The part allowseasy communications to a micropro-cessor by means of a four-wire interface.LT brings the future of switching regu-lators to the forefront by introducingthe new LT1372 constant-frequency,current-mode switching regulator.The LT1372 saves board space, weight,and cost by combining a 500kHzswitching frequency with an on-chippower switch (0.45 ohms) that cancarry 1.5 amps.

Linear Technology has entered thePCMCIA market with a variety of so-lutions. The LT1312 and LT1313 aremicropower, single and dual 120mAPCMCIA VPP regulators/drivers topower and protect the VPP pins ofPCMCIA slots. Also featured in thearticle on PCMCIA are the LTC1314and LTC1315 single and dual VPPswitching-matrix ICs with internalVCC N-channel gate drivers. TheLTC1164-8 merits a feature article asit is the first completely integrated,high-Q, bandpass filter offered by Lin-ear Technology. The part requires onlyan external operational amplifier andtwo resistors to implement very selec-tive narrow bandpass filters with gainsup to one thousand.

We also present some timely infor-mation on thermal analysis. Thearticle “Simple Thermal Analysis—AReal Cool Subject for LTC Regulators”gives thermal information and meth-ods of calculating heat dissipation,heat sinking, and temperature rise.

The Design Ideas section is full, asusual. The LT1585 regulator is fea-tured in a circuit developed to powerthe new PentiumTM class of micropro-cessor. A novel circuit detailed in thissection directly senses the negativeoutput in a positive-to-negative con-verter. Additionally, we have manyother new design ideas.

Every year, things grow at theirown pace: the beans grow up thetrellis in August and the peppers andtomatoes ripen in September; theGreat Pumpkin may not be readyuntil October.

Linear Technology is different. Theapplications group has grown aboutsixfold in only six years. The IC-design group has grown by a factor ofmore than three. We’ve got stacks andpiles of scratched notes on customerproblems on each applicationsengineer’s desk. Each and every de-signer is scratching his or her headtrying to pack the most functionalityinto the smallest die size. There arenow design centers in Boston andSingapore, in addition to the corpo-rate headquarters in Milpitas. Analogcircuit designers are difficult to findanywhere in the world, let alone whereyou’d like to find them.

Growing a company is not likegrowing a single Jwala pepper, butmore like growing the whole crop forsalsa. Design and applications needfertilization (new parts) and new ideasto produce growth; more fabricationfacilities must be built, packagingneeds to be done, and these facilitiesmust be provided on a worldwide ba-sis in order to attract the type oftalent required to grow the businessin the direction that we’d all like.

This issue is filled with morsels forthe engineering palate. The LT1366family is our first series of dual andquad bipolar operational amplifiersthat combine rail-to-rail input andoutput specifications with precisionoffset-voltage specifications. This fam-ily maintains precision specificationsover a wide range of operating condi-tions and capacitive loads. A featurearticle highlights (and backlights) theLT1182 and LT1183 dual CCFL/LCDcontrast switching regulators. Theseparts are dual switching regulatorsthat drive both the cold cathode fluo-rescent lamp and the liquid crystaldisplay panels in portable computersand hand-held instruments.

LTC in the News...

Linear Technology’s AnnualSales Cross $200 Million

Thanks to the support of all ofyou who rely on the quality andperformance of Linear Technology’sproducts, the company’s annualsales crossed the $200 million markin fiscal 1994. Net sales for thefiscal year ended July 3, 1994 werea record $200,538,000, an increaseof 33% over the previous year. TheCompany also reported record netincome for the year of $56,827,000or $1.51 per share, an increase of56% over the $36,435,000 reportedfor fiscal 1993.

Linear Technology has added LeoT. McCarthy and Richard M. Moleyto its Board of Directors. Mr.McCarthy and Mr. Moley fill exist-ing vacancies on the board.McCarthy is currently LieutenantGovernor of the State of Californiaand has announced plans to retirefrom office at the end of 1994. Rich-ard M. Moley is Chairman of theBoard, President, and CEO ofStrataCom, Inc., a telecommuni-cations company.

In its June 21 issue, FinancialWorld magazine included LinearTechnology on its list of “America’s50 Best Mid-Cap Companies,” rank-ing the company 21st.

In a very exclusive selection,WORTH magazine picked LinearTechnology for its list of “50 NewBlue Chip Stocks.” More than athousand companies were consid-ered before the final selection wasmade. The only other semiconduc-tor company to make the list wasIntel.

LTC also made another presti-gious current list: Financial World’s“America’s Best 200 Growth Com-panies.” We made this list last year,too, but this year LTC was 13thcompared to 22nd on the list in1993.

In June the Entrepreneur of theYear Institute named Linear Tech-nology President and CEO RobertH. Swanson, Jr. “Northern Califor-nia Entrepreneur of the Year inElectronics.”

by Richard Markell

Pentium is a trademark of Intel Corporaton

Linear Technology Magazine • October 1994 3

PCMCIA Card SocketPower ManagementIntroduction

A variety of products are designedwith sockets to accept PCMCIA cards,such as extended memories, fax mo-dems, network interfaces, wirelesscommunicators, and a wide assort-ment of other devices. These productsinclude personal digital assistants(PDAs), bar-code readers, palmtopcomputers, and portable medicalequipment. The Personal ComputerMemory Card International Asso-ciation (PCMCIA) has releasedspecifications that outline the gen-eral power requirements for thesecards. The specification calls foran unusual amount of voltageswitching.

Host power delivery to the PC cardsocket flows through two paths: themain VCC supply pins and the VPPprogramming pins. Both supplies areswitchable to different voltages toaccommodate a wide range of cardtypes. The VCC supply is the mainsupply and must be capable of pro-viding up to 1A at either 3.3V or 5V, aswell as a high impedance state. The1A rating is an absolute maximumderived from the contact rating of500mA per pin for both VCC pins. Oneof the most stringent actual require-ments is during hard disk drivespin-up. Present hard drives require5V at 500–600mA for a short dura-

Table 1. LT1312 truth table

EN0 EN1 SENSE VPPOUT VALID

0 0 X 0V 11 0 X 12V 00 1 3V–3.6V 3.3V 10 1 4.5V–5.5V 5V 11 1 X Hi-Z 1X = Don’t care

Figure 1. Typical single-socket power-management system

tion during spin-up. Current drawdrops to 300–420mA during read andwrite operations. The VPP supply mustsource 12V at up to 120mA, 3.3V or5V at lesser currents, 0V, and realizea high impedance state. The VPP sup-ply is intended solely for flash-memoryprogramming. The 120mA currentrequirement allows erasing two flashdevices and writing to two devicessimultaneously. These diverse, spe-cific requirements call for specializedsolutions.

There are two general approachesto providing VPP power to PCMCIAcard sockets: linear driver/regula-tors and a VPP switch matrix. Thelinear regulator approach utilizes aregulator that incorporates resistorsand switches to program the outputvoltage to the appropriate level. Thissolution requires a voltage of about13V or higher. At first this may seemundesirable but, as will be shownlater, when designed as part of a por-table system’s power supply from thebeginning, this higher voltage is de-rived at little cost. The linear regulatoralso provides current and thermallimiting for short-circuit protection.The switch-matrix approach consistsof several MOSFET switches arrangedto route the input voltages to the VPPpins. This approach requires that all

voltages be present at the matrix in-puts. The switch approach has theadvantages of simplicity and effi-ciency, but lacks built-in currentlimiting and short-circuit protection.

LT1312: Linear VPP Driver/Regulator

Figure 1 shows a space-efficient,cost-effective power-managementsolution for PCMCIA sockets. Powerto the two VPP pins, usually tiedtogether to reduce cost, is providedby the LT1312 single VPP driver/regulator.

The LT1312 is a linear regulatorwith programmable output, designedspecifically for PCMCIA VPP driveapplications. Figure 2 shows the cir-cuit block diagram. The LT1312 takesa raw, unregulated 13–20V inputsupply and produces a clean, regu-lated, selectable output voltage inconformance with the PCMCIA stan-dard. The VPP pins are programmableto provide either 0V, 3.3V, 5V, 12V, ora high-impedance state. Two enableinputs (EN0 and EN1) and the VCCsense inputs select among these fivestates, as shown in Table 1. When aVCC voltage is selected, a comparatorin the LT1312 automatically switchesthe VPPOUT pin to 5V or 3.3V, depend-ing on the voltage present at theSENSE pin. The SENSE pin connectsto the PCMCIA socket VCC pin.

A second comparator monitors theoutput voltage when the VPP pin re-quires 12V. This comparator drives

DESIGN FEATURES

VS

13V TO 20V

EN0

51k

1µF

10µF

VPPOUT VPP1

VPP2PCMCIA CARD

SOCKETVCC

LT1312EN1

VALID SENSE

GND

Q1 1/2 Si9956DY

Q2

Q3

Si9956DY

VS

5V

3.3V PCMCIA_1.eps

IN1 OUT1LTC1165

PCMCIA CONTROLLER

IN2 OUT2

IN3 OUT3

GND

+

+

VCC

VLOGIC

A_VPP_PGM

A_VPP_VCC

A_VCC_5

A_VCC_3

A_VPP_VALID

by Doug La Porte


A dual VPP linear regulator, theLT1313, is also available.

Three N-channel MOSFETs, drivenby an LTC1165 triple inverting MOS-FET gate driver, provide VCC pin powerswitching. P-channel MOSFETs canbe used in place of the driver and N-channel parts, at the expense ofsignificantly higher on-resistance.This is of greater significance as VCCgoes lower, as in the case of the two3.3V switches. These very low-thresh-old P-channel parts are also muchmore costly and more difficult toobtain than the more common N-channel parts. The LTC1165 providesa natural break-before-make actionand smooth transitions due to theasymmetrical turn-on and turn-off

the open collector VALID pin low whenVPP is greater than 11V. ManyPCMCIA controllers require this sig-nal for verification that flash memoryprogramming can proceed safely.When switching the VPP signal to12V, the output is clean and of mod-erate rise time to prevent voltageovershoot and subsequent damage toflash memory parts.

The LT1312 also has a 200mAnominal output current capabilitywith a 250mA short-circuit currentlimit and thermal shutdown. Theseprotection features can be very im-portant when considering the overallreliability and robustness of the mainsystem. An additional feature of theLT1312 is its low 25µA quiescent cur-rent in 0V or high-impedance modes.

Figure 2. LT1312 Block diagram

Figure 3. Deriving 14V power from a 3.3V auxiliary winding

DESIGN FEATURES

–

+

LOW DROPOUT LINEAR

REGULATOR

VOLTAGE CONTROL

LOGIC

11V

–

+

4V

VPPOUTVS

EN1EN0

VCC SENSE

PCMCIA_2.eps

VALID

+

+

++

++

PCMCIA_3.eps

VIN

5.5V TO 14.5V

Q1

Q2

C1 22µF

C2 1000pF

C6 100µF

= AVX 22µF, 25V TPSD226M025R0200 = AVX 100µF, 10V TPSD107M010R0100 = MBRS140 = MBRS1100

C1,C3, C5

C6, C7

D1 D2

= Si9430DY = Si9410DY = DALE LPE-6582-A086 DALE (605) 665-9301

Q1 Q2 T1

C7 100µF

HC86

C3 22µF

Q3 VN7002

R2 100Ω

R3 12k

R1 100Ω

R4 22Ω

C4 1000pF

D1 D3 18V

14V AUXILIARY SUPPLY

3.3V OUTPUT

T1 22µH 3.38T

D2C5 22µF

1µF

TO SOCKET VPP PINS

FROM SOCKET VCC PINS

PDRIVE

NDRIVE

SENSE+

LTC1148-3.3 OR

LTC1142 (3.3V REG)

VPPOUT

SENSE

EN0

EN1

VALIDGND

VS

LT1312PCMCIA

CONTROLLER

SENSE–

R5 0.040Ω

of the MOSFET’s. A noninverting ver-sion, the LTC1163, is available forcontrollers with high asserted logic.

Auxiliary WindingPower Supplies for Usewith the LT1312

Because the LT1312 provides ex-cellent output regulation, the inputvoltage may come from a loosely regu-lated source. One convenient andeconomic source of power is an aux-iliary winding on the main switchingregulator inductor of the system powersupply.

LTC1142 or LTC1148-3.3Auxiliary Winding PowerSupply for Low InputVoltages

An auxiliary winding to the 3.3Vinductor of an LTC1142 or LTC1148-3.3-based 3.3V power supply createsa loosely regulated 14V power sup-ply, as shown in Figure 3. Diode D2rectifies the 11V output from thisadditional winding, adding it to themain 3.3V output. (Note the phasingof the auxiliary winding, as shown inthe figure.) Referencing the auxiliarywinding to the main 3.3V output pro-vides DC current feedback from theauxiliary supply to the main 3.3Vsection. Returning the lead of C5 tothe 3.3V output as shown improvesthe AC transient response.

A TTL logic high on the enable line(EN0) activates the 12V output. Thiswill force the 3.3V section of theLTC1142 (or LTC1148-3.3) into thecontinuous mode of operation. A re-sistor divider, composed of R2, R3,and switch Q3, forces an offset tocounteract the internal offset at the−SENSE input of the part. BurstModeTM operation ceases when thisexternal offset cancels the device’sbuilt-in 25mV offset, forcing theswitching regulator into continuous-mode operation. (See the LTC1142and LTC1148 data sheets for furtherdetail.) In this mode, the LT1312 out-put can be loaded without regard toloading on the 3.3V output of theregulator.Burst Mode is a trademark of Linear TechnologyCorporaton


Table 2. LTC1314 truth table

EN0 EN1 VCC0 VCC1 VPPOUT DRV3 DRV5

0 0 X X GND X X0 1 X X VCC IN X X1 0 X X VPPIN X X1 1 X X Hi-Z X XX X 1 0 X 1 0X X 0 1 X 0 1X X 0 0 X 0 0X X 1 1 X 0 0X = Don’t care

LTC1314: VCC switch driverand VPP switch matrix

Figure 4 shows another approachthat is very space and power efficient.Here the LTC1314 PCMCIA switchmatrix, used in conjunction with theLT1301 DC-DC converter, providescomplete power management for aPCMCIA card slot. The LTC1314 andLT1301 combination provide a highlyefficient, minimal parts count solu-tion. This circuit is especially usefulfor designers who are adding PCMCIAsockets to existing systems that cur-rently have 5V or 3.3V available. Table2 shows the truth table for theLTC1314. A dual version of this part,the LTC1315, is available in the 24-lead SSOP package.

The LTC1314 directly drives threeinexpensive N-channel MOSFETs pro-viding VCC pin power switching. Anon-chip charge pump provides thenecessary voltage to fully enhancethe MOSFETs. With the built in chargepump, the MOSFET drive is availablewithout the need for a constant 12Vsupply. The LT1301 switching regu-lator is normally in shutdown modeand consumes only 10µA, exceptduring the brief flash memory-pro-gramming cycles where VPP0 is set to12V. This method results in low on-

resistance VCC switching while main-taining maximum efficiency.

The VPP switching is accomplishedby a combination of the LTC1314 andLT1301 DC/DC converter. As statedabove, the LT1301 DC/DC converteris in shutdown mode to conservepower until the VPP pins request 12V.When VPP pins require 12V,the LT1301 is activated and theLTC1314’s internal switches routethe VPPIN pin to the VPPOUT pin. TheLT1301 is capable of delivering 12Vat 120mA while maintaining highefficiency. The LTC1314’s break-before-make and slope controlledswitching will control the output volt-age transition to be smooth, of

moderate slope and without over-shoot. This is critical for flash memoryproducts to prevent damaging partsfrom overshoot and ringing exceedingthe 14V part limit.

The LTC1314 with theLT1121 Linear Regulator

Often, systems have supply volt-ages greater than 12V available. TheLTC1314, in conjunction with theLT1121 linear regulator, supplies thePC card socket with all necessaryvoltages. Figure 5 shows this circuit,where Q1 is used to invert the shut-down logic signal. Any spare TTLinverter can also be used. The auxil-iary winding technique shown earlier

Figure 5. LTC1314 with the LT1121 linear regulator


DESIGN FEATURES

Figure 4. LTC1314 switch matrix with the LT1301 boost regulator

VDD

VPPOUT VPP1

VPP2

SHDN VPPIN

VCC(IN)

LTC1314

PC CARD SOCKET

EN0

EN1 DRV5

DRV3

GND

VCC0 VCC

PCMCIA_4.eps

VCC1

+

5V

3.3V

10µFQ2A

Q2B

0.1µF

Q1A 1/2 Si9956DY

Si9956DY

5V

VCC SWSELECT SENSE

LT1301

L1 22µH

D1 MBRS130LT3

SHDN NCILIM

PGND GND

C2 33µF

+C1 47µF

+

L1 = SUMIDA CD75-220K C1 = AVX TPSD476M016R0150 C2 = AVX TPSD336M020R0200 SUMIDA (708) 956-0666

PCMCIA CONTROLLER

PCMCIA_5.eps

VDD

VPPOUT VPP1

VPP2

SHDN VPPIN2N7002

VCC(IN)

LTC1314

PC CARD SOCKET

EN0

EN1 DRV5

DRV3

GND

VCC0 VCC

VCC1

+

5V

3.3V

10µFQ2A

Q2B

0.1µF

Q1A 1/2 Si9956DY

Si9956DY

5V

VIN VOUT

LT1121

SHDN

56.2k

121k

ILIM

PGND GND

1µF+

200pF10µF

100k

5V

13V TO 20V (MAY BE FROM

AUXILIARY WINDING)

+

PCMCIA CONTROLLER


An Ultra-Selective Bandpass Filter withAdjustable Gain

tions. One signal-detection applica-tion occurs when two signals are veryclosely spaced in the frequency spec-trum and only one of the signals hasuseful information. The LTC1164-8can extract the signal of interest andsuppress its unwanted neighbor. Forexample, a small 1kHz, 10mVRMS sig-nal is combined with an unwanted950Hz, 40mVRMS signal. The two sig-nals differ in frequency by only five

center frequency. A capacitor, CF,across resistor RF reduces clockfeedthrough and provides a smoothsine-wave output.

Signal Detectionin a Hostile Environment

An outstanding feature of theLTC1164-8 is its ultra-selectivity. Abandpass filter with ultra-selectivityis ideal for signal detection applica-

Figure 2. LTC1164-8 ultra-narrow, 1kHz bandpass filter with gain (gain = 340k/RIN, 1/2πRFCF =10 fCENTER).

IntroductionThe LTC1164-8 is a monolithic,

ultra-selective, eighth order, ellipticbandpass filter. The passband of theLTC1164-8 is tuned with an externalclock; the clock-to-center-frequencyratio is 100:1. The stopband attenu-ation of the LTC1164-8 is greaterthan 50dB for input frequencies out-side a narrow band defined as ±4% ofthe center frequency of the filter (seeFigure 1).

One Op Amp and TwoResistors Build anUltra-Selective Filter

The LTC1164-8 requires an exter-nal op amp and two external resistors.The filter’s gain at its center frequencyis equal to 3.4RF/RIN. For optimumdynamic range with a gain equal toone, the external resistor RF shouldbe 90.9k and the external resistor RINshould be 340k. For gains other thanone, RIN = 340k/gain. Gains of up to1000 are possible. The complete con-figuration is shown in Figure 2. Notethat programming the filter’s gainwith input resistor R IN is equivalentto providing the LTC1164-8 withnoiseless preamplification, since thefilter’s internal noise is not amplified.The wideband noise of the LTC1164-8 measures 400µVRMS at ±5V and isindependent of the filter’s gain and

DESIGN FEATURES

FREQUENCY (kHz)

–80

–90

–70

–50

–60

–40

–20

–30

–10

0

10

GAI

N (d

B)

11648_1.eps

0.90 0.95 1.00 1.101.05

11648_2.eps

1

2

3

4

5

6

7

VIN

5V 100kHz

–5V

RIN 340k

14

13

12

11

10

9

8

LTC1164-8

RF 90.9k

CF 160pF

2

34

–5V

6

–

+LT1006 VOUT

7

5V

by Philip Karantzalis

Figure 1. LTC1164-8 Gain vs. frequencyresponse

Figure 3. Narrow-band signal extraction showing input to and output from the LTC1164-8filter. Filter center frequency set to 1kHz with gain = 100.


Figure 4. Signal detection in the presence of noise example showing input to and output fromthe LTC1164-8 filter. Filter center frequency set to 1kHz with gain = 100.

percent and the 950Hz signal is fourtimes larger than the 1kHz signal. Todetect the 1kHz signal, the LTC1164-8 is set to a gain of 100 and the clockfrequency is set to 100kHz. At thefiltered output of the LTC1164-8 thefollowing signals will be present: anextracted 1kHz, 1VRMS signal and arejected 950Hz, 2.7mVRMS signal, asshown in Figure 3. In a narrow-bandsignal separation and extraction ap-plication, as described previously, theLTC1164-8 provides a simple andreliable detection circuit solution.

A second signal-detection applica-tion occurs when a small signal is tobe detected in the presence of noise.For example, a 1kHz, 10mVRMS signalis mixed with a wideband noise signalthat measures 5mVRMS in a 400Hzfrequency band. The signal-to-noiseratio is just 6dB. With the LTC1164-8 set for a center frequency of 1kHz(fCLK is equal to 100kHz) and a gain of100, the 1kHz, 10mVRMS signal willbe detected and amplified. The wide-band noise will be band-limited bythe very narrow band gain responseof the LTC1164-8. At the output of aLTC1164-8, the 1kHz signal will be1VRMS, as shown in Figure 4. Thetotal band-limited noise will be70mVRMS, with a signal-to-noise ra-tio of more than 20dB, as shown inFigure 5. In applications of signaldetection in the presence of noise, theLTC1164-8 provides asynchronousdetection. Signal detection circuitssuch as synchronous demodulatorsand lock-in amplifiers require thepresence of a reference or carrier sig-nal to provide phase and frequencyinformation of the signal to be de-tected. With an LTC1164-8, signaldetection is accomplished by select-ing a very narrow signal detectionband around the frequency of thedesired signal, which is defined asfCLK divided by 100 (fCLK is the clockfrequency of the LTC1164-8), and byselecting the filter gain by choosingthe value of a resistor.

Figure 5. Wideband noise input to LTC1164-8 filter. Plots show input to and output from thefilter. Filter center frequency set to 1kHz with gain = 100.

DESIGN FEATURES


New LT1182/1183 Dual CCFL/LCDContrast Switching Regulator

by Anthony BonteIntroductionThe current generation of portable

computers and instruments use back-lit liquid-crystal displays (LCDs).These displays also appear in otherapplications, including medicalequipment, automobiles, gas pumps,and retail terminals. Cold-cathodefluorescent lamps (CCFLs) providethe highest available efficiency forbacklighting the display. These lampsrequire high voltage AC tooperate, mandating an efficient, high-voltage DC–AC converter. In additionto good efficiency, the convertershould deliver the lamp drive in theform of a sine wave. This is desirableto minimize RF emissions. Such emis-sions can cause interference withother devices, and can also degradeoverall operating efficiency. The sine-wave excitation also provides optimalcurrent-to-light conversion in thelamp. The circuit should also permitlamp intensity control from zero tofull brightness with no hysteresis or“pop-on.”

The LCD also requires a bias sup-ply for contrast control. The supply’soutput should be regulated, and vari-able over a considerable range.Manufacturers now offer an array ofeither positive or negative contrast-voltage displays. These displays differin operating-voltage range, contrast-adjust range and power consumption.

The small size and battery-pow-ered operation associated withLCD-equipped apparatus mandatelow component count and high effi-ciency. Size constraints place severelimitations on circuit architecture,and long battery life is usually a pri-ority. Laptop and hand-held portablecomputers offer an excellent example.The CCFL and its power supply areresponsible for almost 50% of thebattery drain. Displays found in newercolor machines can have contrastpower-supply battery drains as highas 20%. Additionally, all components,

including PC board and hardware,usually must fit within the LCDenclosure with a height restrictionof 0.25".

New LTC PartsLinear Technology has invested an

enormous amount of time, resources,and technical capability in providingthe premier backlight/contrast solu-tions for system designers. Incontinuing this commitment, LinearTechnology introduces the LT1182/LT1183. These new devices achieve ahigh level of system integration for abacklight/contrast solution. In addi-tion, they reduce system powerdissipation, maintain high efficiencyoperation, require fewer external com-ponents, and reduce overall systemcost.

The LT1182/LT1183 aredual, fixed-frequency, current-modeswitching regulators that provide thecontrol function for cold-cathode fluo-rescent lighting and liquid-crystaldisplay contrast. Two high-current,high-efficiency switches are includedon the die, along with an oscillator,reference, output-drive logic, controlblocks, and protection circuitry. Sepa-rate analog and power grounds forthe regulators minimize interaction.Combining both control functionsonto one die and maximizing perfor-mance allows the IC to fit into anarrow-body 16-pin SOIC. Boardspace and insertion cost are reducedin comparison with devices that re-quire two independent regulators.Also, external components requiredwith previous solutions have beenintegrated onto the IC, providing amore economical solution. TheLT1184 is also available and providesonly the CCFL function.

The LT1182/LT1183 operate withsupply voltages from 3V to 30V anddraw only 9mA quiescent current. Anactive low shutdown pin reduces total

supply current to less than 50µA forstandby operation. A 200kHz switch-ing frequency minimizes the size ofrequired magnetic components. Theuse of current-mode switching tech-niques with cycle-by-cycle limiting giveshigh reliability and simple loop fre-quency compensation.

The CCFL switching regulator typi-cally drives an inductor that acts as aswitched-mode current source for acurrent-driven Royer-class converterwith efficiencies as high as 90%. Thecontrol loop forces the regulator topulse-width modulate the inductor’saverage current to maintain constantcurrent in the lamp. The constantcurrent’s value, and thus lamp inten-sity, is programmable. This drivetechnique provides a wide range ofintensity control. A unique lamp-current programming block permitseither grounded-lamp or floating-lamp configurations. Grounded-lampcircuits directly control one-half of ac-tual lamp current. Floating-lampcircuits directly control the Royer’sprimary-side converter current.Floating-lamp circuits provide differ-ential drive to the lamp and reduce theloss from stray lamp-to-frame capaci-tance, extending illumination range.

The LCD contrast switching regula-tor is typically configured as a flybackconverter and generates a bias supplyfor contrast control. The supply’s vari-able output permits adjustment ofcontrast for the majority of availabledisplays. Newer types of displays re-quire a constant supply voltage andprovide contrast adjustment through aseparate digital control pin. A unique,dual-polarity error amplifier and theselection of a flyback converter topol-ogy allow either positive or negativeLCD-contrast voltages to be generatedwith minor circuit changes. The LT1182and LT1183 differ in their pinouts forthe LCD-contrast error amplifier. TheLT1182 brings out the individual

DESIGN FEATURES


Figure 1. LT1182/LT1183 CCFL/LCD contrast top-level block diagram

error-amplifier inputs for setting uppositive- and negative-polarity con-trast capability, whereas the LT1183ties the error-amplifier inputs to-gether and brings out an internalreference. The reference may be usedin generating negative contrast volt-ages or in programming lamp current.

OperationFixed-frequency, current-mode

switchers control switch duty cycledirectly by switch current rather thanby output voltage. Referring to theblock diagram for the LT1182/LT1183in Figure 1, the switch for each regu-lator is turned ON at the start of eachoscillator cycle. The switches areturned OFF when switch currentreaches a predetermined level. Thecontrol of output lamp current isobtained by using the output of aunique programming block to setcurrent trip level. The contrast volt-age is controlled by the output of a

dual-input-stage error amplifier,which sets current trip level. The cur-rent-mode switching technique hasseveral advantages. First, it providesexcellent rejection of input voltagevariations. Second, it reduces the 90°phase shift at mid-frequencies inthe energy storage inductor. Thissimplifies closed-loop frequency com-pensation under widely varyinginput-voltage or output-load condi-tions. Finally, it allows simple,pulse-by-pulse current limiting toprovide maximum switch protectionunder output overload or short-cir-cuit conditions.

The LT1182/LT1183 incorporatea low-dropout internal regulator thatprovides a 2.4V supply for most of theinternal circuitry. This low-dropoutdesign allows input voltage to varyfrom 3V to 30V with little change inquiescent current. An active low shut-down pin reduces total supply currentto less than 50µA and locks out

switching action for standby opera-tion. The LT1182/LT1183 incorporateundervoltage protection by sensingregulator dropout and locking outswitching below about 2.5V. The regu-lator also provides thermal-shutdownprotection and locks out switching inthe presence of excessive junctiontemperatures.

A 200kHz oscillator is the basicclock for all internal timing. The oscil-lator turns on each output switch bymeans of its own logic and drivercircuitry. Adaptive anti-saturationcircuitry detects the onset of satura-tion in each power switch and adjustsbase-drive current instantaneouslyto limit switch saturation. This mini-mizes driver dissipation and providesrapid turn-off of the switch. The CCFLpower switch is guaranteed to pro-vide a minimum of 1.25A and the LCDpower switch is guaranteed to pro-vide a minimum of 0.625A. Theanti-saturation circuitry provides a

DESIGN FEATURES

FBN

10

–

+ILIM

AMP2

COMP2COMP1

UNDER- VOLTAGE LOCKOUT

THERMAL SHUTDOWN

2.4V REGULATORSHUTDOWN

200kHz OSC

DRIVE 2

SHUTDOWN

9

6

LCD VSW

LCD PGND

LCD VC

FBP ICCFL AGND DIO BULB CCFL VC

CCFL PGND

CCFL VSW

ROYERBATVIN

LOGIC 2

ANTI- SAT2

LOGIC 1 DRIVE 1

LCD

GAIN = 4.4GAIN = 4.4

Q3 2 ×

Q5 1 ×

Q4 5 ×

Q8 1 ×

Q10 2 ×

R1 0.125Ω

R4 0.1Ω

LT1183: FBP AND FBN ARE TIED TOGETHER TO PIN 10 REFERENCE IS BROUGHT OUT TO PIN 11

R2 0.25Ω

Q6 2 ×

R3 1k

D2 6V

D1

Q11

Q1Q2

131412

Q7 9 ×

Q9 3 ×

8 7 11 2 5 3 15 4

16

1

V2 1.24V

V1 0.45V

–

+CCFL

0µA

TO 1

00µA

–10m

V

– +gm

– –+ +

–

+ILIM

AMP1

ANTI- SAT1

1182_1.eps

+

–


ratio of switch current to driver cur-rent of about 40:1.

Simplified Lamp-Current Programming

A programming block in theLT1182/LT1183 controls lampcurrent, permitting either grounded-lamp or floating-lamp configurations.Grounded configurations controllamp current by directly controllingone-half of actual lamp current andconverting it to a feedback signal toclose a control loop. Floating configu-rations control lamp current bydirectly controlling the Royer’s pri-mary-side converter current andgenerating a feedback signal to closea control loop.

Previous backlighting solutionshave used a traditional error ampli-fier in the control loop to regulatelamp current. This approach con-verted an RMS current into a DCvoltage for the input of the error am-plifier. This approach used severaltime constants to provide stable loopfrequency compensation. This com-pensation scheme meant that the loophad to be fairly slow and that outputovershoot with start-up or overloadconditions had to be carefully evalu-ated in terms of transformer stressand breakdown voltage requirements.

The LT1182/LT1183 eliminate theerror-amplifier concept entirely andreplace it with a novel lamp-currentprogramming block. This block pro-vides an easy-to-use interface toprogram lamp current. The program-mer circuit also reduces the numberof time constants in the control loopby combining the error-signal con-version scheme and frequencycompensation into a single capacitor.The control loop thus exhibits theresponse of a single-pole system, al-lows for faster loop transient response,and virtually eliminates overshootunder start-up or overload conditions.Finally, these parts include open-lamp protection circuitry, with usercontrol by means of a simple, exter-nal RC network. This significantlyeases the breakdown requirementsfor the transformer and lowers the

cost associated with winding a high-voltage transformer.

Lamp current is programmed atthe input of the programmer block,the ICCFL pin. This pin is internallyregulated to 450mV and accepts a DCinput current signal of 0–100µA. Thisinput signal is converted to a 0–500µAsource current at the CCFL VC pin.By regulating the ICCFL pin, the inputprogramming current can be set withDAC, PWM, or potentiometer control.

In a grounded-lamp configuration,the low-voltage side of the lamp con-nects directly to the LT1182/LT1183DIO pin. This pin is the commonconnection between the cathode andanode of two internal diodes. In pre-vious grounded-lamp solutions, thesediodes were discrete units; in theLT1182/LT1183, they are integratedonto the IC. Bidirectional lamp cur-rent flows in the DIO pin and thus thediodes conduct on alternate halfcycles. Lamp current is controlled bymonitoring one-half of the lamp cur-rent. The diode conducting onnegative half cycles has one-tenth ofits current diverted to the CCFL pinand nulls against the source currentprovided by the lamp-current pro-grammer circuit. The compensationcapacitor on the CCFL VC pin pro-vides both loop compensation and anaveraging function to the rectifiedsinusoidal lamp current. Therefore,input programming current is relatedto one-half of average lamp current. Ifa floating-lamp configuration is used,the DIO pin is grounded.

In a floating-lamp configuration,the lamp is fully floating with nogalvanic connection to ground. Thisallows the transformer to provide sym-metric, differential drive to the lamp.Balanced drive eliminates the fieldimbalance associated with parasiticlamp-to-frame capacitance and re-duces “thermometering” (unevenlamp intensity along the lamp length)at low lamp currents. Display de-signs should be carefully evaluatedin relation to construction shape,materials, and asymmetric lampwiring, as energy leakage terms de-grading efficiency up to 20% havebeen noted in practice. Maintaining

closed-loop control of lamp currentnow necessitates deriving a feedbacksignal from the primary side of theRoyer transformer. Previous solutionshave used an external precision shuntand high-side sense amplifier con-figuration. This approach has beenintegrated onto the LT1182/LT1183for simplicity of design and ease ofuse. Primary-side Royer-convertercurrent is related to lamp current bythe turns ratio of the transformerand its reflected impedances. TheRoyer-converter current is monitoredacross an internal 0.1Ω resistor,which is connected across the inputterminals of a high-side sense amp-lifier. The input terminals arerepresented by the BAT and Royerpins. A 0–1A Royer primary-side, cen-ter-tap current is translated to a0–500µA sink current at the CCFL VCpin to null against the current sourceprovided by the programmer circuit.Once again, the compensation ca-pacitor on the CCFL VC pin providesboth loop compensation and an aver-aging function to the error sinkcurrent; therefore, input-program-ming current is related to averageRoyer-converter current. Floating-lamp circuits operate similarly togrounded-lamp circuits, except forthe derivation of the feedback signal.However, floating-lamp circuitspermit the lamp to operateover a 40:1 intensity range with-out “thermometering,” whereasgrounded-lamp circuits are usuallylimited to a 10:1 range.

Open-lamp protection is providedby an internal, 7V-threshold com-parator connected between the BATand Lamp pins. This circuit sets amaximum voltage level across theprimary side of the Royer converterand limits the maximum transformeroutput voltage under start-up oropen-lamp conditions. This easestransformer voltage-rating require-ments. The Lamp pin is connected tothe junction of an external resistordivider network. The divider networkconnects from the center tap of theRoyer transformer to the top side ofthe Royer inductor. A capacitor acrossthe top of the divider network filters

DESIGN FEATURES


C11 2.2µF

35V

BAT 8V TO 28V

POSCON

NEGCON

1182_2.eps

EITHER NEGCON OR POSCON MUST BE GROUNDED. GROUNDING NEGCON GIVES 15V FIXED. GROUNDING POSCON GIVES VARIABLE NEGATIVE CONTRAST FROM –10V TO –30V.

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

ICCFL

DIO

CCFL VC

AGND

SHDN

LCD VC

CCFL VSW

BULB

BAT

ROYER

VIN

FBP

FBN

LCD VSW

CCFL PGND

LCD PGND

LT1182

LAMP

UP TO 6mA

10 6

L1

L3

R1 750Ω

L2 100µH

1 2 3 5 4

+

+

+

+

ALUMINUM ELECTROLYTIC IS RECOMMENDED FOR C3B WITH AN ESR ≥ 0.5Ω TO PREVENT LT1182 HIGH-SIDE SENSE RESISTOR DAMAGE DUE TO SURGE CURRENTS AT TURN-ON.

C7, 1µF

C1* 0.033µF

D1 1N5818

D2 1N914

D4 1N914

V (CCFL) 0V TO 5V kHz PWM

C6 4.7µF

R4 34k 1%

SHUTDOWN

C8 1µF

R7, 1k

R5, 40.2k, 1%

C5 1000pF

R2 220k

R3 100k

C3A 2.2µF 35V

C10 10µF 35V

R11 54.9k

1%

R12 4.99k

1%

V (CONTRAST) 0V TO 5V

R10, 120k, 1%R9, 20k, 1%

C4 2.2µF

VIN ≥ 3V

D3 1N5934A 24V

C3B 2.2µF 35V

C2 27pF 3kV

+

C1 MUST BE A LOW LOSS CAPACITOR, C1 = WIMA MKP-20

Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001

L1 = SUMIDA EPS-207 OR COILTRONICS CTX110605. PIN NUMBERS SHOWN FOR COILTRONICS UNIT (C1 VALUE MAY REQUIRE ADJUSTMENT WITH COILTRONICS).

L2 = COILTRONICS CTX100-4

L3 = COILTRONICS CTX02-12403

*DO NOT SUBSTITUTE COMPONENTS

COILTRONICS (407) 241-7876 SUMIDA (708) 956-0666

Q2* Q1*

N = 1:2

NOTE: SET V (CONTRAST) TO 5V IF ONLY POSITIVE CONTRAST VOLTAGES ARE GENERATED.

0µA TO 50µA ICCFL CURRENT GIVES ≈ 0mA TO 6mA BULB CURRENT.

out switching ripple and sets a timeconstant that determines how quicklythe clamp activates. When the com-parator activates, current is generatedto pull the CCFL VC pin down. Thisaction transfers the entire regulatorloop from current-mode operationinto voltage-mode operation.

Dual-PolarityContrast Capability

The LCD contrast regulator can beoperated in many standard switchingconfigurations. A dual-input-stageerror amplifier can regulate eitherpositive or negative contrast voltages.In fact, if a flyback configuration ischosen, divider networks can beimplemented for both polarities ofcontrast. Output polarity is then de-termined by which side of thetransformer secondary the outputconnector grounds. If only positivecontrast voltages are to be generated,one can also use a boost convertertopology for higher efficiencyoperation. The FBN pin is the nonin-verting terminal for the negative

Figure 2. 90% Efficient floating CCFL configuration with dual-polarity LCD contrast

contrast-control error amplifier. Theinverting terminal is offset fromground by −10mV to define the error-amplifier output state under startupconditions. The FBN pin acts as asumming junction for the resistordivider network. FBN input bias cur-rent is typically −1µA. The FBP pin isthe inverting terminal for the positivecontrast-control error amplifier. Thenoninverting terminal is tied to aninternal 1.24V reference. FBP inputbias current is typically 0.5µA. TheLCD VC pin is the high-impedancecurrent output (gm) for the contrasterror amplifier. A series RC networkto ground typically performs loop fre-quency compensation.

Floating CCFL withVariable-Negative/Fixed-Positive Contrast

Figure 2 is a complete floating CCFLcircuit with variable-negative/fixed-positive contrast voltage generationbased on the LT1182. Q1, Q2, L1, andC1 form the core of the Royer con-verter. The LT1182 CCFL VSW pin and

L2 form the switch-mode currentsource that drives the Royer. The200kHz switching frequency allowsthe use of only 100µH for L2. Hookupfor a floating-lamp configuration isas simple as placing the high-sidesense resistor in series with the cen-ter tap of the Royer transformer andfloating the lamp on the transformersecondary.

The feedback signal generated bythe Royer-converter current is con-verted to an error current at the CCFLVC pin. This current is then averagedby compensation capacitor C7 to nullagainst the programming currentsource. C7’s value of 1µF providesstable loop compensation and excel-lent transient response under startupconditions. It is important to notethat the catch diode D1 is returned tothe BAT side of the high-side senseresistor. This connection maintainscurrent flow in L2 when the switch isoff, improves operating efficiency byreturning current to the input sup-ply, and ensures that the feedbacksignal is generated only by the Royerconverter current. The transfer func-tion between primary-side convertercurrent and programming current

DESIGN FEATURES


DESIGN FEATURES


C11 2.2µF 35V

BAT 8V TO 28V

POSCON

NEGCON

1182_3.eps

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

ICCFL

DIO

CCFL VC

AGND

SHDN

LCD VC

CCFL VSW

BULB

BAT

ROYER

VIN

FBP

FBN

LCD VSW

CCFL PGND

LCD PGND

LT1182

LAMPUP TO 6mA

2 10

L1

L3

R1 750Ω

L2 100µH

1 2 3 5 4

+

+

+

C7 1µF

C1* 0.033µF

D1 1N5818

D2 1N914

D4 1N914

N = 1:2 V (CCFL)

0V TO 5V

SHUTDOWN

C8 1µF

R7, 1k

R5 84.5k 1%

C5 1000pF

R2 220k

R3 100k

C2 27pF 3kV

C10 10µF 35V

R11 54.9k

1%

R12 4.99k

1%

V (CONTRAST) 0V TO 5V R10

120k 1%

R9 20k 1%

C4 2.2µF

VIN ≥ 3V

D3 1N5934A 24V

C3 4.7µF 35V

+

MUST BE A LOW LOSS CAPACITOR WIMA MKP-20

ZETEX ZTX849 OR ROHM 2SC5001

SUMIDA EPS-207 OR COILTRONICS CTX110602 PIN NUMBERS SHOWN FOR COILTRONICS UNIT CTX100-4 COILTRONICS CTX02-12403

C1 C1 =

Q1, Q2 =

L1 =

L2 = L3 =

Q2* Q1*

NOTE: SET V (CONTRAST) TO 5V IF ONLY POSITIVE CONTRAST VOLTAGES ARE GENERATED.

0µA TO 50µA ICCFL CURRENT GIVES ≈ 0mA TO 6mA BULB CURRENT.

EITHER NEGCON OR POSCON MUST BE GROUNDED. GROUNDING NEGCON GIVES 15V FIXED. GROUNDING POSCON GIVES VARIABLE NEGATIVE CONTRAST FROM –10V TO –30V.

*DO NOT SUBSTITUTE COMPONENTS

COILTRONICS (407) 241-7876 SUMIDA (708) 956-0666 THE ICCFL CURRENT REQUIRED FOR A GIVEN RMS BULB CURRENT IS: ICCFL ≈ (9 × 10–3)(IBULB)

must be empirically determined andis dependent upon a myriad of factorssuch as lamp characteristics, displayconstruction (which dominates radi-ated loss terms), transformer turnsratio, and the tuning of the Royeroscillator.

The method used to generate theinput programming current is shownin Figure 2. A 1kHz, 0–5V pulse-width-modulated logic signal isconverted to DC by the R4/C6 filternetwork. R5 then converts the DCvoltage into programming current atthe ICCFL pin. 0µA to 50µA ICCFL cur-rent produces 0mA to 6mA lampcurrent.

R2, R3, and C5 form the externaldivider network for the open-lampprotection circuit. The divider moni-tors the voltage across the Royerconverter and compares it with theinternal clamp voltage. In this ex-ample, the divider is set to limit themaximum Royer primary voltage toabout 10V. Capacitor C5 filters outripple at the emitters of Q1 and Q2,which have a ripple component oftwice the Royer frequency.

Efficiency with typical CCFL lampsranges from 85–90% at full load, withlamp current being variable from200µA to 6mA. Higher electrical effi-ciencies are obtainable by increasingthe harmonic content in the wave-

forms (lowering the value of C1) andincreasing the magnetics size to de-crease copper loss. The componentvalues chosen were an interactivecompromise in maximizing photomet-ric output versus input power. Basedrive for Q1 and Q2 is provided bythe tickler winding and the value ofR1. R1’s value must be chosen toguarantee sufficient base drive withminimum betas for Q1 and Q2. Thebattery supply range can extend from8V to 28V, with a nominal batteryvoltage of 12V. The LT1182 is typi-cally powered from a 5V or 3.3V logicsupply. The LT1182 can also bepowered from the battery supply, but

Figure 3. 90% Efficient grounded CCFL configuration with dual-polarity LCD contrast


IntroductionThe LT1366 family is Linear

Technology’s first series of dual andquad bipolar operational amplifiersto combine rail-to-rail input and out-put specifications with precision VOSspecs. The LT1366 family maintainsprecision specifications over a widerange of operating conditions. Thedevices will operate with supply volt-ages as low as +1.8V, and are fullyspecified for +3V, +5V, and ±15V op-eration. Offset voltage is typically150µV when operating from a single5V supply. Supply rejection is 115dB.Open loop gain, AVOL, is 2 milliondriving a 2k load. Common-mode re-jection ratio is typically 86dB over thefull rail-to-rail input range. The com-bination of precision specificationsand rail-to-rail operation makes theLT1366 series versatile amplifiers,suitable for signal processing tasksthat demand the widest possiblecommon-mode range.

The amplifiers are available in twoversions, which differ in their abilityto drive capacitive loads. The LT1366dual and LT1367 quad have conven-

put stages. Figure 1 shows a simpli-fied schematic of the LT1366. Theinput stage consists of two differen-tial amplifiers, a PNP stage Q1–Q2and an NPN stage Q3–Q4, which areactive over different portions ofthe input common-mode range. Lat-eral devices are used in both inputstages, eliminating the needfor clamps across the input pins.Each input stage is trimmed for offsetvoltage. A complementary output con-figuration (Q23–26) is employed tocreate an output stage with rail-to-rail swing. The device is fabricated onLinear Technology’s proprietarycomplementary bipolar process,which ensures very similar DC andAC characteristics for the output de-vices Q24 and Q26.

Since two separate input stagesare used to obtain the rail-to-rail in-put range, there are also two sets ofinput specifications for the device.Linear Technology fully specifies theinput characteristics for each stageand the difference in each parameterbetween stages. The specification for

tional compensation, and are stablewith load capacitances of 1000pF orless. For use in low-frequency or DCapplications, the LT1368 dual andLT1369 quad are compensated foruse with a 0.1µF capacitor at theoutput. In a noisy environment, thelarge output capacitor cleans up theoutput signal by improving the sup-ply rejection and providing a lowoutput impedance at highfrequencies.

The LT1366/LT1368 dualamplifiers are available with indus-try-standard pin-out in either 8-pinSO or 8-pin mini-DIP packages. TheLT1367/LT1369 quad amplifiers areavailable in the 16-pin narrow (150mil width) SO16. The pinout for theSO16 is the same as the industrystandard, but with two end pins (8and 9) left open.

ObtainingRail-to-Rail Operation

As might be expected, the LT1366differs from a conventional op amp inthe design of both the input and out-

Figure 1. LT1366 Simplified schematic diagram

DESIGN FEATURES

1366_1.eps

(VS –300mV)

IN–

V–

IN+V–

V–V–

Q12

Q4 Q7

Q13 Q18 Q19

Q20 Q25

Q26

Q24Q10

V+V+

Q6

D3

D1

D2Q3 Q15Q14

Q8

Q2Q1

Q5Q16

Q17

Q23

OUT

C2CC

C1

Q22

D8

Q21I1

Q9

D7

Q11

D4 D5D6 D7

V–

V+V+

V+

The LT1366 Family: Precision,Rail-to-Rail Bipolar Amplifiers by William Jett

and Sean Gold


Two circuits prevent the outputfrom reversing polarity when theinput voltage exceeds the common-mode range. When the noninvertinginput exceeds the positive supply byapproximately 300mV, device Q12turns on, pulling the output of thesecond stage low, which forces theoutput high. For inputs below thenegative supply, diodes D1–D2 turnon, overcoming the saturation of theinput pair Q1–Q2.

Improved SupplyRejection—the LT1368

The LT1368 is a variation of theLT1366 offering greater supply rejec-tion and lower high-frequency outputimpedance. The LT1368 requires a0.1µF load capacitance for compen-sation. The output capacitance formsa filter, which reduces pickup fromthe supply and lowers the outputimpedance. This additional filteringis helpful in mixed analog/digitalsystems with common supplies, or insystems employing switchingsupplies.

Figures 2a and 2b show the out-puts of the LT1366 and the LT1368with a 200mVP–P, 100kHz square waveadded to the positive supply. Notethat the signal on the output of theLT1368 is only 20mVP–P.

PerformanceTable 1 summarizes the guaran-

teed DC performance of the LT1366.As mentioned before, the device isfully tested and guaranteed for +3V,+5V, and ±15V operation. In additionto the traditional op amp specs, Lin-ear Technology also guarantees the

input bias-current shift properly ac-counts for the opposite directions ofthe input current in the PNP and NPNstages.

First, looking at the input stage,Q5 switches the current from currentsource I1 between the two inputstages. When the input common modevoltage VCM is near the negative sup-ply, Q5 is reverse biased, so thecurrent from I1 becomes the tail cur-rent for the PNP differential pairQ1–Q2. At the other extreme, whenVCM is near the positive supply, thePNPs Q1–Q2 are biased off. The cur-rent from I1 then flows through Q5 tothe current mirror D3–Q6, furnish-ing the tail current for the NPNdifferential pair Q3–Q4. The switch-over point between stages occurs whenVCM is equal to the base voltage of Q5,which is biased approximately 1.3Vbelow the positive supply.

The collector currents of the twoinput pairs are combined in the sec-ond stage, consisting of Q7–Q11. Mostof the voltage gain in the amplifier iscontained in this stage. Differentialamplifier Q14–Q15 buffers the out-put of the second stage, convertingthe output voltage to differential cur-rents. The differential currents passthrough current mirrors D4–Q16 andD5–Q17, and are converted to differ-ential voltages by Q18 and Q19. Thesevoltages are also buffered and ap-plied to the output Darlington pairsQ23–24 and Q25–26. Capacitors C1and C2 form local feedback loopsaround the output devices, loweringthe output impedance at highfrequencies.

DESIGN FEATURES

Figure 2a. LT1366 Power-supply rejection test Figure 2b. LT1368 power-supply rejection test

maximum shift in offset voltage andinput currents between the NPN andPNP input stages.

ApplicationsThe LT1366 series opens up a new

range of possibilities for low-powerand rail-to-rail applications. The fol-lowing circuits demonstrate theLT1366’s versatility and precision insolving problems ranging from simplebuffers to continuous-time activefilters.

Potentiometer BufferIt is often attractive in automotive

and position-sensing applications toconnect a potentiometer directlyacross a reference supply. Bufferinga potentiometer connected in this wayrequires an op amp with rail-to-railinput and output swing (Figure 3).Despite the circuit’s simplicity, caremust be taken in component selec-tion. The potentiometer shouldpresent a load commensurate withthe op amp’s supply current, yet besmall enough to assure negligibleinput-bias-current induced offset er-rors. In the example shown, a 10kΩpotentiometer draws 500µA from theinput supply and causes less than50µV of error due to bias current. Theoffset-voltage error changes with thepotentiometer setting, with the maxi-mum error at mid-scale and theminimum error near the supply rails.

When the op amp’s output satu-rates, the offset-voltage error becomesthe difference between the input sig-nal and the saturation voltage. Thesaturation voltage as a function of

Figure 3. Rail-to-rail potentiometer buffer

–

+

1366_3.eps

VCC

RP 10k

RL

1/2 LT1366


Table 1. LT1366 guaranteed DC performance—25°C

VS = 3V VS = 5V VS = ±15V

Max offset voltage VCM = VCC 450µV 450µV 650µV= VEE 450µV 450µV 650µV

Max input bias current VCM = VCC 35nA 35nA 35nA= VEE 35nA 35nA 35nA

Max input offset current VCM = VCC 6nA 6nA 6nA= VEE 6nA 6nA 6nA

Max offset voltage shift over CMR 400µV 400µV 500µVMax input bias current shift 70nA 70nA 70nAMax input offset current shift 6nA 6nA 6nAMin open-loop gain (RL = 2k) 500k 500k 2000kMin channel separation 120dBMax output voltage—LOW no load 12mV 12mV 12mV

ISINK = 2.5mA 200mV 200mV 200mVMax output voltage—HIGH no load 8mV 8mV 8mV

ISOURCE = 2.5mA 250mV 250mV 250mVMin output current ±10mA ±15mA ±30mAMax supply current per amp 500µA 500µA 550µA

load current is shown in Figure 4. Forlight loads, such as an A/D converter,the op amp can swing to within 5mVof each rail. At 1mA of load current,the offset voltage error is less than75mv; at 10mA of load current, it isless than 500mV.

Topside Current SourceThe circuit shown in Figure 5 takes

advantage of the LT1366’s rail-to-railinput range to form a wide-compli-ance current source. The LT1366adjusts Q1’s gate voltage to force thevoltage across the sense resistor(RSENSE) to equal the voltage from the

DESIGN FEATURES

In this example, the circuit deliv-ers 1A at 200mV of sense voltage.With a 5V input supply the powerdissipation is 5W. For operation at70°C ambient temperature, theMOSFET’s heat sink must have athermal resistance of:

θHS = θJA SYSTEM - θJC FET= 55°C/5W - 1.25°C/W= 9.75°C/W.

This is easily achievable with asmall heat sink. When input voltagesare greater than 5V, the use of alarger heat sink or derating of theoutput current is necessary.

The circuit’s supply regulation isabout 0.03%/V. The output imped-ance is equal to the MOSFET’s outputimpedance multiplied by the op amp’sopen-loop gain. Degradations in cur-rent-source compliance occur whenthe voltage across the MOSFET’s on-resistance and the sense resistordrops below the voltage required tomaintain the desired output current.This condition occurs when VCC − VOUT< ILOAD × [RSENSE + R ON].

High Side Current-Sense Amplifier

In power control, it is sometimesnecessary to sense load current atlow loss near the input supply. Thecurrent-sense amplifier shown in Fig-ure 6 amplifies the voltage across asmall value sense resistor by the ratioof the current-source resistors (R2/R1). The LT1366 forces the low-powerMOSFET’s gate voltage such that thesense voltage appears across a cur-rent-source resistor R1. The resultingcurrent in Q1’s drain is converted to

Figure 6. High side current-sense amplifier

–

+

V I RRR

I

O IN S

IN

=

= •

21

20Ω1366_6.eps

VCC

RS 0.2Ω

R1 200Ω

Q1 TPO610L

IIN

1/2 LT1366

R2 20k

supply to the potentiometer’s wiper. Arail-to-rail op amp is needed becausethe voltage across the sense resistormust drop to zero when the dividedreference voltage is set to zero. Q2acts as a constant current sink tominimize error in the reference volt-age when the supply voltage varies.

The circuit can operate over a widesupply range (5V < VCC < 30V). At lowinput voltage, circuit operation is lim-ited by the MOSFET’s gate-driverequirements. At high input voltage,circuit operation is limited by theLT1366’s absolute maximum ratingsand the output power requirements.

–

+

1366_5.eps

Q2 2N4340

5V < VCC < 30V 1A > ILOAD > 160mA

RP 10k

40k

LT1004 -1.2

RSENSE 0.2Ω

VCC

1k

100Ω1/2 LT1366

0.0033µF

Q1 MTP23P06

IOUT

Figure 4. LT1366 VSAT vs. load current

LOAD CURRENT (mA)

10

SATU

RAT

ION

VO

LTAG

E (m

V)

100

0.001 0.1 1 10

1366_09.eps

10.01

1000

TOP SIDE

BOTTOM SIDE

Figure 5. Topside current source


–

+–

+–

+–

+

VIN

3.3V

VOUT

R1 29.5k*

R2 8.6k*

29.5k*

20k13k

* = 1% RESISTORS1µF

A1 1/4 LT1367

C1 10,000pF

11.8k*A2 1/4 LT1367

C2 10,000pF

21.5k*

11.8k*

A3 1/4 LT1367

10,000pF

SECTION 2SECTION 1

A4 1/4 LT1367

10,000pF

1366_7.eps

DESIGN FEATURES

a ground-referred voltage at R2. (VO =IIN RS [R2/R1])

The circuit takes advantage of theLT1366’s ability to sense signals upto the supply rail, which permits theuse of small-value, low-loss senseresistors. The LT1366 and the gainsetting resistors are also biased atlow current to reduce losses in thecurrent sense.

Single-Supply, 1kHz, Fourth-Order Butterworth Filter

The circuit shown in Figure 7 takesadvantage of all four op amps in theLT1367 to form a fourth-order But-terworth filter. The filter is a simplified

state-variable architecture consist-ing of two cascaded second-ordersections. Each section uses the 360degree phase shift around the two opamp loop to create a negative sum-ming junction at A1’s positive input.1The circuit has two-thirds the powerdissipation and component count asthe classic three op amp biquad,2 yetit has the same low component sensi-tivities for center frequency, w0,and Q.

For cutoff frequencies other thanthe 1kHz example shown, use thefollowing formula for each section:

w02 = 1/(R1 C1 R2 C2),

where R1 = 1/(w0 Q C1) and R2 = Q/(w0C2)

Figure 7. 1kHz fourth-order Butterworth filter

Figure 8. Frequency response of fourth-order Butterworth filter

The DC bias applied to A2 and A4for single-supply operation is notneeded when split supplies are avail-able. The circuit’s output can swingrail-to-rail and displays the maxi-mally flat amplitude response with a1kHz cutoff frequency with 80dB/decade rolloff (Figure 8).

References:

1. Hahn, James. 1982. State Variable Filter TrimsPredecessor’s Component Count. Electronics,April 21, 1982.

2. Thomas, L.C. 1971. The Biquad: Part I—SomePractical Design Considerations. IEEE Transac-tions on Circuit Theory, 3:350–357, May 1971.


LTC1325 Battery Management SystemOffers Unparalleled Flexibility

all the functional blocks needed toimplement a simple but sophisticatedbattery charger (see Figure 1). Themain features of the LTC1325 may besummarized as follows:

It has all the functional blocksneeded to build a charger,including a 10-bit ADC, fault-detection circuitry, a switchingbuck regulator controller, a P-channel MOSFET driver, a timer,a 3V regulator for poweringexternal temperature sensors,and a programmable batterydivider.

The functional blocks are placedunder the control of an externalmicroprocessor for maximumflexibility and adaptability todifferent battery chemistries orcharge rates.

Communication with the micro-processor is via an easy-to-use,four-wire serial interface.

It has stand-alone fault detectioncircuitry to protect the batteryagainst temperature or voltageextremes.

In addition to charging, the chipcan discharge batteries forbattery conditioning purposesand the part includes an accu-rate capacity monitoring function(gas gauge).

It charges batteries using aswitching buck regulator forhigher efficiency and lower powerdissipation.

The supply range is 4.5 to 16V,so that the LTC1325 can bepowered from the chargingsupply for up to ten cells.

A shutdown mode drops thesupply current to 50µA

Charging CircuitUnlike most other charger ICs,

which employ linear regulators, theLTC1325 charges batteries using aswitching buck regulator. This ap-proach improves efficiency andminimizes power dissipation, espe-cially when charging high-capacitybatteries. The only external compo-nents required are an inductor, aP-channel MOSFET switch, a senseresistor, and a catch diode (see Figure2). A programmable battery dividerthat accommodates one to sixteencells removes the need for an externaldivider. All the circuitry for control-ling the loop is integrated on-chip; noexternal ICs are needed.

The LTC1325 operates from 4.5Vto 16V, so that it can be powereddirectly from the charging supply.The wide supply range makes it pos-sible to charge up to ten cells withoutthe need for an external regulator todrop the charging supply down topower the LTC1325. When chargingis completed and the charging supplyis removed, the chip does not loaddown other system supplies, becausethe microprocessor can program theLTC1325 into shutdown mode, inwhich the quiescent current drops to

IntroductionNiCad and NiMH batteries may be

classified according to how quicklythey are designed to be charged. Fastcharge rate batteries can be chargedin as little as fifteen minutes, com-pared to an overnight wait for standardrate batteries. However, nothingcomes for free. With fast batteries,overcharging must be limited or bat-tery life will be adversely affected. Therequirement to terminate charge com-plicates the design of fast chargers.

There are basically two ways tolimit overcharging: a battery can becharged in several stages, i.e., withthe charging current reduced in laterstages, or the proper charge-termina-tion technique can be used. Thecomplete charging algorithm dependson the battery type and takes intoaccount the battery manufacturer’sspecifications and recommendations.The charger circuit should also pro-vide protection against faults (suchas excessively high battery tempera-ture or voltage) to safeguard thebattery when normal terminationfails.

The LTC1325 is new fast chargerIC designed to provide the user with

by Anthony Ng, Teo Yang Long, and Robert Reay

Figure 1. LTC 1325 fast-charger circuit

DESIGN FEATURES

1325_1.eps

VDD

PGATE

DIS

VBATT

TBAT

R2

R3CREG 4.7µF

R4

TA

VIN

SENSEHTF

MCV

LTF

CLK

CS

DIN

DOUT

REG

FILTER

p1.3

p1.2

p1.4

MPU (e.g. 8051)

GND

RSENSE

THERM 2 RDIS

IRFZ94

RTRKL1 50µH

D1 1N5818

VDD (4.5V-16V)

P1 IRF9Z30

THERM 1

R5 R6R1

+

CF 1µF

C2 10µF

+


comparators that monitor TBAT andVCELL to detect low temperature faults(LTF), high temperature faults (HTF),low cell voltages (BATR) and high cellvoltages (MCV). The LTF, HTF, andMCV thresholds are set by an exter-nal resistor divider to maximizeflexibility. The LTC1325 also includesa 32-bit counter that permits themicroprocessor to limit the maximumcharging time to one of eight time-outvalues (5, 10, 20, 40, 80, 160, 320minutes or no time-out). It is possibleto disable timer faults by selecting notime-out.

Battery ConditioningUnder some operating or storage

conditions, NiCad and NiMH batter-ies may lose full capacity. It is oftennecessary to subject such batteriesto repeated deep discharge and chargecycles to restore them to full capacity.The LTC1325 can be programmedinto discharge mode, in which it au-tomatically discharges VCELL to 0.9V.This voltage is defined as the end ofdischarge voltage (EDV). Fault pro-tection is also active in dischargemode to protect the battery againsttemperature extremes (LTF, HTF) and

measure TBAT, TAMB and VCELL. Bykeeping track of elapsed time, themicroprocessor will have the meansto calculate all existing terminationtechniques (including dTBAT/dt andd2BAT/dt2) and perform averaging toreduce the probability of false termi-nation. This flexibility also meansthat a single circuit can charge bothNiCad and NiMH batteries. TheLTC1325 has an on-chip, 10-bit, suc-cessive-approximation ADC with afive-channel input multiplexer. Threechannels are dedicated to TBAT, VCELL,and the gas gauge (see the section oncapacity monitoring); the other twochannels can be used for other pur-poses, such as sensing TAMB oranother external sensor. The LTC1325can be programmed into idle mode, inwhich the charge loop is turned off.This permits measurements to bemade without the switching noisepresent when charging.

Fault ProtectionThe LTC1325 monitors battery

temperature, cell voltage, and elapsedtime for faults, and prevents the ini-tiation or the continuation of chargingshould a fault arise. The fault detec-tion circuit (see Figure 3) consists of

50µA. In shutdown mode, the digitalinputs stay alive to await the wake-up signal from the microprocessor.

The buck-regulator control circuitmaintains the average voltage acrossthe sense resistor (RSENSE) at VDAC. Inaddition, a programmable duty cyclemodulates the P-channel MOSFETdriver output PGATE to reduce averagecharging current. The average charg-ing current is given by:

ICHARGE = VDAC × (duty cycle)/RSENSE

The microprocessor can set VDAC toone of four values(150mV, 50mV,30mV or 15mV) and the duty cycle toone of five values (1/16, 1/8, 1/4, 1/2,1) giving 20 possible ICHARGE valueswith a single RSENSE resistor.

Charge TerminationA plethora of charge termination

techniques are used in the LTC1325.These are based on battery tempera-ture (TBAT), cell voltage (VCELL), time(t), ambient temperature (TAMB) or acombination of these parameters.Unlike other fast charging ICs, theLTC1325 does not lock the user intoa particular termination techniqueand its shortcomings. Instead, it pro-vides the microprocessor a means to

Figure 2. LTC1325 charge, discharge, and gas-gauge circuit

DESIGN FEATURES

1325_2.eps

DUTY RATIO GENERATOR

CHARGE

DISCHARGE

TO ADC MUX

111kHz OSCILLATOR

ONE SHOT

Q S

R

DR0-DR23

–

+

–

+

A2

A1

DAC

VR0, VR1 GAS-GAUGE (GG) CHIP BOUNDARY

REG (3.072V)

C1 16pF

CF

RSENSE

S1

GG

S2CL

R1 500k

2VDAC

S3 S4

PGATE

DIS

L1 50µH

D1 1N5818

VDD (4.5V-16V)

SENSEBATTERY

FILTER

R2 125k

RF 1k

RDIS

RTRKP1 IRF9Z30

N1 IRFZ94

GG 0 0 0 0 1

VR1

0 0 1 1 X

VR0

0 1 0 1 X

DAC VOLTAGE

15mV 30mV 60mV

160mV 0mV


to detect the EDV discharge termina-tion point.

Capacity MonitoringThe LTC1325 may be programmed

into gas gauge mode (GG = 1 in Figure2). In this mode, the sense resistorsenses the battery load current. Thesense voltage is filtered by an 1kΩ ×CF lowpass filter. Amplifier A1 is con-figured as an inverting amplifier witha gain of four. The output of A1 is theaverage sense voltage and is con-verted by the ADC when the gas-gaugechannel is selected by the micropro-cessor. By accumulating gas-gaugemeasurements over time, the micro-processor can determine how muchcharge has left the battery and whatcapacity remains.

ConclusionsBy leaving all decisions (except fault

detection) to the microprocessor, theLTC1325 does not lock the user intoany charge algorithm or charge ter-mination technique. Multistagecharging algorithms or several charg-ing algorithms can be implementedin software, so that a wide range ofbattery types can be charged with thesame circuit. The design of the exter-nal charging circuit is made as simpleas possible by incorporating all thefunctional blocks needed and mini-mizing external component count. Inaddition to charging batteries, theLTC1325 has provisions for condi-tioning batteries and measuringbattery capacity. Each of these threefunctions is under the full control ofthe microprocessor.

shutdown protection at some cost inefficiency. This small loss in efficiencycan be balanced by considering thesystem efficiency and utilizing anauxiliary winding on the inductor ofthe primary power supply. The switch-matrix approach is the ultimate insimplicity but it requires all voltagesto be present the inputs, since novoltages are generated. The bestmethod depends on the application.

ConclusionThe PCMCIA specification requires

a significant amount of voltage switch-ing on both the primary supply (VCC)and the flash-memory programmingsupply pins (VPP). The complexity ofthese requirements mandates spe-cific integrated solutions. Each of thetwo approaches for VPP switching,linear regulation and switch matrix,has its merits. Linear regulation pro-vides current limiting and thermal-

Figure 3. Fault-detection circuitry

DESIGN FEATURES

PCMCIA, continued from page 5

with the LT1312 can be used to pro-vide a rough 13V input to the LT1121for an efficient system design. TheLTC1314 will enable the LT1121 lin-ear regulator only during flashmemory programming intervals. Inall other modes the LT1121 is inshutdown mode and consumes only16µA. This LT1121 also provides ther-mal- and current-limiting features toincrease the robustness of the socket.

1325_3.eps

FLTF

FHTF

BATR

FEDO

FMCV

–

+C6

–

+C5

–

+C4

–

+C3

–

+C2

BATP

1.6V

–

+C1

LTF

HTF

TBAT

MCV

VBATT

REG

SENSE

R4

R3

RT

RL

R2

R1

VDDVDD

100mV

900mV

REG

BATTERY DIVIDER

3.072V LINEAR

REGULATOR

+

–


the switch is turned off and the drivecurrent is reduced to the idle level forthe duration of the cycle. At T4, thecycle is complete and repeats fromT0.

Figure 3 is a simplified diagram ofthe switch circuitry used when theswitch is being turned off. The pri-mary path to turn the switch off iscurrent source I1 and transistor Q1.I1 turns on Q1, which turns the switchoff by pulling down on its base. Turn-off time is determined by the currentQ1 pulls from the switch’s base. Q1’scollector current is limited to I1 timesQ1’s hFE. To increase Q1’s collectorcurrent by simply increasing I1 wouldbe a waste of power. This is becausean increase in Q1’s collector currentis beneficial only during the switchturn-off transition time and not theentire time the switch is off.

In order to reduce switch turn-offtime, a turn-off enhancement net-work is used. Referring to Figure 3,the network comprises C1, R1, D1,and D2. When the switch voltage be-gins to rise, its dV/dt feeds anadditional current, I2, into Q1’s base,via R1 and D1. This increases Q1’scollector current and reduces switchturn-off time. This action is sustainedthrough the entire switch turn-offtransition time. R1 limits the amountof current fed back to Q1, and D2discharges C1 when the switch isturned on.

Figure 2. Switch drive current

stray capacitance, turn-on time, turn-off time, and control of drive currentall contribute to AC losses. Althoughthere is an inherent benefit with ahigh-speed process, there are circuitdesign techniques that can furtherdecrease AC losses.

Switch turn-on time can be re-duced by increasing the switchdrive-current level, but having exces-sive drive current after the switch hasturned on is a waste of power. TheLT1372 boosts drive current at switchturn-on time, senses when turn-onhas occurred, and then reduces drivecurrent to a lower level. Figure 2shows the LT1372’s control of switchdrive current. At T0, a new cycle be-gins with the start of oscillator“deadtime” (the time when the switchis guaranteed to be off). Deadtimeprovides a “look ahead” prior to switchturn-on time and allows the drivecurrent to reach the boosted levelbefore the switch is turned on. At T1,the drive current has attained theboosted level and the switch turn-onsignal is set. At T2, the switch hasturned on. The switch drive circuitrysenses this condition and reducesdrive current to the lower level for theduration of switch on time. The lowerlevel of drive current more closelyapproaches the minimum currentrequired to keep the switch on. At T3,

LT1372, continued from page 1

Block DiagramA block diagram of the LT1372 is

shown in Figure 1 (see page 1). Thecircuit includes an internal, low-drop-out regulator, trimmed oscillator,trimmed reference, error amplifier,current amplifier, current compara-tor, and bipolar switch. Other uniqueLT1372 features shown in the dia-gram include a combined shutdownand synchronization function, 5:1 os-cillator frequency shifting network,negative regulation feedback ampli-fier, and switch-drive boost circuitry.

The SwitchIn order to operate effectively at

high frequencies, the switch musthave low AC switching losses. Switch

DESIGN FEATURES

1372_2.eps

OSCILLATOR VOLTAGE

DEAD-TIME

BOOSTED

NOMINAL

SWITCH ON

IDLEIDLE

T0 T1 T2 T3 T4

SWITCH OFF

3-STATE SWITCH

DRIVE CURRENT

SWITCH VOLTAGE

Figure 3. Simplified switch diagram

1372_3.eps

C1R1

D1D2

Q1

SWITCH

VSW

I2

L1VIN

I1

Figure 4. Schematic diagram: LT1372 boostconverter

+

+

1372_4.eps

S/S

5

C1 47µF

VIN 2.7V TO 11V

VOUT 12V

8

2NFB

C2 0.047µF

N/C3

4

OFF

ON

R1 2k

*SUMIDA CD73-100KC SUMIDA (708) 956-0666

R3 1.24k

C3 33µF X2

R2 10.7k

D1 MBRS130T3

VSW

FB

LT1372

VIN

L1* 10µH

VC

1 6, 7

GND


Figure 6. LT1372 Positive-to-negative converter with direct feedback

Boost ConverterThe boost converter in Figure 4

shows a typical LT1372 application.This circuit converts an input volt-age, which can vary from 2.7V to 11V,into a regulated 12V output. Using allsurface-mount components, the en-tire boost converter consumes only0.5 square inches of board space.Figure 5 shows the circuit’s efficiency,which can reach 87% on a 5V input.

The reference voltage on the FB pinis trimmed to 1.24V, and the outputvoltage is set by the R2/R3 resistordivider ratio (VOUT = VREF × (R2/R3 +1). R1 and C2 frequency compensatethe circuit.

Positive-to-Negative Flybackwith Direct Feedback

A unique feature of the LT1372 isits ability to directly regulate nega-tive output voltages. As shown in thepositive-to-negative flyback converter

in Figure 6, only two resistors arerequired to set the output voltage.The reference voltage on the NFB pinis −2VREF, making VOUT = −2VREF ×(R2/R3 + 1). Efficiency for this circuitreaches 72% on a 5V input.

Dual-Output Flyback withOvervoltage Protection

Multiple-output flyback convert-ers offer an economical means ofproducing multiple output voltages,but the power-supply designer mustbe aware of cross-regulation issues,which can cause electrical overstresson the supply and loads. Figure 7 is adual-output flyback converter withovervoltage protection. Typically, inmultiple-output flyback designs, onlyone output is voltage sensed and regu-lated. The remaining outputs are“quasi-regulated” by the turn ratiosof the transformer secondary. Cross

regulation is a function of the trans-former used, and is a measure of howwell the quasi-regulated outputsmaintain regulation under varyingload conditions. For evenly loadedoutputs, as shown in Figure 8, crossregulation can be quite good, butwhen the loads differ greatly, as in

Figure 7. LT1372 Dual-output flyback converter with over-voltage protection

DESIGN FEATURES

OUTPUT CURRENT (mA)

60

90

85

80

75

70

65

95EF

FICI

ENCY

(%)

1000

1372_5.eps

10 100

VOUT = 12V VIN = 9V

VIN = 5V

VIN = 3V

++

1372_6.eps

S/S

FBN/C

5

C1 22µF

VIN 2.7V TO 16V

D1 MBRS130LT3

8

2

1 3

4

3

C2 0.047µF

4

2

OFF

ON

R1 2k

R2 2.49k

R3 2.49k

T1 = COILTRONICS CTX10-2P COILTRONICS (407) 241-7876

C3 47µF

–VOUT –5V

VSW

D2 P6KE-15A

D3 MUR110

NFB

LT1372

T1

VIN

VC

1 6, 7

GND IOUT 0.3A 0.5A

0.75A

VIN 3V 5V 9V

IOUT (mA)

–30

30

25

20

15

10

5

0

–5

–10

–15

–20

–25

V OU

T (V

)

100

1372_9.eps

1 10

VIN = 5V

VOUT

–VOUT

IOUT (mA)

–30

30

25

20

15

10

5

0

–5

–10

–15

–20

–25

V OU

T (V

)

100

1372_8.eps

1 10

VIN = 5V

VOUT

–VOUT

+

++

1372_7.eps

S/S

5

100µF

VIN 2.7V TO 13V

1.24k 13.3k

MBRS140T3

MBRS140T3

8

2, 3

6, 7

1

84

5

3

C2 0.047µF

4

OFF

ON

R1 2k

12.1k

2.49kT1 = DALE LPE-4841-A038 DALE (605) 665-9301

47µF

47µF

–VOUT –15V

VOUT 15V

VSW

P6KE-20A

1N4148

NFB

LT1372

T1

VIN

2

FB

VC

1 6, 7

GND

Figure 5. Efficiency of boost converter shownin Figure 4

Figure 9. Cross regulation of Figure 7’scircuit. −VOUT unloaded; only +VOUT voltagesensed.

Figure 8. Cross regulation of Figure 7’scircuit. +VOUT and −VOUT evenly loaded.


to isolate high-current paths fromlow-current signal paths. We recom-mend that separate power traces andground traces be provided for eachregulator to provide further isolation.High-current ground paths for eachswitch should meet only at the groundstar. The compensation componentsfor each regulator VC pin and theground-referred feedback divider net-works for the LCD contrast convertershould connect directly to the analogground pin for best regulation.

Grounded CCFL withVariable-Negative/Fixed-Positive Contrast

Figure 3 (see page 12) is a completegrounded CCFL circuit withvariable-negative/fixed-positive con-trast-voltage generation based on theLT1182. The circuit is almost identi-cal to that of Figure 2, except that thelamp connects directly to the LT1182DIO pin, thereby closing a feedbackloop at the CCFL VC pin. Also, the BAT

and ROYER pins have been tied to-gether. This ensures that the high-sidesense amplifier is disabled and doesnot generate any feedback signal.Performance is similar to that of thefloating circuit, with efficienciesranging from 85–90% for typicalCCFL lamps at full load. However,the lamp-current range is limited to1–6mA. Thermometering becomespronounced at currents below about1mA. The battery supply range canextend from 8V to 28V, with a nomi-nal battery voltage of 12V. The LT1182is typically powered from a 5V or 3.3Vlogic supply. Programming of lampcurrent in this example has beensimplified to a resistor and a variable0–5V supply.

with a corresponding loss in operat-ing efficiency of about 3%. Lampintensity is smoothly variable fromminimum to full intensity.

The LCD contrast regulator is con-figured as a flyback converter withdivider networks set up to controleither positive or negative contrastvoltages. Grounding the negative sideof the secondary produces a fixed+15V output. Grounding the positiveside of the secondary produces a vari-able −10V to −30V output range at30mA maximum load current. Out-put voltage varies in direct proportionto changes in V(CONTRAST). LCD con-trast efficiency is about 80% at fullpower. The flyback converter runs indiscontinuous mode to minimize thesize of the magnetics.

The LT1182 has separate powergrounds for each switch and an ana-log ground in order to minimizeinteraction between the regulators.Pay special attention to power andground traces on the PC board layout

CCFL, continued from page 12

in Figure 7 eliminates this require-ment. This circuit senses both the+15V and −15V outputs and preventseither from going beyond its regulat-ing value. Figure 10 shows theunloaded −15V output being held con-stant. The circuit’s efficiency, whichcan reach 79% on a 5V input, isshown in Figure 11.

the case of a load disconnect, theremay be trouble. Figure 9 shows thatwhen only the +15V output is voltagesensed, the −15V quasi-regulatedoutput exceeds −25V when unloaded.This can cause electrical overstresson the output capacitor, output di-ode, and the load when reconnected.Adding output-voltage clamps is oneway to fix the problem, but the circuit

ConclusionThe LT1372 does not exhibit the

usual trade-offs between increasedswitching frequency and efficiency.In Figure 12, the efficiency of aLT1372 is compared to an LT1172,switching at 100kHz. The LT1372has comparable efficiencies anduses less board space.

DESIGN FEATURES

References:

1. Williams, Jim. 1992. Illumination Circuitry forLiquid Crystal Displays. Linear Technology Cor-poration, Application Note 49.

2. Williams, Jim. 1993. Techniques for 92% Effi-cient LCD Illumination. Linear TechnologyCorporation, Application Note 55.

1372_12.eps

ILOAD

10mA50

EFFI

CIEN

CY (%

)

60

70

80

90

100

100mA 1A

VIN = 5V VOUT = 12V

LT1372

LT1172

OUTPUT CURRENT (mA)

60

85

80

75

70

65

EFFI

CIEN

CY (%

)

200

1372_11.eps

5 10 100

VOUT = ±15V VIN = 9V

VIN = 5V

VIN = 3V

IOUT (mA)

–30

30

25

20

15

10

5

0

–5

–10

–15

–20

–25

V OU

T (V

)

100

1372_10.eps

1 10

VIN = 5V

VOUT

–VOUT

Figure 10. Cross regulation of Figure 7’scircuit. −VOUT unloaded; both −VOUT and+VOUT sensed.

Figure 11. Efficiency of dual-output flybackconverter in Figure 7.

Figure 12. LT1372 versus LT1172 efficiency


Simple Thermal Analysis — A RealCool Subject for LTC Regulators

Some typical LTC regulators andtheir thermal characteristics areshown in Table 1.

There are several other commonthermal resistance terms:

θCS — thermal resistance from thecase of the package to the heatsink.

θSA — thermal resistance from aheat sink surface to the ambienttemperature.The last two terms are determined

by how a regulator is mounted to theheat sink and by the properties of theheat sink. Heat sinks are used todecrease the thermal resistance andtherefore lower the temperature riseof the regulator.

Temperature is a term with whichwe are all very familiar. All thermalcalculations will use the Centigradescale or °C.

TJ — temperature of the junction ofthe regulator die.

TC — temperature of the case of theregulator.

TA — ambient temperature.The maximum operating junction

temperature, TJ MAX for LTC regula-tors is shown on the device data sheet.

What is Thermal Analysis?The goal of any thermal analysis is

to determine the regulator junctiontemperature, TJ, to ensure that thistemperature is less than either theregulator rating or a design specifica-tion. In the simplest case, temperaturerise is calculated by multiplying thepower times the total of all thermalresistance:

TR = P × θtotal

θTOTAL includes the thermal resis-tance junction-to-case (θJC), thermalresistance case-to-heat sink (θCS), andthermal resistance heat sink-to-am-bient (θSA).

TR represents the temperature riseabove the ambient temperature; there-fore, to determine the actual junctiontemperature of the regulator, theambient temperature must be addedto TR:

Regulator junction temperature =Ambient Temperature + TR

For example, consider a circuitusing an LT1129CT operating in a50°C enclosure with an input voltageof 8VDC, an output voltage of 5VDC,and a load current of 1 ampere1.

The power dissipated by theLT1129CT is:

P = (VIN - VOUT) × ILOAD= (8V - 5V) × 1 amp= 3 watts

The first question is, does this cir-cuit need a heat sink?

Since we have assumed no heatsink on the LT1129CT for the pur-pose of this calculation, we must usethermal resistance from junction toambient, θJA = 50°C/watt.

TJ = P × θJA + TA= 3 watts × 50°C/watt + 50°C= 150°C + 50°C = 200°C

The junction temperature, TJ thatwe just calculated is greater than theLT1129CT’s maximum junction tem-perature specification of 125°C;therefore this circuit must use a heatsink.

Now the task at hand is to calcu-late the correct heat sink to use. Theselected heat sink must hold the junc-tion temperature at less than 125°Cfor the LT1129CT.

As the temperatures go up...so go the problems with voltageregulators.

IntroductionLinear Technology Corporation

applications engineers get lots of callssaying, “that $X%#@& voltage regu-lator is so hot I can’t touch it!” Thepurpose of the article is to show you,the design engineer, how to performsimple thermal calculations to deter-mine regulator temperature and selectthe proper package style and/or heatsink. In addition, it will show an alter-nate method of specifying thermalparameters on LTC voltage regula-tors.

Definition of TermsPower dissipation is the param-

eter that causes a regulator to heatup; the unit for power is watts. Poweris the product of the voltage across alinear regulator times the load cur-rent (see Figure 1).

Thermal resistance is a measure ofthe flow of heat from one surface toanother surface; the unit of thermalresistance is °C/watt. Common termsfor thermal resistance that show upon most LTC data sheets are:

θJC—thermal resistance from thejunction of the die to the case ofthe package.

θJA—thermal resistance from thejunction of the die to the ambienttemperature.

Figure 1. Typical linear regulator circuit

Table 1. θJC and θJA for three LTC regulators.

Device θJC (°C/watt) θJA (°C/watt)

LT1005CT 5.0LT1083MK 1.6LT1129CT 5.0 50

DESIGN INFORMATION

VOUTVIN

ILOAD

PDISS = (VIN – VOUT) ILOAD (WATTS)Cool_1.eps

by Alan Rich


TJ = P × θTOTAL + TA125°C = 3 watts × θTOTAL + 50°CθTOTAL = 25°C/watt and,θTOTAL = θJC + θCS + θSA

For this configuration:

θJC = 5°C/watt (LT1129CT data sheet)θCS = 0.2°C/watt (typical for heat sinkmounting)θSA = heat sink specification

Plugging in these numbers:

25°C/W = 5°C/W + 0.2°C/W + θSAθSA = 19.8°C/watt

Therefore, the heat sink selectedmust have a thermal resistance ofless than 19.8°C/watt to hold theLT1129CT junction temperature atless than 125°C. Obviously, the lowerthe heat sink thermal resistance,the lower the LT1129CT junctiontemperature. A lower junction tem-perature will increase reliability.

Now, let’s consider a circuit usingan LT1129CT operating in a 50°Cenclosure with an input voltage ofonly 6VDC, an output voltage of 5VDC,and a load current of 1 ampere.

The power dissipated by the LT1129CTis:

P = (VIN - VOUT) × ILOAD= (6V - 5V) × 1 amp= 1 watt

Does this circuit need a heat sink?Again, for the purposes of the cal-culation, we must use thermalresistance from junction to ambient,θJA= 50°C/watt for the LT1129CT.

TJ = P × θJA + TA= 1 watt × 50°C/watt + 50°C= 50°C + 50°C = 100°C

The junction temperature, TJ thatwe just calculated is now less thanthe LT1129CT’s maximum junctiontemperature specification of 125°C.Therefore this circuit does not need aheat sink. This illustrates the advan-tage of a low dropout regulator likethe LT1129CT.

An AlternativeMethod for SpecifyingThermal Parameters

Linear Technology Corp. has intro-duced an alternative method to specifyand calculate thermal parameters ofvoltage regulators. Previous regula-tors, with a single thermal resistancejunction-to-case (θJC), used an aver-age of temperature rise of the controland power sections. This could easilyallow excessive junction temperatureunder certain conditions of ambienttemperature and heat sink thermalresistance.

Several LTC voltage regulatorsinclude thermal resistance andmaximum junction temperaturespecifications for both the control andpower sections, as shown inTable 2.

As an example, let’s calculate thejunction temperature for the sameapplication shown before, using anLT1085CT instead of the LT1129CT.Once again, we are operating in a50°C enclosure; the input voltage is8VDC, the output voltage is 5VDC,and the load current is 1 ampere.

The power dissipated by theLT1085CT is the same as before, 3watts. We will assume we have se-lected a heat sink with a thermalresistance, θSA of 10°C/watt. Firstcalculate the control section of theLT1085CT:

θJC = 0.7°C/watt (LT1085CT data sheet)θCA = 0.2°C/watt (typical)θSA = 10°C/watt

θTOTAL = θJC + θCA + θSA= 0.7°C/watt + 0.2°C/watt + 10°C/watt= 10.9°C/watt

To determine the control section junc-tion temperature:

TJ = P × θTOTAL + TA= 3 watts × 10.9°C/watt + 50°C= 82.7°C (TJ MAX = 125°C)

To calculate the power section of theLT1085CT:

θJC = 3°C/watt (LT1085CT data sheet)θTOTAL = θJC + θCA + θSA= 3°C/W + 0.2°C/watt + 10°C/watt= 13.2°C/watt

To determine the power section junc-tion temperature:

TJ = P × θTOTAL + TA= 3 watts × 13.2°C/watt + 50°C= 89.6°C (TJ MAX = 150°C)

In both cases, the junction tem-perature is below the maximum ratingfor the respective section; this en-sures reliable operation.

ConclusionThis article is an introduction to

thermal analysis for voltage regula-tors; however, the techniquesalso apply to other devices, includingoperational amplifiers, voltage refer-ences, resistors, and the like. For themore advanced student of thermalanalysis, it can be shown that there isa direct analogy between electroniccircuit analysis and thermal analysisas shown in Table 3.

All standard electronic networkanalysis techniques (Kirchhoff’s laws,Ohm’s law) and computer circuitanalysis programs (SPICE) can beapplied to complex thermal systems.

DESIGN INFORMATION

1 The LTC1192CT is guaranteed for 700mA, butcould be selected to output 1 Amp.

Thermal World Electrical WorldPower Current

Temperature differences VoltageThermal resistance Resistance

Table 3. Analogy between thermal analysisand electronic circuit analysis

Control PowerDevice θJC TJ MAX θJC TJ MAXLT1083MK 0.6°C/W 150°C 1.6°C/W 200°CLT1085CT 0.7°C/W 125°C 3.0°C/W 150°C

Table 2. Two examples showing thermalresistance of control and power sections ofLTC regulators.


Low-Noise Portable CommunicationsDC-to-DC Converter

DESIGN IDEAS

Portable communications productspack plenty of parts into close prox-imity. Digital clock noise must beeliminated not only from the audiosections, but also from the antenna,which, by the very nature of the prod-uct, is located only inches from activecircuitry. If a switching regulator isused in the power supply, it becomesanother potential source of noise. TheLTC1174 stepdown converter is de-signed specifically to eliminate noiseat audio frequencies while maintain-ing high efficiency at low outputcurrents.

Figure 1 shows an all-surface-mount solution for a 5V, 120mAoutput derived from five to sevenNiCad or NiMH cells. Small input andoutput capacitors are used to con-serve space, without sacrificingreliability. In applications where it isdesired, a shutdown feature is avail-able; otherwise short this pin to VIN.

The LTC1174’s internal switch,which is connected between VIN andVSW, is current controlled at a peakthreshold of approximately 340mA.This low peak threshold is one of thekey features that allows the LTC1174

DESIGN IDEAS

Low-Noise PortableCommunications DC-to-DCConverter....................... 25Mitchell Lee

LT1585: New LinearRegulator Solves Load Tran-sients ............................ 26Craig Varga

Luma Keying with theLT1203 Video Multiplexer...................................... 27Frank Cox

A Tachless Motor-SpeedRegulator ...................... 28Mitchell Lee

Figure 1. Low-noise, high-efficiency, stepdown regulator for personal communications devices

to minimize system noise comparedto other chips that carry significantlyhigher peak currents, easing shield-ing and filtering requirements anddecreasing component stress.

To conserve power and maintainhigh efficiency at light loads, theLTC1174 uses Burst ModeTM opera-tion. Unfortunately, this controlscheme can also generate audio-fre-quency noise at both light and heavyloads. In addition to electrical noise,acoustical noise can emanate fromcapacitors and coils under these con-ditions. A feedforward capacitor (C3)shifts the noise spectrum up out ofthe audio band, eliminating theseproblems. C3 also reduces peak-to-peak output ripple, which measuresapproximately 30mV over the entireload range.

The interactions of load current,efficiency, and operating frequencyare shown in Figure 2. High efficiencyis maintained at even low currentlevels, dropping below 70% at around800µA. No-load supply current is lessthan 200µA, dropping to approxi-mately 1µA in shutdown mode. Theoperating frequency rises above thetelephony bandwidth of 3kHz at a

load of 1.2mA. Most products drawsuch low load currents only in standbymode with the audio circuitssquelched, when noise is not anissue.

The frequency curve depicted inFigure 2 was measured with a spec-trum analyzer, not a counter. Thisensures that the lowest-frequencynoise peak is observed, rather than afaster switching frequency compo-nent. Any tendency to generatesubharmonic noise is quickly exposedusing this measurement method.

dI1174_1.eps

5-7 CELL INPUT

SHUTDOWN

IPGM

C1 = PANASONIC SP SERIES (201-348-4630) C3 = AVX TPS SERIES (803-946-0690) L1 = COILTRONICS OCTAPAK (407-241-7876)

MBR0520L C3 33µF/20V

L1 CTX33-1

ON

OFF C2 6.8nF

5V/120mA OUTPUT

LTC1174CS8

VSW

FB

VIN

GND

C1 15µF/12.5V

R1 91k

R2 30k

+

+

OUTPUT CURRENT (mA)

0.1

1

10

100

50

90

80

70

60

100

FREQ

UEN

CY (k

Hz) EFFICIEN

CY

100

dI1174_2.eps

0.1 1 10

by Mitchell Lee

Figure 2. Parameter interaction chart forFigure 1’s circuit.


LT1585: New Linear RegulatorSolves Load Transients

and the equivalent series inductance(ESL) from the output capacitor(s)terminal(s) to the load connection.Note that these contributions alsoinclude the lead trace parasitics fromthe capacitor(s) to the load.

The resistive component is simply∆I × ESR. The droop to point A,23.6mV, is the ESR contribution.Calculating ESR:

23.6mV/3.8 Amps = 6.2mΩ

full load current. This is the inflectionpoint in the curve. Since the regulatornow measures the output voltage asbeing too low, it overshoots the loadcurrent and recharges the outputcapacitors to the correct voltage.

Faster RegulatorMeans Fewer Capacitors,Less Board Space

The regulator has one major effecton the system’s transient behavior.The faster the regulator, the less bulkcapacitance is needed to keep thedroop from becoming excessive. It ishere that the advantage of the LT1585shows up. The response time of theLT1585 is about one-half that of thelast generation three-terminalregulators.

The response in the first severalhundred nanoseconds is controlledby the careful placement of bypasscapacitors. Figure 3 is a schematicdiagram of the circuit, but the layoutis critical, so consult the LTC factoryfor circuit and layout information.

The latest hot new microproces-sors have added a significantcomplication to the design of the powersupplies that feed them. These de-vices have the ability to switch fromconsuming very little power to requir-ing several amps in tens ofnanoseconds. To add a furthercomplication, they are extremely in-tolerant of supply voltage variations.Gone are the days of the popcornthree-terminal regulator and the0.1µF decoupling capacitor. TheLT1585 is the first low-dropout regu-lator specifically designed for tightoutput voltage tolerance (optimizedfor the latest generation processors)and fast transient response.

Figure 1 shows the kind of re-sponse that can and must be achievedif these microprocessors are to oper-ate reliably. Figure 2 details the firstseveral microseconds of the transientin Figure 1. The load change in thiscase is 3.8 Amps in about 20ns. Twoparasitic elements dominate the tran-sient performance of the system. Bothare controlled by the type, quantity,and location of the decoupling ca-pacitors in the system.

Anatomy of aLoad Transient.

The instantaneous droop at theleading edge of the transient is theresult of the sum of the effects of theequivalent series resistance (ESR)

Figure 3. Schematic Diagram: LT1585 responding to fast transients

Figure 2. Detailed sketch of first few microseconds of transient

The effects of inductance are pre-dicted by the formula V = LdI/dt. Thevoltage from point A to the bottom ofthe trough is the inductive contribu-tion (13.4mV). ESL is calculated to be0.07nH. After the load current stopsrising the inductive effects end, bring-ing the voltage to point B. At thispoint the curve settles into a gentledroop. The droop rate is dV/dt = I/C.There is about 1300µF of useful ca-pacitance on the board in this case(see Figure 3). As the regulator out-put current starts to approach thenew load current, the droop rate less-ens until the regulator supplies the

DESIGN IDEAS

Figure 1. Transient response of 200mA to 4Amp load step

dI1585_2.eps

23.6mV

13.4mV

“A”

NOMINAL VO

“B”

dI1585_3.eps

VOUT5V

C1 TO C2 220µF

10V SANYO OSCON

× 2

GND

U1 LT1585CT

VIN3

2

1

LOAD+ C9 TO C18

1µF SMD × 10

C3 TO C8 220µF 10V AVX TYPE TPS × 6

+

by Craig Varga


Luma Keying with theLT1203 Video Multiplexer

In video systems, the action ofswitching between two or more activevideo sources is referred to as a “wipe”or a “key.” When the decision to switchvideo sources is based on an attributeof the active video itself, the action iscalled keying. A wipe is controlled bya non-video signal, such as a ramp.The circuit presented in Figure 1 isreferred to as a “luma key” because itswitches between two sources whenthe luminance (“luma”) of a mono-chrome key signal reaches a set level.It is also possible to key on the colorof the video source and this, not sur-prisingly, this is called “chromakeying.”

Figure 1’s operation is very straight-forward. A monochrome video sourceis used to generate the key signal. TheLT1363 is used as a buffer and maynot be needed in all applications. If

coherent with the color reference ofthe sources. The key-source videoshould be monochrome to preventthe key comparator from switchingon the color subcarrier.

the key signal is to be used as one ofthe switched signals, it is convenientto “loop through” the input of thisbuffer. The LT1016 comparatorswitches when the video level exceedsthe DC reference on its inverting in-put, which is controlled by the “keysensitivity” control. The TTL keysignal controls an LT1203 video mul-tiplexer. Any two video sources maybe connected to the inputs of theLT1203, as long as they are gen lockedand within the common-mode range(on ±5 volt supplies this is ±3 voltsover 0–70°C) of the multiplexer. TheLT1203’s fast switching speed, lowoffset, and clean switching make it anatural for an active video switchingapplication like this one. Compositecolor signals can be used, but thebest results will be obtained if thekey signal’s horizontal sync is phase

Figure 1. Schematic diagram: luma keyer

DESIGN IDEAS

COMPOSITE BLANKING (TTL LEVELS)

dI1203_1.eps

–

+

–

+

KEY VIDEO

10k LT1363

LT1016

3

2

2

3

46

5

7

1000Ω

1000Ω

6

–5

15V

LT1004 - 2.5

+5

INPUT

300Ω232pF36.5pF

10k

KEY SENSITIVITY

100k

SOURCE 1 IN1

GRD

IN2 NC

1

LT1203

75Ω

75Ω

1000Ω

–

+LT1363

3

2

6 75Ω

2

SOURCE 23

LOGIC

OUT

EW

5

7

6

IN1

GRD

IN2 NC

1

LT12032

3

LOGIC

OUT

EW

5

7

6

1000Ω

1000Ω

510Ω

VIDEO OUT

510Ω2.2k

5kCOMPOSITE SYNC (TTL LEVELS)

300Ω74AC04

9.3µH

by Frank Cox

Figure 2. Two-level image of IC designer


A common requirement in manymotor applications is a means ofmaintaining constant speed with vari-able loading or variable supply voltage.Speed control is easily implementedusing tachometer feedback, but thecost of a tach may be prohibitive inmany situations, and adds mechani-cal complications to the product. Alower-cost solution with no movingparts is presented here.

Motor speed changes under condi-tions of varying loads because of theeffects of series loss terms in themotor. The effects of the predominant

contributors to loss, copper andbrush/commutator resistance (col-lectively known as RM), are bestunderstood by considering the cir-cuit model for a motor (see Figure 1).A motor’s back-EMF (VM) is propor-tional to speed (n), and the motorcurrent (IM) is proportional to theload torque (T). The following equa-tion predicts the speed of the motor,for any given condition of loading:

where KV and KT are constants ofproportionality for rotational velocityand torque. For a fixed terminal volt-age, the speed of the motor mustdecrease as increasing load torque isapplied to the shaft. For a fixed load,the speed of the motor will also changeif the supply (terminal) voltage ischanged.

A voltage regulator fixes the prob-lem of a varying terminal voltage, butthe only way to eliminate torque fromEquation (1) is by reducing RM to

Nonstandard video signals can beused for the inputs to the LT1203.For instance, it is possible to selectbetween two DC input levels to con-struct a two-level image. Figure 2 isan example of an image constructedthis way. A monochrome video signalis sliced and used to key betweenblack (zero volts) and gray (approxi-mately 0.5 volt) to generate this imageof a famous linear IC designer. Animage formed in this way is not astandard video output until the blank-

ing and sync intervals are recon-structed. The second LT1203 blanksthe video and an LT1363 circuit sumscomposite sync to the video and drivesa cable. For more information on thispart of the circuit, see AN 57, page 7.A clamp is not used, as the DC levelsare arbitrarily set by the inputs, butone could be used, as in the figure onpage 7 of AN57, if the sources werevideo. As another option, Figure 3shows the same key signal used asone of the inputs to the multiplexer.

Figure 3. Key signal used as input to the MUX

A Tachless Motor-Speed Regulator

zero. Physically this is impractical,but an electrical solution exists.

If a motor is driven from a regu-lated source whose output impedanceis opposite in sign and equal in mag-nitude to RM (see Figure 1), the resultis a motor that runs at a constantspeed—regardless of loading andpower-source variations. Figure 2shows a circuit that does it all. TheLT1170 is configured as a buck-boostconverter, which can take a wide-ranging 3-to-20V input source andproduce a regulated output of, say,6V. The circuit shown can deliver 1Aat 6V with a 5V input, adequate formany small, permanent-magnet DCmotors.

To cancel the effects of the motorresistance, a negative output imped-ance is introduced with an op ampand a current-sense resistor (RS). Asthe motor current increases, theLT1006 responds by increasing themotor terminal voltage by an amountequal to IM × RM. Depending on thevalue of R3, the speed can be made toincrease, decrease, or stay the sameunder load. If R3 is just right, the

Figure 1. On the right is shown an equivalentcircuit for a motor. On the left is the modelfor a circuit which will stabilize the motor’sspeed against changes in supply voltage andloading.

VTERMINAL RM

KV KT × KV

T ×n = − (1)

by Mitchell Lee

DESIGN IDEAS

dImotr_1.eps

VTERMINAL

MOTORSPEED REGULATOR

RM

IM = T/KT

VM = KV •nVREF

+–

+–

R RR

1 21

+

− •R RR

S23


motor speed will remain constant untilthe LT1170 reaches full power andthe circuit runs out of steam.

Many small motors in the 1-to-10Wclass are not well characterized. Inorder to choose proper componentvalues for a given motor, figures forRM and VM are necessary. Fortunately,these are easily measured using aDVM and a motor-characterizationtest stand. If you don’t have a motor-characterization test stand, it is alsopossible to use a lathe or drill press todo the job.

Chuck-up the candidate motor’sshaft in a variable speed drill press orlathe, which is set to run at the samespeed you’re intending to operate themotor. Clamp down the motor frameso it won’t spin. Turn on the bigmachine, and measure the open-cir-cuit motor terminals with a DVM.This is the motor voltage, VM, as shownin Figure 1. Switch the meter to mea-sure the motor’s short circuit current,ISC. Motor resistance RM = VM/ISC.With these figures the other compo-nent values can be calculated:

IMAX = motor current at full loadVREF = 1.244VR1 = series combination of 619Ω + 619Ω =1238ΩRS ≤ 1/IMAX (drops less than 1V atmaximum load)

R2 = (VM × R1/VREF) - R1 (2)

R3 = (R2 × RS)/(RM + RS) (3)

loading from the motor. Check themotor’s unloaded speed, and adjustR2 if necessary to set it precisely.

With the motor driving a nominalload, decrease R3 until the motorcommences “hunting.” R3 will be nearthe nominal calculated value. Thisthreshold is very close to optimummotor-resistance cancellation. R5and C5 offer a convenient means ofcompensating for frictional and iner-tial effects in the mechanical system,eliminating instabilities. System sta-bility should be evaluated under avariety of loading conditions. The ef-fect of R5 is to reduce the negativeoutput impedance of the circuit athigh frequencies. Systems with a netpositive impedance are inherentlystable.

When the system stability is satis-factory, a final adjustment of R3 canbe made to achieve the desired speedregulation under conditions of vary-ing loads. These final values can beused in production. Note that R2 de-fines the regulated speed value andmay be production trimmed in preci-sion applications.

Figure 2. Tachless motor-speed regulator

The component values shown inFigure 2 are for a small motor withthe following characteristics mea-sured at 360RPM: VM = 7.8V, ISC =3.7A, RM = 2.1Ω, IMAX ≈ 1A.

RS, a copper resistor, is either lo-cated close to or wound around themotor to assist in tracking changes inarmature resistance with tempera-ture. Copper has a strong, 3930ppm/°C temperature coefficient, matchingthe TC of the motor winding.

Setup ProcedureInitial tests should be performed

with a potentiometer in place of, andtwice the value of, R3. R5 and C5should be disconnected; remove all

DESIGN IDEAS

–

+

dImotr_2.eps

+

–

3V TO 20V INPUT

C1 1000µF

C3 1000µF

C2 330µF

C4 1µF

L1 = L2 = 50-2-52 COILTRONICS (407-241-7876). CAN BE COUPLED AS INDICATED BY PHASING DOTS. LT1006 POWER SUPPLY PINS CONNECTED TO INPUT SUPPLY.

L1 50µH

LT1170

LT1006

FB

VSW

MBRD340CT

R3 309Ω 1%

VIN

VCGND

R2 6490Ω 1%

R1a 619Ω 1%

10k 1%

R5 1k

C5 2.2µF

R1b 619Ω 1%

10k 1%

R4 1k

L2 50µH

RS 100mΩ Cu

RM

VMM

+

++


New Device Cameos NEW DEVICE CAMEOS

LTC1147L-3.3 and LTC1148L-3.3: Extremely Low DropoutStepdown Controllers

The LTC1147L-3.3 and LTC1148L-3.3 are the industry’s first 3.3V,low-dropout, stepdown switchingregulator controllers. The LTC1148L-3.3 is a synchronous switchingregulator controller availablein a compact 14-lead SOIC. TheLTC1147L-3.3 is a non-synchronousswitching regulator controller, avail-able in a space-saving 8-lead SOIC.Both controllers operate down to aminimum input voltage of less than3.6V, providing extremely low drop-out operation. Battery life is extendedby providing high efficiencies at loadcurrents from milliamps (when thedevice is in standby or sleep mode) toamps (under full power conditions).The 100% duty cycle characteristic indropout mode allows maximum en-ergy to be extracted from the batterypack, further extending operatingtime. Both devices use current-modearchitecture with Burst ModeTM op-eration to provide extremely highefficiency over the entire load rangedemanded by the next generation ofnotebook, pen-based, and handheldcomputers.

LT1312/LT1313Micropower Single andDual 120mA PCMCIA VPPRegulators/Drivers

The LT1312 single and LT1313 dualregulators power and protect the VPPpins of standard PCMCIA card slots.Built-in current limit and thermalshutdown ensure that a short-circuitcondition on either side of a standardPCMCIA socket will not damage thehost computer or PC card throughthe VPP pins. Further, the LT1312and LT1313 produce clean, tightlyregulated 0V, 3.3V, 5V, 12V, andHi-Z outputs from any available un-regulated 13V to 20V supply, thuseliminating the cost of an extra 12Vregulator to generate clean VPP power.Direct connection with industry-

standard PCMCIA interface control-lers is provided through compatibledigital inputs.

Both regulators boast micropoweroperation, with the quiescent currentfalling to just 30 microamps for theLT1312 and 60 microamps for theLT1313, when programmed to the 0Vor Hi-Z modes. A single 1 microfaradcapacitor is the only external compo-nent required to ensure smoothtransitions between operating volt-ages. Both parts have no overshootwhen supplying 12V to sensitive flashmemory cards.

Full 3.3V/5V compatibility is pro-vided by an internal comparator thatcontinuously monitors the PC cardVCC supply and adjusts the regulatedVPP output to match VCC when theVPP = VCC mode is selected. Bothdevices include VPP valid status out-puts to indicate when the outputvoltage is in regulation at 12V.

The combination of micropoweroperation and full card and socketprotection make these parts idealsolutions for battery powered appli-cations. The LT1312 and LT1313 willfind application in palmtop, note-book, and desktop computers,instruments, handi-terminals, andbar-code readers.

LTC1314/LTC1315 Singleand Dual PCMCIA VPPSwitching Matrix with Built-in VCC N-Channel Gate Drivers

The LTC1314 single and LTC1315dual power controllers provide all theswitching necessary to control one ortwo PCMCIA card slots. Both devicesboast 120mA VPP output currentcapability and provide 0V, VCC, 12V,and Hi-Z output states. VCC switchingbetween OFF, 3.3V, and 5V is accom-plished with inexpensive low RDS(ON)N-channel switches driven by an in-ternal voltage multiplier, whichgenerates 12V gate drive from the 5Vlogic supply. Both the LTC1314 andLTC1315 have logic-compatible in-puts that interface directly with

industry standard PCMCIA cardcontrollers.

Both the LTC1314 and LTC1315are designed for micropower opera-tion and draw only 1 microamp whenprogrammed to the Hi-Z or 0V oper-ating modes. Built-in charge pumps,which drive both the internal andexternal N-FET switches, eliminatethe need for continuous 12V power.In fact, the external 12V supply isonly required when the VPP = 12Vmode is selected. When not needed,the external 12V supply is shut downby a logic-compatible signal from theLTC1314, further reducing systempower consumption.

Internal break-before-make switchaction and ramped gate drive ensurethat the outputs of the LTC1314 andLTC1315 move smoothly betweenoperating voltages, with no switchoverlap or VPP overshoot. Small by-pass capacitors are the only externalcomponents required.

The combination of efficient opera-tion and full card and socket controlmake the LTC1314/LTC1315 idealsolutions for battery-powered appli-cations. These parts will findapplication in palmtop, notebook, anddesktop computers, instruments,handi-terminals, and bar -codereaders.

LTC1390: Serial-Controlled,Eight-channel AnalogMultiplexer

The LTC1390 is a low cost, eight-channel analog multiplexer featuringeight individual switches controlledby a three-wire digital interface.It has a bidirectional data retrans-mission capability, allowing it to bewired in series with a serial A/D con-verter using only one serial port. Theinterface also allows multi-input ex-pansion with the connection of severalLTC1390s, using only a single digitalport.

The specifications for each analogchannel achieved at bipolar ±5V sup-ply are full analog signal range, 75Ω


NEW DEVICE CAMEOS

maximum on-resistance (RON), ±5nanoamp maximum ON and OFFchannel leakage current, and ±10picocoulomb maximum charge injec-tion. For dynamic switching betweenchannels, a guaranteed break-before-make operation is ensured. All pinscan withstand repeated ±2kV ESDstrikes and 100mA fault currentswithout latching-up. All digital in-puts are TTL and CMOS compatible.The LTC1390 is also provided with afull set of specifications for operationon a +3V supply. The LTC1390 isavailable in 16-lead plastic DIP andSOIC packages.

LTC1348 Ultra-low-power,3.3V-Powered Three-driver, Five-receiverTransceiver withTrue RS232 Output Levels

The LTC1348 is a new three-driver,five-receiver RS232 interface trans-ceiver that supplies true RS232output-voltage levels when operatedfrom a single 3.3V power supply. Theintegral charge-pump power genera-tor uses three 0.1µF commutatingcapacitors to generate ±8V powersupplies for the driver stages. Thecharge pump provides sufficient out-put current to drive a serial-portmouse and 3kΩ2500pF loads atdata rates up to 120kbaud.

Power consumption is only 500µAin normal operation. Driver-enableand receiver-enable controls provideflexible power management capabil-ity to reduce power to as little as0.2µA when the circuit is shut down.Receiver and driver outputs are highimpedance when disabled or withpower off. Driver outputs and receiverinputs are tolerant of the full ±25 voltlevels that can occur on RS232 datalines.

Like all Linear Technology RS232transceivers, the LTC1348 is protectedagainst ±10kV ESD strikes to theRS232 inputs and outputs. This inte-grated ESD resistance saves theexpense and space of external protec-tion devices.

The circuit is available in 28-leadDIP, SOIC, and SSOP packages. All

applications such as High-bit-rateDigital Subscriber Line (HDSL).Whereas the recently introducedLTC1278-4’s 400ksps conversion rateeasily covers T1 data rates, theLTC1279’s 600ksps is ideal for HDSL’sfaster E1 data rates. The parallel databus provides an excellent, yet flexibleand easy, interface to popular DSPs.The analog-signal input range is 0Vto 5V when operating on a single 5Vpower-supply voltage or ±2.5V whenoperating on a split ±5V power-sup-ply voltage.

The LTC1279’s higher speed isachieved without sacrificing dynamicconversion characteristics. The im-proved S/(N+D) (SINAD) is a minimumof 70dB at fS = 600ksps and fIN =100kHz. The high-performance con-version characteristics ensure thatthe bit-error rate is extremely low(<<10−12).

Other key LTC1279 specificationsinclude differential and integral lin-earity errors of ±1LSB (max), THD of−78dB (max), and a typical 4MHzfull-power bandwidth, ideal forundersampling applications.

Key features include low power dis-sipation of just 75mW (fS = 600ksps,+5V or ±5V supply voltage) duringoperation and just 5mW when powershutdown is active. The LTC1279 in-cludes an internal voltage referenceand conversion clock with synchro-nizing circuitry. The LTC1279 isavailable in a small-footprint SOICpackage that results in efficient useof circuit-board space.

package configurations use a flow-through pinout to simplify PC boardlayout.

LTC1261: RegulatedSwitched-Capacitor Inverter

Designed to generate a regulatednegative supply from a low positivevoltage, the LTC1261 is ideal for pro-viding bias voltages for GaAs FET RFtransmitters and other circuits thatrequire a precisely regulated negativesupply. The heart of the LTC1261 is acharge pump that can both doubleand invert the input voltage, allowingit to generate a −5V output from a +3Vinput. An on-board PWM regulationloop keeps the output within 5% ofthe set value at all times, with up to15mA load current, even with varia-tions in the input voltage. Outputripple is typically below 5mV.

An open-drain output monitors theregulation loop and pulls down toindicate that the LTC1261 is in regu-lation, allowing external circuitry tomonitor the output without additionalcomponents. A shutdown pin bringsthe output voltage to zero and cutsthe quiescent current to only 5 mi-croamps. An on-board resistor stringallows the LTC1261CS14 to provide−3.5V, −4V, −4.5 V and −5V outputswith simple pin connections; otheroutput voltages can be set using ex-ternal resistors. The LTC1261CS8 isavailable with adjustable and fixed−4V and −4.5V options.

The LTC1261CS14 is available in a14-pin narrow SO package and re-quires only one or two external 0.1µFcapacitors for operation, dependingon the input voltage; input and out-put bypass capacitors are alsorequired. The LTC1261CS8 offers allthe same features with a standardinverting-only charge pump in an SO8package.

LTC1279: 12-Bit,600ksps Sampling Analog-to-Digital Converter

The LTC1279 is the second in aseries of high-speed, high-resolutionanalog-to-digital converters optimizedfor telecom digital-data transmission

Burst ModeTM is a trademark of LinearTechnology Corporation.

Information furnished by Technology Cor-poration is believed to be accurate and reliable.However, Linear Technology makes no repre-sentation that the circuits described hereinwill not infringe on existing patent rights.

For further information on theabove or any of the other devicesmentioned in this issue of LinearTechnology, use the reader servicecard or call the LTC literature ser-vice number: 1-800-4-LINEAR. Askfor the pertinent data sheets andapplication notes.


AppleTalk is a registered trademark of Apple Computer, Inc.

© 1994 Linear Technology Corporation/ Printed in U.S.A./27K

LINEAR TECHNOLOGY CORPORATION1630 McCarthy BoulevardMilpitas, CA 95035-7487

(408) 432-1900Literature Department 1-800-4-LINEAR

DESIGN TOOLSApplications on DiskNOISE DISKThis IBM-PC (or compatible) progam allows the user to calculate circuit noiseusing LTC op amps, determine the best LTC op amp for a low noise application,display the noise data for LTC op amps, calculate resistor noise, and calculatenoise using specs for any op amp. Available at no charge.

SPICE MACROMODEL DISKThis IBM-PC (or compatible) high density diskette contains the library of LTCop amp SPICE macromodels. The models can be used with any version ofSPICE for general analog circuit simulations. The diskette also containsworking circuit examples using the models, and a demonstration copy ofPSPICETM by MicroSim. Available at no charge.

Technical Books1990 Linear Databook — This 1440 page collection of data sheets covers opamps, voltage regulators, references, comparators, filters, PWMs, data con-version and interface products (bipolar and CMOS), in both commercial andmilitary grades. The catalog features well over 300 devices. $10.00

1992 Linear Databook Supplement — This 1248 page supplement to the1990 Linear Databook is a collection of all products introduced since then.The catalog contains full data sheets for over 140 devices. The 1992 LinearDatabook Supplement is a companion to the 1990 Linear Databook, whichshould not be discarded. $10.00

1994 Linear Databook, Volume III — This 1826 page supplement to the 1990Linear Databook and 1992 Linear Databook Supplement is a collection ofall products introduced since 1992. A total of 152 product data sheets areincluded with updated selection guides. The 1994 Linear Databook Volume IIIis a supplement to the 1990 and 1992 Databooks, which should not bediscarded. $10.00

Linear Applications Handbook — 928 pages full of application ideascovered in depth by 40 Application Notes and 33 Design Notes. This catalogcovers a broad range of “real world” linear circuitry. In addition to detailed,systems-oriented circuits, this handbook contains broad tutorial contenttogether with liberal use of schematics and scope photography. A specialfeature in this edition includes a 22 page section on SPICE macromodels.

$20.00

1993 Linear Applications Handbook Volume II — Continues the streamof “real world” linear circuitry initiated by the 1990 Handbook. Similar in scopeto the 1990 edition, the new book covers Application Notes 41 through 54 andDesign Notes 33 through 69. Additionally, references and articles from non-LTC publications that we have found useful are also included. $20.00

Interface Product Handbook — This 336 page handbook features LTC’scomplete line of line driver and receiver products for RS232, RS485, RS423,RS422 and AppleTalk applications. Linear’s particular expertise in thisarea involves low power consumption, high numbers of drivers and receiversin one package, 10kV ESD protection of RS232 devices and surface mountpackages. Available at no charge.

Monolithic Filter Handbook — This 234 page book comes with a disk whichruns on PCs. Together, the book and disk assist in the selection, design andimplementation of the right switched capacitor filter circuit. The disk containsstandard filter responses as well as a custom mode. The handbook containsover 20 data sheets, Design Notes and Application Notes. $40.00

SwitcherCAD Handbook — This 144 page manual, including disk, guidesthe user through SwitcherCAD—a powerful PC software tool which aids in thedesign and optimization of switching regulators. The program can cut days offthe design cycle by selecting topologies, calculating operating points andspecifying component values and manufacturer's part numbers. $20.00

1994 Power Solutions Brochure — This 52 page collection of circuitscontains real-life solutions for common power supply design problems. Thereare over 45 circuits, including descriptions, graphs and performance specifi-cations. Topics covered include micropower DC/DC, step-up and step-downswitching regulators, off-line switching regulators, linear regulators, switchedcapacitor conversion and power management. Available at no charge.

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