LINEAR TECHNOLOGY LINEAR TECHNOLOGY...The LT1248 Power-Factor Corrector by Carl Nelson 1248_1.eps...

JUNE 1993 VOLUME III NUMBER 2

The LT1206:A 60MHz, 250mACurrent-Feedback Amplifier

IN THIS ISSUE . . .

COVER ARTICLEThe LT1206: A 60MHz,250mA Current-FeedbackAmplifier ......................... 1William Jett

Editor's Page ................... 2Richard Markell

DESIGN FEATURESThe LT1248Power-Factor Corrector.... 3Carl Nelson

LTC1174: A High-EfficiencyBuck Converter ................ 6San-Hwa Chee and Randy Flatness

The New SO8 LTC1147Switching-RegulatorController OffersHigh Efficiency in aSmall Footprint ............... 9Randy Flatness

DESIGN INFORMATIONUltra-Low-Power CMOSRS232 Transceiver Achieves10kV ESD Protection,Eliminates Latch-Up ...... 14Ricky Chow and Robert Reay

DESIGN IDEAS ........... 15-25(complete listing on p.15)

New Device Cameos ....... 26

LTC in the News ............ 27

Design Tools .................. 28

Sales Offices ................. 28–

+

RF

RG RC NETWORK (SNUBBER)

VIN

SERIES R

CL

OUTPUT

1206_1.eps

IntroductionThe LT1206 current feedback

amplifier is Linear Technology’shigh-speed solution for driving low-impedance loads. The part combines60MHz bandwidth with a guaranteed250mA output current, operation with±5V to ±15V supplies, and optionalcompensation for capacitive loads,making it well suited for driving mul-tiple cables and other difficult loads.A shutdown feature drops the supplycurrent to less than 100µA when thepart is turned off. Thermal shutdownand short-circuit protection are in-cluded in the device. The LT1206 isavailable in a variety of packagesranging from the surface-mount SO8to the 7-pin TO220 power package.

Capacitive LoadsDriving capacitive loads presents

two challenges to the IC designer:maintaining the stability of the am-plifier loop and supplying enoughcurrent to slew the capacitor. Theeffect of a load capacitance can be

thought of in terms of the stability ofthe output stage and that of theentire loop. A capacitive load cancause the output stage of an amplifierto peak or even to oscillate. The effectof this peaking on the overall re-sponse depends on the feedbackcomponents. Sometimes additionalnetworks are used to reduce the peak-ing; a resistor added in series with theload capacitance will isolate the out-put stage, or a series RC snubbernetwork added in parallel with theload will swamp out the load capaci-tance (Figure 1). The first approachcauses an increase in output imped-ance and reduction of swing. In thesecond approach, power is wasted inthe snubber network, reducing thecurrent available to slew the loadcapacitance. The LT1206 uses anoptional internal network connectedto an additional pin to compensatefor capacitive loads. Adding a 0.01µFbypass cap between the COMP pinand the OUTPUT pin connects the

by William Jett

continued on page 11

Figure 1. Conventional approaches to driving capacitive lines

LINEAR TECHNOLOGY LINEAR TECHNOLOGY LINEAR TECHNOLOGY

2 Linear Technology Magazine • June 1993

DESIGN FEATURES

has extensive experience in servo-control systems.

Hasti once designed the entirepower management circuit of a palm-top computer on a flight from SanJose to Orange County. The wholecircuit worked the first time it waspowered up on the customer’s printedcircuit board. This design was pow-ered from four AA cells and consistedof a dual output (5 volts and 3.3 volts)switching supply for the logic, a con-stant-current battery charger, a –12Vsupply for VPP flash-memory program-ming, –24 volts for LCD contrast, pluslow-battery detect and microproces-sor watchdog and reset functions.

Hasti is our first female FAE and,as might be expected, a number ofeyebrows are ever so slightly raised

when she first visits customers. Hastiviews this as a challenge to be over-come and feels that once she gainsthe customer’s confidence, any pre-conceptions disappear.

Hasti enjoys traveling and is, infact, a world traveler. She has been tomany countries in Europe, NorthAfrica, and the Middle East, and hasvisited exotic cities in Greece andTurkey. At home in sunny southernCalifornia Hasti enjoys tennis andhorseback riding, but she keeps all ofus at the factory honest by attendingfinance and business seminars in herspare time. Hasti can be reachedthrough LTC’s Southwest RegionalSales Office as listed on the back ofthis magazine.

I first came to Linear Technologyfive years ago. In those days we hadtwo applications engineers (not in-cluding Jim Williams, who playedguru to all) and about 300 people.Since my first days at Linear Technol-ogy, the emphasis has been on quality.We don’t publish circuits unless theyhave been breadboarded and exten-sively tested. The customer is ourprimary concern. Phone calls must bereturned as soon as possible. Circuitbreadboards are sometimes sent backand forth across the country severaltimes before the customer is satisfiedthat the circuit works correctly.

The word got out. Customers, com-petitors, and even the stock marketdiscovered LTC. Customers discov-ered that we mean what we say aboutcustomer support and quality; that is,we don’t take it for granted.

Now we have several more applica-tions engineers as well as many moreFAEs. They all work for you, thecustomer. If you don’t believe that,give it a try.

This issue features a bevy of newproducts from LTC. LTC’s first power-factor corrector (PFC) circuit is high-

lighted and, to make life easier for “yeolde editor,” Carl Nelson answers theage-old question, “What is Power Fac-tor, Anyway?” The LT1248 power-factor corrector can provide powerfactors of greater than 99.5%over a 10:1 load range—quite anachievement!

Next are two more switching regula-tors, which warp through the 90%efficiency barrier. The LTC1147 is an8-pin, current-mode buck DC/DCconverter with output current to 1amp and efficiencies greater than90% over two decades of output cur-rent. The LTC1174 is an easy-to-usestep-down converter, which requiresonly a few external components toconstruct a complete high-efficiencyconverter. The LTC1174 requires only130µA quiescent current with no loadand a tiny 1µA in shutdown. An inter-nal current limit is pin selectable to340mA or 600mA. Both of the newswitchers make use of Burst ModeTM

operation to maximize efficiency atlow current levels.

The high-speed area continues toblossom with the introduction of theLT1206, a 60MHz current-feedback

amplifier with a guaranteed 250mAof output current. The LT1206is optimized to drive multiplevideo cables. It features excellent dif-ferential gain and phase performancespecifications. Additionally, theLT1206 draws less than 100µA whenturned off.

This issue features a bumper cropof Design Ideas. A high-current syn-chronous switcher circuit converts 5volts to 3.3 volts at 10 amps andanother switcher converts 12–36 voltsto 5 volts at 5 amps output current.The LT1228 makes a reappearance inthis issue in an article on how tooptimize a video gain control. Thegrowing world of RF and wirelesscommunications is represented witha circuit to generate protected gatebias for GaAs power amplifiers for RFtransmissions. Noise generation andnoise generators are here with shortsections on noise generation and noisefor communications channel testing(i.e., filters). Finally, there is a circuitthat provides a constant 5 volt outputfrom 3.5 to 40 volt inputs without theuse of a transformer. The circuit hack-ing continues.

FAE Cameo: Hasti ForoutanLTC now has nineteen Field Appli-

cation Engineers (FAEs) spreadthroughout the world to assist cus-tomers in the design and selection ofcircuits available from LTC. All of ourFAEs are available by phone and, incertain situations, in person, to helpyou design your circuitry. This spacewill profile one FAE per issue.

Hasti Foroutan works out of ourSouthwest Area office. She coverspart of southern California, includ-ing Orange and San Diego Countiesand parts of Los Angeles and River-side Counties. She has BSEE andMBA degrees from the University ofCalifornia at Irvine. Hasti’s expertiseis in the areas of switching powersupply and interface design. She also

Service, Customer Satisfaction,and All That by Richard Markell

EDITOR'S PAGE

Linear Technology Magazine • June 1993 3

DESIGN FEATURES

The LT1248 Power-Factor Correctorby Carl Nelson

1248_1.eps

IEC 555-2 CLASS D LIMITS

CURRENT PULSE “IN ENVELOPE”

95% OF TIME

LINE VOLTAGE

UNCORRECTED LINE CURRENT

LINE CURRENT AFTER POWER- FACTOR CORRECTION

Figure 1. Typical waveforms for an electronic load

What is Power Factor,Anyway?

Power factor, as it relates to AC-mains usage, is basically the ratio ofRMS load power to the product of linevoltage and RMS current. The two arenot the same because the line currentor its harmonics may be phase shiftedwith respected to line voltage.

Figure 1 shows typical waveformsfor an electronic load (a TV, stereo,computer, or the like) that full-waverectifies the line voltage to feed a largefilter capacitor. Line current flowsonly at the peak of the line voltage,generating large harmonic contentand making the RMS current muchhigher than would be predicted byload power. After power-factor correc-tion, the current waveform mimicsthat seen with a resistive load.

Clever engineers will note that aswitching power supply driving a con-stant load actually looks like a negativeresistance to the mains. As the mainsvoltage decreases, current increasesto maintain constant load power. Powerfactor, however, is concerned with whathappens in one cycle of the line. Thefact that RMS currents increase withdecreasing line voltage when averagedover many cycles is a separate issue. Itis interesting to note, however, thatthis may become a thorn in the side ofpower companies as electronic loadsbecome a higher percentage of total

load. The use of rolling brownouts toreduce peak load will actually result inhigher mains current.

The LT1248 was designed to handlepower-factor correction for off-lineswitching supplies by generating a380–400V DC output using a simpleboost topology. This DC supply isthen used as a raw input to a conven-tional switching supply, which isolatesand steps down the voltage to providethe final load voltages. The chip candirectly drive an external MOSFET atpower levels up to 300W with AC inputvoltages varying over the full interna-tional range of 90 to 270VAC. Theboost topology was chosen because itis very efficient and uses a minimum ofparts, and because the inductor actsas a natural filter to isolate switchingripple from the AC line. Nevertheless,the LT1248 is versatile enough to alsoimplement inverting and flyback to-pologies with very little change in thecontrol circuitry.

One way to implement power-factorcorrection is to use constant switchon-time and an inductance lowenough to ensure discontinuous-modeoperation under all load andinput-voltage conditions. If switchon-time does not vary during onecycle of the line, the average currentdrawn from the line will vary directlywith voltage and will, therefore, look

resistive. This approach has the dis-advantage of very high peak currents,which lower efficiency, require a largerswitch, and create more EMI.A second approach forces the inductorto operate just at the border of discon-tinuous mode and uses a multiplier tomake peak current track the line-voltage waveform. This methodreduces peak currents but has theadditional disadvantage of variablefrequency.

The LT1248 uses a fixed-frequencyapproach that measures actual linecurrent and forces switch duty cycleto adjust itself dynamically to makeline current track line voltage. Thistechnique gives high power factorwith low harmonic distortion andworks independently of continuous-or discontinuous-mode operation, sothat peak currents are kept low andwide variations in load can be easilyaccommodated. A multiplier insertedbetween the error amplifier and thePWM logic has line voltage as oneinput, forcing current to track voltage.

The circuit in Figure 2 shows theLT1248 generating a power-factorcorrected 382VDC bus suitable as aninput for a standard, off-line switchingregulator. The line filter replaces thefilter normally used in switchingregulators; it can be simpler andlower in cost than the normal filterbecause of the filtering action of theboost inductor. A standard bridgeprovides full-wave rectified power tothe boost input. Note that no capacitoris used at the output of the bridge.A capacitor here would createharmonic line currents. A boost archi-tecture is created by inductor L1,diode D1, and switch Q1. The basicDC equations for a boost circuit are:

VIN

1–DC(VOUT)(IOUT)

VIN

(DC = duty cycle)

VOUT = IIN =


DESIGN FEATURES

If the loop functioned as a normalboost converter and held the outputvoltage constant with a constant load,input current would be highest at lowinput voltage and lowest at high in-put voltage. This is exactly what wedon’t want—the voltage and currentshould track. Remember, however,that power factor is concerned withwhat happens in one cycle of the line,i.e., whether current tracks voltageover a 16ms time frame. The fact thataverage current (over many cycles)does not track voltage is irrelevant.This special boost converter cannotviolate the basic equations shown onpage three for averaged conditions,but it can force current to track volt-age during one line cycle. It does thisby making the response time of theerror amplifier very slow. Over manyline cycles, the error amplifier cancorrect the switch drive to keep out-put voltage constant, but within one

cycle the multiplier forces line currentto follow line voltage.

One consequence of power-factorcorrection is that the current deliv-ered to the output capacitor isequivalent to full-wave rectified 60HzAC, and therefore the capacitor has ahealthy amount of 60Hz ripple on it.This is the direct result of power-factor correction and is independentof the topology used to achieve correc-tion. From a practical standpoint, thisis not usually a problem, but itdoes have one important implication:It is theoretically impossible to com-bine the power-factor function withthe switching-regulator function toachieve a power-factor corrected out-put that does not contain largeamounts of 60Hz ripple.

Loop DescriptionThe basic purpose of the loop is to

sense line current and force it to track

line voltage, while maintaining a con-stant DC output voltage. First I willdescribe the circuit that forces linecurrent to follow line voltage. A cur-rent proportional to line voltage isgenerated with R5. This current islightly filtered with C4 to remove anyswitching ripple and then fed into amultiplier. The output of the multi-plier (IM) is the product of this currentand a second current proportional toerror-amplifier output squared. (Thesquaring of error amplifier outputdoes not affect power-factor correc-tion per se; it is done for loop-stabilityconsiderations—more on this topiclater.) Over one line cycle, the erroramplifier output is assumed to besteady, so IM is a full-wave rectified60Hz sine wave with an amplitudedependent on error-amplifier output.This rectified sine wave current is fedthrough R2 to develop a voltage thatbecomes the reference to which line

1248_2.eps

–

+–

+LOGIC GATE

DRIVER

MULTIPLIER IM = K(IAC)(IE)2

–

+ 7.9V

–

+EA7.5V

REFERENCE

+

–

C2 100pF

R4 20k

C3 1nF

C4 0.01µF

D1 MUR850

R3 4k

BRIDGELINE FILTER

90VAC TO

270VAC

C1 180µF

L1 750µH

382VDC

R1 0.2Ω

IAC

IE

PWM COMP

IM

OSCILLATOR

R2 4k

+R5 1M

R6 5Ω

R10 20k

C5 0.1µF

R9 20k

R8 20k

R7 1M

R11 15k

C6 1nF

VCC

CSET RSET GND VA OUT

VSENSE

OVP

GATE

IACCUR AMP OUTISENSE

MULT OUT

CA

Q1

OVP COMP

C7 0.1 µF

REF

Figure 2. 25W–300W power-factor corrected supply schematic


DESIGN FEATURES

1248_3.eps

MAIN INDUCTOR

VCC

90k 1W

56µF 18V

1µF

MAIN INDUCTOR

VCC

90k 1W

1µF

56µF

CAPACITOR DRIVEN EXTRA INDUCTOR WINDING

+

+

++

1000pF


current. This was annoying becausethe resistor had to be about 40kΩ toensure start-up at low line voltage,but at high line voltage, the resistorhad about 250V across it and dissi-pated almost 2W. The LT1248 has astart-up current of approximately250µA, so much lower trickle cur-rents can be used. Supply efficiency isimproved and heat buildup is signifi-cantly reduced.

Auxiliary windings on boost-converter inductors do not generateconstant output voltages like thoseon flyback transformers. The voltageacross the inductor is equal to theinput voltage in one polarity and isequal to output voltage minus inputvoltage in the opposite polarity. Bothof these signals are unregulated andvary widely. The solution is to peakdetect both waveforms and sum them.The result is a voltage proportional tothe regulated output voltage. Figure3 shows how this is done. The 90kresistor supplies trickle start-upcurrent.

Also shown in Figure 3 is a newtechnique for IC supply voltage thateliminates the need for an extra in-ductor winding. It uses capacitorcharge transfer to generate a constantcurrent source, which feeds azener diode. Current to the zener isequal to (VOUT – VZ)(C)( ƒ ), where VZis the zener voltage and ƒ is theswitching frequency. For VOUT = 382V,VZ = 18V, C = 1000pF, and ƒ = 100kHz,zener current will be 36mA. This isenough to operate the IC, includingthe extra current drawn by gate-chargedrive to the FET.

Protection FeaturesAt Linear Technology, we pride

ourselves on making “bustproof”ICs, so we have included severalfeatures on the LT1248 to handlepotentially destructive conditions,regardless of their origins. Thereis a peak-current-limiting comparatorthat overrides the current regulatingloop if, for some reason, the loop at-tempts to increase switch currents todestructive levels. This is a backup tothe main current limiting function

current is compared. Notice that thecurrent amplifier CA has its inputsconnected to this reference voltageand to a second resistor (R3), which isconnected to the opposite side of R1.If CA operates as a true op amp, itmust somehow force the voltageacross R1 to be equal to the referencevoltage across R2. There is no voltagedrop across R3 other than amplifierbias current. C2, C3, and R4 areincluded to filter switching ripple fromCA’s output.

How does the current amplifierforce line current? Like any standardPWM loop, comparator C1 comparesa triangle-wave signal from theoscillator to the CA output togenerate a square wave whose dutycycle is proportional to CA output.This signal flows through some logicto drive a power MOSFET. Linecurrent will be a function of switchduty cycle, so the loop is closed andCA can force the voltage across R1 tofollow the voltage across R2. Thisaction must have a bandwidth manytimes greater than 60Hz to avoid dis-tortion, but much less than the typical100kHz switching frequency to rejectswitching ripple.

The second part of the loop re-quirement is to maintain a constantDC output voltage. Voltage is sensedwith the R7/R8 divider and comparedto a 7.5 volt reference by theerror amplifier. Local AC feedbackaround the error amplifier, consistingof R10 and C5, gives the amplifiervery slow response, but over severalcycles of the line it can adjustthe magnitude of the line current ref-erence IM to whatever value is neededto maintain the output voltage at

382VDC. The fact that it takes severalcycles of the line to correct for anyload or line variations is importantand necessary. If the amplifiercould respond significantly in one linecycle, it would create distortion in theIM reference current with resultingline-current distortion.

The output of the multiplier ismade proportional to the square of theerror amplifier output to minimizevariations in voltage control-loopgain. The worst case for loop stabilitytends to be at high line voltage withlight loads. The error amplifier outputis low in this condition and the squar-ing term produces lower loop gainwith improved response.

Chip Start-Upand Supply Voltage

During normal operation, powerfor the LT1248 can be drawn froman auxiliary winding on the powerinductor or transformer. Start-up,however, is a problem, because theauxiliary power is not available untilswitching starts. The solution is tomake the IC have a low supply currentuntil its input voltage reachessome threshold—say, 16V. Then thechip turns on to normal operationand draws normal operating current.A large supply bypass capacitor holdsthe supply voltage up long enough toallow power from the auxiliary wind-ing to develop. The trickle currentneeded to charge the capacitor istaken from a large resistor connectedto the rectified mains. This is oftenreferred to as “burp” starting.

In older designs, burp starting re-quired several milliamps of trickle

Figure 3. Generating the LT1248’s supply voltage


DESIGN FEATURES

4

1174_1.eps

VIN

SHUTDOWN

SWIPGM

LBIN

LBOUT

GND

LTC1174-5

3

2

7

8

1

6

+

VOUT

5

+VIN 9V

1N5818

100µH†

5V/175mA

220µF* 10V

100µF* 20V

+0.1µF

*SANYO OS-CON †COILTRONICS CTX100-4 COILTRONICS (305) 781-8900

LTC1174: A High-EfficiencyBuck ConverterIntroduction

Not long ago, linear regulators werethe choice for low-power applicationsand applications where heat dissipa-tion was not a problem. Recently,with the trend towards portability,users started to take a hard look atswitching regulators for these appli-cations. The main advantage ofswitching regulators over linear regu-lators is higher efficiency, whichtranslates to longer battery life.

The LTC1174 is a 8-pin SOIC,“user-friendly” step-down converter.(A DIP package is also available.) Onlyfour external components are neededto construct a complete, high-efficiencyconverter. With no load, it requiresonly 130µA of quiescent current;this decreases to a mere 1µA uponshutdown. The LTC1174 is protectedagainst output shorts by aninternal current limit, which is pinselectable to either 340mA or 600mA.This current limit also sets theinductor’s peak current. This allowsthe user to optimize the converter’sefficiency depending upon the outputcurrent requirement.

In dropout conditions, the internal0.9Ω (at a supply voltage of 9V) powerP-channel MOSFET switch is turnedon continuously (DC), thereby maxi-mizing the life of the battery source.

(Who says a switcher has to switch?)In addition to the features alreadymentioned, the LTC1174 boasts alow battery detector. Moreover, theLTC1174 comes in three versions: theLTC1174-5 (5V output), the LTC1174-3.3 (3.3V output), and the LTC1174(adjustable). All versions functiondown to an input voltage of 4V andwork up to an absolute maximum of13.5V.

EfficiencyFigure 1 shows a practical

LTC1174-5 circuit with a minimumof components. Efficiency curves forthis circuit at two different inputvoltages are shown in Figure 2. Notethat the efficiency is 94% at a supplyvoltage of 6V and load current of175mA. This makes the LTC1174attractive to all power sensitive appli-cations and shows clearly whyswitching regulators are gainingdominance over linear regulators inbattery-powered devices.

If higher output currents are de-sired, pin 7 (IPGM) can be connected toVIN. Under this condition, the maxi-mum load current is increased to450mA. The resulting circuit and effi-ciency curves are shown in Figures3 and 4, respectively.

4

1174_3.eps

VIN

SHUTDOWN

SWIPGM

LBIN

LBOUT

GND

LTC1174-5

3

2

7

8

1

6

+

VOUT

5

+VIN 9V

1N5818

100µH†

5V/425mA

220µF* 10V

100µF* 20V

+0.1µF

*SANYO OS-CON †COILTRONICS CTX100-4 COILTRONICS (305) 781-8900

What isBurst ModeTM Operation?

To maximize efficiency, theLTC1174 was designed to go intosleep mode once the output voltagereaches regulation. When this occurs,a majority of the internal circuitryis turned off, reducing the quiescentcurrent from 0.45mA to 130mA.Under these conditions, all theload current is supplied by theoutput capacitor. When the outputvoltage drops by an amount equiva-lent to the hysteresis of the comparator,the LTC1174 wakes up and startsswitching again until the outputreaches regulation. The process then

by San-Hwa Cheeand Randy Flatness

Figure 3. Typical application for higher outputcurrents

LOAD CURRENT (mA)

170

EFFI

CIEN

CY (%

)

80

90

95

100

10 100 500

1174_4.eps

95

85

75

VIN = 6V VIN = 9V

100µH 5V VIN CTX100-4

L = VOUT = IPGM =

COILTRONICS =

Figure 1. Typical application for low outputcurrents

LOAD CURRENT (mA)

170

EFFI

CIEN

CY (%

)

80

90

95

100

10 100 200

1174_2.eps

95

85

75

VIN = 6V VIN = 9V

L = 100µH VOUT = 5V IPGM = 0V

Figure 4. Efficiency vs. load currentFigure 2. Efficiency vs. load current

Burst ModeTM is a trademark of Linear Technology Corporation


DESIGN FEATURES

LOAD CURRENT (mA)

00

DR

OPO

UT

VOLT

AGE

(mV)

100

150

250

350

450

550

100 400

1174_6.eps

50 150 200 250 300 350

50

500

400

300

200 100µH 5V VIN CTX100-4

L = VOUT = IPGM =

COILTRONICS =

The minimum value of L can be deter-mined by the following:

4 × 10–6(VOUT + VD)2(IPEAK – IOUT)

L (Henrys) =

where IPEAK = 340mA or 600mA, de-pending on the condition of IPGM andVD = diode forward drop. As the induc-tance is increased from this minimumvalue, the ripple current decreases.This also increases the maximum loadcurrent, since the inductor’s peakcurrent is fixed.

Another consequence of constantoff-time architecture is that theswitching frequency is related to theinput voltage. For an input voltagerange of 6V to 12V, with an outputvoltage of 5V, the operating frequencyvaries from about 42kHz to 146kHz.For example, with an input voltageof 9V, the switching frequency is ap-proximately 110kHz. At this highoperating frequency, a small inductorvalue can be used.

100% Duty Cyclein Dropout Conditions

When the input voltage decreases,the switching frequency decreases.With the off-time constant, the on-time is increased to maintain the samepeak-to-peak ripple current in theinductor. Ultimately, a steady state

repeats. The inductor’s current andoutput voltage waveforms (Figure 5)reveal how the LTC1174 utilizesBurst ModeTM operation.

As the load current increases, thetime spent in sleep decreases.Eventually the LTC1174 goes intocontinuous-mode operation, nevergoing to sleep. Any increase in loadcurrent past this point will cause theoutput to drop out of regulation.

Constant Off-Time ArchitectureAnother technical point to note is

that the LTC1174 uses a constant off-time architecture. This off-time is setto 4ms when the output is in regula-tion; otherwise it is inversely propor-tional to the output voltage. Inthe extreme condition, the P-channelis turned on at a 100% duty cycle tomake the LTC1174 a low-dropoutregulator. The advantage of thisscheme is that the inductor’s ripplecurrent is predictable and well con-trolled, making the selection ofinductor much easier. The inductor’speak-to-peak ripple current is in-versely proportional to the inductance.With a 50mH inductor and a 5V out-put voltage, the ripple current is440mA. If a lower ripple current isdesired, a larger inductor can be used.

The inductance value is governedby the current requirement of the load.

condition will be reached whereKirchoff’s Voltage Law determines thedropout voltage. When this happens,the P-channel power MOSFET isturned on at DC (100% duty cycle).The dropout voltage is then governedby the load current multiplied by thetotal DC resistance of the MOSFET,the inductor, and the internal 0.1Ωcurrent-sense resistance. Figure 6shows the dropout voltage as a func-tion of output current. Note that for aload current of 400mA, the dropoutvoltage is only 0.51V. This is compa-rable to what linear regulators canoffer. Unlike linear regulators, whereground current varies with load cur-rent, the LTC1174’s ground current isa constant 450µA.

Figure 5. LTC1174 inductor current and output voltage at medium load

Figure 6. Dropout voltage vs. output current

1174_5.eps

BURSTING

SLEEPING OFF-TIME 4µs

TIME

IND

UCT

OR

CU

RR

ENT

AC C

OU

PLED

OU

TPU

T VO

LTAG

E

TIME

HYSTERESIS OF VOLTAGE

COMPARATOR


DESIGN FEATURES

+

LBIN3

4

1174_9.eps

VIN

SHUTDOWN

SW

IPGM

LBOUT

GND

LTC1174-5

7

2

8

1

6

+

VOUT

5

INPUT VOLTAGE 4V - 7.5V

1N5818

50µH†

VOUT = –5V/150mA

2 X 33µF* 16V

+0.1µF

AVX TPSD336K016 COILTRONICS CTX50-4 COILTRONICS (305) 781-8900

* †

270k

39k

4.7k

LOW BATTERY INDICATOR

2 X 33µF* 16V

X = 20µs/DIV

LOAD CURRENT 300mA/DIV

1174_8.eps

INDUCTOR CURRENT 0.5A/DIV

AC COUPLED OUTPUT VOLTAGE

50mV/DIV

LTC1174 COMING OUT OF SHORT

OUTPUT VOLTAGE

1V/DIV

GND

1174_7.eps

OUTPUT SHORTED

X = 50µs/DIV

INDUCTOR CURRENT

100mA/DIV

LOAD CURRENT (mA)

150

EFFI

CIEN

CY (%

)

100

10 100 500

1174_11.eps

VIN = 5V90

80

70

60

50µH 3.3V VIN CTX50-4

L = VOUT = IPGM =

COILTRONICS =

A 5V to 3.3V ConverterThe LTC1174-3.3 is ideal for appli-

cations that require 3.3V at less than450mA. A minimum board area,surface mount 3.3V regulator is shownin Figure 10. Figure 11 shows thatthis circuit can achieve efficiencygreater than 85% for load currentsbetween 5mA and 450mA.

Good Start-Upand Transient Behavior

The LTC1174 exhibits excellentstart-up behavior when it is initiallypowered-up or recovering from ashort-circuit. This is achieved by mak-ing the off-time inversely proportionalto the output voltage when the outputis still in the process of reachingits regulated value. When the outputis shorted to ground, the off-time isextended long enough to prevent anypossibility of a build up in theinductor’s current. When the short isremoved, the output capacitor beginsto charge and the off-time graduallydecreases. Note the absence of over-shoot when the LTC1174 comes outof a short-circuit, as shown in Figure7. When the output reaches its regu-lated value, the off-time is fixed at 4µs.The initial power-up waveform issimilar to Figure 7.

In addition, the LTC1174 hasexcellent load-transient response.When the load current drops sud- Figure 11. Efficiency vs. load current

Figure 9. Positive to –5V converter with low-battery detection Figure 10. 5V to 3.3V output application

LBOUT2

4

1174_10.eps

VIN

SHUTDOWN

SW

IPGM

LBIN

GND

LTC1174-3.3

7

3

8

1

6

+

VOUT

5

INPUT VOLTAGE 4V - 12.5V

1N5818

50µH†VOUT = 3.3V/450mA

2 X 33µF** 16V

3 X 15µF* 25V

+0.1µF

(3) AVX TPSD156K025 (2) AVX TPSD336K016 COILTRONICS CTX50-4 COILTRONICS (305) 781-8900

* **

†

Figure 7. Short-circuit and start-up response of the LTC1174 Figure 8. Load transient response

denly, the feedback loop respondsquickly by turning off the internalP-channel switch. Sudden increasesin output current will be met initiallyby the output capacitor, causing theoutput voltage to drop slightly. Asmentioned above, tight control ofinductor’s current means that out-put-voltage overshoot is virtuallyeliminated (see Figure 8).

Typical Applications

Positive-to-Negative ConverterThe LTC1174 can easily be set up

for a negative output voltage. TheLTC1174-5 is ideal for –5V outputs,as this configuration requires thefewest components. Figure 9 showsthe schematic for this applicationwith low-battery detection capability.The LED will turn on at input voltagesbelow 4.9V. The efficiency of this cir-cuit is 81% at an input voltage of 5Vand output current of 150mA.


DESIGN FEATURES

LOAD CURRENT (A)1mA

60

EFFI

CIEN

CY (%

)

80

90

100

10mA 100mA 1A

1147_3.eps

70

VIN = 5V

LTC1147-3.3

IntroductionThe LTC1147 switching-regulator

controller is the latest addition toLinear Technology’s high efficiency,step-down DC/DC converter family.This 8-pin controller uses the samecurrent-mode architecture and BurstModeΤΜ operation as the LTC1148/LTC1149 but without the synchro-nous switch. Ideal for applicationsrequiring up to 1 amp, the LTC1147shows 90% efficiencies over two de-cades of output current.

The LTC1147, like the other mem-bers of the LTC1148/LTC1149 family,automatically changes from current-mode operation at high outputcurrents to Burst ModeΤΜ operationat low output currents. The wideoperating range is illustrated by thetypical efficiency curve of Figure 1. Allmembers of the LTC1147/LTC1148/LTC1149 family are capable of opera-tion at 100% duty cycle, providingvery low dropout operation, and allhave built-in current limiting.Load and line regulation are excellentfor a wide variety of conditions,including the transition from BurstModeΤΜ operation to continuous modeoperation.

The New SO8 LTC1147Switching-Regulator Controller OffersHigh Efficiency in a Small Footprint

by Randy Flatness

1147_2.eps

KRL SP-1/2-A1-0R100 COILTRONICS CTX100-4 COILTRONICS (305) 781-8900 KRL/BANTRY (603) 668-3210

RS = L =

ITH

CT

GND

VIN

SENSE +

SENSE –

+

1000pF

+

VIN (4V-12V)

RC 1k

CC 3300pF

2

8

5

4

D1 MBRD330

CIN 15µF x 2 25V

L 100µH

1

7

SHUTDOWN6

0.1µF

3

PDRIVE

+

0V = NORMAL >1.5V = SHUTDOWN

CT 560pF

LTC1147-3.3

P-CH Si943ODY RSENSE

100mΩ

VOUT 3.3V/1A

COUT 220µF 6.3V

High Efficiency in a Small AreaThe 8-pin SOIC package and a few

external components make high effi-ciency DC-to-DC conversion feasiblein the extremely small board spaceavailable in today’s portable electron-ics. (A DIP package is also available.)An ideal application for the LTC1147is dropping 5V to 3.3V locally on a PCboard. If a linear regulator with a1 amp output current is used, thepower dissipation will exceed 1.7 Watts.This is unacceptable when there is noway to remove the heat from an en-closed space. The LTC1147 5V to 3.3Vconverter shown in Figure 2 has 85%efficiency at 1A output, with efficien-cies greater than 90% for load currentsup to 500mA. Using the LTC1147reduces the power dissipation to lessthan 500mW. The efficiency plotted asa function of output current is shownin Figure 3.Figure 1. Greater than 90% efficiency is obtained

for load currents of 20mA to 2A (VIN = 10V)

Figure 2. This LTC1147 5V to 3.3V converter achieves 92% efficiency at 300mA load current

Giving Up theSynchronous Switch

The decision whether to use a non-synchronous LTC1147 design or afully synchronous LTC1148 designrequires a careful analysis of wherelosses occur. The LTC1147 switching-regulator controller uses the same

Figure 3. The LTC1147 5V to 3.3V converterprovides better than 90% efficiency from20mA to 500mA of output current

LOAD CURRENT (A)1mA

70

EFFI

CIEN

CY (%

)

85

90

95

100

10mA 100mA

1147_1.eps

80

1A

75

VIN = 10V

LTC1147-5

VIN = 6V


DESIGN FEATURES

IOUT (A)10mA80

EFFI

CIEN

CY/L

OSS

(%)

100

3A

1147_4.eps

95

90

85

30mA 0.1A 0.3A 1A

LTC1147 I Q

GATE CHARGE I 2R

SCHOTTKY DIODE

MOSFET is on at DC or at a 100% dutycycle.

With the switch turned on at a100% duty cycle, the dropout is lim-ited by the load current multiplied bythe sum of the resistances of theMOSFET, the current shunt, and theinductor. For example, the low drop-out 5V regulator shown in Figure 6has a total resistance of less than200mΩ. This gives it a dropout voltageof 200mV at 1A output current.At input voltages below dropout theoutput voltage follows the input. Thisis the circuit whose efficiency is plot-ted in Figure 1.

Figure 4. Low current efficiency is enhancedby Burst ModeTM operation. Schottky diodeloss dominates at high output currents

INPUT VOLTAGE (V)4

60

EFFI

CIEN

CY (%

)

80

90

100

6 12 14

1147_5.eps

70

8 10

ILOAD = 100mA

ILOAD = 1A

LTC1147-5 LTC1148-5

Figure 5. At high input voltages combinedwith low output currents, the efficiency ofthe LTC1147 exceeds that of the LTC1148

1147_6.eps

KRL SL-1-C1-0R050J COILTRONICS CTX100-4 COILTRONICS (305) 781-8900 KRL/BANTRY (603) 668-3210

RS = L =

ITH

CT

GND

VIN

SENSE +

SENSE –

+

1000pF

+

VIN (5.5V-12V)

RC 1k

CC 3300pF

2

8

5

4

D1 MBRD330

CIN 15µF x 3 25V

L 62µH

1

7

SHUTDOWN6

0.1µF

3

PDRIVE

+

0V = NORMAL >1.5V = SHUTDOWN

CT 470pF

LTC1147-5

P-CH Si943ODY RSENSE

50mΩ

VOUT 5V/2A

COUT 220µF x 2 10V

Figure 6. The LTC1147 architecture provides inherent low dropout operation. This LTC1147-5circuit supports a 1A load with the input voltage only 200mV above the output.

loss-reducing techniques as the othermembers of the LTC1148/LTC1149family. The non-synchronous designsaves the N-channel MOSFET gate-drive current at the expense ofincreased loss due to the Schottkydiode.

Figure 4 shows how the losses in atypical LTC1147 application are ap-portioned. The gate-charge loss(P-channel MOSFET) is responsiblefor the majority of the efficiency lost inthe mid-current region. If BurstModeΤΜ operation was not employed,the gate-charge loss alone would causethe efficiency to drop to unacceptablelevels at low output currents. WithBurst ModeΤΜ operation, the DC sup-ply current represents the only losscomponent that increases almost lin-early as output current is reduced. Asexpected, the I2R loss and Schottky-diode loss dominate at high loadcurrents.

In addition to board space, outputcurrent and input voltage are the twoprimary variables to consider whendeciding whether to use the LTC1147.At low input-to-output voltage ratios,the top P-channel switch is on mostof the time, leaving the Schottkydiode conducting only a smallpercentage of the total period. Hence,the power lost in the Schottky diodeis small at low output currents. Thisis the ideal application for theLTC1147. As the output current in-creases, the diode loss increases. Athigh input-to-output voltage ratios,

the Schottky diode conducts most ofthe time. In this situation, any loss inthe diode will have a more significanteffect on efficiency and an LTC1148might therefore be chosen.

Figure 5 compares the efficienciesof LTC1147-5 and LTC1148-5 cir-cuits with the same inductor, timingcapacitor, and P-channel MOSFET.At low input voltages and 1A outputcurrent, the efficiency of the LTC1147differs from that of the 1148 by lessthan two percent. At lower outputcurrents and high input voltages, theLTC1147’s efficiency can actuallyexceed that of the LTC1148.

Ideal for LowDropout Applications

Because the LTC1147 is so wellsuited for low input-to-output voltageratio applications, it is an idealchoice for low dropout designs. Allmembers of the LTC1148/LTC1149family (including the LTC1147) haveoutstandingly low dropout perfor-mance. As the input voltage on theLTC1147 drops, the feedback loopextends the on-time for the P-channelswitch (off-time is constant)thereby keeping the inductor ripplecurrent constant. Eventually the on-time extends so far that the P-channel



DESIGN FEATURES

–

+ eIN

1206_4.eps

LT1206

RF

RG

75Ω

RF = RG = 560

75Ω

75Ω

75Ω

75Ω

75Ω CABLE

FREQUENCY (MHz)1

– 8

VOLT

AGE

GAI

N (d

B)

2

12

10 100

1206_2.eps

10

8

6

4

0

–2

– 4

–6

RF = 1.2k COMPENSATION

RF = 2k NO COMPENSATION

RF = 2k COMPENSATION

VS = ±15V

Figure 3. LT1206 large signal response Figure 6. Differential gain vs. supply voltage

network. This network has the effectof smoothing the output peaking sothat, with the correct feedback resis-tor, the overall response is flat. Figure2 shows the effect of the optionalcompensation network on frequencyresponse with CL = 200pF. Using a1.2kΩ feedback resistor and the com-pensation, the overall response is flatwithin 0.35dB to 30MHz.

Although the optional compensa-tion works well with capacitive loads,it simply reduces the bandwidth whenit is connected with resistive loads.For instance, with a 30Ω load, thebandwidth drops from 55MHz to35MHz when the compensation isconnected. Hence, the compensationwas made optional.

Although the small signal responsewith a capacitive load depends on thefeedback components and the optionalcompensation network, the largesignal response is limited by themaximum output current. In the

fastest configuration, the LT1206 iscapable of a slew rate of 1V/ns. Thecurrent required to slew a capacitorat this rate is 1mA per picofarad ofcapacitance, so a 10,000pF cap wouldrequire 10A. The large signal behaviorwith CL = 10,000pF is shown inFigure 3. The slew rate is about40V/µs, determined by the currentlimit of 400mA.

Buffers and Cable DriversThe combination of a 60MHz

bandwidth, 250mA output-currentcapability, and low output impedancemake the LT1206 ideal for drivingmultiple video cables. One concernwhen driving multiple transmissionlines is the effect of an unterminated(open) line on the other outputs. Sincethe unterminated line creates a re-flected wave that is incident on theoutput of the driver, a non-zero ampli-fier output impedance will result incrosstalk to the other lines. Figure 4shows the LT1206 connected as adistribution amplifier. Each line isseparately terminated to minimizethe effect of reflections. For systems

using composite video, the differentialgain and phase performance arealso important and have been consid-ered in the internal design of thedevice. The differential phase anddifferential gain performance versussupply is shown in Figures 5 and 6for 1, 3, 5, and 10 cables. Figure 7shows the output impedance versusfrequency. Note that at 5MHz theoutput impedance is only 0.6Ω.

Although the wide bandwidth andhigh output drive capabilities of theLT1206 make it a natural for videocircuits, these characteristics are alsouseful for audio applications. Figure8 shows the LT1206 combined withthe LT1115 low-noise amplifier toform a very low-noise, low-distortionaudio buffer with a gain of 10. With a32Ω load and a 5Vrms output level(780mW), the THD+noise for the cir-cuit is 0.0009% at 1kHz, rising to0.004% at 20kHz. The frequencyresponse is flat to 0.1dB from DCto 600kHz, with a –3dB bandwidthof 4MHz. The circuit is stable withcapacitive loads of 250pF or less.

Figure 2. Frequency response, AV = +2, CL =200pF

LT1206, continued from page 1

SUPPLY VOLTAGE (±V)5

0.00

DIF

FER

ENTI

AL P

HAS

E (D

EG)

0.50

15

1206_5.eps

0.40

0.30

0.20

0.10

7 9 11 13

RL = 15Ω (10 CABLES)

RF = RG = 560Ω N PACKAGE



RL = 150Ω (1 CABLE)

Figure 5. Differential phase vs. supply voltage

SUPPLY VOLTAGE (±V)5

0.00

DIF

FER

ENTI

AL G

AIN

(%)

0.10

15

1206_6.eps

0.08

0.06

0.04

0.02

7 9 11 13

RF = RG = 560Ω N PACKAGE


RL = 30Ω (5 CABLES) RL = 50Ω (3 CABLES)

RL = 150Ω (1 CABLE)

Figure 4. LT1206 distribution amplifier

AV = +2, RF = RG = 3k, VS = ±15V, CL = 0.01µF1206_3.eps


DESIGN FEATURES

Circuit DescriptionThe LT1206 uses a current-

feedback topology and LTC’s comple-mentary bipolar process to obtainexcellent speed and power gain withsimple circuitry. Since both the NPNsand PNPs are fast devices, both areused in the signal path. Figure 9shows a simplified schematic. Theinput stage consists of devices Q1–Q4. Devices Q5–Q8 form very fastcurrent mirrors. The output stageconsists of complementary emitterfollowers Q9 and Q12, followed bycomplementary Darlington followersQ10–Q11 and Q13–Q14. Diodes D1and D2 act as level shifts, providingthe proper AB bias levels for the

output devices Q11 and Q14. Theoptional compensation networkfor capacitive loads consists of CCand RC. When the COMP pin is leftopen, devices Q15 and Q16 act asa bootstrap, preventing the com-pensation network from loadingthe output of the current mirrors.When used, a bypass capacitor(0.01µF) AC shorts the COMP pinto the OUTPUT pin. A DC shortcannot be used because of thelarge currents that would flow inthe COMP pin during an outputshort circuit.

The shutdown pin provides adual function; first, it can be usedto turn off the biasing for theamplifier, reducing the quiescentcurrent to less than 100µA, andsecond, it can be used to controlthe quiescent current in normaloperation. All of the biasing for theLT1206 is derived from the collectorcurrent of Q18, which results in thesupply current being proportional tothe shutdown pin current. With theshutdown pin grounded, Q17 limitsQ18 current to about 500µA, result-ing in a supply current of approxi-mately 20mA. Supply current can bereduced by putting a resistor from theshutdown pin to ground. The voltage

across the resistoris then VS–3Vbe.The supply currentis approximately 40times the current

that flows in the shutdown pin. Forexample, to set the supply current to10mA on ±5V supplies, use a 12kΩresistor.

PerformanceTable 1 summarizes the major per-

formance specifications of the LT1206on ±5V and ±15V supplies.

ConclusionsThe LT1206 combines high output

current with wide bandwidth to forman effective solution for driving low-impedance loads. Stability problemswith capacitive loading have beensolved by a novel compensationscheme. Package options range fromthe surface mount SO8 to the 7-pinTO220.

FREQUENCY (Hz)100k

OU

TPU

T IM

PED

ANCE

(Ω)

1M 100

1206_7.eps

100.01

0.1

100

1

10

VS = ±15V RF = RG = 560Ω N PACKAGE

1206_9.eps

V –

OUTPUT

V+

Q11

Q14

Q13

Q10

50Ω

Q15

Q16

D2

Q12

Q9

D1

Q6

Q8

Q7

Q5

Q2

Q4

Q3

Q1

V–

V+

V+

V–

CC RC COMP–IN+IN

SHUTDOWN

Q18

1.25k

TO ALL CURRENT SOURCES

Q17 Parameter Conditions VS = ±5V VS = ±15VBandwidth AV = +2, RL = 50 45MHz 60MHzBandwidth AV = +2, RL = 30 40MHz 55MHzBandwidth AV = +2, RL = 10 30MHz 38MHzSlew rate AV = –1, RL = 100 500V/µs 1000V/µsSlew rate AV = –1, RL = 50 450V/µs 700V/µsSlew rate AV = +2, RL = 50 300V/µs 600V/µsMinimum output current 250mA 250mAMaximum inputoffset voltage 10mV 10mVMaximum invertinginput current 60µA 60µANominal supplycurrent 18mA 20mA

Table 1. LT1206 Performance

Figure 9. Simplified schematic

+15V

–

+

+

+

–

+

+

+

1µF

LT1115

LT1206

+15V1µF

0.01

1µF

560560

909

1µF

68pF–15V

–15V

100

INPUT

1206_8.eps

Figure 8. Low noise × 10 buffered line driver

Figure 7. Output impedance vs. frequency


DESIGN FEATURES

LTC1147-3.3 LTC1147-5 LTC1148-3.3 LTC1148-5 LTC1148 LTC1149-3.3 LTC1149-5 LTC1149Continuous input voltage ≤ 48VContinuous input voltage ≤ 13.5VLow dropout 5V

Adjustable/ multiple output5V to 3.3V

Minimum board area

High EfficiencyStep-Down Regulator Family

Table 1 shows the members of theLTC1147/LTC1148/LTC1149 familyand several of their applications. TheLTC1147 is available in both fixed3.3V and fixed 5V versions and isavailable in both 8-pin DIP and 8-pinSOIC surface mount packages. Forother output voltages, the LTC1148

adjustable is suggested. (If the syn-chronous switch is not needed in alow current application, the N-chan-nel MOSFET can be eliminated and aSchottky catch diode can be used.)

ConclusionThe LTC1147 adds even more ver-

satility to Linear Technology’s family

of high efficiency step-down regula-tor controllers. Optimized for lowcurrent applications, the LTC1147saves board space and cost over syn-chronous designs with only a smallreduction in efficiency. The high per-formance of this controller is ideal forextending the battery life of the new-est portable electronics.

LTC1147, continued from page10

provided by a current clamp on theIM output.

To guard against output overvoltage conditions, an over-voltage-comparator monitors DC output volt-age and immediately stops switchingaction if the output voltage rises morethan 10% above its intended value.Over-voltage is a real possibility withpower-factor designs because of theslow error amplifier response time.Remember that power delivered tothe output increases as the square ofinput voltage for frequencies abovethe bandpass of the control loop(20Hz). A sudden load shedding coin-cident with increased line voltage cancreate a situation where the erroramplifier cannot reduce currentquickly enough to prevent overshootat the output. The over-voltage com-parator provides an extra level ofprotection against such unforeseenconditions.

An internal gate clamp limits gatedrive if the supply voltage climbsabove 16V, ensuring reliable MOSFEToperation during extended periods ofhigh chip supply voltage.

Additional FeaturesThe enable/sync pin does double

duty as an on/off input and as asynchronizing port. The threshold forshutdown is 2.5V, and that for syncis 5.5V, so the two functions can bemade non-interactive. The LT1248 issometimes synchronized to the mainswitcher or to a system clock to pre-vent beat frequencies or to eliminatespecific time-slot noise spikes.

The SS pin acts as clamp on thereference input to the error amplifierduring startup. By connecting a ca-pacitor to SS, the output can beprogrammed to ramp up in a con-trolled fashion.

PerformanceFigure 4 shows the performance of

the LT1248 both in terms of powerfactor and harmonic distortion. Powerfactor remains above 99.5% over aten-to-one load-current range. Thisallows the supply to be used in “green”applications, where it will be used inboth full-power and low-power modes.Harmonic distortion also remains lowover a wide load range, even whenmeasured past the 20th harmonic.Agency guidelines may actually be

more concerned with harmoniccontent than power factor, so a well-designed power-factor corrector mustperform well in both areas.

The LT1248 fits in 16-pin, narrow-body, dual-in-line and surface mountpackages. It uses a high-frequency ICprocess to achieve fast switching withonly 8mA supply current. Switchingfrequency can extend up to 300kHz.A flexible topology, low external partscount, and the ability to handle bothlow- and high-power applicationsmake this new IC attractive for a widevariety of applications. An 8-pin ver-sion with frequency set internally at100kHz (LT1249) will also be available.

Power Factor, continued from page 5

Table 1. LTC1147/LTC1148/LTC1149 family applications

1248_4.eps

LOAD CURRENT (%)0

95%

96%

97%

98%

99%

100%

20 40 80 10060

TRANSITION BOUNDARY

SET BY INDUCTOR

VALUE

REGION OF CONTINUOUS OPERATION

Figure 4. Power factor vs. load current


DESIGN FEATURES

The LTC1337 is an ultra-low-powerRS232 transceiver with supply cur-rent up to 20 times lower than otherCMOS devices. Using proventechnology borrowed from LTC’s low-power RS485 products, the LTC1337achieves ultra-low power consump-tion without sacrificing ESDprotection and latch-up immunity.

The single 5 volt, three-driver/five-receiver RS232 transceiver draws only300µA of supply current in the no-load condition and 1µA in theshutdown mode. The charge pumprequires only four 0.1µF capacitorsand can supply up to 12mA of extracurrent to power external circuitry.The transceiver can operate up to120k baud with a 1000pF cap inparallel with a 3kΩ load, and bothdriver outputs and receiver inputscan be forced to ±25V and withstandmultiple 10kV ESD strikes.

Latch-Up EliminatedThe main sources of latch-up in

CMOS RS232 transceivers have beeneliminated by incorporating circuittechniques borrowed from the LTC485family. Latch-up can occur in conven-tional CMOS output stages whenthe output of the driver is forcedbeyond the supply rails. If the outputof the traditional CMOS output stageof Figure 1 is driven above V+ or

below V–, the parasitic P+/N-well/P-substrate PNP or the N+/P-substrate/N-well NPN (Q1, or Q2,respectively) will turn on.

The collector current of Q1 or Q2will then turn on the SCR structureformed by R1, R2, Q3, and Q4. TheSCR structure forms a short-circuitpath between V+ and V–, and remainson even though the output isbrought back within the supply rails(i.e., Q1 or Q2 turns off). This willcause the charge pump to collapseand excessive current to flow. Thelatch-up state can only be stopped byturning off the power.

By incorporating Schottky diodesinto the output stage of the LTC1337

Ultra-Low-Power CMOS RS232Transceiver Achieves 10kV ESDProtection, Eliminates Latch-Up

1337_1.eps

BIAS

BIAS

INPUT

P2

N2

N3

C1

N1

P1

V+

V–

OUTPUT

P1

N1

V+

V–

Q1

Q2

Q3

Q4

OUTPUT

R2 P-SUBSTRATE

N-CHANNEL SOURCES

R1 N-WELL

P-CHANNEL SOURCES

driver (simplified in Figure 2), thetriggering source of the latch-up iseliminated. When the output is forcedabove V+ or below V–, Schottky diodeD1 or D2 will reverse bias and preventcurrent flow. By preventing Q1 or Q2from turning on, the trigger source oflatch-up is eliminated.

ESD ProtectionESD protection is provided by a

specialized ESD protection circuit con-nected to the receiver inputs and driveroutputs (Figure 3). The protectioncell is designed to turn on andabsorb the ESD pulse energy, but toremain off during normal operation.Figure 4 shows the I-V curve of theESD protection cell.

When the voltage reaches about23V with respect to ground, the ESDcell starts to turn on; when it reachesabout 28V the cell snaps back andconducts a large amount of currentwhile limiting the voltage to 5V. Dur-ing an ESD pulse, the ESD cell turnson and clamps the pin at 5V until allof the energy has been dissipated,then returns to its off state. With thevoltage clamped at 5V, the powerdissipation is kept to a minimum and

Figure 1. Conventional CMOS output stage and parasitic devices

1337_2.eps

BIAS

BIAS

INPUT

P2

N2

N3

C1

N1

P1

V+

V–

OUTPUT

P1

N1

V+

V–

Q1

Q2

Q3

Q4

OUTPUT

R2 P-SUBSTRATE

N-CHANNEL SOURCES

R1 N-WELL

P-CHANNEL SOURCES

D1D1

D2

D2


DESIGN INFORMATION

Figure 2. Simplified LTC1337 driver output stage and parasitic devices

by Ricky Chowand

Robert Reay


DESIGN FEATURESDESIGN IDEAS

DESIGN IDEAS

Switching Regulator Pro-vides Constant 5V Outputfrom 3.5–40V Input, withouta Transformer ................ 15Brian Huffman

Optimizing A Video-Gain-Control StageUsing the LT1228 ........... 17Frank Cox

Noise Generatorsfor Multiple Uses ............ 20Jim Williams, Richard Markell,and Bent Hessen-Schmidt

High-Current SynchronousSwitcher Converts 5V to3.3V at 90% Efficiency ... 23Milt Wilcox

Protected Bias forGaAs Power Amplifiers ... 24Mitchell Lee

High-Current,Synchronous, Step-DownSwitching Regulator ...... 25Brian Huffman

A common switching-regulator re-quirement is to produce a constantoutput voltage from an input voltagethat varies above or below the outputvoltage. This is particularly importantfor extending battery life inbattery-powered applications. Figure1 shows how an LT1171 switchingregulator IC, two inductors, and a“flying” capacitor can generate aconstant output voltage that is inde-pendent of input voltage variations.This is accomplished without the useof a transformer. Inductors are pre-ferred over transformers because theyare readily available and moreeconomical.

The circuit in Figure 1 uses theLT1171 to control the output voltage.A fully self-contained switching

regulator IC, the LT1171 contains apower switch as well as the controlcircuitry (pulse-width modulator,oscillator, reference voltage, erroramplifier, and protection circuitry).The power switch is an NPN transistorin a common-emitter configuration;consequently, when the switch turnson, the LT1171’s VSW pin is connected toground. This power switch can handlepeak switch currents of up to 2.5A.

Figure 2 shows the operating wave-forms for the circuit. In thisarchitecture, the capacitor C2 servesas the single energy transfer devicebetween the input voltage and outputvoltage of the circuit. While the LT1171power switch is off, diode D1 is forwardbiased, providing a path for the cur-rents from inductors L1 and L2. TraceA shows inductor L1’s current wave-form and trace B is L2’s current wave-form. Observe that the inductor currentwaveforms occur on top of a DC level.

The waveforms are virtually identicalbecause the inductors have identicalinductance values and the same volt-ages are applied across them. Thecurrent flowing through inductor L1 isnot only delivered to the load, but isalso used to charge C2. C2 is chargedto a potential equal to the input volt-age.

When the LT1171 power switchturns on, the VSW pin is pulled toground and the input voltage isapplied across the inductor L1. At thesame time, capacitor C2 is connectedacross inductor L2. Current flowsfrom the input-voltage source throughinductor L1 and into the LT1171.Trace C shows the voltage at the VSWpin, and Trace D is the current flowingthrough the power switch. Thecatch diode (D1) is reversed biased,and capacitor C2’s current also flowsthrough the switch, through ground,and into inductor L2. During this

Switching Regulator ProvidesConstant 5V Output from 3.5 – 40V Input,without a Transformer by Brian Huffman

Figure 1. LT1171 provides constant +5V output from 3.5V to 40V input. No transformer is required

1171_1.eps

= 1.25V (1 + R2/R3) = NICHICON (AL) UPL1H560MEH, ESR = 0.250Ω, IRMS = 360mA = NICHICON (AL) UPL1H151MPH, ESR = 0.100Ω, IRMS = 820mA = NICHICON (AL) UPL1C471MPH, ESR = 0.090Ω, IRMS = 770mA = COILTRONICS CTX50-4, DCR = 0.090Ω, COILTRONICS (305) 781-8900

EQ. 1: VOUT C1 C2 C3

L1, L2

VIN

FB

VIN (3.5V-40V)

R1 1k

4

2

D1 MBR350

C2 150µF

50V

5

VOUT +5V 0.5A

VSW

VCGND

3

L1 50µH

L2 50µH

LT1171

1

C4 1µF

+ C1 56µF 50V

+ C3 470µF 16V

R2 3.01k 1%

R3 1.00k 1%

+


DESIGN FEATURES

interval, C2 transfers its stored energyinto inductor L2. After the switchturns off, the cycle is repeated.

Another advantage of this circuit isthat it draws its input current in atriangular waveshape (see Trace A inFigure 2). The current waveshape ofthe input capacitor is identical to thecurrent waveshape of inductor L1,except that the capacitor’s currenthas no DC component. This type ofripple injects only a modest amountof noise into the input lines becausethe ripple does not contain any sharpedges.

Figure 3 shows the efficiency of thiscircuit for a 0.5A load and maximumoutput current for various inputvoltages. The two main loss elements

are the output diode (D1) and theLT1171 power switch. A Schottkydiode is chosen for its low forwardvoltage drop; it introduces a 10%loss, which is relatively constant withinput-voltage variations. At low inputvoltages the efficiency drops becausethe LT1171 power switch’s saturationvoltage becomes a higher percentageof the available input supply.

This circuit can deliver an outputcurrent of 0.5A at a 3.5V inputvoltage. This rises to 1A as inputvoltage is increased. Above 20V,higher output currents can beachieved by increasing the values ofinductors L1 and L2. Larger induc-tances store more energy, providingadditional current to the load. If 0.5A

of output current is insufficient, use ahigher current part, such as theLT1170.

The output voltage is controlled bythe LT1171 internal error amplifier.This error amplifier compares a frac-tion of the output voltage, via theR1–R2 divider network shown in Fig-ure 1, with an internal 1.25V referencevoltage, and varies the duty cycleuntil the two values are equal. (Theduty cycle is determined by multiply-ing the switch-on time by theswitching frequency.) The RC network(R1 and C4 in Figure 1) connected tothe VC pin provides sufficient compen-sation to stabilize this control loop.Equation 1 (on the schematic) can beused to determine the output voltage.

the CMOS devices are kept below theoxide-rupture voltage. To providefurther protection, the receiver inputgoes through a 9-to-1 resistive

divider before going to any CMOSdevices. The ESD cell providesprotection against over 10kV ofhuman-body-model ESD strikes.

1337_3.eps

BIAS

BIAS

INPUT

P2

N2

N3

C1

N1

P1

V+

V–

OUTPUT

ESD CELL

R1 150Ω

ESD CELL

R2 10kΩ

R2 1kΩ

R1 8kΩ

BIAS

INPUTV–

Figure 3. LTC1337 ESD protection

LTC1337, continued from page 14

Figure 3. Efficiency and load characteristics forvarious input voltages

DESIGN IDEAS

Figure 2. LT1171 switching waveforms

VOLTAGE (V)–30

CUR

REN

T (m

A)

30

1337_4.EPS

200–10–20

0

10

Figure 4. ESD protection cell IV curve

INPUT VOLTAGE (V)0

0.0

I OU

T (M

AX) (

A)

0.6

1.2

20 40

1171_3.eps

0.2

0.4

0.8

1.0

5 10 15 25 30 35

IOUT (MAX) EFFICIENCY

50

65

80

55

60

70

75

EFFICIENCY (%

)

5µs/DIV

D = 1A/DIV ISW

1171_2.eps

A = 1A/DIV IL1, IC1

B = 1A/DIV IL2

C = 10V/DIV VSW


DESIGN FEATURES

Video automatic-gain-control (AGC)systems require a voltage- orcurrent-controlled gain element. Theperformance of this gain-control ele-ment is often a limiting factor in theoverall performance of the AGC loop.The gain element is subject to several,often conflicting restraints. This isespecially true of AGC for compositecolor video systems, such as NTSC,which have exacting phase- and gain-distortion requirements. To preservethe best possible signal-to-noise ratio(S/N),1 it is desirable for the inputsignal level to be as large as practical.Obviously, the larger the input signal,the less the S/N will be degradedby the noise contribution of the gain-control stage. On the other hand, thegain-control element is subject todynamic-range constraints, and ex-ceeding these will result in risinglevels of distortion.

Linear Technology makes ahigh-speed transconductance (gm) am-plifier, the LT1228, which can be usedas a quality, inexpensive gain-control element in color video andsome lower-frequency RF applica-tions. Extracting the optimum

performance from video AGC systemstakes careful attention to circuitdetails.

As an example of this optimization,consider the typical gain-controlcircuit using the LT1228 shown inFigure 1. The input is NTSC compositevideo, which can cover a 10dB range,

Optimizing A Video-Gain-ControlStage Using the LT1228

to limit distortion in the transconduc-tance stage. The gain of this circuit iscontrolled by the current into the ISETterminal, pin 5 of the IC. In a closed-loop AGC system the loop-control cir-cuitry generates this current bycomparing the output of a detector2 toa reference voltage, integrating thedifference and then converting to asuitable current. The measured per-formance for this circuit is presentedin Table 1.

All video measurements were takenwith a Tektronix 1780R video-measurement set, using test signalsgenerated by a Tektronix TSG 120.The standard criteria for characteriz-ing NTSC video color distortion arethe differential gain and the differen-tial phase. For a brief explanation ofthese tests see the sidebar “Differen-tial Gain and Phase.” For this designexercise the distortion limits were setat a somewhat arbitrary 3% for differ-ential gain and 3° for differential phase.Depending on conditions, this shouldbe barely visible on a video monitor.

Figures 2 and 3 plot the measureddifferential gain and phase, respec-tively, against the input signal level

Extracting the optimumperformance from video

AGC systems takes carefulattention to circuit details

from 0.56 volt to 1.8 volt. The out-putis to be 1 volt peak-to-peakinto 75 ohms. Amplitudes were mea-sured from peak negative chroma topeak positive chroma on an NTSCmodulated ramp test signal (seesidebar).

Notice that the signal is attenu-ated 20:1 by the 75 ohm attenuatorat the input of the LT1228, so thevoltage on the input (pin 3) rangesfrom 0.028 to 0.090 volt. This is done

by Frank Cox

DESIGN IDEAS

Figure 1. Schematic diagram

1228_1.eps

–

+ 75

–

+

TEKTRONIX 1780R VIDEO

MEASUREMENT SET

75

75082.5

8

365 RSET

2k

gm

75

3

2

TEKTRONIX TSG 120

75

VARIABLE ATTENUATOR

75Ω ATTENUATOR

37.5

5

1

6

VARIABLE ISET GENERATOR

20:1

3750

BIAS GENERATOR

LT1228


DESIGN FEATURES

(the curves labeled “A” show theuncorrected data from Table 1). Theplots show that increasing the inputsignal level beyond 0.06 volt results ina rapid increase in the gain distor-tion, but comparatively little changein the phase distortion. Further at-tenuating the input signal (andconsequently increasing the set cur-rent) would improve the differentialgain performance but degrade theS/N. What this circuit needs is agood tweak!

Optimizing forDifferential Gain

Referring to the small signaltransconductance versus DC inputvoltage graph (Figure 4), observe thatthe transconductance of the amplifieris linear over a region centeredaround zero volts.3 The 25°C gm curvestarts to become quite nonlinear above0.050 volt. This explains why the dif-ferential gain (see Figure 2, curve A)degrades so quickly with signals above

this level. Most RF signals do not haveDC bias levels, but the composite videosignal is mostly unipolar.

Video is usually clamped at someDC level to allow easy processing ofsync information. The sync tip, thechroma reference burst, and somechroma-signal information swingnegative, but 80% of the signal thatcarries the critical color information(chroma) swings positive. Efficient useof the dynamic range of the LT1228requires that the input signal havelittle or no offset. Offsetting the videosignal so that the critical part of thechroma waveform is centered in thelinear region of the transconductanceamplifier allows a larger signal to beinput before the onset of severedistortion. A simple way to do this isto bias the unused input (in thiscircuit the inverting input, pin 2) witha DC level.

In a video system it might be con-venient to clamp the sync tip at amore negative voltage than usual.

Clamping the signal prior to the gain-control stage is good practice becausea stable DC reference level must bemaintained.

The optimum value of the bias levelon pin 2 used for this evaluation wasdetermined experimentally to beabout 0.03 volt. The distortion testswere repeated with this bias voltageadded. The results are reported inTable 2 and Figures 2 and 3(curves B). The improvement to thedifferential phase is inconclusive,but the improvement in the differen-tial gain is substantial.

DESIGN IDEAS

Input Differential Differential(volts) ISET(ma) Gain Phase S/N0.03 1.93 0.5% 2.7° 55dB0.06 0.90 1.2% 1.2° 56dB0.09 0.584 10.8% 3.0° 57dB

VIDEO INPUT LEVEL (V)0.020

DIF

FER

ENTI

AL G

AIN

(%)

11

0.1

1228_2.eps

10

9

8

7

6

5

4

3

2

1

0.03 0.04 0.05 0.06 0.07 0.08 0.09

A, UNCORRECTED

B, CORRECTED

VIDEO INPUT LEVEL (V)0.02

1.0

DIF

FER

ENTI

AL P

HAS

E (D

EG)

3.5

0.1

1228_3.eps

3.0

2.5

2.0

1.5

0.03 0.04 0.05 0.06 0.07 0.08 0.09

A, UNCORRECTED

B, CORRECTED

INPUT VOLTAGE (mVDC)

–2000

TRAN

SCO

ND

UCT

ANCE

(µA/

mV)

0.2

0.4

1.4

2.0

–150 –100 –50 200

1228_4.eps

0 100 150

1.8

1.6

1.2

0.6

0.8

–55°C

VS = ±2V TO ±15V ISET = 100µA

50

1.025°C

125°C

Figure 4. Small-signal transconductancevs. DC input voltage

Input Bias Differential Differential(volts) Voltage ISET(ma) Gain Phase S/N0.03 0.03 1.935 0.9% 1.45° 55dB0.06 0.03 0.889 1.0% 2.25° 56dB0.09 0.03 0.584 1.4% 2.85° 57dB

References

1. Signal to noise ratio, S/N = 20 x log(RMS signal/RMS noise).

2. One way to do this is to sample the colorburstamplitude (the nominal peak-to-peak amplitudeof the colorburst for NTSC is 40% of thepeak luminance) with a sample-and-hold andpeak detector.

3. Notice also that the linear region expandswith higher temperature. Heating the chip hasbeen suggested.

Table 1. Measured performance data (uncorrected) Table 2. Measured performance data (corrected)

Figure 3. Differential phase vs. input levelFigure 2. Differential gain vs. input level


DESIGN IDEAS

superimposed on a linear ramp (orsometimes on a stair step). The ramphas the duration of the active portionof a horizontal line of video. Theamplitude of the ramp varies fromzero to the maximum level of theluminance, which, in this case, is0.714 volt. The gain error correspondsto compression or expansion by theamplifier (sometimes called “incre-mental gain”) and is expressed as apercentage of the full amplitude range.An appreciable amount of differen-tial gain will cause the luminance tomodulate the chroma, causing vi-sual chroma distortion. The effect ofdifferential gain errors is to changethe saturation of the color being dis-played. Saturation is the relative de-gree of dilution of a pure color withwhite. A 100% saturated color has0% white, a 75% saturated color has25% white, and so on. Pure red is100% saturated, whereas pink is redwith some percentage of white and istherefore less than 100% saturated.

Differential phase is a measure ofthe phase shift in a linear amplifier atthe color subcarrier frequency when

Differential Gain and PhaseDifferential gain and phase are

sensitive indications of chroma-signal distortion. The NTSC sys-tem encodes color informationon a separate subcarrier at3.579545MHz. The color subcarrieris directly summed to the black andwhite video signal. (The black andwhite information is a voltage pro-portional to image intensity and iscalled luminance or luma.) Eachline of video has a burst of 9 to 11cycles of the subcarrier (so timed thatit is not visible) that is used asa phase reference for demodulationof the color information of that line.The color signal is relatively immuneto distortions, except for thosethat cause a phase shift or anamplitude error to the subcarrierduring the period of the video line.

Differential gain is a measure ofthe gain error of a linear amplifierat the frequency of the colorsubcarrier. This distortion is mea-sured with a test signal called amodulated ramp (shown in Figure5). The modulated ramp consists ofthe color subcarrier frequency

Figure 5. NTSC test signal

1228_br.eps

+0.1429V

–0.1429V

0V

0µs 7µs 10µs 11.5µs

–0.286V

0V BLANKING

+0.714V 100% WHITE

3.58 MHz COLOR SUBCARRIER SUMMED TO LINEAR RAMP

References

4. From the preceding discussion, the limits onvisibility are about 3° differential phase, 3%differential gain. Please note that these are nothard and fast limits. Tests of perception can bevery subjective.

the modulated ramp signal is used asan input.

The phase shift is measured rela-tive to the colorburst on the testwaveform and is expressed in de-grees. The visual effect of the distortionis a change in hue. Hue is the thatquality of perception which differen-tiates the frequency of the color, redfrom green, yellow-green from yel-low, and so forth.

Three degrees of differential phaseis about the lower limit that canunambiguously be detected by ob-servers. This level of differential phaseis just detectable on a video monitoras a shift in hue, mostly in the yellow-green region. Saturation errors aresomewhat harder to see at theselevels of distortion—3% of differen-tial gain is very difficult to detect on amonitor. The test is performed byswitching between a referencesignal, SMPTE (Society of MotionPicture and Television Engineers)75% color bars, and a distortedversion of the same signal, withmatched signal levels. An observer isthen asked to note any difference.

In professional video systems(studios, for instance) cascades ofprocessing and gain blocks can reachhundreds of units. In order to main-tain a quality video signal, thedistortion contribution of eachprocessing block must be a smallfraction of the total allowed distor-tion budget4 (the errors arecumulative). For this reason, high-quality video amplifiers will havedistortion specifications as low as afew thousandths of a degree fordifferential phase and a few thou-sandths of a percent for differentialgain.


DESIGN IDEAS

Noise Generators for Multiple UsesA BroadbandRandom Noise Generator

by Jim WilliamsFilter, audio, and RF-communica-

tions testing often require a randomnoise source. Figure 1’s circuit pro-vides an RMS-amplitude regulatednoise source with selectable band-width. RMS output is 300 millivolts,with a 1kHz to 5MHz bandwidth,selectable in decade ranges.

Noise source D1 is AC coupledto A2, which provides a broadbandgain of 100. A2’s output feeds a gain-control stage via a simple, selectablelowpass filter. The filter’s output isapplied to A3, an LT1228 operationaltransconductance amplifier. A1’soutput feeds LT1228 A4, a current-

A Diode Noise Generatorfor “Eye Diagram” Testing

by Richard MarkellThe circuit that Jim Williams de-

scribes evolved from my desire tobuild a circuit for testing communi-cations channels by means of “eyediagrams.” (See Linear Technology,Volume I, Number 2 for a shortexplanation of the eye diagram.) Iwanted to replace my pseudo-ran-dom code generator circuit, whichused a PROM, with a more “analog”design—one that more people couldbuild without specialized compo-nents. What evolved was a noisesource sampled by a very fast com-parator (see Figure 5). The compara-tor outputs a random pattern of 1’sand 0’s.

The noise diode (an NC201) is fil-tered and amplified by the LT1190high-speed operational amplifier (U1).The output feeds the LT1116 (U2), a12ns, single-supply, ground-sensingcomparator. The 2kΩ pot at the in-verting input of the LT1116 sets thethreshold to the comparator so that aquasi-equal number of 1’s and 0’s areoutput. U3 latches the output fromU2 so that the output from the com-parator remains latched throughoutone clock period. The two-level out-put is taken from U3’s Q0 output.

The additional circuitry shown inthe schematic diagram allows thecircuit to output four-level data forPAM (pulse amplitude modulation)testing. The random data from thetwo-level output is input to a shiftregister, which is reset on every fourthclock pulse. The output from the shiftregister is weighted by the three 5kΩresistors and summed into theLT1220 operational amplifier fromwhich the output is taken. The filternetwork between the 74HC74 outputand the 74HC4094 strobe input isnecessary to ensure that the outputdata is correct.

feedback amplifier. A4’s output, whichis also the circuit’s output, is sampledby the A5-based gain-control con-figuration. This closes a gain controlloop to A3. A3’s ISET current controlsgain, allowing overall output levelcontrol.

Figure 2 plots noise at a 1MHzbandpass, whereas Figure 3 showsRMS noise versus frequency in thesame bandpass. Figure 4 plots similarinformation at full bandwidth(5MHz). RMS output is essentiallyflat to 1.5MHz, with about ±2dBcontrol to 5MHz before sagging badly.

+ –

D1 NC201

16k1µF

+15V

1k

1k

10Ω

–

+

1k

1.6k

0.1(1kHz)

0.01(10kHz)

0.001(100kHz)

100pF(1MHz)

NC (5MHz)

3k

910Ω

0.1

+

–

15V

–15V

A5 LT1006

–

+

15V

–15V510Ω

10Ω

0.5µF

22µF

– +

22µF

10k

1M

LT1004 1.2V

4.7k–15V

1N4148 THERMALLY

COUPLED

10k

1µF NON POLAR

NC 201= NOISE COM=

NOISE COM CORP. (201) 261-8797

A4 LT1228

CFA

A2 LT1226

–

+A3

LT1228 gm

Noise_1.eps

Figure 1. Broadband random noise generator schematic


DESIGN IDEAS

Symmetrical WhiteGaussian Noise

by Bent Hessen-Schmidt,NOISE COM, INC.

White noise provides instanta-neous coverage of all frequencieswithin a band of interest witha very flat output spectrum. Thismakes it useful both as a broad-band stimulus and as a power-level reference.

Symmetrical white Gaussiannoise is naturally generated in re-sistors. The noise in resistors isdue to vibrations of the conduct-ing electrons and holes, as de-scribed by Johnson and Nyquist.1,2

The distribution of the noise volt-age is symmetrically Gaussian,and the average noise voltage is:

Vn = 2 √kT∫ R(f) p(f) df (1)

where:k = 1.38E–23 J/K

(Boltzmann’s constant)T = temperature of the

resistor in Kelvinf = frequency in Hzh = 6.62E–34 Js (Planck’s

constant)R(f) = resistance in ohms as a

function of frequency

kT[exp(hf/kT) –1]hf

p(f) = (2)

p(f) is close to unity for frequenciesbelow 40GHz when T is equal to290°K. The resistance is often as-sumed to be independent of fre-quency, and ∫df is equal to thenoise bandwidth (B). The availablenoise power is obtained when theload is a conjugate match to theresistor, and it is:

Vn2

4R= kTBN = (3)

where the “4” results from thefact that only half of the noisevoltage and hence only 1/4 of thenoise power is delivered to amatched load.

Figure 2. Noise amplitude at 1MHz bandpass

Figure 3. RMS noise vs. frequency at 1MHz bandpass

AMPL

ITU

DE

VAR

IAN

CE (d

B) 0

–3

–6

–9

–12

6

3

Noise_4.epsFREQUENCY (MHz)

0 1 2 3 4 5 6 7 8 9 10

–15

–18

–21

9

Figure 4. RMS noise vs. frequency at 5MHZ bandpass

Noise_2.eps10.0µs/DIV

NO

ISE

AMPL

ITU

DE

500m

V/D

IV

Noise_3.eps

AMPL

ITU

DE

VAR

IAN

CE (d

B)

FREQUENCY (MHz)0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0

0

–3

–6

–9

9

6

3

–12

–15

–18

12


DESIGN IDEAS

non-reflecting load exceeds the noisepower available from a load held atthe reference temperature of 290°K(16.8°C or 62.3°F).

The importance of a high ENRbecomes obvious when the noise isamplified, because the noise contri-butions of the amplifier may bedisregarded when the ENR is 17dBlarger than the noise figure of theamplifier (the difference in total noisepower is then less than 0.1dB). TheENR can easily be converted to noisespectral density in dBm/Hz orµV/√Hz by use of the white noiseconversion formulas in Table 1.

Equation 3 shows that the avail-able noise power is proportional tothe temperature of the resistor; thusit is often called thermal noise power.Equation 3 also shows that whitenoise power is proportional to thebandwidth.

An important source of sym-metrical white Gaussian noise is thenoise diode. A good noise diode gen-erates a high level of symmetricalwhite Gaussian noise. The level isoften specified in terms of excessnoise ratio (ENR).

ENR (in dB) = 10Log (Te –290)290

(4)

Te is the physical temperature thata load (with the same impedance asthe noise diode) must be at to gener-ate the same amount of noise.

The ENR expresses how manytimes the effective noise powerdelivered to a non-emitting,

Figure 5. Pseudo-random code generator schematic diagram

Table 1. Useful white noise conversion

dBm = dBm/Hz + 10log(BW)dBm = 20log(Vn) – 10log(R) + 30dBdBm = 20log(Vn) + 13dB for R = 50ohmsdBm/Hz= 20log(µVn√Hz) - 10log(R) – 90dBdBm/Hz= –174dBm/Hz + ENR for ENR > 17dB

15 OE

1 STR

11

LE

10k

1µF

5V

100k

1M

1k

1k

10pF

–

+U2

LT1116

5V

–5V

100pF

NC 201= NOISE COM=

NOISE COM DIODE (201) 261-8797

–5V

NC201 NOISE DIODE

3

2

7

6

4–

+U1

LT1190

100k

100k

+10µF

5V

–5V

3

2

8

67

+1µF TANT.

2k–5V5V

4

5

20

U3 74HC373

VCC3

DO

2

Q0

10

LE

1

0E

2 LEVEL OUTPUT

CLOCK5 6

U5 74HC04

3

1CLK

4

2

1D

5

1Q

6

1Q

15V

1SD1RD

11

2CLK

10

12

2D

9

2Q

8

2Q

135V

1k

74HC74 VCC = PIN 14 GND = PIN 7

2 D

3 CP

5V

5k x 3

3-STATE OUTPUTS

8-BIT STORAGE REGISTER

8-STAGE SHIFT REGISTER

QS210

QS19

QP0 QP1 QP2 QP3 QP4 QP5 QP6 QP7

4 5 6 7 14 13 12 11

+

–LT1220

8V

2

3

8

– 8V

4

1.7k

7

4 LEVEL OUTPUT

Noise_5.eps

74HC4094 VCC = PIN 16 GND = PIN 8

2SD2RD

When amplifying noise it is im-portant to remember that thenoise voltage has a Gaussian distri-bution. The peak voltages of noise aretherefore much larger than the aver-age or RMS voltage. The ratio of peakvoltage to RMS voltage is called crestfactor, and a good crest factor forGaussian noise is between 5:1 and10:1 (14 to 20dB). An amplifier’s 1dBgain-compression point should there-fore be typically 20dB larger than thedesired average noise-output powerto avoid clipping of the noise.

For more information about noisediodes, please contact NOISE COM,INC. at (201) 261-8797.

References

1. Johnson, J.B. “Thermal Agitation of Electricityin Conductors,” Physical Review, July 1928,pp. 97-109.

2. Nyquist, H. “Thermal Agitation of ElectricCharge in Conductors,” Physical Review,July 1928, pp. 110–113.


DESIGN IDEAS

High-Current Synchronous SwitcherConverts 5V to 3.3V at 90% Efficiency

by Milt Wilcox

The next generation of micropro-cessors used in desktop computersand workstations will consume pro-digious amounts of current at 3.3V.Often a high-current, offline 5V supplyis already available. The problemthen becomes how to generate 3.3Vfrom the available supply without theneed for costly and space-consumingheat sinks. For example, dropping 5Vto 3.3V in a linear regulator wouldconsume a minimum of 17W.

The circuit in Figure 1 efficientlyconverts 5V to 3.3V at output currentsup to 15A and features soft-start andshort-circuit protection. The efficiencyat 10A output is 91% (see Figure 2),meaning that only 7.25A is drawnfrom the 5V input. Because the 3.3W-conversion power loss is fairly evenlydivided among RS, L1, and the fourMOSFETs, heat sinking is not requiredfor operation at 10A. At 15A output,total losses are still only 7W, requiringminimal heat sinking.

At start-up, the circuit in Figure 1uses the charge pump built into theLT1158 synchronous, N-channelMOSFET driver to pump the gates ofthe top MOSFET switch above VIN.Start-up current is controlled by thesame LT1158 protection loop thatprovides short-circuit protection. TheDC level of the CMOS 555 triangle-wave oscillator relative to the LT1158input threshold sets the PWM dutycycle during normal operation. TheLT1431 contains the reference anderror amplifier that controls the DClevel of the triangle wave via the CMOS555 supply pin.

When the output is shorted, theLT1158 current-sense comparatoroverrides the LT1431 output to createa current-mode protection loop.Current limit is approximately100mV/RS, or approximately 20A forthe Figure 1 circuit. During a short,the duty cycle drops to a very low

value, increasing the dissipation ineach bottom MOSFET to approxi-mately 3W. Therefore, 20°C/Wheat sinking should be provided forthe bottom MOSFETs if continuousshort-circuited operation is required.

Figure 2. Efficiency for 5V to 3.3V high-currentsynchronous switcher

SEN+

LT1158

V+

BST

T DR

T FB

SRC

B DR

B FB

SEN–

V+

BIAS

IN

FAULT

+10µF 0.01µF

0.01µF

CMOS 555

4

8

2

61 7

3.3k

24k

500pF

0.01µF

16k

1000pF

V+ COLLREF

GND-F

GND-S

4.99k+ COUT

220µF 10V OS-CON

RS 5mΩ

LT1431

3.3V/15A OUT

0.1µF 500k

BAT85

– +

L1 20µH

+ CIN (4) 150µF/16V OS-CON

(2) IRLZ44

5VIN

1.62k

1158_1.eps

COILTRONICS CTX 02-12061 COILTRONICS (305) 781-8900 DALE WSC-2-4 (SURFACE MOUNT) LVR-3 (THROUGH HOLE)

L =

RS =

3

f = 50kHz

(2) IRLZ44

Figure 1. 5V to 3.3V/15A high-efficiency switching regulator

IOUT (A)0

80

EFFI

CIEN

CY (%

)

90

96

100

4 16 20

1158_2.eps

84

8 12

98

94

92

88

86

82

2 6 10 14 18


DESIGN IDEAS

Portable communications devicessuch as cellular telephones and an-swer-back pagers rely on small,GaAsFET-based 0.1 to 1.0W RF am-plifiers as the transmitter outputstage. The main power device re-quires a negative gate-bias supply,which is not readily available in abattery-operated product. The circuitshown in Figure 1 not only developsa regulated negative gate bias, italso switches the positive supply,protects against the loss of gatebias, limits power dissipation inthe amplifier under high standing-wave ratio (SWR) conditions, andprotects against amplifier failuresthat might otherwise short-circuit thebattery pack.

Negative bias is supplied by anLTC1044 charge-pump inverter, andthe amplifier’s positive supply isswitched by an LTC1153 electroniccircuit breaker. An open-collector

switch can be used to turn theLTC1044 inverter off by groundingthe OSC pin (7). When off, the LTC1044draws only 2µA.

The negative output from theLTC1044 is sensed by a 2.5V referencediode (IC2) and Q2. With nonegative bias available, Q2 is off andQ3 turns on, pulling the LTC1153’scontrol input low. This shuts off theGaAs amplifier. Total standby power,including the LTC1044, is approxi-mately 25µA.

If the LTC1044’s OSC pin (7) isreleased, a negative output, nearlyequal in magnitude to the batteryinput voltage, appears at VOUT (pin 5).The negative bias is regulated by R1,IC2, and Q2’s base-emitter junction.Q2 saturates, shutting Q3 off andthereby turning the LTC1153 on.

The LTC1153 charges theN-channel MOSFET (Q4) gate to 10Vabove the battery potential, switching

Protected Bias forGaAs Power Amplifiers

Q4 fully on. Power is thus applied tothe GaAs amplifier.

The nominal negative bias is –3.2V,comfortably assuring the –2.5V mini-mum specified for the amplifier. Totalquiescent current, exclusive of theGaAs amplifier drain supply, is ap-proximately 1.5mA in the “on” state.

Short circuits or over-current con-ditions in the GaAs amplifier candamage the circuit board, the batter-ies, or both. The LTC1153 senses theamplifier’s supply current and turnsQ4 off if it is over 2A. After a timeoutperiod set by C6 (200ms) the LTC1153tries again, turning Q4 on. If theamplifier’s supply current is still toohigh, the LTC1153 trips off again.This cycle continues until the faultcondition is cleared. Under fault con-ditions the LTC1153’s STATUS pin (3)is low. As soon as the fault is cleared,the LTC1153 resets and normal op-eration is restored.

1044_1.eps

+

VS

DS

GATE

SD

IN

CT

STATUS

GND

V+

OSC

LV

VOUT

BOOST

CAP+

GND

CAP–

+

+

1

2

3

4

8

7

6

5

IC1 LTC1044CS8C1

1µF

Q1

C2 100nF

C3 1µF

C4 1µF

R1 3.3kΩ

R2 10kΩ

Q2*

IC2 LT1004CS8-2.5

R4 1MΩ

R3 1MΩ

C5 100nF

C6 220nF

FAULT

1

2

3

4

8

7

6

5

C7 1nF

C8 10µF

R5 1kΩ

R6 5.1MΩ

Q4 IRFR024

VDD

GATE BIAS

RF OUT

GaAsFET AMPLIFIER

R8 1MΩ

R7 50mΩ

Q3*

* ZETEX ZTX 384 ZETEX (516) 543-7100 OR MOTOROLA MMBT3904

7.2V (6 NiCd CELLS)

IC3 LTC1153CS8

OFF

ON

+by Mitchell Lee

Figure 1. Schematic diagram


DESIGN IDEAS

High-Current, Synchronous, Step-DownSwitching Regulator

The new LTC1149 can drive anexternal, N-channel MOSFET toachieve high-efficiency, high-currentpower conversion. The standardP-channel approach is preferred atoutput currents of under 2A; however,P-channel MOSFETs become adominant loss element at higher out-put currents, limiting overall circuitefficiency. Consequently, N-channelMOSFETs are better suited for use inhigh-current applications, since theyhave a substantially lower on-resistance than comparably pricedP-channels. The best P-channels havean on-resistance of 60mΩ, whereasN-channels with on-resistances lessthan 25mΩ are readily available. Thecircuit shown in Figure 1 uses the low-loss characteristics of N-channelMOSFETs, providing efficiency inexcess of 90% at an output currentof 5A.

The circuit’s operation is as follows:the LTC1149 provides a PDRIVE

output (Pin 4) that swings betweenground and 10V, turning Q3 on andoff. While Q3 is on, the N-channelMOSFET (Q4) is off because its gate ispulled low by Q3, through D2. Duringthis interval, the NGATE output (pin13) turns the synchronous switch(Q5) on, creating a low-resistancepath for the inductor current.

Q4 turns on when its gate is drivenabove the input voltage. This isaccomplished by bootstrappingcapacitor C2 off the drain of Q4. TheLTC1149 VCC output (Pin 3) suppliesa regulated 10V output that is used tocharge C2 through D1 while Q4 is off.With Q4 off, C2 charges to 5V duringthe first cycle in Burst ModeTM opera-tion and to 10V thereafter.

When Q3 turns off, the N-channelMOSFET is turned on by theSCR-connected NPN-PNP network(Q1 and Q2). Resistor R2 supplies Q2with enough base drive to trigger theSCR. Q2 then forces Q1 to turn on,

(TA) LOW ESR NICHICON (AL) UPL1J102MRH, ESR = 0.027Ω, IRMS = 2.370A SANYO (OS-CON) 10SA220M, ESR = 0.035Ω, IRMS = 2.360A PNP, BVCEO = 30V NPN, BVCEO = 40V SILICONIX NMOS, BVDSS = 60V, RDSON = 5Ω

C3 CIN

COUT Q1 Q2 Q3

VIN 12V TO 36V

VIN

PGATE

SENSE–

NGATE

PDRIVE

SENSE+

ITH

CT

SD2

SD1

VCC

VCC

CAP

C3 3.3µF

+

SGND PGND RGND

LTC1149-5

3

5

16

10

15

7

6

11

1

4

9

8

13

2

0V = NORMAL >2V = SHUTDOWN

R1 1kC4

3300pF X7R

CT 820pF

NPO 12 14

C4 0.001µF

Q3 VN2222LL

D2 1N4148

R5 100Ω

R6 100Ω

R3 470Ω

D1 1N4148

C1 0.1µF

+

Q1 2N3906

R2 10k

Q2 2N2222

R4 220Ω

C2 0.1µF

D3 MBR160

Q4 MTP30N06EL

L1 50µH

RSENSE 20Ω

COUT 220µF X 2 10V

+5V/5A

CIN 1000µF 63V

Q5 IRFZ34

NMOS, BVDSS = 60V, RDSON = 0.05Ω SILICON, VBR = 75V MOTOROLA SCHOTTKY, VBR = 60V KRL NP-2A-C1-0R020J, PD = 3W COILTRONICS CTX50-5-52, DCR = 0.21Ω, IRON POWDER CORE ALL OTHER CAPACITORS ARE CERAMIC

Q4, Q5 D1, D2

D3 RSENSE =

L1 =

+

1149_1.eps

Figure 2. LTC1149-5 (12V–36V to 5V/5A)high-current buck

supplying more base drive to Q2. Thisregenerative process continues untilboth transistors are fully saturated.During this period, the source of Q4is pulled to the input voltage. While Q4is on, its gate-source voltage isapproximately 10V, fully enhancingthe N-channel MOSFET.

Efficiency performance for this cir-cuit is quite impressive. Figure 2shows that for a 12V input the effi-ciency never drops below 90% overthe 0.6A to 5A range. At higher inputvoltages efficiency is reduced due totransition losses in the powerMOSFETs. For low output currents,efficiency rolls off because of quiescentcurrent losses.

Figure 1. LTC1149-5 (12V–36V to 5V/5A) using N-channel MOSFETs

OUTPUT CURRENT100mA50

EFFI

CIEN

CY (%

)

80

100

1A 5A

1149_2.eps

60

70

90

36V

24V

12V

by Brian Huffman


DESIGN IDEAS

New Device Cameos

been designed to be very efficient. Thestandby current with the three inputsswitched off is typically 0.01microamps. The quiescent currentrises to 95 microamps per channelwith the input turned on and thecharge pump producing 11V from a3.3V supply.

Micropower operation, coupledwith a power supply range of 1.8V to6V, makes the LTC1163 ideal for 2-to 4-cell battery-powered applications.The LTC1163 is also well suited for3.3V and 5V nominal supply applica-tions. The LTC1163 is available inboth 8-pin DIP and 8-pin SOIC pack-aging. The pin-out has been optimizedas a “flow-through” configuration, withthe three inputs and ground pin onone side and the three outputs andsupply pin on the other.

LT1204 Four-InputVideo Multiplexer withCurrent-Feedback Amplifier

The LT1204 is a four-input videomultiplexer designed to drive 150Ωcables and to expand easily into largerrouting systems. Wide bandwidth,high slew rate, and low differentialgain and phase make the LT1204ideal for broadcast-quality routing.Gain flatness is 0.1dB at 40MHz forAV = 2. Channel separation and dis-able isolation are 90dB at 10MHz. Thechannel-to-channel output switchingtransient is only 40mV, with a50ns duration, making the transi-tion imperceptible on high-qualitymonitors.

A unique feature of the LT1204 isits ability to expand into larger routingmatrices. This is accomplished bybootstrapping the feedback resistorsin the disable condition, raising thetrue output impedance of the circuit.This feature negates the effects ofpoor cable terminations in largesystems.

A large input range of ±6V makesthe LT1204 ideal for general-purposeanalog signal selection and multi-plexing. The new multiplexer operateson ±5V to ±15V supplies and a shut-down feature reduces the supplycurrent to just 1mA. The part is avail-able in a 16-lead plastic DIP (N)package or a 16-lead wide SO.

LT1381 5V-Powered RS232Transceiver in a 16-Lead,Narrow-Body SO Package

The LT1381 is a new two-driver/two-receiver RS232 interface trans-ceiver with an integral charge-pumppower generator. The circuit is avail-able in 16-lead narrow SOIC pack-ages, reducing board space by 27%over existing devices. The circuit isalso available in a 16-lead plastic DIP.

The circuit contains an on-chipcharge-pump generator, which usessmall 0.1µF ceramic chip capacitorsto supply RS232 output drive levelswhile operating from a single 5V powersupply. Power consumption is a low13mA max. RS232 I/O pins are pro-tected from ESD transients in excessof ±10kV, eliminating the need forexpensive TranZorbs. Driver outputsare capable of driving 2500pF loads todata rates in excess of 120k baud.Driver outputs are high impedancewhen powered down, and do not loadthe RS232 line as do CMOS devices.

LT1332 Wide-Supply-Range,Low Power RS232 Transceiver

The LT1332 is an extremely lowpower RS232 transceiver intendedfor systems that operate at low voltageyet require true RS232 outputlevels. The LT1332 is designed to bepowered from an external switchingregulator, which may be used else-where for power conditioning. In atypical application, the LT1332 shares

LT1180A, LT1181A,and LT1130A FamilyRS232 Transceivers

Three new RS232 interface trans-ceivers provide enhanced performanceand fault tolerance compared to ex-isting devices. The LT1180A andLT1181A are two-driver/two-receivertransceivers, pin compatible withthe existing LT1080/LT1081 andLT1180/LT1181. The LT1130A fam-ily consists of eleven devices that arepin compatible with the LT1130 fam-ily. Members of the LT1130A familyprovide up to five drivers and fivereceivers with a charge pump in asingle package.

All of the new circuits feature ±10kVESD protection on the RS232 linepins and operate to 120k baud whiledriving up to 2500pF loads. The on-chip charge-pump power generatorsare capable of using low-cost 0.1µFcapacitors to generate RS232 levelsfrom standard, 5V power supplies.Power consumption is reduced by asmuch as 43% from the earlier devices,while SHUTDOWN and DRIVERDISABLE operating modes allow fur-ther power savings in some systems.

All of these circuits are available inPlastic DIP or SOIC packages. TheLT1137A and LT1138A are also avail-able in 28-lead SSOP packages.

The LTC1163 Triple1.8V to 6V MOSFET Driver

The LTC1163 triple 1.8V to 6V gatedriver makes it possible to switcheither supply- or ground-referencedloads through low RDS(ON) N-channelswitches from as little as 1.8V (twodischarged cells). The LTC1163 con-tains three on-chip charge pumps sothat less expensive, lower RDS(ON)N-channel MOSFETs can be used toreplace high-side P-channel switches,which generally perform poorly below5V and cannot operate down to 1.8V.

The three charge pumps requireno external components and have

NEW DEVICE CAMEOS


DESIGN IDEAS

reduce board space. The LT1332 isavailable in SO, SSOP, and DIPpackages.

LT1259 Dual/LT1260 TripleCurrent-Feedback Amplifiers

The LT1259 contains two indepen-dent 100MHz current-feedbackamplifiers, each with a shutdownpin. These amplifiers are designed forexcellent linearity while driving cablesand other low impedance loads. TheLT1260 is a triple version, especiallysuited to RGB video applications.These amplifiers operate on all sup-plies from single 5V to ±15V and drawonly 5mA per amplifier when active.

When shut down, the LT1259/LT1260 amplifiers draw zero supplycurrent and their outputs becomehigh impedances. The amplifiers’ turn-on times are only 70ns and theirturn-off times are 140ns. In portableequipment and systems using severalamplifiers, the shutdown featurereduces system power without theproblems normally associated withpower supply switching. The shut-down feature also allows the outputsof several amplifiers to be wired to-gether to make a video mux amp.

Dual and triple amplifiers signifi-cantly reduce costs compared withsingles; board space and the numberof insertions are reduced and fewersupply-bypass capacitors are re-quired. The wide bandwidth and highslew rate of these amplifiers makedriving RGB signals easy. Only twoLT1260s are required to make a com-plete, two input RGB MUX and cabledriver.

The LT1259 is available in 14-pinDIPs and narrow S14 surface mountpackages. The LT1260 is available in16-pin DIPs and narrow S16 surfacemount packages.

the regulator’s positive output, whilecharge is capacitively pumped fromthe regulator’s switch pin to the nega-tive supply. Schottky rectifiers builtinto the LT1332 simplify the charge-pump design.

The LT1332/switcher combinationgenerates fully compliant RS232 sig-nal levels from inputs as low as 2V(e.g., two dead AA batteries) or as highas 6V (e.g., a poorly regulated 5Vsupply). The LT1109A can delivergreater than 100mA of output cur-rent, making it an excellent choice forpowering FLASH memory, while thesurplus power available to the LT1332is sufficient to drive long cables, heavyloads, or mice. When operated un-loaded, the LT1332 draws 1mA ofsupply current. While shut down, theLT1332 typically draws 40µA of sup-ply current which keeps one receiveractive for detecting start-up signals.

If external RS232 supplies are avail-able (6.6V < V+ < 13.2V; –13.2V < V–< –6.6V), the LT1332 can be used asa stand-alone unit. When used in thisway, the LT1332’s low supply currentmakes it an attractive alternative tothe LT1141.

The LT1332 is a complete RS232serial port with three drivers and fivereceivers, arranged in a flow-througharchitecture similar to that of thepopular LT1137. Advanced driveroutput stages operate up to 120kbaud while driving heavy capacitiveloads. New ESD structures on thedriver outputs and receiver inputsmake the LT1332 resilient to multiple±10kV strikes, eliminating costly tran-sient suppressors.

The LT1332 requires two externalcharge pump caps for the negativesupply, but only small decoupling capsare needed on the positive supplyinputs. The low parts count helps

For further information on theabove or any other devicesmentioned in this issue of LinearTechnology, use the reader servicecard or call the LTC literature-service number: (800) 637-5545.Ask for the pertinent data sheetsand application notes.

Information furnished by Linear TechnologyCorporation is believed to be accurate and reliable.However, no responsibility is assumed for its use.Linear Technology makes no representation thatthe circuits described herein will not infringe onexisting patent rights.

NEW DEVICE CAMEOS

LTC in the News . . .Third Quarter Earnings

Thanks to the continued patronageof our customers and acceptance ofour products and application solutionsin the marketplace, Linear TechnologyCorporation recorded its 30th consecu-tive quarter of increased sales andincome levels. Linear TechnologyCorporation net sales for the thirdquarter of 1993 reached a record$38,806,000, a 27% increase overthe third quarter of 1992. Third quar-ter net income also reached a record$9,571,000 or $0.26 per share, anincrease of 46% over last year (ad-justed to reflect the two-for-one stocksplit distributed on November 24, 1992).

Rankings & RatingsMagazines and newspapers annu-

ally rank leading companies on thebases of sales growth, profitability,return on equity and other criteria.Linear Technology has traditionallydone very well in all of the ratingsfor which the company qualified. Thisyear is no exception. Three of theyear’s recent evaluations are summa-rized below.

In Business Week ’s “Top 1000:America’s Most Valuable Companies,”based on its market value, Linearranked 812th in the U.S. compared to853rd last year, when LTC first ap-peared on this list. This ranking placedLinear Technology Corporation best inits class and 10th among all semicon-ductor manufacturers.

Financial World magazine rankedLinear Technology Corporation 1stamong all semiconductor companiesand 22nd among its “Top 200Growth Companies in the U.S.” rank-ing for 1993.

Upside magazine is a national maga-zine published in San Francisco thatfocuses on high technology companies.In this year’s “Upside 100” list, LinearTechnology placed 5th in market valuegrowth among all semiconductor com-panies and 31st within the 800 tech-nology companies in the survey.

Industry statistical analysts atIn-Stat in Scottsdale, Arizona, namedLinear Technology “the best financiallymanaged manufacturer in the semi-conductor industry” in its annualKachina Awards.

Bob Swanson, President of LinearTechnology, shares the honor of Finan-cial World “CEO of the Year” with theCEO of Loral and Andy Grove of Intel.


DESIGN IDEAS

DESIGN TOOLS

© 1993 Linear Technology Corporation/ Printed in U.S.A./20K

Applications on DiskNOISE DISKThis IBM-PC (or compatible) progam allows the user tocalculate circuit noise using LTC op amps, determine thebest LTC op amp for a low noise application, display thenoise data for LTC op amps, calculate resistor noise, andcalculate noise using specs for any op amp.Available at no charge.

SPICE MACROMODEL DISKThis IBM-PC (or compatible) high density diskette containsthe library of LTC op amp SPICE macromodels. Themodels can be used with any version of SPICE for generalanalog circuit simulations. The diskette also contains work-ing circuit examples using the models, and a demonstrationcopy of PSPICETM by MicroSim.Available at no charge.

Technical Books1990 Linear Databook — This 1440 page collectionof data sheets covers op amps, voltage regulators,references, comparators, filters, PWMs, data conversionand interface products (bipolar and CMOS), in both com-mercial and military grades. The catalog features well over300 devices. $10.00

1992 Linear Databook Supplement — This 1248 pagesupplement to the 1990 Linear Databook is a collection ofall products introduced since then. The catalog contains fulldata sheets for over 140 devices. The 1992 Linear DatabookSupplement is a companion to the 1990 Linear Databook,which should not be discarded. $10.00

Linear Applications Handbook — 928 pages full ofapplication ideas covered in depth by 40 Application Notesand 33 Design Notes. This catalog covers a broad range of“real world” linear circuitry. In addition to detailed, systems-oriented circuits, this handbook contains broad tutorialcontent together with liberal use of schematics and scopephotography. A special feature in this edition includes a 22page section on SPICE macromodels. $20.00

1993 Linear Applications Handbook Volume II —Continues the stream of “real world” linear circuitry initiatedby the 1990 Handbook. Similar in scope to the 1990 edition,the new book covers Application Notes 41 through 54 andDesign Notes 33 through 69. Additionally, references andarticles from non-LTC publications that we have founduseful are also included. $20.00

Interface Product Handbook — This 200 page handbookfeatures LTC’s complete line of line driver and receiverproducts for RS232, RS485, RS423, RS422 and AppleTalk

applications. Linear’s particular expertise in this area in-volves low power consumption, high numbers of driversand receivers in one package, 10kV ESD protection ofRS232 devices and surface mount packages.Available at no charge.

Monolithic Filter Handbook — This 232 page book comeswith a disk which runs on PCs. Together, the book and diskassist in the selection, design and implementation of theright switched capacitor filter circuit. The disk containsstandard filter responses as well as a custom mode. Thehandbook contains over 20 data sheets, Design Notes andApplication Notes. $40.00

SwitcherCAD Handbook — This 144 page manual, in-cluding disk, guides the user through SwitcherCAD—apowerful PC software tool which aids in the design andoptimization of switching regulators. The program can cutdays off the design cycle by selecting topologies, calculat-ing operating points and specifying component values andmanufacturer's part numbers. $20.00

AppleTalk is a registered trademark of Apple Computer, Inc.

LINEAR TECHNOLOGY CORPORATION1630 McCarthy BoulevardMilpitas, CA 95035-7487

(408) 432-1900Literature Department (800) 637-5545

FRANCELinear Technology S.A.R.L.Immeuble “Le Quartz”58 Chemin de la Justice92290 Chatenay MallabryFrancePhone: 33-1-46316161FAX: 33-1-46314613

GERMANYLinear Technology GMBHUntere Hauptstr. 9D-8057 EchingGermanyPhone: 49-89-3194710FAX: 49-89-3194821

JAPANLinear Technology KK5F YZ Building4-4-12 Iidabashi Chiyoda-KuTokyo, 102 JapanPhone: 81-3-3237-7891FAX: 81-3-3237-8010

KOREALinear Technology Korea BranchNamsong Building, #505Itaewon-Dong 260-199Yongsan-Ku, SeoulKoreaPhone: 82-2-792-1617FAX: 82-2-792-1619

SINGAPORELinear Technology Pte. Ltd.101 Boon Keng Road#02-15 Kallang Ind. EstatesSingapore 1233Phone: 65-293-5322FAX: 65-292-0398

TAIWANLinear Technology CorporationRm. 801, No. 46, Sec. 2Chung Shan N. Rd.Taipei, Taiwan, R.O.C.Phone: 886-2-521-7575FAX: 886-2-562-2285

UNITED KINGDOMLinear Technology (UK) Ltd.The Coliseum, Riverside WayCamberley, Surrey GU15 3YLUnited KingdomPhone: 011-44-276-677676FAX: 011-44-276-64851

InternationalSales Offices

CENTRAL REGIONLinear Technology CorporationChesapeake Square229 Mitchell Court, Suite A-25Addison, IL 60101Phone: (708) 620-6910FAX: (708) 620-6977

NORTHEAST REGIONLinear Technology CorporationOne Oxford Valley2300 E. Lincoln Hwy., Suite 306Langhorne, PA 19047Phone: (215) 757-8578FAX: (215) 757-5631

NORTHWEST REGIONLinear Technology Corporation782 Sycamore Dr.Milpitas, CA 95035Phone: (408) 428-2050FAX: (408) 432-6331

SOUTHEAST REGIONLinear Technology Corporation17060 Dallas ParkwaySuite 208Dallas, TX 75248Phone: (214) 733-3071FAX: (214) 380-5138

SOUTHWEST REGIONLinear Technology Corporation22141 Ventura BoulevardSuite 206Woodland Hills, CA 91364Phone: (818) 703-0835FAX: (818) 703-0517

U.S. AreaSales Offices

World HeadquartersLinear Technology Corporation1630 McCarthy BoulevardMilpitas, CA 95035-7487Phone: (408) 432-1900FAX: (408) 434-0507

LINEAR TECHNOLOGY LINEAR TECHNOLOGY...The LT1248 Power-Factor Corrector by Carl Nelson 1248_1.eps...

Documents

Transcript of LINEAR TECHNOLOGY LINEAR TECHNOLOGY...The LT1248 Power-Factor Corrector by Carl Nelson 1248_1.eps...