Interleaving of a Soft-Switching Boost Converter Operated in Boundary Conduction Mode

D. Gerber, J. Biela Power Electronic Systems Laboratory, ETH Zürich

Physikstrasse 3, 8092 Zürich, Switzerland

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3374 IEEE TRANSACTIONS ON PLASMA SCIENCE, VOL. 43, NO. 10, OCTOBER 2015

Interleaving of a Soft-Switching Boost ConverterOperated in Boundary Conduction Mode

Dominic Gerber and Juergen Biela, Member, IEEE

Abstract— This paper presents the interleaved operation of asoft-switching boost converter operated in boundary conductionmode. First, the operating principle of the converter as wellas the basic concept of the interleaving is presented. Then, thedynamic behavior is modeled using the z-transform to obtain aconverter model that is independent of the switching frequency.With the model, the stability of the closed-loop system with aproportional–integral (PI) controller is analyzed. It is shown thatan adaptive PI controller can be easily implemented to achievea minimal settling time over a wide operating range. Finally,the controller is validated with two converters with a 40-kWnominal output power and an output voltage of 3 kV. The testsat different output voltages under different load conditions showa stable interleaved operation.

Index Terms— Boost converter, capacitor charging,interleaving, soft switching.

I. INTRODUCTION

ACOMPACT and cost-effective X-ray free-electron lasersystem (SwissFEL) is currently built at the Paul Scherrer

Institute in Switzerland [1]. This laser system requiresmodulators with a pulse power of 127 MW for 3 μs. The highpulse power is provided by a capacitor bank that is rechargedto 3 kV by two 40-kW boost converters with a 1.25-kV inputvoltage between two consecutive pulses.

In the considered system, two interleavedconverters (Fig. 1) [2] are used to charge the capacitorbank. The interleaved operation results in a reduced inputcurrent ripple. In pulse-width modulation (PWM) controlledsystems with fixed switching frequency, the interleaving canbe easily controlled by shifting the PWM signals relative toeach other. However, this method does not work for convertersoperating in boundary conduction mode (BCM) that resultsin a variable switching frequency. Different open-loop andclosed-loop control methods for the operation in BCM havebeen investigated [3], [4]. However, those methods do notguarantee a soft-switched operation when the system isperturbed.

In this paper, a control strategy for interleaved boostconverters operated in BCM is presented. The proposedcontroller guarantees zero voltage switching during the

Manuscript received November 27, 2014; revised February 4, 2015;accepted April 6, 2015. Date of publication May 1, 2015; date of currentversion October 7, 2015.

The authors are with the Laboratory for High Power ElectronicSystems, ETH Zurich, Zurich 8092, Switzerland (e-mail: gerberdo@ethz.ch;jbiela@ethz.ch).

Color versions of one or more of the figures in this paper are availableonline at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPS.2015.2422476

Fig. 1. Prototype of the 1.25–3-kV and 40-kW boost converter.

interleaved operation including startup and the synchronizationof the converters.

The basic operation principle of the converter and theinterleaving is presented in Section II. In Section III, theconverter model and the controller model are introduced.Two different controllers are investigated, and the imple-mented controller is shown. Finally, measurements with theproposed controller and two 40-kW and 3-kV interleaved boostconverters are presented.

II. CONVERTER OPERATION

The basic circuit of the converter including the snubbercircuits is shown in Fig. 2. The details of the converter and thefeedback controller are presented and analyzed in [2] and [5].In the following, the converter operation in BCM and theinterleaved operation are explained.

A. Converter Operation

The series-connected switches S1a–S1n are turned ON at thebeginning of interval T1 (see Fig. 2). The input voltage Vinis applied across inductor L1, and inductor current iL startsto rise. When inductor current iL reaches the level iLp,switches S1a–S1n are turned OFF at the beginning of interval T2(peak current control). Diodes D1a–D1n are not conducting atthat time since capacitors CSn,D1a–CSn,D1n are still charged.Inductor L1 and snubber capacitors CSn form a resonantcircuit. Since inductor current iL is positive, the capacitors inparallel to the diodes D1a–D1n are discharged to zero and thecapacitors across the switches are charged to Vout. As soon as

0093-3813 © 2015 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.

GERBER AND BIELA: INTERLEAVING OF A SOFT-SWITCHING BOOST CONVERTER OPERATED IN BCM 3375

Fig. 2. Boost converter circuit and the corresponding BCM waveforms.

the voltage across the diodes reaches zero, they start to conduct(interval T3). At that point of time, the difference betweenVout and Vin is applied across L1 and current iL decreases.At the beginning of interval T4, iL reaches zero and the diodesstop conducting. The snubber capacitors and the inductor formagain a resonant circuit. The voltage across the inductor is thedifference between Vout and Vin. Hence, the inductor currentbecomes negative and the capacitors across the switches arecharged to zero if the difference between Vout and Vin is largeenough. At the end of T4, the body diodes of the switchesstart to conduct. At that point of time, the switches have to beturned ON before the inductor current becomes positive andthe next switching cycle starts.

The operation in BCM results in a switching frequency thatdepends on the input and output voltage, the output power, aswell as the inductance value.

B. Interleaved Operation

The inductor current waveforms for two interleavedconverters are shown in Fig. 3. Since the switching fre-quency depends on iLp, the phase shift ϕ = (�Tn/Ts,0)can be controlled by reducing or increasing iLp,1 relativeto iLp,0. The phase shift can be measured by detecting theinductor current zero crossings. One converter is used as a

Fig. 3. Interleaved inductor current waveforms.

Fig. 4. Current balancing of the average inductor currents depending oninductance mismatch �L = Lslave − Lmaster/Lmaster .

reference (master), whereas the other is used to adjust thephase shift (slave).

In an ideal system with identical converters, the switchingfrequencies for both converters are identical for the samecurrent iLp. This is not the case in a real system because ofcomponent tolerances, temperature drifts, and so on. In BCM,nonequal inductor values result in unbalanced inductorcurrents. Fig. 4 shows the relative deviation of the averageinductor current between the master and the slave dependingon the inductor mismatch at the nominal operating point of theinvestigated system (Table II). The deviation is smaller if theslave’s inductor is larger than the one of the master. Therefore,the converter with the smallest inductor value should beselected as the master.

In order to model the interleaved operation, a general modelwith nonidentical converters has to be developed.

III. FEEDBACK CONTROLLER

In this section, the interleaving model and the controllermodel are derived. In addition, the optimal controller para-meters are calculated. Afterward, the implemented phase shiftmeasurement and the controller are presented.

A. System Model

1) Plant Transfer Function: The inductor currentwaveforms are shown in Fig. 3. It is assumed that theinput and output voltage as well as iLp,0 remain constant.The dynamic behavior of the time offset between thetwo inductor currents is modeled by calculating the timeoffset for the next switching cycle �Tn+1 depending on the

3376 IEEE TRANSACTIONS ON PLASMA SCIENCE, VOL. 43, NO. 10, OCTOBER 2015

Fig. 5. Closed-loop control system of the interleaving model.

offset �Tn and the switching period lengths of the convertersTs,0(iLp,0) and Ts,1(iLp,1)

�Tn+1 = �Tn − Ts,0(iLp,0) + Ts,1(iLp,1). (1)

The switching periods Ts,k(iLp,k) can be analyticallycalculated by solving the differential equations describing thedynamic behavior of the converter. These expressions arenonlinear because of the resonant transitions during the timeintervals T2 and T4. In order to simplify the model, Ts,1(iLp,1)is linearized at iLp = iLp,0

Ts,1(iLp,1) ≈ Ts,1(iLp,0) + dTs,1(iLp)

diLp

∣∣∣∣iLp,0

�iLp (2)

with �iLp = iLp,1 − iLp,0.Since the continuous offset �Tn is discrete in time,

the z-transform is applied

Z{�Tn+1} = zZ{�Tn} = z�T (z). (3)

It is assumed that iLp,0 is constant. Hence, Ts,0(iLp,0),Ts,1(iLp,0), and (dTs,1(iLp)/diLp)|iLp,0 are constant.Using (1)–(3), one obtains

�T (z) = Gd(z) + G p(z)�ILp(z) (4)

with the disturbance transfer function

Gd (z) = 1

z − 1[Ts,1(iLp,0) − Ts,0(iLp,0)] = 1

z − 1�Ts

and the plant transfer function

G p(z) = 1

z − 1

dTs,1(iLp)

diLp

∣∣∣∣iLp,0

= 1

z − 1T ′

s,1(iLp,0).

2) Closed-Loop Transfer Function: In this section, theclosed-loop system shown in Fig. 5 is analyzed. The systemmodel including the controller is

�T (z) = Gc(z)G p(z)

1 + Gc(z)G p(z)�Tset(z)+ Gd(z)

1 + Gc(z)G p(z)= Gcl(z)�Tset(z) + Gdis (5)

where Gcl(z) represents the plant closed-loop transfer functionand Gdis the disturbance closed-loop transfer function.

B. Proportional–Integral Controller

Since a proportional (P) controller is not able to drive thesystem to the desired phase shift without a static deviation,a proportional–integral (PI) controller is investigated. It isknown from the control theory that a system is stable if andonly if its poles are inside the unit circle of the complexplain. In the following section, the stability of the controlleris investigated.

Fig. 6. Settling time in cycles of the closed-loop system with PI controller.

1) Stability: The closed-loop transfer function for aPI controller Gc(z) = K P + KI (z/z − 1) is

Gcl,PI(z) = (KI,n + K P,n)z − K P,n

z2 + ((KI,n + K P,n) − 2)z − K P,n + 1(6)

K P,n = K P T ′s,1(iLp,0) (7)

KI,n = KI T ′s,1(iLp,0). (8)

Hence, the dynamic behavior of the system depends onK P,n and KI,n , providing an operating point-independentdynamic behavior. Gcl,PI(z) has two poles at

p1,PI = − K P,n +KI,n −√

(K P,n + KI,n)2 − 4KI,n

2+ 1 (9)

and

p2,PI = − K P,n + KI,n + √

(K P,n + KI,n)2 − 4KI,n

2+ 1.

(10)

The system is stable for

0 < KI,n < 4 − 2K P,n (11)

and

0 < K P,n <4 − KI,n

2. (12)

2) Settling Time: The settling time of the system withPI controller is shown in Fig. 6. The plot shows a minimumsettling time of two switching cycles around K P,n = 1 andKI,n = 1. The step response of the controller is shownin Fig. 7. The overshoot of 100% does not matter in that casesince the model is a small-signal model.

3) Disturbance Transfer Function: The static controllererror can be investigated by applying a unit stepU(z) = (1/1 − z−1) and using the final value theorem.The static error of the disturbance transfer function

Gdis,PI(z) = (z − 1)�Ts

z2 + ((KI,n + K P,n) − 2)z − K P,n + 1(13)

is

limz→1

(z − 1)Gdis,PI(z)U(z) = 0. (14)

GERBER AND BIELA: INTERLEAVING OF A SOFT-SWITCHING BOOST CONVERTER OPERATED IN BCM 3377

Fig. 7. Step response for K P,n = 1 and K I,n = 1.

Fig. 8. Phase shift measurement.

Hence, a PI controller is able to drive the system to the desiredpoint independent of the period deviation �Ts .

C. Phase Shift Measurement

As already mentioned before, the inductor current zerocrossings are used to measure �Tn and Ts,0. However, a stableinterleaved operation is only possible, when the measurementrange of �Tn is larger than the setpoint range. Otherwise,setpoints close to the boundary of the measurement rangemight result in large steps in the error e = �Tset −�Tn when�Tn exceeds the measurement range. This in turn results inan unstable operation since these large steps are not presentin the real system. For the presented system, the followingsetpoint and measurement range are selected:

− Ts,0

2< �Tset <

Ts,0

2(15)

− Ts,0 < �Tn < Ts,0. (16)

With this measurement range, a method is required, whichis able to measure positive and negative values of �Tn andprovides a unique �Tn for every switching cycle. All possiblecases are shown in Fig. 8. All other cases indicate a toolarge deviation of the switching frequencies, which results ina controller reset.

D. Linearization

Since the plant transfer function is a linearized model,it has to be taken into account that the model is only anapproximation of the real system behavior. Fig. 9 shows theperiod length depending on the peak inductor current at thenominal operating point of the presented converter. The period

Fig. 9. Period length Ts depending on ilp at the nominal operating point.

Fig. 10. Block diagram of the implemented controller.

length shows a linear behavior at peak inductor currentsbigger than 15 A. Hence, the critical operating conditions areat low peak inductor currents.

E. Feedback Controller Implementation

The closed-loop transfer functions Gcl,P and Gcl,PI showthat the system dynamics depend on the product of T ′

s,1(iLp,0)and the controller coefficients. Hence, the optimal controllerparameters depend on the operating point. They can becalculated with

K P = K P,n

T ′s,1(iLp,0)

(17)

and

KI = KI,n

T ′s,1(iLp,0)

. (18)

The term T ′s,1(iLp,0) can be calculated analytically. This means

that it is possible to calculate the controller coefficients online.Since the sampling frequency of the system model is equalto the switching frequency, the integral part of the controlleris sampled with the switching frequency of the converters.A block diagram of the controller is shown in Fig. 10.

Since the derived model depends on the linearized systemmodel, it is necessary to limit the controller output to assure

3378 IEEE TRANSACTIONS ON PLASMA SCIENCE, VOL. 43, NO. 10, OCTOBER 2015

TABLE I

CONVERTER HARDWARE SPECIFICATIONS

that the linearized model is a good approximation of thesystem dynamics. This also makes sure that the switchingfrequencies of both converters are close enough to each otherin order to prevent one converter to skip switching cycles orswitch multiple times during one switching cycle of the otherconverter. The slave setpoint iLp,1 is limited after the controlleroutput to ensure zero voltage switching (ZVS) and to avoidexcessive thermal stress.

IV. MEASUREMENTS

In the following, the converter and the test setup aredescribed, and the test results are presented.

A. Converter

1) Semiconductors and Snubber Circuits: Since the con-verter is operated under ZVS conditions, MOSFETs are thebest choice as semiconductor switches. However, the outputvoltage of 3 kV requires the series connection of multipleMOSFETs since there are no devices available with therequired blocking voltage. The presented system uses 650 VMOSFETs of the type STY139N65M5. As an alternative,MOSFETs of the type IPW65R019C7 could also be used.Eight switches are connected in series, resulting in a blockingvoltage of 5200 V, providing enough margin to ensure areliable operation. Two switches are connected in parallel toreduce the conduction losses, resulting in 16 switches in total.A snubber capacitance of 11 nF is connected in parallel toeach switch to ensure a voltage balancing between the switchesduring the switching transients.

The voltage balancing of the switches is determined bytwo effects: 1) component tolerances of the snubber capac-itances and 2) switching signal jitter of the individualMOSFETs. The voltage balancing between n capacitorsconnected in series is given by the capacitance values. Thevoltage vi of a capacitor Ci in series with a capacitor Ck isgiven by

vi = Ck

Civk . (19)

At the beginning (T1 in Fig. 2), all snubber capacitors aredischarged.

The total blocking voltage is simply the sum of allvoltages vi

n∑

i=1

vi =n∑

i=1

Ck

Civk = vtot. (20)

Ideally, vk is vtot/n. The resulting ratio of vk and vtot/n isvkvtotn

= n∑n

i=1CkCi

. (21)

The most interesting case is when vk is higher than the idealvoltage vtot/n. This occurs, when the capacitor Ck is thesmallest capacitor. If it is assumed that all capacitors are withinCmin and Cmax, the worst case is

vk,maxvtotn

= n

1 + (n − 1) CminCmax

. (22)

By assuming a capacitance tolerance of ±5%, the resultingvoltage imbalance is approximately 9.1% or 34 V for thepresented converter.

The nonsynchronous switching of the MOSFETs alsoresults in an unequal voltage distribution of the switches. Themain sources for the nonsynchronous switching are differentpropagation delays of the individual switching signals and gatedrive circuits. The voltage difference between two switches isestimated by

�V = i0�T

Csn,S1x(23)

where i0 is the charging current of the snubber capacitorsat the turn-OFF instance, �T is the difference between theturn-OFF time instance, and Csn,S1x is the snubber capacitancein parallel to the switch. The propagation delay difference �Tfor the presented system was measured and determined to beless than 20 ns. For a current i0 of 100 A and a capacitancetolerance of ±5%, the resulting voltage �V is 96 V. Takingthe capacitance tolerance and the switching signal jitter intoaccount, this results in a worst case blocking voltage of 505 V.The maximum blocking voltage of the MOSFETs is 650 V ata 25 °C junction temperature. Since the occurring overvoltagecaused by the inductance in the commutation path during theturn-OFF transients is small if the snubber capacitances areconnected as low inductive as possible to the switches, theselected snubber capacitance values provide enough margin.

Four 1200 V diodes of the type APT175DQ120BGconnected in series are used at the output side. A 1-nF snubbercapacitor is connected in parallel to each diode. An additionalcapacitor of 4 nF is connected in parallel to all diodes,resulting in an equivalent capacitance of 7 nF. This additionalcapacitance is used to reduce the influence of the nonlinearcapacitance of the semiconductors, resulting in a wider ZVSoperating range.

In addition, a resistor is connected in parallel to eachsemiconductor to assure a balanced voltage at dc.

All semiconductors are mounted on a water-cooled heatsink. Because the semiconductors are at a different electricalpotential during operation, a thermally conductive insulationis required. The switching node is switched between 3 kVand 0 V at high frequencies of up to 200 kHz. This leads tocapacitive currents that are heating up the insulating material,reducing the voltage withstand capability [6] at highfrequencies. Film materials are not suitable in this appli-cation since the small thickness results in large capacitivecurrents. Hence, the presented system uses thermallyconductive ceramic.

GERBER AND BIELA: INTERLEAVING OF A SOFT-SWITCHING BOOST CONVERTER OPERATED IN BCM 3379

Fig. 11. Block schematic of the control.

2) Inductor: The inductor is split into two inductorsconnected in series. Four U/I cores of the type I 93/28/30 andU 93/76/30 manufactured by EPCOS are used per inductor.A litz wire with 4000 strands and a core diameter of 50 μmis used. The inductor consists of seven turns and an air gapof 5.6 mm. The inductors are air cooled by three 120-mm fans.

3) Control: The control of the converter is implementedin an field-programmable gate array. Each converter has itsown control, which assures ZVS as well as several protectionmechanisms. The feedback control consists of four blocks: 1)the output voltage controller; 2) the interleaving controller;3) a master converter selector; and 4) a state machine thatcontrols the number of operating converters (Fig. 11). Theoutput voltage controller sets the peak inductor current ofthe master converter, whereas the interleaving controller setsthe peak current for the slave converter. The operation ofboth converters is controlled by a superordinate state machine.The interleaving is disabled toward the end of the chargingcycle to achieve a high repetition accuracy. In addition, thesuperordinate state machine features a trigger function. If thetrigger function is activated, a charge retention mode is enteredand the last switching cycle is suppressed. A feedback signalindicates to the superior control that the converter is readyto execute the last switching cycle. When the output voltagefalls below a level where more than one switching cycle isrequired to reach the targeted output voltage, the capacitorsare charged again until only one switching cycle is left. Assoon as the superior control unit sends the trigger signal, thelast switching cycle is immediately executed. This assures adefined time instance when the capacitors are fully charged,further increasing the precision of the complete system. Thehardware specifications are shown in Table I.

B. Test Setup

The interleaved operation is tested with two 40-kWand 3-kV converters. The converter parameters are shownin Table II. The two converters are connected in parallel toan ohmic load, as shown in Fig. 12. Although the converteris designed to be operated with a capacitive load, an ohmicload has been chosen to test the interleaved operation. Theinterleaving works properly only when the operating pointremains more or less constant. This is the case for a largecapacitive load as well as for an ohmic load. Hence, an ohmicload is used since it simplifies the tests.

TABLE II

SPECIFICATIONS OF THE PROTOTYPE SYSTEM.THE NOMINAL OPERATING POINT IS BOLD

Fig. 12. Setup for interleaving measurements.

Fig. 13. Measured inductor current waveforms and phase shift at an inputvoltage of 1020 V, an output voltage of 2.5 kV, and an output power of 4.5 kW.

C. Results

The interleaved operation was successfully tested withtwo different ohmic loads of 1.4 and 215 �. The outputvoltages was set between 750 V and 3 kV. The operationat low output power is more critical than the operation withhigh output power because of the higher switching frequencyand because the relative error of the linearized term is biggerat input currents close to the free-wheeling current. As anexample, the measured current waveforms of the inductorcurrent and the resulting phase shift at 2.5 kV with a 1.4-k�load are shown in Fig. 13.

The deviation of the setpoint during the operation canbe explained by the high switching frequency and the jitterof the zero-crossing detection and the accuracy of theinternal current measurement of the converter. A phase shift of10° corresponds to a shift of approximately 200 ns at a140-kHz switching frequency. The used current sensor has ameasurement range of 100 A. An accuracy of 0.5% alreadycorresponds to a deviation of 50 ns of the ON-time of theswitch at the measured operating point.

Fig. 14 shows the inductor current and the waveform at aninput voltage of 1250 V and an output voltage of 3 kV after the

3380 IEEE TRANSACTIONS ON PLASMA SCIENCE, VOL. 43, NO. 10, OCTOBER 2015

Fig. 14. Measured inductor current waveforms and phase shift afterturning ON the slave converter at an input voltage of 1250 V, an output voltageof 3 kV, and an output power of 6.5 kW.

slave converter has been enabled. The phase shift is adjustedby the controller within a few cycles. The deviation after thephase shift has been adjusted is caused by the output voltagecontroller since it changes the setpoint of the master converterbecause of the doubled output power after the slave converterhas been turned ON.

V. CONCLUSION

In this paper, the interleaved operation of a soft-switchingboost converter operated in BCM is investigated. The soft-switching operation as well as the interleaving concept ispresented.

A general model for interleaved operation is derived usingthe z-transform in order to obtain a switching frequency-independent model. Afterward, the stability of the closed-loopsystem with a PI controller is investigated. In addition, it isshown that the dynamic behavior of the small-signal model ofthe system is independent of the operating point for normalizedcontroller coefficients. An adaptive interleaving controller ispresented.

Finally, the controller has been implemented and success-fully tested with two converters at different loads at an outputvoltage ranging from 750 to 3000 V.

ACKNOWLEDGMENT

The authors would like to thank Ampegon AG andAmpegon PPT GmbH for the construction of the convertersand the tests with the 215-� load.

REFERENCES

[1] (Mar. 2014). SwissFEL. [Online]. Available: http://www.psi.ch/swissfel/[2] D. Gerber and J. Biela, “Charging precision analysis of a 40-kW 3-kV

soft-switching boost converter for ultraprecise capacitor charging,” IEEETrans. Plasma Sci., vol. 42, no. 5, pp. 1274–1284, May 2014.

[3] L. Huber, B. T. Irving, and M. M. Jovanovic, “Open-loop controlmethods for interleaved DCM/CCM boundary boost PFC converters,”IEEE Trans. Power Electron., vol. 23, no. 4, pp. 1649–1657, Jul. 2008.

[4] H. Choi and L. Balogh, “A cross-coupled master–slave interleavingmethod for boundary conduction mode (BCM) PFC converters,” IEEETrans. Power Electron., vol. 27, no. 10, pp. 4202–4211, Oct. 2012.

[5] D. Gerber and J. Biela, “Charging precision analysis of a 40 kW, 3 kVsoft-switching boost converter for ultra precise capacitor charging,” inProc. 19th IEEE Pulsed Power Conf., Jun. 2013, pp. 1–8.

[6] W. Pfeiffer, “High-frequency voltage stress of insulation. Methodsof testing,” IEEE Trans. Elect. Insul., vol. 26, no. 2, pp. 239–246,Apr. 1991.

Dominic Gerber received the M.Sc. degree inelectrical engineering and information technologyfrom ETH Zurich, Zurich, Switzerland, in 2010,where he is currently pursuing the Ph.D. degree withthe Laboratory for High Power Electronic Systems.

He was involved in power electronics, drivesystems, and high voltage technology. His currentresearch interests include solid-state modulators,high accurate capacitor charging, and currentmeasurement based on the Faraday effect.

Juergen Biela (S’04–M’06) receivedthe Diploma (Hons.) degree from theFriedrich-Alexander-Universität Erlangen-Nürnberg,Nuremberg, Germany, in 1999, and the Ph.D. degreefrom ETH Zurich, Zurich, Switzerland, in 2006,where he is currently pursuing the Ph.D. degreewith a focus on optimized electromagneticallyintegrated resonant converters with the PowerElectronic Systems Laboratory (PES).

He dealt in particular on resonant dc-link inverterswith the University of Strathclyde, Glasgow, U.K.,

and the active control of series-connected Integrated gate-commutatedthyristor with the Technical University of Munich, Munich, Germany.He joined the Department of Research, Siemens Automation and Drives,Erlangen, Germany, in 2000, where he was involved in inverters withvery high switching frequencies, SiC components, and electromagneticcompatibility. He was a Post-Doctoral Fellow with PES and a GuestResearcher with the Tokyo Institute of Technology, Tokyo, Japan, from2006 to 2007. He was a Senior Research Associate with PES from 2007to 2010. He has been an Associate Professor of High-Power ElectronicSystems with ETH Zurich since 2010. His current research interests includedesign, modeling, and optimization of power factor correction, dc–dc andmultilevel converters with an emphasis on passive components, the designof pulsed-power systems, and power electronic systems for future energydistribution.