Implementation of a Solar Power Battery Energy Storage System with Maximum Power Point Tracking

Yu-Lung Ke Senior Member, IEEE

National Penghu University 300 Liu-Ho Road, Makung City Penghu County, Taiwan, R.O.C.

benjamin.ke@npu.edu.tw

Ying-Chun Chuang Kun Shan University 949 Da-Wan Road

Yung-Kang City, Tainan County Taiwan, R.O.C.

chuang@mail.ksu.edu.tw

Bo-Tsung Jou Kun Shan University 949 Da-Wan Road

Yung-Kang City, Tainan County Taiwan, R.O.C.

Yuan-Kang Wu National Penghu University

300 Liu-Ho Road, Makung City Penghu County, Taiwan, R. O. C.

wuyuankang@npu.edu.tw

Abstract -- This work implements a solar power battery

energy storage system (BESS) with maximum power point tracking (MPPT) under substantial variation in temperature and intensity of illumination. A tracker is also designed based on the perturbation and observation method to track rapidly the maximum power point of the energy output of the solar cells. The power generation data are then transmitted to computers via RS232 for analysis. Next a battery charger is developed using a dsPIC30F chip. Additionally, the maximum power is output following a photovoltaic transformation by varying the illumination intensity. Moreover, the power generated by the solar cells is fed back to the lead-acid battery to supply the load.

Index Terms--maximum power point tracking (MPPT), battery energy storage system (BESS), battery charger, snubber.

INTRODUCTION Taiwan imports over 97% of its energy due to a scarcity of

natural resources. Thermal power generation continues to be the major power generation source in Taiwan, where thermal power generation, including coal, oil, and gas, exceeds 67% of overall capacity. However, as global warming and the greenhouse effect pose an environmental threat despite the relatively inexpensive power generation cost of nuclear energy, a significant amount of nuclear waste materials are generated through processing and security. Additionally, generating thermal power is rather difficult owing to public opposition. Whether thermal power or nuclear energy influences the environment and security reflects the increasing importance worldwide to adopting green energy power generation technologies.

Commonly referred to as renewable energy, green energy includes wind power, hydraulic power, terrestrial heat, solar energy, difference in temperature of seawater, as well as tide and fuel cells, implying the feasibility of using energy repeatedly without possible deficiency. As a latecomer to developing green energy, Taiwan is extremely limited in qualified personnel and financial resources. Additionally, the power generation cost of renewable energy exceeds that of a

thermal power and nuclear power without considering the pollution costs of generating thermal and nuclear power. Other energy applications are limited natural resources in addition to fuel cells. Wind power and solar power are the most promising renewable energy technologies under development in Taiwan. However, sparsely populated regions are the only appropriate areas for establishing a wind power generation facilities due to noise levels. Wind power generation requires a large scale installation capacity and yields low economic benefit. A notable example is the Miaoli wind power generation facility of Formosa Plastics Group. Solar power generation thus warrants further attention, with an emphasis on geographic environmental conditions and large scale installation.

French scientist A. E. Becquerel pioneered the photovoltaic effect in 1839, largely via energy conversion from the semiconductor photovoltaic effect. Bell laboratory invented the solar cell in 1954 with the intension of providing electricity to remote areas. At that time, solar cell efficiency was only around 6%. When the former Soviet Union launched its first man-made satellite in 1957, solar cells profoundly impacted space flight. The landing of American astronauts on the moon in 1969 streamlined the use solar cells. After 1990, advances in semiconductor technology led to the integration of power generation from solar cells with power consumption in daily live. The retail price and conversion efficiency of solar energy module represent the major limitations in developing solar cells.

II. CIRCUIT CONFIGURATION AND CIRCUIT ANALYSES OF FLYBACK CONVERTER

While attempting to charge a battery with a voltage of 12V and utilize output power from solar panels, this work adopts the flyback converter to control the charging process. Figure 1 schematically depicts the disposition diagram of the main circuit configuration of the proposed system.

978-1-4244-6395-4/10/$26.00 ©2010 IEEE

Fig. 1. Disposition diagram of the main circuit configuration of the developed system

mLiΔ

mLi

si

Di

Ci

RVbt−

⎟⎠⎞

⎜⎝⎛−

21

NNVbt

sT

sTd1 sTd2

Fig. 2. Waveforms of related components operated in a continuous current mode

Several assumptions are made before the circuit analyses: 1. An extremely high output capacitance occurs such that

output voltage Vo is a constant. 2. The circuit is operated at steady state, in which all voltages

and currents are periodic.

3. Duty ratio of the switch is D with a closed time DT and open time (1–D)T; and

4. Switches and diodes are ideal. Stage I:0 < t < t1, SW is turned ON, Dm is turned OFF

While solar panels charges the battery, the microchip outputs a set of pulse width modulation (PWM) signal to control the switch. Solar power energy is stored into the inductor Lm based on the duty cycle of the switch SW through means of conduction or cut off by high frequency switching on the power switch SW to control the charging current while charging. Figure 3 indicates that the inductor Lm stores energy when the switch SW is turned on.

Fig. 3. Current direction when the switch is turned on.

Voltage and current rate of the power source side are expressed as follows when the switch is turned on.

dtdiLm

mpv LVV ==1

L

V

m

sLmLmLm

DTi

ti

dtdi =Δ=

ΔΔ=

Current variation in the magnetized inductor of the transformer is calculated by Eq. (1).

( )L

V

m

pvconductedLm

DTi =Δ

Circuit equations on the load side of the transformer are summarized as follows.

⎟⎠⎞

⎜⎝⎛=⎟

⎠⎞

⎜⎝⎛=

N

NV

N

NVV pv

1

2

1

212

01

2 <⎟⎠⎞

⎜⎝⎛−−=

N

NVVV pvbtd

i2=0 i1=0

Because the diode is cut off, i2 equals zero, i.e. i1 equals zero. Consequently, the current flowing through the magnetized inductor Lm increases linearly when the switch is turned off and no current flows through the winding in an ideal transformer. In an actual transformer, this phenomenon implies that the current in the primary winding increases linearly and the secondary winding contains no current. Stage II:t1 < t < Ts, SW is turned OFF, Dm is turned ON

When the switch is turned off (Fig. 4), the current flowing through the inductor Lm cannot change instantaneously.

(1)

Therefore, the conduction path must pass through the primary side of the ideal transformer. By assuming that the output voltage still maintains a constant Vbt, the voltage at secondary side of the transformer becomes –Vbt.

Fig. 4. Current direction when the switch is turned off

The voltage across inductor Lm is generated.

N

NVV bt

2

11 −=

The voltage and current are expressed as follows when the switch is turned off.

VV bt−=2

N

NV

N

NVV bt

2

1

2

121 −==

N

NVVL bt

Lmm

dtdi

2

11 −==

( ) N

N

L

V

m

btLmLmLm

TDi

ti

dtdi

2

1

1−=

−Δ=

ΔΔ=

Current variation in the magnetized inductor of transformer is obtained when the switch is open.

( ) ( )N

N

L

V

m

btcutoffLm

TDi2

11−−=Δ

Before the next period, the net current change in the inductor at steady-state operations must be zero, and the following equations are obtained from Eq. (1) and (2).

( ) ( ) 0=Δ+Δ cutoffLmconductedLm ii

( ) 012

1pvV =−−N

N

L

V

L m

bt

m

TDDT

Next, Vbt is calculated by Eq. (3).

N

NVV

DD

pvbt1

2

1−=

The relationship between the input and the output of a flyback converter resembles that of a buck-boost converter, except for that a term of turn ratio of a transformer.

The other voltage and current expressions when the switch is turned off are listed as follows.

⎟⎠⎞

⎜⎝⎛=⎟

⎠⎞

⎜⎝⎛−=

N

N

N

NLmD iii

2

1

2

11

⎟⎠⎞

⎜⎝⎛+=−=

N

NVVVVV btpvpvsw

2

11

Ri V bt

R =

Riiii V

N

N btLmRDC −⎟

⎠⎞

⎜⎝⎛=−=

2

1

Voltage VSW across the switch that is turned off is larger than the power source voltage. A situation in which the output voltage equals the input voltage with a turn ratio of unity implies that the voltage across the switch is twice that of the power source.

III. DESIGN OF BASIC CIRCUIT CONFIGURATION Figure 5 illustrates the basic configuration of flyback

converter with elements including a transformer, diode Dm, transistor SW, output filtering capacitor C and output load R. Energy is stored in the transformers if the power transistor is conducted. Additionally, energy is transferred to the output terminal when the power transistor is turned off and energy is transferred incompletely before the next operation. Energy is still stored in the transformer, which is the current in the transformer that does not decline to zero. The circuit that is operating is called continuous current mode (CCM) or an incomplete energy conversion mode. Otherwise, the transformer current decreases to zero before the next conduction and then operated in discontinuous current mode (DCM) or a complete energy conversion mode.

sV

LmI mL

mD

C R

1N 2N

1V

SWV

2V

Fig. 5. Basic configuration of a flyback converter. A. Transformer Design

Under ideal circumstances, power absorbed by the load resistor must equal the power delivered by the source, i.e.

OS PP =

or

RV

IV btspv

2

=

Equation (5) shows the relationship between average power source current IS and average magnetized inductor current ILm.

( ) ( )DI

TDTII Lm

LmS ==

Average magnetized inductor current ILm is determined by

incorporating IS into Eq. (4).

(2)

(3) (4)

(5)

RDI V

V btLmpv

2

=

DRI

V

V

pv

btLm

2

=

Average current flowing through the inductor can be expressed as Eq. (7) by using Vpv in Eq. (3).

( ) ( ) ⎟⎠⎞

⎜⎝⎛

−=⎟

⎠⎞

⎜⎝⎛

−=

N

NV

N

NV

RDDDI btpv

Lm1

22

1

22 11

The maximal and minimal current through the inductor can be obtained from Eq. (7) and (1).

2max,Lm

LmLmiII Δ+=

( ) L

V

N

NV

m

pvpv DTRD

D21

2

1

22 +⎟

⎠⎞

⎜⎝⎛

−=

2min,Lm

LmLmiII Δ−=

( ) L

V

N

NV

m

pvpv DTRD

D21

2

1

22 −⎟

⎠⎞

⎜⎝⎛

−=

Notably, ILm,min in Eq. (9) must exceed zero in the continuous current operations. The boundary conditions for continuous and discontinuous current operations are listed as follows, where f denotes the switching frequency.

0min, =LmI

( ) fDDT

RDD

L

V

L

V

N

NV

m

pv

m

pvpv

221 1

22 ==⎟

⎠⎞

⎜⎝⎛

−

The minimal value of Lm at continuous current operations is determined as Eq. (10).

( ) ( ) 2

2

12

min 21

⎟⎠⎞

⎜⎝⎛−=

N

N

fRDLm

B. Output Capacitance Design

In the above analysis, the output voltage remains constant under the assumption of an extremely large capacitance. In practice, the output voltage does not remain constant for a capacitor with a finite value. The capacitor current can be determined based on the output voltage fluctuation or ripple via the relationship between the voltage and current of the capacitor.

RDC iii += Ripple of the output of flyback converter can be

determined from the current waveform of the capacitor in Fig. 5(a).

VV

btbt CDT

RQ Δ=⎟

⎠⎞

⎜⎝⎛=Δ

Next, ΔVbt is obtained by arranging Eq. (12).

RCfD

RCDT VV

Vbtbt

bt ==Δ

In Eq, (13), ΔVbt represents the ripple of peak-to-peak output voltage, as shown in Fig. 5(b), and can be obtained by arranging the above equation.

RCfD

V

V

bt

bt =Δ

Finally, the required capacitance can be obtained according to Eq. (14), and Eq. (14) can be rewritten as Eq. (15).

Rf

DC

V

V

bt

btΔ=

The required capacitance can be determined based on Eq.

(15), where the general expectation value of V

V

bt

btΔ

is

within 0.2%.

QΔ

VbtΔ

Fig. 5.(a)current in the capacitor(b)ripple voltage of the capacitor

IV. PERIPHERY CIRCUIT DESIGN

A. Photo Coupling Driving Circuit This work adopts the photo coupling amplifier TLP250 as

a driving signal of isolation and amplification to drive the switch component. Figure 6 shows the photo coupling driving circuit.

Fig. 6. Photo coupling driving circuit

B. Voltage Detection Circuit

The voltage of the battery entering the dsPIC must be reduced since analog/digital (A/D) conversion of the dsPIC accepts only an input voltage within 0~5V. The simplest approach is to divide the terminal voltage of the battery and then deliver it into the A/D port of the dsPIC, as shown in Fig. 7.

(6)

(10)

(8)

(9)

(11)

(12)

(13)

(14)

(15)

Fig. 7. Voltage detection circuit

C. Current Detection Circuit

The current feedback circuit of solar energy converts the current to a voltage analog signal and, then, converts an analog signal to a digital signal for internal operations via built-in 10 bits A/D converter in the digital signal processor. This study utilizes Hall IC sensor HX 05-P as the current detection circuit, as shown in Fig. 8. A situation in which the current flows from IP+ to IP- implies not only that the VIout generates a corresponding voltage signal with 0~5V, but also that a signal can be delivered to the A/D port of the dsPIC. The sensor can also detect the current direction. Additionally, a situation in which VIout equals 2.5V implies that no current flows. VIout is between 2.5V and 5V, i.e. the current direction is positive. Otherwise, the current direction is negative when VIout ranges between 0V and 2.5V. This study utilizes the above characteristics to evaluate whether it is a charging or discharging current.

Fig. 8. Current detection circuit

D. Snubber Circuit on the Transistor

The Snubber circuit on the transistor protects the components away from a high voltage and high current switching stress during the switching period. The switching state-transition analysis of the developed circuit depends on Kirchhoff’s law.

The diode is turned off and the transistor bears the weight of the load current when the transistor is in a conduction state. When the transistor is turned off, the diode maintains a reverse-biased situation till the transistor voltage VQ increases to equal the power source voltage VS and the load voltage VL decreases to zero. While the transistor voltage reaches VS, the diode current increases to IL when the transistor current decreases to zero.

The diode bears the entire load current when the transistor is turned off. During the turn-on period, the transistor voltage is not lower than VS until the diode is turned off. At this

moment, the transistor bears the entire load current and the diode current is zero.

Consequently, a triangular instantaneous power waveform can be obtained, as shown in Fig. 9(b) and 9(c). The voltage and current of the transistor are at a high level simultaneously when the switch is turned off.

LI

Qi

QV

Qi

QV

LI

St ft

Fig. 9. (a) converter model for switching the inductor load (b) voltage and current waveform during the switching period (c) instantaneous power waveform of the transistor

Figure 10 (a) shows a typical snubber circuit, where the

snubber provides another load current path when the transistor is turned off. The voltage across the transistor increases when the transistor is turned off. Next, the snubber diode DS becomes forward-biased and the capacitor CS starts to charge. The voltage rate of the transistor voltage decreases due to the capacitor, subsequently delaying the transition from a low-level voltage to a high-level one. When the transistor is turned off (Fig. 10(c)), the capacitor CS charges until the voltage of the turned off state appears. When the transistor is conducted (Fig. 10(b)), the capacitor discharges via resistor RS of the snubber and the transistor.

LI

QiSD SR

SC

Ci

LI

QiSD SR

SC

Ci

LI

QiSD SR

SC

Ci

Fig. 10. Description of snubber circuit and operations The capacitance value of the snubber determines the rising

rate of a voltage across the transistor when the transistor is turned off. The transistor bears the weight of the load current before the switch is turned off and the transistor current decreases approximately linearly to zero during the turned off period. The snubber capacitor bears else a load current until the switch is turned off completely. The transistor current and

snubber current during turned off period can be expressed as follows.

( )⎪⎩

⎪⎨

⎧⎟⎟⎠

⎞⎜⎜⎝

⎛−

=

0

1f

LQ t

tIti

f

f

tt

tt

≥

≤≤0

( )

( )

⎪⎪⎪

⎩

⎪⎪⎪

⎨

⎧ =−

=

0

L

f

LQL

C I

ttItiI

ti

x

xf

f

tt

ttt

tt

≥

<≤

≤≤0

Where xt denotes the time that it takes for the capacitor voltage to reach the final value. Figure 11 shows the waveforms of a current and voltage of the transistor at various value of capacitance C. A small damping capacitance causes the voltage to reach Vs before the transistor current reaches zero. However, the voltage requires a longer time to reach Vs for a larger capacitance. According to Fig. 11, the energy absorbed by the transistor is decreased during the switching period with an increased capacitance of the snubber.

Qi

Qi

Qi

CV

CV

CV

xt ft t

t

t

xf tt =

ft xt

fV

( )tp

( )tp

( )tp

Fig. 11. Waveforms of the snubber circuit with various capacitances during

turned off period, where the capacitance(a)＜(b)＜(c)

The capacitor is selected based on the required voltage

once the transistor current reaches zero. The voltage across the capacitor in Fig. 11(c) can be expressed as Eq. (18).

( ) ( ) ( )

⎪⎪⎪⎪

⎩

⎪⎪⎪⎪

⎨

⎧

+−=+

=

= ∫

∫

s

fLf

Lfc

t

t L

t

f

L

f

L

C

V

CtI

ttCItvdtI

C

CttIdt

ttI

C

tvf 2

1

21

0

2

x

xf

f

tt

ttt

tt

≥

≤≤

≤≤0

A situation in which the switch current reaches zero before the capacitor is fully charged implies that the capacitor voltage can be decided by the first part of Eq. (18). Let

( ) ffc Vtv = and Eq. (18) can be arranged as Eq. (19).

( )CtI

CttI

V fL

f

fLf 22

2

==

Next, C is determined using Eq. (18), where Vf denotes the required voltage when the transistor current reaches zero and Vf ( sf VV ≤ ).

f

fL

VtI

C2

=

The capacitor discharges before the transistor is turned off by selecting resistor Rs. The capacitor discharge requires three to five times of a time constant. The conduction time of

the transistor is then RCton 5> or C

tR on

5< , assuming that

there are five time constants. The capacitor discharges via the resistor and the transistor when the transistor is turned on and the energy stored in the capacitor is determined by Eq. (21).

2

21

sCVW =

The energy is almost entirely transferred to the resistor during the conduction period of the transistor. Additionally, the power absorbed by the resistor is determined based on Eq. (22), where T denotes the switching period and f represents the switching frequency.

fCVT

CVP s

s

R2

2

212

1

==

Obviously, power loss in the snubber is directly proportional to the snubber capacitance. Despite decreasing the power losses of the transistor, the high capacitance increases the dissipation of power losses in the snubber resistor.

V. EXPERIMENTAL RESULTS Figure 12 schematically depicts the configuration of the

developed circuit containing SIEMENS SP75 polysilicon solar module. The module with solar spectrum air mass (AM) 1.5 and the parameters at 25 Celsius degree temperature of the solar cell are listed as follows.

short circuit current:4.8A open circuit voltage:21.7V maximum power:75W voltage at maximum power:17.0V current at maximum power:4.4A

(16)

(17)

(18)

(19)

(20)

(21)

(22)

Figure 13 shows the circuit of a burning circuit panel of the dsPIC. The circuitry panel integrates RS-232, LCD module, isolation and amplification circuit. Figure 14 displays the flyback main circuit that this work adopts the rear stage of maximum power tracking. Therefore, the Hall Sensor and voltage sensing circuit are placed in the rear terminal of the circuitry. Because the signal is sent by the current and the voltage sensing circuit contains a certain noise level, the output signal of the current and voltage sensing circuit have a second-order filtering circuit to filter the noise, which are then inputted into the A/D terminal of the dsPIC for operations.

mLi mL

SWV

Fig. 12. Entire circuit configuration.

Fig. 13. dsPIC burning circuit

Fig. 14. Flyback main circuit

Fig. 15. Circuit diagram (connecting all circuits)

Figure 16 show the waveform of the voltage across the G-S

terminal of the MOSFET switch and waveform of capacitor current. Figure 17 displays the waveform of the voltage across the G-S terminal of the MOSFET and voltage waveform at the primary side of the transformer.

Vsw

ic

CH1:X-axis:25μs/div Y-axis:10V/div CH4:X-axis:25μs/div Y-axis:2V/div

Fig. 16. waveform of the voltage across the G-S terminal of the MOSFET switch and waveform of the capacitor current

CH1:X-axis:25μs/div Y-axis:20V/div CH3:X-axis:25μs/div Y-axis:20V/div

Fig. 17. waveform of the voltage across the G-S terminal of the MOSFET and voltage waveform at the primary side of the transformer

Figure 18 displays the waveform of the voltage across the

G-S terminal of the MOSFET and voltage waveform at the secondary side of the transformer. Figure 19 shows the measured curve of MPPT power.

CH1:X-axis:25μs/div Y-axis:20V/div CH2:X-axis:25μs/div Y-axis:20V/div

Fig. 18. waveform of the voltage across the G-S terminal of the MOSFET and voltage waveform at the secondary side of the transformer

Fig. 19. Measured curve of MPPT power.

VI. CONCLUSIONS This work presents a solar energy battery energy storage

system with maximum power point tracking, in which a microcontroller (dsPIC30F4011) is used to retrieve the voltage and current in the flyback circuit for increasing or decreasing the duty of the MOSFET switch to achieve maximum power output. Additionally, this work adopts the frequently used perturbation observation method with a simple mathematic formula in maximum power point tracking, despite the power losses incurred by perturbation around the maximum power point. However, a microchip with a less expensive and moderately difficult calculation capability can be selected to lower the costs of overall circuit, making it highly promising for industrial use.

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