COMPARISON OF HALF-BRIDGE, ZCS-QRC AND ZVS-MRC FOR OFF-LINE APPLICATIONS*

Milan M. Jovanovid, Richard Farrington, and F r e d C. Lee

Virginia Power Electronics Center The Bradley Department of Electrical Engineering Virginia Polytechnic Institute 8 State University

Blacksburg, Virginia 24061

ABSTRACT

Performance of the half-bridge (H8) zero-current-switched (ZCS) quasi-resonant (QR) and zero-voltage-switched (ZVS) multi- resonant (MR) converters are compared, with respect to their er%- ciency, input voltage range, semiconductor stresses, power density, and reliability. The efficiency of the HB ZVS-MRC at a given nominal input is shown to be highly dependent on the range of fhe input-voltage, and i t suffers when the converter has to be designed to cover a wide range. However, this is not the case for the HB ZCS-QRC. An experimental HB ZCS-QRC and H 8 ZVS-MRC, each with different input voltage ranges, were designed under the same constraints to facilitate their comparison.

1. INTRODUCTION

The zero-current-switching quasi-resonant technique and zero- voltage-switching quasi-resonant and multi-resonant techniques have been successfully implemented in the megahertz range for off-line power supplies [1]-[5], [l 11. Among the zero-current- switched (ZCS) quasi-resonant converters (QRCs), the half-bridge (HB) topology (Fig. I) operating inhalf-wave mode with secondary side resonance is preferred (61. Although it has more compo- nents than the single-ended topologies, such as the forward and flyback converters, it was shown to be more efficient and to have better core utilization. These features makes the HB topology more desirable for applications with high-input voltages and high-output power [6]. However, the conversion frequency of the half-wave HB ZCS-QRC at light load is usually several (5-10) times lower than the maximum conversion frequency (at low line and full load). The wide conversion frequency range penalizes the filter size and hampers the design of the control loop with high gain and wide bandwidth 161. The frequency range can be im- proved by operating the converter in full-wave mode [SI. How- ever, the full-wave converter is difficult to implement at high frequencies due to the slow recovery of the antiparallel diode [SI. [El.

One major limiting factor of the maximum conversion frequency of the ZCS-QRCs is the presence of the turn-on switching loss due to the discharge of the power MOSFET output capacltance [SI. Presently, the maximum conversion frequency attained in off-line ZCS-QRCs is around 2 MHz [2], [ill.

The zero-voltage-switching technique eliminates the turn-on switching loss completely by turning on the switch at zero voltage. As a result, the zero-voltage-switched (ZVS) converters are ca- -blP of operating at a higher conversion frequency (around 10

MHz) [4]. Recently, a 75 W ZVS HB QRC was designed to operate in the frequency range of 1 to 4.2 MHz with input voltage ranging from 250 V to 350 V [4]. The power stage achieved an efficiency of 83.5% at full load and 250 V input. However, the converter was only capable of operating, while maintaining zero-voltage switch- ing, in a very limited load range, from full load down to 70% of full load. Furthermore. undesirable parasitic oscillations between the resonant inductor and the capacitance of the rectifiers were ob- served. These oscillations further limit the converter’s load range and cause instabilities in the closed-loop system. To alleviate the above-mentioned difficulties, the multi-resonant (MR) technique was proposed [7]. It allows the rectifier‘s junction capacitance to resonate in a controlled fashion with the resonant inductor. Em- ploying this technique, a 75 W ZVS HB MRC for off-line applica- tions was implemented (Fig. 2) 141. While attaining a maximum efficiency of 81.7% at full load, the converter was capable of op- eration from full load down to no load and at an input voltage ranging from 250V to 350V. The conversion frequency varied from 1.5 to 8 MHz.

The objective of this paper is to assess the merits and limitations of the HB ZCS-QRC and the HB ZVS-MRC with respect to their efficiency, input-voltage range, stresses on the semiconductor components, power density and reliability.

2. REVIEW OF HALF-BRIDGE ZCS-QRC AND ZVS-MRC

2.1 Half-Bridge ZCS-QRC

The secondary-side resonant HB ZCS-QRC. shown in Fig. I. uti- lizes the leakage inductance of the transformer and capacitance C, to resonate the switch current, in order to achieve a lossless turn off. A detailed analysis of the converter’s operation is pre- sented in 121. Figures 2 and 3 show the key waveforms and the dc voltage-conversion-ratio characteristics of the ideal HB ZCS-QRC.

To maintain a constant output voltage under variable input and load, it is necessary to modulate the conversion frequency. The conversion frequency increases as the input voltage decreases andlor the load current increases. The frequency modulation scheme used here can be regarded as a constant on-time vari- able off-time control.

From Fig. 2 it can be seen that, to achieve zero-current switching. the amplitude of the resonant current (Vs/2NZ,) must be greater than the output current (lo), i.e.,

* This work was supported by Digital Equipment Corporation and Virginia Center for Innovative Technology.

445 C H Z ? l 9 - 3 / 8 9 / ~ - 0 4 4 5 $1.00 0 1989 IEEE

RL

Fig. 1 Half-bridge, half-wave zero-current-switched quasi- resonant converter (HB ZCS-QRC)

where V, is the input voltage, N is the turns ratio of the transfor- mer, /o is the output current, and Z,, = m is the character- istic impedance of the resonant tank.

From Eq. (1). the maximum characteristic impedance, which en- sures the zero-current switching for the entire input-voltage and load range, is determined at low line (Vyl") and full load (I&"""). Once the resonant frequency is selected, the resonant tank com- ponents are uniquely determined. The maximum conversion fre- quency is limited mainly by the turn-on switching loss due to the discharge of the MOSFET output capacitance. Generally, power MOSFET's with lower output capacitance are accompanied with a larger on-resistance. However, it is possible to select MOSFET switches so that the total loss (switching + conduction) is mini- mized [SI. It was found to be impractical to design a 100W HB ZCS-QRC with a conversion frequency greater than 2MHz [2]. (111.

Because of a relatively high operating frequency (2MHz) and the use of thick-film hybrid packaging technique, a power density greater than 3OWlcu.in. was achieved [Ill

M =

CO"

6

Fig. 3 DC vonage-conversion ratio of half-wave, HB ZCS-QRC as a function of conversion frequency for different nor- malized output currents (IoN = loZ,/(Vs/2N)).

I 'GS2 I

1 I I

iL I

4.2

[prim

Fig. 2 Key waveforms of hagwave HB ZCS-QRC

The main shortcoming of the HB ZCS-QRC is its relatively low minimum conversion frequency, which occurs at high line and llght load. The frequency range increases as the range of input- voltage and load increase, For example, for a converter designed to operate from full load to 10% of full load and with a 100% input voltage change, the minimum conversion frequency is around one tenth of the maximum conversion frequency. A relatively low minimum frequency severely constrains the optimal design of the output filter and the bandwidth of the closed-loop system [6].

2.2 Half-Brldge ZVS-MRC

The circuit diagram and the key waveforms of the HB ZVS-MRC are shown in Figs. 4 and 5. Unlike the HB ZCS-QRC, where the

446

COR1

COR?

Fig. 4 Half-brldge zero-voltage-switched muiti-resonant con- verter (HB ZVS-MRC)

'GSI 1

"GS21 II I I II I I I I1 7 4 ! I , I I , I I I I I I

I l l I I I I I I vCl 1 11 I I I I I I I

, , I I I l l I I

t

.t I , ! I I l l I I

vc2

"S

'prim

I 1 I I , I , I

t I I I I I l l I I I l l I I l l I I 1

-t

M =

.. . . .. is* I ;; ; II I I I

, I / ( I

t I l l I I I I I I

Fig. 5 Key waveforms of HB ZVS-MRC

resonance occurs only during a portion of the resonant cycle, the inductor L, of the HB ZVS-MRC continuously resonates with the output capacitances of the switches (C=C, = C,) and the capacitances across the rectifiers (CDR = C,,, = CDR2). As a re- sult, both the primary switches and the rectifiers are switched with no abrupt changes of the voltage across them. The turn-on

Fig. 6 DC voltage-converslon ratio of HB ZVS-MRC as a func- tion of conversion frequency: a) xc=5,

Normallzed output current I,, = 2Z,4,,/NVS and X, = 2CDR/N2C are parameters. X, represents ratlo of capacitance across rectmers reflected into prlmary c ~ ~ / ( N / 2 ) 2 and resonant capacitance of primary X .

of the primary switches and the turn-off of the rectifiers occur at zero voltage.

The dc voltage-conversion ratio as a function of the conversion frequency is shown in Fig. 6 [4]. The characteristics are plotted with two parameters: = 2Z,,/0/NV,, the normalized output cur- rent, and X, = 2C,,/(N2C) , the ratio of the capacitance across the rectifiers reflected to the primary (C,,KN/2), and the resonant capacitance of the primary (2C). From Figs. 6.(a) and 6.(b) it can

b) Xc=fO.

447

be seen that the load range of the converter improves as COR in- creases and, thus, X, increases. However, the peak current of the switches (/FE$ also increases, since it is governed by [9]:

Prlmery Switch Q1, Q2

Turns Ratio (N) # of Prlm. Turns (N1) # of Sec. Turns (N1,

K Core

e 5 cn

where the characteristic impedance is defined as Z, =

Therefore, to minimize the current stress and to maximize the conversion efficiency, it is desirable to select the smallest capacitance, CDR, necessary to maintain zero-voltage switching at minimum load and high line (worst-case condition). Generally, lower values of C,, are required as the characteristic impedance increases 141. As a result, the HB ZVS-MRC requires the use of a relatively large resonant inductance (usually > 5pH) and switches with extremely low output capacitance (around 100 pF or less). The detailed design guidelines are given in 141.

From Fig. 6, it can be seen that frequency modulation is required to maintain a constant output voltage. Since the resonance of the switch voltage takes place only when both switches are off (Fig. 5), the frequency modulation technique implemented in this case is constant off-time variable on-time control. The conversion fre- quency increases when the input voltage or the load current de- creases.

IRF730 IRF730 TDK LP 2318 ( H 7 4 TDK LP 22113 ( H 7 d

12 10 12T, Lltz 100142 l T , 2 mll Cu foll

30T, Lltz 175144

3T, 3 x Lltz 175146

3. EXPERIMENTAL RESULTS

The HB ZCS-QRC and HB ZVS-MRC, were designed for the input voltage range from 150 V to 350 V and a maximum output power of IOOW. The converters were designed using the design proce- dures described in [6] and 141, which maximize the efficiency and minimize semiconductor stresses of the corresponding convert- ers. Table I contains the component lists for the designs, and Table I1 summarizes the key performances of the two bread boards.

Measured efficiencies, as functions of the input voltage, are shown in Figs. 7 and 8 for full load and 50% of full load, respec- tively. At full load (Fig. 7), the maximum efficiency of the HB ZCS-QRC occurs at 250 V. On the other hand, the maximum effi- ciency of the HB ZVS-MRCs occurs at low line (150 V). As the input voltage increases, the efficiency of the HB ZVS-MRC de-

creases monotonically At high line, the efficiency of the HB ZCS-QRC is slightly higher (73.2%) than the efficiency of the HB ZVS-MRC (72.4%). However, at the nominal line (250 V), the effi- ciency of the HB ZCS-QRC is 5% higher. For lower load currents (Fig. 8), both converters are shown to have the maximum em- ciencies at low line. The efficiency of the HB ZCS-QRC is always higher than that of the HB NS-MRC.

In the following sections, the performances of the two breadboards are compared in a more comprehensive manner. A qualitative loss analysis of the two converters is performed, which underlines the key parameters that affect the efficiency profile of the two converters.

TABLE I t

Performances of Experimental Converters

I I Efficiency at 20 A & 300 V I 78 % I 73.4%-

r

4. PERFORMANCE COMPARISONS

4.1 Efficiency vs Input Voltage

As mentioned earlier, design of both the HB ZCS-QRC and HB ZVS-MRC is constrained by the input-voltage range. The input- voltage range, along with the maximum output power, dictates the selection of the major circuit components such as the semicon- ductor switches and diodes, the resonant tank, the power trans- former, and the output filter. In addition, the input-voltage range also dictates the conversion frequency range and, consequently, the crossover frequency and bandwidth of the converter. The ef- ficiency of the converter is also strongly related to the input volt- age. This is illustrated by performing a qualitative loss analysis of both converters.

TABLE I

Component Values for Experimental Half-Bridge Converters Designed for 150-350 Vdc Input

1 COMPONENT I t HBZCS-QRC I HBNS-MRC I

I . .- .. I

Resonant Rec. Cap ( C d

Filter lnd. (L$ Filter Cap (C$

28 nF (NPO ceram.) Rectifiers IR60CNQ045 IR60CNQ045

Input Cap (CJ 0.1 pF I400 V 0.1 pF I400 V

448

EFFICIENCY - X 4.1.1 Haki3rMge ZCS-QRC

To maintain a constant output voltage in the HB ZCS-QRC when the input voltage is increased, it is necessary to decrease the conversion frequency [6]. Therefore, the input voltage not only affects various circuit losses, which are voltage dependent, but also losses which are switching-frequency-dependent. Losses which are voltage-dependent are the turn-on switching loss (Po,), due to the discharge of the output capacitance of the switches, and the hysterisis loss ( f H ) 151, [SI:

84

82

00

78

76

74

72

70

HB ZCS-QRC (0.1-1.4 MHz)

.-_ - - - _ _ _ _ _ HB NS-MRC .-*..

(1.7-10 MHz)

r., = 5 v I I 1

150 200 250 300 350

INPUT VOLTAGE - V

FIg. 7 Emclencles at full power (5 V I 20 A) of experimental HB ZCS-QRCs and HB ZVS-MRCs as functions of input volt- age (150-350 V)

EFFICIENCY - %

a4

82

80

78

76

74

72

70

68

66

\ HB ZCS-QRC (0.1-1.4 MHz)

.. 8 .

HB ZVS-MRC"-.. (1.7-10 MHz) .-.

I I I

150 200 250 300 350

INPUT VOLTAGE - V

Fig. 8 Emclencles at 50% of fu// power (5 V I 10 A) of exper- lmental HB ZCS-QRCs and HB ZYS-MRCs as functlons of Input voltage (150-350 V)

Turn-on switching loss = Po, oc V~f,,, (3)

Hysterisis loss = PH oc S,f,,, = Vsfonfcon (4)

where V, is the input voltage, f,,,= 2 4 is the conversion fre- quency, f, is the switching frequency, B, is the maximum flux density in the transformer, and to, is the on-time.

Although f,,, decreases as V, increases, f,,, decreases at a slower rate than Vs increases and, consequently, Po, and PH in- crease as the input voltage increases [2], [6].

Besides the hysterisis loss, other loss factors in the transformer are strongly frequency dependent. The eddy-current loss (PEC) and the residual loss ( f R ) occur in the core, whereas the losses due to skin-effect (PSE) and proximity-effect (PPE) occur in the copper [ e ] :

Eddy current loss = PEc oc ton ( 5 )

Residual loss = P, oc fcon (6)

Skin-effect loss = P,, oc con (7)

Proximity loss = PPE OC con (8)

where n is in the range from 0.5 to 2, depending on the operating frequency and the diameter of the conductor IS].

Losses given in Eq. (5)-(8) always decrease as the input voltage increases.

For a converter with a low-output voltage, the major losses are the conduction losses of the rectifiers and the freewheeling diode (around 50% of the total loss for a 5 V output). As V, increases, f,,, decreases and the conduction time of the rectifiers de- creases. As a result, the conduction loss of the rectifiers de- creases. However, as Vs increases, the average current of the freewheeling diode increases. Therefore, the total conduction losses of the rectifiers and freewheeling diode are almost inde- pendent of the input voltage variation.

It should be noted that, as the input voltage increases, the RMS current in the primary of the transformer decreases slightly (Fig. 7 in 151). Therefore, the conduction loss of the primary switches also decreases.

Since P,,and f" increase and PE,, P,. fsE, and P,, decrease as the input voltage increases, the converter efficiency can be opti- mized at a particular chosen input voltage. This can be achieved by a proper selection of the primary switches, and the design of the power transformer and the resonant tank.

449

4.1.2 Half-Bridge ZVS-MffC

Contrary to the HB ZCS-QRC, to maintain a constant output volt- age, the conversion frequency of the HB ZVS-MRC must be in- creased as the input voltage increases. Therefore, all the voltage and frequency dependent losses (foff, PE,, PR, fsE, and PpE) in- crease as the input voltage increases. The hysterisis loss as given by Eq. (2) is a function of V, , f,,,. and to , . Since the MRC is controlled with a variable frequency and constant off-time con- trol, when f,,, decreases, ton increases. As a result, P, is es- sentially a function of V,.

Zero-voltage switching technique eliminates the turn-on switching loss. However, as the input voltage increases, the conduction loss increases because the RMS current through the power switches increases (91.

The transformer secondary currents also increase as the input voltage increases. As a result, for a wide input voltage range, the copper loss in the secondary windings increases significantly at high line.

The turn-off switching loss is proportional to [IO]:

Turn-off switching loss = Poff oc fconl$ (9)

6

2 f E c n .

EFFICIENCY - %

Core TDK LP 2318 (H,& TDK LP 32113 (H7&

Turns Ratio (N) 12 16

# of Prim. Turns (N,) # of Sec. Turns (Nd

12T, Litz 100142 lT , 2 mil Cu foll

Leakage Ind. 30 nH @! secondary 25 pH @! primary

16T, Litz 60140

lT , Litz 135135

where /&,is the switch current at the moment of turn off. Thus, it increases when the input voltage increases.

Since all the major losses in the HB ZVS-MRC increases as the input voltage increases, the efficiency of the converter is always maximum at low line and monotonically decreases as the line voltage increases. Thus, if the converter is designed for a wide input-voltage range, the efficiency at nominal input voltage can be substantially lower than the maximum efficiency. On the other hand, for the MRC designed for a relatively narrow input voltage range, the efficiency can surpass that of its QRC counterpart.

4.1.3 Efficiency for a Narrow Input-Voltage Range

The performance of the HB ZVS-MRC is expected to improve considerably for a narrower input-voltage range. To verify this, a HB ZCS-QRC and HB ZVS-MRC were designed for an input volt- age from 250 to 350 V, and a 5V output with for maximum power of 1OOW. The component lists for these designs are given in Table 111.

Resonant Ind. (L,J

Resonant Cap (C,J Resonant Rec. Car, ICnm)

84

02

80

70

76

74

72

70

leakage inductance of the transformer

180 nF (polypropy1ene)l C t l p 30 pF I 20 nF (NPO ceram.)

HB ZVS-MRC (1.5-8 MHz)

".._._ '..__ ... '.. .. __,

Rectifiers

Filter Cap I C J fllter Ind. (L$

-7- - \

IR60CNQ045 IR60CNQ045 3 PH 1 rH

100 UF (chip tant.) 25 UF (Z5U chip)

HB ZCS-QRC (0.1-1.4 MHz)

(1.7-10 MHz)

v, = 5 v

150 200 250 ' 300 350

INPUT VOLTAGE - V

FIg. 9 E/riclencies at full power (5 V I 20 A) of experimental HB ZCS-QRCs and HB ZVS-MRCs as functlons of input volt- age (250-350 V)

Figures 9 through 10 show the measured efficiencies superim- posed on the measured efficiencies of previous designs with wide input voltage ranges (150V-35OV). For a narrower voltage range (250-350 V). the efficiency of the HB ZVS-MRC is shown to be higher than that of the HB ZCS-QRCs for the entire input-voltage and load ranges. Although the maximum efficiency of the HB ZVS-MRC still occurs at low line, it falls off at a slower rate, compared to the previous design. This can be attributed primarily to the higher turns ratio of the transformer (16 vs IO. Table I). higher characteristic impedance of the resonant tank (645 n vs 200 n). and lower resonant capacitance across the rectifiers (20

TABLE 111

Component Values for Experlmental Half-Brldge Converters Designed for 250.350 Vdc Input

C 0 M P 0 NE NT 11 HBZCS-QRC I HB ZVS-MRC

Prlmarv Switch Q.. Q, 11 IRF730 I IRF710

lnout Car, IC,) 11 0.1 uF 1400 V I 0.1 uF I400 V

450

EFFICIENCY - %

84 1

82 i- HB ZVS-MRC (1 .58 MHz)

80

78

76

74

70 72 I 68 1 v, = 5 v

HB ZVS-MRC'., (1.7-10 MHz) **.

66 I I I I 150 200 250 300 350

INPUT VOLTAGE - V

Fig. 10 Emciencles at 50% of full power (5 v I I O A) of exper- imental HB ZCS-QRCs and HB ZVS-MRCs as funct/ons of input voltage (250-350 V )

nF vs 28 nF). Due to the higher turns ratio and higher character- istic impedance, the conduction loss in the primary switches and the copper loss in the primary winding are reduced. In addition, the use of smaller resonant capacitance across the rectifiers re- sults in a reduced peak (Eq. (2)) and reduced RMS currents in the transformer.

4.2 Maximum Conversion Frequency vs Maximum output Power

From the condition for zero-current switching, Eq. (I), it follows that if the maximum output current is increased, the characteristic impedance must be reduced, either by reducing L, or increasing C,. To avoid decreasing the resonant frequency ( W O = I/=), it is necessary to simultaneously increase C, and reduce L,. Since L, is the leakage inductance of the trans- former, its minimum value is constrained by the transformer de- sign. Generally, the leakage inductance increases as the size of the transformer increases to accommodate higher power. As a result, the maximum attainable conversion frequency for the ZCS-QRC is limited for a given load current

Figure I 1 shows the maximum resonant frequency (fa) of the HE ZCS-QRC as a function of the output current for different resonant (leakage) inductance. It should be noted that the practical limita- tion of the maximum conversion frequency is the loss in the magnetics, rather than the leakage inductance.

The I-iB ZVS-MRC requires a much larger characteristic Impedance than the HB ZCS-QRC. To achieve a high character- istic impedance and high resonant frequency simultaneously. it is necessary to have a large resonant inductance and small reso- nant capacitance. Therefore, the leakage inductance of the transformer does not impose any limitation on the maximum out- put power of the converter. However, the minimum value of the resonant capacitance is limited by the output capacitance of MOSFET switches. and, consequently, the maximum conversion

0 N

u, -

0 -

LD

0

10 [AI

Fig. 11 Maxlmum conversion kequency vs maximum load cur- rent for ditlerent leakage Inductances of HB ZCS-QRC transformer

frequency of the HB ZVS-MRC is also limited. In spite of this limitation, the HB ZVS-MRC is capable of delivering much higher power than the HB ZCS-QRC operating in the same frequency range.

4.3 Stresses on the Semiconductor Components

Due to the half-bridge topology, the voltage stresses on the pri- mary switches for both converters are identical. The maximum voltage across the nonconducting switch is equal to the maximum input voltage Vyax.

The maximum reverse voltage across the rectifiers in the HB ZCS-QRC is slightly lower than that of the HB ZVS-MRC. For the HB ZCS-QRC it is (61

Due to the resonance of the rectifier voltage, the maximum re- verse voltage in the HB ZVS-MRC is

In both converters the primary currents are quasi-sinusoidal. The maximum, primary current in the HB ZVS-MRC is given in Eq. (2) for Vyax, whereas the corresponding current in the HE ZCS-QRC is [6]

Figure 12 shows the maximum, peak primary current for the ex- perimental converters as a function of input voltage. As can be seen, the HB ZVS-MRC has lower current stress.

The maximum peak current through the rectifiers in the HB ZCS-QRC is N times higher than the maximum peak primary cur- rent IS). The same conclusion is approximately valid for the HB ZVS-MRC.

4.4 Power Density

451

vs IVI

Fig. 12 Current stress of primary switches in experimental con- verters

The only difference between the HB ZCS-QRC and HB ZVS-MRC power stages is the design of the resonant tank components. The HB ZCS-QRC uses a resonant capacitor on the secondary side of the transformer to resonate with the leakage inductance of the transformer, while the HB ZVS-MRC employes the output capacitors of the primary switches and the resonant capacitors across the rectifiers. A relatively large resonant inductance in the HB ZVS-MRC can be realized either by the leakage inductance of the transformer or an external inductor or a combination of both. Regardless of how the resonant inductance is realized, the size of the magnetics in the HB ZVS-MRC is larger than that in the HB ZCS-QRC. Since the size of the other components does not differ significantly. the power density of the HB ZCS-QRC is somewhat higher. This is always the case at lower output power (around 100 W; 20 A / 5 V). A recently fabricated I00 W HB ZCS-QRC attains a power density (including heatsink and control circuit) of 35 W/cu.in [ I l l . The projected density of the 100 W HB ZVS-MRC being presently hybridized is approximately 30 W/cu.in. As discussed earlier, the maximum output power of the HB ZCS-QRC can not be increased without lowering the frequency and, therefore, increasing the size of the transformer. On the other hand, the HB ZVS-MRC is able to operate at higher output power and higher frequencies. Thus, at higher output power, it is anticipated that the power density of the HB ZVS-MRC can be higher than that of the HB ZCS-QRC.

4.5 Reliability

Due to the relatively large resonant inductance in series with the primary switches, the HB ZVS-MRC is less prone to failure under overload and short-circuit conditions. By simply limiting the min- imum operating frequency, the peak current through the primary switches can be limited. For example, if the minimum frequencies of the experimental HB ZVS-MRCs (Tables I and 111) are limited to 1.5 MHz, the calculated peak primary currents when the outputs are shorted are 10 A (for the 150-350 V design) and 2.5 A (for the 250-350 V design), respectively. These currents are much lower than the rated peak currents of the employed switches (22 A for IRF730 and 6 A for IRF'IIO).

In the HB ZCS-QRC, the overloadlshort circuit protection cannot be implemented easily because of a much smaller inductance in series with the switches. Furthermore, during a short circuit at

the output. the reflected inductance into the primary is only one half of the reflected inductance during the resonant stage, due to the shorted secondaries. Even if a maximum frequency is im- posed, it is usually not enough to effectively limit the primary current and protect the power switches. For example, if the maximum frequencies of the experimental HB ZCS-QRCs are limited to 2 MHz. the peak primary current for the shorted outputs is around 22 A.

5. CONCLUSIONS

Performance comparisons of the half-bridge (HB), half-wave zero-current-switched (ZCS) quasi-resonant converter (QRC) and zero-voltage-switched (ZVS) multi-resonant converters (MRC) were presented. The two converters were compared with respect to their efficiency, input voltage range, semiconductor component stresses, power density and reliability.

A qualitative loss analysis revealed that the efficiency of the HB ZVS-MRC is always maximum at low line. On the other hand, the efficiency of the HB ZCS-QRC can be maximized at any desired input voltage. As a result, the efficiency of the HB ZVS-MRC can be significantly lower at a nominal operating voltage when the input-voltage range is wide. The maximum efficiency of the HB ZCS-QRC is almost independent of the input-voltage range. These findings are verified experimentally by measuring efficien- cies of 100 W, off-line HB ZCS-QRC and ZVS-MRC designed for different input-voltage ranges.

Trade-off analysis was performed between maximum outpuf power (load) and maximum conversion frequency due to the lim- itations of the resonant tank components (leakage inductance of the transformer and output capacitance of the switches). It was shown that the HB ZVS-MRC is capable of delivering a higher power and is capable of operating at high frequencies than the HB ZCS-QRC. As a result, the power density of the HB ZVS-MRC is expected to surpass that of the HB ZCS-QRC as the output power is increased. At lower output power (around 100 W), the power density of the HB ZCS-QRC is slightly higher due to a rel- atively large resonant inductance employed in the HB ZVS-MRC.

The voltage stresses on the switches and rectifiers are similar for both converters. The current stresses are lower in the HB ZCS-QRC. The stresses are dependent on the input voltage and they increase as the input-voltage range increases.

The HB ZVS-MRC is shown to be capable of effectively limiting the maximum peak current of the switches under overload and short-circuit conditions. In the HB ZCS-QRC, this self-protecting feature can not be implemented as easily.

The HB ZVS-MRC is most suitable for off-line applications where the input voltage range Is relatively narrow. With a relatively nar- row input voltage range, all the positive attributes associated with the multi-resonant operation, such as high efficiency. reliability, high power density and multiple outputs can be utilized to their fullest extent.

REFERENCES

[I] P. Vinciarelli, "Forward Converter Switching at Zero Current," US. Patent, 4,415,595, 1983.

"Hybridized Off-Line 2-MHz Zero-Current-Switched Quasi- Resonant Converter," IEEE Transactions on Power Electronics, vol. 4, no. I. January 1989.

[2] D.C. Hopkins, M.M. Jovanovid, F.C. Lee, F.W. Stephenson,

452

[3] A. Hayman, "Low-Profile High-Frequency Off-Line Quasi-Resonant Con- verter." IEEE Applied Power Electronics Conference Proceedings, pp.157-165, 1987.

[4] M.M. Jovanovid, W.A. Tabisz. F.C. Lee, "Zero-Voltage-Switching Technique in High-Frequency Off- Line Converters," IEEE Applied Power Electronics Conference Proceedings, pp.23-32, 1988.

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[Ill D.C. Hopkins. M.M. Jovanovid. F.C. Lee, F.W. Stephenson, M. Hayes, "Design and Hybridization of a 100 W, 300 V ZCS-QRC," to be presented at the High Frequency Power Conversion Conference (HFPC'89), Naples, FL.. May 15-18, 1989.

[9] M.M. Jovanovid,

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