High Throughput Millimeter-Wave MIMO Beamforming System for Short Range

Communication

Eran Pisek, Shadi Abu-Surra, Jordan Mott, Thomas Henige, Rachit Sharma Samsung Research America – Dallas, Samsung Electronics, Dallas, TX, USA

{episek, sasurra, jmott, thenige, r3.sharma}@sta.samsung.com

Abstract — The interest in short range communication has significantly increased in the last decade with the introduction of different wireless connectivity standards and their evolution such as IEEE802.11n/ac and IEEE802.11ad. New demands for higher resolution multimedia applications such as UHD push for higher data rates to exceed 30Gbps. This high data rate demand requires short range standards to adopt a new approach to accommodate these rates. Millimeter wave enable high bandwidth usage which implies higher capacity to accommodate 10-100Gbps data rates. However, the high data rate pose big challenges to both the system algorithm and architecture. In this paper, we propose a novel FPGA-based high-throughput beamforming MIMO system for millimeter wave short range communication. We present our first realization of a complete millimeter wave short range communication system, which includes RF frontend, ADC/DAC, beamforming control, synchronization, channel estimation, MIMO precoding/detection, and channel encoder/decoder. The design works at millimeter carrier frequency with 500MHz bandwidth and 2-channel MIMO Beamforming. We show real-time application test results as captured in the millimeter wave system.

Index Terms—millimeter wave communication, beamforming, MIMO, LDPC, baseband, OFDM, AGC, FPGA.

I. INTRODUCTION Short range wireless communication has been successfully

adopted in a wide range of communication standards. Most of them are applied for home computing and entertainment where multimedia applications are mainly used. Recent advances of the multimedia applications suggest higher resolutions such as Ultra High Definition (UHD) that require higher data rates. Many of these high throughput applications are currently migrating to wireless devices and mobile devices, which drive the demand for high throughput wireless link between them and with a very high resolution screen such as the UHD screen.

Millimeter wave band, 30GHz – 300GHz, allows high frequency bandwidths (in GHz range) that can enable 10s of Gbps of capacity. This capacity can be used to wirelessly transfer compressed and uncompressed UHD resolutions of video between devices and screens. In addition, fast downloads, cloud computing, and internet browsing can be easily achieved using these high rates. Due to the millimeter waves unique propagation characteristics [1][2][3] of high propagation loss, and high reflection from objects such as buildings and walls,

they are a perfect fit for indoor wireless communication. In addition, due to their small wavelengths between 1 mm to 10 mm, the form factor of the antenna system can be very small compared to conventional IEEE802.11n/ac [5] 5GHz band antennas. With small antenna form factors the directivity of the Tx and Rx beams can be controlled by using smart antenna arrays enabling high-gain antennas. For indoor communication where the devices assume stationary to low mobility behavior, directivity of the transmitter antenna and the receiver antenna is crucial to achieve maximum power efficiency. Multiple standards were already formed to support Indoor/In-room wireless communication and specifically the 60GHz band with up to 10 meters distance and 7 Gbps data rate [4], such as IEEE 802.15.3c [6], IEEE 802.11ad [7], and others. However, in these standards the spectral efficiency is low compared to others operating in lower frequency band (e.g., IEEE802.11ac [5]). Increasing the data rates to support higher rates such as 30 Gbps can be achieved in multiple ways such as increasing the modulation scheme (above 64 QAM), increasing bandwidth (above 2GHz), or MIMO schemes. Note, each one of these options and definitely any combination of the three pose a big challenge to the Antenna/RF and baseband systems in the transmitter side as well as in the receiver side. In this work we present a new revolutionary wireless communication system that enables high bandwidth millimeter wave together with MIMO and beamforming.

The remainder of this paper is organized as follows. In Section II we give a quick overview of the system, in Section III we describe in detail the MIMO beamforming transceiver, in Section IV real-time over the air test results are shown, and we conclude the paper in Section V. Included are details of the synchronization algorithm, beamforming control, automatic gain control (AGC), channel estimation, MIMO detection, and channel decoder.

Figure 1. AP and UE Slices

978-1-4799-2355-7/14/$31.00 ©2014IEEE

The 11th Annual IEEE CCNC- Wireless Communications Track

537

Figure 2. A novel millimeter wave MIMO beamforming OFDM transceiver (2-user example)

II. SYSTEM OVERVIEW In this section we explain the millimeter wave MIMO

system. This system uses transmit and receive beamforming to overcome the unfavorable propagation characteristics of millimeter waves, achieve higher energy efficiency (lower energy per bit), and provide higher spatial channel order. The access point (AP) serves several sub-sectors (called slices). Each slice corresponds to a beam and is outlined by a synchronization channel (SCH). Similarly, the user equipment (UE) also has multiple slices (see Figure 1). In this millimeter wave MIMO system, the AP-to-UE-link and the UE-to-AP-link are time multiplexed, and each direction carries data and control information to such as beam training results and channel parameters for optimizing the link throughput.

The system can be configured in different ways based on the required data rate and the number of frequency bands. The AP, in the MIMO frequency-division multiple access (FDMA) mode, can support simultaneous AP-to-UE-links, each UE on a different band. Multiple data streams can be sent to the user over spatially separated beams. Alternatively, in the Single-Band Space-Division Multiple Access (SDMA) Mode, the AP can serve multiple users on the same band by using simultaneous beams separated in space. In a third mode, Multi-Band MIMO Mode with Carrier Aggregation, the AP serves multiple UEs over aggregated bands. Here we assume that the aggregated bands can be supported by the same antenna array. The AP has multiple antenna arrays with MIMO precoding across the arrays.

III. MIMO BEAMFORMING SYSTEM BLOCK DIAGRAM The System block diagram is shown in Figure 2. The AP

has multiple (two) antenna arrays, each consisting of several (two) sub-arrays, and each sub-array is driven by one digital chain. The elements of the same antenna array are closely spaced (e.g., half wave length apart), while the elements of the different antenna arrays are several wave lengths apart. The antenna elements are connected to digital phase shifters, which are controlled by the beamforming processor in the baseband to form different radio frequency beams. The UE is similar to the

AP, but usually has fewer antenna arrays and antenna elements. The transceiver’s baseband (BB) includes LDPC encoder/decoder, M-QAM modulator/demodulator, MIMO precoder/detector, Sync, AGC, and IFFT/FFT. Novel algorithms and optimized architectures were developed to enable high data-rate transmission.

A. System Structure and Numerology

We will explain first the structure of the baseband system. The system is Orthogonal Frequency Division Multiplexing (OFDM) based and the numerology is described in Table I.

TABLE I. mmWave OFDM system numerology.

mmWave OFDM Configuration

System Bandwidth (MHz) 500

Sampling rate (MHz) 552.96

Subcarrier spacing (KHz) 270

OFDM symbol length (FFT size) 2048

No. of useful subcarriers 1728

Total Guard band (MHz) 86

OFDM symbol duration (us) 3.70

CP length 512

CP duration (us) 0.93

Slot duration (us) 125

Number of OFDM symbols per slot 27

Subframe duration (ms) 1

Number of slots per subframe 8

Frame duration (ms) 10

Number of subframes per frame 10

Superframe duration (ms) 40

Number of frames per superframe 4

The system is multi-band with 500MHz bandwidth per band.

Each band is divided into 2048 subcarriers with 270KHz of subcarrier spacing and total of 86 MHz guard band between

The 11th Annual IEEE CCNC- Wireless Communications Track

538

bands to support simple analog image rejection filters between bands. Therefore we assume in our system only 1728 useful subcarriers. The baseband system is sampled at 552.96 MHz and the Cyclic Prefix (CP) is 0.25 of the symbol (0.93 us) duration. Shorter CPs can be applied depending on the Inter-Symbol Interference (ISI) of the channel. The symbols are grouped into slots (125 us) where there are 27 symbols per slot. The slots are grouped into subframes (1 ms). There are 8 slots per subframe where one slot is synchronization slot and one is reserved for Reverse Link (Uplink). The rest of the 6 slots per subframe are used for Forward Link (Downlink). The subcarriers are centered around the carrier frequency ((-864)-to-(-1) and 1-to-864)×270KHz). The system described below is an enhancement to the system described in [8] enabling Time Division Duplex (TDD) MIMO beamforming for short range communication.

B. Beam Selection and Synchronization Channel (SCH)

The Synchronization Channel (SCH) is responsible for slice identification, beam selection, timing, and frequency synchronization. There are two types of SCH, one for Forward Link (DSCH) and one for Reverse Link (USCH).

As described in [8], the Forward Link Synchronization Channel is sent by the AP over the first slot of the subframe. In our system the DSCH occupies only 256 subcarriers out of the 2048 that are centered around the carrier frequency. As explained above, the DSCH is primarily used for timing synchronization of the UEs in Forward Link so each UE will sync its received subframes, slots and symbols based on the detected DSCH. The DSCH is also used for Slice Identification (Slice-ID); a unique Zadoff-Chu (ZC) Slice-ID sequence is used for different AP slice. The DSCH sends each AP Slice-ID sequence on different OFDM symbol in the Sync slot occupying only the even subcarriers. For example, as described in Figure 3, with four AP slices, DSCH spans on four (consecutive) OFDM symbols with Z1, Z2, Z3, and Z4, ZC Slice-ID sequences with corresponding transmission through four slices, respectively. In order to guarantee the fastest synchronization by the UE at any point of time, similar approach as the cyclic prefix is proposed, the first symbol in the DSCH is copied and added to the end of DSCH. By doing this, AP transmits DSCH with 5 OFDM symbols as Z1, Z2, Z3, Z4, and Z1 for the first subframe. In the following subframes, the AP transmits the DSCH with Z2, Z3, Z4, Z1, and Z2, and so on.

In each UE, after detecting one or more DSCH symbols, the AP Slice-ID is determined by correlating the received DSCH symbol with all the reference DSCH symbols of all the AP Slice-IDs (known by the UEs) and finding out the maximum one.

The same method is used in the reverse link (Uplink) from the UE to the AP using the USCH channel in the Uplink slot period (Figure 3). However, there may be multiple UEs that send their sync sequences, channel information and data to the AP and they all use the same Uplink slot. Therefore, the Uplink slot is divided into multiple Time Division Multiplexing) TDM sub-slots. Each UE in the system is assigned with either a single or multiple sub-slots to transmit its USCH sequence, channel information and data to the AP. The division between the UEs in the Uplink slot can be also done in FDM within a symbol (divide the subcarriers between the UEs) or any combination between the two multiplexing

methods. In the following section we will continue and describe the beam pairing procedure between the AP and UE.

Figure 3. Synchronization channel between AP and UE.

C. Beam Pairing

The UE receives the DSCH Slice-ID sequences from different AP slices in the Sync slot period through a certain Rx beam (see Figure 3). Once the UE detects a DSCH Slice-ID sequence from one or more AP slices, it compare it to the previous detected IDs within the same Rx beam period (1 ms) and selects the slice with highest correlation value as described in Figure 3. Then the UE can change its own Rx beam and repeat the process described above for the new Rx beam. Each Uplink slot the UE can send back the AP its best received AP Slice-ID in its USCH assigned location in the Uplink slot period. The UE compare each subframe the best AP Slice-ID energy with the current AP Slice-ID energy and can decide whether to ask for AP slice change or keep the previous AP slice. The AP can use the corresponding slice to send data to this particular UE. If the AP does not detect any Slice-ID in the USCH, then it assumes that this particular UE does not require any change of slice. However, the AP can decide to keep the previous slice and not change the slice as instructed by the UE until a certain number of subframes are passed and the UE does not request to change Slice-ID within this time. The same is done for all the UEs; each can send its best AP Slice-ID back to the AP in its location in the Uplink slot. A multi-stage beam pairing algorithm is proposed. Once the best AP Slice-ID is determined by both the AP and UE, a second set of narrower AP beams within the best slice can be used with same or different Slice/Beam-IDs respectively, instead of the slices and Slice-IDs used on the first stage. The quality of the link can be significantly improved in a second stage by scanning a set of narrower beams within that selected slice in similar procedure and so on for next stages. Then the data is transmitted on the paired narrow beams.

Although it is straightforward for both transmitter and receiver to scan all combinations of transmission and receiving beams to find out the strongest beam direction, this direct scanning is a very time consuming and not energy efficient. For example, in each round, the transmitter fixes its beam at one transmission direction, and the receiver scans all receiving directions. This whole procedure in total uses SCH OFDM symbols to scan all combinations, where q is the number of

The 11th Annual IEEE CCNC- Wireless Communications Track

539

narrow beams in each sector (assuming same number of antennas per sector in both Tx and Rx sides).

The multi-stage beam pairing approach, proposed in this paper, tremendously reduces the number of required SCH OFDM symbols to complete the beam pairing, which is much suitable for our short range communication. The pairing approach is based on successive approximation method. In the first stage, the receiver and transmitter use a wide beam, or slice, each covering narrower beams. The receiver and transmitter can detect the strongest beam directions for this wide beam in OFDM symbols. Then the receiver and transmitter switch to narrower beams to scan inside that strongest wider beam. This takes another OFDM symbols to find out the strongest beam direction for the narrower beam. And so on for where , till beams left. In total, this algorithm uses

OFDM symbols. The total OFDM symbols required for beam pairing is

much smaller than the straightforward approach of OFDM symbols described above. It is also smaller than the 2-stage approach [8] with 2q OFDM symbols for , and equal for . This algorithm can be generalized to any number of antennas in either side. In the case where the Tx has q antennas per sector and Rx has m antennas per sector, the total number of OFDM symbols required is:

Other sub-optimal methods can be applied as well. In addition, in case where the energy difference between two AP/UE slices/beams pairs is below a certain threshold, the decision can be postponed to later stages by increasing the number of OFDM symbols involved in the beam pairing, or to keep the decision based on current beam pairing. For example, if the current beam pairing is part of one of the candidate group of beams with smaller difference than the threshold, then it will be also the surviving group.

After finding the best beam pair, the scanning still keeps running each subframe to track any changes due to movements and obstructions. Once a better quality pair is found, the AP and UE will update the recorded pair for data transmission. In the next section we describe how the beams are formed and the gain is controlled.

D. Analog Beamforming & AGC Control

In order to allow both the AP and the UE to form their beams with different angles, the antenna consists of an array of elements. Each antenna element is fed by different phase shifts of the RF modulated signal. Each antenna element can also have its own Power Amplifier (PA). In our system there is only one PA per antenna array. The phase shifts for the antenna arrays and all other RF configuration such as Variable Gain Amplifiers (VGAs), T/R switch, PLL control, PA gain, LNA gain and other controls are digitally configured and provided by the Baseband platform. In the Baseband platform there are predefined configurations for the different beams and beam widths. In addition, for proper beam selection, it is important to set the signal power at the receiver input of the analog-to-digital converter (ADC) for each beam pair to a fixed value which will use the maximum ADC dynamic range. This is achieved by adjusting the VGAs in the UE Rx RF unit. The

VGA adjustment is done by the AGC algorithm running at the UE baseband. AGC adjusts the gain of the RF VGAs according to the received power, E, computed in the synchronization module.

The AGC control mechanism that varies the VGA gain based on the received signal energy can also be a good indication for the received beam signal reliability. The higher the gain required from the VGA by the AGC control mechanism the smaller the received signal energy which indicates the received beam signal to noise ratio is low which leads to low signal integrity. Hence, high energy received beams which require low VGA gain indicate strong beam energy. Figure 4 describes the AGC gain control decision.

Figure 4. AGC Scheme

For OFDM systems as described in Figure 4, we assume 20 dB head room for Peak-to-Average Power Ratio (PAPR), fading and beam forming variations. Signal target is set at 32.5 dB above ADC noise floor.

At system initialization (or loss of sync), the front end attenuators are set to their minimum value. This setting is advantageous to the capture of weak signals and provides consistent capture and initialization for the expected worst case conditions. The DSCH must be reliably received in order to maintain timing at the receiver.

Once the DSCH has been captured, power estimates of the pilot symbol are used by the AGC algorithm to determine the required adjustment of the front end attenuators. Power estimates greater than 37.5 dBm above the noise floor increase the front end attenuators. Power estimates below 32.5 dBm above the noise floor decrease the front end attenuators. Power estimates between 32.5 and 37.5 dBm do not change the attenuators.

Loss of the DSCH can occur and if the duration of the loss is greater than 1 sec, the AGC algorithm resets the front end attenuators to their minimum value to acquire the sync signal again.

E. Channel estimation

After pairing the transmitter and the receiver beams, setting the AGC levels, syncing the time-domain received signal and transforming it to frequency-domain through FFT, the MIMO channel can be estimated through the pilot bits sent in selected symbols for both the transmitters.

The 11th Annual IEEE CCNC- Wireless Communications Track

540

In the transmitter, pilots from each transmission chain are first interleaved in the frequency domain, and then are inserted into the OFDM every 6 subcarriers. At the receiver [8], with the received pilot, the corresponding channel coefficient

in the channel matrix for subcarrier j is estimated by using least square estimator , where

is the reference pilot for subcarrier j from transmitter

chain n to the receiver chain m; , is the received pilot. The estimated channel coefficients are then linearly interpolated to compute the channel coefficients of the subcarriers without pilot for each channel (frequency-domain estimation). In addition, linear time-domain estimation is performed for matching sub-carriers in neighboring symbols ( symbols).

Besides the channel coefficients, complex noise variance is also estimated in the channel estimator. This is computed

by calculating the signal energy on the guard band (in place of non-existent sub-carrier).

F. MIMO detection

With the estimated channel matrix and noise variance , the a posteriori probability (APP) detector calculates log-likelihood ratio (LLR) L of bit k as described in [8]. By performing maxLogAPP approximation, LLR computation is reduced to

, 1

, 1

1

1

1min { ( ) ( )}

21

min { ( ) ( )}2

k

k

Tk

x X

T

x X

L−

+

−

∈

−

∈

≈ − −

− − −

y Hx C y Hx

y Hx C y Hx

(2)

where means the sets of , and means the sets of ; C is the covariance matrix.

By assuming the received signals are independent from each other, in our 2x2 MIMO receiver, we further reduce the complexity of the first term

, 1

1min ( ) ( )k

T

x X −

−

∈− −y Hx C y Hx to

, 1

* * * *1 11 2 21 1 1 12 2 22 22 2 2 2

1 2 1 2

* * *1 2 11 12 21 222 2

1 2

1 1 1 1min {-2 (( ) )-2 (( ) )

1 1 2 ( ( ))}

kx Xy h y h x y h y h x

x x h h h h

σ σ σ σ

σ σ

−∈ℜ + ℜ +

+ ℜ +

(3)

where is the real part of a complex number; indicates the conjugate of ; is the element in row i and column j of the channel matrix; is noise variance for receiver chain i. The same simplification is also performed to the other term:

, 1

1min ( ) ( )k

T

x X +

−

∈− −y Hx C y Hx

(4)

The MIMO detection block is divided into 3 pipelined sub-

blocks: the ML computation shared for the two channels, the Norm calculation for all possible combinations (16 combinations for 2x2 MIMO QPSK) and the Log-Likelihood Ratio (LLR) computation that finds the minimum distance of all combinations and output LLR info to the channel decoder.

G. Channel Coding

Due to the fading characteristics of the MIMO channel, the system requires also a strong channel coding in order to correct any errors introduced by the channel. Like other connectivity standards such as IEEE802.11n/ac/ad, the proposed MIMO beamforming system incorporates Low Density Parity Check (LDPC) coding. The transmitted raw data is first LDPC encoded in the transmitter. The LDPC code is a block code, and similar to IEE802.11ad, we define the block size of the LDPC to be an exact division of the number of useful subcarriers. From Table I, the number of useful subcarriers per symbol is 1728. Hence, we defined the basic LDPC block size to be 1728/4 = 432 bits and we designed two lifting factor hierarchies that construct the 1728-bit H-Matrix as in [9]. The LDPC code is based on a 432-bit block with Z=27 lifting factor. A higher level Zp=4 lifting factor interleaves four blocks of 432-bits. Figure 5 shows the LDPC R=1/2 H-Matrix based on these two lifting factors. The R=1/2 code is four-layer decodable with four independent row pairs which means that the LDPC decoder processing can be divided into two parallel machines each support four-layer decoder. Hence, in the receiver, the MIMO-detected LLR values are fed into the LDPC decoder that supports the required data rate.

Figure 5. 1728-bit LDPC code H-Matrix

Figure 6 shows the LDPC decoder block diagram. We use the layered decoding algorithm for reducing the amount of required memory and reducing the number of iterations. For higher throughput with low complexity, the decoder is implemented with 8-bit LLR precision.

Figure 6. LDPC Decoder structure

As enabled by the code depicted in Figure 5 with Zp=4, four LDPC decoders using the Base Code from Figure 5(a) simultaneously process four sub-codes of N=432 comprising the full 1728-bit block. However, to support the data throughput of 1.52 Gbps, only one 432-bit decoder is required.

The 11th Annual IEEE CCNC- Wireless Communications Track

541

H. RF/BB Interface

The RF/BB interface, shown in Figure 7, consists of two parts: signal and control. The signal path between the RF and BB units is passing through digital-to-analog converters (DACs) in the Tx signal path and analog-to-digital converters (ADCs) in the Rx signal path. The BB transceiver platform supports four High Speed FMC connectors. Each FMC connector supports a module with dual channel 16-bit DAC (I and Q) and dual channel 12-bit ADC (I and Q). The sampling frequency of the ADC/DAC modules is 552.96MHz. The ADC/DAC clock is derived by the RF unit (1105.92MHz with FMC board internal division by 2).

The control interface allows the BB unit to control the operating parameters of the RF unit, including phased-array beamforming and beamwidth, PA gain, LNA gain, VGA gain, and RF attenuation. The physical connection of the RF control is through an HDMI cable connected between the BB ADC/DAC module and the RF unit.

The system also provides a method for directly controlling the RF unit without the BB unit. This method of control uses a PC connected to the RF unit using a USB port. The PC can control all of the same RF parameters as the BB unit. PC-based control can be implemented using a graphical user interface (GUI) running on the PC, or by software which automates the functions such as beamforming and gain control. We use the PC-based control to test, calibrate, and analyze the performance of the RF units.

Figure 7. RF/BB Interface

Figure 8 shows the RF unit control interface. An FPGA module integrated in the RF unit provides the two methods of control previously described (BB unit control via HDMI cable, and PC control via USB port). The FPGA dispatches the control commands to the corresponding RF components within the RF unit.

Figure 8. RF/BB I/F Block diagram

I. Multiple Antenna configurations

As said above, the system can be configured in different ways: FDMA, SDMA, or multi-band MIMO mode. To support these modes, the AP is equipped with antenna arrays. Each antenna array consists of sub-arrays, where each sub-array is connected via digital chain to the baseband. The UE, however, is equipped with only one antenna array, which consists of sub-arrays. Note, each sub-array is capable of forming an analog narrow beam to transmit data. Also, each antenna arrays has bandwidth that covers all supported bands in the system (this is practically achievable in mmWaves band).

In the FDMA mode, the AP can support up to min( , number of bands) UE simultaneously, and each UE can has up to min( , ) spatial streams. The AP serves each UE on a separate antenna array and frequency band. In the SDMA mode, the AP can support up to UEs simultaneously, where up to UE can share the same frequency and time resource, but only separated spatially by different analog beams. Similar to the FDMA mode, in the SDMA mode, each antenna array transmits signal in only one frequency band. Finally, in the Multi-band MIMO mode, each user can be served on aggregated bands with up to streams. In this mode, an antenna array can transmit signal on multiple frequency bands concurrently.

IV. REAL-TIME TEST RESULTS Indoor over-the-air tests are performed on the final

integrated system with AP and UE. Each side has 28GHz RF unit, 552.96MHz ADC/DAC, and baseband running on FPGA [8]. The basic design can run up to 1.52Gbps on two Xilinx Virtex-6 FPGAs of the multi-FPGA system. The system is partitioned into a few clock domains to maximize the throughput of each separate module. The receiver on Virtex-6 FPGAs uses 98168 Slices, 279361 LUTs, 79387 Registers, and 212 DSP48s. The system is scalable to support the desired throughput. An SDMA MIMO system was demonstrated in the lab as shown in Figure 9. A real time dual video transmission was sent from the AP to the 2 UEs. The video application software at the AP sends uncompressed video packets. The video frame at the transmitter PC is packetized and sent via Gigabit Ethernet using the WinPcap API to the AP Baseband Tx FPGA where the video packets are transformed into an OFDM BB signal which sent to the Tx RF unit that transmit the two videos in two separate beams. After beam pairing process, each receiving endpoint UE, captures the incoming beam and down converts the RF signal to BB OFDM, The UE Rx BB FPGA sync, demodulates and decodes the data and generates the recovered video packets which are then sent over the Gigabit Ethernet to the corresponding Rx PC. The depacketization process in the Rx PC assembles the packets as a frame and renders it using DXVA 2.0 hardware acceleration.

Figure 10 describes the Error Vector Magnitude (EVM) test results done on the mmWave Tx/Rx RF units. The measured EVM for full Tx/Rx RF units with 500MHz OFDM QPSK modulation was -19 dB at distance of 10 m. The OFDM signal shown in Figure 10 included 1728 useful subcarriers between ±233.28 MHz at 552.96 MHz sampling frequency. From Figure 10 it can be seen that due the RF/IF frequency response, there

�

The 11th Annual IEEE CCNC- Wireless Communications Track

542

was some degradation of 5 dB between the positive OFDM frequencies to the negative OFDM frequencies.

Figure 9. Indoor SDMA demo setup with 10m separation between AP and

UEs.

Figure 10. EVM measurements show the performance of the RF units used in

the demo.

Figure 11 shows the error rate performance of the LDPC decoder block. Table II describes the corresponding LDPC decoder performance once synthesized and implemented in TSMC 65nmLP process. We can see that at high rates (3/4 and 13/16) a single LDPC decoder block can easily surpass 1 Gbps with low power consumption of only 60 mw which implies that the total decoding power consumption to support higher date rates, for example 10 Gbps, should not exceed 300 mw in TSMC 65nmLP process. Smaller dimension processes such as 32nm, and 28nm should reduce the power consumption even more.

TABLE II. ASIC Throughput and power consumption of the LDPC code decoder.

Figure 11 describes the LDPC decoder performance at

different rates and modulation scheme. It is shown in Figure 11 that the 16-QAM is worse than the QPSK by 6 dB. In addition, the Rate=3/4 LDPC code is worse than Rate=1/2 code by 3 dB. Because of the huge complexity of the system, the current receiver is partitioned onto 2 FPGAs, one for LDPC decoder, and the other for all other modules. The data are transferred between them through 6.6 Gbps GTX transceivers. The

expected overall throughput of our integrated design can reach 1.52 Gbps for the uncoded data. With the rate-13/16 LDPC code, this corresponds to 1.235 Gbps information bit rate. The maximum clock frequency in FPGA LDPC module is 28.8MHz, and the throughput is 1.61 Gbps.

Figure 11. Performance of the 1728-bit Quasi-Cyclic LDPC decoder over

AWGN channel.

V. CONCLUDING REMARKS In this paper, we presented algorithms and architectures to

realize high-data rate millimeter wave MIMO communication system. The system was implemented on FPGA and tested to prove that it is feasible to design and implement a high data rate MIMO beamforming system for millimeter wave short range communications with current technologies. Future work will focus on higher rates, lower power consumption, and wider band system realization to efficiently approach 30 Gbps rate.

VI. REFERENCES [1] C. R. Anderson and T. S. Rappaport, “In-Building Wide-band

Partition Loss Measurements at 2.5 and 60 GHz,” IEEE Trans. Wireless Communications., vol. 3, no. 3, May 2004.

[2] H. Xu et al., “Measurements and Models for 38-GHz Point-to-Multipoint Radiowave Propagation,” IEEE JSAC, vol. 18, no. 3, March 2000.

[3] “Millimeter wave propagation: Spectrum management implications”, Federal Communications Commission, Office of Engineering and Technology, Bulletin Number 70, July 1997.

[4] F. Gutierrez, S. Agarwal, K. Parrish, and T. S. Rappaport, “On-Chip Integrated Antenna Structures in CMOS for 60 GHz WPAN systems,” IEEE Journal on Selected Areas in Communications, vol. 27, no. 8, pp. 1367–1378, October 2009.

[5] Van Nee, R., "Breaking the Gigabit-per-second barrier with 802.11AC," Wireless Communications, IEEE , vol.18, no.2, pp.4,4, April 2011

[6] Baykas, T.; Chin-Sean Sum; Zhou Lan; Junyi Wang; Rahman, M.A.; Harada, H.; Kato, S., "IEEE 802.15.3c: the first IEEE wireless standard for data rates over 1 Gb/s," Communications Magazine, IEEE , vol.49, no.7, pp.114,121, July 2011

[7] Perahia, E.; Cordeiro, Carlos; Minyoung Park; Yang, L.L., "IEEE 802.11ad: Defining the Next Generation Multi-Gbps Wi-Fi," Consumer Communications and Networking Conference (CCNC), 2010 7th IEEE , vol., no., pp.1,5, 9-12 January 2010

[8] B. Yin, S. Abu-Surra, G. Xu, T. Henige, E. Pisek, Z. Pi, J. R. Cavallaro, "High-Throughput Beamforming Receiver for Millimeter Wave Mobile Communication," Accepted Globecom 2013, December 2013.

[9] S. Abu-Surra, E. Pisek, T. Henige, "Gigabit rate achieving low-power LDPC codes: Design and architecture," WCNC 2011, pp.1994-1999, March 2011.

The 11th Annual IEEE CCNC- Wireless Communications Track

543