Comparison of Reverse Recovery Behavior ofSilicon and Wide Bandgap Diodes in High

Frequency Power ConvertersDaniel Costinett andDragan Maksimovic

Colorado Power Electronics CenterECEE Department

University of Colorado BoulderBoulder, CO 80309

Email: Daniel.Costinett@Colorado.edu

Regan ZaneDepartment of Electrical

and Computer Engineering,Utah State University

North Logan, UT 84341

Alberto Rodrıguez andAitor Vazquez

Departamento de Ingenierıa Electrica,Electronica, de Computadores y Sistemas

University of OviedoGijon, SPAIN 33204

Abstract—The nature of reverse recovery losses is exam-ined in hard-switched and soft-switched converters, usingsilicon (Si), silicon carbide (SiC), or gallium nitride (GaN)devices. A loss model and experimental results with aprototype 150-to-400 V, 150 W, boost converter operatedat switching frequencies between 500 kHz and 2 MHz areused to characterize and quantify losses related to diodereverse recovery. It is found that reverse-recovery relatedlosses with Si diodes cannot be neglected even when theconverter is soft switched, with zero-current switching ofthe diode and zero-voltage switching of the transistor. Theswitching losses with SiC or GaN diodes are substantiallysmaller in all cases considered, and can be reduced tonegligible values when the converter is soft switching.

I. INTRODUCTION

Loss modeling of the boost DC-DC converter ofFig. 1 has been studied extensively in the field ofpower electronics. Indeed, the boost converter is oftenused as an example in introductory power electronicscourses to develop notions of conduction and switchinglosses, extending to concepts such as soft switching. Oneparticularly challenging step in the boost converter lossmodeling is the accounting of diode reverse recovery,which has been studied extensively in e.g. [1]–[5]. De-spite its prevalence, analytical or empirical estimation ofreverse recovery losses remains an area where extensiveexperimentation is often required, particularly when op-erating waveforms do not match closely the values orplots given in the device datasheets.

Because of the nature of reverse recovery calcula-tions and their impact on efficiency, solutions that caneliminate or mitigate their effects on circuit operation

CswQ1

Vg

idL Dil

vsw

+

_

vd +_ +

–Vout

Fig. 1. Circuit diagram of the boost converter. Csw is the lumpedswitched node capacitance, including MOSFET output capacitance,diode capacitance and inductor winding capacitance.

are attractive. Commercially available power diodes areusually designated as both “fast” and “soft” recoverydevices. Both of these characteristics are desirable inmitigating engineering challenges related to dealing withreverse recovery losses. Recent work, however, hassuggested that “snappy” recovery diodes could performbetter in applications designed with specific accountingfor peak reverse-voltage [6]. When device selectionis unable to achieve suitable performance, techniquessuch as snubbing circuits or synchronous and/or soft-switching are employed, with associated increases incircuit and control complexity, cost, and possibly stressesof circuit components [7]–[11].

Recent trends in power electronics have led to twosimple approaches to converter design which are as-sumed to result in significant reductions of diode reverserecovery related losses. First, it is commonly assumedthat reverse recovery losses can be reduced substantiallyif the inductor current ripple is increased and the diode

978-1-4673-4916-1/13/$31.00 ©2013 IEEE

current is allowed to drop to zero prior to MOSFETturn-on. This behavior is most easily seen in a boostconverter operating in discontinuous conduction mode.Of particular interest is soft switching operation such aszero-voltage-switching quasi-square-wave (ZVS-QSW),in which increased inductor current ripple allows theconverter to be operated with both zero current turn-offof the diode and zero voltage turn-on of the MOSFET,thus addressing all major switching loss mechanisms.Second, in contrast to silicon minority-carrier p-n junc-tion diode, majority-carrier device implementations inwide bandgap materials such as gallium nitride (GaN)[12]–[15] and silicon carbide (SiC) [16]–[18] have beenintroduced with claims of essentially no reverse recovery.

Trends in power electronics continue to push towardshigh switching frequency operation for improved powerdensity, dynamic response, and harmonic content [19]–[22]. As switching frequency increases, the impact ofdevice switching behavior becomes increasingly im-portant, challenging the commonplace assumptions re-garding mitigation of losses related to diode reverserecovery. As discussed further in this paper, in spite ofsignificantly slower ramp rates in the soft-switched ZVS-QSW boost, reverse recovery behavior and losses are stillobserved experimentally. Furthermore, previous studieshave shown the presence of reduced but nonzero reverserecovery waveforms with wide bandgap SiC devices[18], [23]–[25].

In light of the state of reverse recovery loss modeling,it is worthwhile to look again at the behavior of diodereverse recovery. The behavior is examined for the testcase of a high frequency, 150-to-400 V, 150 W boostconverter operating in both traditional, hard-switched(i.e. CCM) and ZVS-QSW modes, and with Si, SiC, andGaN diode devices. It is desired to understand when,and to what extent, reverse recovery effects are trulynegligible, taking into account varying operating modesand devices. Reverse recovery dynamics are reviewedbriefly in Section II. In Section III, the boost converteris examined in ZVS-QSW operation, with hard-switchedoperation examined in Section IV. Section V concludesthe paper.

II. REVERSE RECOVERY DYNAMICS

Typical minority-carrier silicon diode reverse recoverywaveforms during hard-switched operation are given inFig. 2. The total recovery charge Qrr = Qa + Qb

and time trr are parameters that are commonly givenin datasheets for silicon power devices; typically alongwith their dependence on the current ramp rate did/dt

t

id(t)vd(t)

diddt

QaQb

IRRM

ta tb

trr

IF

MOSFET turn-on

Fig. 2. Simplified diode current id and voltage vd during turn offwith reverse recovery parameters shown explicitly.

and the peak forward current IF . However, data iscommonly given for did/dt in the range of hundredsto thousands of ampere per microsecond, which is anappropriate range for hard-switched operation, but whichmay be orders of magnitude higher than the ramp ratesobserved in a ZVS-QSW converter and in other soft-switching configurations. It is understood that both trrand Qrr decrease (albeit nonlinearly) with lower ramprates, but as switching frequencies increase in the MHzrange, even reduced amounts of reverse recovery chargemay lead to significant losses and decrease in efficiency.Unfortunately, in the absence of relevant datasheet pa-rameters, modeling of reverse recovery related losses insoft switched converters is more difficult.

In the hard-switched waveforms of Fig. 2, the reverserecovery process is initiated when the Boost converterMOSFET is turned on. As the MOSFET channel forms,inductor current is shunted away from the diode andinstead flows through the MOSFET. However, until thestored charge is removed, the diode remains forwardbiased and vsw ≈ Vout causing significant instantaneouspower loss on the MOSFET. Eventually, all current isredirected to the MOSFET and id reaches zero. At thispoint, the diode current reverses direction in order toremove the stored charge Qa, after which the diode junc-tion may begin to become reverse biased, and the voltagevd begins to rise at a rate limited by the equivalent nodecapacitance Csw, which consists of components fromboth MOSFET output capacitance Coss, diode junctioncapacitance Cj and inductor winding capacitance. Dur-ing time interval tb, an additional amount of charge Qb

must be removed from the quasi-neutral region of thediode as vd is rising. In the case of a soft-recovery diode,

t

il (t)

vsw(t)

dildt

Qa Qb

ta

Qc

MOSFET turn-on

Fig. 3. Simplified inductor current il and switched-node voltagevsw during soft-switched operation with ZCS diode turn off and ZVStransistor turn on.

in order to reduce the voltage spike associated with anyparasitic package inductance in the diode during thereverse recovery process, the slope of the diode currentduring the removal of Qb is decreased in magnitude,causing a greater portion of the charge to be removedacross the increasing junction barrier voltage. If thetransition is a hard-switching one, as in the boost CCMmode, this simply leads to a redistribution of lossesbetween the diode and the transistor; all of Qb is removedthrough the output voltage Vout = vsw + vd.

If the transition is zero-voltage switching, as is thecase in the boost ZVS-QSW, the pertinent inductorcurrent il and switched-node voltage vsw waveforrmsare shown in Fig. 3. In this case, the diode turn-off isinitiated passively when the diode current reaches zero,with the MOSFET turn on delayed until the voltagetransition has been completed. This leads to ideally zerolosses in the MOSFET, as all current flowing into thedrain of the device is deposited on its output capacitance,including the charge Qa. Once Qa is removed and thediode turnoff is initiated, two charges, Qb and Qc due todiffusion and junction charges are required to completethe ZVS transition. The charge Qc includes effects ofboth the transistor output capacitance Coss and the diodecapacitance Cd which are charged through resonancewith L in order to increase the depletion region anddevelop the reverse voltage on the diode. The charge Qb

is once again removed from the diode through graduallyincreasing barrier voltage, with associated losses on thediode itself. Since a “soft” recovery diode has a slowerthe removal of Qb, a “snappy” recovery diode may

actually be more efficient in the soft switching (ZVS-QSW) case by allowing Qb to be removed at a lower bar-rier voltage. Further, the time taken to remove junctioncharge and allow diode turnoff to begin may constitute asignificant portion of converter switching period, forcingincreased duty cycle and therefore increasing RMS cur-rents in order to maintain output power, thus increasingconduction losses.

III. BOOST CONVERTER SOFT-SWITCHED

OPERATION

The three devices listed in Table I are used to testthe accuracy of loss modeling in the ZVS-QSW boost.Devices include a SiC Schottky, a GaN-on-Si Schot-tky, and a hyperfast Si diode. For each device, thedatasheet-reported curves for forward-biased conductionand reverse-biased capacitance are given in Fig. 4. Whileboth SiC and GaN diodes are 4 A rated devices, theSi diode was selected as a 15 A part due to supe-rior performance and speed of the ETX device seriescompared to alternate devices available with rated 4 Acurrent. Note that Si and SiC have similar capacitance,and therefore should exhibit similar switching behaviorin the absence of any reverse recovery, while the GaNdevice has elevated capacitance which allows favorableforward conduction characteristics. In all cases, the sameMOSFET is used, which is the TPH3006, a cascodeconnection of depletion mode GaN-on-Si HEMT andlow voltage Si MOSFET to achieve enhancement modeequivalent behavior. The boost is operated at 150 W,150-to-400 V conversion ratio, with switching frequencyvaried between 500 kHz, 1 MHz, and 2 MHz.

TABLE IDIODES USED IN PROTOTYPE CONVERTER

Part # Type IF,avg [A]

15ETX06 Si 15†

TPS3411PK‡ GaN 4

C3D04060A SiC 4†

Losses in 15 A device were lower than losses

in lower current rated A-FRED and HEXFRED

devices

A loss model of the converter is developed usingstate plane analysis to solve for all voltage and currentwaveforms, taking into account the zero-voltage turn-onof the MOSFET and zero-current turn-off of the diodewhich are inherent in ZVS-QSW operation. The analysisis then used to solve for the inductance L necessaryto obtain full soft-switching with minimum conduction

0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 20

2

4

6

8

10

VF [A]

I F [A

]15ETX06TPS3411C3D04060A

(a)

0 50 100 150 200 250 300 350 400 450 10

100

1000

VR [V]

Cd [p

F]

15ETX06TPS3411C3D04060A

(b)

Fig. 4. Diode characteristics: (a) forward current IF as a function of forward bias voltage VF and (b) diode capacitance as a function ofreverse-bias voltage VR.

losses on all devices, effectively placing operation nearthe hard-switched CCM boundary. This inductance isadjusted during experimentation to ensure ZVS-QSWoperation, with different values used at all switchingfrequencies and small variations among the devices dueto the varying diode capacitance and losses. Lossesare then calculated based on the state plane analysis,including conduction loss due to the full nonlinear diodeforward voltage in Fig. 4(a), MOSFET ron, inductorAC resistance, and inductor core losses. In all cases,the inductor is a 0L42020UG core implemented with17 turns of 138-strand 46-AWG Litz wire to minimizecopper loss. In loss modeling, switching losses includingreverse recovery losses are neglected because the con-verter is soft switched in ZVS-QSW mode of operation.Experimental results are then used to examine validityof the assumption that reverse recovery losses can beneglected in these cases.

Independent of switching losses, inductor copper andcore losses also vary with switching frequency. Inductorcore losses are calculated using the NSE/iGSE, withSteinmetz parameters adjusted for individual curve fits ateach switching frequency [26], [27]. Note that losses arelowest at 1 MHz due to the tradeoff in variation betweenswitching frequency and flux density at each operatingpoint. Due to the use of a single layer of Litz wire, verylittle variation is expected in winding resistance, whichis confirmed by finite element simulations. Additionally,converter RMS currents and therefore conduction lossesin all devices increase with switching frequency due tothe larger relative portion of the switching period takenby switching transitions. As an example, this effect isdiagrammed in Fig. 5 for the GaN diode current andvoltage waveforms at all three switching frequencies,with a normalized time axis. When denormalized, thecurrent ramp rates will be significantly larger in thehigher frequency operating conditions, though still wellbelow typically considered values in datasheet-reported

0

100

200

300

400

V sw [V

]

0 Ts/4 Ts/2 3Ts/4 Ts

−2

0

2

4

I l [A

]

time

fs = 0.5 MHzfs = 1 MHzfs = 2 MHz

Fig. 5. Inductor current il and switched-node voltage vsw duringZVS-QSW operation of the boost converter at varying frequency.The time axis is normalized to demonstrate the effect of increasedswitching frequency on RMS currents.

reverse recovery characteristics.Analytical predictions and experimental results for

each switching frequency and device implementation aresummarized in Table II. Note that at low frequency losscalculations are well matched to experimental results forall devices. As the switching frequency increases, onemay observe that neglecting reverse recovery could notbe fully justified in the case of GaN and SiC devices,although the efficiency predictions could be consideredadequate. However, in the soft-switched (ZVS-QSW)case with the Si diode, in spite of both MOSFET ZVSand diode ZCS, the modeling error is more significant,and it increases with switching frequency, clearly indicat-ing that switching losses are not eliminated. Additionally,despite the Si diode having capacitance well matchedto the SiC diode, frequency dependent losses are muchmore significant in the Si diode even with zero currentturn-off.

Unfortunately, due to the capacitor currents presentin soft switching operation, the diode reverse recovery

TABLE IILOSSES IN ZVS-QSW BOOST CONVERTER

fs [MHz] Part Type L [µH] Pcond [W] Pcore [W] ηpredicted [%] ηmeasured [%] Loss Error [%]

0.5 15ETX06 Si 60 0.98 2.97 97.4 97.1 9

0.5 TPS3411 GaN 50 0.77 2.91 97.6 97.4 5.6

0.5 C3D04060A SiC 66 0.84 2.97 97.5 97.5 -1.9

1 15ETX06 Si 26 1.05 2.28 97.8 96.5 36.5

1 TPS3411 GaN 23 0.85 2.22 98.0 97.6 14.6

1 C3D04060A SiC 26 0.90 2.28 97.9 97.8 3.48

2 15ETX06 Si 7.5 1.16 2.77 97.4 89.4 75.3

2 TPS3411 GaN 8.7 1.02 2.68 97.5 96.1 36.8

2 C3D04060A SiC 6.7 1.01 2.76 97.5 96.6 26.0

(a)

(b)

(c)

Fig. 6. ZVS-QSW inductor current il (Ch4), switched-node voltagevsw (Ch3), and transistor gate voltage vgs (Ch2) for Si, GaN, andSiC devices in (a)-(c), respectively.

current cannot be characterized external to the device,as charges Qb and Qc in Fig. 3 cannot be distinguished.However, using the model of Fig. 2, the time ta duringwhich a charge Qa must be removed from the diodebefore the voltage vsw can begin to drop can be observed.Thus, the transition is shown in more detail in thewaveforms of Fig. 7 for operation at 1 MHz, measuredusing a Tektronix DPO2014 oscilloscope with TCP0030current probe and P5205 differential voltage probe. Notethat although the ta values found for SiC and GaN diodesare nonzero, they are within measurement margins oferror for the given experimental setup. The value for theSi diode, however, is significant, indicating that reverserecovery behaviors are still present even when the silicondiode is zero-current switched followed by a zero-voltageswitching transient.

IV. BOOST CONVERTER HARD-SWITCHED

OPERATION

Next, the converter is operated in hard-switched CCMwith the same three devices of Table I. Inductance isincreased to L = 67µH and only the 1 MHz switchingfrequency is considered, but the circuit is otherwiseunaltered. Experimental waveforms for each device aregiven in Fig. 8. Because no significant resonant intervalsare included in hard-switched operation, state plane anal-ysis is not necessary. Losses are calculated in the samemanner as previously, now including switching lossesdue to the lack of ZVS. Losses are incurred from thenonlinear output capacitance of both transistor Coss(vds)and diode Cj(vd) are predicted using the method in[28]. Taking into account the voltage dependence of thecapacitances, the total switching losses due to device

(a)

(b)

(c)

Fig. 7. Comparison of ta for Si, GaN, and SiC devices (a-c).Waveforms are given for inductor current il (Ch4), switched-nodevoltage vsw (Ch3), and transistor gate voltage vgs (Ch2).

output capacitances are given by

Esw =

∫ Vout

0vxCoss(vx)+(Vout − vx)Cj(vx)dvx . (1)

Reverse recovery losses are again neglected. This isknown to be a very poor approximation for the Si diode,but it is desired to test whether this model remainsaccurate for the GaN and SiC devices. The resulting cal-culations are compared to experimental data in Table III.As expected, Si diode modeling results, without takinginto account reverse recovery, are a poor prediction of ex-perimental efficiency. Both GaN and SiC results remainfairly accurate, with GaN efficiency being significantlyworse due to the larger diode capacitance. Note, however,that despite the use of wide bandgap materials for bothMOSFET and diode, significant efficiency improvements

(a)

(b)

(c)

Fig. 8. CCM operation for Si, GaN, and SiC devices (a-c).Waveforms are given for inductor current il (Ch4), switched-nodevoltage vsw (Ch3), and transistor gate voltage vgs (Ch2).

are still obtained through employment of soft switching.At 1 MHz switching frequency and 150 W output power,the additional conduction losses incurred from the largercurrent ripple in the ZVS-QSW case are less than 6% ofthe losses saved through soft switching.

V. CONCLUSIONS

The nature of reverse recovery losses is reviewed,and common assumptions of cases in which the lossesmay be neglected are examined. A loss model andexperimental results with a prototype 150-to-400 V,150 W, boost converter operated at switching frequenciesbetween 500 kHz and 2 MHz are used to characterize andquantify losses related to diode reverse recovery. The ex-periments show that switching losses remain significant

TABLE IIILOSSES IN CCM BOOST CONVERTER

Part Type Pcond [W] Pcore [W] Psw [W] ηpredicted [%] ηmeasured [%] Loss Error [%]

15ETX06 Si 1.60 2.59 8.7 91.4 80.3 56.5

TPS3411 GaN 1.33 2.59 14.3 87.8 86.0 13.1

C3D04060A SiC 1.45 2.59 8.4 91.7 90.1 16.6

in high frequency operation with silicon (Si) devices,even under soft-switching conditions: zero current turn-off of the diode and zero voltage turn-on of the transistor.An important conclusion is that reverse-recovery relatedlosses with Si diodes cannot be neglected even whenthe converter is soft switched. Furthermore, althoughthe switching losses are significantly reduced with widebandgap devices (silicon carbide (SiC) or gallium nitride(GaN)), the switching losses are not negligible in thehard-switched prototype. The best results are obtainedwhen soft switching is applied in the prototypes with SiCor GaN devices, with modeling and experimental resultsdemonstrating that switching losses can be reduced tonegligible values at switching frequencies in the MHzrange.

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