46 IEEE ANTENNAS & PROPAGATION MAGAZINE1045-9243/19©2019IEEE A U G U S T 2 0 1 9

Digital Object Identifier 10.1109/MAP.2019.2920664Date of publication: 27 June 2019

In this article, we introduce a novel planar chipless radio-frequency identification (RFID) tag. The tag is composed of multiple miniaturized reflecting resonators based on a slow-wave structure. To validate the proposed approach, a tag with

a coding capacity of 16 b was designed with a compact size of .15 21 mm2# Tags with different pattern configurations were

fabricated using a Rogers RO4003 substrate, and their radar cross-section (RCS) responses were measured. Compared to conventional multiresonator tags, the proposed tag offers a good miniaturization ratio and spectral coding efficiency. In addition, the measurements revealed a high quality factor (Q-factor) and coding robustness, which demonstrates the efficiency of the used approach to develop high-performance chipless tags.

BACKGROUNDToday, RFIDs and optical bar codes are the most used tech-niques in wireless data capturing. However, given the advan-tages of the RFID technology over the bar code, namely, the possibility of item detection without face-to-face communi-cation, the viability of simultaneous multiple detection, the long detection range, and the feasibility of detection through

obstacles, this recent technology is increasingly developing. Although conventional RFID systems based on the use of sil-icon RFID chips are very common and widely exploited, the chip has many limitations, such as delicacy and a high cost of approximately US$0.3. Therefore, researchers invented an alternative solution, called the chipless RFID, which is locat-ed on the border between the traditional RFID and the opti-cal bar code [1], [2]. This technology has the aforementioned mentioned advantages of the conventional RFID while offer-ing a unit cost (approximately US$0.005) comparable to that of an optical bar-code tag (approximately $US0.0015) [2].

Chipless RFID tags are fully passive microwave circuits that use electromagnetic signals for encoding data. Accordingly, they can be referred to as electromagnetic bar-code tags. There are two families of chipless RFID tags: RCS based and retransmission based. RCS-based chipless RFID tags use reflecting resonators that receive the signal and send it back with the tag signature [1]–[16]. In [1], a compact RCS-based tag using reflecting resona-tors in is presented. The reported tag offers a coding capacity of 20 b operating in the frequency range of 2–4 GHz with an overall size of .25 70 mm2#

Retransmission-based chipless RFID tags [17]–[24] utilize single or double reflecting resonators to receive and transmit the signal. The spectral signature of the tag is obtained by a multiple

Using a slow-wave approach for miniaturized structure.

High-Performance Chipless Radio-

Frequency Identification Tags

Maha Added, Noureddine Boulejfen, Milan Svanda, Fadhel M. Ghannouchi, and Tan-Phu Vuong

47IEEE ANTENNAS & PROPAGATION MAGAZINE A U G U S T 2 0 1 9

resonator circuit where each resonator creates a notch or a peak around a given frequency point. The chipless tag introduced in [22] is a retransmission-based tag using spiral resonators that allow the coding of 35 b in a frequency band ranging from 3.1 to 7 GHz. In [23] and [24], a retransmission-based tag that uses microelectro-mechanical-systems-based switches to pattern the tag backscat-tered signal and support communication between the tag and RFID reader is presented. Even if the retransmission-based tags have a robust bit pattern and ensure a large reading range due to the use of independent antennas for both transmission and recep-tion, this type of chipless tag usually has a relatively large size, which limits their use. On the contrary, RCS-based tags are able to reach a high coding capacity with a relatively compact size.

THEORYThe RCS-based tag presented in this article consists of 16 slow-wave-structure-based reflecting resonators excited by a linearly polarized planar wave. The code pattern of the tag, or the tag code, is the spectral signature created by the reflect-ing resonators. A basic structure for the designed reflecting resonator used in the proposed tag is based on the well-known half-wave dipole reflecting resonator, as shown in Figure 1. The length of the dipole reflecting resonator is equal to / ,2gm where gm is the wavelength of its corresponding resonant frequency fres in the substrate. As stated in [19], fres is inde-pendent from the linewidth. However, a reduced linewidth leads to an enhanced resonance Q-factor. A half-wave dipole reflecting resonator of the length .I 10 5 mmhwl = and the width .w 0 8 mmhwl = was designed for the resonant frequency

.f 8 45 GHzres = on a Rogers RO4003 substrate with the thick-ness . ,h 0 83 mm= a dielectric constant . ,3 55rf = and a loss tangent . ,0 0027tgd= simulated using Computer Simulation Technology (CST) Studio Suite.

As shown in Figure 2, the RCS response and the radiation pattern of the simulated half-wave dipole reflecting resonator reveal a resonant frequency of approximately 8.4 GHz and an omnidirectional radiation pattern at the plane perpendicular to the dipole axis. This pattern allows the detection of the reflected wave regardless of the resonator orientation, which is essential for the RFID applications. Starting from the basic structure described previously, a slow-wave structure was designed to reduce the length of the half-wave dipole reflecting resonator while keeping the same electrical behavior.

As shown in Figure 3 and based on the slow-wave concept, a conventional microstrip line with a given length is substituted by a shorter microstrip line loaded with equally spaced capacitances terminated to ground [26]–[29]. The role of the loading capaci-tances is to slow down the wave propagation along the microstrip line. Hence, this exchange results in a longer effective electrical length compared to that of an unloaded microstrip line. Based on

this approach, the values of the loading capacitances Cp and their spacing d have to be accurately determined to obtain the desired electrical behavior using a reducing line length.

Let’s consider an unloaded lossless transmission line with the characteristic impedance Zc_un and the phase velocity Vp_un given by [28], such as

Z CL

_c un = (1)

,VLC1

_p un = (2)

lhwl = 10.5 mm

11.5 mm

5 m

m whwl = 0.8 mm

FIGURE 1. The simulated basic structure of a half-wave dipole reflecting resonator.

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S (

dBm

2 )

2 4 6 8 10Frequency (GHz)

(a)

(b)

12 14 16 18 20

030

60

90

120

150180

150

120

90

60

30φ = 0 φ = 180

–80 –60 –40

θ/° Versus dBm2

Bistatic Scattering RCS Abs (φ = 0)

FIGURE 2. The simulation results of the basic half-wave dipole reflecting resonator. (a) The RCS response and (b) the radiation pattern in the dipole plane.

48 IEEE ANTENNAS & PROPAGATION MAGAZINE A U G U S T 2 0 1 9

where L and C are the line inductance- and capacitance-per-unit length of the transmission line, respectively. Periodically loading the transmission line with equally spaced shunt capaci-tors Cp allows the reduction of its characteristic impedance and phase velocity. For the spacing d between the capacitors infe-rior to the signal wavelength, the obtained characteristic imped-ance Zc_hwl and phase velocity Vp_hwl are given by [29] such that

ZC d

CL

_cp

hwl =

+

(3)

.VL C d

C1

_pp

hwl =

+c m (4)

Equation (4) clearly shows the reduction of the phase veloc-ity Vp_hwl compared to that of the unloaded line. This means that an effective electrical length can be achieved using a transmission line with a shorter physical length. The effective electrical length of the loaded line is expressed as follows:

,VNd Nd L C d

C _p

p00lo

lo4

~~= = +c m (5)

where N is the number of loading capacitors and 0~ is the angular frequency of interest. Equations (1)–(5) enable us to determine the values of the spacing d and the capacitance Cp of the loading capacitors such that

,d N Z

Z V_

_ _

c

c p

0 un

hwl lo un4

~= (6)

( ) .CN Z Z

Z Z _ _

_ _p

c c

c c

0

2

2

2

un hwl

lo un hwl4

~=

- (7)

To obtain an entirely planar circuit, the loading capacitances Cp can be realized using open-circuit stubs by applying the fol-lowing formula [29]:

C Z Vl

_ _p

c p stub stub

stub= (8)

for ( ) ,/V 1_p0 stub %~ where ,lstub ,Z _c stub and V _p stub are the length, characteristic impedance, and phase velocity of the stub, respectively.

DESIGN CONSIDERATIONSDepending on the planar technology to be used to implement the chipless RFID tag, some limiting constraints have to be taken into account. Based on [26]–[30], for microstrip technolo-gy, d and lstub should be less than 1/10 of the signal wavelength at the highest frequency of interest. Additionally, the reduction factor determined by the ratio /Z Zc c_hlw _un must be less than one for a slow-wave behavior. Thus, to achieve the highest possi-ble length reduction, the difference between the loaded ( )Zc_hlw and unloaded ( )Zc_un should be at its highest possible value.

Obviously, the length of the unloaded line is given by .l N d#= Thus, the values of d and Cp are inversely propor-

tional to the number of capacitors .N When N increases, the values of d and Cp decrease; this means that for a given ,Zstub the value of lstub is reduced. However, to minimize the cross-talk among stubs, the spacing d must be greater than ,h3 where h is the height of the used substrate. This condition ( )d h3#$ can be relaxed by placing the stubs on both sides of the line. Moreover, it is possible to further minimize the length of the stub by reducing its characteristic impedance, thus increasing its width to a certain limit to avoid coupling between stubs. Taking into account all of these design consid-erations, a slow-wave-structure-based reflecting resonator with a reduced size can be developed to substitute the basic half-wave dipole reflecting resonator described previously while maintaining the same electrical and radiating behaviors.

To obtain the highest possible line-length reduction, the characteristic impedance Zc_hlw of the basic half-wave dipole resonator should be at its lowest possible value; this explains the choice of the width .w 0 8 mmhwl = (Figure 1), which offers an acceptable resonance Q-factor with the relatively reduced char-acteristic impedance . .Z 78 1c_hwl X= In addition, the unloaded microstrip line impedance Zc_un should be at its highest value. Therefore, the unloaded microstrip linewidth wun is fixed to

.w 0 2 mm,un = allowing a characteristic impedance of approxi-mately .Z 113c_un X=

To calculate the parameters of the slow-wave-based reflect-ing resonator that offers the same behavior as the basic half-wave dipole, (1)–(7) were used. Thus, the unloaded microstrip line has to be loaded by N 6= capacitances of .C 0 74 pFp = spaced by . .d 1 2 mm= Therefore, its length l is equal to

. ,l N d 7 2 mm#= = giving a length reduction of 31% com-pared to the basic half-wave dipole resonator.

For a planar structure, the capacitances Cp were imple-mented using open stubs. The stub width is fixed to

.w 0 2 mm,stub = and its length lstub was calculated using (8) to

lhwl, whwl, Zc_hwl, Vp_hwl

l

wun, Zc_un, Vp_un

Cp Cp Cp

d

d/2 d/2

l stu

b

Zstub, Vstub

FIGURE 3. The slow-wave approach principle.

49IEEE ANTENNAS & PROPAGATION MAGAZINE A U G U S T 2 0 1 9

obtain . .l 1 7 mmstub = The designed slow-wave-based reflect-ing resonator is described in Figure 4. To avoid mutual coupling between stubs, as shown in Figure 4, the stubs are alternatively placed on both sides of the line in such a way to obtain a spacing of .d2 2 4 mm# = between stubs, which satisfies the condition of .d h3#$

To verify the efficiency of the used slow-wave approach, the simulation of the developed slow-wave-based reflecting resonator was performed. The RCS responses and the radiation pattern of the basic half-wave dipole reflecting resonator and its equivalent slow-wave-based one are illustrated in Figure 5. According the RCS response of the designed slow-wave-based reflecting resonator shown in Figure 5(a), a resonant frequency of approximately 8.4 GHz was noted, which corresponds to the resonant frequency of the basic half-wave dipole reflecting resonator. Thus, a good match is observed between the RCS responses of the two resonators.

Similarly, an omnidirectional radiation pattern is offered by the slow-wave-based reflecting resonator [see Figure 5(b)], which can be well compared to that of the basic half-wave dipole reflecting resonator shown in Figure 2(b). Moreover, the slow-wave structure exhibits a resonance with a higher Q-factor, which improves the detection and recognizability of the corre-sponding bit value in the tag pattern. Thus, the effectiveness of the used slow-wave approach was proved, and a similar electri-cal and radiating behavior to that of the half-wave dipole resona-tor with a length reduction of 30% was reached.

TAG DESIGNAfter demonstrating the effectiveness of the slow-wave approach in reducing the length of one reflecting resonator, an RCS-based chipless tag of 16 slow-wave reflecting resonators operat-ing in the X frequency band was designed. An overall length reduction is expected for the designed tag. However, its width is expected to increase due to the presence of the shunted stubs. To reduce this effect, the different reflecting resonators of the tag were overlapped to each other while relying on a tuning process and a rearrangement method, described in the “Experi-mental Validation” section, to avoid the mutual coupling and to ensure a high tag coding robustness. The structure of the pro-posed tag is presented in Figure 6.

As revealed by Figure 6, the tag contains 16 slow-wave-based reflecting resonators placed in two columns of eight resonators, from res1 to res8 in Figure 6(a) and from res9 to res16 in Figure 6(b). Tables 1 and 2 illustrate the resonant

wun

wstub

lstub

8 mm

5 m

m

l

d

2 × d

FIGURE 4. The designed slow-wave-based reflecting resonator.

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S (

dBm

2 )

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(a)

(b)

12 14 16 18 20

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60

90

120

150180

150

120

90

60

30φ = 0 φ = 180

–70–60

–40–50

θ/° Versus dBm2

Bistatic Scattering RCS Abs (φ = 0)

Slow-Wave-Based Reflecting ResonatorHalf-Wave Dipole Reflecting Resonator

FIGURE 5. The simulation results of the slow-wave-based and half-wave dipole reflecting resonators: (a) the RCS responses of the two resonators and (b) the radiation pattern of the slow-wave-based reflecting resonator in the dipole plane.

res9

res16

res1

(a) (b)

res8

FIGURE 6. A slow-wave-based structure for a 16-b chipless RFID tag with the tag code set to 1111111111111111; (a) from res1 to res8 and (b) from res9 to res16.

50 IEEE ANTENNAS & PROPAGATION MAGAZINE A U G U S T 2 0 1 9

frequencies as well as the parameters of each basic half-wave dipole reflecting resonator and those of their corresponding slow-wave-based ones. For the half-wave dipole reflecting resonators, lengths depend on the desired resonant frequen-cies, while widths are determined in such a way to obtain the characteristic impedance ,Zc_hwl allowing, after the slow-wave transformation, a fixed spacing d between stubs for all of the resonators. This is relevant in the sense that fixing a constant spacing d between the stubs allows the overlap of resonators, which further reduces the size of the tag. Thus, the spacing d for all resonators is fixed at . .d 1 2 mm= Furthermore, the unloaded linewidth wun and stub width

wstub are the same for all of the resonators and equal to .w w 0 2 mm.un stub= =

The resonators belonging to Figure 6(a) and (b) have a corresponding common unloaded microstrip line length, l1 and ,l2 and number of sections N1 and N2, respectively. Table 1 illustrates the parameters of the reflecting resona-tors belonging to Figure 6(a) from res1 to res8, where the number of sections is equal to N 61 = and the unloaded line length is equal to . .l 7 2 mm1 = Table 2 illustrates the parameters of the reflecting resonators of Figure 6(b) from res9 to res16, with the number of sections N2 5= and the unloaded line length .l 6 mm2 =

A tuning process is performed to adjust the spacings between reflecting resonators so as to minimize the cou-pling effect and to ensure the stability of the resonant frequencies when changing the configuration pattern of the tag. The dimensions of this tag are .15 21 mm2# Based on the on–off keying coding tech-nique used by the proposed tag, the presence of a resonant frequency cor-responds to the corresponding bit set to 1, while its absence corresponds to a bit set to 0. To modify the tag code and set certain bits to 0, it is required to simply remove the unloaded lines of the corre-spondent reflecting resonators. Figure 7 shows the layouts of the tags with codes 1111111111111111, 1010101010101010, and 1000000010000001, respectively.

The simulated RCS responses of the developed tags for the different described tag codes can be seen in Figure 8. For the structure with a tag code set to 1111111111111111, the red curve in the figure, each reflecting resonator resonates at a precise frequency corresponding to 1 b of the tag code. Thus, 16 resonances are detected in the spectral signature of the developed tag, which offers a tag coding capacity of 16 b. The resonant frequen-cies range from 8.36 to 11.48 GHz, with a frequency spacing of approximately 230 MHz. The dotted black curve repre-sents the spectral signature of the tag with a tag code set to 1010101010101010, while the blue one corresponds to the tag code 1000000010000001. As is clearly shown from these two curves, the resonant fre-quencies corresponding to the bits that are set to 0 have been removed, and those corresponding to the bits that are set to 1 are always present.

Although high Q-factors are revealed by Figure 8 for all of the

TABLE 1. PARAMETERS OF THE REFLECTING RESONATORS IN FIGURE 6(a).

Half-Wave Dipole Reflecting Resonator

Slow-Wave Equivalent Structure

Length Reduction (%)

Resonator (From Top to Bottom)

Frequency of Resonance (GHz) lhwl (mm) w (mm) Zc_hwl (Ω) Cp (pF) lstub (mm)

res1 8.451 10.56 0.82 72.65 0.76 1.7 31.8

res2 8.528 10.47 0.78 74.23 0.71 1.6 31.2

res3 8.785 10.31 0.75 75.48 0.67 1.5 30.1

res4 8.918 10.1 0.72 76.78 0.63 1.4 28.71

res5 9.151 9.97 0.68 78.59 0.58 1.3 27.78

res6 9.356 9.74 0.65 80.02 0.54 1.2 26.07

res7 9.55 9.55 0.61 82.01 0.49 1.1 24.6

res8 9.772 9.39 0.58 83.6 0.45 1 23.3

N1 = 6, l1 = 7.2 mm, and d = 1.2 mm.

TABLE 2. PARAMETERS OF THE REFLECTING RESONATORS IN FIGURE 6(b).

Half-Wave Dipole Reflecting Resonator

Slow-Wave Equivalent Structure

Length Reduction (%)

Resonator (From Top to Bottom)

Frequency of Resonance (GHz) lhwl (mm) w (mm) Zc_hwl (Ω) Cp (pF) lstub (mm)

res9 9.926 8.82 0.82 72.92 0.63 1.7 31.9

res10 10.141 8.69 0.79 74.13 0.6 1.6 30.9

res11 10.336 8.58 0.75 75.79 0.56 1.5 30

res12 10.565 8.44 0.72 77.12 0.52 1.4 28.9

res13 10.776 8.36 0.68 78.95 0.48 1.3 28.3

res14 11.023 8.17 0.65 80.41 0.45 1.2 26.6

res15 11.256 8 0.61 82.42 0.41 1.1 25

res16 11.526 7.85 0.58 84.04 0.37 1 23

N2 = 5, l2 = 6 mm, and d = 1.2 mm.

51IEEE ANTENNAS & PROPAGATION MAGAZINE A U G U S T 2 0 1 9

resonances, a small frequency shift can be observed for certain resonances when changing the tag code. This can affect the coding robustness of the proposed tag. This is because of the residual mutual coupling between the reflecting resonators. The mutual coupling is due to the reduced distance between resona-tors, which detunes the response of neighboring resonators in the tag. Thus, the logical zeros that are coded by the absence of the corresponding resonance are difficult to detect as the neigh-boring resonances are detuned from their original positions (see Figure 9). This reduces the reliability of reading the coded information or even makes it impossible to do so [9]. The influ-ence of the mutual coupling is crucial on the proper behavior of the tags composed of arrays of resonators, as shown in [9].

To palliate this problem, a technique presented in [9] and aimed at minimizing the inter-resonator mutual coupling was employed. As described in Figure 10, the reflecting resona-tors were rearranged so as to keep away each two reflecting resonators with neighboring resonant frequencies. The original ascending order of resonators res1 to res8 and res9 to res16 presented in Figure 10(a) is modified to res7 res2 res5 res8 res3 res6 res1 and res15 res10 res13 res16 res11 res14 res9 res12 [see Figure 10(b)]. In this layout, the resonators with neighboring resonance frequencies are then located further apart. Thus, their coupling is substantially reduced.

To prove the efficiency of the employed technique, the simu-lation of the new designed tags with rearranged reflecting reso-nators was performed for the three tag codes 1111111111111111,

1010101010101010, and 1000000010000001. A comparison among the RCS responses of the tags with the initial resonators arrangement and those with rearranged resonators is illustrated in Figure 9. It’s quite clear from the figure that from one tag code to another, the frequency shifts observed for the first and the last resonances are considerably reduced, and the Q-factors of the resonances are improved by rearranging the reflecting resonators’ positions. This demonstrates that the used rearrangement method enhances the robustness and the detectability of the RCS responses.

EXPERIMENTAL VALIDATIONThe tag with the rearranged design was fabricated for a fre-quency range of 8–12 GHz with all of the three configuration patterns, as shown in Figure 11. The RCS responses of the tags were measured in an anechoic chamber using the monostatic

(a) (b) (c)

FIGURE 7. The tag layout for the tag code set to (a) 1111111111111111, (b) 1010101010101010, and (c) 1000000010000001.

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11 11.5 12

111111111111111110101010101010101000000010000001

FIGURE 8. The simulated RCS responses of the designed tags with the tag codes set to 1111111111111111, 1010101010101010, and 1000000010000001.

–25–30–35

–40

–25–30–35

–40–45

–45

RC

S (

dBm

2 )R

CS

(dB

m2 )

8 8.5 9 9.5 10 10.5Frequency (GHz)

11 11.5 12

8 8.5 9 9.5 10 10.5Frequency (GHz)

(b)

(a)

11 11.5 12

111111111111111110101010101010101000000010000001

FIGURE 9. The simulated RCS responses of the designed tags representing three different tag codes, 1111111111111111, 1010101010101010, and 1000000010000001. (a) Tags with initial resonators arrangement. (b) Tags with rearranged resonators.

res1res2res3res4res5res6res7res8

res9res10res11res12res13res14res15res16

res7res2res5

res3res6res1res4

res15res10res13res16res11res14res9res12

res8

(a) (b)

FIGURE 10. The layout of the 16-b chipless RFID tag with the 1111111111111111 configuration: (a) the initial resonators arrangement and (b) the rearranged resonators.

52 IEEE ANTENNAS & PROPAGATION MAGAZINE A U G U S T 2 0 1 9

measurement by the R&S ZVA40 network analyzer (see Fig-ure 12). The tag was based on the evaluation of the reflection coefficient of a double-ridged horn antenna DRH20 [31], in front of which a tag at a distance of 15 cm was placed. The far field for the horn aperture diagonal of the size 0.1 m is approxi-mately 0.65 m at 10 GHz ( / ),2 D2

0m where 0m is the free-space wavelength, so that the condition for evaluation of the RCS by measurement in the far-field zone is not fulfilled. However, the measurement should verify only the RCS resonant peaks’ dis-tinction for tag codes. The calculation of the tag response was performed by the relation used in [9]:

,S SS S

11 11

11 11tagref iso

tag isorefv v=

-

- (9)

where S11tag and S11

ref symbolize the reflection coefficients in the case where the measured tag and a reference metallic plate are

used as scatterers, respectively. S11iso is the reflection coefficient

of the antenna itself if no scatterer is used, and it comprises the residual reflection from the experimental surroundings. The

tagv stands for the RCS of the measured tag, and refv repre-sents the RCS of the reference scatterer, which is the rectan-gular metal plate of a b 25 30 mm2# #= in size (near the tag size) and 0.3 mm in thickness. It is given by [33] as

.a b4 2

2 2refv r

m= (10)

The monostatic measurement arrangement enables the avoid-ance of the use of an angular-dependent formula for the refer-ence scatterer and eliminates the influence of mutual coupling of the transmitting and receiving antennas in the case of bistatic measurement. Figure 13 illustrates the measured RCS respons-es of the three different 16-b chipless tags with the tag codes 1111111111111111, 1010101010101010, and 1000000010000001. As is clearly noted, 16 resonances with high Q-factors, ranging from 8.57 to 11.8 GHz, were detected in the case of the tag with the 1111111111111111 code, which meets the required specifica-tions of the designed tag. The RCS levels of the resonant peaks range from −40 to −25 dBsm. The decision threshold for bit evaluation can be set as (in parts) monotonously constant or as an increasing curve interleaving the resonant peaks. It is conspicuous in both the simulated and measured results.

Equations (11) and (12) represent the spectral bit capacity (SBC) and the spatial bit density (SBD), respectively, to evaluate the demand on the frequency allocation and spatial requirements:

( ) ( )bit/GHz frequency bandwith GHznumber of bitsSBC = (11)

( )

SBD bit/cmtag size cm

number of bits .22=^ h (12)

By applying (9) and (10) on the measured results, the SBC offered by the proposed tag is equal to 4.95 b/GHz, and the SBD is equal to 5.08 b/cm .2

For the modified tag codes, shown in the blue and black curves, Figure 13 shows that the bits set to 0 were completely

(a) (b)

(c) (d)

FIGURE 11. The realized tags with different tag configuration patterns: (a) and (b) the tag with the 1111111111111111 code, (c) the tag code set to 1010101010101010, and (d) the tag code set to 1000000010000001.

The Tag

FIGURE 12. The measurement setup in an anechoic chamber.

–22–24–26–28–30–32–34–36

RC

S (

dBm

2 )

8 8.5 9 9.5 10 10.5Frequency (GHz)

11 11.5 12

111111111111111110101010101010101000000010000001

FIGURE 13. The measured RCS responses of the designed tags with the tag codes set to 1111111111111111, 1010101010101010, and 1000000010000001.

53IEEE ANTENNAS & PROPAGATION MAGAZINE A U G U S T 2 0 1 9

removed and that those set to 1 are always presents. It’s clear that for all of the realized tags, the resonant frequencies are sta-ble even if the tag code changes, which demonstrates the cod-ing robustness and the high detectability of the proposed tag.

To highlight the effectiveness of the miniaturization tech-nique used in this article, the slow-wave approach, a compara-tive study between the proposed tag and the tag presented in [1] and taken over by [9] was established. Vena et al. [1] proposed an RFID chipless tag operating in the frequency range of 2–3.9 GHz based on 20 U-shaped reflecting resonators, leading to a coding capacity of 20 b for a size of ,70 25 mm2# which presents an important surface coding density in the literature.

For a consistent comparison between the tag of Vena et al. [1] and the proposed tag, a 20-b slow-wave-based tag operat-ing in the same frequency band was developed. The tag motif was etched on the low-loss dielectric substrate Rogers RO4003 ( . , . )tan3 55 0 002rf d= = with a thickness of 0.83 mm. The size of the proposed 20-b slow-wave structure is equal to

.45 28 mm2# To validate the designed tag, a simulation of the RCS response was performed using CST Studio Suite and is illustrated in Figure 14 which reveals 20 resonance frequencies with high Q-factors. All of the resonances are distributed on a frequency band ranging from 2.59 to 3.88 GHz.

To summarize, for the same operating frequency band of the tag of Vena et al., the proposed tag allows a coding capacity of 20 b for a size of ,45 28 mm2# which offers a size reduction of 35.7% compared to the tag of Vena et al. More-over, an SBC of 15.5 b/GHz and an SBD of 1.59 b/cm2 are achieved through the proposed tag over an SBC of 10 b/GHz and an SBD of 11. b/cm2 for the tag of Vena et al. Furthermore, the sig-nificant advantage of the proposed tag in comparison with [1] is due to the relatively high RCS level as well as the depth and height of the drops and peaks (from 3 to 15 dB versus 0.5 to 6 dB), leading to higher SBD and SBC. Table 3 depicts the comparison of sev-eral typical representatives of previ-ously published frequency-domain chipless RFID transponders. SBD is measured as follows:

SBD ( )( )

bit/transponder size

number of bits .22mm

=

(13)

Taking into account the operating frequency band, a well-balanced ratio between the SBC and the SBD as well as a satisfactory RCS level can be observed in our design. The obtained results confirm the efficiency of the proposed slow-wave structure to design miniaturized RFID chipless tags with high performances.

CONCLUSIONSIn this article, a slow-wave structure was used to design min-iaturized RCS-based RFID chipless tags operating in the X frequency band. The designed tags allow a coding capac-ity of 16 b for a compact size of .15 21 mm2# The reflecting resonator’s positions were arranged to minimize the mutual coupling between reflecting resonators. The measured RCS response of the tag when all bits are set to 1 reveals 16 reso-nant frequencies with high Q-factors that are well detectable. Moreover, the two tags with the tag codes 1010101010101010 and 1000000010000001 were fabricated. The measured RCS responses of the three tags show a high coding robustness, stability, and detectability. The proposed tag design exhib-ited a considerable improvement of the resonant Q-factors and

TABLE 3. A PERFORMANCE COMPARISON OF FREQUENCY-DOMAIN, POLARIZATION-INDEPENDENT

CHIPLESS RFID TAGS.

ReferenceFrequency Range (GHz)

RCS Level (dBsm)

Minimum Peak Height (dB)

SBC (b/GHz)

SBD (b/cm2)

SBD (b/λ2)

Verification of 0-b Coding

[1] 2–4 −35 0.5 10 1.1 110 No

[3] 6–12 −35 2 1.3 3.7 41.1 Yes

[4] 3.5–6 −30 5 2 2.4 95.7 No

[5] 2–5 −35 7 1 0.3 22 No

[6] 5–15.5 −25 2 3.4 5.8 49.7 No

[7] 5.5–15 −27 5 0.95 3.8 32.6 Yes

[8] 5–7 −32 20 1.5 0.3 7.5 No

[9] 2.3–4 −35 2 13.3 1.1 99.8 Yes

[10] 8–14 −35 3 3.2 10 74.4 Yes

[11] 1.8–3.6 −33 4 12.5 0.7 86.4 Yes

[12] 3–9 −20 1 4 2.6 65 Yes

[13] 6.5 −40 5 4.5 18.9 435.5 Yes*

Our design

8.3–12 −26 3 4.3 5.1 44.6 Yes

* A negative result of verification: frequency shifts in the positions of the resonances are demonstrated.

–18

–20

–22

–24

–26

RC

S (

dBm

2 )

2 2.5 3 3.5Frequency (GHz)

4

FIGURE 14. The simulated RCS response of the proposed 20-b slow-wave-based tag with the tag code set to 11111111111111111111.

54 IEEE ANTENNAS & PROPAGATION MAGAZINE A U G U S T 2 0 1 9

spectral coding density, making it a good candidate for high-performance miniaturized RFID chipless tags.

ACKNOWLEDGMENTThis work was supported by the Czech Science Foundation under project 17-02760S.

AUTHOR INFORMATIONMaha Added (maha.added@gmail.com) is pursuing a Ph.D. degree in electrical engineering at the Microwave Electronics Research Laboratory, the Faculty of Sciences of Tunis, Tunisia. Her current research interests include chipless radio-frequency identification technology and the design of passive microwave components.

Noureddine Boulejfen (nboulejf@ucalgary.ca) is a professor at the Centre for Research on Microelectronics and Nanotech-nology, Technopole of Sousse, Tunisia. He is also with the iRadio Lab, Department of Electrical and Computer Engineering, Uni-versity of Calgary, Canada. His current research interests include the characterization, modeling, and linearization of radio-fre-quency power amplifiers and wireless transmitters as well as the design of microwave passive and active components/subsystems.

Milan Svanda (milan.svanda@fel.cvut.cz) is a research sci-entist in the Department of Electromagnetic Field at the Czech Technical University in Prague, Czech Republic. His current research interests include low-profile and wearable antennas, ultrahigh frequency radio-frequency identification, and antennas and sensors for special applications.

Fadhel M. Ghannouchi (fghannou@ucalgary.ca) is a pro-fessor, an Alberta Innovates Strategic Chair in Intelligent Radio-Frequency Technology, a Canada Research Chair in Green Radio Systems, and the director of the iRadio Lab at the Uni-versity of Calgary, Canada. His current research interests include microwave, wireless, and optical communications.

Tan-Phu Vuong (tan-phu.vuong@minatec.inpg.fr) is a pro-fessor at Phelma, the Grenole Institute of Technology, France. His current research areas are the design of small antennas and printed antennas for mobile, radio-frequency identification, and the design of passive and active millimeter-wave components.

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