Final Patch Antenna

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ABSTRACT

The concept of microstrip radiators was first proposed by Deschamps as early as

1953. The first practical antennas were developed in the early 1970¶s by Howell and Munson.

Since then, extensive research and development of microstrip antennas and arrays, exploiting

the new advantages such as light weight, low volume, low cost, low cost, compatible with

integrated circuits, etc., have led to the diversified applications and to the establishment of the

topic as a separate entity within the broad field of microwave antennas.

The Aim of this LRP is to study microstrip antennas, various Feed Techniques,

Radiation Mechanism, Broadbanding Patch Anntenna techniques, and Slot Antennas. It will

be a basic building block for the designing of the Fractal Antenna.



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MICROSTR IP PATCH ANTENNA

Introduction

1. In high performance aircrafts, spacecrafts, satellites, missiles and other aerospaceapplications where size, weight, performance, ease of installation and aerodynamics profile

are the constraints, a low or flat/conformal profile antenna is required. In recent years various

types of flat profile printed antennas have been developed such as Microstrip antenna (MSA),

strip line, slot antenna, cavity backed printed antenna and printed dipole antenna. When the

characteristics of these antenna types are compared, the micro strip antenna is found to be

more advantageous.

2. Microstrip antenna are conformable to planar or non planar surface, simple and

inexpensive to manufacture, cost effective compatible with MIC designs and when a

particular patch shape and excitation modes are selected, they are very versatile in terms of

resonant frequency, polarization, radiation patterns and impedance. Microstrip antennas have

several advantages compared to conventional microwave antennas and therefore have many

applications over the broad frequency range from 100MHz to 50GHz.

Characteristics desired in Antenna

3. An Antenna should exhibit the following characteristics

(a) Antenna should be Compact and small in size.

(b) It should be Conformal to the surface.

(c) It should be Light in Weight and Volume.

(d) It should give value of return loss less than 10 db for desired BW.

(e) It should have circular polarization.

(f) It should have high axial ratio BW.

(g) It should have high efficiency.

(h) It should have omnidirectional radiation pattern.

(j) It should have high gain.

4. R easons f or Using Microstrip Patch Antenna

(a) Light weight, low volume, low profile planar configurations which can be

made conformal.

(b) Low fabrication cost, readily amenable to mass production.

(c) Can be made thin, hence, they do not perturb the aerodynamics of host

aerospace vehicles.



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(d) The antennas can be easil mounted on missiles, rockets and satellites without

ma jor alterations.

(e) These antennas have low scatter ing cross section.

(f) Linear, circular (lef t hand or r ight hand) polar i ations are possi ble with simple

changes in the f eed positions.

(g) Dual f requency and dual polar i ation antennas can be easily made.

(h) No cavity back ing required.

( j) Can be easily integrated with microwave integrated circuits.

(k ) Microstr i p antennas are compati ble with modular designs (solid state devices

such as oscillators, amplif iers, var iable attenuators, switches, modulators, mi ers etc.

can be added directly to the antenna substrate boar d).

(l) Feed lines and matching networks are f abr icated simultaneously with the

antenna structure.

Patch Antenna

5. A microstr i p antenna in its simplest f orm consists of a radiating patch on one side of

a dielectr ic substrate and a ground plane on the other side. The top and side views of a

rectangular microstr i p antenna are shown in f ig 1. However other shapes, such as the square,

circular, tr iangular, semicircular, sectoral and annular r ing shapes as shown in f ig 2 are also

used.

6. R adiation f rom the microstr i p antenna can occur f rom the f r inging f ields between the

per i phery of the patch and the ground plane. The length L of the rectangular patch f or the

f undamental TM10 mode excitation is slightly smaller than /2, where is the wavelength in

the dielectr ic medium, which in terms of f ree space wavelength 0 is given as 0/eff ere eff is

the eff ective dielectr ic constant of a microstr i p line of width w. The value of eff is slightly

less than the dielectr ic constant of the substrate because the f r inging f ields f rom the patch to

Fi 1: A Schematic Diagram of Microstri Patch Antenna



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the ground plane are not conf ined in the dielectr ic only, but are also spread in the air. To

enhance the f r inging f ields f rom the patch, which account f or the radiation, the width w of the

patch is increased. The f r inging f ields are also enhanced by decreasing the dielectr ic constant

or by increasing the substrate thickness h.

7. Applications of Microstrip Antennas

(a) The telemetry and communications antennas on missiles need to be thin and

conf ormal and are of ten microstr i p antennas.

( b) R adar altimeters are small arrays of microstr i p radiators.

(c) Other aircraf t related applications include antennas f or telephone and satellitecommunications.

(d) Microstr i p arrays have been used f or satellite imaging systems.

(e) Patch antennas have been used on communication links between shi ps and

satellites.

(f) Smar t weapon systems use microstr i p antennas because of their thin prof ile.

(g) Pagers, the Global system f or mobile communication, and the global

positioning system (GPS) are ma jor users of microstr i p antennas.

Feed Techniq es

8. Microstr i p patch antennas can be f ed by a var iety of methods. These methods can be

classif ied into two categor ies- contacting and non-contacting. In the contacting method, the

R F power is f ed directly to the radiating patch using a connecting element such as a

microstr i p line. In the non-contacting scheme, electromagnetic f ield coupling is done to

transf er power between the microstr i p line and the radiating patch. The f our most popular

f eed techniques used are the micro str i p line, coaxial probe ( both contacting schemes),

aper ture coupling and proximity coupling ( both non-contacting schemes).

Fig 2: Different Shapes of Microstrip Patches



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(a) Microstrip Line Feed In this type of f eed technique, a conducting str i p

is connected directly to the edge of the microstr i p patch as shown in Figure 3. The

conducting str i p is smaller in width as compared to the patch and this k ind of f eed

arrangement has the advantage that the f eed can be etched on the same substrate to

provide a planar structure. The purpose of the inset cut in the patch is to match the

impedance of the f eed line to the patch without the need f or any additional matchingelement. This is achieved by proper ly controlling the inset position. Hence this is an

easy f eeding scheme, since it provides ease of f abr ication and simplicity in modeling

as well as impedance matching. However as the thickness of the dielectr ic substrate

being used, increases, sur f ace waves and spur ious f eed radiation also increases, which

hampers the bandwidth of the antenna. The f eed radiation also leads to undesired

cross polar i ed radiation.

( b) Coaxial Feed The Coaxial f eed or probe f eed is a very common technique

used f or f eeding Microstr i p patch antennas. As seen f rom Figure 4, the inner

conductor of the coaxial connector extends through thedielectr ic and is soldered to

the radiating patch, while the outer conductor is connected to the ground plane.The

main advantage of this type of f eeding scheme is that the f eed can be placed at any

desired location inside the patch in or der to match withits input impedance. This f eed

method is easy to f abr icate and has low spur ious radiation. However, its ma jor

disadvantage is that it provides narrow bandwidth and is diff icult to model since a

hole has to be dr illed in the substrate and the connector protrudes outside the ground

plane, thus not mak ing it completely planar f or thick substrates(h > 0.02ë o ). Also,

f or thicker substrates, the increased probe length makes the Input impedance more

inductive, leading to matching problems. It is seen above that f or a thick dielectr ic

substrate, which provides broad bandwidth, the microstr i p line f eed and the coaxial

Figure 3 : Microstrip Line Feed



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f eed suff er f rom numerous disadvantages. The non-contacting f eed techniques which

have been discussed below, solve these problems.

(c) Aperture Coupled Feed In this type of f eed technique, the radiating patch

and the microstr i p f eed line are separated by the ground plane as shown in Figure 5.

Coupling between the patch and the f eed line is made through a slot or an aper ture in

the ground plane. The coupling aper ture is usually centered under the patch, leading to

lower cross-polar i ation due to symmetry of the conf iguration. The amount of

coupling f rom the f eed line to the patch is determined by the shape, si e and location

of the aper ture. Since the ground plane separates the patch and the f eed line, spur ious

radiation is minimi ed. Generally, a high dielectr ic mater ial is used f or the bottom

substrate and a thick, low dielectr ic constant mater ial is used f or the top substrate to

optimi e radiation f rom the patch. The ma jor disadvantage of this f eed technique is

that it is diff icult to f abr icate due to multi ple layers, which also increases the antenna

thickness. This f eeding scheme also provides narrow bandwidth.

Figure 5: Aperture-coupled feed

Figure 4 : Probe fed Rectangular Microstrip Patch Antenna



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(d) Proximity Coupled Feed This type of f eed technique is also called as the

electromagnetic coupling scheme. As shown in Figure 6, two dielectr ic substrates are

used such that the f eed line is between the two substrates and the radiating patch is on

top of the upper substrate. The main advantage of this f eed technique is that it

eliminates spur ious f eed radiation and provides very high bandwidth (as high as 13%)

, due to overall increase in the thickness of the microstr i p patch antenna. This schemealso provides choices between two diff erent dielectr ic media, one f or the patch and

one f or the f eed line to optimi e the individual per f ormances. Matching can be

achieved by controlling the length of the f eed line and the width-to-line ratio of the

patch. The ma jor disadvantage of this f eed scheme is that it is diff icult to f abr icate

because of the two dielectr ic layers which need proper alignment. Also, there is an

increase in the overall thickness of the antenna.

Characteristics Microstrip

Line

Feed

Coaxial Feed Aperture

coupled Feed

Proximity

coupled Feed

Spurious feed

radiation

More More Less Minimum

Reliability Better Poor due to solder ing Good Good

Ease of fabrication Easy Solder ing & dr illing needed Alignment reqd Alignment reqd

Impedance

Matching

Easy Easy Easy Easy

BW(achieved by

Z matching)

2-5% 2-5% 2-5% 13%

Figure 6 : Proximity-coupled Feed

Table 1: Summarizes the characteristics of the different feed technique



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RADIATION MECHANISM

9. When the patch is exited by a feed, a charge is established on the undersize of the

patch is positively charged and the ground plane is negatively charged. The attractive forces

between these sets of charges tend to hold a large percentage of charge between the two

surfaces. However, the repulsive force between positive charges on the patch pushes some of this charge towards the edges, resulting in large charge density at the edges. These charges

are the source of fringing fields and the associated radiation. The movement of these charges

from the bottom of the patch, around its edges, to its top surface creates corresponding

current densities Jb and Jt at the bottom and top surface of the patch, respectively. For most

practical microstrip antenna, the height to width ratio is very small. Therefore the attractive

force between the charges dominates and most of the charge concentration and the current

flow remain underneath the patch.

10. A small amount of current flows around the edges of the patch to its top surface and is

responsible for weak magnetic field tangential to the edges. Hence we can make a simple

approximation that the tangential magnetic field is zero and one can place magnetic wall

around the periphery of the patch. This assumption has greater validity for thin substrates

with high dielectric constant. Also since the substrate used is very thin compared to wave

length, in the dielectric the field variations along the height can be considered to be constant

and the electric field nearly normal to the patch to the surface of the patch. Consequently, the

patch can be modeled as a cavity with electric walls on top and below and four magnetic

walls along the edges of the patch. Only TM modes are possible in the cavity. For the

fundamental TM10 mode, the length L should be slightly less than /2, where is the space

wavelength in the dielectric medium. Hence is equal to 0/ eff , where 0 is the free space

wavelength and eff is the dielectric constant of the patch. The fundamental TM10 mode

implies that the field varies one /2 cycle along the length, and there is no variation along thewidth of the patch. Along the width of the patch, the maximum voltage is and current is

minimum due to open end.

11. It may be observed from fig 7 that the vertical component of the electric field (E

field) at the two edges along the width are in opposite directions and hence cancel each other

in the broadside direction, whereas the horizontal components are in the same direction and

hence combine in the broadside direction. Therefore, the edges along the width are termed as

radiating edges. The fields due to the sinusoidal distribution along the length are known as

non radiating edges.



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12. Important Parameters

(a) The Resonance Frequency is given by

Where

( b) The Radiation Eff iciency is def ined as

Where Psp is the power radiated into space,

Ptotal the total input power is given as the sum of

Pc - the power dissi pated by conductor loss,

Pd - the power dissi pated by dielectr ic loss, and

Psw - the sur f ace-wave power.

Fig 7 : Top and side view of the patch showing f ield variation and fringing f ields



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(c) The Bandwidth is given as

(d) The Input Impedance is given as

(e) The Effective loss tangenteff is given as: eff = 1/Q T

(i) QT is the total antenna quality f actor and has been expressed

(ii) Q d represents the quality f actor of the dielectr ic and is given

where

r is the angular resonant f requency.

W T is the total energy stored in the patch at resonance. P d is the dielectr ic loss.

tan is the loss tangent of the dielectr ic.

(iii) Q c represents the quality f actor of the conductor and is given

Figure 8 : Equivalent Circuit of Patch Antenna for Input



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where P c is the conductor loss.

is the skin depth of the conductor.

h is the height of the substrate.

( iv

¡ ¢

represents the quality factor for radiation and is given as

Where P £ is the power radiated from the patch.

Substituting equations i, ii, iii, iv

Method of Anal sis

13. The analysis methods for microstrip antennas can be broadly divided into two groups.

(a) In the first group, the methods are based on equivalent magnetic current

distribution around the patch edges (similar to the slot antennas). There are three

popular analytical techniques:

(i) The Transmission Line Model In this the microstrip radiator

element is viewed as a transmission line resonator with no transverse field

variations (the field only varies along the length), and the radiation occurs

mainly from the fringing at the open ends. The patch is represented by two

slots that are spaced by the length of the resonator.

(ii) The Cavity Model In the cavity model, the region between the path

and the ground plane is treated as a cavity that is surrounded by magnetic

walls around the periphery and electric walls from top and bottom sides. Since

thin substrates are used, the field underneath the patch for regular shapes such

as rectangular, circular, triangular and sectoral shapes can be expresses as a

summation of the various resonant modes of the two dimension resonator.



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(iii) The MNM The MNM for analyzing the MSA is an extension of the

cavity model. In this method, the electromagnetic fields underneath the patch

and outside the patch are modeled separately. The patch is analyzed as a two

dimensional planner network, with a multiple number of ports located around

the periphery.

(b) In the second group, the methods are based on the electric current distribution

on the patch conductor and the ground plane (similar to dipole antennas, used in

conjunction with full wave simulation /numerical analysis methods). Numerical

methods have come into predominance and become more attractive with the advent of

fast digital computers. Some of the numerical methods for analyzing MSAs are listed

as follows:

(i) FDTD Method In this Technique, spatial as well as time grid for the

electric and magnetic fields are generated over which the solution is required.

The spatial discretizations along three Cartesian coordinates are taken to be

same.

(ii) FEM Finite Element Method) This method is suitable for

volumetric configurations. In this method, the region of interest is divided into

any number of finite surfaces or volume elements depending upon the planner

or volumetric structures to be analyzed. These discretized units, generally

referred to as finite elements, can be any well defined geometrical shapes.

(iii) Method of Moment In the MOM, the surface currents are used to

model the microstrip patch, and the volume polarizations currents in the

dielectric slab. An integral equation is formulated for the unknown currents on

the microstrip patches and the feed lines and their images in the ground plane.

The integral equations are transformed into algebraic equations that can be

easily solved using the computer, This method takes in to account the fringing

field outside the physical boundary of the two dimensional patch, thus

providing a more exact solution.

Broad banding patch antenna techni ues

14. Dual-frequency techniques for patch antennas .In principle, dual-frequency planar

antennas should operate with similar features, both in terms of radiation and impedance

matching, at two separate frequencies. Obtaining these features by using planar technologies

is not a straightforward matter; the simplest way to operate at dual frequencies is to use the

first resonance of the two orthogonal dimensions of the rectangular patch, i.e. the TMloo and

the TMolo modes. In this case, the frequency ratio is approximately equal to the ratio

between the two orthogonal sides of the patch. The obvious limitation of this approach is that

the two different frequencies excite two orthogonal polarizations. Anyway, this simple

method is very useful in low-cost short-range applications, where polarization requirements



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are not pressing. The above approach characterizes a first category of dual-frequency patch

antennas, which will be identified as

(a) Orthogonal mode dual f re uency patch antennas As mentioned before,

these antennas are characterized by two resonances with orthogonal polarizations.

These may be obtained, in the simplest case, by a rectangular patch. An interestingfeature of these antennas is their capability of simultaneous matching of the input

impedance at the two frequencies with a single feed structure (denoted by ³single-

point´ in Figure 9). This may be obtained with a probe-fed configuration, which is

displaced from the two principal axes of the patch. As demonstrated in, the

performance of this approach in terms of matching level and bandwidth is almost

equal to that of the same patch fed separately on the two orthogonal principal axes.

This provides the possibility of using the well-known design formula for standard

feeds. It is also worth noting that the simultaneous matching level for structures that

provide the same polarizations at the two frequencies is, in general, worse with

respect to the case relevant to orthogonal polarization. Single-feed dual matching may

be obtained by using slot coupling, in which the slot is inclined with respect to the

microstrip feed line. The required slot length and inclination angle can be

approximately obtained by projecting the slot onto the two orthogonal directions. The

two projections can be thought of as the length of two equivalent slots that excite the

patch at the two separate polarizations. The inclination of the slots may also be

adjusted, in order to compensate for error introduced by the matching stub, which is

designed to be a quarter of a wavelength for only one frequency. Orthogonal modes

may be excited by separated microstrips. This solution cannot provide flexibility in

designing the frequency ratio. The patch rim is composed of two intersecting portions

of circles of the same radius, with their center displaced by a distance that is designed

to have a given frequency ratio. Good isolation between the orthogonal ports can also

be obtained with electromagnetically coupled microstrip feed lines (27 dB).

(b) Multi-patch dual-f re uency antennas In these structures, the dual-

frequency behavior is obtained by means of multiple radiating elements, each of them

supporting strong currents and radiation at the resonance. This category includes

multi-layer stacked patches. These antennas operate with the same polarization at the

two frequencies, as well as with a dual polarization. The same multi layer structures

can also be used to broaden the bandwidth of a single-frequency antenna, when the

two frequencies are forced to be closely spaced. In this latter case, the lower patch can

be fed by a conventional arrangement and the upper patch by proximity coupling withthe lower patch. In order to avoid disappearance of the upper resonance, the sizes of

the two patches should be close, so that only a frequency ratio close to unity may be

obtained. A direct probe feed for the upper patch may also be used. In this case, the

probe passes through a clearance hole in the lower patch, and is electrically connected

to the upper patch. This kind of configuration insures one more degree of freedom

(the hole radius) in designing the optimum matching at the two frequencies, and

allows a wider range of the frequency ratio with respect to the structure in which the



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upper patch is electromagnetically coupled. In comparison with the resonant

frequencies of the two isolated patches, the frequency of the upper (smaller) patch

increases, and the frequency of the lower (larger) patch decreases. In any case, due to

the strong coupling between the two elements, simple design formulas cannot be

found, so that a full-wave analysis is, in general, required in the first phase of the

design. Multi-frequency antennas can also be obtained by printing more resonators onthe same substrate. Radar applications, such as SAR and multi-spectral scatter meters,

often require a large separation between the frequencies, So that the multi-resonator

structure must involve patches of very different sizes. It consists of a cross-shaped

patch for the lower frequency, and a sub array of four patches for the upper frequency.

The patches and the relevant feed networks for the two frequencies can be stacked on

two different substrates, thus obtaining two almost-independent antennas. This

arrangement has the advantage that the two sub-antennas can be designed almost

independently, provided that orthogonal polarizations at the two frequencies are

imposed. In this configuration, one should expect parallel plate mode excitation

between the slotted ground plane and the lower patch. This can produce spurious

radiation and coupling between the array transmit-receive modules.

(c) R eactively-loaded patch antennas The most popular technique for

obtaining a dual-frequency behavior is to introduce a reactive loading to a single

patch. The simplest way is to connect a stub to one radiating edge, in such away as to

introduce a further resonant length that is responsible for the second operating

frequency. This may easily be understood by resorting to the transmission-line model.

As is shown in Figure 1, other kinds of loading can be used, including notches, pins

and capacitors, and slots. The reactive-loading approach was first used in , where an

adjustable coaxial stub was employed. This structure may provide both tuning and

design of the frequency ratio in a simple manner; on the other hand, it is encumbering

and not well-suited for high frequencies. In a more practical configuration is

presented, in which the stub is constituted by a microstrip. Loading the radiating edge

with an inset or a spur-line (³notch loading´) is an alternative way to introduce a dual

frequency behavior that creates the same effect as the microstrip loading effect, with

the advantage of reduced size. However, both with stubs and notches, the frequency

ratio cannot be designed to be higher than 1.2 without introducing strong cross-

polarization levels or pattern distortion at the additional frequency. To obtain higher

values of the frequency ratio, different approaches have been proposed. There, the use

of pin diodes is proposed for changing the loading configuration, thus allowing

frequency agility. Very high values of the frequency ratio (4-5) can be obtained bymeans of two lumped capacitors, connected from the patch to the ground plane.

Another kind of reactive loading can be introduced by etching slots on the patch. The

slot loading allows for a strong modification of the resonant mode of a rectangular

patch, particularly when the slots are oriented to cut the current lines of the

unperturbed mode. In particular, as shown in the simultaneous use of slots and short-

circuit vias allows a frequency ratio of from 1.3 to 3, depending on the number of

vias. Other kinds of slot-loaded patches have been independently introduced and



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consist of a rectangular patch with two narrow slots etched close to and parallel to the

radiating edge. The same conf iguration has been investigated and extended to dual

polar i ation. More details are given in the next section. Bef ore proceeding f ur ther, it

is wor th noting that some or iginal dual-f requency structures cannot be easily f ramed

into the categor ies.

Fig 9 : various methods of broadbanding patch antenna



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Introduction to Slot Antenna

15. Slot antennas are popular omnidirectional microwave antennas. These antennas

feature omnidirectional gain around the azimuth with horizontal polarization. Waveguide slot

antennas, usually with an array of slots for higher gain, are used at frequencies from 2 to 24

GHz, while simple slotted-cylinder antennas are more common at the UHF and lower microwave frequencies where the size of a waveguide becomes unwieldy. The Alford slot is

an enhanced form of the slotted-cylinder antenna with somewhat higher gain. The concept of

microstrip slot antennas has evolved from slot antennas exited by a strip line. These types of

slot antennas have been extensively used. They have numerous promising features, but they

suffer from undesired modes such as parallel plate mode exited between the ground planes of

the strip line.

16. Microstrip slot antennas (MSAs) have the advantage of being able to produce

bidirectional and unidirectional radiation patterns with larger bandwidth. Strip and slot

combination offer an additional degree of freedom in the design of microstrip antennas. Acombination of strip conductors and slots arranged along the sides of a microstrip feed can

produce circularly polarized radiation. Antennas with desired polarization can be produced

and they are less sensitive to manufacturing tolerances than are microstrip patch antennas.

Annular slot antennas can be used as vehicular antenna. Tapered slot antenna produces end

fire radiation and finds a number of applications at millimeter wave frequencies.

17. Slots and Dipoles A thin slot in an infinite ground plane is the complement to a

dipole in free space. The slot have the same radiation pattern as a dipole with the same

dimensions as the slot, except that the E- and H-fields are swapped, as illustrated in Figure

the slot is a magnetic dipole rather than an electric dipole. As a result, the polarization is

rotated 90º, so that radiation from a vertical slot is polarized horizontally. For instance, a

vertical slot has the same pattern as a horizontal dipole of the same dimensions and we are able

to calculate the radiation pattern of a dipole. Thus, a longitudinal slot in the broad wall of a

waveguide radiates just like a dipole perpendicular to the slot.

18. Comparison of Slot Antenna with MSAs- Different shapes: MSAs are easy to

design and fabricate, and can have square, circular, triangular, semicircular, sectoral and

annular ring shapes and so on. Linear or circular polarization can be achieved by nearly

changing the feed position. Loading of the patch can be used to obtain dual-frequency

operation, circular polarization, change in the radiation pattern, and increase in bandwidth.

On the other hand, slot antennas can produce unidirectional or bidirectional radiation. Aninteresting feature of slot antenna is their greater bandwidth because of bidirectional

radiation. Slot antennas are less sensitive to manufacturing tolerances compared with patch

antenna.



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Feeds used in Slot Antenna

19. The var ious types of f eeding in slot antennas are-:

(a) Microstrip-fed Rectangular Slot Antennas A microstr i p slot antenna

compr ises a slot cut in the ground plane of the microstr i p line such that the slot is

perpendicular to the str i p conductor of the microstr i p line. The f ields of microstr i p

lines excite the slot. For eff icient excitation of the slot, the str i p conductor is either

shor t circuited through the dielectr ic substrate to the edge of slot.The microstr i p

excited rectangular slot antenna has the advantage of very low cross polar i ation as

compared to microstr i p patch antennas. Its drawback is the inherently bidirectional

radiation, which can be corrected by using a metallic cavity or a metallic ref lector at one end.

Fig 10 : Slots and D ipoles

f ig 11 : Microstrip fed rectangular slot antenna



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( b) Coplanar Waveguide Feed A coplanar waveguide (CPW) is the pref erred

txmn line f or microwave monolithic integrated circuits (MMICs). Both the CPW and

microstr i p antenna belong to the planner geometry, theref ore sometime it is desirable

to f eed antenna with CPW. It is shown in f ig. Using coplanar waveguides off ersthe

advantage of ease of integration with active devices due to their uni planar design,

eliminating the need f or vias. In the literature, some f eeding methods f or single-layer microstr i p antennas using coplanar waveguides, mainly inductive or capacitive

coupling via a rectangular slot, have been explored. Fur thermore, since the CPW

conductors are also used as the ground plane f or the microstr i p patch, the f eed

substrate used in conventional microstr i p aper ture coupling is no longer needed. Three

possi bilities with this excitation are as given below -:

(i) Inductive coupling-

(ii) Capacitive coupling

(iii) Coupling through annular slot

(c) This coupling arrangement is somewhat similar to the aper ture coupling; the

only diff erence between the two is that the slot in the ground plane is f ed by a

microstr i p line in aper ture coupled microstr i p antennas.Due to its low radiation losses

and easy inser tion of the active and passive components, the coplanar waveguide

(CPW) has been of ten used as an alternate to microstr i pline f or f eeding the pr inted

antennas. CPW f ed wide slot antennas have the advantages of wide bandwidth and

easy integration with monolithic microwave integrated circuit.

Fig12 :. CPW Feed with inductive, capacitive, and loop slot coupling.



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Selection of Substrate

20. The selection of a substrate material is a balance between the required electrical,

mechanical and environmental performance required by a design versus economic

constraints. When a dielectric substrate is selected, one is interested in a material with the

lowest tangent (tan ) available. The loss tangent is a metric of the quantity of electricalenergy which is converted to heat by a dielectric. The lowest possible loss tangent maximizes

the antenna efficiency. The relative dielectric constant r of the substrate determines the

physical size of a patch antenna. The larger the dielectric constant the smaller the element

size, but also the smaller the impedance, bandwidth and directivity and the surface wave loss

increases. The use of the substrates with higher dielectric constants also tightens fabrication

tolerances.The tolerance of the dielectric value is also of significant importance in

manufacturing yield.

21. Generally, dielectric constant r and loss tangent tan increase with temperature. In

space applications moisture outgassing produces a lower dielectric constant and loss tangent.

(a) Tef lon (Polytetrafluoroethylene - PTFE) has very desirable electrical qualities

but is not recommended for many space applications.

(b) R exolite is a very good material for space applications and has many desirable

mechanical properties. Rexolite is easily machined and its dielectric constant

remains stable up to 100 GHz.

(c) Noryl is suitable for many commercial microwave applications. It has a much

lower loss than FR4 and is relatively cost effective, but it is soft and melts at a

relatively low temperature which can create soldering complications, and sometimeshas unsuitable mechanical properties for some applications.

(d) FR4 is inexpensive and find use in many commercial applications below 1

GHz. The material can be used for some wireless applications, but great care must

be taken to budget and minimize the losses when it is used as a substrate of PTFE

and Epoxy glass (FR4) which has the desirable properties of FR4 with lower loss.

(e) Alumina has desirable microwave properties for applications which require a

relatively high dielectric constant r ~ 10.0 and low loss tangent. Its drawbacks are

the difficulty involved in machining it and its brittleness. Alumina has good thermal

conductivity and in some aerospace applications it more readily dissipates heat and

remains cooler than other common microwave substrates. In some missile

applications where high temperatures may compromise solder joints alumina is a

viable option for the dissipation of heat. Alumina¶s dielectric constant is very

sensitive to the processing used to produce the alumina.



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Material r Tan

Tef lon (PTFE) 2.1 0.0005

R exolite 1422 2.55 0.0007

Noryl 2.6 0.0011

FR 4 4.1 0.02

Alumina 9.8 0.0003

Microwave Coaxial Connector

22. For high f requency operation the average circumf erence of a coaxial cable must be

limited to about one wavelength, in or der to reduce multimodal propagation and eliminate

erotic ref lection coeff icients, power losses and signal distor tion. The standar di ation of

coaxial connectors dur ing Wor ld War II was mandatory f or microwave operation to maintain

a low ref lection coeff icient or a low voltage standing wave ratio (VSWR ). Since that time

many modif ications and new designs f or microwave connectors have been proposed and

developed. Seven types of microwave coaxial connectors are descr i bed below.

(a) APC-3.5 The APC-3.5 (Amphenol Precision Connector-3.5mm) was

or iginally developed by Hewlett-Packar d. The connector provides the repeatable

connections and has very low voltage standing-wave ratio (VSWR ). Either the male

or f emale end of this 50 connector can mate with the opposite type of SMA type

connector. The APC-3.5 connector can work at f requencies up to 34 GHz.

( b) APC-7 The APC-7 (Amphenol Precision Connector-7mm) was also

developed by Hewlett-Packar d in the mid 1960s. The connector provides a coupling

mechanism without male or f emale distinction and is the most repeatable connecting

device used f or very accurate 50 measurement applications. Its VSWR is extremely

low, in the range of 1.02 to 18 GHz.

Figure 13 : APC-7 Connector

Table 2 : Dielectric and Loss tangent for different materials



21

(c) BNC The BNC (Bayonet Navy Connector) was originally designed for

military system applications during World War II. The connector operates very well at

frequencies up to about 4GHz, beyond that it tends to radiate electromagnetic energy.

The BNC can accept flexible cables with diameters of up to 6.35mm (0.25inches) and

characteristic impedance of 50 to 75. It is now the most commonly used connector

for frequencies under 1 GHz.

Figure 14 : BNC Connector

(d) SMA The SMA (Sub-Miniature A) was originally by Bendix Scintilla

Corporation. The main application of SMA connector is on component for microwave

systems. The connector is seldom used above 24 GHz because of higher order modes.

Figure

15 : SM

A Connector

(e) SMC The SMC (Sub Miniature C) is a 50 connector that is smaller than the

SMA. The connector can accept flexible cables with diameters of up to 3.17mm

(0.125 inches) for a frequency range of up to 7 GHz.

Figure 16 : SMC Connector

(f) TNC The TNC (Threaded Navy Connector) is merely a thread BNC. The

function of thread is to stop radiation at higher frequencies, so that the connector can

work at frequencies up to 12GHz.



22

Figure 17 : TNC Connector

(g) Type N The Type N (Navy) connector was originally designed or military

systems during World War II and is the most popular measurement connector for the

frequency range of 1 to 18GHz. It is 50 or 75 connector and its VSWR is extremely

low, less than 1.02.

Figure 18 : Type N Connector

Conclusion

23. In this LRP, the basic properties of Microstrop Patch Antennas have been covered. A basic set of specifications has been defined allowing the reader to understand and write a set

of requirements for a specific application. A couple of patch antenna feeding techniques have

been described. Besides the patch antennas covered here, many more design options and

different implementations are possible It will be Further utilized in the design of Fractal

Antennas for various Applications.



23

REFERENCES

1. Constantine A. Balanis, ³ Antenna t he ¤ ry analy¥ i ¥ and d e¥ ig n´, 2nd edition, JohnWiley & sons, Inc, 1997

2. D. Orban and G.J.K. Moernaut, The Basics of Patch Antennas,

www.orbanmicrowave.com.

3. http:// www.wikipedia.com

4. S. Maci and G. Bif J i Gent ili , Dual-Fre uency Patch Antennas, Department of Electronic Engineering, University of Florence, Via S. Marta 3, 50135, Florence, Italy

5. Vibha Rani Gupta And Nisha Gupta, Gain And Bandwidth Enhancement In

Compact Microstrip Antenna, Birla Institute Of Technology, Mesra, Ranchi

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