FIELD EFFECT TRANSISTORS - TDL

FAST TRANSIENT SWITCHING IN BIPOLAR JUNCTION

TRANSISTORS AND METAL-OXIDE-SEMICONDUCTOR

FIELD EFFECT TRANSISTORS

by

STEVEN MENHART, B.Sc.

A THESIS

IN

ELECTRICAL ENGINEERING

Submitted to the Graduate Faculty of Texas Tech University in

Partial Fulfillment of the Requirements for

the Degree of

MASTER OF SCIENCE

IN

ELECTRICAL ENGINEERING

Approved

Accepted

December, 1985

<? ACKNOWLEDGMENTS

I would like to thank Dr. w. M. Portnoy for

his professional expertise and overall guidance

during the course of this work. Also, I would

like to thank Mr. John Power of Los Alamos

National Laboratory, New Mexico, for his techni

cal advice, which helped solve many problems. I

am also indebted to Drs. J. F. Walkup, E. Jobe,

and T. F. Krile for agreeing to serve on my

committee.

I would especially like to thank my parents,

Hans and Christine Menhart, for their encourage

ment and financial support, without which this

work would not have been possible.

ii

TABLE OF CONTENTS

Page

ACKNOWLEDGMENTS ii

LIST OF TABLES V

LIST OF FIGURES vi

I. INTRODUCTION 1

11. THEORY OF THE MEASUREMENTS 3

III. EXPERIMENTAL PROCEDURES 8

Test Devices 8

Test Circuits 10

Repetitive Triggering 22

Measurements 24

IV. EXPERIMENTAL RESULTS 3 0

Base Drive Effects 30

Circuit Effects 43

Base-Triggered Bipolar Transistor

Measurements 49

The MOS Transistor 54

Collector-Triggered Measurements 58

Repetitive Measurements 64 v. CONCLUSIONS 70

LIST OF REFERENCES 71

111

APPENDICES

A. Theoretical Load Voltage Calculation for Constant dv/dt Switch 7 2

B." Photographs of the Representative Behavior of a Base-Triggered RS3500, as a Function of Load Resistance and Capacitance 74

C. Photographs of the Representative Behavior of a Gate-Triggered BS170, as a Function

of Load Resistance and Capacitance 92

D. Specifications for the RS350 Transistor liO

E. Specifications for the 2N2222 Transistor 113

F. Specifications for the BS170 Transistor 118

iv

LIST OF TABLES

Page

Table 1.

Table 2.

Table 3.

Table 4.

Table 5.

Fall Times and Peak voltages for the Output Pulses Obtained with the Mercury-Wetted Relay as a Substitute for the TUT 44

Theoretical Peak Voltages Obtained with the Mercury-Wetted Relay as a Substitute for the TUT 47

Fall Times and Peak Voltages for the Output Pulses of the Base-Triggered RS3500 50

Fall Times and Peak Voltages for the Output Pulses of the Base-Triggered 2N2222 51

Fall Time and Peak voltages for the Output Pulses of the Gate-Triggered BS170 56

LIST OF FIGURES

Page

Figure 1. Representative characteristic curves for a bipolar transistor 4

Figure 2. Emitter geometries 9

Figure 3. Circuit schematic for base-triggered measurements on the RS3500 and the 2N2222 11

Figure 4. Circuit schematic for gate-triggered measurements on the BS170 12

Figure 5. Circuit schematic for the collector-triggered measurements on the RS3500 and the 2N2222 14

Figure 6. Circuit schematic for the drain-triggered measurements on the BS170 15

Figure 7. Physical layouts of the RS3500 circuits 17

Figure 8. Physical layouts of the 2N2222

circuits 18

Figure 9. Physical layouts of the BS170 circuits 19

Figure 10. Typical pulse obtained from the line pulser into a matched 50 load 21

Figure 11. Circuit schematic for the RS3500 repetitive collector-triggered measurements 25

Figure 12. Circuit schematic for the constant-base drive measurements on the RS3500 31

Figure 13. Base current, collector current and collector-emitter voltage curves: V is 250 V and the load is 500 U 34

VI

Figure 14. Base current, collector current and collector-emitter voltage curves: v^c is 250 V and the load is 47 k« 37

Figure 15. Base current, collector current and collector-emitter voltage curves: VQQ is 100 V and the load is 47 kfl 41

Figure 16. Gate trigger pulse 57

Figure 17. Circuit schematic illustrating the simplified form of the collector-triggered circuit 59

Figure 18. Collector-triggered RS3500 waveforms; the amplitude of the trigger pulse is 750 V, CL is 20 pF and RL is 50 Q 61

Figure 19. Collector-triggered RS3500 waveforms; the amplitude of the trigger pulse is 750 V, CL is 20 pF and RL is 480 « 62

Figure 20. Collector-triggered RS3500 waveforms; the amplitude of the trigger pulse is 750 V, CL is 20 pF and RL is 4.8 k« 63

Figure 21. The 750 V trigger pulse used for repetitive collector-triggering 67

Figure 22. Observed load voltage for a RS3500 device repetitively triggered at the collector 69

Vll

CHAPTER I

INTRODUCTION

This work was conducted to investigate the effects of

load resistance and capacitance on the fast transient

switching performance of both bipolar junction transistors

(BJTs) and metal-oxide-semiconductor field effect transis

tors (MOSFETs). As a consequence of this work, it was

hoped to determine the suitability and reliability of these

devices as solid state switches in pulse forming circuits,

particularly Marx banks (sometimes called generators).

These networks essentially act as voltage multipliers for

pulsed voltages. The fundamental principle is to charge

several capacitors in parallel, then switch them into a

series configuration. The output voltage then becomes

equal to the sum of the voltages across the capacitors [IJ.

The width of the output pulse is determined by the values

of resistors and capacitors used; the pulse rise time, by

the switches used and by circuit parasites. Obviously, in

order to obtain rapid pulse rise times, fast switches are

required.

Three devices were investigated: the Raytheon RS3500

avalanche transistor, the 2N2222 (obtained from TRW as

samples from a single wafer) and the Siliconix BS170. The

first two devices are BJTs; the latter is a MOSFET. All

three devices were tested in two modes: based-triggered and

collector-triggered by way of an overvoltage (in a Marx

bank utilizing solid state switches, the first stage

requires a trigger, e.g., base-trigger; the latter stages

are usually switched by means of an overvoltage). Testing

was performed single-shot and repetitively for four samples

of the RS3500.

CHAPTER II

THEORY OF THE MEASUREMENTS

Figure 1 illustrates a representative family of

characteristic curves for an n-p-n bipolar transistor.

Four breakdown voltages are shown on the v^- axis: BV^- <

®^CER ^ ^^CEX ^ ®^CBO* ®^CEO ^^ ^^® open-base collector-

emitter breakdown voltage; BV _-, is the breakdown voltage

with the base tied to the emitter through a resistance; and

BVp„^, with the base-emitter junction reverse biased.

BV QQ is the open emitter collector-base breakdown voltage.

Associated with the base-emitter junction is a parasitic

resistance which shunts the junction. The BV breakdown

voltage is established when the collector-base leakage

current becomes high enough to create a voltage drop across

the shunting resistor, which is sufficient to forward bias

the base-emitter junction. (For small values of R, in the

order of the internal shunt resistance, BV ^ ^ will be near

BV-_^; this equality was in fact observed in this work when CBO

R was around 50 J2.)

Normally, the transistor is operated along a load

line, such as A, within the safe operating region bounded

by the saturated collector voltage curve, the BV^^^ curve,

and some power limited boundary such as the dashed line in

the figure. Such conventional switching in which the

^^CEO ^^CER ^^CEX CBO

Figure 1. Representative characteristic curves for a bipolar transistor.

device behaves normally as a transistor can be relatively

fast, particularly if the transistor is kept out of

saturation. However, the total voltage switched is limited

by BVp_Q. If the transistor is biased at BV , and a load

line such as B is used, the transistor will switch from its

quiescent operating point (the intersection of the load

line and the BV curve) to a second stable point, the

intersection of the load line and the BV___ curve. This CEO

switching is enhanced by avalanche effects and is very

fast. Here also, the total voltage switched is limited,

being the difference between BV-„ and BV , it is likely

that the above phenomenon is dependent upon the base

current, i.e., for base currents less than those corre

sponding to the intersection of the load line, and the

BV-_^ curve the collector-emitter voltage will probably

remain at BV^„-.. For base currents greater than this CEO

initial value, it should be possible to drive the transis

tor directly into a region of low collector-emitter

voltage.

There is a switching mode, however, which is not only

fast, but large in voltage swing. If the base is open

circuited and V__ is increased, the transistor quiescent

point will move in a stable fashion along the BV^^^ curve

until it reaches a point at which an instability occurs.

At this point (P in Figure 1), very rapid switching occurs

to a low voltage state in which the current is limited only

by external circuit resistance. This instability is also

known as second breakdown, which is associated with high

collector-emitter voltages and collector currents f2]. It

is usually a damaging event unless the total power dis

sipated in the resistance is kept small. Using a dynamic

load line such as D in a circuit which contains small

stored energy, say in a capacitor, will then permit

extremely fast switching of reasonably high voltages with

reasonable safety. Note that the load line, D, must lie

above P for such operation. At higher resistances (load

line B), the transistor will switch to a stable point on

the BV__^ curve. Even at lower resistances (load line C), CEO

if the load line lies below P, although close to it,

switching will occur to the intermediate voltage BV even

if the intermediate point is not stable. In this case, the

output will consist of a double pulse with a low amplitude

front edge. This phenomenon is probably subject to the

same base current constraints discussed earlier.

Such behavior suggests a procedure for testing the

performance of a bipolar transistor under fast, high

voltage switching conditions. Switching speed can be

examined as a function of the load line, and vulnerability,

as a function of stored energy. In this work, the point P

was determined for the RS3500 on a curve tracer by setting

the bias supply to 250 V (just below BV^^^) and varying the

collector resistance; under these bias conditions, P

corresponded to a load line of around 4000 Q. Test

conditions were then established for a logarithmic sequence

of resistances: 47, 470, 4700 and 47,000 12. Measurements

were also made at 4.7 a, but there was considerable ringing

in the output pulse at this load resistance. Stored energy

was controlled by using a sequence of capacitance values:

20, 200, 2000 and 20,000 pF. To permit a direct comparison

of performance between devices, these values of resistance

and capacitance were used in every measurement, on each

device, and during collector-triggering as well.

CHAPTER III

EXPERIMENTAL PROCEDURES

Test Devices

The RS3500 and 2N2222 are bipolar transistors, the

RS3 5 00 having the epitaxial structure and gold-doping of a

power transistor, although smaller in size. The 2N2222 is

a small signal device. Figure 2 illustrates the emitter

geometries of both the RS3500 and the 2N2222 (the center

connection is the emitter in both cases). The avalanche

device is capable of dissipating approximately 2.25 times

more power than the 2N2222 for a given case temperature.

The large emitter-based periphery of the RS3500 is typical

for power devices, to ensure uniform emitter current flow.

The BS170 utilizes metal oxide technology and is an

n-channel enhancement device. MOSFETs are majority carrier

semiconductor devices and their construction and principles

of operation are fundamentally different from those of

traditional bipolar transistors, which are minority carrier

semiconductors. A bipolar device is current controlled,

whereas a MOSFET is a voltage controlled device and conse

quently MOSFETs are often modeled as a voltage controlled

resistor. The hermetic package of the BS170 makes it a

physically robust device.

8

b.

Figure 2. Emitter geometries for: (a) the RS3500 (MdQnifica-tion 102X); (b) the 2N222: (Magnification 154X).

10

Test Circuits

The design of the test circuits for base-triggering

was constrained by the measurements to be performed: the

dynamic load line must correspond to the load resistance

and the energy stored in the capacitor must be dissipated

in the load following each trigger event. Figure 3 is a

schematic of the circuit ultimately used for the base-

triggered measurements on the RS3500 and the 2N2222; both

circuits are identical except for the collector supply

(V^^), which is 250 V for the RS3500 and 100 V for the

2N2222. Figure 4 illustrates the circuit schematic for the

gate-triggered measurements on the BS170; it is essentially

the same as that of Figure 3, except that there is no

negative gate bias and the drain supply (Vj. .) is 100 V.

The operation of both circuits may be understood by

considering the circuit of Figure 3. The load capacitance,

C-./ charges up to the supply voltage through R^ (100 kiJ),

the collector resistor, and the load resistor, R_. When a LI

trigger voltage sufficient to forward bias the base-emitter

junction of the bipolar device (approximately 0.7 V) or

greater than the gate-source threshold voltage for the

MOSFET (approximately 2.3 V), is applied, the transistor

switches along a load line established by R^, as R is

isolated by a 10 uH inductor. The transistor will remain

turned on until C has discharged, since once unstable

switching has occurred it ceases only when the energy

11

V cc 0

R^ < 100 Kfi

10 yH

n 47 Q

R,

-2 V

•oVj^(t)

OSCILLOSCOPE (50 ^)

Figure 3. Circuit schematic for base-triggered measurements on the RS3500 and the 2N2222. Vcc is 250 V for the RS3500 and 100 V for the 2N2222.

12

n

(I

100 kfi

10 yH

47

T UT

R,

t» Vj^(t)

OSCILLOSCOPE (50 n)

Figure 4. Circuit schematic for gate-triggered measurements on the"BSl70.

13

stored in C. has been dissipated; i.e., the collector

current falls below some holding current value.

The collector-triggered (or drain-triggered) measure-

meats involve a much higher voltage (from 400 to 1000 v

peak) pulse at the collector. Figures 5 and 6 are schema

tics of the collector-triggered circuits for the RS3500 and

2N2222 and the drain-triggered circuit for the BS170,

respectively. The circuits for the bipolar transistors

differ only in the value of collector voltage (V^^) used,

which are the same as the respective base-triggered

circuits. Again, v for the BS170 is 100 v.

The load capacitor is charged through R and a second

100 ka resistor; these are isolated from the trigger pulse

by 10 uH inductors. Several base-emitter (or drain-gate)

bias configurations were considered, but the shorted base

(or gate) was chosen because it is the configuration used

for overvoltage collector-triggering in a Marx bank. A high

voltage pulse summed with the capacitor voltage was the

collector (or drain) trigger. The load resistor R was

located in the emitter or source circuit.

Several arrangements of the load were considered to

minimize stray inductance; all of these involved configura

tions of resistors to achieve the required value of load

resistor in such a manner that the individual magnetic

fields created by the current flow in each resistor

combined in such a way that they effectively cancelled.

14

C > 100 k^

10 yH

TUT

10 yH

R,

V^(t) ^

OSCILLOSCOPE

100 KQ

(50 fi)

Figure 5. Circuit schematic for the collector-triggered measurements on the RS3500 and the 2N2222. Vcc is 250 V for the RS3500 and 100 V for the 2N2222. The trigger pulse has a magnitude of three times VQQ,

^DD (100 V) 15

100 k^

10 yH

TUT

10 yH

V^it) CK

OSCILLOSCOPE (50 fi)

100 K^

Figure 6. Circuit schematic for the drain-triggered measurements on the BS170.

16

resulting in a low inductance configuration. One arrange

ment consisted of a parallel array of 1/8 W composition

resistors. This configuration was investigated; however,

it did not produce the quality of output pulse obtained

with a single 1/4 W composition resistor and hence was not

pursued further. A second configuration was an attempt to

construct what is referred to as a 'current viewing

resistor' (CVR), This type of resistor has a coaxial form;

those fabricated consisted of five resistors connected in

parallel, directly at one end, and by means of a circular

ring at the other. When incorporated into a circuit, this

configuration is arranged such that the return current path

is through the center of the ring connecting the resistors

together. It was hoped that this would yield a configura

tion with reduced inductance. Fall-time measurements did

not confirm this and hence it was concluded that a lower

inductance configuration other than a single 1/4 composi

tion resistor could not be obtained.

The principal considerations in the physical layout of

the circuits were parasitics and mismatches. All external

connections were made via coaxial lines and connectors.

The circuits themselves were fabricated, as compactly as

possible, from double-sided, copper-clad laminate; a large

ground plane was employed, to which the extensions of the

coaxial connectors were soldered. The physical layouts of

the circuits are illustrated in Figure 7, 8 and 9.

1 /

a.

b.

Figure 7. Physical layouts of the RS3500 circuits: (a) base triagered; (b) collector-triggered.

a.

b.

Figure 8. Physical iayouts of the 2N2222 circuits: (a) base triQgered; (b) collector-triggered.

a.

b.

Figure 9 1 layouts of the :ircuits: (a) gats-Phvsica.

^ ' ' ""-^^ (b) drain-triggered; VD) '-^-^ triggered.

20

Initially, base-triggered measurements utilized a

HP-214A pulse generator as the trigger source. However,

this was superseded by a line pulser which produced a more

sharply defined rectangular pulse with decreased rise and

fall times (approximately 1 ns) and constant amplitude as

compared to the trapezoidal pulse obtained from the pulse

generator. The line pulser consisted of approximately one

meter of RG-58A/U coaxial cable in series with a Magnecraft

Model W134MIP-3 mercury wetted relay and delivered a 10 ns

pulse when the relay was closed. The line was charged to

40 V through a 1.5 Mfi resistor and delivered a 20 v

rectangular pulse to a matched 50 fl load. This pulse is

shown in Figure 10. The cable pulser was used in all base-

and gate-triggered measurements; however, mismatch at the

base and gate resulted in considerable ringing. A 47 Q

resistor at the gate to the source reduced the ringing in

the MOSFET measurements, but very little improvement could

be obtained under any conditions at the base.

This line pulser was also used in the collector and

drain-triggered measurements, except that it was charged to

1500 V for the RS3500 and 600 V for the 2N2222 and the

BS170. Under these conditions, 750 V and 300 V pulses were

provided for summing with the respective voltage stored

across the capacitor in the three measurements. The

capacitor was charged in each case to v or V .

0 1

Figure 10. Typical pulse obtained from the line pulser into a matched 50 load; the line was charged to 40 V. The vertical and horizontal scales are 5 V and 10 ns per large division, respectively.

22

The coaxial cable from which the line pulser is

constructed is essentially a transmission line consisting

of distributed series inductance and parallel capacitance.

The pulse width is determined by the time required for the

voltage wave to propagate the length of the cable, which in

turn is governed by the physical dimensions of and the

materials from which the cable is manufactured. Pulse

width is given by x^ = 2ZQC , where ZQ is the character

istic impedance and C is the total capacitance of the

line.

Repetitive Triggering

Initial repetitive measurements were performed by

rep-rating the relay coil directly from a pulse generator.

However, excessive relay bounce did not permit the con

tinued use of this simple arrangement. It was determined

that a command charging scheme was required; i.e., the

bounce itself could not be eliminated. However, its

manifestations could be eliminated. The command charging

system operated in the following manner: while the relay

contacts were open, the command charging element was

activated, causing the line to be charged; before the relay

contacts were closed, the command charging element discon

nected the line. Although the relay contacts continued to

bounce, a second voltage spike was not produced as the line

was discharged and recharge was inhibited.

23

Several command charging elements were considered,

including a series combination of SCRs; these were found to

self-trigger as a result of the high dv/dt produced when

the relay contacts closed? this caused the line to recharge

in an uncontrolled fashion. High voltage MOSFETs were also

considered. However, their leakage current proved too

great, i.e., the leakage current was sufficient to charge

the line to a voltage such that a second voltage spike of

some 100 V was observed approximately 30 us after the

initial discharge. It was determined that a solid state

switch could not be used and that a second relay, operating

ISO** out of phase with the first, would be most effective.

The repetition rate at which a relay can be operated

is principally governed by the L/R time constant of the

drive circuit. The inductance of a W134MIP-3 relay coil is

fixed at approximately 230 mH; however, its overall

resistance, R, can be increased by adding external series

resistance. Reducing the L/R time constant results in a

more rapid rise of current through the relay coil, which

reduces the delay between the application of a pulse to the

relay coil and the contacts closing. However, adding

series resistance reduces the magnitude of the coil

voltage. In order to maintain the nominal voltage across

the relay coils, these were driven by placing them in the

drain circuit of a common source FET circuit. In this

manner, by operating the MOSFETs at the rated voltage

24

(100 V), an additional 4.7 kfi of resistance could be added

in series with the 1400 ft internal resistance of the coil.

This reduced the L/R time constant by 77%, enabling the

maximum rep-rate to be increased to 350 Hz.

The complete circuit used for the repetitive measure

ments is illustrated in Figure 11. The relative timing of

the relays was assured by adjusting the respective FET gate

drives, which were obtained from pulse generators.

Measurements

Switching waveforms were displayed on a Tektronix

Model 7834 400 MHz storage oscilloscope, using a 600 MHz

7A19 plug-in amplifier. The trigger source was a 7B80 time

base. A storage oscilloscope was used rather than a

sampled measurement for several reasons; first, because

single-shot testing was desirable, at least initially, in

order not to stress the devices and also because jitter in

the trigger pulse sequence introduced time spread in the

sampled waveform display. Single-shot measurements on the

7834/7A19 combination, which has a specified rise time of

0.9 ns (although the results obtained here suggest that

this rise time is conservative), were compared with and

found to agree with sampled measurements on a base-

triggered RS3500, so that the system rise time was con

sidered adequate to obtain an accurate representation of

the phenomenon occurring.

o o in

25

o o CO Q)

tn u

 0 0 ) 0 : CQ > 0)

-H e •P 4-> 0) •H -H J-i 3 4J 13 U 0) CO M P i (TJ

•H 0) (U o M e

26

Several different procedures were considered for

obtaining the waveforms. A current coil produces a voltage

proportional to the current it is measuring. A current is

induced in the current coil as a result of magnetic

coupling between the current to be measured and current

coil. Consequently, current transformers offer the

advantage of being able to measure currents (and hence

voltage across a resistive load) without any physical

connection to the circuit, eliminating mismatch and loading

considerations. Unfortunately, this contrivance is, at

present, limited to a risetime response of 2 ns or greater,

which is inadequate for the sub-nanosecond fall-times

inherent in this measurement.

Having determined that a physical connection was

required, source-load matching became the primary consid

eration. When frequency components of the order of 500 MHz

are involved (as they are here), slight mismatches can

result in reflections and waveform distortion which ulti

mately yields a waveform that conveys dubious information--

if any. Since the bandwidth constraint necessitates the

use of a 50 ft input impedance plug-in amplifier, it was

decided to observe the voltage across the load by means of

direct coupling via 50 ft coaxial cable, utilizing 50 ft

attenuation at the oscilloscope input where necessary.

Initial measurements used this scheme, which produced

very "clean" waveforms without any ringing being evident.

27

However, it was determined that this scheme severely

restricted the amount of information obtainable owing to

loading of the high resistance load values by the 50 ft

parallel connection. Hence, this scheme was ultimately

considered impracticable.

A Tektronix P6057 1.4 GHz lOOX probe was considered as

a means of measuring the voltage. Certainly the frequency

response is more than adequate and, with an effective

impedance of 5 kft, it could be used across resistors of

4.8 kft, or less, without significant loading. However, it

could not be used across loads greater than 5 kft. This

limitation aside, the probe proved unsatisfactory owing to

excessive ringing of the observed pulse. Ringing could not

be eliminated even by connecting the ground-case of the

probe directly to the circuit board ground plane, i.e., by

the elimination of the probe grounding lead. As a result

of the above problems and the fact that a fast, high

voltage, high impedance probe is simply not commercially

available, the load voltage could not be measured directly.

A voltage divider, using two resistors in series as

the load, was finally considered. Inasmuch as direct

measurement using coaxial cable provided the cleanest

pulses, it was decided to use this scheme by making one of

the load resistors several ohms and to use it as a scaling

resistor across which the voltage could be observed. In

this manner a scaled version of the total load voltage was

28

obtained. Various values of the scaling resistor were

examined; unfortunately, all produced waveforms which

exhibited severe ringing.

At this point it was decided to use the plug-in

amplifier input resistance as the scaling resistor, i.e.,

to connect the oscilloscope in series with the load

resistor. The oscilloscope was used alone for load values

of 50 ft (with 50 ft Pasternack 1.5 GHz attenuators where

necessary). This scheme produced waveforms with essen

tially zero ringing; consequently, it was decided to use

this method, mutatis mutandis, for the collector-triggered

measurements.

The adoption of this scheme revealed what finally

appeared as a formidable problem. During the course of the

measurements, it was observed that the output voltage for

the higher values of load was considerably greater than its

predicted value. Because the oscilloscope impedance is

fixed at 50 ft, the remaining load impedance in that case

must be less than its nominal value. This is probably

attributable to the parasitic capacitance shunting the

load and all resistors at sufficiently high frequencies

[31.

In order to minimize the p a r a s i t i c capacitance, a l l

loads were constructed of four 1/4 w res i s tors connected in

s e r i e s (except for the 50 ft l o a d ) . In t h i s manner, the

p a r a s i t i c capac i tance of the load was reduced by a factor

29

of four, which considerably reduced the peak overshoot

voltage. No series inductance was introduced by this

arrangement; measurements made with up to ten resistors in

series, which was considered as a limiting case, provided

the same results as those obtained with four.

CHAPTER IV

EXPERIMENTAL RESULTS

Base Drive Effects

Preliminary measurements raised the question of the

effect of the magnitude of the base drive on the switching

behavior of the bipolar transistors. No intermediate

voltage levels were observed at any of the high resistance

loads. It was not clear if the dynamic load line was

affected by parasitic capacitance shunting the load

resistance, or if the high voltage at the base drove the

devices into saturation, obscuring the unstable switching

behavior. In order to determine the effects of base

current upon the switching performance of the avalanche

transistor when operated in the BV regime, a constant

base current drive, pulsed-collector supply circuit was

constructed. Figure 12 shows a schematic of the circuit.

The pulsed-collector supply was necessary in order to avoid

device damage or failure owing to excessive power dissipa

tion. The collector supply begins to turn off 375 us after

the leading edge of the base current pulse, which was 50 us

wide. The collector supply is connected to the load

resistor through a normally closed mercury-wetted reed

relay (Magnecraft W134MIP-3); the coil of this is activated

by means of an ECG 955M timer, which results in the relay

30

V^^ (250 V) , CC 15 kft

100 V 0

100 ft

IRF5

.47 ft

31

220 Kft

50 uF

0.01 yF

(. -2 V

PULSE GENERA

TOR

EXTERNAL TRIGGER

DELAYING INPUT EXTERNAL

TRIGGER OUTPUT

Figure 12. Circuit schematic for the constant-base drive measurements on the RS35O0. Ml and M2 represent Pearson current transformers used to the base and collector currents, respectively.

32

contacts opening. The relative timing between the base

current pulse and the collector supply is adjusted by using

two pulse generators and adjusting the delays between their

outputs. When the timer (commonly known as a 555) is

triggered, a pulse approximately 15 s long is applied to

the coil of the relay, during which time the collector

supply is turned off; the supply is simply disconnected and

reconnected manually.

The IRF 520 MOSFET provides a current pulse of

constant amplitude to the base of the transistor under test

(TUT). The amplitude of this pulse is varied by changing

the drain supply voltage, Vj ^ , up to a maximum of 100 V,

providing pulses of up to 1 A peak current. Very small

base currents (Ig < 1 mA) required adjustment of both V

and the amplitude of the gate trigger pulse in order to

avoid an overshoot of the leading edge of the current

pulse.

It was decided to use the same sequence of load

resistors which was established for the base and collector-

triggered measurements. However, use of a 47 ft load

resistor resulted in catastrophic device failure on the

first attempt to switch the device; consequently, values of

500 ft, 4.7 kft and 47 kft were used. (The high currents

obtained with a 47 ft load melted the chip connections.)

Three values of collector supply were used, viz., 250 v,

160 V and 90 v, the latter being within the region bounded

33

by the BV curve. Upon the application of a base current

pulse to the RS3500 device, both the collector current and

collector-emitter voltage were measured. All currents were

measured using a Pearson Model 411 current transformer.

Measurement of currents less than 50 mA necessitated the

use of a plug-in voltage amplifier capable of sensing

microvolts. For this purpose, a 7A22 differential ampli

fier was employed.

The RS3500 device used in this measurement had a BV

of 116 V at 1 mA. According to the theory, it would be

expected that v__ values greater than BV___ cannot stably

be obtained, as these voltages lie outside the transistor's

stable region of operation. For a collector supply of

2 50 V and a load of 500 ft, the device switched to a v^^ of

40 V, which was independent of base drive (above the

threshold for triggering which was found to be approxi

mately 500 uA). This is illustrated in Figure 13:

Figure 13(a) shows the base current pulse; Figure 13(b)

shows the collector-emitter voltage (lower trace), and the

collector current (upper trace). It is to be noted that

changes in V__ are reflected in the collector current C £

waveform. Forty volts is higher than voltages normally

considered to correspond to the V^^ saturation curve;

furthermore, V__ remains low even after the base drive has C£

been removed, and after some time falls to an even lower

value, around 10 v. The second reduction in collector

36

voltage is accompanied by an increase in collector current,

although this does not persist as long as the collector

voltage remains low. This result is quite reproducible and

is not understood. However, if while base current is

flowing the device is held in a quasi-stable state, upon

this current becoming zero, the operating point of the

device is undefined and hence the device is free to move

along the load line until a stable state is reached, the

current being limited only by the external resistance. The

device cannot turn off as the collector current has passed

through a point that is higher than the point of insta

bility on the BV__^ curve; i.e., the device is in second

•^ CEO

breakdown. The device would remain on indefinitely,

eventually failing owing to excess power dissipation in the

device, if the V supply were not disconnected. Although

somewhat different, these results are reminiscent of a

stepped second breakdown previously observed i4J.

For load values of 4.7 kft and 47 kft, with a V of

250 V, it was not possible to switch the device to voltages

of the order of BV^^^ and for v^^ to remain at that value,

even momentarily. Base currents of 500 uA < I^ < 1 mA,

resulted in V__ oscillating between several volts and 130 v

CE

(4.7 kft) and 180 V (47 kft). Figure 14 illustrates the base

current and corresponding collector current and collector-

emitter voltage, for a 47 kft load and a v^^ of 250 v. it

is unclear as to the mechanism by which the device is able

39

to oscillate to voltages that are greater than BV .

Large values of load resistance (e.g., 47 kft) severely

restrict the values of collector current obtainable and

inhibit the device going into second breakdown. This is

evident from the fact that the device turns off, for all

values of base current, upon this current being removed.

It may possibly be the case that, at low values of base

current, the device is in a borderline state, between being

turned on and turned off, resulting in the oscillations

observed. Base current pulses greater than 1 mA caused the

transistor to be driven directly into saturation, approxi

mately 5 V (Vpg); the device turned off upon removal of the

base current pulse.

Reducing the collector voltage moves the load line

(for any given load) closer to, or further into the stable

operating region. For any given value of load, the load

lines obtainable are always parallel to each other,

regardless of the value of V^^. The waveforms observed for

the 500 ft load at 160 V V are essentially the same as

those obtained for the 250 V V^^ measurement. This is as

expected, for again the load line does not enter any stable

region.

The results obtained for load values of 4.7 kft and

47 kft, at 160 v V-p, are interesting because the device

appears to operate in an almost conventional manner: v^^

can be stably controlled by the base current from 0 V to

40

125 V (4.7 kft) and from 0 V to 150 V (47 kft). Unlike the

results obtained for the 250 V measurement, no oscillations

were observed. Figure 15 illustrates the base current

pulse, corresponding collector current and collector-

emitter voltage for a V^^ of 160 V and load of 47 kft. (The

slow decay in V is the result of the relay contacts

opening; the contacts appeared to be damaged after conduct

ing the high currents involved in these measurements.) v__

falls from 160 V to 150 V for the duration of the base

current pulse. The high frequency noise existent on the

current waveforms is not produced by the circuit or the

Pearson current coil, but is a consequence of the sensitive

setting that had to be used on the Tektronix 7A22 ampli

fier, in order to measure the small currents present. The

fact that the device can operate at voltages in excess of

its breakdown voltage is troublesome, and raises the

question as to whether, in fact, BV^j,^ is the defining

breakdown voltage. Measurements of BV^^^ and BV ^ , for a

range of resistors, showed that for small values of R

(50 ft), BV -^ and BV^^j^ are essentially both equal to

BV . The value of BV ^ ^ begins to deviate from BV^^^ as

R is increased from approximately 20 kft. It is interesting

to note that, for a given value of R, BV^^^ is only, at

most, several volts greater than BV ^ . This indicates

that the value of R is the primary determining factor,

regarding both the BV ^ ^ and BV^^^ breakdown voltages; also

-t i .

a.

b .

43

that breakdown voltages up to the maximum value of BV-g^

can be obtained without recourse to a negative bias.

Clearly, by definition, the device cannot sustain voltages

greater than BV-,„Q with the base open circuited; this

suggests that it is most probable that the breakdown

voltage is determined by BV ^^^cBO ^°^ * ^^ ^^'

Although the device is operated in the BV-_^ mode when the

base-emitter lunction is forward biased, the negative base

potential need not be considered; however, the resistor

remains in the circuit. It is believed that this is the

mechanism by which the device is able to sustain these

higher voltages.

Circuit Effects

Early single-shot base-triggered line pulser measure

ments suffered from loading effects; consequently, these

measurements were repeated using the voltage divider

technique described earlier, in which the oscilloscope

input resistance acted as a sampling resistor. However,

the role of parasitics was still not clear, so a sequence

of measurements was performed using the mercury-wetted

relay as a substitute for the TUT to determine if para

sitics were still a problem in the circuits. Table 1 shows

the circuit fall times, t , which is the time required for

the output pulse amplitude to increase from 10% to 90% of

its maximum output value (V ^ = 250 V). The voltages shown

44

0) rH

4J

4.)

O



<d 0)

•d c <d

CO

<u

43 4J

<d

<u

(0

o CO 4) CO r-»

04

D ^

<U 43 4J

O tM

0)

.P •H 4J CO

45 3 en td

(0

<d rH <U Oi

H tt II U U U Ii II It R H N n 1 H N n • n n N 1 R R H H M H R R n R R II H n H R R n R R R R H

B— U K R

n H n H R u n n n R R H R R H R R H R R H R R R H R R tt R R H R H II n n N u u u a u t .

/ * N

>

a >

X"s. (0 C

V.*-

M •M

Ft« a

o o o

» o CM

h a

o o o CVJ

(C4

a o o CN

FL«

a o CM

•

{S4

a o o o

» o CM

b a

o o o CN

Pl a

o o CM

Pl4 a

o CM

o «

m en CM

o •

in CM CM

o •

o c 1—1

o •

0 0 l—i 1—1

f i ft

CM

o •

CM

tn •

1—(

o ft

f—l

c: o l A

O •

^ u-> CM

O •

• ^ in CM

o •

IT)

\o CM

m •

0 0

m CM

<7\ •

O

oo •

o

0 0 •

O

r^ •

O

C3

O 0 0

CM «

CM m CM

vO ft

I — •

r* CM

VO ft

l-H

r*. CM

o\ ft

t—1

vA CM

r«. •

O

r» •

O

r-» »

O

r«. •

O

c: M

0 0 •

CM •

ON m CM

o »

0 0 0 0 CM

o •

0 0 ON CM

o •

0 0 ON CM

r^ •

O

r^ ft

O

!>• ft

o

r«. •

O

c; M

0 0 sf

45

represent the voltage across the entire load, which was

obtained by multiplying the measured voltage by R-/50 and L

adding it to the observed voltage. The values of tabulated

voltage are not the absolute peak values, which include an

overshoot caused by the parasitic capacitance which shunts

the load and changes its effective value at high frequen

cies. However, after the initial fast transient switching

of the device, the load is essentially its nominal value.

It is believed that the calculated voltages represent the

true load voltage, the voltage at which the capacitor (Cj)

begins to discharge, i.e., the beginning of the exponential

discharge curve. On an expanded time scale the voltage

spike created by the parasitic capacitance is not visible,

while the exponential curve is well defined. Despite this,

some tabulated load voltages slightly exceed 250 v. This

is physically unreasonable and is believed to be attributa

ble to estimation errors in extracting the data from the

photographs.

From Table 1 it can be seen that the relay must close

in approximately 0.7 ns. This closure time is independent

of the coil drive and of the circuit to which it is

connected. The above information is sufficient to enable

the form of the load voltage to be calculated if it is

assumed that the voltage fall across the relay is linear.

This calculation is given in Appendix A and the final

46

result shown below as Equation (1). V_(t) is the calcu-a.

lated voltage across the input impedance of the oscillo

scope, V^^ is 250 V, C^ is the load capacitance, R is the

load resistance; R represents the oscilloscope input

impedance which is 50 ft and t^ is the closure time of the

relay (0.7 ns).

v^it) = ^CC<^L^

-PlVlf^ - 1 (1)

Equation (1) is valid for 0 < t < t, only, setting

t = t^ enables the peak voltage value to be calculated.

Table 2 was constructed in an analogous manner to Table 1,

the complete load voltage being calculated using the

formula discussed earl ier . (If the relay contacts are

considered to close as an ideal voltage step function, this

yields calculated load voltages of 250 V, irrespective of

the values of load capacitance and resistance used.)

From a theoretical standpoint, Table 2 shows that the

peak load voltage is maximized for large C R products. L LI

The empirical results of Table 1 show a similar, but weaker

trend; there are other factors influencing these results,

but non-ideal switching behavior clearly affects the output

voltage. At high frequencies, the parasitic capacitance

shunts the load, resulting is an effective impedance which

is less than the nominal load value. This increases the

47

Table 2

Theoretical Peak voltages Obtained with the Mercury-Wetted Relay as a

Substitute for the TUT

50 ft 480 ft 4.8 kft

48 kft

20 pF

179.8 241.7 250.0 250.0

V (V) P

200 pF 2000 pF

241.5 249.1 250.0 250.0

249.1 250.0 250.0 250.0

20,000 pF

249.9 250.0 250.0 250.0

48

proportion of voltage that is dropped across the oscillo

scope.

The smallest magnitudes of load voltage are obtained

for the 50 ft load; Equation (1) predicts that the magnitude

of these voltages should be the smallest. However, the

voltage obtained for the 50 ft load and 20 nF capacitor is

smaller than predicted. More generally it appears that the

voltages obtained for the 50 ft load, irrespective of the

value of capacitance used, are smaller than predicted.

Equation (1) ignored parasitics: parasitic capacitance has

been discussed, but cannot be a significant influencing

factor for the 50 ft load as this measurement utilizes the

input impedance of the oscilloscope and 50 ft attenuators,

all of which are designed for high frequency operation. The

effects of parasitic inductance are proportional to the

rate of change of current with time (dl/dt). Assuming that

the parasitic inductance of the circuit is constant then

its effects will be most evident for the lowest values of

load, because these produce the highest dl/dt's. Conse

quently, the effects of parasitic inductance are most

noticeable at low values of load. As the effects of

parasitic inductance become greater, the rate of current

rise is further decreased, which results in the observed

rate of voltage fall decreasing, i.e., the switching time

increases. The peak voltage decreases as a proportion of

the voltage is dropped across the parasitic inductance.

49

Parasitic capacitance becomes more of a problem at large

values of load resistance, since the reactance it creates

has to be less than the value of load resistor in order to

create loading problems.

It is not clearly evident as to which parts of the

circuit are primarily responsible for the parasitic

inductance. However, from measurements made utilizing a

load comprised of series resistors (experimental proce

dure), it appears probable that the packaging of the relay

contributes significantly. The above measurement has

served to illuminate the effects of parasitics. It is

expected that the device packages of the various TUTs will

cause parasitic inductance effects to be manifested in

various degrees. However, results that differ substan

tially from those of Table 1 can most probably be attrib

uted to actual device phenomena, e.g., a change in switch

ing speed, etc.

Base-Triggered Bipolar Transistor Measurements

Tables 3 and 4 contain the summary of results of the

single-shot measurements on the RS3500 and the 2N2222,

respectively. Appendix B contains a complete characteriza

tion for the RS500; the results of the 2N2222 measurements

are similar. The fall time and peak voltage are as defined

in the previous section. (The peak values are estimated.

50

CO

0)

A (d H

4 J

3 P t

4 J

3 o o <=>

0) x :

I D CO CO

- P u*

U Ti o **-4

CO 

^ (d

CO

(d CQ

 0) Q

R B H H

n H R

« R H R R R R R R R

a R H H R U R R 11 H II tt H • R H M II H II 11 U H N H

R H

H M H N R B H R R R H tt II n H tt U H

u B tt B B tt II tt U R B H R tt tt n n H H tt R H H tt N R tt

- ^ V

>

a >

CO

c ^>m^

u p

1 f**

a o o o

• * •

1 o CM

Cbi

a o

1 o o CM

(^ a

o o CM

f*^

a o Cv*

Ft4

a o o o o CM

^ a

o o o CM

^ ^ ^ • ^ ^ • • ^

pbl

200

p

u* a

o CM

O m O O ft ft ft •

m 0 0 v r ON i—t r^ r*. m CM CM f - t CM

O NO O O • • • •

o o «n ON o •<»• « n i n CM CM r-H CM

O ON o o ft ft ft ft

O f ^ CO o NO CM ' ^ m r H CM CM CM

o m o o • • • •

vo in CO o 0 0 0 0 • * < f ^ CM CM

O N CO s o f ^ ft ft ft ft

CM CM CM O

CO ><t NO 0 0 ft ft ft ft

CM CM CM O

ON CM i n 0 0

• • • • r-H CM CM O

\ o o i n OO • ft • ft

r-t CM CM O

50 f

t 48

0 ft

4f

t8

kft

48 k

ft

ON f—l

 0)

Q

H tt B B B B H tt B B B N H B B B R tt H tt B B U tt H tt R tt B H

n n B II II M B tt R N B R R tt H

R— tt R H H tt B tt tt tt B tt fl tt tt tt B B B B R B B B fl B N H tt R B II H II B tt H U H tt

a tl H

n II tt U

^^^ >

^s«^

a >

. ^ 1 CQ 1 C I

•s—'

U 1 • P

1 Pb,

1 a 1 o 1 o 1 o 1 » 1 o 1 CM

fa a

o o o CM

fa a

o o CM

fa a

o CM

fa 1 a

o O 1 O 1

» o CM

fa a ; o i O 1 O 1 CM

fa a i

o O 1 CM

fa a

o CM

1 o 1 • 1 i n 1 '—• 1 CM

1 o 1 •

1 o 1 o CvJ

O •

o NO f—t

o •

CM ON

0 0

• CM

<r •

CM

NO

• r-4

CM ft

r - l

a o i n

i n

« 0 0 CO CM

NO

• o •<t CM

ON

• r^ CM CM

CM

• o 0 0 f — . I—(

o •

i n i n 1—t

o •

CO >:f CM

O

• CO

>* CM

r - t

• CM

m •

CM

CM

• CM

ON

• 1—t

c; M

oo •

O •

ON i n CM

O •

ON

m CM

o •

o i n CM

o •

o s i -CM

r« .

• O

r^

• O

r *

» O

r^ •

O

c: ^ 0 0 sr

51

^

f H

<d H

4J

a - p 3 CM O CM

CM

0) CN

1 > / ^ i 4 ^ V " .

H T3 O «M

CO



> ; <d 0)

M &H 1

<U CO (d CQ

0) 04 4?

1 1

•d C * H

(d o (0 <u s •H E H

i H f H (d (X4

CO 0) CO

f H

3 04

m f—l

0 0 CM

u 0)

s 9 sz: 9

- H

Dev

n tt B B B B H R R H H B U tt B B H R R H R R H N H tt B tt B H H tt B B B B tt B B B H tt II tt B

tt B H B tt B tt tt tt B R H B II tt H B B B B tt tt tt n H R H H tt H tt tt U tt tt tt tt B tt R » B H n tt • I

y - s

>

. ^ ^ >

• ' ^ \

CQ

c >^/'

Wi •p

fa a

o o o

o CM

fa a

o 1 o

1 o CM

fa a

o o CM

fa a

o CM

1 fa

a o o o

1 o CM

fa

a o o o CM

fa a

o o CM

fa a

o CM

o • o 0 0

O •

1 • * f ^

o ' V

I CM I i n

o .

f - H

CM

ON •

CO

- * •

CO

oo •

CO

vo •

l - H

50 ft

o • 0 0 0 0

o •

S t 0 0

o •

i n ON

O .

r—l

0 0

- * 4

CM

CO •

CM

f—l

CM

r~« ft

i—t

480 ft

o • r^ CJN

O •

i n <T^

o •

r * 0 0

o ft

r * CJN

<r • I—t

NO ft

t—t

1 ^ ft

1—1

S t •

^-*

4.8 kft

o ft

«-H

o I—t

o ft

f^

o f—l

o ft

1—1

o f—l

o •

NO ON

ON •

o

ON •

o

o • 1—t

<7N •

o

48 kft

S t 285

i-i 0) qm

n

2

a> u

•H

Dev

tt— n u II II H B H II M B B B H H tt B tt H tt u R B H R tt B B B H tt II H II Ii II It n n R B B B R U B

B B B B B U U tt B U B tt B tt H B B n B B B B H tt H R B tt B U H M fl tt B tt II n II 11 n II H M H

z * ^

> 's**'

a >

•^"^ 1 CO 1 C 1

>—>'

U i u

1 fa \ a

o 1 o

o e

o CM

fa a

o o o CM

fa a

o O 1 CM

fa 1 a

o CM 1

fa 1

a o O 1 o 1 O 1 CM 1

fa

a o O 1 O 1 CM 1

fa a

o O 1 CM

fa a

o CM

o o • • 1 o o 1 00 O

f—t

o o • • S t - * f^ ON

o o • •

vo i n i n ON

o o ft •

f-H f—l CM 0 0

o o • • r ^ l-H

ON O f — t

o o • • i n —1 <t\ o

I—t

o o • •

r > * i - t 00 O

1—I

o o • •

oo NO ON ON

CM CO • •

CM CM

CM CO • •

m CM

O f - i • •

CO CM

0 0 r«>. • •

l-H l-H

50 ft

480 fi

< t ON • •

-< o

vO o • •

.—1 r—l

• •

,-. o

i n o • «

l-H l-H

4.8 kft

48 kft

52

based on the criteria discussed previously.) Both of the

bipolar device types switch fastest for the 48 kft load--the

effects of parasitic capacitance are again most noticeable

at this value of load. This is consistent with the

theoretical predictions made earlier, i.e., those concern

ing parasitic inductance; however, the switching speed

obtained for the RS3500 device is not in general slower for

the 50 fi load as compared to the 480 ft and 4.8 kft loads.

The RS3500 device type has exhibited switching speeds of

the order of 0.7 ns and the measurement using the 4.8 kft

load, with the relay in circuit, has demonstrated that

switching speed is not diminished for this value of load as

compared to the 48 kft load. Hence, it must be assumed that

the slower switching speed obtained for the 4.8 kft load is

attributable to the device itself.

The slower switching times exhibited for loads other

than 48 kft, may possibly be attributed to the parasitic

inductance of the device package, although this is specula

tion and again one would expect its effects to be most

noticeable for low values of R. . The extent to which this

phenomenon may be attributed to actual device physics is

unknown. A general trend is evident from Table 3: higher

peak voltages are obtained for larger values of the C R

product. A similar trend in switching times is not

apparent, i.e., there is no clear dependence upon R ,

although there is a weak dependence on C . Generally the

53

fall time either increases or remains constant as C. is LI

increased. It can be seen from Appendix B that the smooth

R^ exponential charge-up is not obtained for the 480 ft

load and values of C ^ of 200 uF, 2000 pF and 20 nF; the

reasons for this are unknown.

The preceding analysis is also applicable to the

2N2222. However, unlike the RS3500 device, there is a

clear dependence of the switching speed upon the value of

Rj ! for a given value of C^ the fall time decreases as the

value of R^ increases. As discussed earlier, the 2N2222

device is designed for small signal operation; hence, it is

possible that its deterioration in switching speed, for low

values of R., is the result of device packaging (small

diameter wires connecting the chip to its external pack

age), i.e., parasitic inductance. This suggests that the

2N2222 may be suitable for fast switching applications

provided that the current is restricted to hundreds of

mllliamperes.

The anticipated dependence of the shape of the

switched output pulse on load resistance was not observed

for either device type. There were in general no inter

mediate values of voltage to which the device switched; a

single device exhibited some discontinuity on the leading

edge of the waveform for the 48 kft load. This behavior is

not representative and is considered anomalous.

54

Two explanations for the absence of the phenomenon

discussed were considered: driving the base with a 20 V

pulse results in large base currents (on average, approxi

mately 500 mA) which may be sufficient to drive the device

directly into saturation. However, no "hang-up" of the

leading edge could be created by reducing the base drive.

Voltages of approximately 2 V were required to trigger the

devices. This trigger voltage produced a reduced-amplitude

pulse (approximately one order of magnitude smaller) but

did not exhibit any "hang-up." The effect of base drive

was discussed earlier; another possible explanation is the

shunting of the load by the parasitic capacitance. It is

possible that the effective load impedance created is less

than the 4000 ft considered necessary in order for a

discontinuity to be observed. A value of the parasitic

capacitance cannot be calculated owing to the number of

undetermined quantities incorporated in the circuit, viz.,

the actual voltage across the load, parasitic inductance,

etc.

The MOS Transistor

MOS power transistors exhibit a latch-up behavior

which is similar to the unstable switching observed in

bipolar devices. The principal model is a transition

between BV^„g (emitter-base short) and BV^^^, which occurs

when emitter current begins to flow in the parasitic n-p-n

55

bipolar transistor by way of a parasitic internal resis

tance. It is, however, possible for the parasitic bipolar

to exhibit the instability employed for switching the

avalanche transistor; in that case, a large voltage swing

is possible.

The gate-triggered results for the BS170 are listed in

Table 5. In this mode of operation, the speed at which a

MOSFET turns on is determined by how quickly the gate-

source capacitor can be charged; this has a typical value

of 60 pF for the BS170 (see Appendix F for specifications).

Figure 16 illustrates the 20 V gate trigger pulse, matched

into a 50 ft load and as observed between the gate and

source by means of a Tektronix P6057, 5 kft probe. The

degradation of the leading edge probably reflects the time

required to change the gate input.

The magnitude of the trigger voltage and v j are the

same as those used to evaluate the base triggered perform

ance of the 2N2222. This facilitates a direct comparison

between these two devices. The BS170 switches faster than

the 2N2222 for all values of C ^ and R^; from 7% (C ^ =

20 pF, R, = 4.8 kft) to 52% (C ^ = 200 pF, R^ = 50 ft) faster,

for device E as contrasted to device number 2815, although

with the absence of any apparent pattern. As with the

bipolar devices, the effects of parasitic capacitance are

evident; again this distorts the apparent value of the load

voltage. Hence, some degree of estimation was involved in

56

in

0) r H JQ (d EH

4J

3 P< 4J 3 O O

0) f C • P

. H CO CQ

wt -d

o tM

CO 0)

<D U

gge

tn-H

Ita

o

-Tr

 - p

(d (d (i> 0) 04 f C

TJ <4->

C M-l <d CO (D

• H EH

r H r H

(d CLI

O

CO

(0 i H 3 O4

B B R R R tt B B R R « fl B B « B H R H tt H N tt R R tt R R R R R R R H B B B B R R R R H B B

a B B H II H u N tt H tt 11 B B B B B B tt R tt B tt M tt tt tt B tt B tt R tt H tt R tt tt tt B fl U B H B

y f - V

>

a >

CQ c

>^^

U

u

fa 1 a 1

0 1 0 1 0 1

# 1 0 1 CM

fa a ;

0 0 0 1 CM

fa a

200

fa a

0 CM

1 fa a

0 1 0

0

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0

200

1 fa 1 a 1 0 1 0 1 CM

1 fa 1 a 1 0 1 CM

0 0 • •

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00 00 00 ON

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ON »n CO 00

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C7N f—l 0^ 0

I—1

0 0 • •

ON f.^ ON 0

f - l

0 0 ft ft

r^ 1-4 ON 0

0 0 • •

ON f t a^ 0

r-t

r«» CO

CM i-i

• ^ CO ft *

CM f t

1 00 (^ \ * *

I - I «-•

1

1 CM CM 1 * * 1 f—t <—'

50 f

t 48

0 ft

CM 00 ft ft

r-t 0

CO 00 • •

r s 0

CO 00 • •

r s 0

CO 00 • •

r- 0

4.8

kf

t 48

kf

t

b.

Figure 16. Gate trigger pulse: (a) into a matched 50 ft load, (b) observed from the gate to source by means of a P6056 Tektronix probe, vertical and hor^-ontal scales are 5 V and 5 ns large division, respectively, for both photographs .

The

per

58

extracting the peak voltages from the photographs.

Appendix C shows a complete characterization for device E,

as a function of R ^ and C . It was concluded that the

gate-triggered BS170 was operating in the active mode, so

that only one gate-triggered measurement was performed.

Despite the distortion of the load voltage, the BS170 also

appears superior to the 2N2222 in terms of the peak load

voltage. The voltage switched is either comparable to, or

greater than, the corresponding value obtained for the

2N2222 device. It is greatest for the 50 ft load, for which

it is from 25% {C^ = 20 pF) to 52% (C ^ = 200 pF) greater,

for the same pair of devices as compared earlier. This may

possibly be attributable to the TO-18 package of the 2N2222

as compared to the TO-92 MOSFET package (see Appendices E

and F).

Collector-Triggered Measurements

Some early collector triggered measurements were made

on all three devices; however, these measurements suffered

from loading effects. It was attempted to take further

collector-triggered data on the RS3500 using the modified

circuit illustrated in Figure 5, in which the oscilloscope

input resistance was used as a sampling resistor. Most of

the circuit is isolated from the trigger pulse by means of

inductors; hence, the trigger pulse sees a simplified

circuit as shown in Figure 17. Collector-triggered

59

TUT

V W134MIP-3

CC 11 1 M of RG58A/U CABLE « ».

^ .

nm

PULSE GENERATOR

1.5 Mft

150 V

•«v^(t)

OSCILLOSCOPE (50 ft)

Figure 17. Circuit schematic illustrating the simplified form of the collector-triggered circuit.

60

measurements indicate that the device again switches very

rapidly when triggered from an overvoltage at the collec

tor. The circuit of Figure 17 terminates a transmission

line with a complex impedance. Large values of load result

in a considerable mismatch, creating considerable waveform

distortion at the collector of the TUT. For the largest

values of load, the pulse amplitude is essentially doubled,

decaying in an exponential manner; small values of load

create an undershoot which decays exponentially. These

phenomena are observed in the load voltages at the oscillo

scope. Again, this is a scaled version of the true load

voltage. Superimposed onto these waveforms are reflected

waveforms that propagate to the source from the load and

then back again. (The source has a high value of resis

tance in series with it; this is necessary to create the

pulse .)

Figures 18, 19 and 20 show load voltages for a 20 pF

load capacitor and resistor values of 50 ft, 480 ft and

4.8 kft, respectively. Notice that even the 50 ft load

produces an undershoot, for which the capacitor is respon

sible. The reflections discussed earlier are clearly

evident in Figure 19(a). Figures 18(b), 19(b) and 20(b)

show that the device is capable of achieving switching

speeds, in this mode of operation, comparable to those of

the base-triggered measurements.

61

^ ^ ^ 1

^ • ^ X ^ \ / ^

'idfl

fca^ -• T-r— — 1

1 a.

20(ky Mtm IrS

XfFSOB

Figure 18.

b .

Collectcr-triggered RS3500 waveforms; the amplitude of the triaaer pulse is ' 50 v, CL is 2 0 pF and R^ is 50 ft. The vertical scale is 200 V per large division for both photographs. The horizontal scales are: (a) 10 ns and (b) 1 ns per larae division.

62

a.

Figure 19.

b.

Collector-triggered RS3500 waveforms; the amplitude of the triacer pulse if 750 V,

j_i J- a 0 pF and R^ i scale 460 ft. Tne vertica.

is 50 V per large division for both photographs. The horizontal scales are: (a) 10 ns and (b) i ns per larae division.

63

a ,

a»M li

• • ^ • * . .

20aFi.X!

Figure 20 Collector-triggered RS3500 waveforms; the amplitude of the trigger pulse is 750 v, CL is 20 pF and Rr is 4.8 kft. The vertical scale is 5 V per large division for both photographs. The horizontal scales are: (a) 50 ns and (b) 1 ns per larae iiv'ision.

64

Collector triggered measurements were terminated, as

no qualitative information could be obtained from them,

owing to the level of stress applied to the device being

dependent upon circuit component values. The same phe

nomena were observed for the 2N2222 and the MOSFET;

consequently, no collector (drain) triggered measurements

were repeated for these devices. However, early measure

ments, despite suffering from loading effects, demonstrated

that these devices are capable of fast switching when

operated in this configuration.

Repetitive Measurements

All the devices which were stressed single-shot

survived with no apparent degradation, the criterion for

which was a change in certain characteristic voltages. For

the bipolar devices, these were BV^^^ at 10 uA, BV^^^ at

10 uA and BV-_- at 1 mA. Characteristic curves were also CEO

obtained to see if there were any significant changes in

the forward current gain. The BS170 voltages were BV g ^

V„„ = 0 and I^ = 100 uA, and the threshold voltage, v_„, GS D " for V = V ^ and I^ = 1 uA. No characteristic curves were

DS GS D

obtained for the MOSFETs because of curve tracer induced

oscillations which interfered with the measurement and

which could not be suppressed. Pre- and post-stress

characterizations are contained in Appendices B and C.

65

Subsequently, measurements were begun to test the devices

under repetitive collector-triggered stress.

The line pulser relay would not close if the coil

pulse was shorter than 1.5 to 2 ms; an interval of this

duration was required for the contacts to open again, so

that initially the devices were stressed at a repetition

rate of 200 Hz. The load resistance and capacitance were

50 ft and 200 pF. This combination was calculated to give

the least amount of trigger pulse distortion. Two RS3500

transistors were tested (collector bias was the same as for

the single-shot measurements). Each device was initially

characterized, stressed, then recharacterized. The first

was stressed continuously for ten hours (7.2 million

shots); after stress, the characterization parameters, with

the exception of BV^^Q, remained essentially unchanged.

Originally, BV^„^ was found to be 122 v at 1 mA. Follow-C£«0

ing the test, BV__^ could no longer be measured at its CEO

specified current because snap-back (unstable switching)

occurred at very low collector currents. However, BV

could be increased to 130 V before snap-back took place.

The second device was tested the same way, except that

stress was interrupted at one, two and five hours to

measure BV-_^. The initial value, at 1 mA, was 120 V; CEO

after one hour, and at subsequent measurements (two, five

and ten hours), snap-back at low currents appeared; the

66

respective values of BV^^^ at the specified times were

128 V, 130 V, 142 V and 142 V.

The repetitive tests were stopped when relay contact

bounce was observed. The W134MIP-3 relay is rated at a

1000 v DC switching voltage, a 1 A switching current, and a

2 A carry current, with the maximum volt-ampere product

being 50 VA. Under the test conditions existing here

(1500 V on the line, 20 A peak current, 200 Hz), the

average VA product is 0.06 VA, which is well within the

rating. However, exceeding the maximum rated switching

voltage while switching 20 A current spikes can be expected

to affect the contacts.

The circuit used for the resumption of the repetitive

measurements is illustrated in Figure 11 and was discussed

previously. Figure 21 shows the 750 V pulse (into a 50 ft

load) used to trigger the devices. Figure 21(a) shows the

trigger pulse on an expanded time scale, and Figure 21(b),

over a period of 100 us. Figure 21(b) clearly illustrates

the absence of contact bounce (the principal bounce

previously occurred from 20 to 30 microseconds after the

initial closure). The modified repetitive test circuit

allowed the maximum rep-rate to be increased to 350 Hz.

This was desirable as a 200 Hz stress level did not result

in unambiguous device degradation. Two other RS3500

devices were tested at this higher level of stress,

resulting in degradation of the form previously described.

a.

b.

Figure 21.

i ne

The 750 v trigger pulse used for repeoitive ooilecoor-origgering; the rep-rate is 350 Hz. vertical scale is 200 both photographs, the horizontai scales are: (a) 10 ns and (b) 10 us per larae iivision.

V tor

68

Each device was recharacterized after one, two, five and

finally ten hours. Again, the collector-emitter voltage

required to initiate breakdown increased with the total

time of stress.

For device A22, the measured values of BV--, were: CEO

116 V (initial), 128 V, 130 V, 136 v and finally 138 V. For device A23, the measured values of BV__ were: 114 v

CEO

(initial), 126 V, 129 V, 134 V and finally 139 V. Device

A23 was further tested for another 10 hours without

interruption and BV „ was found to increase by another

volt. The rate at which BV-_- increases with continued CEO

stressing appears to decrease with time. It is not known whether the value of BV^-. will eventually saturate or

CEO ^

continue to increase. It is unclear whether the change in

BV-_,Q characteristics represents true degradation, inasmuch

as the output pulse (Figure 22) which was measured at

intervals during and at the end of stress, did not change.

Perhaps the low current snap-back increases the vulnera

bility of the transistor, but this cannot be established

without longer term testing.

o9

a.

b.

Figure 22 Observed load voltage for a RS3500 device repetitively triggered at the collector; the rep-rate is 350 Hz, CL is 2 00 pF and R- is 5 0 it. The vertical ocale for both photographs is 00 V oer large di'/ision, the hori zontal scales are: (a) 10 ns and (h) i ns.

CHAPTER V

CONCLUSIONS

The 2N2222 is inferior to the RS3500 with respect to

pulse rise time and switched voltage. However, the 2N2222

is a general purpose small-signal transistor which is not

designed to operate in the avalanche mode, whereas the

RS3500 is specifically designed and fabricated for enhanced

switching. Still, both devices operate in an unstable

mode, raising the question of long-term reliability.

Switching in the BS170, on the other hand, may or may not

involve instabilities, and its switching characteristics

are comparable to those of the RS3500, certainly for

drain-triggering. Both power bipolar transistors and power

FETs are available at higher collector and drain breakdown

voltages, so that higher voltage Marx banks are possible

with a small number of stages. However, there is some

indication that bipolar devices switch more slowly at

higher collector voltages than at low collector voltages.

There is no similar information respecting drain-triggered

MOSFETs.

70

LIST OF REFERENCES

fll J. F. Francis, "High Voltage Pulse Techniques," AFOSR report number AFOSR-74-2639-5, Texas Tech University, December 1, 1972.

[21 S, A. McMullen, "Energy Considerations in Second Breakdown," M.S. thesis, Texas Tech University, 1984.

[3] J, Power, Los Alamos National Laboratory, New Mexico, private communication, March, 1984.

[4] W. M. Portnoy and F. R. Gamble, "Fine Structure and Electromagnetic Radiation in Second Breakdown," IEEE Transactions on Electron Devices ED-11, 470 (1964).

71

APPENDIX A

THEORETICAL LOAD VOLTAGE CALCULATION FOR CONSTANT dv/dt SWITCH

72

73

Voltage across the switch is given by v(s) = v /s -

V^^/t^S^ 0 < t 1 t^.

CO CO = I(S) t^S' SO,

•I- R . + R

v CC

-v => 1(3) = CC

1^ [so, + R + R. .]

-V C V- C > I(S) = ^^ ^ + ^^ ^

I(t) = lexp

1

v^(t) ^CC^'^L f / -t \

where Vo(t) is the voltage observed at the oscilloscope A

input.

APPENDIX B

PHOTOGRAPHS OF THE REPRESENTATIVE BEHAVIOR OF A BASE-TRIGGERED RS3500, AS A FUNCTION OF

LOAD RESISTANCE AND CAPACITANCE

74

a.

b.

Characteristic curves for device A19: (a) initial; (b) final. The vertical and horizontal scales are 10 V and 1 mA, respectively, per large division; base current is 0.005 mA per step. The ^nitiai characterization, which remained unchanged, is: B- cEO ^ " ^ " 104 V; BV^Bo (-0 »-^ = -"^ ''' '-'EBO (10 uA) = 10.6 V.

7 6

b.

C L is 20 pf and R L is 50 ft. The horizontal scales are (a) 5 ns and (b) 1 ns; the vertical scale is 20 V (all per large division).

b.

CL is 20 pF and RL is 480 ft. horizontal scales are (a) 10 (b) 1 ns; the vertical scale (all per large division).

The ns ma is 5

a.

b.

CL is 20 pF and RL is 4.8 kftft horizontal scales are fa) 20 (b) 1 ns; the vertical scale (all per large division).

ns and is 1 V

a.

20tf 4SB?

b.

CL is 20 pF and RL is horizontal scales are and (b) 1 ns; the 100 mv (all per large

vertical

48 kft. The (a) 100 ns

scale division).

so

a,

SOOtV

mf sss

CL is 200 pF and RL is 50 ft. The horizontal scales are (a) iO ns and (b) 1 ns; the vertical scale is 50 V (all per large division).

1

500t¥ f l-'S

Tooifm

b.

CT is 200 pF and RL i-? 480 17. The horizontal scales are (a) 50 ns and (b) 1 ns; the vertical scale is 5 V (all per large

J^ \^ - ^ -w --fc .b >..>'

ivision)

8 2

a.

b.

CT is 200 pF and RL is 4.3 KL'. The horizontal scales are (a) 200 ns and (b) 1 ns; the vertical sca.e is 500 mV (all per large division).

8

a,

b.

CL is 200 pF and RL is 48 ki"!. The horizontal scales are (a) 200 ns and (b) 1 ns; the vertical scale is 100 mV (all per lar^e division).

84

b.

CL is 2 nF and RL is 50 ft. The horizontal scales are (a) 100 no and (b) 1 ns; the vertical scale is 50 V (all per larae division).

35

a.

b.

CL is 2 nF and RL IS 4eO horizontal scales are (a) _ and (b) 1 ns; the vertical :. 5 V all per large division).

ft. The 500 ns

icale

r^ •'-^•0

o X 1 i C CL is 2 nF and RL is 4.8 horizontal scales are (a) 2 us and (b) 1 ns; the vertical scale is 1 v (all per larae division).

87

a.

b.

C is 4 8 kft. The . is 2 nF ana .-.L horizontal scales are (a) 200 ns and (b) 1 ns; the vertica. scale is 100 (all per iarue division).

88

a.

CL is 20 nF and RL is 50 ft. The horizontal scales are (a) 500 ns and (b) 1 ns; the vertical scale is 50 V (all per large division)

89

a.

Than

CL is 20 nF and RL i- 4?0 ft. horizontal scales are (a) 5 u (b) 1 ns; the vertical scale is 5 (all per large division).

V

90

a.

CL is 20 nF and RL is 4.8 kft. The horizontal scales are (a) 10 us and (b) 1 ns; the vertical scale is 1 V (all per large division).

91

a.

b.

4 8 kft. The CL is 20 nF and RL is horizontal scales are (a) 200 uS and (b) 1 ns; the vertical scale is 100 V (all per large division'.

APPENDIX C

PHOTOGRAPHS OF THE REPRESENTATIVE BEHAVIOR OF A GATE-TRIGGERED BS170, AS A FUNCTION OF

LOAD RESISTANCE AND CAPACITANCE

92

BS170 Device Number E

Initial characterization: BV___(v^_ = 0, I-

OSS GS u

= 100 uA) = 130 V

^GS<^^><^DS = GS' D

= 1 mA) = 2.3 V.

Recharacterization: Unchanged.

93

94

a.

b.

CL is 20 pF and R^ is 50 ft. The horizontal scales are (a) 10 ns and (b) i ns; the vertical scale is 10 V (ail per large division).

95

a .

200Mf m^^

zmt m

4

)

^

b .

i s 480 (a )

ft. 10

The ns and C. i s 20 pF and R L ^

L ^4-aT s c a l e s -^^^ ^ , ^- •-) n o r i o o n t a . ^ s c a ^^^^^^^ s c a l e lo (b) i ^^ ' .^ t^Xe d i v i s i o n ) , ( a l l p e r l - r , e a i

96

a.

b.

CL is 2 0 pF and RL is 4.8 Kft, Tne horizontal scales are {a) 50 ns and (b) 1 ns; the vertical scale is 200 mv (all per larae division*.

97

a.

f-at i

»pF m

kft. 'N

The CL is 2 0 pF and RL IS -O horizontal scales are (a) 200 ns and (b) 1 ns; the vertical scale 50 mV (all per large iivision).

Q3

a.

200M RTt40S InS

mf SB

b .

CT is 200 pF and RL is 50 horizontal scales are (a) (b) 1 ns; the vertical sc (ail per large aivision).

ft. Tne 10 n3 and

ale is 2 V

99

b.

Cj is 200 pF and RL is 480 ft. The horizontal scales are (a) 10 ns and (b) 1 ns; the vertical scale is 200 mV (all per large division).

100

a.

2(M T-mi

mF4.9

b.

CL is 200 pF and RL IS 4.8 kft. The horizontal scales are (a) 50 ns ana (b) 1 ns; the vertical scale is 200 mV (all per large division).

1 0 1

a.

r-aw f ms

-S»t5*.

aOkFOD

b .

200 p F a n d R] CL is 20U PF ana KL is 4 8 kft. Tne horizontal scales are (a) 200 ns and (b) 1 ns; tne vertical scaxe is 50 mV (all per large division).

102

a.

axM 2)r{«tt IrS

if 509

b .

CL is 2 nr and RL is 5 0 ft. The horizontal scales are va) 10 ns and (b) 1 ns; the vertical scale is 20 V (all per large aivision).

103

a.

^ 7 CL IS / nF and RL is horizontal scales are (b) 1 ns; the vertical (all n-er large division).

4 80 Q. (a) 10 scale is

Tne n 3 a n >

2 V

103

a .

200M 29Hm IrS

ifm

b .

2 nF a n d Ri i s 480 ft. CL IS Z nt ana K.L horizontal scales are (a) 10 (b) 1 ns; the vertical scale is

Tne n s and

V 1 ^

(all per large division).

104

a.

CL is 2 nF and RL is 4.3 kft. The horizontal scales are (a' 50 ns and (b) 1 ns; the vertical scale is 500 mV (all per large division).

05

JrS

in.

af«tf

CT is 2 nF and RL is 48 kft. The horizontal scales are (a) 200 ns and (b) 1 ns; the vertical scaie 50 mV (all per large division).

13

106

CJ is 2 0 nF and RL is 5 0 ft. The horizontai scales are (a' 10 ns and (b) 1 ns; the vertical -caie is 20 V (all per large division).

107

480 Q. The (a) 10 ns and

(b) 1 ns; the vertical scale is 2 V (all per large division).

C is 20 nF and R^ is horizontal scales are

108

is 2 0 nF and RL is 4.8 kft. Tne hSrizontal scales are la) 50 ns an (b) 1 ns; the vertical scale is 500 mV (all per large division;.

109

CL is 20 nF and RL is 48 kft. The hori'ontal scales are (a) 200 ns and (b) 1 ns; the vertica^ scale is 5 0 mV (all per large division).

APPENDIX D

SPECIFICATIONS FOR THE RS350 TRANSISTOR

110

I l l

RAYTHEON Silicon Planex* Avalanche Transistor

RS3500

Description The Raytheon NPN Silicon Planex* transistor type RS3500 is designed for high breakdown voltages, and fast avalanche switching speed, useful in applications requiring rise times under 1 nanosecond.

Absolute Maximum Ratings Collector to Base Voltage Vcso Collector to Emitter Voltage VCEO

Emitter to Base Voltage VEBO

Total Device Dissipation @Case Temperature 25 'C @Case Temperature 100°C @Free Air Temoerature 25° C

Junction Temperature (Operating) Storage Temperature

200 Volts 50 Volts

7 Volts

4.0 Watts 2.0 Wans O.a Watts

—€5°C -200" C —65" C -200° C

Eiectrical Characteristics (TA =

Collector to Emitter Sreakdown Voltage

Emitter to Base Breakdown Voltage

Collector to Base Leakage Current

Emitter to Base

Leakage Current

Collector to Emitter Saturation Voltage

Base to Emitter Saturation Voltage

Collector to Emitter Breakdown Voltage

Rise Time

Pulse Width

Pulse Amplitude

Sym.

BVCES

BVE30

ICBO

lEBO

VCEfs)

VBE(s)

BVCEO

tR

PW

V pulse

25° C unless otherwise noted)

Conditions

Ic = IOMA

Ic = lOuA

VC3 = 100V

VE3 = 5.0V

10mA. 1mA

10mA. 1mA

lc= 10mA

(Figure 1)

(Figure 1)

(Figure 1)

Min.

200

7.0

50

Typ. Max.

300

Units

Volts 1 1

I Volts

! ^0

1 0.8 3.0

190

1.0

0.3

0.9

MA

1

Volts

Volts

Volts

1 nS

nS 1 Volts

•Pi«n«x—fltytntoni Owignation For Pi«n«r Eoitaxiai

112

RS3500

•300V

t f l<1nS PWslOnS

25V

100K

tOOOpF

22pF 5in -WV-

< ^ T.U.T

Sin:

-1.8V o-

lOOOoF i 1

lOOn Pot.

-OQut

Vp » 190V Typical tR » O.SnS Typical PW * 3nS Typical

Hgure 1 . Avalanche Switching Circuit

Packaging Information

•370 •" 3 3 5

DIA. ! .335 r .305 "* i OIA.

3-Lead TO-5 Package

.040 MAX. .260 h ^40

f 0 Ii U f !o.A. ^ny °- r"

3 LEADS -

^ 0 0 TYP

Mechanical Data

Case:

J ED EC TO-5

Terminal Connections:

Lead 1 Emitter Lead 2 Base L e a d 3 C o l l e c t o r (atcctneaiiy connvctM to M M )

TH« .nfonnwton eom»io«l m tfx. aata lAMt n u B M « ear^ l iv eomo.i»0: n o - * * * ' •! «n«ii not Dy .moHOi.on or <jtf»«n«.»« owom« M n e« tn« l»n»» .no eooait.OM o« any totuwiuwtt !••«. «»ivtn«on s liMinrv »n«(l M o«i«rTT,.n,o loiwv ov t» tt»na»ro i»tw« ana cenaiuon* o» tatu. NO f»0f««it inon n to (oo«>caf ion or u»« or tnm '.n* c:.«uit» ar* •.tn«r «f^Ma or trM from e«t»ni

APPENDIX E

SPECIFICATIONS FOR THE 2N2222 TRANSISTOR

113

114 TYPES 2N2217 THRU 2N2222. 2N2218A, 2N2219A. 2N2221A. 2N2222A

N-P-N SILICON TRANSISTORS B U L L E T I N N O . DL-S 7 3 1 1 9 1 6 . M A R C H 1973

DESIGNED FOR HIGH-SPEED, MEDIUM-POWER SWITCHING AND GENERAL PURPOSE AMPLIFIER APPLICATIONS

• *^FE • • . Guaranteed from 100 M A to 500 mA

• High f j at 20 V. 20 mA . . . 300 MHz (2N2219A, 2N2222A)

250 MHz (all others)

• 2N2218, 2N2221 for Complementary Use with 2N2904,2N2906

• 2N2219,2N2222 for Complementary Use with 2N2905, 2N2906

*mechanical data

Devic* types 2N2217, 2N2218, 2N2218A. 2N2219, and 2N2219A are in JEDEC TO-5 packages. Device types 2N2220. 2N2221, 2N2221A, 2N2222. and 2N2222A are in JEDEC TO-18 packages.

THE COLLECTOI IS IN E l E a i l U L CONUCT WITH THE USE

0 100—1 y-t COlUCTOt

H»-— • ^ " • l

I r « o u t 0 too MIN - - ^ — t - 000* > i»*os 1 • * " >

DdAKS Of ouniNc iw ' • 001* TMIl 20Nt OTTtONAl LjJATINC ' " ' * IMITtlt

PtANt

0e4» i n . «m 001* n * •"• OOX j j ^

^ mm ~ %~%tM

T0.5 DIMENSIONS ARE IN INCHES UNLESS OTHEIWISE SrECIFIED TO-18 I " TO-S TO-II

'absolute maximum ratings at 25°C free-air temperature (unless otherwise noted)

Collector-Base Voltage Collector-Emitter Voltage (See Note 1)

Eminer-Base Voltage Continuous Collector Current Continuous Device Dissipation at (or below) 25''C Free-Air Temperature (See Notes 2 and 3) Continuous Device Dissipation at (or below) 25*'C Case Temperature (See Notes 4 and 5) Operating Collector Junction Temperature Range

Storage Temperature Range Lead Temperature 1/16 Inch from Case for 10 Seconds

2N2217 2N2218 2N2219

60

30 5

0.8

0.8

3

2N2218A 2N2219A

75 40

6 0.8

0.8

3

2N2220 2N2221 2N2222

60 30

5 0.8

0.5

1.8

2N2221A 2N2222A

75 40

6 0.8

0.5

1.8

- 6 5 to 175 - 6 5 to 200

230

UNIT

V V V A

W

W

°C ^C ^C

N O T E S : Th«M varuM apply between 0 and 500 mA collector current when tfie bate-ennitter diode n open^ircuited. Oerata 2 N 2 2 1 7 . 2 N 2 2 1 8 , 2 N 2 2 1 8 A , 2 N 2 2 1 9 , and 2 N 2 2 1 9 A linearly to 175°C free-air temperature at the rate o« 5 33 myft/ Derate 2 N 2 2 2 0 . 2 N 2 2 2 1 , 2 N 2 2 2 1 A. 2 N 2 2 2 2 . and 2 N 2 2 2 2 A linearly to 176°C free-air temperature at the rate of 3.33 mw ' Oarate 2 N 2 2 1 7 , 2 N 2 2 1 8 . 2 N 2 2 1 8 A . 2 N 2 2 1 9 . and 2 N 2 2 1 9 A linearly to 175°C ca»e temoerature at the rate of 20 0 m w ' c . Derate 2 N 2 2 2 o ! 2 N 2 2 2 1 , 2 N 2 2 2 1 A , 2 N 2 2 2 2 , and 2 N 2 2 2 2 A linearly to 175°C ca»e temperature at iha rale of 12.0 m W / ' C .

• J E D E C regifiered data. T h u d a t a theet containt all applicable regntered data in affect at the time of publication. USES CHIP N24

TYPES 2N2217 THRU 2N2222. 2N2218A. 2N2219A, 2N2221A. 2N2222A NPN SILICON TRANSISTORS

115

2N2217THRU2N2222

'electrical characteristics at 25°C free-air temperature (unless otherwise noted)

PARAMETER

V(BR)CBO

V«BR)CE0

V(BR)EBO

•CBO

>EBO

"FE

V B E

VCE(sst)

N«i

• T

Cote

> i«(rcsl)

Coilector-BaM

Breakdown Voltage

Collector-Emitter

Breakdown Voltage

Emitter-BaM

Breakdown Voltage

Collector Cutoff

Current

Emitter Cutoff Current

Static Forward Current

Transfer Ratio

Base-Emitter Voltage

Collector-Emitter

Saturation Voltage

Small-Signal

Common-E miner

Forward Current

Transfer Ratio

Transition Frequency

Common-Base

Open-Circuit

Output Capacitance

Real Part of

Small-Signal

Common-Emitter

Input Impedance

TEST COI

I c - 10 MA,

I C " 10 mA,

I g - 10 MA,

V c B - S O V .

VcB - 50 V.

V E B - 3 V,

V c E - l O V ,

V c E - l O V ,

VcE - 10 V,

V C E - 1 0 V ,

VcE - 10 V,

VcE - 1 V, I g - 15mA.

l B - 5 0 m A ,

Ig • 15 mA,

Ig - 50 mA,

VcE • 20 V,

VcE " 20 V,

V c B - l O V ,

VcE • 20 V,

yiDITIONS

T O - 5 -

T O - 1 8 -

I g - O

I B ' O , See Note 6

i c - 0

l E - 0

' E " 0 . T A - 1 5 0 * C

I C " 0

I c - 100 MA

Ic - 1 mA

Ic - 10 mA

I c - 150 mA

I c - 500mA

I c * 150 mA

I c - 150 mA

Ic - 500 mA

I c - 150 mA

Ic - 500 mA

See Note 6

See Note 6

S*a Note fi

I C - 2 0 m A , f - 1 0 0 MHz

I C - 2 0 m A . See Note 7

l g - 0 , f - I M H z

I C - 2 0 m A , f - 3 0 0 MHz

2N2217

2N2220

MIN MAX

60

30

5

10

10

10

12

17

20 60

10

^J3

0.4

2.5

250

8

60

2N2218

2N2221

MIN MAX

60

30

S

10

10

10

20

25

35

40 120

20

20

1.3

2.6

0.4

1.6

2.5

250

8

60

2N2219

2N2222

MIN MAX

60

30

5

10

10

10

35

50

75

100 300

30

SO

1.3

2.6

0.4

1.6

2.5

250

8

60

UNIT

V

V

V

nA

MA

nA

V

V

MHz

pF

n

N O T E S : 6 . Theee parameters mutt be maaaured ut int pulte techniques, t ^ - 300 MS. Outy cycle < 2%.

7 . To obtain f x . the h f , l rewonse with frequency it extrapolated at the rete of - 6 d B per octave from f -

frequency at which h f , I " 1 .

switching characteristics at 25"C free-air temperature

100 MHz to the

P A R A M E T E R

Delay T ime

_«f_ Rise T ime

Storage T ime

Fall T ime

TEST C O N D I T I O N S ^

V C C - 3 0 V , I c - 1 5 0 m A , l B ( i ( - 1 6 m A .

V s E l o f f ) • ~ 0 - 5 V , See Figure 1

V c c - 3 0 V ,

1 0 ( 2 ) - - 1 5 m A ,

I C " 1 5 0 m A , l B ( i ) - 1 5 m A ,

See Figure 2

T Y P

15

190

23

U N I T

ns

'Voltaoe and current values thown are nominal, e.ect values vary .lightly with t r .nt i . tor pararr^tar..

• JEDEC reyitrered data

TYPES 2N2217 THRU 2N2222. 2N2218A, 2N2219A, 2N2221A. 2N2222A

N-P-N SILICON TRANSISTORS

116

2N2218A, 2N2219A. 2N2221A, 2N2222A

•electrical characteristics at 25°C free-air temperature (unless otherwise noted)

PARAMETER

V(BR)CB0

V(BR)CEO

V(8R)EB0

>CBO

'CEV

'BEV

•EBO

hFE

VBE

VcE(sat)

' 'ie

Nt

»»r«

^'oe

^ f e |

fr

Cote

Cjte

^ietreal)

^b'Cc

Collector-Base Breakdown Voltage

Collector-Emitter Breakdown Voltage

Emitter-Base Breakdown Voltage

Collector Cutoff Current

Collector Cutoff Current

Base Cutoff Current

Emitter Cutoff Current

Static Forward Current

Transfer Ratio

Base-Emitter Voltage

Collector-Emitter Saturation Voltage

Small-Signal Commort-Emitter

Input Impedance

Small-Signal Forward Current

Transfer Ratio

Small-Signal Common-Emitter

Reverse Voltage Transfer Ratio

Small-Signal Common-Emitter

Output Admittance

Snull-Signal Common-Emitter

Forward Current Transfer Ratio

Transition Frequency

Common-Base Open-Circuit

Output Capacitance

Common-Base Open-Circuit

Input Capacitance

Real Pan of Small-Signal

Common-Emitter Input Impedance

Collector-Base Time Constant

TEST CO

I c - I O M A ,

I C " 10mA.

I E - I O M A ,

VcB - 60 V.

VcB - 60 V,

VcE - 60 V,

VcE • 60 V,

V E B - 3 V ,

VcE • 10V,

V c E - l O V ,

V c E - 10 V,

V c E - l O V ,

VcE - 10 V,

VcE - 1 V,

V c E - l O V ,

T A - - 5 5 * C

IB - 15mA.

I B - 50 mA,

I B - 15mA,

Ig - 50 mA,

V c E - l O V ,

V c E - l O V ,

V c E - 10 V,

V c E - 10 V,

VcE - 10 V,

V C E - 1 0 V ,

V c E - 10 V,

VcE - 10 V,

VcE " 20 V,

VcE - 20 V,

V C B - 1 0 V ,

V E B - 0 . 5 V,

VcE - 20 V,

VcE - 20 V,

NOITIONS

T O - B -

T O - 1 8 -

l E - 0

I B ' 0, See Note 6

i c - 0

lE-0 l E " 0 , T A - . 150°C

V B E - - 3 V

V B E - - 3 V

I c - 0

I c - I O O M A

I C " 1 mA

I C " 10 mA

I c - 150 mA

Ic - 500 mA

I c - 150 mA

I c - 10 mA,

Ic " 150 mA

Ic - 500 mA

Ic - 150 mA

Ic - 500 mA

Ic • 1 mA

• C - 10 mA

I c - 1 mA

•C - 10 mA

I c * 1 mA

Ic - 10 mA

I c - 1 mA

Ic " 10 mA

See Note 6

See Note 6

See Note 6

f - 1 kHz

I C " 20 mA, f - 100 MHz

I C " 2 0 m A , See Note 7

I E - 0, f - 100 kHz

I c - 0 , f - 100 kHz

Ic - 20 mA, f - 300 MHz

I C - 2 0 m A , f - 3 1 . 8 MHz

2NZ218A

2N2221A

MIN MAX

75

40

6

10

10

10

- 2 0

10

20

25

35

40 120

25

20

15

0.6 1.2

2

0.3

1

1

0.2

30

50

3.5

1

150

300

5 x 1 0 - *

2 .5x10-*

3

10

15

100

2.5

250

8

25

60

150

2N2219A

2N2222A

MIN MAX

75

40

6

10

10

10

- 2 0

10

35

50

75

100 300

40

50

35

0.6 1.2

2

0.3

1

2

0.25

50

75

8

1.25

300

375

8x10~*

4x10—*

5

25

35

200

3

300

8

25

60

150

UNIT

V

V

V

nA

MA

nA

nA

nA

V

V

kn

*imho

MHz

pP

pF

n

PS

N O T E S : 6. These parametert mutt be meatured uting pulte techniquet. tyy - 300 M». duty cycle < 2%.

7. To obtain fy. the ^ f , | retoonte with frequency ii extrapolated at the rate of - 6 d B oer octave from f - 100 M H i to the

frequency at which ^ f , | - 1 .

' J E D E C registered data

117 TYPES 2N2217 THRU 2N2222. 2N2218A. 2N2219A. 2N2221A. 2N2222A N-P-N SILICON TRANSISTORS

•operating characteristics at 25*'C free-air temperature

PARAMETER

F Spot Noise Figure

TEST CONDITIONS

T O - S -

T O - 1 8 -

VcE - 10V. I c - I O O M A , R Q • 1 kn , f - 1 kHz

2N2218A

2N2221A

MAX*

2N2219A

2N2222A

MAX

4

UNIT

dB

^switching characteristics at 25°C free-air temperature

PARAMETER

t(j Delay Time

tf Rise Time

TA Active Region Time Constant^

tf Storage Time

tf Fall Time

TEST CONDITIONSf

T O - 5 -

TO-18 -

V c c - 3 0 V. I c - 1 5 0 m A , l B ( , ) - 1 5 m A .

^BEJoff) " -^-^ V, See Figure 1

V c c - 3 0 V, I c - 1 5 0 mA. l B ( i ) - 1 5 m A ,

'8(2) ' — 1 5 ' " A . See Figure 2

2N2218A

2N2221A

MAX

10

25

2.5

225

60

2N2219A

2N2222A

MAX

10

25

2.5

225

60

UNIT

ns

ns

ns

ns

ns

^Voltage and current values thown are nominal; exact values vary slightly with transistor parametert.

t\jnamr the given conditions r A it equal to — . 10

APPENDIX F

SPECIFICATIONS FOR THE BS170 TRANSISTOR

118

BS107HBS170

200V N-Channel Enhancement-Mode

MOSPOWER

119

Siliconix

FEATURES • High Voltage • No Second Breakdown • High Input impedance • internal Drain-Source Diode • Very Rugged: Excellent SOA • Extremely Fast Switching • Reduced Component Count • Improved Performance • Simpler Designs • improved Reliability

APPLICATIONS • Telephone i-landsets • Switching Regulators • Solid State Switching

ABSOLUTE MAXIMUM RATINGS

Product Summary Part

Number

BS107

BS170

B V M . Volts

200

60

''OSION) lOMM)

28

5

Package

TO-92

TO-92

'-^4

PsrsnMlsf

vos VQGR

>0

VQS

P D @ T C - 25*0

T j

T«g

Drain - Souros Voitag*

Drain- Oat* Vottage ( R Q S - 1 Mn)

Continuous Drain Currant

Gala - Sourea Voitaga

Max. Po«*ar Dissipation

Opal aivig and

Storaga Tamparatura Ranga

Laad Tampaniijra

BS170

60

60

500

±2SV

0833

-5510 150

300

BS107

200

200

120

±25V

0500

-5510 150

300

UnKs

V

V

mA

V

w

•c

•c

PACKAGE DIMENSIONS

PINJ —SottfM Hfi't ^ Qate PIN I — Drain

fAu omtrntommmiujiitTam

T 0 «

1 a •

ELECTRICAL CHARACTERISTICS (Tc = 25X unless otherwise noted) ±zo

Paramaiar

- Dram-Sourca Brsakoown "'OSS

vonaga

^os<eii Qata-ThrashoW Vottaga

IQSS Gata-Body Laakaga

IQSS Zaro Gala Vottaga Dram

Currant

>OS>l

V Q 3 ion) Static Drain-Source On-Stats

Vottagel

Rgg^, Static Orain-Sourca On-Stata

Rasotancal

Typ«

BS170

BS107

BS170

BS107

8S17aBSl07

BS170

BS107

BS107

BS170

BS107

BS170

BS107

Mn.

60

200

0-B

Typ.

1-8

Max.

3 0

10

0-5

0O3

1

1

056

5

1

Unlta

V

V

nA

MA

nA

nA

V

V

n n

TMt Conditions

V G S - O . I Q - I O O M A •

Vos-Vas. 'o-• '"»*

V o s - l 5 V . V o s . 0

V O S - 2 S V , V Q S - 0

V o s - 1 » V . V 6 5 - 0

Vos-70V.Vos-0-2V

V O S - 1 0 V . I D - 0 - 2 A

VQS " 2-8V. lo - 20mA

V Q S - ' ' 0 . I O - 0 - 2 A

VQ5-2-8V.I(,-20mA

DYNAMIC

THERMAL RESISTANCE

gls

c*

^{an)

^im

Forward Transductancel

Input Capacitanca

Turn-On Time

Tum-OtfTima

BS170

BS170

BS107

BS170

BS107

BS170

BS107

200

60

58

10

10

10

10

mS(n)

pF

pF

ns

ns

ns

ns

Vos"10^' 'D = 0-2A

Vos- lOV

V0S-20V

IO-0-2A

I O - 0 2 A

V G S - 0

t - i M H z

f^njA Junction-to-Air AH 150 •CW

1. Pulse Test

FIELD EFFECT TRANSISTORS - TDL

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