Experimental Demonstration of OCDMA and OTDMA …digitool.library.mcgill.ca/thesisfile19245.pdf ·...

Experimental Demonstration of OCDMA and

OTDMA PONs with FEC and Burst-Mode

Reception

Noha Kheder

Department of Electrical and Computer Engineering McGill University, Montréal, Canada

February 2008

A thesis submitted to McGill University in partial fulfillment of

the requirements of the degree of Master of Engineering

© Noha Kheder, 2008

i

To my parents….

ii

Abstract

Passive optical networks (PONs) are a promising economic solution in

delivering data to the end-user. We demonstrate experimentally the uplink

of a spectral-amplitude-coded optical code division multiple access (SAC-

OCDMA) and an optical time division multiple access (OTDMA) PON, with

burst-mode reception. The receiver performs clock and data recovery

(CDR), phase acquisition and forward-error-correction (FEC).

Using FEC we demonstrate an error-free 7x622 Mbps uplink of an

incoherent SAC-OCDMA PON, while operating at a relatively low power of

around -24 dBm. In going to from a back-to-back architecture to a local

sources PON configuration the penalty introduced is less than 1 dB.

We show that the burst-mode functionality of the receiver enables

instantaneous phase acquisition and zero packet loss. However, it

introduces a power penalty of around 1dB, which is the price to pay to

accommodate bursty traffic and achieve instantaneous phase acquisition

using zero bits of preamble.

iii

Résume

Les réseaux optiques passifs (PONs - “passive optical networks”) sont

une solution économique pour la livraison de données à l'utilisateur final.

Nous démontrons expérimentalement l'uplink d'un lien CDMA (accès

multiples à répartition des codes) et TDMA (accès multiples à répartition

dans le temps) PON, avec réception qui supporte un trafic de paquets en

rafale (“burst-mode traffic“). Le récepteur peut rétablir rapidement les

impulsions d’horloge et les données (CDR “clock-and-data recovery“) et

effectuer la correction d’erreurs sans voie de retour (FEC - “forward error

correction”).

En utilisant un FEC, nous démontrons une transmission ‘uplink’ sans-

erreur à 7x622 Mbps d'un lien SAC-OCDMA incohérent PON, opérant à

une puissance relativement basse d'environ -24 dBm. D’une configuration

jumelée à une configuration de PON, la pénalité de puissance est de

moins de 1 décibel.

Nous montrons que le récepteur permet l’acquisition de phase de

façon instantanée sans perte de paquets donnant une pénalité de

puissance d'environ 1 décibel, le prix à payer pour supporter le trafic de

paquets en rafale.

iv

Acknowledgements

First and foremost, I would like to thank my supervisor Professor David

Plant, for his guidance and support. Besides teaching me the skill of conducting

research, he has taught me a great deal on a personal level. I thank him for his

always insightful, valuable and genuine advice. I very much appreciate his

support and encouragement in every step of the way. It has been an enriching

experience working with him, as I have learned from a great researcher, leader

and person.

Next I would like to thank those who have contributed to the research

carried out in this thesis. I thank Bhavin Shastri for his collaboration on the SAC-

OCDMA and GPON projects; Ziad El-Sahn for putting together the SAC-OCDMA

system; Ming Zeng for her collaboration on the SAC-OCDMA project; Nick Zicha

for his help on the GPON project; Professor Leslie Rusch for letting me work in

her lab; Professor Lawrence Chen, for letting me use some of his equipment and

for his valuable technical advice; Julien Faucher for having designed the burst-

mode receiver, and for his help; Odile Liboiron-Ladouceur for her help in the lab,

and for aiding in translating the Absract; Christian Habib, Irina Kostko, Varghese

Baby, Dragos Cotruta, Nikolaos Gryspolakis and Rhys Adams for their help with

using equipment, technical advice and stimulating discussions; Joshua Schwartz,

Carrie Serban, Kay Johnson and Christopher Rolston for their valuable and

timely administrative support.

v

This research has been the product of working in a very friendly

environment, to which I first thank Professors David Plant, Andrew Kirk,

Lawrence Chen and Martin Rochette for creating such a dynamic and interesting

group, and I also thank all my colleagues who have made the past two years an

unforgettable experience.

I would like to express my thanks and great appreciation to my dear friend

Marija Nikolic and who has been very attentive and supportive. I thank her for her

valuable genuine advice, for her loyal listening ears and kind words that kept me

motivated. I would also like to thank my friend and colleague Alaa Hayder for all

his help and support and for the great conversations we shared while working in

the lab. I also thank Jelena Jevtic, Sarah Abdel Hameed and Ruba Kayali for all

their support, and for their great frendships.

Finally, I end this by thanking my parents Mohamad Kheder and Mervat

Faheem for their endless love and support over the years. I thank them for

believing in me and inspiring me to get to the point where I am today. It is

because of their love, dedication and encouragement that I am able to achieve

and to them I am eternally grateful. I also thank my brother Ahmed Kheder for

being a great friend, for his love and support, and for his youthful wisdom and

great sense of humor.

vi

Contents

Chapter 1 - Introduction................................................................................1

1.1 Motivation......................................................................................................... 1

1.2 Thesis objectives and contributions .................................................................. 4

1.3 Thesis overview ................................................................................................ 5

Chapter 2 - Literature Review.......................................................................7

2.1 OCDMA............................................................................................................ 7

2.2 Burst-mode receivers: amplitude and phase recovery .................................... 12

2.3 Forward error correction ................................................................................. 14

Chapter 3 - Demonstration of a 7 user SAC-OCDMA Uplink with FEC

and Burst-Mode Reception.........................................................................16

3.1 Introduction..................................................................................................... 16

3.2 SAC-OCDMA burst-mode receiver ............................................................... 18

3.3 SAC-OCDMA uplink and experimental setup ............................................... 23

3.4 Results and discussion .................................................................................... 30

3.4.1 BER Performance ............................................................................ 31

3.4.2 PLR Performance............................................................................. 37

3.4.3 CID Immunity.................................................................................. 40

3.4.4 Eye Diagrams................................................................................... 42

3.5 Conclusion ...................................................................................................... 45

vii

Chapter 4 - Performance Analysis of a Burst-Mode Receiver in GPON

Uplink ...........................................................................................................47

4.1 Introduction..................................................................................................... 47

4.2 GPON burst-mode receiver............................................................................. 48

4.3 GPON test-bed................................................................................................ 50

4.4 Results and discussion .................................................................................... 53

4.4.1 BER Performance ............................................................................ 53

4.4.2 PLR Performance............................................................................. 56

4.4.3 CID Immunity.................................................................................. 61

4.4.4 Eye Diagrams................................................................................... 62

4.5 Conclusion ...................................................................................................... 63

Chapter 5 - Conclusion ...............................................................................66

References ...................................................................................................68

viii

List of Figures and Tables

Figures Fig. 1.1 Passive Optical Network……………………………………………...2

Fig. 1.2 Bursty uplink traffic in a passive optical network…………...………3

Fig. 2.1 Spectral Amplitude Coded OCDMA…………………………..……10

Fig. 2.2 Burst-mode receiver functionalities. (a) AGC performing amplitude

recovery. (b) BM-CDR handling phase recovery…………………………..13

Fig. 3.1 SAC-OCDMA receiver block diagram……..………………………18

Fig. 3.2 Test signal emulating bursty uplink traffic…………………………20

Fig. 3.3 Local sources PON architecture……………………………………24

Fig. 3.4 SAC-OCDMA PON uplink experimental setup……………………25

Fig. 3.5 Balanced receiver…………………………..………………………..27

Fig. 3.6 Illustration of balanced detection…………………………..............29

Fig. 3.7 BER vs. useful power for different number of users using a global

clock…………………….............................................................................31

Fig. 3.8 BER vs. useful power for different number of users: comparing

global clock and recovered clock………………….……………….………...33

Fig. 3.9 BER vs. useful power with and without FEC ......…………..…………35

Fig. 3.10 BER vs. useful power for a single user and fully-loaded systems:

comparing PON and back-to-back architectures..…………………………36

ix

Fig. 3.11 PLR vs. phase difference without CPA for different number of

users………...............................................................................................38

Fig. 3.12 PLR vs. number of users……………………………....................40

Fig. 3.13 PLR vs. length of CID………………………………………………41

Fig. 3.14 Eye diagrams showing the response of the CDR to bursty

traffic…………………………………………………………………………….44

Fig. 4.1 Receiver block diagram……………………………………………..49

Fig. 4.2 GPON uplink experimental setup…………………………………..51

Fig. 4.3 BER vs. useful power of the GPON uplink: experimental results

with and without FEC and FEC simulation results…………………………54

Fig. 4.4 PLR vs. phase difference (a) Back-to-back configuration with CDR

for different preamble lengths. (b) Comparison between back-to-back and

PON configurations with and without CPA with 0 bit preamble…………..57

Fig. 4.5 PLR vs. useful power for continuous and burst-mode reception..59

Fig. 4.6 PLR vs. useful power for CDR (with 0 bit preamble), CDR (with 28

bit preamble) and burst-mode receiver (with 0 bit preamble)……………..60

Fig. 4.7 PLR vs. number of CIDs…………………………...………………..61

Fig. 4.8 Eye diagrams showing the response of the CDR to bursty

traffic…………………………………………………….………………………62

Tables Table 3.1 BIBD Codes Used………………………………………………….23

x

Abbreviations

AGC Automatic Gain Control

ASIC Application specific integrated circuit

BER Bit error ratio

BERT Bit error rate tester

BIBD Balanced incomplete block design

BM-CDR Burst-mode clock and data recovery

CPA Clock phase aligner

CDR Clock and data recovery

CIDs Consecutive identical digits

DCF Dispersion compensation fiber

DFB Distributed feedback

EAM Electro-absorption modulator

EDFA Erbium-doped fiber amplifier

EOM Electro-optic modulator

FBG Fiber Bragg Grating

FEC Forward error correction

FPGA Field-programmable gate array

FTTx Fiber-to-the-home/curb/neighborhood

GPON Gigabit passive optical network

IPCC In-phase cross-correlation

MAI Multiple access interference

xi

NRZ Non-return-to-zero

OCDMA Optical code division multiple access

ODN Optical distribution network

OLT Optical line terminal

OTDMA Optical Time division multiple access

ONU Optical network unit

PLL Phase-locked Loop

PLR Packet loss ratio

PON Passive optical network

PPG Pulse pattern generator

PRBS Pseudo-random bit sequence

QoS Quality of service

RF Radio frequency

RS Reed-Solomon

SAC-OCDMA Spectral amplitude coded OCDMA

SMF Single-mode fiber

SNR Signal-to-noise ratio

SONET Synchronous optical network

TIA Trans-impedance amplifier

VOA Variable optical attenuator

WDMA Wavelength division multiple access

WHTS wavelength-hopping time-spreading

xii

1 Introduction

Chapter 1

Introduction

1.1 Motivation

Today the demand for bandwidth is growing with the increasing usage

of multimedia applications, such as video streaming, voice-over-IP and

gaming, among others. Optical fibers can accommodate this growth. In

delivering data to the end-user, passive optical networks (PONs) are a

promising economic solution to alleviate the bandwidth problem in the

access network [1 - 4].

A PON is a point-to-multi-point network with a physical tree topology as

illustrated in Fig 1.1. The optical line terminal (OLT) is located at the root,

being the central office (CO), and optical network units (ONUs) are located

at the branches near the end users. The optical distribution network

(ODN) is composed of passive components, such as passive couplers and

splitters. Since they use no powered equipment, PONs offer a cost-

effective solution for fiber-to-the-home/curb/neighborhood (FTTx) that

require little network management and infrastructure upgrades.

1

1 Introduction

Fig. 1.1 Passive Optical Network

Since in a PON, the upstream direction is multi-point-to-point, several

users (ONUs) may transmit simultaneously on a single channel towards a

single OLT; therefore a channel separation technique is necessary to

2

1 Introduction

avoid collisions. The three general multiple access techniques are: optical

time-division-multiple-access (OTDMA), wavelength-division-multiple-

access (WDMA) and optical code-division-multiple-access (OCDMA).

Existing PON standards are based on OTDMA and can serve up to 32 or

64 users at an aggregate bit rate of 1.25 Gbps [1]. To meet future

bandwidth demands, WDMA [5 - 8] and OCDMA [9 – 11] PONs are

proposed to increase the capacity of the existing OTDMA PONs.

Since in a PON uplink packets may travel through different lengths of

fiber they undergo different attenuation and delays. As a result packets

may arrive at the OLT with varying amplitudes and phases as depicted in

Fig 1.2. This imposes a requirement on receivers at the OLT to acquire

the phase and amplitude of an incoming packet in order to accommodate

the bursty nature of uplink traffic.

Fig. 1.2 Bursty uplink traffic in a passive optical network.

3

1 Introduction

1.2 Thesis objectives and contributions

The objective of this thesis is to study the performance of PONs using

burst-mode reception. We focus on OCDMA and OTDMA multiplexing

techniques for PONs. The research is carried out as two experimental

demonstrations 1) an experimental demonstration of a 7x622 Mbps

incoherent spectral amplitude coded OCDMA (SAC-OCDMA) PON using

burst-mode reception, and 2) a performance analysis of burst-mode

receivers in Gigabit PON (GPON). The burst-mode receiver performs

clock-and-data (CDR) recovery, forward-error-correction (FEC) and clock-

and-phase alignment (CPA). The original contributions of this thesis are

• Investigating and analyzing the performance of a burst-mode

receiver with CDR and CPA in GPON and in a SAC-OCDMA PON.

It is found that the burst-mode functionality of the receiver allows for

instantaneous phase acquisition, but introduces a slight power

penalty. However, this power penalty is a reasonable compromise

to accommodate the bursty nature of uplink PON traffic while

leaving all the delimiter bits to be used for amplitude recovery or

increasing the information rate.

• An experimental demonstration of a 7x622 Mbps SAC-OCMDA

PON without a global clock, and with FEC. Error-free operation

using FEC is achieved for a fully-loaded system while using a

recovered clock. This shows that SAC-OCDMA, with balanced

4

1 Introduction

detection and cancellation of multiple access interference (MAI), is

a good candidate for employment in PONs.

The burst-mode receiver used in the SAC-OCMDA and OTDMA

experiments is designed by Julien Faucher; and the SAC-OCDMA test-

bed is assembled by Ziad El-Sahn. The author’s contributions are the

assembly of the OTDMA testbed; the testing and investigation of the

OTDMA setup with Bhavin Shastri; and the testing and investigation of the

SAC-OCMDA setup with Ziad El-Sahn, Bhavin Shastri and Ming Zeng.

This work has contributed in the following conference papers and journals

[12 - 16], of which [12] is a first authorship.

1.3 Thesis overview

The remainder of this thesis is organized as follows. Chapter 2

presents an overview of OCDMA, FEC and burst-mode receivers in three

sections. Each section presents background theory and fundamentals, as

well as recent progress in the respective fields.

In chapter 3, we present the demonstration of an incoherent SAC-

OCDMA system with a standalone receiver. The receiver performs CDR,

FEC and CPA; the design and functionality of each module is explained

and the SAC-OCDMA system demonstrator is described in detail. The

performance of the system is quantified in terms of bit-error-rate (BER) to

assess the impact of adding users on the network. We use the coding gain

5

1 Introduction

from FEC to increase the number of supported simultaneous users and

achieve error-free operation for a fully loaded system. Packet-loss-ratio

(PLR) and consecutive-identical-digits (CID) immunity results, as well as

the obtained eye diagrams, are also presented.

In chapter 4, we present a performance analysis of a burst-mode

receiver in GPON. A brief overview of the receiver is presented, along with

the GPON experimental test-bed. The PLR performance of the system is

analyzed to characterize the functionality of the CPA and quantify its

performance and the power penalty incurred by burst-mode reception

using twice over-sampling. BER measurements and the CID immunity of

the receiver, as well as eye diagrams, are also presented.

Finally in chapter 5, the obtained results are summarized.

6

2 Literature Review

Chapter 2 Literature Review

2.1 OCDMA

The two main multiple access techniques that are currently widely

used are OTDMA and WDMA [18]. In OTDMA PONs each ONU can only

transmit in its specified transmission window (time slot), whereas in

WDMA PONs each ONU operates on a different wavelength [19]. An

alternative is OCDMA PON, in which users are assigned a specific code in

the wavelength-time space. The receiver is able to de-multiplex the

received signal in the presence of other channels by knowing the code

used at the transmitter.

In OTDMA the performance of the system is limited by the time-serial

nature of the multiplexing scheme. It offers a large number of node

addresses; however, this requires that the receiver operates at the total bit

rate of the system, which is roughly equal to the number of nodes

connected times the data rate per node. OTDMA also requires strong

centralized control introducing additional latency and overhead [18, 20].

WDMA require a significant amount of coordination between the nodes or

a dedicated control channel which wastes bandwidth and introduces

latency [20].

7

2 Literature Review

OCDMA, on the other hand, can provide multiple access without the

need for very high-speed electronic data processing device and

wavelength-sensitive components, as are needed in OTDMA and WDMA

networks respectively [21]. In OCDMA, all active users share the same

wavelength and time domain space, hence providing a fair division of the

bandwidth, as opposed to OTDMA and WDMA where only a small portion

of the bandwidth is allocated to each user. Moreover, OCDMA systems

can operate asynchronously without centralized control, and allow for all-

optical processing. They can achieve very low latencies since they do not

suffer from packet collisions [9, 18, 20]. The capacity of OCDMA is soft-

limited since the BER is dependent on the number of users supported by

the network, as opposed to the hard-limited nature of OTDMA and WDMA

networks where capacity is limited by the number of time or wavelength

slots respectively. This provides flexibility in controlling the Quality of

Service (QoS) and the possibility of providing soft-capacity on demand [9,

18, 20]. It is a promising approach in upgrading existing PONs since the

PON infrastructure need not be upgraded; it has simple OLT and ONU

configurations and requires no synchronization [9 - 11].

OCDMA systems can be divided into two main categories: incoherent

OCDMA and coherent OCDMA. Incoherent schemes achieve encoding

through intensity modulation. As for coherent OCDMA the encoding is

performed on the phase of the signal; this is done through the use of a

highly coherent wideband source. To recover the user’s data coherent

8

2 Literature Review

reconstruction of the signal is required. Within each type, we can further

classify OCDMA systems based on the way in which the coding is applied

as follows [18, 22]:

• Incoherent OCDMA Approaches

o Spectral Amplitude Coding

o Spatial Coding

o Temporal Coding

o Hybrid Coding

• Coherent OCDMA Approaches

o Spectral Phase Coding

o Temporal Phase Coding

Spectral amplitude coding involves spectrally decomposing a

broadband light source into spectral slots, and the intensity of each slot is

modulated such that the slot is either ‘on’ or ‘off’ depending on the user

code being applied [11, 23, 24]. In chapter 3 of this thesis, we focus on

this type of encoding, which is depicted in Fig. 2.1.

Spatial coding uses multi-core fibers or multi-fiber systems to create

spatial patterns; each user transmits patterns of optical pulses distributed

over the fibers. This scheme is suitable for parallel transmission and

access of images [25 – 27].

9

2 Literature Review

Fig. 2.1 Spectral Amplitude Coded OCDMA. The figure shows two codes of weight 3 and length 8; (a) user’s code: 01101000 (b) user’s code: 01010001

Temporal coding involves splitting each bit into N smaller time

intervals, called chips. Each time chip is then transmitted as a ‘1’ (i.e.

contains an optical pulse) or a ‘0’ depending on the user code being

applied [28 – 30]. This is historically one of the first approaches to be

implemented [18].

Hybrid encoding is a multi-dimensional encoding in which light is

encoded in a combination of time, space and frequency. There has been

10

2 Literature Review

research in the space-wavelength [31], space-time [32], and the space-

wavelength-time [33] domains; however, the more common scheme is

wavelength-hopping time-spreading (WHTS) which spreads the codes in

the wavelength-time domain. In this method channels of different

wavelengths are delayed differently using optical delay lines, so that each

user has a unique two-dimensional code in the wavelength-time space

[34, 35].

As for coherent approaches to OCDMA, the optical phase can be

controlled in the frequency domain or time domain. Spectral phase coding

involves splitting the signal into N spectral bins, where the phase of each

bin is manipulated according to the desired user’s code. Spectral phase

coding results in a spreading of the signal in the time domain which makes

it appear noise-like [36 – 37]. Temporal phase coding, on the other hand,

creates N copies of the optical pulse through optical delay lines to create

equally spaced pulses in the time domain. A relative phase is then added

to each delayed pulse [38 – 39].

Fiber non-linearity leads to the deterioration in information capacity for

both coherent and incoherent OCDMA systems; however it acts differently

on each type. The performance deterioration in coherent systems is

mainly due to cross-phase modulations, whereas in incoherent systems it

is mainly due to four-wave mixing [18]. In incoherent OCDMA, if the

frequency channels are strongly separated, the non-linearity will have little

or no effect, therefore a properly designed incoherent OCMDA system

11

2 Literature Review

will show a better BER performance than an coherent one [18]. In our

demonstration in Chapter 3, an incoherent OCDMA system is used.

OCDMA can be synchronous, where synchronization is achieved

through a system global clock throughout the network or alternatively

asynchronous (no synchronization). Most OCDMA demonstrations have

used a global clock [10, 40 - 42]. In our demonstration, asynchronous

transmission is used, whereby the clock from the incoming data is

recovered at the receiver side using a commercial SONET CDR module.

Whichever approach to OCDMA is used, the proper choice of a

suitable all-optical technology to implement the coding and decoding is

critical [18]. Advances in writing Fiber Bragg Gratings (FBGs) [11, 43 – 45]

have made this a low-cost, compact and scaleable solution for OCDMA

encoding and decoding, that are suitable for coherent and incoherent

OCDMA. FBGs can be used as simple optical filters for encoding schemes

such as SAC-OCDMA, and in generating phase-encoded signals. Using

the combined wavelength selective and dispersive properties of FBGs 2D

wavelength-time coding schemes can also be implemented.

2.2 Burst-mode receivers: amplitude and phase recovery

In a PON uplink, as traffic travels towards the OLT, packets originating

from different ONUs travel through different lengths of fiber thus getting

attenuated and delayed by different amounts. Therefore receivers at the

OLT must be able to handle such bursty traffic. Receivers must include an

12

2 Literature Review

Automatic Gain Control (AGC) module to handle amplitude acquisition [46

– 50]. The AGC module adjusts the threshold at the beginning of every

incoming burst according to the received power level, in order to sample

the data correctly. In addition to performing AGC, burst-mode receivers

must be able to lock to the frequency and acquire the phase of the

incoming packet in a short time; this is done through a CDR module. The

functionality of burst-mode receivers’ AGC and CDR modules is depicted

in Fig 2.2.

Fig. 2.2 Burst-mode receiver functionalities. (a) AGC performing amplitude recovery. (b) BM-CDR handling phase recovery.

PONs are meant to provide long physical reach and a large splitting

ratio (i.e. support a large number of users). This imposes requirements of

a large dynamic range, fast response time and high sensitivity on the AGC

modules of burst-mode receivers at the OLT [46, 47]. AGC ICs based on

trans-impedance amplifiers (TIAs) are implemented in feedback or feed-

forward architectures [47]. In feedback architectures, the threshold is

13

2 Literature Review

determined completely from the preamble field, and is then held constant

for the rest of the packet [51, 52], whereas in the feed-forward

architectures the threshold is adaptively determined according to the input

data [53 – 55]. Feedback architectures are more stable due to the

feedback, but they require a differential input/output preamplifier, whereas

feed-forward architectures can use a conventional DC coupled amplifier,

but need to be carefully designed to avoid oscillation [46].

As for the frequency and phase acquisition module in a burst-mode

receiver, several approaches have been proposed. Burst-mode CDRs

based on wide-band phase-locked loops (PLLs) can achieve fast phase

acquisition by adaptively controlling the loop bandwidth with phase, so that

when the phase error is large the loop bandwidth is increased [56, 57]. A

higher PLL bandwidth translates into shorter settling time, hence faster

phase acquisition. As an alternative technique to achieve fast phase

acquisition, gated oscillators have been widely used [58 – 61]. This

method achieves lock on the first data transition instantaneously. Another

method for burst-mode CDRs is using over-sampling; each bit is over-

sampled, and the best sample is chosen through a phase-picking

algorithm [62 – 65].

2.3 Forward error correction

Borrowed from the wireless world, FEC in optical communications is

used to correct errors that may occur during transmission, thus improving

14

2 Literature Review

system performance by reducing the BER. FEC involves adding

redundancy to the information, which is used to detect and correct errors.

The first FEC codes proposed are block codes [66, 67]. Block coding

performs the coding on one block at a time; this family of codes includes

Reed-Solomon (R-S) code which is adopted in the ITU-T Rec. G.709

standard [68]. The following demonstrate implementations of R-S codes

in OCDMA systems [40, 56, 69 – 71]. Convolutional FEC codes [72 – 74]

and concatenated codes [75 – 78] in which convolutional codes are

concatenated with short R-S codes are also reported.

Recently, researchers are investigating the use of soft decision [n 79 –

82], where receivers can recognize intermediate levels between a ‘1’ and

a ‘0’ for each received bit. With soft-decision, the decoder provides an

integer indicating how likely it is that the received bit is a ‘1’ or a ‘0’ based

on how far the received power is from the threshold; one such coding

scheme is Turbo-coding.

15

3 Demonstration of a 7 user SAC-OCDMA Uplink with FEC and Burst-Mode Reception

Chapter 3 Demonstration of a 7 user SAC-OCDMA Uplink with FEC and Burst-Mode Reception

3.1 Introduction

Since in a PON uplink multiple ONUs transmit to a single OLT at the

CO, a multiple access scheme is required to avoid collision and govern

the traffic. OCDMA is a promising candidate for deployment in PONs

carrying bursty and asynchronous traffic. It combines the large bandwidth

of the fiber medium with the flexibility offered by CDMA encoding to realize

high speed connectivity. Moreover, among the advantages of OCDMA is

that it can support a large number of simultaneous active users in an

asynchronous environment without centralized control [18]. Unlike

OTDMA and WDMA, OCDMA requires no time and wavelength

management respectively at all nodes, therefore it has simple ONU and

OLT configurations [9 - 11, 18]. There is a compromise between the

network capacity, in terms of the number of users it can support, and the

QoS since the BER is dependent on the number of users. However, the

soft-capacity of OCDMA allows growing the client base without extensive

16


upgrades to the infrastructure which gives flexibility in network design and

upgrade.

In this demonstration, we focus on incoherent SAC-OCMDA because

of its ability to cancel multiple access interference (MAI) using balanced

detection from a normal decoder and its complementary decoder when

codes with fixed in-phase cross-correlation (IPCC) are used [83]. It also

permits the use of low-speed electronics operating at the bit rate,

compared to 2-D OCDMA which requires electronics operating at the chip-

rate [84, 87]. Moreover, advances in writing FBGs have made possible the

design of low cost and compact encoders/decoders well adapted to PONs

[18].

In this chapter, a 7 user spectral-amplitude-coded OCDMA uplink is

demonstrated using a burst-mode receiver that performs CDR, CPA and

FEC. Section 3.2 describes the design of the receiver used. Section 3.3

presents the implementation of the 7-user SAC-OCMDA uplink and the

experimental set-up. The BER and PLR performance, as well as the CID

immunity of the system using the receiver are measured and the results

obtained are presented and analyzed in section 3.4. Results are shown for

a back-to-back configuration and briefly compared to a local sources PON

architecture. Finally, section 3.5 provides a summary of obtained results

and conclusions.

17


3.2 SAC-OCDMA burst-mode receiver

The main building blocks of the SAC-OCMDA burst-mode receiver are

illustrated in Fig 3.1. The receiver includes a quantizer, a multi-rate

SONET CDR, a 1:8 deserializer, and the following blocks which are

implemented on a field-programmable gate array (FPGA): comma

detector (the role of which will be made clear later) and framer, a CPA, a

R-S(255, 239) FEC decoder, PLLs and a custom BERT. The multi-rate

CDR is from Analog Devices (part # ADN2819); the deserializer is from

Maxim-IC (part# MAX3885), and the FPGA board is from Xilinx.

Fig. 3.1 SAC-OCDMA receiver block diagram.

A quantizer is used before the CDR to apply a threshold on the

incoming signal in order to filter out intensity noise and other channel

impairments. The threshold can be manually adjusted to sample in the

18


middle of the eye opening in order to obtain optimum BER measurements.

The multi-rate CDR recovers the clock and data of the incoming signal.

The system is tested at OC-12 rates, and the CDR supports the following

data rates of interest: 622.08 Mbps for operation without FEC, 666.43

Mbps to account for the 15/14 FEC overhead, and 1.25 Gbps for burst-

mode operation with the CDR sampling the incoming data at twice the bit-

rate.

At the used data rate of 622.08 Mbps, when the CPA is employed the

data and clock outputs of the CDR have a frequency of 1.25 Gbps. This is

higher than the maximum rate that can be supported by the LVDS buffers

of the FPGA which is 840 Mbps. Therefore, a 1:8 deserializer is used

before the FPGA to reduce the frequency of the incoming signal, by

parallelizing the data and clock. The deserializer outputs use SMB

connectors, while high-speed QSE connectors are used to bring data onto

the FPGA board, therefore a SMB-to-QSE interface PCB designed by

Julien Faucher is used between the deserializer and the FPGA board.

When packets arrive with a phase difference, once the CDR has

locked according to the phase of one packet it takes time for it to re-

acquire lock so that the clock is properly aligned with the data from the

second packet. Hence, until lock is achieved according to packet 2, some

of packet 2 bits may not be correctly detected. To overcome this problem,

without resorting to the use of preamble bits, the CPA is used to correctly

receive all the bits, even the ones sent before CDR achieves lock. It uses

19


a twice over-sampling SONET CDR and a phase picking algorithm

implemented on an FPGA board. Instantaneous phase acquisition is

achieved using zero preamble bits for any phase step between the

packets. The CPA is turned ON for the PLR measurements with phase

acquisition, otherwise it is bypassed.

Fig. 3.2 Test signal emulating bursty uplink traffic.

To clearly understand the operation of the CPA, we first present the

test signal used. The signal, which is shown in Fig. 3.2, is a typical bursty

uplink signal that complies with PON standards. It is composed of two

packets with a silence period in between. Packet 1 is an alternating

sequence of ones and zeros (‘1010…’); it is a dummy packet used to lock

the CDR to the desired frequency before the arrival of the packet 2, on

which the measurements are actually made. Packet 2 can be seen to be

made of n preamble bits, 20 delimiter bits, 215 – 1 payload bits and 48

comma bits. The preamble and the delimiter bits correspond to the

physical layer upstream burst-mode overhead at 622 Mbps, as specified

by the ITU-T G.984.2 standard [86]. The preamble is normally used for

20


amplitude and phase recovery; in this demonstration the length of the

preamble is set to zero and the phase picking algorithm is used to recover

the phase. The delimiter and comma are unique patterns that mark the

beginning and end of a packet respectively, and hence are used to for

synchronization and determining packet loss. The payload is a 215 – 1

pseudo random binary sequence (PRBS). The lock acquisition time

corresponds to the number of preamble bits (n) needed in order to get a

zero PLR for over three minutes at 622 Mbps (>106 packets received, i.e.

PLR<10-6), at a BER<10-10, and for any phase step (-2π ≤ Δφ ≤ +2π)

between consecutive packets [62]. It is found that zero bits of preamble

are needed for phase recovery when the CPA is employed.

The silence period between the two packets includes m consecutive

identical digits (a sequence of m zeros) and a phase difference ∆φ (-2π <

∆φ < 2π) which can each be separately controlled. The random phase

step between the two packets represents the asynchronous nature of

OCDMA traffic. This signal is generated by combining data from two ports

of an HP80000 pulse pattern generator using a radio frequency (RF)

power combiner. The phase steps between the consecutive packets can

be set anywhere between ±2 ns on a 2 ps resolution, corresponding to a

±1.25 unit interval (UI) at 622 Mbps, where 1 UI corresponds to 1 bit

period.

On the receiver, automatic detection of the payload is achieved

through the comma detector and framer, as well as the byte synchronizer

21


which is responsible for detecting the delimiter. The idea behind the phase

picking algorithm is to replicate the byte synchronizer twice in an attempt

to detect the delimiter on either the odd and/or even samples of the data

respectively. The functionality of the CPA is explained in more detail when

the eye diagrams obtained are presented in the results section.

The realigned data at the output of the CPA is then sent to the R-

S(255, 239) decoder, which is turned ON when BER measurements with

FEC are made, otherwise it is by-passed. The RS decoder is an IP core

from Xilinx LogiCORE portfolio. It is followed by a custom bit-error-rate-

tester (BERT) implemented on the FPGA, which performs BER

measurements on the payload bits of the received packets; it compares

the received data with a pre-stored 215 – 1 PRBS. This custom BERT

allows BER measurements to be made on bursty data, which is not

possible using a commercial BERT since they require continuous

alignment between the incoming pattern and the reference pattern not to

lose synchronization. The custom BERT is also capable of making PLR

measurements to assess the performance of the CPA. Moreover, making

the measurements on the FPGA board eliminates the need to up-convert

the frequency back to 622.08 Mbps or 666.43 Mbps using a 8:1 serializer

and avoids the use of a commercial BERT. On the receiver, BER

measurements are made on the payload bits of the received packets only.

PLR measurements are determined by packet loss count; at the receiver,

22


if a delimiter is not received, but a comma is correctly received, the packet

is declared lost.

3.3 SAC-OCDMA uplink and experimental setup

The SAC-OCDMA system used is based on balanced incomplete block

design (BIBD) codes of length 7 and weight 3 that have with a fixed IPCC

of 1 [83], hence allow elimination of MAI using balanced detection. These

codes are illustrated in Table 3.1, along with the corresponding decoder

(DEC) and complementary decoder (C-DEC) codes with respect to

desired user #1. The optical band for the 7 wavelengths is 9.6 nm [11].

Table 3.1

BIBD Codes Used

User Code

# 1 1101000

# 2 0110100

# 3 0011010

# 4 0001101

# 5 1000110

# 6 0100011

# 7 1010001

DEC 1101000

C-DEC 0010111

23


In this demonstration the PON architecture used is a local sources

architecture [13], in which a directly modulated light source is placed at

each ONU; this is illustrated in Fig. 3.3 which shows the PON uplink.

Balanced detection is implemented to cancel MAI. Although balanced

detection cancels MAI, it does not eliminate the intensity noise added by

interferers.

Fig. 3.3 Local sources PON architecture.

The experimental set-up of the SAC-OCDMA uplink is shown in Fig.

3.4. A single incoherent broadband source is filtered around 1542.5nm

using two cascaded FBG band-pass filters to remove out of band intensity

noise, providing an optical band of 9.6nm. The light is modulated with a

non-return-to-zero (NRZ) 215 - 1 pseudo random binary sequence (PRBS)

using a polarization independent electro-absorption modulator (EAM). The

data rate of the modulating signal is 622 Mbps when FEC is not employed,

and 666.43 Mbps with FEC to account for the 15/14 overhead introduced

by the R-S (255, 239) codes.

24


Fig. 3.4 SAC-OCDMA PON uplink experimental setup. BBS: broad-band source;

EAM: electro-absorption modulator; ENC: encoder; DCF: dispersion-

compensation fiber; EDFA: erbium doped fiber amplifier; VOA: variable optical

attenuator; DEC: decoder; LPF: low-pass filter.

The modulated light is then split up using a 1x8 power coupler to

obtain 7 users, the desired user and 6 interferers. The modulated signals

are spectrally encoded using FBGs; each user’s FBG has a spectral

response that yields one of one of the codes illustrated in Table 3.1. It can

be seen that any two users always overlap in a single wavelength. The

FBGs, which are optimized for operation around 622 Mbps [11], are

working in transmission. After encoding, the signals from different users

are passed through different lengths of optical delay lines to de-correlate

the data; the lengths of optical delay lines are 0, 3, 6, 9, 12, 15 and 18

meters for the 7 users. The desired user has 0 delay and the other delays

25


are randomly distributed among the remaining 6 users. The spectrally-

encoded de-correlated outputs represent 7 ONUs; these are then

combined on to a single fiber using an 8x1 coupler, which is followed by

20km of uplink single-mode fiber (SMF) representing the ODN and

dispersion compensation fiber (DCF).

At the OLT, the signal is amplified using an erbium-doped fiber

amplifier (EDFA) to compensate for the losses through the network and

the splitting losses. A variable optical attenuator (VOA) is the used to

control the received power before the balanced receiver. Balanced

detection is used to decode the incoming data with respect to user 1. Two

FBGs working in transmission are used as the decoder and its

complementary decoder. For the BIBD codes used, every interferer has

one wavelength in common with DEC, but two wavelengths in common

with C-DEC (Table 3.1). Therefore, for balanced detection to be achieved,

a 3dB attenuation is introduced using a VOA in the second arm after the

complementary decoder. To ensure perfect MAI cancellation, the optical

lengths of the two branches of the balanced receiver, must be perfectly

equal. The optical length of the second arm includes the optical delay

introduced by the VOA (3dB attenuation) hence an equivalent optical

delay must be introduced to the first arm to ensure equal optical path

lengths; this is illustrated in Fig. 3.5. The output of each arm is then

passed to one of the two inputs of a balanced photo-detector from New

Focus (model 1617) that has a bandwidth of 800 MHz. However, since

26


the 3 dB attenuation and the optical delay introduced cannot be perfectly

set, receiver balancing, hence the elimination of MAI, cannot be perfectly

achieved.

Fig. 3.5 Balanced receiver. ODL: Optical Delay Line.

To understand how this balanced receiver eliminates MAI, consider the

illustration of Fig. 3.6. We will consider the case of the desired user

transmitting a ‘1’ and verify that it passes through with its full power, and

then consider the case of an interferer transmitting a ‘1’ and verify that

zero power gets transmitted in this case. A ‘1’ is represented by three

units of power (3u), one unit per wavelength. It is easy to see from Fig. 3.6

that the ‘1’ of a desired user passes through the decoder completely, and

gets completely blocked by the complementary decoder. Therefore, the

output of the balanced receiver will have a power of 3u, same as the input.

Now consider the case of an interferer transmitting a ‘1’; in this case the

interferer is user #2 (Table 3.1). The decoder passes through only one unit

of power 1u that corresponds to wavelength λ2, while the complementary

27


decoder passes through two units of power 2u, those corresponding to λ3

and λ5. After the 3-dB attenuation, the lower arm will also have one unit of

power 1u; the powers in the two arms cancel out and the output of the

balanced photo-detector is zero, hence MAI is removed. In practice

however, due to imperfect balancing, MAI is not completely eliminated and

there is some noise in the system due to MAI.

This applies for any of the interferers with respect to any of the desired

users, since the IPCC is always one, which means that any user will

always overlap with the decoder at one wavelength, and overlap with the

complementary decoder at two wavelengths. Notice that although

balanced detection eliminates most of the MAI, it does not remove any of

the intensity noise coming from the interferers; this intensity noise passes

through the system, and it is the limiting factor to SAC-OCDMA systems’

performance. The intensity noise in an incoherent system is inversely

proportional to the effective optical bandwidth, and proportional to the

electrical bandwidth [11].

28


Fig. 3.6 Illustration of balanced detection. DEC: decoder of desired user

(1101000); DEC’: complementary decoder of desired user (0010111).

29


The electrical signal at the output of the photo-detector is then

amplified and low-pass filtered using a 4th order Bessel-Thomson filter

whose 3-dB cut-off frequency is 467 MHz to remove the out-of-band high-

frequency electrical noise. Such a filter reduces intensity noise from the

incoherent broadband source [11], while keeping inter-symbol interference

to a minimum [85]. The electrical output of the filter is then passed to

either the error detector to make BER measurements using the global

clock or to the receiver for CDR and further processing.

3.4 Results and discussion

In this section we present the performance of the system in terms of

BER, PLR and CID immunity. The obtained eye diagrams are also

presented, and they are used to explain the functionality of the CPA with

twice over-sampling and phase picking algorithm. The results are

presented for a back-to-back configuration and then the BER

measurements are compared with the local sources PON architecture.

The BER measurements are made on continuous upstream traffic;

they are made either using a commercial BERT for the global clock

measurements, or using the custom BERT on the receiver for all other

measurements after clock and data recovery. The PLR measurements

quantify packet loss performance of the system, and they use bursty

uplink traffic (Fig 3.2) with packets of 215 - 1 PRBS length. With the CPA

error-free operation is achieved using a preamble length of zero. Today’s

30


PON standards provide a preamble length of maximum 28 bits [86], to

allow the receiver enough time for phase and amplitude recovery.

3.4.1 BER Performance

This subsection presents the BER performance of the SAC-OCDMA

system using the receiver. Initially, the BER measurements versus power

are presented in Fig. 3.7 for 1, 3, 5 and 7 users for the back-to-back

configuration. The horizontal axis represents the useful power, in other

words the received power from the desired user. The corresponding eye

diagrams at -18 dBm are also shown as insets on the plot. These

measurements are made with the global clock using a commercial error

detector. The phase difference between the packets is kept at zero.

1 user3 users5 users7 users

1 user

3 users

5 users

7 users

BER = 10-9

ERROR FREE

-30 -28 -26 -24 -22 -20 -18 -16Useful power [dBm]

-12

-10

-8

-6

-4

-2

0

10

10

10

10

10

10

10

Fig. 3.7 BER vs. useful power for different number of users using a global clock.

31


It can be seen that starting from a single user, the results show a

classic waterfall curve. As the number of simultaneous users in the system

is increased, BER floors begin to appear, starting from the introduction of

the 5th user. More specifically, it can be seen that the system supports

error-free operation (BER < 10-9) for up to 5 simultaneous users. However,

the seventh user is not error-free due to the error-floors residing just below

a BER of 10-6. The BER floors for 5 to 7 users are created due to the

intensity noise added by the interferers. Although balanced detection

eliminates most of the MAI, it does not remove the intensity noise added

by MAI. This can be seen from the eye diagrams captured at -18 dBm

which are added as insets on the figure. The eye diagram for a single user

is very open; however, as the number of users increases the eye becomes

more closed, despite the cancellation of MAI.

Next we quantify the system performance using the clock recovered by

the receiver’s CDR module. Fig. 3.8 shows BER versus useful power for

the back-to-back architecture for 1, 3, 5 and 7 simultaneous users. For

ease of comparison, the global clock measurements are repeated on this

figure.

32


Fig. 3.8 BER vs. useful power for different number of users: comparing global

clock and recovered clock (Dashed lines: using the global clock, solid lines: using

CDR).

It can be seen that there is a slight penalty introduced due to the non-

ideal sampling of the recovered clock, compared to that of the global

clock; however, this penalty is negligible as we can see the proximity of

the global clock and CDR curves for each user. It is important to note that

for each user the global clock and recovered clock curves intersect at

around -26 dBm. At the lower power levels, there is a slight improvement

in performance with the CDR compared to using the global clock, despite

the non-ideal sampling of the recovered clock. This improvement is due to

the accurate adjustment of the quantizer’s threshold at low power levels.

We were able to manually control the decision threshold using a DC

33


power supply; this positioned the threshold more accurately in the middle

of the eye compared to the use of an automated decision threshold in the

commercial BERT for the global clock measurements. The manual

threshold optimization explains the slight improvement in BER

performance when using the recovered clock at power levels smaller than

-26 dBm.

To remove the BER floor for 5 to 7 users, FEC using RS(255, 239)

codes is employed. To determine the impact of adding FEC, we plot in

Fig. 3.9 the BER versus useful power when using the CDR and FEC,

compared to using only the CDR. After FEC error-free operation is

achieved for all 7 users. A coding gain of more than 2.5 dB, 3 dB, and 5.5

dB (measured at a BER = 10-9) for 1 user, 3 users, and 5 users,

respectively, is achieved. This could not be measured for 7 users due to

its BER floor without FEC residing above 10-9. Furthermore, all BER floors

are eliminated and the plots obtained are classic waterfall curves, for all

users. The penalty in moving from a single user to a fully loaded system is

around 3.2 dB with FEC. Note that despite the closure of the eye-

diagrams for a fully-loaded system (can be seen on Fig 3.7) FEC corrects

transmission errors allowing for error-free operation.

34


-30 -28 -26 -24 -22 -20 -18 -1610

-12

10-10

10-8

10-6

10-4

10-2

100

Useful power [dBm]

1 user3 users5 users7 users

CDR + FEC

CDR

BER = 10-9

ERROR FREE

Fig. 3.9 BER vs. useful power with and without FEC (Dashed lines: using CDR,

solid lines: using CDR and FEC).

Finally, back-to-back and PON architectures are compared in Fig.

3.10. The results are shown for CDR, and CDR and FEC being employed.

It can be seen that the 20km of uplink fiber have introduced a penalty of

less than 1 dB at a BER of 10-9 for all users.

35


Fig. 3.10 BER vs. useful power for a single user and fully-loaded systems:

comparing PON and back-to-back architectures (Dotted lines for back-to-back;

dashed lines for LS architecture).

36


Therefore, while operating at a relatively low power of around -24 dBm

we demonstrate an error-free 7x622 Mbps uplink of an incoherent SAC-

OCDMA PON using a standalone receiver with CDR and FEC.

3.4.2 PLR Performance

This subsection presents the PLR performance of the SAC-OCDMA

system. This measure pertains to the functionality of the CPA, and the

results are presented for a back-to-back configuration. In Fig. 3.11 the

PLR is plotted versus phase step using only the SONET CDR without the

phase picking algorithm; this measure is repeated for 1, 4 and 7

simultaneous users. The power level is kept at -18 dBm for these

measurements. The horizontal axis ranges from 0 to 1600 ps, which

corresponds to 0 to 2π phase difference at the desired bit rate (~622

Mbps). We did not consider the interval from -2π to 0, since theoretically it

gives the same performance as the interval from 0 to 2π [62]. It can be

seen that for all users, the curve has a bell-shape indicating that the PLR

performance is worst at a phase difference of 800ps which is equivalent to

∆φ = π. This makes sense since at ∆φ = π with the CDR sampling at the

data rate, the data is sampled very close to the transitions, making this the

worst-case phase difference. If jitter had been prevalent in the system, the

worst case would be displaced from 800ps. This is not the case in our

37


measurements, from which we conclude that jitter is not significant in the

system.

Fig. 3.11 PLR vs. phase difference without CPA for different number of users

It can be seen that as the number of users supported by the system is

increased from a single user to a fully loaded system of 7 users, the PLR

performance generally deteriorates. For 1 and 4 users the performance is

very similar; however for a fully-loaded system of 7 users the performance

is much worse. Whereas users 1 and 4 have error-free operation

(PLR<10-6, corresponding to a BER<10-10) for small phase differences

(less than 400 ps or greater than 1100 ps), with 7 users the system never

achieves error-free operation. The degradation in the PLR in going from 4

to 7 users can be explained by the corresponding degradation in the BER.

38


At a power level of -18dBm, users 1-4 operate error-free; however, for

users 5-7 there are BER floors (CDR plot on Fig. 3.8). As the BER

performance degrades, there is a higher chance of having erroneous bits

in the packet delimiter. With the delimiter not being correctly detected, a

packet is declared lost, hence contributing to the packet loss count and

the PLR. It can also be seen that for 7 users, the PLR curve experiences a

slightly odd behavior, since the PLR slightly decreases for increasing the

phase error between 0 and 200 ps, before increasing as expected. This is

probably due to measurement inaccuracy especially since for 7 users at

this power level, the BER performance is poor, and the eye is much

degraded (Fig. 3.7).

To see the impact of using the CPA, the upper bound of the packet

loss ratio i.e. the maximum PLR value that occurs at π phase shift, is

plotted versus the number of users with and without CPA in Fig. 3.12. With

the CDR only, the worst-case PLR is near 1, indicating that most of the

packets are lost since the CDR is sampling at the edge of the eye-diagram

at this phase difference. When the phase picking algorithm is employed

with twice over-sampling, the PLR performance improves greatly. Error-

free operation (PLR<10-6, corresponding to a BER<10-10) is achieved for

up to 4 users. Beyond 4 users, some packets are still lost; again this

degradation in the PLR is due to the degradation in the BER for 5 and 7

users at -18 dBm. Therefore, for a fully loaded system of 7 users, the CPA

improves the PLR performance by more than two orders of magnitude.

39


Fig. 3.12 PLR vs. number of users

3.4.3 CID Immunity

The immunity of the CDR to silence periods is examined by increasing

the number of CIDs between the two packets (refer to Fig. 3.2) and

monitoring the packet loss. This is depicted in Fig. 3.13 where the PLR is

plotted versus the number of CIDs. The measurements are made for a

single user at a useful power level of -18 dBm. From the CDR plot on Fig

3.8 it can be seen that at this power level, for a single-user BER operation

is error-free, hence the PLR performance is not affected by the BER. No

preamble bits are used in making these measurements.

40


Fig. 3.13 PLR vs. length of CID

The general trend of the curve shows that as the number of CIDs

between the packets is increased, more packets are lost. This is because

once the CDR has locked to packet 1, if the silence period between the

two packets becomes too long, the CDR loses lock and by the time packet

2 arrives it need to re-acquire lock. As the CDR is trying to acquire lock to

packet 2, the delimiter of packet 2 may be incorrectly sampled in the

mean-time, resulting in the packet being lost. It can be seen that the

maximum number of silence bits that the receiver can withhold is around

600 CIDs, which satisfies existing PON standards.

41


3.4.4 Eye Diagrams

The obtained eye diagrams and recovered clocks are illustrated in Fig

3.14, for a conventional CDR and for the CDR in twice over-sampling

mode. The CPA is based on a twice over-sampling CDR with a phase

picking algorithm. Recall that the idea behind the phase picking algorithm

is to try to detect the delimiter in either the odd/even samples of the data.

That is, regardless of any phase step ∆φ between consecutive packets,

there will be at least one clock edge (either todd or teven) that will yield an

accurate sample. In Fig 3.14 three specific phase differences between

packets are considered: (a) Δφ = 0 rad (0 ps), (b) Δφ = π/2 rads (400 ps),

and (c) Δφ = π rads (800 ps). Whereas Δφ = π rads (800 ps) represents

the worst case phase step for the CDR operated at the bit rate, Δφ = π/2

rads (400 ps) phase step is the worst case scenario for the over sampling

CDR at 2× the bit rate. The worst-case phase step is the phase difference

at which the second packet will be sampled very close to the transitions.

To understand how the CPA works consider the worst case scenario

for twice over-sampling, of Δφ = π/2 rads. As packet one arrives at the

receiver, the CDR locks at the phase of this incoming packet 1. As soon

as packet 2 arrives, the CDR’s lock remains as was acquired for packet 1.

Hence, the CDR would initially sample packet 2 with the samples marked

as ‘1’ corresponding to clock samples as locked to packet 1. With the

conventional CDR, the ‘1’ sample lies close to the transition of packet 2.

42


Comparing this with twice over-sampling, there are two ‘1’ samples in this

case, the odd and even samples (todd and teven). It can be seen that the

odd sample (todd) lies very close to the packet 2 transition. In this situation,

the byte synchronizer of path O will likely not detect the delimiter at the

beginning of the packet. On the other hand, the even sample (teven) lies

close to the middle of the eye-opening, and path E will have more

accurate samples of the data, and is likely to detect the delimiter. The

phase picker then uses feedback from the byte synchronizers to select the

correct path from the two possibilities. Once the selection is made, it

cannot be overwritten until the comma is detected, indicating the end of

the packet. This process repeats itself at the beginning of every packet.

The result is that the CPA achieves instantaneous phase acquisition (0 bit)

for any phase step (±2π rads). That is, no preamble bits at the beginning

of the packet are necessary.

43


Fig. 3.14 Eye diagrams showing the response of the CDR to bursty traffic rence,

Also from the eye diagrams it can be seen that after the CDR the eye

looks much cleaner compared to the input to the CDR. The reason for this

is that the quantizer filters out a lot of the intensity noise coming from the

broad-band source, making the eye more open.

(packets with different phases): (a) no phase difference, (b) π/2 phase diffe(c) π phase difference.

44


3.5 Conclusion

In this chapter, we demonstrated an incoherent SAC-OCDMA PON

uplink supporting 7 simultaneous users using a standalone burst-mode

receiver. The system is tested under continuous and bursty uplink traffic,

for the BER and PLR measurements respectively. BER, PLR and CID

immunity results have been presented for a back-to-back configuration,

and BER measurements for a local sources PON architecture [13] are

presented an compared to the back-to-back results. We see that in going

to from a back-to-back architecture to local sources PON configuration the

penalty introduced is less than 1 dB.

The receiver used performs CDR, FEC and CPA. We show that the

non-ideal sampling of the recovered clock has a negligible penalty

compared to the global clock. Using the RS(255, 239) codes, we achieve

error-free transmission for a fully-loaded system, compared to 4 users

before FEC employment. More specifically, we obtain a coding gain of

more than 2.5 dB, and elimination of the BER floors for 5 - 7 users.

Therefore, while operating at a relatively low power of around -24 dBm we

demonstrate an error-free 7x622 Mbps uplink of an incoherent SAC-

OCDMA PON using a standalone receiver with CDR and FEC.

We studied the performance of the system using bursty uplink traffic

through the PLR measurements. Using the twice over-sampling and a

phase picking algorithm, we measure a zero PLR using instantaneous

45


phase acquisition (no preamble bits) for up to 4 users. Beyond that, up to

a fully-loaded system, the CPA improves PLR performance by more than

two orders of magnitude for the worst-case phase difference.

Finally the CID immunity of the receiver in this SAC-OCDMA PON

environment is measured for a single user. It is found that the receiver can

take up to 600 CIDs while maintaining PLR error-free operation which

complies with existing PON standards.

This is the first time, to our knowledge, that an OCDMA system has

been tested in a bursty environment. Moreover, most previous work in

OCDMA does not demonstrate performance with electronic receivers

using a recovered clock. A similar system has been reported in [87],

where the authors demonstrate a 2-D OCDMA system with recovered

clock using an electronic receiver. However since 2-D OCDMA requires

electronics operating at the chip rate, while SAC-OCMDA has the

advantage of operating at the data rate, our results are at a higher data

rate of 622 Mbps (with 7 error-free users), as compared to 155 Mbps (with

5 error-free users) for the 2-D OCMDA results. The results obtained show

that SAC-OCDMA is a promising candidate for PONs, when FEC is

employed to improve soft capacity, and burst-mode reception is used to

reduce packet loss.

46

4 Performance Analysis of a Burst-Mode Receiver in GPON Uplink

Chapter 4 Performance Analysis of a Burst-Mode Receiver in GPON Uplink

4.1 Introduction

In the previous chapter we demonstrated the use of SAC-OCMDA in a

PON uplink. In this chapter, the use of OTDMA with burst-mode reception

in a PON uplink is investigated. OTDMA is the most widely deployed is

scheme in PONs; it is used in today’s GPON standards [86]. In a PON, to

save optical fiber and reduce repair costs, a single fiber can be used for

downlink and uplink transmission; therefore for OTDMA PON two

wavelengths can be used to share the fiber; one for downlink and the

other for uplink. In GPON, 1310 nm is the uplink wavelength, and 1550 nm

is the downlink wavelength [86].

The performance of a burst-mode receiver in a GPON uplink is

studied. The receiver used in this chapter is very similar to that of chapter

3; it performs CDR, CPA and FEC. Section 4.2 presents a brief overview

of the receiver. In section 4.3, the GPON uplink test-bed is presented.

Section 4.4 presents the results in terms of BER performance, PLR

performance, CID immunity and eye diagrams of the system; however,

focus is placed on analyzing packet loss in order to assess the

47


functionality of the CPA and assess the penalty of burst-mode reception

using twice over-sampling. Finally section 4.5 concludes the chapter.

4.2 GPON burst-mode receiver

The burst-mode receiver to study the performance of a bursty GPON

uplink is illustrated in Fig 4.1. It is similar to that described in the previous

chapter (section 3.2), but with a slight difference. Whereas in the

demonstration of the SAC-OCDMA uplink, a quantizer was used before

the CDR to manually set the threshold, it is not used in this set-up. In the

case of OCDMA, multiple users were supported by the network; as the

number of users was increased beyond 4 users, the intensity noise was

very significant creating BER floors. The quantizer was used to set a

threshold on the signal, in order to get the most optimum BER

measurement. This threshold filters out some of the intensity noise, hence

enabling a more optimum BER measurement to be taken. In this case of a

GPON uplink, only one user is transmitting at a time. Therefore the

intensity noise level is not as high as in the OCDMA uplink with 7

simultaneous users, and the use of the quantizer was not necessary.

48


Fig. 4.1 Receiver block diagram.

The main building blocks of the receiver are a multi-rate CDR, 1 1:8

deserializer, a CPA and an RS(255, 239) FEC decoder. The CDR

recovers the clock and data from the incoming signal. The 1:8 deserializer

reduces the data rate of the data and clock to a rate that can be

processed by the digital logic that follows. The CPA and RS(255, 239)

decoder are implemented on an FPGA board, along with a custom BERT.

For a description of the separate components of the receiver, and detailed

explanation of their functionalities refer to section 3.2 of this thesis. The

data rate is 622.08 Mbps or 666.43 Mbps depending on whether FEC is

turned OFF or ON.

49


4.3 GPON test-bed

A block diagram of the uplink GPON experimental setup is illustrated in

Fig 4.2. Since the uplink is considered, a 1310 nm distributed feedback

(DFB) laser is used as the light source. The laser is modulated using a

Mach Zehnder electro-optic modulator (EOM) with data coming from a

HP80000 pulse pattern generator (PPG). The bursty test signal used is

composed of two packets, separated by a silence period made of m

consecutive identical digits (a sequence of m zeros) and a phase

difference ∆φ (-2π < ∆φ < 2π) which can each be separately controlled.

This signal is illustrated in Fig. 3.2, and explained in detail in section 3.2.

The phase difference between the packets is varied for the measurements

with the CPA otherwise it is kept at zero. The payload on which BER and

PLR measurements are actually made is a NRZ 215 – 1 PRBS. The EOM

takes in a voltage swing of around 5 V, which is too large to be supplied

directly by the PPG. Therefore, an EOM driver, which is essentially an RF

amplifier, is used to amplify the output of the PPG, to provide a high

enough voltage swing to drive the EOM.

50


Fig. 4.2 GPON uplink experimental setup. PC: polarization controller; LPF: low-

pass filter

The EOM is polarization dependent therefore to eliminate this

dependence and be able to get reliable optical data for BER and PLR

measurements at the output of the EOM, a polarization controller is used

before the EOM. The arms of the polarization controller are moved until

optimum data at the output of the EOM is obtained; the optimum output

signal is one that has the best extinction (on-off) ratio.

The EOM is biased using a DC power supply. The bias point is very

important in obtaining optical data reliable for making accurate and

repeatable measurements. It was adjusted to optimize the extinction ratio

to get the cleanest eye possible. The issue with EOMs is that their bias

point drifts with time therefore during experimentation the bias point may

shift rendering the measurements inaccurate, and unrepeatable. To avoid

this problem, an EAM can be used instead. However, due to unavailability

51


of equipment, we used an EOM while keeping this issue in mind. To

ensure that the bias point has not deviated during the capture of results,

random points that have already been obtained on the BER or PLR plots

were measured again to check repeatability. If the points agree, it

indicates that the bias point has not shifted. This was repeated regularly

during the capture of measurements, to ensure the results correspond to a

fixed bias point, hence are meaningful. If a point did not yield a repeatable

result, and the previously obtained BER or PLR curve has shifted, this

indicates that the bias point has drifted. When this happens, the bias point

is re-adjusted for the optimum extinction ratio, and the repeatability of

previously obtained results is verified before more points are taken.

The modulated optical signal at the output of the EOM is then passed

through 20 km of uplink SMF fiber, representing the ODN. At the OLT

side, a VOA is used to control the received power. The optical signal at

the output of the VOA is converted to an electronic signal using a photo-

detector from New Focus (model # 1617). To remove out-of-band high

frequency electrical noise, a 4th order Bessel-Thomson filter whose 3-dB

cut-off frequency is 467 MHz is used prior to the receiver. Such a filter

reduces intensity noise from the incoherent broadband source while

keeping inter-symbol interference to a minimum [85]. The electrical output

of the filter is then passed to the receiver for further processing and

measurements.

52


4.4 Results and discussion

The performance of a burst-mode receiver in GPON is measured and

presented in this section. The BER performance is presented, showing the

impact of employing FEC. Then a thorough study of PLR using the burst-

mode receiver is given. Finally the CID immunity of the receiver under

these settings is measured.

4.4.1 BER Performance

The BER measurements for the PON architecture with and without

FEC are plotted in Fig. 4.3; the horizontal axis represents the received

power at the photo-detector. Comparing initially the curves of experimental

results with and without FEC, one can see that the coding gain is around 3

dB measured at a BER of 10-10. This coding gain obtained through FEC

can have several uses in GPON; it can be used to reduce the transmitter

power by the amount of the gain, or increase the minimum receiver

sensitivity by the same amount. Alternatively, this effective gain can be

used to achieve a longer physical reach or a higher split ratio.

53


-20 -19 -18 -17 -16 -15 -14 -13 -12-10

-9

-8

-7

-6

-5

-4

-3

-2

-1

0

Useful power [dBm]

Log

(BE

R)

Without FECWith FECTheoretical

Fig. 4.3 BER vs. useful power of the GPON uplink: experimental results with and without FEC and FEC simulation results.

The theoretical plot for FEC based on the BER measurements made

without FEC are simulated and illustrated on the same plot. Let pe be the

measured BER without FEC. The theoretical results with FEC are

calculated with [17]

( ) jS

jS

tj

m

mFECS

mm

ppj

jp −−−

+=

−⎟⎟⎠

⎞⎜⎜⎝

⎛ −−

≈ ∑ 1212

11

1212

1 (4.1)

where ps and psFEC are the symbol error probabilities before and after FEC

decoding respectively, m is the number of bits per symbol (8 in this case

54


for the RS(255, 239) codes), and t is the error-correction capability of the

code given by

⎥⎦⎥

⎢⎣⎢ −

=2knt (4.2)

where represent the largest integer smaller than or equal to x. The

symbol error probabilities ps are calculated from the bit error probabilities

pe that are experimentally measured using the following formula, which

assumes purely random bit errors.

⎣ ⎦xt =

( )meS pp −−= 11 (4.3)

The lower bound of the bit error rate with FEC is peFEC calculated by

mp

pFECSFEC

e = (4.4)

Comparing the experimental results with the theoretical lower bound, it

can be seen that the experimental curve agrees closely with the

simulations for a BER less than 10-4, above which the measured results

deviate from the theoretical predictions. Since the theoretical predictions

assume completely random bits, this deviation indicates that the system

has deterministic errors that become more significant at higher power

levels. This may be explained by the memory added in the channel

through deterministic jitter and the CDR making errors statistically

55


dependent in this system. To remove this dependency an interleaver may

be used.

4.4.2 PLR Performance

In this section the PLR performance of the burst-mode receiver is

thoroughly studied; particularly the functionality of the receiver’s CPA is

highlighted. A comparison of PLR performance for the back-to-back

configuration and PON configurations is presented; the rest of the results

are made for a back-to-back configuration, without the 20km of uplink

SMF. Fig 4.4(a) shows the PLR versus phase difference for the CDR (no

CPA) for different lengths of preamble (0, 16 and 28). The expected bell-

curve can be seen with the PLR improving as more preamble bits are

used; recall that the preamble is an alternating sequence of ones and

zeros (‘10101…’) that aids in phase acquisition. Error-free (PLR < 10-6

and

BER < 10-10

) operation is observed when 32 preamble bits are used.

56


0 200 400 600 800 1000 1200 1400 1600-6

-5

-4

-3

-2

-1

0

Phase difference, Δφ [ps]

Log

(PLR

)

01628

(a)

0 200 400 600 800 1000 1200 1400 1600-6

-5

-4

-3

-2

-1

0

Phase difference, Δφ [ps]

Log

(PLR

)

20-km GPONBack-to-Back

(b)

Fig. 4.4 PLR vs. phase difference (a) Back-to-back configuration with CDR for different preamble lengths. (b) Comparison between back-to-back and PON configurations with and without CPA with 0 bit preamble.

Fig. 4.4(b) shows the PLR curves with and without CPA for a 0

preamble length, for both, back-to-back and PON configurations. It can be

seen that the introduction of the 20 km of fiber degrades the PLR

57


performance. However, more importantly, we observe that once the CPA

is used, error-free operation is achieved for both configurations at any

phase step with no preamble bits, allowing for instantaneous (0 preamble

bit) phase acquisition. This is well below the maximum preamble length of

28 bits specified in the GPON standards for amplitude and phase recovery

[86].

To determine the burst-mode penalty of the receiver, the PLR versus

received power is plotted in Fig. 4.5 for two cases to be compared: 1) the

CDR sampling continuous data (no phase difference) at the data rate of

622 Mbps and 2) the burst-mode receiver (2x over-sampling CDR and

CPA) sampling bursty-data with the worst-case phase difference of π/2

rads. Both measurements are made for a 0 bit preamble. With the 2×

over-sampling and the phase picking algorithm employed on incoming

bursty data, we observe a penalty of less than 1 dB compared to sampling

continuous data. The respective eye diagrams at the input of the CPA are

shown as insets.

58


-18 -17 -16 -15 -14 -13 -12-6

-5

-4

-3

-2

-1

0

Useful power [dBm]

Log

(PLR

)

ContinuousBursty

Fig. 4.5 PLR vs. useful power for continuous and burst-mode reception.

When there is a phase difference between the packets, the CDR alone

is unable to recover the phase without the use of preamble bits,

regardless of the signal power, resulting in almost all the packets being

lost. This is illustrated in Fig. 4.6. To enhance the PLR performance either

preamble bits should be used while sampling with the CDR, or the

proposed burst-mode receiver can be employed. We compare the PLR

performances using a 0 bit preamble and employing the burst-mode

receiver, as opposed to using the GPON standard of 28 bit preamble while

using only the SONET CDR. The results obtained are shown in Fig. 4.6. It

can be seen that the two sets of results agree closely. Therefore, although

59


using twice over-sampling introduces a power penalty compared to

sampling continuous data, it allows phase acquisition using zero bits of

preamble. A power penalty of 1 dB is a reasonable compromise to

accommodate the bursty nature of GPON uplink while leaving all the

preamble bits to be used for amplitude recovery or increasing the

information rate.

-18 -17 -16 -15 -14 -13 -12-6

-5

-4

-3

-2

-1

0

Useful power [dBm]

Log

(PLR

)

CDR (0 bit preamble)CDR (28 bit preamble)BMRx (0 bit preamble)

Fig. 4.6 PLR vs. useful power for CDR (with 0 bit preamble), CDR (with 28 bit preamble) and burst-mode receiver (with 0 bit preamble)

60


4.4.3 CID Immunity

The immunity of the CDR to silence periods is examined by increasing

the number of CIDs between the two packets (refer to Fig. 3.2 for packet

structure) and monitoring the packet loss; the observed PLR for the PON

configuration is plotted versus the number of CIDs in Fig 4.7. It can be

seen that the receiver can support more than 800 CIDs with error-free

operation. This is more than ten times the maximum allowed number of

CIDs specified by GPON, which is 72 bits [86].

500 600 700 800 900 1000 1100-6

-5

-4

-3

-2

-1

0

Length of CID

Log

(PLR

)

Fig. 4.7 PLR vs. number of CIDs

61


4.4.4 Eye Diagrams

The obtained eye diagrams are illustrated in Fig 4.8, for a conventional

CDR and for the CDR in twice over-sampling mode. Three cases of phase

difference are shown (∆φ = 0, π/2 and π rad).

Fig. 4.8 Eye diagrams showing the response of the CDR to bursty traffic

(packets with different phases): (a) no phase difference, (b) π/2 phase difference,

(c) π phase difference.

62


The eye diagrams illustrate that the CDR removes some of the

intensity noise added on the optical channel, since the eye at the output of

the CDR is cleaner than at its input. Also the functionality of the CPA in

the case of bursty input can be understood as follows. Since it takes time

for the CDR to acquire lock once a new packet is received, initially with the

conventional CDR, the data may be sampled close to the transitions.

However a with twice over-sampling CDR, there are two samples; todd and

teven; at least one of the samples will sample the data in the eye opening.

The phase picking algorithm implemented on the FPGA board then

determines the correct sample based on correctly detecting the delimiter

on of the paths. A more detailed explanation of the CPA’s functionality can

be found in subsection 3.4.4.

4.5 Conclusion

In this chapter, a GPON uplink is demonstrated with a burst-mode

receiver. The receiver performs CDR, CPA and FEC. The system

performance is analyzed through BER measurements, CID immunity

measurements and focus is placed on PLR measurements to assess the

functionality of the CPA module in a bursty uplink environment, and

investigate the burst-mode penalty of the receiver.

It is found that employing FEC using R-S(255, 239) codes yields a

coding gain of around 3 dB. This coding gain can be used to reduce the

63


power budget, extend the physical reach of the PON, increase the split

ratio, or reduce the receiver sensitivity in a GPON environment. A

comparison between theoretical simulations and experimental results for

BER performance with FEC shows that the system has deterministic

errors that become more significant at higher power levels. The presence

of deterministic errors is likely due to the memory added in the channel

through deterministic jitter and memory incurred by the CDR.

The PLR results obtained show that twice over-sampling and the

phase picking algorithm enable instantaneous phase acquisition and zero

packet loss. It is found, however, that burst-mode reception through twice

over-sampling introduces a power penalty of around 1dB, which is the

price to pay to accommodate bursty traffic and achieve instantaneous

phase acquisition using zero bits of preamble. This leaves the preamble

bits to be used for purposes of amplitude recovery or increasing the

information rate.

Finally assessing the CID immunity of the receiver shows that more

than 800 CIDs can be supported with error-free operation. This is more

than ten times the maximum allowed number of CIDs specified by GPON,

which is 72 bits [86].

Compared to other approaches that have been proposed which

attempt to reduce the phase acquisition time, the receiver used in this

demonstration achieves instantaneous phase acquisition using zero

preamble bits. The price to pay for instantaneous phase acquisition is

64


faster electronics and the incurred power penalty due to twice over-

sampling. However, this receiver design makes use of off-the-shelf

components offering a cost-effective alternative to the design of a custom

ASIC and leveraging the commercial deployment of GPON.

65

5 Conclusion

Chapter 5 Conclusion

PONs are a promising technology in bringing fiber to the home. The

bursty nature of a PON uplink imposes burst-mode requirements of

receivers at the OLT. In this thesis the performance of a SAC-OCDMA

and a OTDMA PON are experimentally investigated using a burst-mode

receiver that performs CDR, CPA and FEC.

The SAC-OCDMA uplink demonstrated supports 7 asynchronous

users at 622 Mbps. The system makes use of FEC to remove BER floors

that are present because of the intensity noise added by MAI. Error-free

operation is obtained for a fully loaded system.

The use of FEC is found to remove BER floors in the case of multiple

simultaneous users in OCDMA, and introduce a coding gain. Hence it

enables error-free operation for simultaneous users in an OCDMA system

which is limited by intensity noise. Alternatively, the coding gain introduced

by FEC can be used to reduce the power budget, extend the physical

reach of the PON, increase the split ratio, or reduce the receiver

sensitivity, such as in a OTDMA uplink.

In assessing the phase acquisition capability of the receiver, it is found

that the use of a twice over-sampling SONET CDR with a phase picking

algorithm implemented in digital logic, provides a means of instantaneous

66

5 Conclusion

phase acquisition using zero preamble bits. Although the use of twice

over-sampling introduces a power penalty, it allows the preamble bits to

be used for amplitude recovery or increasing the information rate. It is also

found that the non-ideal sampling of the recovered clock introduces a

negligible penalty compared to the global clock.

67

References

References

[1] A. Girard, FTTx PON technology and testing, Quebec City, Canada:

Electro-Optical Engineering Inc., ISBN 1-55342-006-3, 2005.

[2] C.-H. Lee, W. V. Sorin, and B. Y. Kim, “Fiber to the home using a

PON infrastructure,” IEEE J. Lightwave Technol., vol. 24, no. 12, pp.

4568-4583, Dec. 2006.

[3] M. Abrams, P. C. Becker, Y. Fujimoto, V. O’Byrne, and D. Piehler,

“FTTP deployments in the United States and Japan-equipment

choices and service provider imperatives,” IEEE J. Lightwave

Technol., vol. 23, no. 1, pp. 236-246, Jan. 2005.

[4] R. E. Wagner, J. R. Igel, R. Whitman, M. D. Vaughn, A. B. Ruffin,

and S. Bickham, “Fiberbased broadband-access deployment in the

United States,” IEEE J. Lightwave Technol., vol. 24, no. 12, pp. 4526-

4540, Dec. 2006.

[5] R. Feldman, E. Harstead, S. Jiang, T. Wood, and M. Zirngibl, “An

evaluation of architectures incorporating wavelength division

multiplexing for broad-band fiber access,” IEEE J. Lightwave

Technol., vol. 16, no. 9, pp. 1546-1559, Sep. 1998.

[6] D. Jung, S. Shin, C.-H. Lee, and Y. Chung, “Wavelength-division-

multiplexed passive optical network based on spectrum slicing

techniques,” IEEE Photon. Technol. Lett., vol. 10, no.9, pp. 1334-

1336, Sep. 1998.

68

References

[7] C. Arellano, C. Bock, J. Prat, and K.-D. Langer, “RSOA-based optical

network units for WDM PON,” in Proc. OFC, Anaheim, CA, Mar.

2006.

[8] J. Prat, C. Arellano, V. Polo, and C. Bock, “Optical network unit

based on a bidirectional reflective semiconductor optical amplifier for

Fiber-to-the-Home netwoks,” IEEE Photon. Technol. Lett., vol. 18,

no. 1, pp. 250-252, Jan. 2005.

[9] T. Hamanaka, X. Wang, N. Wada, A. Nishiki, and K.-I. Kitayama,

“Ten-user truly asynchronous gigabit OCDMA transmission

experiment with a 511-chip SSFBG en/decoder,” IEEE J. Lightwave

Technol., vol. 24, no. 1, pp. 95-102, Jan. 2006.

[10] K.-I. Kitayama, X. Wang, and N. Wada, “OCDMA over WDM PON-

solution path to gigabitsymmetric FTTH,” IEEE J. Lightwave

Technol., vol. 24, no. 4, pp. 1654-1662, Apr. 2006.

[11] J. Penon, Z. A. El-Sahn, L. A. Rusch, and S. LaRochelle, “Spectral-

amplitude-coded OCDMA optimized for a realistic FBG frequency

response,” IEEE J. Lightwave Technol., vol. 25, no. 5, pp. 1256-

1263, May 2007.

[12] N. Kheder, Z. A. El-Sahn, B. J. Shastri, M. Zeng, L. A. Rusch, and D.

V. Plant, “Performance of Incoherent SAC-OCDMA Using a Burst-

Mode Receiver with CDR and FEC,” in Proc. LEOS Annual Meeting,

Orlando, FL, pp. 610 – 611, Oct. 2007.

69

References

[13] Z. A. El-Sahn, B. J. Shastri, M. Zeng, N. Kheder, D. V. Plant, and L.

A. Rusch, “Experimental Demonstration of a SAC-OCDMA PON with

Burst-Mode Reception: Local Versus Centralized Sources,” IEEE J.

Lightwave Technol., accepted for future publication.

[14] B. J. Shastri, Z. A. El-Sahn, M. Zeng, N. Kheder, L. A. Rusch, and D.

V. Plant, “A Standalone Burst-Mode Receiver with Clock and Data

Recovery, Clock Phase Alignment, and RS(255, 239) Codes for

SAC-OCDMA Applications,” IEEE Photon. Technol. Lett. 2008, Vol.

20, no. 5, pp. 363 – 365, Mar. 2008.

[15] Z. A. El-Sahn, M. Zeng, B. J. Shastri, N. Kheder, D. V. Plant, and L.

A. Rusch, “Dual Architecture Uplink Demonstration of a 7×622 Mbps

SAC-OCDMA PON Using a Burst-Mode Receiver,” in Proc. OFC,

San Diego, CA, Feb 2008, OMR3.

[16] B. J. Shastri, N. Kheder, and D. V. Plant, “Effect of Channel

Impairments on the Performance of Burst-Mode Receivers in Gigabit

PON,” accepted in the IEEE Midwest Symposium on Circuits and

Systems (MWSCAS)/ Northeast Workshop on Circuits and Systems

(NEWCAS) 2008.

[17] B. Sklar, Digital Communications: Fundamentals and Applications,

2nd Ed., Upper Saddle River, NJ: Prentice Hall, 2001.

[18] Paul. R. Prucnal, Optical Code Division Multiple Access:

Fundamentals and Applications, Chapter 3 – 6, Taylor and Francis,

2006.

70

References

[19] Glen Kramer, Ethernet Passive Optical Networks, Chapter 2,

McGraw Hill, 2005.

[20] Stok & Sargent, “Lighting the local area: Optical Code-Division

Multiple-Access and Quality of Service Provisioning,” IEEE Network,

Nov/Dec. 2000, pp. 42 – 46.

[21] Gerd Keiser, Optical Fiber Communications, 3rd ed., Chapter 12,

McGraw Hill, 2000.

[22] V. Baby, “Subsystems, Systems and Interfaces for Flexible

Bandwidth Allocation in Optical CDMA Networks,” PhD dissertation,

Dept. of Elec. Eng., Princeton University, Princeton, NJ, 2006.

[23] M. Kavehrad and D. Zaccarin, “Optical code-division-multiplexed

systems based on spectral encoding of noncoherent sources,” IEEE

J. of Lightwave Technol., vol. 13, no. 3, March 1995, pp. 534-545.

[24] D. Zaccarin, K. Kavehrad, “An Optical CDMA system based on

spectral encoding of an LED,” IEEE Photon. Technol. Lett. Vol. 4, no.

4, pp. 479 – 482, Apr. 1993.

[25] K. Kitayama, “Novel spatial spread spectrum based fiber optic CDMA

networks for image transmission,” IEEE J. of Selected Areas in

Communications, vol. 12, no. 4, pp. 762-772, May 1994.

[26] A. A. Hassan, J. E. Hershey, and N. A. Riza, “Spatial Optical CDMA”,

IEEE J. of Selected Areas in Communications, vol. 13, no. 3, pp.

609–613, Apr. 1995.

71

References

[27] G.-C. Yang and W. C. Kwong, “Two-dimensional spatial signature

patterns,” IEEE Transactions on Communications, vol. 44, no. 2, pp.

184-191, Feb. 1996.

[28] P. R. Prucnal, M. A. Santoro, and T. R. Fan, “Spread Spectrum

Fiber-Optic Local Area Network Using Optical Processing,” IEEE J. of

Lightwave Technol., vol. 4, no. 5, pp. 547-554, May 1986.

[29] J. A. Salehi, “Emerging Optical Code-Division Multiple Access

Communications Systems,” IEEE Network, vol. 3, no.2, pp. 31-39,

Mar. 1989.

[30] W. C. Kwong, P. R. Prucnal, and Y. – L. Liu, “All-serial Coding

Architecture for Ultrafast Optical Code-Division Multiple Access,” ICC

Technical Program, vol. 1, pp. 552 - 556, May 1993.

[31] C.-C. Yang and J.-F. Huang, “Two-Dimensional M-Matrices Coding in

Spatial / Frequency Optical CDMA Networks,” IEEE Photon. Technol.

Lett., vol. 15, no. 1, pp. 168-170, Jan. 2003.

[32] E. Park, A. J. Mendez, and E. M. Garmire, “Temporal/Spatial Optical

CDMA Networks-Design, Demonstration and Comparison with

Temporal Networks,” IEEE Photon. Technol. Lett., vol 4, no. 10, pp.

1160 – 1162, Oct. 1992.

[33] S. Kim, K. Yu, and N. Park, “A New Family of Space / Wavelength /

Time Spread Three-Dimensional Optical Code for OCDMA

Networks,” IEEE J. of Lightwave Technol., vol. 18, no. 4, pp. 502-

511, Apr. 2000.

72

References

[34] L. Tanceski, and I. Andonovic, ‘Wavelength-Hopping Time-Spreading

Code-Division Multiple Access Systems’, Electronics Letters, vol. 30,

no. 17, pp. 1388–1390, Aug. 1994.

[35] L. Tanceski, and I. Andonovic, ‘Wavelength Hopping/Time Spreading

Code Division Multiple Access Systems’, Electronics Letters, vol. 30,

no. 9, pp. 721–723, Aug. 1994.

[36] A. M. Weiner, J. P. Heritage, and J. A. Salehi, “Encoding and

Decoding of femtosecond pulses,” Optics Letters, vol. 13, no. 4, pp.

300-302, Apr. 1988.

[37] J. A. Salehi, A. M. Weiner, and J. P. Heritage, “Coherent ultrashort

light pulse code-division multiple access communication systems,”

IEEE J. of Lightwave Technol., vol. 8, no. 3, pp. 478-491, Mar. 1990.

[38] H. Sotobayashi, W. Chujo, K. Kitayama, “1.6–b/s/Hz 6.4 Tb/s QPSK-

OCDM/WDM (4OCDM x 40 WDM x 20Gb/s) transmission experiment

using optical hard thresholding,” IEEE Photon. Technol. Lett., vol. 14,

no. 4, pp.555-557, Apr. 2002.

[39] H. Sotobayashi, W. Chujo, K. Kitayama, “Highly Spectral Efficient

Optical Code Division Multiplexing Transmission System,” IEEE J.

Selected Top. Quant. Electron. Vol. 10, no. 2, pp. 250-258, Mar. –

Apr. 2004.

[40] V. J. Hernandez, W. Cong, J. Hu, C. Yang, N. K. Fontaine, R. P.

Scott, Z. Ding, B. H. Kolner, J. P. Heritage, and J. B. Yoo, “A 320-

Gb/s Capacity (32-User × 10 Gb/s) SPECTS O-CDMA Network

73

References

Testbed With Enhanced Spectral Efficiency Through Forward Error

Correction,” IEEE J. Lightwave Technol., vol. 25, no. 1, pp. 79 – 86,

Jan. 2007.

[41] C.–S. Bres, I. Glesk, and P. R. Prucnal, “Demonstration of an 8-user

115Gchip/s incoherent OCDMA system using supercontinuum

generation and optical time gating,” IEEE Photon. Technol. Lett., vol.

18, no. 7, pp. 889 – 891, Apr. 2006.

[42] V. Baby, C.–S. Bres, L. Xu, I. Glesk, and P.R. Prucnal,

“Demonstration of a differentiated service provisioning with 4-node

253 Gchip/s fast frequency-hopping time-spreading OCDMA,”

Electron. Lett., vol. 40, pp. 755 – 756, Jun. 2004.

[43] L. R. Chen, S. D. Benjamin, P. W. E. Smith, J. E. Sipe, “Wavelength-

encoding/temporal-spreading optical code division multiple-access

system with in-fiber chirped moire gratings,” Appl. Opt. Vol. 38, no.

21, pp. 4500 – 4508, 1999.

[44] S. Kutsuzawa, N. Minato, S. Oshiba, A. Nishiki, K. Kitayama, “10

Gb/s x 2 ch signal unrepeated transmission over 100 km of data rate

enhanced time-spread/wavelength-hopping OCDM using 2.3-Gb/s-

FBG en/decoder,” IEEE Photon. Technol. Lett. Vol. 15, no. 2, pp. 317

– 319, Feb. 2003.

[45] D. Pastor, W. Amaya, and R. Garcia-Olcina, “Design of high

reflectivity superstructured FBG for coherent OCDMA employing

74

References

synthesis approach,” IEEE Electron. Lett., vol. 43, no. 15, pp. 824-

825, Jul. 2007.

[46] M. Nakamura, Y. Imai, Y. Umeda, J. Endo, Y. Akatsu, “1.25-Gb/s

Burst-Mode Receiver ICs With Quick Response for PON Systems,”

IEEE J. of Solid State Circuits, vol. 40, no. 12, pp. 2680 – 2688, Dec.

2005.

[47] C. Su, L.-K. Chen, and K.-W. Cheung, “Theory of burst-mode

receiver and its application in optical multi-access networks,” IEEE J.

Lightwave Technol., vol. 15, no. 4, pp. 590-606, Apr. 1997.

[48] K. Schneider, H. Zimmermann, “Three-Stage Burst-Mode

Transimpedance Amplifier in Deep-Sub-/spl mu/m CMOS

Technology ,” IEEE Transactions on Circuits and Systems I: Regular

Papers, vol. 53, no. 7, pp. 1458 – 1467, Jul. 2006.

[49] Q. Le, S. G. Lee, Y. H. Oh, H.-Y. Kang, T.-H. Yoo, “A Burst-Mode

Receiver for 1.25-Gb/s Ethernet PON with AGC and Internally

Created Reset Signal,” IEEE J. of Solid State Circuits, Vol. 39, no.

12, Dec. 2004.

[50] P. Ossieur, D. Verhulst, Y. Martens, W. Chen, J. Bauwelinck, X. Z.

Qiu, J. Vandewege, “A 1.25-Gb/s burst-mode receiver for GPON

applications,” IEEE J. of Solid State Circuits, Vol. 40, no. 5, pp. 1180

– 1189, May 2005.

75

References

[51] C. A. Eldering, “Theoretical determination of sensitivity penalty for

burst mode fiber optic receivers,” IEEE J. Lightwave Technol., vol.

11, no. 12, pp. 2145-2149, Dec. 1993.

[52] P. M. Valdes, “Performance of optical direct receivers using noise

corrupted decision threshold,” IEEE J. Lightwave Technol., vol. 13,

no. 11, pp. 2202-2214, Nov. 1995.

[53] Y. Ota and R. G. Swartz, “Burst-mode compatible optical receiver

with large dynamic range,” IEEE J. Lightwave Technol., vol. 8, no.12,

pp. 1897-1903, Dec. 1990.

[54] Y. Ota and R. G. Swartz, “DC 1-Gb/s burst mode compatible receiver

for optical bus application,” IEEE J. Lightwave Technol., vol. 10, no.

2, pp. 244-249, Feb. 1992.

[55] C.-H. Yu and D.-U. Li, “A 2.5 Gb/s CMOS Burst-Mode Limiting

Amplifier for GPON System,” ISCAS, pp. 2538 – 2541, May 2008.

[56] A. Li, J. Faucher, and D. V. Plant, “Burst-mode clock and data

recovery in optical multi-access networks using broad-band PLLs,”

IEEE. Photon. Technol. Lett., vol. 18, no. 1, pp. 73-75, Jan. 2006.

[57] J. Lee and B. Kim, “A low-noise fast-lock phase-locked loop with

adaptive bandwidth control,” IEEE J. Solid State Circuits, vol. 35, no.

8, 1137 – 1145, Aug. 2000.

[58] C.-F. Liang; S.-C. H.; S.-I. Liu, “A 10Gbps Burst-Mode CDR Circuit in

0.18ýým CMOS” Proc. CICC, pp. 599 – 602, Sep. 2006.

76

References

[59] L.-C. Cho; C. Lee; S.-I. Liu; “A 33.6-to-33.8Gb/s Burst-Mode CDR in

90nm CMOS,” Proc. ISSCC, pp. 48 – 49 and 586, Feb. 2007.

[60] M. Banu and A. E. Dunlop, “Clock recovery circuits with

instantaneous locking,” Electronic Lett., vol. 28, no. 23, pp. 2127 –

2130, Nov. 1992.

[61] S. Kobayashi and M. Hashimoto, “A multi bitrate burst-mode CDR

circuit with bit-rate discrimination function from 52 to 1244 Mb/s,”

IEEE Photon. Technol. Lett, vol. 13, no. 11, pp. 1221-1223, Nov.

2001.

[62] J. Faucher, M. Y. Mukadam, A. Li, and D. V. Plant, “622/1244 Mb/s

burst-mode CDR for GPONs,” in Proc. IEEE LEOS Annual Meeting,

pp. 420 – 421, Oct. 2006.

[63] C.-F. Liang; S.-C. Hwu and S.-I. Liu, “A 2.5Gbps Burst-Mode Clock

and Data Recovery Circuit,” Asian Solid-State Circuits Conference,

pp. 457 – 460, Nov. 2005.

[64] S. Lee, M. Hwang, Y. Choi, S. Kim, Y. Moon, B. Lee, D. Jeong, W.

Kim, Y. June Park, and G. Ahn, “A 5Gb/s 0.25μm CMOS jitter-

tolerant variable-interval oversampling clock/data recovery circuit”,

IEEE J. of Solid-State Circuits, vol. 37, no. 12, pp. 1822- 1830, Dec.

2002.

[65] Y.-H. Moon, and J.-K. Kang, “2× oversampling 2.5 Gbps clock and

data recovery with phase picking method,” Current Applied Physics,

vol. 4, no. 1, pp. 75-81, Feb. 2004.

77

http://ieeexplore.ieee.org/xpl/RecentCon.jsp?punumber=4017511

http://ieeexplore.ieee.org/xpl/RecentCon.jsp?punumber=4017511

References

[66] I. S. Reed and G. Solomon, “Polynomial codes over certain finite

fields,” SIAM J. Appl. Math., vol. 8, pp. 300–304, Jun. 1960.

[67] Y. Sugiyama, M. Kasahara, S. Hirasawa, and T. Namekawa, “A

method for solving key equation for decoding Goppa codes,”

Information and Control, M. Eden, Ed. New York: Academic, vol. 27,

pp. 87–99, 1975.

[68] Interfaces for the Optical Transport Network (OTN), ITU-T Rec.

G709, Mar. 2003.

[69] G. A. Magel, G. D. Landry, R. J. Baca, D. A. Harper and C. A.

Spillers, “Transmission of 8 channels x 622 Mb/s and 15 channels x

155 Mb/s using spectral encoded optical CDMA,” Electron. Lett., vol.

37, pp. 1307 – 1308, Oct. 2001.

[70] W. Xu, N. Wad, G. Cincotti, T. Miyazaki, K. Kitayama,

“Demonstration of 12-user, 10.71 Gbps truly asynchronous OCDMA

using FEC and a pair of multi-port optical encoder/decoders,” in Proc.

ECOC 2005, vol. 5, pp. 53 – 54, Glasgow, Scotland, Sep. 2005.

[71] V. J. Hernandez, W. Cong, R. P. Scott, C. Yang, N. K. Fontaine, B.

H. Kolner, J. P. Heritage, and S. J. B. Yoo, “320 Gb/s capacity (32

users x 10 Gb/s) SPECTS O-CDMA local area network test-bed,” in

Proc. OFC 2006, Anaheim, CA, pp. 1 – 3, Mar. 2006.

[72] A.J. Viterbi, “Convolutional codes and their performance in

communication systems,” IEEE Trans. Commun. Technol., vol. 19,

no. 5, pp. 751–772, Oct. 1971.

78

References

[73] G. Jr. Forney, “Convolutional codes I: Algebraic structure,” IEEE

Trans. Inform. Theory, vol. 16, no. 6, pp. 720–738, Nov. 1970.

[74] A. Viterbi, “An intuitive justification and a simplified implementation of

the MAP decoder for convolutional codes,” IEEE J. Select. Areas

Commun., vol. 16, no. 2, pp. 260–264, Feb. 1998.

[75] G. Vareille, O. A. Sab, G. Bassier, J. P. Collet, B. Julien, D.

Dufournet, F. Pitel, and J. F. Marcerou, “1.5 terabit/s submarine 4000

km system validation over a deployed line with industrial margins

using 25 GHz channel spacing and NRZ format over NZDSF,” in

Proc. OFC, Anaheim, CA, pp. 293 – 295, Mar. 2002.

[76] A. Puc, F. Kerfoot, A. Simons, and D. L. Wilson, “Concatenated FEC

experiment over 5000 km long straight line WDM test bed,” in Proc.

OFC, San Diego, CA, vol. 3, pp. 255 – 258, Feb. 1999.

[77] J.-X. Cai, M. Nissov, A. N. Pilipetskii, A. J. Lucero, C. R. Davidson, D.

Foursa, H. Kidorf, M. A. Mills, R. Menges, P. C. Corbett, D. Sutton,

and N. S. Bergano, “2.4 Tb/s (120_20 Gb/s) transmission over

transoceanic distance using optimum FEC overhead and 48%

spectral efficiency,” in Proc. OFC, Anaheim, CA, vol. 4, pp. PD20-1 –

PD20-3, Mar. 2001.

[78] Y. Yamada, S. Nakagawa, T. Kawazawa, H. Taga, and K. Goto, “2

Tbit/s (200_10 Gbit/s) over 9200 km transmission experiments using

C-band EDFA and VSB format with 53% spectral efficiency,” in Proc.

SubOptic, Kyoto, Japan, May 2001.

79

References

[79] B. Vasic and I. B. Djordjevic, “Low-density parity check for long-haul

optical communication systems,” IEEE Photon. Technol. Lett., vol.

14, no. 8, pp. 1208–1210, Aug. 2002.

[80] I. B. Djordjevic and B.Vasic, “Projective geometry LDPC codes for

ultra long-haulWDMhigh-speed transmission,” IEEE Photon. Technol.

Lett., vol. 15, no. 5, pp. 784–786, May 2003.

[81] T. Mizuochi, Y. Miyata, T. Kobayashi, K. Ouchi, K. Kuno, K. Kubo, K.

Shimizu, H. Tagami, H. Yoshida, H. Fujita, M. Akita, and K.

Motoshima, “Forward Error Correction Based on Block Turbo Code

With 3-Bit Soft Decision for 10-Gb/s Optical Communication

Systems,” IEEE J. of Selected Topics in Quan. Electronics, vol. 10,

no. 2, pp. 376 – 386, Mar/Apr. 2004.

[82] H. Tagami, T. Kobayashi, Y. Miyata, K. Ouchi, K. Sawada, K. Kubo,

K. Kuno, H. Yoshida, K. Shimizu, T. Mizuochi, and K. Motoshima, “A

3-bit Soft-Decision IC for Powerful Forward Error Correction in 10-

Gb/s Optical Communication Systems,” IEEE J. of Solid-State

Circuits, vol. 40, no. 8, pp. 1695 – 1705, Aug. 2005.

[83] Z. Wei and H. G. Shiraz, “Unipolar Codes with Ideal In-Phase Cross-

Correlation for Spectral Amplitude-Coding Optical CDMA Systems,”

IEEE Transactions on Communications, vol. 50, no. 8, pp. 1209 –

1212, Aug. 2002.

80

References

[84] J. Faucher, S. Ayotte, L.A. Rusch, S. LaRochelle and D.V. Plant,

“Experimental BER performance of 2D λ-t OCDMA with recovered

clock,” Electronic Letters, vol. 41, no. 12, pp. 713 – 715, Jun. 2005.

[85] J. Faucher, “Burst-mode clock and data recovery circuits for optical

multi-access networks,” PhD dissertation, Dept. of Elec. Eng., McGill

University, Montreal, Canada, 2006.

[86] ITU-T, “Broadband optical access systems based on passive optical

networks,” Recommendation G.984.2, 2003.

[87] J. Faucher, S. Ayotte, Z. A. El-Sahn, M. Mukadam, L. A. Rusch, and

D. V. Plant, “A standalone receiver with multiple access interference

rejection, clock and data recovery, and FEC for 2-D λ-t OCDMA,”

IEEE Photonics Technol. Lett., vol. 18, pp. 2123-2125, Oct. 2006.

81

Experimental Demonstration of OCDMA and OTDMA …digitool.library.mcgill.ca/thesisfile19245.pdf ·...

Documents

Transcript of Experimental Demonstration of OCDMA and OTDMA …digitool.library.mcgill.ca/thesisfile19245.pdf ·...