Eindhoven University of Technology MASTER Horizontale ...Eindhoven University of Technology MASTER
Transcript of Eindhoven University of Technology MASTER Horizontale ...Eindhoven University of Technology MASTER
Eindhoven University of Technology
MASTER
Horizontale afbuiging met symmetrisch heen- en teruglopende afbuigstroom voor TV-ontvangers
Frensch, A.J.
Award date:1984
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AFDELING DER ELEKTROTECHNIEK
TECHNISCHE HOGESCHOOL EINDHOVEN
VAKGROEP TELECOMMUNICATIE EC
HORIZONTALE AFBUIGING MET SYMMETRISCH
HEEN- EN TERUGLOPENDE AFBUIGSTROOM VOOR
TV-ONTVANGERS
(HORIZONTAL DEFLECTION IN TV-RECEIVERS
USING A SYMMETRIC CURRENT)
door: AdJ. Frensch
van januari 1983 tot maart 1984
Afstudeerhoogleraar: Prof. Dr. I.C. Arnbak
Begeleiders: Dr. Ing. U.E. Kraus,
Hoofdindustriegroep Video,
Nederlandse Philips Bedrijven B.V.,
TV-Lab., Signal Processing Groep.
Ir. A.P. Verlijsdonk.
De afdeling der elektrotechniek van de Technische Hogeschool
Eindhoven aanvaardt geen verantwoordelijkheid voor de inhoud
van stage- en afstudeerverslagen.
CONTENTS
SAMENVATTING
ABSTRACT
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LIST OF SYMBOLS AND ABBREVIATIONS
1 • INTRODUCTION 1. I The horizontal deflection circuit in the con
ventional colour TV receiver 1.2 Power dissipation, high voltage and radiation
problems in the standard horizontal deflection circuit at different line frequencies
1.3 Symmetrie horizontal deflection currents
2. THEORY 2.1 The effects of sinusoidal horizontal deflection
currents on the geometry of the picture 2.1.1 Introduetion 2. 1.2.Bounds on the deflection angle ~D 2.1.3 Bounds on the deflection current ih 2.1.4 Sinusoidal currents for horizontal
deflection
page
3
5
7
10 12
19
24
28 30
30 32 42 44
2.2 Generation of the sinusoidal current 48 2.2.1 The ser resonance circuit 49 2.2.2 The parallel resonance circuit 50 2.2.3 Removal of pin-cushion effect: Qualitative 52
consideration of an amplitude modulating signal.
2.3 Deflection current phase modulation due to 54 cross-coupling between the deflection circuits 2.3.1 Variation of e 1n the series resonance 55
circuit 2.3.2 Variation of e in the parallel resonance 56
circuit 2.4 Alignment of the lines scanned in opposite direction 58 2.5 The effects of the symmetrie deflection current 67
on the scanning raster
3. HARDWARE REALISATION 71 3.1 Design of a breadboard model for sinusoidal 71
horizontal deflections currents 3.2 The line memories and blanking control (CONTROL) 75 3.3 The memory enabling (VIE/VOE) unit 81 3.4 The clock and state signals (CLOCK/ST) unit 87 3.5 The sinusoidal deflection current (SIN) unit 91 3.6 The horizontal deflection current-measurement 93
processing (IID•i PROC) unit
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4. DISCUSSION AND EXPERIMENTAL RESULTS 4.1 Choice of amplitude and generation methad
of the sinusciclal deflection current 4.2 Choice of alignment 4.3 Power dissipation, high voltage and radiation
in the sinusciclal driven horizontal deflection circuit
CONCLUSIONS
REPERENCES
page
96 96
100 102
105
107
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SAMENVATTING
Het aftastschema dat wordt gebruikt in de huidige TV-ontvangers wordt tot stand gebracht door een zaagtandvormige stroom in zowel de horizontale als verticale afbuigspoel. Voor het horizontale afbuigcircuit heeft deze golfvorm de volgende nadelen: - Grote verrnogensdissipatie; - hoge spanningsbelasting van de componenten in het horizontale
afbuigcircuit; - straling. Deze nadelen zullen nog toenemen als de lijnfrequentie wordt verhoogd om grootvlakflikker en interlijnflikker te elimineren.
Het functiemodel dat hier wordt beschreven gebruikt een sinusvormige horizontale afbuigstroom gegenereerd met behulp van een serieresonantiecircuit. Omdat zowel de stijgende als dalende gedeeltes van deze stroom worden gebruikt om het scherm af te tasten, moet, om een hornogeen aftastschema te verkrijgen waarin de aftastgedeeltes van het A-raster en het B-raster elkaar niet snijden, de verticale afbuigstroom worden veranderd in een trapjesvormige stroom. Doordat het scherm in tegengestelde richtingen wordt afgetast, zal de video-informatie om de andere lijn moeten worden omgedraaid. De omdraaioperatie en het richten van de in tegengestelde richting afgetaste lijnen is digitaal gerealiseerd op kleinsignaal-nivo. De correctie van de speldenkussenvervorming is ook op kleinsignaal-nivo gerealiseerd, door de samples van het amplitudegemoduleerde signaal, dat gebruikt wordt om de eindversterker van het horizontale afbuigcircuit te sturen, op te slaan in een EPROM. De hoge spanning voor de beeldbuis wordt onafhankelijk van het horizontale afbuigcircuit opgewekt.
Bij een lijnfrequentie van 31,25 kHz, zoals wordt gebruikt in een 100Hz-TV-ontvanger (26 inch beeldbuis), is het totale in de horizontale spoel gedissipeerde vermogen teruggebracht van 15,2 watt (in het geval van de zaagtandvormige afbuigstroom) tot 8.75 watt voor de sinusvormige afbuigstroorn. De spanning over de horizontale afbuigspoel is teruggebracht van ca. 2500 volt piekpiek tot 700 volt piek-piek met gebruik van dezelfde zelfinduktie als in de conventionele ontvanger bij 15,625 kHz. Ook de door het horizontale circuit veroorzaakte straling is aanzienlijk verminderd. De scheiding van de opwekking van de hoge spanning voor de beeldbuis en de opwekking van de horizontale afbuigstroom heeft als voordeel dat beide onafhankelijk van elkaar kunnen worden geoptimaliseerd (b.v. verlaging van de inwendige weerstand van de hoogspanningsbron). Voordat een volledige vergelijking, betreffende beeldkwaliteit en vermogensdissipatie, kan worden gemaakt, moet het functiemodel worden afgebouwd. Op dit moment blijkt echter al, dat het vermogen dat wordt gedissipeerd in een ontvanger die een sinusvormige afbuigstroorn gebruikt, veel minder is dan het vermogen dat wordt gedissipeerd in een conventionele ontvanger, die een zaagtandvormige
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afbuigstroom gebruikt, vooral bij lijnfrequenties van 31,25 kHz en hoger. Deze vermindering in vermogensdissipatie, die ook veroorzaakt wordt door een vermindering van het aantal grootsignaal-circuits, gaat ten koste van extra kleinsignaal-circuits. Door monolithische integratie kunnen de afmetingen en kosten van deze circuits worden gereduceerd.
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ABSTRACT
The scanning raster 1n most TV-sets today is produced by means of sawtooth-current wavefarms in both the vertical and horizontal deflection coils. For the horizontal deflection circuit this kind of wavefarm has the following disadvantages: - High power dissipation; - high voltages over the components in the horizontal deflection
circuit; - radiation. These disadvantages will become even larger the line frequency is increased in order to eliminate large-area flicker and interline flicker.
The breadboard model discribed here uses a sinusoidal horizontal deflection current generated with the aid of a series resonance circuit. Since both the rising and traili:ng parts of this current are used to scan the screen, the vertical deflection current waveform has to be changed into a staircase current wavefarm in order to get a homogeneaus scanning pattern, in which the scanparts of the A- and B-field do not intersect. The scanning in opposite directions involves the reversal of the video information in every other line. The reversing operation and the alignment of the lines scanned in opposite directions, has been realized digitally, at small-signal level. The correction of the pin-cushion distartion has also been realized at small-signal level, by storing the samples of the amplitude modulated signal, used to drive the amplifier end-stage of the horizontal deflection current, in an EPROM. The high vol for the picture tube is realized 1n dependently from the horizontal deflection circuit.
At a line frequency of 31.25 kHz, as used in a 100Hz TV-set (26 inch picture tube) the total power dissipated in the horizontal deflection yoke has been reduced from 15.2 watt (in case of the sawtooth current waveform) to 8.75 watt for the si:nusoidal deflection current. The voltage over the horizontal deflection yoke has been reduced from ca. 2500 volt peak-to-peak to 700 volt peak-to-peak with the same self inductance (1.35 mH) as at the 15.625 kHz line frequency in standard TV-sets. The radiation caused by the horizontal deflection circuit has also been reduced. The separation of the high voltage souree and the horizontal deflection circuit makes it possible to optimize them separtely (e.g. reduction of the internal resistance of the high voltage source).
Before a full comparison concerning picture quality and power dissipation can be made, the breadboard model has to be completed. Already at this moment, however, it appears that the power dissipated in a TV-set which uses a sinusoidal horizontal deflection current is for less than the power dissipated in a conventional TV-set, which uses a sawtooth current waveform, especially at line frequencies of 31.25 kHz and higher.
This reduction in power dissipation which is also caused by a reduction in large signal circuitry, will be at the expense of extra small signal circuitry. Monolithic integration is a possibility for reducing this circuitry in size and cost.
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LIST OF SYMBOLS AND ABBREVIATIONS
ff field frequency;
fh frequency of the horizontal deflection current;
f 1 line frequency;
h viewing screen height;
~h horizontal deflection current;
~ souree current for the parallel resonance circuit generating s
the horizontal deflection current;
~ vertical deflection current; V
k
s
s s
the sealing factor with which the active time (T ) a
is reduced, the video signals compressed, and the spot
velocity (v) increased;
humer of windings of the horizontal deflection coil; 6 Ta . = ~ , fract~on of the line time (T
1) carrying video
1 information;
coordinate of the spot along the path on the screen;
souree signal for the resonance circuit, generating the
horizontal deflection current (s is v or i ); s s s t time variable;
v spot velocity;
vh voltage over the horizontal deflection coil;
v souree voltage for the series resonance circuit generating s
the horizontal deflection circuit;
v voltage over the vertical deflection coil V
w v~ew~ng screen width
A blanking signal in TV-sets which have ff = 50 Hz;
Ah amplitude of the actual sinusoidal horizontal deflection
current;
A normalised amplitude of the sinusoidal deflection current; n
A amplitude of the sinusoÏdal souree signal (s) driving the s s
resonance circuit;
B "Blue" video information signal for ff 50 Hz or ff 100 Hz;
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B re-arranged vers~on of B used for scanning with a symmetrie sym
horizontal deflection current;
C normalisation constant for the horizontal deflection current
(ih);
c a
capacitor ~n the resonance circuit tuned to Lh such that I
w =---h
LhCa
G "green" video information signal for ff = 50 Hz or ff = 100 Hz;
G re-arranged version of G used for scanning with a symmetrie sym
horizontal deflection current;
H line synchronisation signal for f 1 = 15.625 kHz;
Lh self-inductance of the horizontal deflection coil;
1 self-inductance of the vertical deflection coil; V
"Red" video information signal for ff = 50 Hz or ff
resistance of the horizontal deflection coil;
R magnetic resistance; m
100 Hz;
R re-arranged version of R used for scanning with a symmetrie sym
horizontal deflection coil;
RD distance between the centre of the deflection system and the
screen;
R screen radius; s
T time period during which the spot ~s actually visible on the a
screen (active time);
Tf period of one field;
Tfb time period during which fly-back takes place;
Th period of the horizontal deflection current,in case of the
conventional system Th= T1
, in case of the symmetrie de
flection system described here Th= 2T1 ;
T1
period of one line;
V field synchronisation signal for ff = 50 Hz;
2A blanking signal in TV-sets which have ff 100 Hz;
2H line synchronisation signal for f1
= 31.25 kHz;
2V field synchronisation signal for ff = 100 Hz
al
e
p
T
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phase difference between the r1s1ng edge of the ST
signal and the following zero-crossing of the s1nus-di
oidal deflection current 1h' such that ~ 1S positive; dt
phase difference between the souree signal s and the s
horizontal deflection current ih; 6 I . = ~ RS' normal1sed screen curvature;
ê vt/~, norrnalised time variable (or normalised dis
tance) used for calculating the desired horizontal de
flection current;
Wh angular velocity of the sinusoidal horizontal deflec-
tion current;
norrnalised angular velocity of the sinusoidal horizontal
deflection current;
line duet a sequence of one LR line and one RL line including the
overscan parts;
LR line the piece of the video signal (R G B ) which syrn' syrn' syrn
will be used to reproduce the picture with the reprodu-
C1ng spot rnov1ng frorn Left to !ight on the screen;
RL line the piece of the video signal (R G B ) which syrn' syrn' syrn
will be used to reproduce the picture with the repro-
ducing spot rnaving frorn !ight to Left on the screen;
ST signal the logic signal at half the line frequency deciding
whether the line of the i~~2~igg video signal will be
corne an LR line or an RL line; if ST is "low", then the
line of the incorning video signal will becorne an LR line;
if ST is "low", it will becorne an RL line.
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I. INTRODUCTION
In recent years, the quality of the ?icture displayed by consu
mer, television receivers has been improved to the extent that
further improvP.ments are difficult to achieve within the frame-urk
of existing display techniques. Additional impravement could be
obtained through improvements in colour picture tubes, better sig
nal processing, and through modification of current video-signal
transmission standards, i.e. NTSC, PAL and SECAH:[1.2]. Howevermo
dification of these standards may not be acceptable unless the
changes are compatible with the circuitry in the receivers alrea
dy installed in millions of homes and do not make these receivers
obsolete. Alternative approaches are being explored that do not re
quire a change in transmission standards, but provide an imprave
ment in observed picture quality through the elimination of cer
tain artifacts in the picture; i.a. large-area flicker and inter
line flicker. Lare area flicker will disappear completely when
the number of fields per second is dou;.;led; i.e. from 50 to I 00 Hz
for the European systems· (PAL and SECAM), and from 60 to 120 Hz
for the A~erican system (NTSC).
For the PAL system this has already been worked out by
Kraus [3]. In fact, the system which will be considered bere, is to
be used 1n combination with a 100 Hz colour TV-set according
to [3]. The increase in field frequency from 50 to 100Hz
implies an increase in line frequency from 15.625 kHz to 31.25 kHz.
A technique under evaluation for the NTSC system is the impravement
of picture quality by a change in the horizontal scanning system
from the current 525 line interlaced system to a 525 line non-1n
terlaced system (progressive-scan) system. This would involve an
increase in the line frequency from 15.75 kHz to 31.5 kHz.
Bath quality-improvement techniques described here need a
doubling of the line frequency, but do not require any change 111
the transmitted signal. Note that a combination of both techniques
for the NTSC system would involve an increase in line frequency
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from 15. 7 5 kHz to 63 kHz, and for the PAL sys tem would invo 1 ve an
increase in line frequency from 15.625 kllz to 62.5 kHz, respecti
vely. The standard horizontal deflection circuit used in televi-
sion-receivers has the following disadvantages:
High power consumption;
high voltage over the deflection coil and other elements ~n the
circuit;
- radiation problems.
These disadvantages become even worse if the line frequency ~s ~n
creased [ 4l, as will be the case with the impler.1entation of the
described improvements. The disadvantages arise from the big chan
ges in the first derivative of the deflection current due to the
short fly-back time [ 4}.
A horizontal deflection system, in which both the r~s~ng and trai
ling part of a syrnmetr.ic deflection current, e.g. a triangular or
sinusoidal current, are used to scan the screen horizontally, can
be used as an alternative. The frequency of such a symmetrie cur
rent ~s half the line frequency. Such a deflection syste~ will not
have the disadvantages of the conventional deflection system but
there is now a new problem:
The three video signals (R, G and B), userl to drive a colour catho
de~ray tube, have to be turnen around every other line with shi~t
registers, and at the same time the lines "written" on the screen
from left to right have to be aligned with the lines "written" on
the screen from right to left.
This alignment problem can be solved fundamentally by determining
the starting moment of the read-operation using a direct measure
ment of the deflection current and its first derivative.
In this report a horizontal deflection system ~s considered, which
uses a sinusoidal deflection current. SDecial attention has to be
payed to:
- The power dissipation;
- the exactness of the control circuitry;
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the possibility of monolithic intergration of the total deflec
tion circuit.
In chapter 1 a conventional colour TV system and its standard ho
rizontal deflection system at the receiver is reviewed. The disad
vantages of this system as a function of the line frequency are
considered, and the system is compared with the alternative system,
which uses a symmetrical deflection current. The theoretical foun
dations for a breadboard model of the alternative horizontal deflec
tion system are treated in chapter 2. Chapter 3 gives the hardware
re al ion of this model. In chapter 4 the balance will be drawn
up, to see in what way the theoretical aspects, concerning the
breadboard model, correspond to the practical results. The exact
ness of the control circuitry will be considered, and the power
d sipation 1n the horizontal deflection circuit, based on a s1nus
oÏdal current, will be compared with the power dissipation in the
conventional horizontal deflection circuit. Finally we will consi
der roughly the possibility of monolithic integration of the total
deflection circuit.
1.1 The horizontal deflection circuit in the conventional colour
TV receiver
The problem of television (the transmission and the reproduetion
of moving scenes) is solved by subdividing the image field into a
sufficiently high number of picture elements and by periodically
transmitting electrical signals representing the light intensities
of the picture elements 1n a sequential way. The sequential elec
trical signals are generated by a raster scan of the image field
by means of a light sensitive probe (exploring element) of
which the aperture determines the size of the picture elements
(see figure 1.1). The reproduetion of the 1mage is realised by
moving a intensity-modulated small light souree (reproducing spot)
1n a raster scan over the display screen. Measures must be taken
to synchronize the recording and reproducing rasters. The probe,which
produces a voltage or current proportional to the light intensity,
starts at point A1
and moves with constant but unequal rates in the horizontal and vertical directions, following the path A1 A2.
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h
Fig. 1.1, [sJ: Scanning raster Cline spacing grossly exaggerated). Solid lines are the A field; dashed lines are the B field.
Upon reaching point A2 , the probe quickly flies back to A3
(the
horizontal fly-back) and proceeds similarly to point A4 . Then
the probe flies back vertically to B1
and follows an interlaced
pattern ending at B2.
The process is then repeated starting aga1n at A1 • The two sets
of lines are called the A and B fields; tagether they constitute
one complete picture or frame. The frame rate 1s just rapid
enough (25 per second) to create the illusion of continuous mo
tion, while the field rate (twice the frame rate) brings the
large-area flicker effect tri a still annoving but in general
tolerable level, the field sequence being ABABAB .... The com
plete system for this type of television transmission and recep
tion can be represented by the five basic elements shown in
figure 1.2. Communication
I channel I
1--- -f--- - f- -1--- --, Scanning >- -
-t._ patterns~ -- -~ -.._, ·'<
Exploring Reproducing spot element
Image field Viewing screen
Figure 1.2, [6]: Functional Representation of a black-and-white TV system.
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.These basic elements are: (I) Exploring element, (2) Scanning
raster on the image field. (3) Communication channel. (4) Repro
ducing spot. (5) Scanning raster on the v~ew~ng screen.
The system shown in figure 1.2 can only serve as a modelfora
black-and-white TV system. Figure 1.3 suggests how the system
might be generalised in case of a colour TV system.
Figure 1.3 depiets the fact that the colour sensation of nearly
any speetral light distribution within the visible range can
be produced by means of three monochromatic sourees (i.e, red,
green, and blue) which stimulate the three different light
sensitive recepters in the eye to the same extent as the origi
nal distribution. In the ~mage piek-up camera the original ~ma
ge is divided into three color components. These separated ~ma
ges are explored synchronously in the same way as the single
image, in the black-and-white system. Three signals (R, G and
~) are sent simultaneously to the receiver.
'1J
Red mq fiek:! R G channel ~
B
--LJ -
Green ·lra:Je fetj Vil?';ciirq screen
,_
- -o -
Blue ir1ldge fiekl
Fig. 1.3: Functional representation of a colour TV system.
"J i-'•
()Q . -. ..". ..
(/) <: 1-'• 1-'•
()Q 0.. ::l (I) Pol 0 1-' (/)
0" H'l 1-' 0 Pol 'i ::l
~ Vl 1-'• 0 ::l
()Q ::r: N Pol
Pol .,5. ::l 0.. 1-'
J-1.• ::l
0 (I) 0
Pol ::r: ::l N 0..
H H'l <: 1-'•
(l) 'i 1-' (l) 0.. (') (l) (/)
1-'• '< < ::l (I) (') 'i ::r' (/) 'i
0 ::l 1-'• (/)
Pol ("1"
1-'• 0 ::l
R, G Of B 50 or 100Hz
composi te bldnking A!SOJ Of 2AI100Hz} n
A field
I -------.!: ~ 1 I
fiE>Id-sync: V Of 2V
I B field :
B field
A field
R,G orB I
comp blanking I '-----:---------------------------J
I ine-synr
fi<?ld-sync
) rïr1r1r1r1r1r.-,rïrïr-lrïr-lr-lr-lr-lr-lr-lr-lr-lr-rlr-lr-lrïr-lrïr-lrïr-lr-lr
I
ll-~~----l
l=Tt = 64~s a=12~
5"0 Hz b cc5,51JS C= 1,51JS d= SIJS I!= fiJS
{
L= Tt = 321JS a= 61JS
100Hz b=2,15~J. s c~o?51JS d= 2 S~s e= l51JS
fig. 1.4a
b
(
d
e
f
g
h
Vl I
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In the rece~ver, as shown in figure 1.3, there are three collo
cated spots which produce the three images.
At a distance of 5 to 6 times the viewing screen height, these
three images are amalgamated by the eye, thus bringing back the
sensation of the original colour picture.
Blanking pulses are inserted before the video signals are sent
over the channel. They are inserted during the fly-back intervals
to blank out the fly-back lines at the receiving tube. The added
synchronisation signals are used to synchronize the horizontal
and vertical scanning mechanism at transmitter and receiver.
Figure 1.4 shows an example of one of the video signals (R, Gor
B), the blanking signal, and the line and field synchronisation
signals as a function of time for 50 and 100 Hz TV sets. The sig
nals used in this 100 Hz TV-set have the same waveferm as the
signals in the conventional 50 Hz TV-set, but they are compressed
in time.
The reproducing spot on the viewing screen (see figure 1.2)
formed by cathode-ray beams deflected over the display screen by
means of a suitable magnetic deflection system. It is the current
through the horizontal deflection coil which plays a major rêle
~n this report.
If the spot size ~s constant and we want the same resolution over
the whole picture, then the spot velocity of the exploring spot
(as well as of the reproducing spot) has to be constant, too.
In the case when the spot velocity is a linear function of the
deflection current, the deflection currents corresponding to the
scanning raster shown in figure 1.1 would have sawtooth wavefarms
for both vertical and horizontal direction. In practice only the
deflection angle is approximately a linear function of the cur
rent through its deflection coil. The spot velocity, however,
is not a linear function of the deflection angle. This can be
shown easily for a flat screen. A constant angular velocity
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will nat give a constant spot velocity s~nce the spot will move
faster at the extremes of the screen than in the middle (see
figure 1.5). Thus the linear deflection current must be changed
óS
óS ) 65 e m
centre of deflection
fig. 1.5: Spot displacement on a flat screen in case of a con
stant angular velocity.
into an S-shaped current in order to obtain a constant spot velo
city over the whole screen. This is especially felt with respect
to the current through the horizontal coil, since the width (w)
of the picture is greater than its height (h). For standard TV,
w/h = 4/3 (see figure 1.1). In its simplest embodiment the hori
zontal de.flection system of a conventional television receiver
consists of a magnetic deflection coil, a power souree and one
or more switches [7]. Figure 1.6 shows such a deflection system,
while figure 1.7 showsits associated current and voltage wave
farms. The switch in this case ~s a combination of a transistor
and a diode (damper). The circuit includes a fly-back capacitor
Cfb' which, tagether with the deflection coil Lh and the fly-back
coil, determines the length of the fly-back interval.
It also includes the capacitor C , which provides the duel func-s
tion of DC blocking and S shaping. An actual deflection circuit
used ~n a receiver would norrnally also include pin-cushion and
linearity correction circuitry plus a winding on the fly-back
coil to provide a scurce of high voltage for the picture tube.
The disadvantages of the described conventional horizontal de
flection circuit are:
- High power consumption;
high voltage over the deflection coil and other elements ~n the circuit;
- radiation problems.
Ie
HOR. OUTPUT
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Eoc
11 11 ISO LA Tl ON l 11 (Fl YBACK) 11
10 I (fb
DAMPER
(fb ~~OR 1~ YOKE
.. Lh Cs
Fig. 1.6, [4]: Basic horizontal deflection circuit 1n the con
ventional receiver.
Fig. 1.7 [4]: Current and voltage wavefarms in the
horizontal deflection circuit.
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1.2 Power dissipation, high voltage and radiation problems in the
standard horizontal deflection circuit at different line fre-
quenc~es
In the circuit of figure 1.6 there are 3 eleoents which dissi~ate
power, Le.: (I) The deflection yoke; (2) the transistor; (3) the
damper.
Power dissipation ~n the deflection yoke results from 3 factors,
[ 4] : (a) r 2
R losses in the windings;
(b) eddy current losses in the windings;
(c) hysteresis and eddy current losses ~n the ferrite yoke r~ng.
The power dissipated in the transistor ~s the sum of the power dis
sipation due to collector current during scan, the dissipation due
to the base drive, and the dissipation during fly-back.
W.E. Babcock and W.F. hledam [4] have detemined the losses ~n the
standard deflection circuit (shown in figure 1.6), at different
line frequencies, under the following conditions:
(a) The DC voltage is the same for all line frequencies;
(b) the peak stared energy in the yoke is constant for all line
frequencies; ~ 4,7 milijoules;
(c) the peak fly-back voltage is about the same (~ 900 volt) at
all line frequencies;
(d) the ratio of scan time and fly-back time ~s constant for all
line frequencies.
The fly-hack voltage is limited by reducing the number of windings
(nh) of the horizontal coil (thereby reducing the self-inductance
Lh; Lh::n~) and at the sametime increasing the current ih through
the coil with the same factor.
Tables I. I a, band c show the horizontal deflection yoke losses
for three ferrite ring materials tested at 15.75, 31.5 and 63kHz.
Figure 1.8 shows theselossesas a function of frequency.
line ency
-20-
Table 1.1, [~: Power losses 1n the horizontal deflection yoke
at d ferent line frequencies
table 1.1a:
LOSSES USING DEFLECTION YOKE lHTH RCA 804 FERRITE
122 total eddy eddy current
frequ- temp rise retrace P-P retrace tot al yoke yoke current & & hysteresis above amb. time voltage loss loss hysteresis loss during
loss retrace
15.75 kHz 19° c 12.3 psec 880 7.6 w 3. 95 1l 3.65 w 3.65 w 31.5 kHz 33° c 6. I psec 875 15.2 H 4.53 w 10.67 1\f 8.19 w 63 kHz 82° c 3.1 psec 865 52 1-1 5.5 w 46.5 w 38.4 w
table l,lb:
LOSSES USING DEFLECTION YOKE WITH PERRITE "A"
total eddy line temp rise retrace P-P retrace total yoke yoke 1
22 current & hys-frequency above amb. time voltage loss loss teresis loss
15.75 kHz 19.4° c 12.2 psec 920 6.7 w 4.05 w 2.55 w 11.5 kHz 14.4° c 6,08 psec 910 13.6 w 4.44 w 9.16 w 61 kHzlt --- --- --- --- ---
ltUnable to run at 63 kHz because core temperature exceeded curie temperature
table 1. Ie:
LOSSES USING DEFLECTION YOKE W!TH FERRITE "B"
line temn rise retrace P-P retrace total yoke yoke !22 total eddy frequency above amb. time voltage loss loss current & hys-
teresis loss
!5.75 kHz 17° c 12.2 psec 920 7.4 j.,r 4.1 w 3.3 w 31.5 kHz 36.6° c 6.05 usec 910 20 w 4.5 w 15.5 w 61 kHz 60° c 3.03 usec 880 41.5 j.] 5.44 w 36.06W
The line frequenc 15.75, 31.5 and 63kHz are related to the
NTSC system. Note that the corresponding line frequenc of the
PAL system (15.625, 31.25 and 62.5 kHz respectively) are approxi
mately the same.
The curves 1n figure 1.8 show that at the standard 15.75 kHz line
frequency the combined hysteresis and eddy current losses are
about equal to the losses in the windings. As the line fre-
quency is increased and the fly-back time is shortened, core losses
and eddy current losses in the copper become a large portion of
the total loss.
lOOr-----------------------~~~~~ Renace T ~ ~ }'
1';<: ·• Scan R.ate
u.n kHz
ll. S kHZ
12. l 118 1040 "'" ~ ;~;,;.:..: - .
6.1 "'' 26o "'"~Ti: :~~ ·~
f • 1.8a (RCA 804 Ferrite)
• IJ ... !1 I
ll 0 ... ... :t ... 111 111 ...
Q
t ~ ....
' . ' i ' ; i l i
1 10 lS
- -- I -1--1..:.. -- ;__ - 1-
.. ,. : ; ' ~ ; . :I·· i . ::1 , .. .... ; •zt· I ., ..
10 50 llO
Scanning R.ate - kllz
• 1 • 8b (Ferrite Haterial "A")
• ... ... !1
Cl 0 ... ... a. ... • • ...
Q
100 Scan R.He Rctrace T Yoke J:: •·····~·r-
----- ·-·- ....... t'l. n klh 12. 2
~· 1068 ~~~~~ 1-- f-·· f-
)1.) kHz 6.0) ~8 26) ~H ;- ·-·· -I- -
6] !<11; ].0) 6). 1 --· -~- - +-~-
~8 ~~~ 0 ... I·
l
1·, -- ---~ ·-· ... I· .
.. ~1- --·-•·-'----'--- -- --- -· --- .v l
0 vvoq V u - .......... : • V I;' I.;._;_.;. -;-:-- -- .,"' v V" . ..,. ... . ., ..
0 .. -.?vL .. "' . :. ' . ... ~ 1/t--~ o"- -- -~- ....
i I ' . ·y .-v ' . --~ .".
··----- --- -;: V ... "'I;' ~ . ·- i·· .
. . t/V ~~V~~ ' l :
• I J 41 •' 7 .... - 1-.
"'. . . I I f-., . . . f o" . ·--...
2
10
-. I .. :; : cl .--v ; ... ~7- ~--. f-":' l--\"
et- f-·· ·--. (.0~ ~ .. .
\1'-\..O,ses
I---'-- -~ ~- -- -· 1-- ·-:
) 1-1- f-f-
. ' ---- --- --· ~-· ~-- 1-- . . - ~- .. . I-. . .
2 --- --1-;-
' __ .........,. __ ... -· ~--- -- -- -- .,. . ·~-- ~-
l . .. ,.
I . I. i .. 1
10 l5 lJ 1 0
Scannlng R<He - IUiz
fig. 1.9c (Ferrite t·1aterial "B")
. 1.8, [4]: Deflection yoke lossesas a function of line frequency (horizontal scanning rate)
I N
l 2 3
-22-
5 lO 20
Figure 1.9 [4J: Deflection yoke lossesas a function of
fly-back (returne) time
Figure 1.9 shows a tynical curve of yoke dissi~ation as a function
of fly-back (retrace) time. At the very short fly-back time used
for the higher scanning rates, the losses increase rapidly. In the
conventional system not only the power dissi~ation in the yoke in
creases as frequency increases, but the nower dissipation in the
transistor and the diode of the circuit shown in fi~ure 1.6 will
also increase as frequency increases. These power losses have als0
-23-
been determined by Babcock and Hedam [ 4]. Table I. 2 glves the
result of these calculations.
Table 1.2, [~: Horizontal output transistor and
damper dissipation at different
line frequencies.
scannlng ra te 15.75 kHz 31.5 kHz 63 kHz
retrace time 12 ~sec 6 ~sec 3 ~sec (T ) r peak retrace 990 990 990 '10 1 ts (V )
s DC supply 128 123 128 volts (EDC)
Peak switch 3 A.rnn. 6 A.n:m 12 A.rn'. current
Peak base 1.2 Ar.!~. 2.4 Amp. 4.3 Amp. current
Transistor I. 4-4. 3H 7. 6-28. 7H 35-165\\T d. . . :t) lSSlpatlOn
Damper 0.4 ~v 0.83 H 1.8 ~v
dissiuation -· ------
:t)Lower value is for fall time Tf of 0.4 ~icroseconds. Higher value is for fall time of I microsecond.
The peak voltage across the deflection yoke lS glven by [4]:
where
V h,peak
fly-back time.
(I. I)
The peak voltage on the deflection switch and fly-back capacitor
is:
V S,peak
V C,peak
V S,neak
-24-
TIL I h h,peak
Tfb + EDC (I • 2)
As shown by equations (1.1) and (1.2) tQe peak voltage on the yoke
and on the deflection s1vitch is inversely pro~ortional to the fly
back time.
Therefore, for a given deflection yoke, if the line frequency is
doubled and the fly-back time is cut in half, the peak yoke volta
~e will double and the peak switch voltage will nearly double. If
these comnonents cannot withstand this high voltage or do not have
adequate safety factor, then yoke inductance must be decreased and
peak current must be increased to keep the peak voltage within the
desired limit and stil have sufficient scan. That is what Babcock
and Nedam did 1n their calculations and measurements: they kept the
fly-back voltage constant at all line frequencies. In this way the
high fly-back voltage contribute indirectly to the power losses in
the horizontal deflection circuit at the higher line frequencies.
It is obvious that, with increasing line frequency and hence decrea
sing fly-back time, the radiation will increase, since radiation is
mainly caused by the high frequency components due to the short fly
back time.
1.3 Symmetrie horizontal de~lection currents
If one of the symmetrie currents, shown in figure 1.10, is used,
then the disadvantages of the horizontal deflection system will get
smaller. During scan time the wavefarm of such a s~etric deflec
tion current should approximate the wavefaro of the conventional
asymmetrie deflection current, in order to get a constant snot ve
locity on the screen. Figure 1.10 shows two symmetrie current wave
farms. Figure I.IOa shows a triangular form, figure !.lOb shows a
sinusoidal form. The parts of the rising-part of the wavefarms in
dicated with "LR-scan" are approximately the same as the scan part
0
-25-
\ LR- SCël'l \ RL-scan over-scan over-scan
~ ~ / ~
t
..... 1,..... -\ LR-scan \ RL-scan
over-scan over-scan
Tl
Th
Fig. 1.10: Two symmetrie current wavefarms suitable for
horizontal deflection: A trangular wavefarm (a)
and a sinusoÏdal wavefarm (b).
of the wavefarm of the conventional horizontal deflection current.
If we want to use the trailing parts of these waveforms, too, then
the video signals displayed in the corresponding time interval have
to be turned around in time.
-26-
If we maintain the vertical deflection current used in the conven
tional receiver, then the scanning pattern on the viewing screen
changes from the one shown in figure 1.2 to the one shown in figure
1.11 in which the viewing screen is alternately scanned from left
to right and from right to left.
w
h ~
v
Fig. 1.11: Scanning raster produced by a conventional verti
cal deflection current and a symmetrie horizontal
deflection current.
The lines scanned from left to right and the lines scanned fro~
right to left should be aligned, that is to say: A straight vert
cal line in the original picture should stay straight in the pic
ture displayed on the viewing screen. If the nlace of the spot on
the screen is determined unambiguously by the current through the
deflection coil, then the rising and trailing parts of the deflec
tion current have to be absolutely symmetrie in order to get this
alignment. Note that with the conventional asymmetrie horizontal
deflection current this alignment is accom~lished much easier, be
cause small deviations fro~ the ideal current wavefarm will not
give disturbing geometrical deformation, since these deviations are
likely to be the same for all lines.
It is obvious that from the two symmetrie currents shown in figure
1.10 the sinusoidal current is preferable, because its first deri-
-27-
vative does not contain any JUmps at all, the rising and trai
ling parts of the sinusoidal current wavefarm are S-shaped just
like the conventional deflection current waveform, and, moreover,
a ser1es or parallel resonance circuit using an additional capaci
tor can be created such that the horizontal deflection amplifier
just has to deliver effective power. That is why the description
of a breadboard model using a sinusoidal horizontal deflection
current will be the main subject of this report.
-28-
2, THEORY
In par. 1.1 we assumed the deflection angle to be a linear function
of the deflection current. For the çlat screen shown in figure 1.5
we then demonstrated that the snot velocity is not a linear func
tion of the deflection angle, but that the deflection current
should have an S-shaned wavefo~ in order to achieve a constant
snot velocity on the screen.
Consicier a cathode-ray tube of a colour TV-receiver (figure 2.1).
x
--z 0
xz-plane
Fig. 2.1: Picture tube ~n the rece~ver
The tube has two planes of s:~etry. Let the z-ax~s of a Carte
sian coordinate system (x,y,z) be the intersection of these two
planes, the xz-nlane and yz-plane being as indicated in figure
2. I. Let the xy-plane touch the tube in point 0 the origin of the
system of axes.
We will determine the horizontal d~flecrion current ~n the ulane
of intersection of the screen with the xz-plane. We might also
determine the horizontal deflection current for an arbitrary
plane narallel to the xz-plane, but this appears to be nuch
more difficult. Xoreover, the calculated deflection current ~s
used for horizontal scanning for all lines in a field. Although
-29-
this will g1ve pin-cushion distartion of the picture 10 East-West
direction, as shown in :igure 2.13, this distartion may be correc
ted by amlJlitude modulatinf, the given horizontal de:!:lection cur
rent with a signal of :ield frequency l~ . In par. 2.1 the exact current wave~orm necessary fora constant
spot velocity, will be determined. This theore.tically and numme
rically determined S-shaped wavefarm will be compared with the
correspondin3 S-shaped part of the sinusoidal deflection current.
The part of the current of interest for this cornparison is called
the "active" part of the deflection current, corresponding to 'Ji
sible spot positions on the screen. If the active parts of the
theoretically deterrnined current waveforn and the sinusoidal cur
rent wavefarm used in nractice do nat coincide, then a geometrie
horizontal distartion o~ the ~icture will result. The magnitude
of this distartion appears to be rather small for the ,icture
tube used bere (A66-540X); this distartion can be minimised
further by signal processing, as we shall see in nar. 2.1. That
opens the way to a horizontal deflection system with a nerfect
ly sinusoidal deflection current. Thi~ sinusoidal current will
be generated by a resonance circuit. In ~ar. 2.2 the two prin
ciual types of resonance circuit will be reviewed, and the am
plitude rnodulating signal of field frequency, necessary for cor
rection of the pin-cushion distortion, will be considered quali
tatively. The influence of the Q-factor of this circuit will be
discussed in par. 2.3.
The lines "written" on the screen from left to right will nat be
aligned with the lines "written" on the screen frorn right to
left. ~o solve these alignrnent problerns, treated 1n ryar. 2.4, we
will only consider the active narts of the video signals; so the
blanking partsof the lines (see figure 1.4) are excluded.
By shifting the active parts of the video signal such, that they
coincide, in time, with the active uarts of the sinusoidal cur
rent, the alignrnent problems can be solved. Finally in nar. 2.5,
the consequences, which the implementation of a sinusoidal hori
zontal deflection current has on the scanning raster, will be con
sidered.
-30-
2.1 The effects of sinusciclal horizontal deflection currents on
the geometry of the picture
2.1.1. Introduetion
The desired deflection current wavefarm g~v~ng a constant spot ve
locity denencis on the place of the observer. An observer ~t a dis
tanee of 5 to 6 times the viewing screen hei~ht (h) is at the opti
mal position for looking at a screen [9]. However, it is very diffi
cult to determine the current in this case. Instead, the current
will be determined in two other c~ses: (I) ~Vhen the spot velocity ~s
constant on the curvature of the screen iffielf; (2) when the spot ve
locity is constant for an infinitely removed observer.
These two currents provide lower and u~ner bounds on the desired
current giving a geometrically correct picture for an observer at a
standard distance of 5 to 6 times the viewing screen height. For
the cathode-ray tubes used ~n modern TV-sets these two bounds will
appear to be so very close to each other, that the desired current
is determined for all practical purposes.
The vacuum inside the cathode-ray tube is very high. To limit the
resulting farces the screen has been curved. The intersectien of
the screen with the xz-plane is a niece of a circle at least for
the nieture tube considered here (A66-540X, see figure 2.2).
x l
xz-plane
Fig. 2.2: Intersectien of the picture tube (A66-540X) with the horizontal plane (xz).
-31-
The instantaneous horizontal deflection current ih 1
will de-, flect the electron-beam over an angle ~D
1, and the snot will ,
move from s0 ( ih = 0, t = 0) to S 1 • Seen from the screen the electron
bearn seerns to come from the ~oint ~~ on the z-axis known as the
centre of the deflection system. Let th~ distance from f~ to the
point s0 on the screen be ~· and let the screen have radius R8 and centre M
8.
Then we want to know the deflection current ih as a function of
~· R8 and t, giving a geometrically correct picture.
The current direction is assumed to be chosen such, that 1h 1s
positive if ~D is positive, and we then will only consider ~osi
tive values of ih. The negative values of 1h can then be deter
mined easily, since ih is an odd function 1n time t.
The phase and the frequency of a sinusciclal deflection current
are already determined, because the sinusciclal current has to be
0 for t = 0, with positive first derivative, and the frequency
is half the line frequency. Thus the sinusciclal current has to be
max1mum at t = ~ T1
, where T1
1s the oeriod of I line. The only
thing which can be changed 1s the arnnlitude of the sinusciclal cur
rent.
For a given spot velocity v, and given ~ and R8 , the horizontal
deflection current giving a geometrically correct picture is
completely detemined. Now let T be the time interval corresponding a with the active 9art of the deflection current and let w be the x width of the picture measured along the x-axis. Then for a given
picture tube the spot velocity v is determined such that during
T seconds the distance w is covered. So the deflection current, a x
which is theoretically determined by this spot velocity (and by
the given Ru and R8), will cause the spot to reach the extreme
right of the picture at t = !Ta. We now have to choose the am
plitude of the sinusciclal current such, that 1n the interval
0 < t < lT the sinusciclal current resembles this theoretically 2 a
deterrnined current as closely as possible. Here the amplitude of
the sînusoidal current will be chosen such that the ~agnitudes
of the sinusciclal current and the theoretically determined cur
rent are the same at t = ~T • a
-32-
Note that in this case the interval lT 2 a < t < !T1 will be used
for overscan.
Normally the spot velocity, which 1s determined by T , cannot be a
changed, because T has to be the same at transmitter and rece a
ver. However, in the breadboard model described here, the video
signals are A/D converted and stared into memory, so they can be
compressed or expanded in time easily, by changing the reacl-out
clock frequency of the memory. By campressing or expanding the
video signals, in fact (at the receiver) is changed. With
each new T a new spot velocity v is determined, and with each a
new snot velocity a new deflection current, giving a geometr
ly correct picture, is determined. This current 1n turn, will
determine a new amplitude for the ap?roximately sinusoirlal cur
rent. The distartion which is found with this improved T might a be less than the distartion which has been found with the stan-
dard
Note that for the case when T is decreased (compressing of the a video signals whereby the spot velocity is increased), a smal-
ler, more linear, part of the sinusoidal current is used; con
versely for the case when T is increased (expanding of the video a
signals whereby the spot velocity is decreased), a larger, more
curved, part of the sinusoirlal current used.
In this way, by changing T , and thus changing the snot velocity, a
the geometrie horizontal distartion mi~ht be minimised further.
2.1.2. ~~~~~~-~~-!~~-~~f!~~!!~~-~~g!~-~D
a. Lower bound (for observer at the screen).
Consicier a screen with radius R8 , and a deflection system D with
radius RD. We are searching for the angle ~D as a function of ~·
Rs and t when the spot-velocity on the curvature of the screen 1s
assumed constant, see figure 2.3! In this figure the following
equation holds:
-33-
with 6s = v 6t, v being the spot veloci~y. We find:
V or
ar ~n the limit 6t ~ 0:
V
Let ~S be a function of ~D' then:
-- = dt
ar with (2.1):
V 1 • d~
s d~D
( 2. 1)
(2.2)
(2.2)
Equation (2.2) gives a relation between ~D' R8, t and ~S
d~D Ta determine -at as an explicit function of ~D' ~ and R
5, we
shall first derive a relation between ~S and ~D. d~D
Now let ~ and R8 be fixed, then with -at known as an explicit
function of ~D' we are able tq determine ~D as a function of t,
numerically. Suppose, for instance, that ~D(t) is known at t 1,
then ~D(t 1 + 6t) with 6t + 0, is determined by:
d<bnj ~D(t 1 +6t) ~ ~D(t 1 ) +-at t=tl .öt, (2,3)
for suitably small. Now take ~D(O) = 0, then with (2.3) ~D
can be determined arbitrarily well for every t.
-34-
,Fig. 2.3: Deflection geofletry for a spherical screen
In triangle MsMUs 1 the following equation holds:
(2.4)
In triangle MbS)S 1 the following equations hold:
-35-
or (2.5)
With (2.4) and (2.5) we find:
2 Omitting the subscript "I" and deviding by RS g1ves for an arbi-
trary point
Defining p
Defining
or
. 2'"' s1n '~'s
. 2'"' S1n '~'D
~2 ~ I + (I - --) - 2(1 - --)costjl
Rs Rs s
(',~ =- the
Rs last equation beco~es after squar1ng
. 2'"' s1n '~'s
{ 2 sin tjJD
2 - I - (1-p) }
2
. 2'"' (', . 2 (', ( 2) (', 2 s1n '~'D p, s1n <Ps q, and 1-p u ,
2 2 (3. - I - \! )
p
2 4 \) ( 1-q)'
2 2 2 2 2 2 2 q - 2q(p + \! p - 2\! p ) + (1-v ) p o.
The roots of the last equation are:
(2.6)
q:. = p[(I+p2) - 2}p:_ 2\!v }p2
+ (I+})p + ;] (2.7)
Using the definitions for p and q
sin2
rps = sin2
<t>D[(l-+U2)- 2U2 ~dn2rpD + 2uVu2
sin4
rpD-(I+U2)sin
2rpD+;]
(2.8)
t.Je shall restriet ourselves to configurations as shown 1n figure
2. 3, in which ~ ~ RS < co, or 0 < p ~ I •
-36-
If we do not consider the configuration with p = I, then it can be !::. shown, that the factor between brackets 1n (2.8), U = V ~ IV, 1s
always positive. Then, sirree sin~S must be positive if sin~D 1s
positive, only 2 of the 4 solutions for sin~S can represent the
configuration in figure 2.3, so:
sin~s = sin~D[(l+1})- 2u2 sin2~D + 2uVu2 sin4~D-l(l+u2 )sin2~D + 1]
(2.9)
Note that equation (2.9) also includes the solutions of sin~S for
P = I (u=O), s1nce sin~S = sin~D for p = I in the configuration of
figure 2.3.
Differentiation of (2.9) with respect to ~D g1ves:
d~s cos~s -d,t,
'~'D
I • ,t, 1 2 S1n'I'D
cos~D [U] 2
+
x d: [u J D
[up (2.10)
1T For symmetry reasans we may restriet ourselves to 0 ~ ~S < 2, so with (2.9) we find:
Now, wi~h (2.10), we find:
[I -sin2
n[u]J 2 ruJ 2
d~s~ with (2.11) we find two possible values for ---d~D ~ = 0
D
[v + wl !J<lb 0 I + u 2 - p
d~s [TIJ!\ I --- =
w] 2
1<lb d~D
·For 0 < p < I
configuration
(jJ =0 D [v - 0 I - u = p
d~s only the latter value will do, sirree --- < I,
d~D of figure 2.3.
1n the
-37-
So finally we find: l l • <P d
[ V-~<71 d<Ps co s<PD [ v-W] 2 2s~n D
d<j)D
d<j)D 2 -[I-sin <PD[v-wJ]
2
+
[I-sin2
<PD[V-ivJ] 2
[v-w] 2
with V~ (I+J2
) - 2v2
sin<j)D
w~ 2 v 2 . 4 c 2) . 2 v v s~n <P - !+V s~n <P D D
_d_[v-w] dcjJD
+ 1
(2.12)
\.!ith (2.2), (2.3) and (2.13) cjJD can be determined numerically as a
function of t.
Noting that if t
cjJD givin by:
dqJD
dt t=O
V l =
0 én cp5 = O, the first derivative of
V I v Rs V = --= --
Rs p Rs ~ ~ d~s~ dcjJD cjJD=O
~ vt will vary if v or RD ~s varied, the dimensconless quantity t
has been introduced.
Hith t we find:
dqJD
Hence, s~nce
dcjJD = dt
dtf!s
cjJ =0 D
dt -= dt
p,
V ~/v
..J?_ (2.13) . dij)= dcjls s
dcjJD dtf!D
Using t, all curves cpD can be olotted in the same figure with the
same resolution. t, may be considered a normalised time variable,
since t can be~egarded as: The ratio between the time variable t
and the time -- , which the spot, moving with a constant velocity V .
v, needs to cover the distance RD. t, however, may also be consi-
dered a normalized distance,since t can be regarcled as: The ratio
between the distance vt, which the spot when moving with a con-
-3~-
stant velocity v, has covered during ti~e t, and the distance ~·
Since vt = s (see figure 2.3), the normalised distance T is also
g1ven by:
T = vt s (2.14)
Usinz (2.12) and (2,13), ~D(T) can he calculated numerically 1n
the sa~e way as ~D(t), see (2.3), with:
(2.15)
Figure 2.4 shows the results of these calculations for the norma-
lized curvatures p l,p=.S,p .1, p = .01 and p -+0 (ATN (TAU)).
Fl
1.0
0.5
0 0
/ /
0.5 1.0 1.5
Fig. 2.4: Lower bound on the deflection angle ~D(T)(FI), for
different normalised screen curvatures, p.
-39-
The extremes for p = I and p ~ 0 can be determined inderyendently
of (2.12), (2.13) and (2.15), and can therefore be used to con
trol the calculations. For p = I that ~s when RS = ~' ~D(T) will
be a linear function, the spot following a circular are with con
stant speed.
For p ~ 0 (that ~s if RS ~ oo) ~D can be determined with figure
2.5.
s=vt
Fig. 2.5: Deflection geo~etry for a flat screen.
In fi8ure 2.5 the following equation holds:
vt
or vt arctan (~) = arctan T (2.16)
The results for these extreme values of the normalized curvature
p found in this way match well with the results found by numeri
cal calculation and thus give confidence in the general results.
In this case we do not want the spot-velocity on the screen to be
constant but rather the projection of the spot velocity on the
x-axis. This situation is shown in fi3ure 2,6,
-4U-
Fig. 2.6: Deflection geometry for a curved screen viewed
by a distant observer.
Let this constant spot-velocity in the x direction be v, then the
following equations hold, see fig. 2.6:
or in the limit ~t + 0:
(2.17)
-41-
dcps Since dt
dcpo we find with (2.17):
t
V
With (2.10) we find:
with the same V, \v as in (2.12).
(2.18)
(2.19)
Figure 2.7 shows the results of the numerical calculations of cpD
for this case with p = I, p = .5, p =.I and p = .OI.
Fl
1.0
0.5
t
0 0
------------F I, RO= 1-:::----~-------,' ____ fi, RO=. ___ fi, ROz. __ FI. RO=.
I
/
/
/ ~ / ~
/ </ ;/,/
/ij //#'
// ;_/
0.5
/
/ /
/
/ /
/ / /
l.O 1.5
Fig. 2.7: Upper bound on the deflection angle~(T) (FI), for different normalised screen curvatures, p.
-42-
Note that now vt =x (see figure 2.6), so if T ~s considered to be
a normalised distance rather than a normalised time, T is also
g~ven by:
vt x (2.20) 1
2.1.3 Bounds on the deflection current
If the relation between the current ~h and the angle ~D ~s known
then ih can be determined by:
(2.21)
The basic relation between ih and ~D ~s g~ven by Donald G.Fink [8]:
or
. 1 ,r nh ~h = , ;---;;:--' V E
jJ V e/2m
~h
c
0
D a . ,+,
l s~n'VD' f
(2.22)
where ih/C is a dimensionless quantity. The normalization factor C
is given by:
with
nh number of windings of the horizontal deflection coil
E electrical field strength in the cathode-ray-tube
the effective diameter of the coil
the effective length of the coil
space permeability) -9 l-l 4n.IO henry/m (free \~ 5 - 1 -1 VeiLID = 3.10 mV 2 sec
With (2.21) and (2.22) one finds:
-43-
This last equation with (2.13) becomes the lower bound:
pcoscfJD
dcps
d<tJD
and with (2.18) the upper bound:
pcos<PD
d<jJS coscp 5 -d'"'
"'D
(2.23)
(2.24)
Figure 2.8 shows the results, for the normalised curvatures p .5
and p
1/C
1.0
0.5
0
0
.2. Note that for p = .2, which almost equal to the P
0.5 1.0 1.5
pper-twnd
Fig. 2.8: Bounds on the normalised horizontal deflection current
ih/C (I/C), for different normalised screen curvatures, p.
-44-
of the picture tube (A66 - 540X) used in the breadboard model
described here, there is hardly any difference between the up
per and lower bound on ih/C.
If (2.22) is differentiated with respect toT, we find:
dih d<jlD
dT c . dT cos<jlD (2.25)
For T 0 (2.25) becornes:
dih d<jlD cos<jlDI C- .
dT T=O dT T=0 T=0 c, (2.26)
in this <Pn = o, and d<jlD
s1nce case dT T=0 I.
For the case when the lower bound on ih/C is calculated, and the
nomalised distance T s C can be deterrnined by (2.26) - RD' as:
dihl dih dx~
dih (2.27) c TT T=O = ( dx ' dT ~ ds T=O s=O
For the case when the lower bound on ih/C is calculated, but the
norrnalised distance T x
Ru' C can be deterrnined in a sirnilar way
c dihl ~ dx x=O·
(2.28)
Using equations (2.27) or (2.28) as appropriate C can be deter
rnined by rneasurernent of the deflection sensitivity coëfficient
dihl dih I ds (or dx ), by rneasuring s (or x) as a function of 1h s=O x=O
for a known ~·
The upper bound on the horizontal deflection current 1s deter
rnined cornpletely by a given spot velocity v, and given ~ and
R5
(see par. 2,13).
.. 0 .. ... • • c .c u • .. .. ...
3
0
~ .. >
i5 t.J z ... a
3 a
:e ... .a ... f .; z ... a .. ...
-45-
The spot velocity v ~s determined by w and T : x a
V
w x T
a (2.29)
With the normalised deflection currents
~ and RS are implied in the normalised
implied in the normalised time T(=~).
calculated in par. 2.1.3,
curvature p(=:U), and v is s
A normalised sinusoidal deflection current can be written as
A sin~T • n
The normalised frequency ~s g~ven by:
~ Tr /2 Tr ,-;:::-zTl Tl
s~nce this current bas to be max~mum
tl v Tl
~
Now let
(2.30)
for T
(2.31)
(2.32)
then s~nce Ta= T1-a, where a ~s the line blanking time (see fi
gure 2.4):
If the magnitude of the theoretically determined current is
(ih/C)E for T = ha' then the amplitude An of the normalised
current is given by:
with (2.30) and (3.32) we will find for A : n
A n sin(p .n/2) a
(2.33)
-46-
Figure 2.9 shows ~D and (ih/C) for the normalised curvature
o = .1, both in the upper-bound case. With 1 = 2.34 we find: a
A n
.855 1.09
This normalised sinusoidal current is also included 1n figure
2. 9.
l/C OR FI
1.0
<Pu
( ih/ c )E t -- --
0.5
I
0 ...J<.--;1--l-+··1 I I I I I I I I ++-+-t--+-t I I I -I I' I I I I I j I b 0.5 1.0 'Ta 1.5
TRU (=V*TIRDJ
Fig. 2.9: Normalised sinusoidal horizontal deflection current,
for O= .1
Suppose now that T 1s reduced by a factor k, a
T I a
T /k a (2.34)
-47-
This means that we have to lncrease the spot velocity by the
same factor k:
v' kv
Then also T changes,into:
T' v't kvt
kT RD ~
Thus T 1 ' will be:
v'T ' 1
kT1 Tl RD
Ho wever T does not change, Slnce a
T v'T ' kv 2:.
' a k T
~ RD T a a
With (2.32), (2.37) and (2.38) pa' can be determined by:
p ' a
(2.35)
(2.36)
(2.37)
(2.38)
(2. 39)
In agreement with the change in T • If the modified normalised a
current is given by A 'sin~'T' then: n
A' n sin( p 1T /2k)
a (2.40)
(2.41)
Figure 2,10 shows the upper bounds for ~D and ih/C if p = .1,
(same ~D and ih/C as in figure 2,9), tagether with a few exam
ples of normalised sinusoÏdal currents.
Figures 2.9 and 2.10 demonstrate clearly the usefulness of the
sinusoÏdal current as a deflection current.
-48-
l!C OR F l
l.O
<~>o,E
0.5
l I
0 I
-f--1--+-t-+-1-+-+-+-l-+---+--+ +++--++--+- + +-+ --t----+':-1 I I I I I + l.O 1'a 1.5 D 0.5
Fig. 2.10: Normal sinusoirlal horizontal deflection currents,
for p = ,1, at different spot velocities.
2.2 Generation of the sinusoÏdal current
It is obvious that the sinusoirlal current can be generated with
the help of a resonance circuit. If the horizontal deflection
coil were to be driven directly by a source, that souree also
would have to exchange the reactive power with the coil. If a
resonance circuit is used, then the souree only has to deliver
the power dissipated in the coil resistance.
-49-
There are two principal types of resonance circuits that can be
used: The seriesresonance circuit and the parallel resonance
circuit.
2.2.1 The series resonance circuit
The series resonance circuit is shown ~n figure 2.11.
+
V
Fig. 2.11: Serie resonance circuit.
For figure 2.11 the following equation holds:
or V(jw) lh (jw)
v(jw)
. ( I Rh + J WL - -) h wc .
a
We want this circuit to be in resonance for w
(2.42) has to be real. Thus
c a T • 2f 2 -h'+TT h
w n
(2.42)
2TTlh' where
(2.43)
-51-
For figure 2.12 the following equation holds:
\(jw) V(jw) V(jw)
Ic(jw) + Ih(jw) = + . I Rh + JWLh
or It(jw)
V(jui)
jwc a
We want this circuit to be in resonance for w
(2.46) has to be real~ Then
c = a R 2
h
2 Normally Rh
2 2 << w Lh
c = a
That is the same value we found ~n (2.43).
(2.46)
(2.47)
The relation between the souree current ~s and the current ~h will
be found with:
I s
V
l.Jîth
IR + IC Ih V . V
+ =- + JWC V R. a Rh + jwLh ~
I s
I -+ jwC + R. a Rh + jwLh ~
V ~--~~--- the last equation changes ~n: ~ + jwLh
I s
( R ~ j wC) (Rh+ j wLh) + I ~
I s
or
..
(2.48)
-52-
So if l_ Assinwht, then s
l_h = Ahsin(wht-8) (2.49)
A Ah
s =
with
2.2.3 B~~~~~l-~~-Ei~:~~~~i~~-~ff~~~~-g~~li~~~iY~-~~~~i~~E~~i~~-~=
~~-~~Eli!~~~-P2~~~l~!i~g-~i~~~l
The souree signals v and i of the described resonance circuits s s .
should be amplitude modulated in order to correct for the pin-
cushion distortion. This amplitude modulation can be realised on
small signal level. The amulitude modulated small signal then
simply is fed to a power amplifier, which directly excites the
resonance circuit.
Note that with the conventional deflection circuit (described in
par. 1.1) the amplitude modulation has to be clone on large signal
level.
The modulating signal can be detemined qualitatively with the help
of figure 2.13 showinga picture with pin-cushion distortion in the
horizontal direction. This picture 1s s~etric with respect to the
x-asis, the modulating signal m(y) 1s determined qualitatively by
the even-order Taylor series in the space domain.
m (y) 2
= ao + a2y
-5.3-
y
Fig. 2.13: A picture with pin-cushion distortien in the hori
zontal direction.
Here v is the spot velocity in the direction of the y-axis, and y
Tf the period of one field.
With y = v t the modulating signal m(t) can be written as: y
m(t) . •. . . . . , (2.50)
1n the time domain. Since the latter function is periodic with
the modulating signal m(t) can also be written as a Fourier
series:
where
m(t) b0 + b 1coswft + b2cos2wft + b3cos3wft + •••• ,
(2.51)
for all t,
2îT w =·
f
With the rnodulating signal m(t) given by (2.50) or (2.51) the
sinusoidal amplitude modulated horizontal deflection current is
given by:
-54-
(2. 52)
Note that equation (2.52) holds for -~Tf < t < !Tf if (2.50)
is used, and for all t if (2,51) is used.
2.3 Deflection current modulation due to cros
between the deflection circuits
Figure 2.4 shows the deflection yoke with the ring of ferronag
netic material. Inside this ring one fincis the yokes for hori-
Fig. 2.14: Deflection yoke with ring of ferromag
netic material.
zontal as well as for vertical deflection. Although the magnetic
fields for horizontal deflection ~ and for vertical deflection
H are orthogonal they influence each ether via the ring of ferrov
magnetic material. The ring serves as a ~agnetic short-circuit for
both fields.
Consicier the magnetic circuit for the horizontal deflection. The
magnetic souree ih.nh sees two magnetic resistances R . and R t' m, ~ m, The farmer the magnet resistance inside the shield; the lat~
ter is the magnetic resistance of the ring itself.
Although the latter is much smaller than the farmer, its variatien
may cause trouble. This variatien in R is due to the variable m,r
field H caused by the vertical sawtooth current ~ . This variable V V
field H saturates the shield thus enlarging R . If R becomes V m, r m, r
-ss-
larger than the flux ~h for the horizontal deflection becomes
smaller. And with ~h' Lh becomes smaller.
Although this variatien in Lh is rather small, it will cause a
noticeable variatien in the phase e between the exciting souree
signal and the current ih of the resonance circuits, described
1n par. 2.3. The impact of this variatien of Lh on the phase 8
will be described below for both the series resonance circuit
and the parallel resonance circuit.
2.3.1. Variatien of 8 in the series resonance case -------------------------------------------The relation between the souree voltage v
5 and the current 1h 1s
given by (2.4S).
Let the variatien on Lh o.ue to variatien 1n 1 be óL , and let v m L h,m be the self indoetanee of the horizontal deflection coil for
i o, so that: V
Lh = L - L.lL h,m m (2.53)
with 0 < óL < óL m m,max
óL = 0 for i 0 m V
óL = L for l = + l m m,max V - v,max
2 In practice resonance will not be attained exactly, so Wh LhC
will be somewhat smaller or somewhat larger than I.
Let L = L + óL h,m res res
(2.54)
with L ~ -2-res wh ca
Then with (2.53) and (2.54) we find:
L + 6L - 6L (2.55) res res m
-56-
T.Jith (2.55) '\ and 8 from (2.45) will change into:
8
A s 2 , A ) 2
+ Wh luL -61 res m
wh(6L -61 ) res m arctan( R )_
t
(2.56a)
(2.56b)
In case of resonance 61 and 61 will bath be small. Suooose: res m --
and
lw(6L -61 )I<< R res m t }2.57)
I6L I has the same order of magnitude asi 61 I m res
Then Ah and 8 of (2.56) change into:
8
A s
Rt
wh(6L -61 ) res m
(2.58a)
(2.58b)
(2.58) show that the assumed variatien of Lh due to variatien 1n
iv' with the supposition of (2.57), will have no impact on Ah but
surely some on 8.
Since 61 > 0, see (2.53), 8 always becomes smaller if m
comes larger; 8 of course remains close to 0.
2.3.2.Variation of 8 in the narallel resonance case ----------------------~----------------------
li I be-v
The relation between the souree current 1 and the current 1h s 1S given by (2.49). With R. -+oo (2.49) changes 1n: 1
if 1 A sinwht s s
then 1h '\ sin(wht-8)
with
A s
-57-
1 + l\.1 r res l\.1m those values of Ah and 8 change into:
8
Now suppose:
arctan
A s
jw(l\.1 -l\.1 )[<< w~ C m res -ba
(2.59a)
(2.59b)
(2.60)
and jl\.1mj has the sameorder of magnitude as jll.1resl
Then Ah and 8 of (2.59) change into:
8
A s (2.6Ia)
(2.6Ib)
The variation of 1h due to variation of iv' with the supposition
of 2.60), will have no impact on Ah but certainly has impact on 8.
Here, too, 8 becomes smallerif ji I becomes larger; 8. V
Although the changes in 8 are small, in both cases, they are
noticeable. Especially in a system, which uses a symmetrie deflec
tion current, as we shall see in par. 2.4.
-ss-
Here no choice is made between the two types of resonance
circuits. Although in the hardware realisation described in
chapter 3 a ser~es resonance circuit is chosen, this choice is
made for practical reasons. In chapter 4 th
discussed.
choice will be
2.4. Alignment of the lines scanned 1n opposite directions
If a symmetrie horizontal deflection current is used the R, G,
B signals, containing the picture information, have to be re
arrangend in time, because the signal coming from the trans~it-
ter corresponds to a scanning raster which
asymmetrie horizontal deflection current.
produced by an
the new types of
1 , necessary for a raster produced by a symmetrie horizontal
deflection current, will be defined.
R G B : The re-arranged vers1ons of R, G resp. B used sym' sym' sym
LR line
RL line
Line duet
LR scan
for scanning with a symmetrie horizontal de
flection current. See figure 2.15 c and d.
The piece of the video signal(R , G , B ) sym sym sym which will be used to reproduce the picture
with the reproducing spot rnaving from ~eft to
~ight on the screen, see figure 2.15d.
The p of the video signal which will be
used to reproduce the picture with the repro
ducing spot mov1ng from ~ight to Left on the
screen, see figure 2.15d.
A sequence of one LR line and one kL line in
cluding the overscan parts, see figure 2.15d.
The scanning piece on the visible part of the
screen with the spot rnaving from Left to ~ight.
See figure 2.15e.
RL scan
-59-
The scanning piece on the visible part of the
screen with the spot rnaving from ~ight to Left.
See figure 2.15e.
If the scann1ng raster 1s produced by an asymmetrie horizontal
deflection current, the lines are written from left to right.
The scanning raster of the receiver described here is produced by
a symmetrie horizontal deflection current and this raster needs a
video signal consisting out of line duets.
The video information of every other 1 ine has to be reversed. \ifl1ich
of the lines of the incoming signal will become an LR line, and
which an RL line, will be determined with the "ST" signal:
ST signal The logic signal at half the line frequency
deciding whether the line of the incoming video
signal will become an LR line or an RL line.
If ST is "low", then the line of the incoming
signal will become an LR line; if ST is "high",
it will become an RL line. Fig. 2.14b shows the
relation between the ST signal and the line syn
chronisation signal (the latter signal has al
ready been shown in figure 1.4).
The resulting video information will be delayed by one line due to
the turning operation so the LR line coincides, ~n time, with the
"high" part of the ST signal, and the RL line with the "low"
part of the ST signal. In figure 2.15 this has been shown.
On the rece1ver screen the active parts of the LR lines have to be
aligned with the active parts of the RL lines, 1n order that a
straight vertical line in the original picture rema1ns straight on
the receiver screen and doesn't become rippled.
There are three ways to align these active parts of the LR and RL
1 ines:
-60-
2H
JI---L.---1 ------!.....1 J__l ~. t (a)
ST
(b)
R,G,orB
(c)
(d)
0
LR-line RL -irne
I i neduet ih
0 (e)_
LR-scan \ RL-scan over-scan OYer-scan
Tl Tl
Th
Fig. 2.15: The line synchronisation signal 2H(a); the ST signal
(b); the incoming video signal (R, Gor B)(c); the
resulting video signal (R , G orB ) (d); the sym sym sym horizontal deflection current (ih) (e).
-61-
( 1 ) By changing the current ih;
(2) by changing the spot position with the aid of 'n auxiliary
coil;
(3) by shifting the active parts of the video signals ~n time.
If the current ih is realised with the help of a resonance circuit,
then it will be very difficult to use methad 1. Methad 2 is possi
ble, but requiers more electrical power than methad 3, and more
over it might disturb the convergence and the purity of the three
electron-beams.
Methad 3 appears to be the one methad to use. That ~s why four
ways of alignment will be described here which all use shifting of
the active parts of the video signal.
These alignment methods will be described with the aid of figure
2. 16. This figure shows two periods of the line sync signal, one
period of the ST signal, a corresponding line duet and one period
of the sinusoÏdal current ih. The LR scan takes place from tLR,s
till tLR,e and the RL-scan from tRL,s till tRL,e" These four va
lues of t deterrnine the active parts of the horizontal deflection
current. If the spot is in the extreme left and the LR scan is
going to take place then t = tLR,s" The other values are deterrnined
in similar ways. The time differences tv0 , tv 1, tv2 and tv3 are
related to the starting and stopping moments of the active parts
of the LR line and the RL line. The active part of the LR line
starts tv0 seconds after the rising edge of the ST signal and it
ends tv0 + tv 1 seconds after the rising edge of the ST signal.
The other values tv2 and tv3 indicate the start and the end of
the active part of the RL line.
Consider the four ways of alignrnent described ~n the following:
Linear alignment: The alignment ~s said to be linear if the re
sulting video signal corresuonds to a linear
function of the incoming videosignal. If this
~s not the case then the alignment is said to
be non-linear.
Static alignment:
Line-duet static
alignment:
Line-duet dynamic
alignment:
ST
c><.l
\. tv
0 I
-62-
The alignment is said to be static if the time
differences tv0
, tv 1 , tv2 and tv3 (see figure
2.16) are the same for all line duets.
The alignment is said to be line-duet static if
the time differences tv1
, tv2 and tv3 are the
same for all line duets and tv0 is not the same
for one or more line duets.
The alignment lS said to be line-duet dynamic
if one of the time differences tv1
, tv2 or tv3 1s not the same for one or more line duets.
[ • I (a)
tv1 I tvz I" tv3 I
I R ,G ar Î sym s,m
I :~ ~: B sym
I I (b) I •t
ih r (c)
· 2 16 Th ST · 1 (a), one corresponding line duet of the re-Flg. • : e s1gna
sulting video signals (R G orB ) (b), and one s)llp.' sym sym
corresponding period of the sinuscaidal horizontal deflec-
tion current (c).
-63-
~~ich of these alignment methods can be used depends upon the
quality of the sinusoidal current ih.
Assume that the current ih satisfies the following conditions:
(a) For a eertaio value i of the vertical deflection the place of V
the spot on the screen is determined unambiguously by the in-
stantaneous value of the sinusoidal current ih through the ho
rizontal coil.
(b) The amolitude of the sinusoidal current is varied such, that
the pin-cushion distortion (in horizontal direction) is correc
ted. However, this variatien of the amnlitude is so small that,
for practical purposes, it can be considered constant, within
one line duet.
(c) The phase difference a 1 (see figure 2.16) between the rising
edge of the ST signal and thaifollowing zero-crossing of the
sinusoidal current ih, with d~ positive, is the same for all
lineduets; with a1
~ 100°.
Now suppose an alignment mechanism, which syncronised to the
positive edge of the ST signal, and which determines when and
with which rate the active parts of the LR and RL line are to
be written on the screen. Then, with conditions a, b and c ful
filled, the intervals of writing on the screen are completely
determined by values of tv0
, tv1
, tv2
and tv3 which are the
same for all line duets. Hence, with conditions a, b and c ful
filled, static alignment can be applied.
But if a resonance circuit used to drive the horizontal coil
then condition c might not be fulfilled anymore (see par. 2.3
and 2.4). It is obvious that a souree signal s , s being v or s s s is, can be made such that the phase difference, ao, between the
ST signal and ss is constant for all lines. (With a 0 being the
phase-difference between the rising edge of the ST-signal and
the following zero-crossing of the sinusoÏdal signal ss' with ds s dt positive).
Now let us for example take a series resonance circuit with souree
signal v . Then, if the supposition of (2.57) is fulfilled, s
-64-
L al = ao + e ce has been discussed in par. 2.4) will change if the
vertical current, iv, changes. If [ivf becomes larger than a1
will become smaller, and ih will be shifted backwarcis in time with
respect to the rising edge of the ST signal. If [i [ becomes smal-v
ler then ih will be shifted forwards 1n time with respect to the
rising edge of the ST signal. ~ig. 2.17 shows ih for two values
I I ,.
I =I / v v.max ---/
/ /
Fig. 2.17: One peri'od of the sinusoidal horizontal current (ih)
for two different values of [i [: [i [ = 1 and v v v,max [i I = o
V
i • So, if a series re-v,max
sananee circuit is used to drive the coil for horlzontal deflec-
tion, and if supposition (2.57) is fulfilled then a 1 will not be
the same for all line duets and the application of static align
ment will give distortion. During the LR scan the instantaneous
value of ih will be larger for the case \vhen [i [ = 1 v v,max than
for the case when [i [ = 0; V
value of ih will be smaller
for the case when [i [ = 0. V
during the RL scan the instantaneous
for the case when [i [ = 1 v v,max than
Figure 2.18 shows this kind of dis-
tortion; the diverging (vertical) lines should remain together,
but that is only true in the centre.
Suppose that condition (c) is not fulfilled and static linear
alignment is applied: This gives the distartion just described.
Then replace condition c by condition
(c') The phase-difference a1
between the ST-signal and ih is not
the same for all lineduets but its variatien is so small
that, for practical purposes, it can be considered constant
within one line duet.
-65-
I I I I . I \i ! I i
'I . I l j
~ I 11 ,' I I i
!-·
i\
I ~~ 1\ 11 11
• 2.18: Distartion due toa change in phase in the
sinusoÏdal horizontal deflection circuit, (a):
undistorted grid, and (b): Distorted grid.
ST
0
I I
I l 1h ' I
t : I 1-:-:--1 I
I
0 I I
!
I
!\
(a)
(b)
(c)
Fig. 2.19: Line-duet static linear align~ent: The resulting
video signals (b) are shifted in time, with res
pect to the ST signal (a), in the same way as the
horizontal deflection current (c).
-66-
t.Jith the conditions (a), (b) en (c') fulf led line-duet stat
alignment can be applied. Fig. 2.19 shows the result of that kind
of alignment for the lineduets with I i I i and I i I = 0. v v,max v How do we de termine tv , which will be different for each 1 ine-
o duet? Within one field, tv0 will be different for each lineduet,
but tv0 will be the same for the corresponding line duets in the
corresponding fields following, since tv0 depends on iv. So the
value tv0 will return periodically. Store these values and let
the line duet number within one period determine the appropr
value of tv0 to be used. Another way of determing tv0 is by de
terming the phase difference a1
• This can Be clone by measuring
the current ih. Figure 2.20 shows a method to measure ih. In the
s resonance circuit used to drive Lh' the horizontal coil,
an extra resistor is inserted. The voltage over this resistor is
proportional to 1h.
F • 2.20: Series resonance circuit including a measuring
resistance Rhm·
Let the sinusoidal current be max1mum for t t , then with ma x the conditions (a), (b) and (c ') fulfilled (and of cour' se also
with (a), (b) and (c) fulfilled), the sinusciclal current, for
practical purposes, can be considered symmetrie with respect
to the axis t = t , within each line duet. That is why, to ma x achieve alignment, tv
1, tv2 and tv3 can betaken the same for
all line duets. If the sinusciclal current were no longer suf
ficiently symmetrie, then line-duet dynamic alignment would
-67-
have to be applied.
Non-linear alignment might be used ~n combination with the above
kinds of alignment to reduce small geometrical distortions in the
horizontal direction.
In the hardware realisation described ~n chapter 3 line-duet sta
tic linear alignment is applied. This choice will be discussed in
chapter 4.
2.5 The effects of the symmetrie deflection current on the scann~ng
Figure 2.22a shows the type of scanning raster which is used in
almast every TV-set at present. For convenience ~n every
frame only 25 lines have been taken; each field thus having 12.5
lines.
(a)
(b)
(c)
Fig. 2.21: The scanning raster (a) produced by two asymmetrie
sawtooth currents, (b) for the vertic al and (c) for
the horizontal deflection.
-68-
For simplicity the time between the A- and B-field has been
taken zero; whereas the time between the B-field from one frame
and the A-field from the next frame is supposed to coÏncidence
with the fly-back of the 25th line.
The deflection currents are shown in 2.2lb and c. These currents
are asymmetrie both for horizontal and vertical scanning, and
will be supposed to be linear. In practice the latter is not
true, as we saw in par. 2.1, but that is of no concern when des
crihing the effects on the scanning raster.
If the horizontal deflection current is changed from asymmetrie
to symmetrie, the scanning raster will change to the one shown
1n figure 2.22a. Figure 2.22c shows the horizontal current as a
function of time for this situation.
-= F-=-
f:::-·
F==--·-=
~:::=--1----
---= f=·
---= F=-:"
_____-\t c======--= r
Vr À À À À À À À À À À À ;, V~ ~ V~~ V'V'V
(a)
(b)
(c)
Fig. 2.22: The scann1ng raster (a) produced by a sawtooth current
(b) for vertical deflection, and a symmetrie triangu
lar current (c) for horizontal deflection.
-69-
If figure 2.22 is cornpared with 2.21 the following things will
attract attention:
In figure 2.22a the active line segrnents of the A- and B-field
interseet and that is sarnething that doesn't happen in the scan
ning-raster of figure 2,2la.
The scanning raster shown in figure 2.22a has "holes" in the left
and right sides of the raster, whereas the active line segments
are hornogeneously spread over the screen in figure 2,2la,
If we campare figure 2.21b with 2,22b then we notice that the
maximurn value of the current ih for horizontal deflection in the
symmetrie case is larger than in the asymmetrie case. The diffe
rence ~s about 6%. This is because, in the symmetrie case, the
fly-back time is "translated" ~n over-scan time and therefore the
current must be made larger.
The only way to get rid of the "holes", while at the sarne time
preventing intersections appears, to be re-arranging the lines such
that the lines of the A- and B-field are parallel. In the case of
symmetrical scanning the only possible way to manage this is by
changing the vertical sawtooth current into a "staircase" current,
such as the one shown in figure 2,23b. The horizontal deflection
current is the sarne as the one shown in figure 2.22b, Figure 2.23a
shows the resulting scanning raster.
The staircase vertical deflection current ~ can be realised by V
superirnposing the "norrnal" sawtooth current i , on a sawtooth curv
rent at line frequency in the vertical deflection coil L . V
This has been shown in diagram in figure 2.24.
Another way to achieve the wanted staircase deflection ~n the ver
tical direction is by superirnposing a "norrnal" sawtooth rnagnetic
field H and a sawtooth rnagnetic V
in the sarne direction. The field
field, H , of line frequency, y,a H should be set up with the
V
"norrnal" sawtooth current through L , and the field H v v,a with a
sawtooth current of line-frequency through an auxiliary coil L v,a
-70-
(a)
r~.m~~ ~! ~ ~ l(b)
fr 1\ A A /\ A 1\ A 1\ 1\ 1\ 1\ G VVV V VVV V\fVVV (c)_
Fig. 2.23: The scann1ng raster (a) produced by a staircase
current (b) for vertical deflection, and a symme
trie triangular current (c) for horizontal deflec
tion.
b4 Fig. 2.24: Staircase current waveform realised by superimposing
two sawtooth current waveforms.
-71-
3. HARDWARE REALISATION
~. 1. Design of a breadboard model for sinusciclal horizontal
deflection currents
In par. 2.4 the lines of the resulting video signals (R , sym
G and B ) were divided into two types: LR lines and RL lines. sym sym Now we shall also devide the lines of the incoming video signals
(R, G and B) into two types, by defining:
PLR line: A line of the incoming video signals which 1s Predesti
nated to become an LR line.
PRL line: A line of the incoming video signals which 1s Predesti-
nated to become an RL line.
The line types of the incoming and outcoming video signals are
determined by the ST signal, see par. 2.4.
Within each line a further subdivision will be made into the
active part and the blanking part. The active part has already
been defined in chapter 2; the blanking part is the complement
of the active part within each line.
The digital hardware described in this chapter has been realised
with TTL components. The two signal levels used are: "0" and ''1 ",
and they correspond with the "low" and "high" level used 1n par.
2.4.
If symmetrie scanning 1s used the video information has to be
reversed in every other line, and the video information of the
lines which are nat reversed has to be delayed one line time in
order to get the same over-all delay. These operations, which are
realised digitally, will be explained with the aid of the block
diagram shown in figure 3.1. The blocks which actually execute
these operations are: (pre) LPF, ADC, LR MEMORY, BLANKING, RL
MEMORY, DAC and (post) LPF. These blocks are present for each video sig-
nal (R, G and B), since the operations are the same for all three
video signals, it suffices to describe them for one video signal:
-72-
r!- LR- MEMORY f-1-M A c R U'
'--
10 r
R,G orB 3 ,....---------1 -1 A 0 C -Î
Cl I
BLANKING ~
[lÜ w I
c!. -1 RL -MEMORY & -
~ A CI.R l'i
I l -ç I
~ 10
A WE OE
,--J-+j-f--l2r CONTROL
I
2 H -----+----!2 H
2V ---,_...~2v
CLI CLCJ ( 1_LR CLRl Sf ST V!E VOE
I I l ' J
I l I I I CLI CLO o...:JR CLLR CLRL S ~ ST VIE VOE CU [lJ] CIDR ST
CLOCK/ST VIE/VOE
Cll ST
L------------1 2V SIN
's
--'-c
IHI1
IHM
PROC m+ii----J
Fig. 3.1: Block diagram of the breadboard model for sinusoidal
horizontal deflection.
-73-
The incoming video signal is low-pass filtered (LPF), AD-converted
(ADC) and stared in a line memory. In the case when the ST signal
is "0", the line of the incoming video signal
and the samples of the ~~ÈiY~ part of this 1
a PLR line,
are written into
the LR MEMORY in order of arrival. In the same time interval the
samples of the ~~!!Y~ part of the RL line are read out the RL
MEMORY in a last-in-f st-out (LIFO) order, and thus the line of
the resulting video s is then an RL line. In the case when
the ST-signal is "I", the line of the incoming video signal is a
PRL line and the samples of the ~~!iY~. part of this line are writ
ten into the RL MEMORY in order of arrival. In the same time in
terval the samples of the ~~tiY~ part of the LR line are read out
the LR MEMORY in a first-in-first-out (FIFO) order, and thus the
line of the resulting video signal is then an LR line. During the
time without read out the line memory, the line of the outcoming
video signal is blanked (=made zero). Finally the samples which
come out the BLANKING unit or the LR or RL MEMORY are DA-conver
ted (DAC) and low-pass filtered (LPF).
The LR and RL MEMORY units are RAMs. Figure 3,2 shows their block
diagram. Figure 3.3 shows the BLANKING unit.
10
RAM f 5. 2l~o7l
WE
i I
\wr ifE fig. 3.3: BLANKING unit
fig. 3.2: MEMORY unit
-74-
In addition to the described basic units, 1.e: LPF, ADC, LR MEMO
RY, RL MEMORY, BLANKING and DAC, five important blocks can be
distinguished, 1.e.:
I. CONTROL unit.
This unit generates the control signals for the LR MEMORY, RL
MEHORY and BLANKING units. Demultiplexing and multiplexing of
the digital video signals are done by an appropriate choice of
the WE and OE signals.
S Signals in: CLI, CLO, CLLR, CLRL, ST, ST, 2H, VIE and VOE.
Signals out:
ALR, Adresses LR-MEMORY;
ARL, " RL-MEMORY;
WELR, Write Enable for LR-MEMORY;
WERL, " " " RL- "
OELR, .Q_utput Enable " LR-"
OERL, " " " RL- "
OEBL, " " " ELANKIN unit.
2. VIE/VOE unit.
The VIE signal determines when the active parts of the incoming
lines are to be written into the line memories. The VOE signal
determines when the active parts of the resulting signals are
to be read from the line memories.
Signals in: CLI, CLO, CLOR, ST, STIHM.
Signals out:
VIE, Video In Enable;
VOE, Video Out ~nable;
3. CLOCK/ST unit.
This unit generates all the clock signals and the ST and ST s1g
nal.
Signals 1n: 2H, 2V, STIHM.
Signals out:
CLI, CLOCK _!_n;
~75-
CLO, CLOCK Qut;
CLOR, 11 11 for Resetting the counter with which the VOE
signal is determinded;
CLLR, I! par. LR operations;
CLRL, 11 I! RL I!
ST, Determines the STate of the processor, see par. 2.4
ST, 11 11 I! " 11 11 11 11 11
4. SIN unit.
5.
This unit generates the sinusoÏdal souree voltage v for the s
ser~es resonance c it described in 2.3. This has been chosen
because it was then possible to use an existing audio power am
plifier to drive the resonance circuit.
Signals in: CLI, ST, 2V.
Signals out: v . s
IHM PROC unit.
This unit uses the voltage IHM over the
If IHM > 0 then "0"; if IHM < 0
The r1s~ng edge of STIHM determines the
and with a 1 also tv0 , see par. 2.4.
Signal in: IHM, I Horizontal ~easured.
resistor Rhm.
then = "1".
phase difference
Signal out: STIHM, STate I Horizontal Measured;
al
These 5 blocks will be described in more detail in the pars~
3.2 through 3.6.
3. 2 The line memorHi!S and blanking control (CONTROL) unit
The addresses for the MEMORY units are generated by counters
(3 for each unit) connected in such a way that they forma 12
bit synchronous up/down counter (of these 12 bits only JO will
be used). These addresses are only generated during the time the
active parts of the video lines are written into or read out the
-76-
line memor~es. The LR COUNTER generates the addresses (ALR) for
the LR MEMORY. The RL COUNTER generates the addresses (ARL) for
the RL MEMORY, see figure 3.4. So to realise the output signals
LR COUNTER IT
CLLR
10
ALR
li CE RL IT
ST UID
CLRL CL
10
ARL
Fig. 3.4: Blockdiagram of the LR and RL COUNTER generating
the addresses for the LR and RL MEMORY units res
pectively.
ALR and ARL of the CONTROL unit we have to realize the input s~g
nals of the counters, i.e.:
LELR (LERL), Load Enable LR (RL) COUNTER;
CELR (CERL), Count Enable LR (RL) 11
U/DLR (U/DRL), ~pjQown LR (RL) COUNTER;
CLLR (CLRL), CLock LR (RL) COUNTER,
The two clock signals CLLR and CLRL need not be realized separate
ly since they are also input signals for the CONTROL unit, so they
can be directly used as input signals for the LR and RL COUNTER
respectively.
Finally we have to realize the following signals: LELR, LERL,
-77-
U/DLR, U/DLR, CELR, , WELR, WERL, OELR, and OEBL.
The first four signals are realised with the ST and the 2H sig
nals. The rest of the signals are realised with the VIE, the VOE
and the ST signals.
The LR and RL COUNTER are loaded on the rising and trailing edge
of the ST signal. The LR COUNTER 1.s loaded with "O"'s at both
edges of the ST signal. The RL COUNTER is loaded with "0"' s at
the rising edge of the ST signal, and with the content of the
RL COUNTER at the trailing edge of the ST signal. Here the
relation between the ST signal and the 2H signal (see par. 2.4)
has been used, and the counters are loaded by a monostable mul
tivibrator (MMV) which is started at the trailing edge of the
2H signal, see figure 3.4
In this way the start and the end of the "O" part of the loading
signals (LE) are not well defined with respect to the edges of
the clock signals. However, this will give no timing problems,
because the "0" part of the loading signal will be a subinter
val of the blanking part of the video lines.
The U/DLR is made "I". So whether there is written into or read
out the LR MEMORY, the LR COUNTER will start with ALR equal to
"0", see figure 3.4, and then count up, in agreement with the
fact that the LR line not reversed. For the U/DRL signal the
ST signal is taken. So when the ST signal is "I" the RL COUNTER
starts with ARL equal to "0" (see figure 3.4), and then counts
When the ST signal 1.s "0" the RL COUNTER starts with ARL equal to
the last ALR, from the preceding ST = "0" part, and then counts
down, in agreement with the LIFO reading sequence for the RL line.
With the VIE signal the active parts of the incoming video signals
are determined. The VOE signal determines the active parts of the
resulting video signals. With the VIE, VOE and ST signals the CE
signals for the LR and RL COUNTER are determined. The CELR and CERL
CLLR
U!IJLR
-78-
signals, in turn, determine, together with the ST signal, the
and OE signals. The WE and OE signals finally carry into effect the
demultiplexing operations on the incoming video signals, and the
multiplexing eperating on the resulting video signals, respective
ly. Let for example the ST signal be "1", then read-out from the
LR MEMORY and writing into the RL MEMORY will take peace. Figure
3.5 shows the time diagram for this situation.
reaJmg in LR-MEMORY
n o o o n n.n !1D ---··--,-------
output
~i~~oo--F-F-+--~_;','_·D~..-~-J_O__j_,L.O__j_,L.L_ AWELR
CLRL
UltîRL
CERL
writ1ng in RL -MEMORY I
output ati:J~ a;~:~tss 0-FF Rl:ti----L-----L
AWERL
Cl
WERL 0 ___ L ___ __j_......J~-'---1----'------ ...1.---J.~-·-'- JODOODCJOC
Fig. 3.5: LR and RL MEMORY control signals during the time
interval in which read-out from the LR MEMORY and
writing into the RL MEMORY take place.
I
-79-
This figure shows that the WERL signal consists of pulses. In
practice these pulses are made with the help of another signal,
AWERL (~id for WERL s ignal).
This signal keeps the sarne value ("!) during the writing period.
The NAND function of the S\mRL signal and a delayed version of the
clock signal will give imRL. So here we will deterrnine the AWERL
(and also AWELR) signal rather than the \{ERL and suppose that both
signals are clocked with edge-triggered flip-flops (D-FF's).
Let the CERL signal change frorn "I" to "0" during clock period I,
and frorn "O" to "I" during clock period 1; in figure 3.5 this has
been shown. The value of ARL just before the RL COUNTER is enabled,
is in fact the first address sent to the RL MEMORY, because of the
D-FF on the RL MEMORY unit, see figure 3.2. This first address will
arrive at the RAM on the RL MEMORY unit during clock period 2.
Hence the AWERL signal has to change forrn "O" to "I" during clock
period 2. The last address will arrive at the RAM on the RL MEMORY
unit during the (l+l)th clock period. Hence the AimRL signal has to
change frorn "I" to "0" during the (1+2)th clock period. So the rela
tion between the AWERL signal and the CERL signal is given by:
AWERL =ST. (CERL(-1) + CERL(-2))
Where "(-k)" denctes a delay of k clock periods,
or AWERL = ST.(CERL(-I).CERL(-2)
or AWERL = ST.(CERL(-I).CERL(-2) (3.I)
In a sirnilar way we find for AWELR, and OERL:
AWELR .(CELR(-J).CELR(-2)) (3.2)(3.2)
OELR ST.(CELR(-I).CELR(-2)) (3. 3)
OERL ST.(CERL(-I).CERL(-2) (3 .4)
-80-
\.Jith and known, OEBL can he determinded, since the three
OE-signals have to be mutual exclusive, so:
OEBL OELR + OERL OELR.OERL. (3. 5)
The relation between the CE-signals and the VIE and VOE-signals
are determined by the ST-signal, since:
CELR VIE.ST + VOE.ST = VIE. .VOE.ST
CERL VIE.ST + VOE.ST = VIE.ST.VOE.ST
or CELR = VIE. (3.6)
CERL = VIE.ST + VOE.ST = VIE.ST.VOE.ST (3. 7)
Now with (3. 1) through (3.7) the signals CELR, CERL, OELR, OERL, OE
, WELR and WERL can all he determined as a function of the sig
nals VIE, VOE, and ST.
Figure 3.6 shows the hardware realisation with NAND-gates and D-FF's.
The WELR and WERL-signals are made with the auxilliary signal AWELR
and AWERL respectively, and a delayed version of the clock.
Fig. 3.6: Hardware realisation of the control signals
WERL, OERL, OEBL, CELR and CERL by means of the signals
VIE, VOE, ST and ST.
-81-
3.3 The memory enabling (VIE/VOE) unit
The time intervals during which the VIE signal is 11 I 11, the 11 1"
parts, determine when the active parts of the incoming video
signals are written into the line memories, whereas the 110" parts
of the VIE signal determine the blanking parts of the incoming
video signals. For the VOE signal the same holds with respect to
the resulting video signals. The VIE and VOE signals are deter
mined with 12-bit up/down counters, see figure 3.4 (of these 12
bits only 11 bits are used). Each of these countersspansaline
duet, since that is the period of the display process with the
sinusoÏdal deflection current. The VIE and VOE signals have to be
present continuously, hence the counters have to be enabled all
the time. That is why special attention was to be payed to the
periodical reset. The counters are reset at the begin of each
line duet, that is at, or just after, the rising edge of the ST
signal.
Fig. 3.6 shows the block diagram of the unit which determines
the VIE signal. The upper part of the diagram shows the part which
determines the VIE signal during normal action.
The lower part used to load the RAM with a function from the
EPROM. This loading will have to be done when the power swit-
ched on, or if another VIE function is wanted. Note that the EPROM
cannot be used directly since it has an access time of 450 ns max.
while I clock period lasts 41~ ns!
The VIE RAM reset with the ST signal, after every line duet.
This reset takes place with an auxiliary counter: The VIE-RAM
RESET COUNTER. When the ST signal "0 11, this counter is loaded
with "!", at every positive edge of the clock. When the ST signal
becomes "!" the RESET COUNTER starts to countdown. At the moment
this counter reaches "0", a ripple carry out (RCO) resets the VIE
RAM COUNTER.
12 BITS VIE RAM SYNCHRON()J
-82-
VIE RAM COUNTER
12 BITS SYNCHRON()JS
VIE I l ~ c-------fl----ro
VIE RAM O-FFs I
+ VIE RAM __1_ UP/DOWN
+ COUNTER RE SET COUNTER
JP/DOWN 11 11 11 f C 0 U N TER l---t---"-i D- FF 1------+-t-t-i
(2x374)
RAM
1 x 4096 WE~
UI
---'r-- I 21471
J_ -
ViE SELECT I ON
SWITCH
VIE EPROM COUNTER VIE EP ROM O- FFs
11 r---- BCD F
MU X f----- ENCO- ~~ Tof--- DER'=".
12 BITS. SYNCHRONOUS
UP/DOWN COUNTER
~ VIE EPROM EPROM
11 8 x 2048 II 1-
11 11 f------+~---t D- FF 1-++--t->-t
_.__ 12716)
-:- 16
LOAD CONTROL
Fig. 3.7: Block diagram of the VIE unit generating the Video
In Enable (VIE) signal.
:I-
During the rest of the time during which the ST signal is "!" the
RESET COUNTER will nat produce a RCO again. If the RESET COUNTER
would be able to count down all the time, the time interval be
tween two RCO's would be 4096 x TCLI" In reality the counter can
only count during I line period, that 1s:
for 20 MHz < fCLI < 30 MHzf (see par. 3.4). The clock frequency of
the VIE EPROM COUNTER is ~~I which is low enough to access the
EPROM succesfully, even if fCLI = 30 MHz.
During normal action the EPROM FF's and theEPROMare disabled and
VIE LOAO
BUTTON
-83-
and the RAM FF's enabled. With the VIE LOAD BUTTON a monostable
multivibrator MMV) is triggered, which produces a load pulse.
During the load time the EPROM FF's and theEPROMare enabled and
the RAM FF's disabled. At the same time wr enable pulses are
sent to the RAM. Note that the VIE EPROM COUNTER is never reset,
so the load pulse has to last at least
2048 seconds.
With the VIE SELECTION SWITCH a choice can be made from the 8 avai
lable VIE functions in the EPROM.
The outcoming (resulting) video-signal enabling (VOE) unit
The VOE unit has the same block diagram as the VIE unit, only the
clock signals and the reset signal differ. The reset signal for
the VOE RAM COUNTER is not determined by the ST signal, since line
duet static linear alignment is applied. Instead of using the ST
signal, the STIHM signal is used to reset the VOE RAM COUNTER.
If the rising edge of the STIHM signal ~s shifted forward in time
with respect to the rising edge of the ST signal, then the reset
moment of the VOE RAM COUNTER and the VOE signal self will also
be shifted forward in time. Thus tv0 changes; however, tv 1, tv2 and tv3 remain constant, just what line-duet static alignment re
quiers. The alignment is also linear because the active parts are
read out with a contstant clock frequency.
But the STIHM signal ~s used the clock signal CLI cannot be used.
The clock CLI is locked to the 2H signal, and hence also to the ST
signal. The phase difference between the clock and the ST signal
therefore is constant. The STIHM signal, however, changes in phase
with respect to the ST signal and so also changes in phase with
respect to the clock CLI (The STIHM signal changes in phase because
the horizontal deflection current i changes in phase, see par. n
2.4. There will be line duets, during which the rising edges of the
STIHM signal and the clock signal CLI coincide or almost coincide.
And since there always will be some jitter on both signals there
-84-
will be differences, of 1 clock period in the reset-time of cor
responding line duets (with the same i ) in fields following. If V
the reset time is shifted 1 clock period, both the LR line ánd
the RL line will be shifted 1 clock period forward ~n time. So on
the screen the LR line will be shifted TCLI'v (v being the spot
velocity) to the right while the RL line will be shifted TCLI'v
to the left, giving a total difference on the screen of 2 TCLI'v
(=1,6 mm with the A66-540X tube used here). That is too much.
Note that a scanning raster produced by an asymmetrie horizontal
current would have a fault of only TCLI'v. Thus with symmetrie
deflection reset timing faults produce faults on the screen, which
are twice as big as with an asymmetr deflection current.
Instead of using the clock signal CLI, the clock signal CLO is
used as a system clock for reading. This clock signal made by
changing the phase of the clock signal CLI. Figure 3.8 shows the
block diagram of the unit which generates the clock signal CLO.
INTERVAL
DEtECTOR CODER
k
MUX
CL')
Figure 3.8: Generation of the clock signal GLO.
One clock period of the clock CLI is subdivided into k intervals
with the aid of delay elements (T). The k new clocks signals
-85-
created this way are used to detect the interval (1 through k),
in which the STIHM signal changes from "0" to "I". The CODER deter
mlnes which of the k clock signals finally will be used (which clock
clock will be chosen and how this is done will be treated in par.
3.4).
Now consider three periods of the ST signal as shown in figure 3.9.
(a)
(b)
(c)
STIHM!
,] (d)
u ~~--~--'-----' ~[.' (e)
Fig. 3.9: Three line duets (a) and three corresponding periods
of the ST signal (b), the sinusoirlal horizontal de
flection current (c), the STIHM signal (d), and the
STIHM signal (e).
In each ST period a new clock signal CLO will be determined.
So the clock signal CLO can be different in each ST period.
If the (i+2)th line duet is read out of the line memories then
the VOE RAM RESET COUNTER is started at the positive edge of
the STIHM signal in the (i+l)th ST period. At the reset moment
of the RAM COUNTER, that is in the begin of the (i+2)th ST
period, the clocks of the RESET COUNTER and the RAM COUNTER
-86-
have to be the same, otherwise reset cannot be done properly.
That is why the following strategy has been used:
The clock signal CLO to be used in the (i+2)th ST period is de
termined in the ith ST period with the rising edge of the STIHM
signal. This clock signal will also be used by the RESET COUNTER
during the second half of the (i+1)th ST period and the first
half of the (i+2)th ST period.
This clock signal used by the RESET COUNTER is called CLOR. This
clock signal is changed at every rising edge of the ST signal.
The complete realisation of the unit producing the clock signals
CLO and CLOR will be given in par. 3.4.
In figure 3.10 the block diagram of the unit producing the VOE
signal has been shown. Note that the RESET COUNTER is loaded
with "0001100000000" (=384 decimal), so the RCO is produced a
quarter of an ST period after the rising edge of the STIHM sig
nal, if fCLO = 24 MHz. VOE ..--~---1 u..l----
RCO VOE RAM COUNTER .----------------+---------t6 VOE RAM O·FFs
VOE RAM 12 BITS
SYNCHRONOO UP/DOWN COUNTER
12 BITS SYNCHRONCIJS VOE RAM
bits Q.6•9 11
RESET COUNTER
CLO
12 UP/DOWN
11 COUNTER t--+--+----i
VOE EPROM COUNTER
11
VOE EPRtJ-1 0-FFs
12 BIT:; SYNCHRONCIJS
UP/DOWN 11 COUNTER 1-----+----;
Fig. 3.10: Block diagram of the VOE unit generating the Video Out Enable (VOE) signal.
RAM
1. 4096 0{
(2147)
EPROM
e. 2048 ll'
11716)
LOAO CONTROL
VOE SELECTION
SWITCH
VOE EPROM
V8E
1- LOAD
.,ç BUTTON
CLO
-87-
To extend the static control possibilities for the alignment of
the LR lines with the RL lines (see par. 2.4), an extra delay unit
has been added. This unit enlarges tv2 , see figure 2.16, by delay
~ng only the RL reading part of the VOE signal, and can be used
in combination with the eight VOE functions in the VOE EPROM.
Tagether they make a variable range of 64 clock periods in tv2 possible. The variation, in tv2 is, of course, a static variation,
that is to say if tv2 is changed then it is changed for all line
duets. Figure 3.11 shows the block diagram of this delay unit.
BCO
CO OER
re ss
fig. 3.11: Block diagram of the RL DELAY unit offsetting the
RL line with respect to the LR line
3.4 The clock and state signal (CLOCK/STl unit
The clock signal CLI has been made with an crystal Phase-Locked
Loop (XPLL) which uses the line synchronisation signal 2H as re
ference. The system has been built with a 24 MHz clock. But it
is also possible to use other clock frequencies between 20 MHz
and 30 MHz, if only the ratio of the clock frequency and the line
frequency is an integer. The video signal in the 100 Hz system
has a bandwith of 10 MHz, twice the bandwith of the video signal
in the 50 Hz system. So the clock frequency of 20 MHz, being
twice the signal bandwith, is a lower limit on fCLI' The upper
limit is determined by the components used.
VOE
-88-
The clock signal CLO has the same frequency as the clock signal
CLI but it differs in phase. The phase of the clock signal is de
termined by the STIHM-signal. tlliy this is necessary and how it
is clone have been explained in par. 3.3. That paragraph (3.3)
also introduces the clock signal CLOR for the RESET COUNTER. The
block diagram of figure 3.7 can be extended to one that gives the
clock signals CLO and CLOR as outputs. Figure 3.12 shows the ex
tended blockdiagram.
Cll k -r[DrE}.F(Drr······· ... --· ................. ----(D-
w0 cu1 cu2 CLs Q.lk- 1
l J ,.......:......~. ............... _ _,L...,
Si"'iii'M INTERVAL f-- COOER i--- MU X
DE1ECTOR
MUX
Fig. 3.12: Block diagram of the CLO/CLOR unit generating the
clock signals CLO and CLOR.
-89-
Figure 3.13 showshow the block diagram of figure 3.12 has been
realised with TTL components. One clock period of the clock sig
nal CLI has been divided in 4, nearly equal, intervals with AND
gates. With the flip-flop's D-FFla through D-FFld the interval
in which the rising edge of the STIHM
ned.
found, will he determi-
Take, for example, the case that the rising edge of the STIHM
signal is in interval I. Then the Q output of D-FFlb will become
"!" and hence the other flip-flops D-FFla, D-FFlc en D-FFld will
remain "0", since they are reset by D-FFla by means of its Q out-
put.
Note that the propagation time of the AND-gate connected to the
R input of the flip-flop, has to he less than 1 quarter of the
clock period TCLI'
Fig. 3.13: Realisation of the CLO/CLOR unit.
-90-
The CODER of figure 3.12 which determines the clock number is for
med by the BCD CODER, the ADDER, MMV2, triggered by the field syn
chronisation signal 2V and FF1a through FF4b. MMV2 divides one
period of the field synchronisation signal into two equal pieces
(see figure 3.14).
2V
·---'----'------' [ ..
Fig. 3.14: Sub-division of one period of the field synchronisation
signal, 2V, into 2 equal intervals.
During the time the upper half of the screen is scanned, the r~s~ng
edge of the STIHM-signal will shift forward in time with respect
to the ST-signal. During the scanning of the lower half of the
screen the STIHM-signal will shift backward in time. The clock s~g
nal CLOR to be used in the secend half of the (i+l)th ST period and
the first half of the (i+2)th ST period determined in the ith ST-
period (see figure 3.9 in par. 3.3).
During the time interval determined by the moment the clock signal
CLOR is determined, and the moment the STIHM signal starts the RESET
COUNTER (see par. 3.3), the signal will shift in time, in a
direction that is different for the upper and lower part of the
screen, see par. 2.4. Figure 3.15 shows this shift in time.
When the rising edge of the STIHM signal in the ith ST-period ~s
found in the jth interval, then in the upper half of the scYeen the
clock mod(j-~+4 ) will be taken and in the lower half of the screen
the clock j will be taken.
-91-
nsmg
t t t t 4
t edges ~loc~~
CLI 1 CLJ2
CLJ3
r1s 1ng
t t ~ ··± ·-S I .. ---·---·· (i•1) ST-period iS T-penod [i+1l ST-pericd
Ja.;er part screen upper part screen
Fig. 3.15: The change in the STIHM signals for the lower
and upper part of the screen.
The clock signals CLLR and CLRL are made by multiplexing the
clock sig~als CLI and CLO with the aid of the ST signal. The clock
signal CLLR is equal to CLI if the ST signal is "0", and equal
to CLO if the ST signal is "I"; the clock signal CLRL is equal
to CLO if the ST signal is "0", and equal to CLI if the ST
s i:gnal "I".
The ST and ignal are obtained by dividing the 2H-signal by
2, by means of a toggle FF.
3.5 The sinusoirlal deflection current unit
The sinusoÏdal deflection current ih is made with the aid of the
series resonance circuit shown in fig. 2.20 of paragraph 2.3.
The voltage-souree has been realised with a small signal voltage
souree followed by an audio power amplifier.
The signal and the small signal souree has to be:
1. Very stabil in frequency and phase
2. Very little distorted
J. Amplitude modulated, with an adjustable modulation function of
field frequency.
[ ll ( 24Miizl
ST
2V
-92-
4. Adjustable ~n frequency and phase.
The first two requirements are necessary because we want to apply
line-duet static linear alignment. The amplitude modulation ~s
necessary to compensate for the pincushion distartion in East-1-Jest
direction, see par. 2.2.3.
The frequency of the signal leaving the signal souree has to be
adjustable, because we also want to use it for a 50 Hz-system,
where a frequency of! 7.5 kHz is required. Adjustment of the
phase is necessary to cornpensate for signal delays in the ampli
fier.
The digitally generated sinusoidal voltage from the signal souree
shown in figure 3.16, meets all these requirements.
I.MHzl _,_,
SIN
EP ROM COUNTER
I
R~SET V
SIN-EC
(2HHz)
13 SIN
a::XJres EP ROM
t I
I
2 I SIN- ~I
CONTROL 9 SIN- ~ EPRO~ ~ CONTROl I
COUNTER EPROH
RESET
I I I I
I SIN-CEC
J L_ __ _ ---
'Z
I I 2 V
16 PARALLEL IN DIGITAL
data SERlAl OUT dot a F lL TER
2
I
ANALOG ~ 1-d-a.,ta--t.__F_IL_TE_R--J~ /-
AOC
data
Fig. 3.16: Block diagram of the SIN unit generating the sinus
aidal voltage driving the series resonance circuit.
-93-
In the SIN-EPROM 16-bit samples necessary to generate an amplitude
modulated sinusoidal voltage are stored. Instead of using a DA-con
verter followed by an reconstruction filter, here a combination of
a digital filter followed by an DA-converter and an analog filter
~s used. This combination ~s also used in the "Compact Disc".
In this way a sinusoidal voltage with a distartion suppression of
more than 80 dB is obtained. Amplitude modulation is done by storing
all the different sample values of the sinusoÏdal voltage within one
field. The frequency and the phase of the sinusoidal voltage can be
changed by changing the samples in the EPROM's. The necessary control
signals are made with the aid of the SIN CONTROL EPROM.
3.6 The horizontal deflection curYent-measure~ent nrocess~ng
(n1M PROC) unit
This unit employs IHM, the voltage drop over the measurement res~s
tor Rhm (see figure 3. I) to obtain the necessary infonmatien for the
applied alignment.
In this case line-duet static linear alignment ~s used and thus the wan
ted information ~s the change in phase (a 1) of the deflection current
ih with respect to the ST signal. The zero crossings of the IHM sig-
nal give this information. The sinusoÏdal IHM signal H converted
into a "TTL" signal, the STIHM signal, by means of a smitt-trigger
(comparator with positive feedback, see figure 3.17). As worked out
in par. 3.3, the rising edge of this STIHM signal can be used to de
termine the phase variation.
Figure 3.17 also shows an adjust'lhle de1ay, unit included for the
following reason. Consicier one period of the sinusoidal current
as shown in figure 3. 18 and two picture elements to be aligned,
x and y, each having a width of wCLO; wCLO being the width corres
ponding with one period of the clock.
The time interval t between the "writing" on the screen of the y middle of picture element x and the middle of picture element y,
is given by:
t y (3. 8)
-94-
-- ....... STIHM
LAY UNIT
75
Fig._3.17: Hardware realisation of the IHM PROC unit, process~ng
the voltage drop IHM over the rneasurement resistor ~·
Fig. 3. 18: The position of two aligned picture elernents x and y
with respect to the sinusoidal horizontal deflection
current.
-95-
where k 1s an integer; given by the way the VOE signal 1s geney rated (see par. 3.3).
In case of perfect alignment
d d x y
or 2t + t x y (3. 9)
TCLI' where kCLI 1s an integer (kCLI is
an integer because the clock signal CLI 1s locked to the line syn
chronisation signal 2H). Now s1nce TCLO = TCLI (see par. 3.3 and
par. 3.4), equation (3.9) changes, us1ng equation (3.8), into:
2t x (3. I 0)
6 where kx = 3kCLI - ky 1s an integer.
Thus to get proper alignment, 2t has to be an integer multiple of x the clock period TCLO
The time between the zero cross1ng
and the rising edge of the clocked
dih . of ih where ~ 1s negative,
STIHM signal resetting the
VOE RAM COUNTER, will in general not be an integer multiple of
TCLO; nor will the time between the moment when the sample x is
read out of the line memory and the moment when the picture ele
ment x is actually written on the screen, be an integer multiple
of TCLO' Thus tx will in general not be an integer multiple on
TCLO' That is why a delay, adjustable between 0 and !TCLO' is
included. This delays range follows from (3.10), and is realised I
in practice, with a seriesof 8 AND gates, each with delay T6 TCLO'
-96-
4. DISCUSSION AND EXPERIMENTAL RESULTS
In chapter 2 the theoretical implecations of the use of a sinus
aidal current for horizontal deflection have been considered.
In chapter 3 a hardware model has been realised based on these
theoretical considerations. Thereby choices have been made concer
ning: (I) The amplitude of the sinusoidal current, and the way
the sinusoÏdal current has been generated; (2) the kind of align
ment applied. These choices will be discussed here. Besides that,
we will consider the power dissipation, coil voltage and radiation
in the resulting deflection circuit and compare these fenomena
with the ones in the conventional deflection circuit.
4.1 g~~i~~-~~-~~E!i~~~~-~~~-g~~~!~~i~~-~~~~~~-~~-~~~-~i~~~~!~~! deflection current
The amplitude of the sinusoïdal current can be determined with the
horizontal deflection current giving a geometrically correct pic
ture, see par. 2.1. This deflection current can be determined the
oretically and experimentally. The theoretical current is deter
mined by the radius ~ of the deflection system and the radius R5 of the screen. For the picture tube used here (A66- 540X), these
values are:
~ = 26.9 cm 114 cm.
The experimental current can be determined by driving the horizon
tal deflection coil with an adjustable DC current, and measuring
the deflection of the spot.
Such a measurement has been clone. To this end, a ruler without pa
rallax has been mounted in the xz-plane along the x-axis (see
figure 2.2) in order to obtain the current giving a geometrically
correct picture for a remote observer. The results of this measure
ment are reproduced in table 4.1.
-97-
Table 4.1: Horizontal deflection current I, as a function
of the spot deflection, x.
1 l.n Ampère x 1.n cm
0 0 • I 05 .85 .206 I. 60 .309 2.40 .407 3.20 .507 4.00 .757 6.05 I. 000 8.10 I. 19 10.00 I. 90 12.00 1. 60 14.00 I. 79 16.00 I. 96 18.00 2. 11 20.00 2.28 22.00 2.41 24.00 2.52 26.00
(extreme right)
With (2.20) and (2.28) the values found 1.n table 4.1 can be used
to plot the measured deflection current 1.n the same way as the
calculated currents are plotted. Figure 4.1 shows the result.
In this figure the theoretically calculated current has also
been shown. For I/C = (ih/C)E the difference in T corresponds
to a difference of circa 5 mrn on the screen (with an A66-540X
tube). Figure 4.1 also shows the normalised sinusoÏdal current
for k=1 correspondig to the measured curve. The norrnalised
amplitude A is 0.7734, corresponding with an amplitude of 2.63 n
ampère in reality. Figure 4.2 shows the normalised sinusoidal
currents for k= 1.05 and k=l.l, both corresponding to the measu
red curve. The latter sinusoidal current with k=l.l appears to
match the measured current well. For k=l.l the normalised ampli
tude A would have to be 0.8071, corresponding with an amplitude n
of 2.75 ampère in reality, and the clock CLO would have to be
26.40625 MHz rather than 24 MHz (k=I); the clock-frequency being
-98-
l/C ~R FI
1.0 ____________ MERS, I IC, R~=. 2359649 I 2, C=3. 40 I 39 I 57~ ______ l/C, Rl'J=.235964912 -~------~---
- ___ 11 I. 293*5 IN (I. 32*TRU )----- - -------·- -----~
0.5
I ,'fa
0 -!L-+--f-+--+--+-+-+---+--+--+--+-+--lr---+--+--+--t-t--t-4--t---t-+' ---+---+-+-+-+-+-+ 0 0.5 1.0 1.5
Fig. 4.1: Comparison of the measured normalised horizontal
deflection current (A66-540X tube) with the norrna
lised sinusoidal current at standard spot velocity.
an integer multiple of the line frequency f1
( = 31,25 kHz).
In the breadboard model described in chapter 3 we have taken k=l,
but with this hardware realisation it is also possible to use different
frequencies for the clock signals CLI and CLO; hence, since the am
plitude of the sinusoidal deflection current can be continuously va
ried, it is possible to use other values for k.
The theoretical current calculated 1n par. 2. I and the s1nus
oÏdal current constructed to match it indicate very clearly that
--99-
1/C OR FI
1.0 T ····· ..... MERS. 1/C. "". 23596'912. c •'-'0 139!57 r ______ 1 / 1 . 2 3 9 * S I N I I . 2 * T RUl. k = 1 1 - -- - -- ---------~ ____ l/!.267*SINI!.258*TRUl, k=105---- --------------\ \
+
0
Fig. 4.2: Comparison of the measured normalised horizontal
deflection current with normalised sinusoÏdal
currents at different spot velocities.
a sinusoÏdal current may be used for horizontal deflection with
out giving unduly large geometrical distortion. This has been
verified in practice. That is why par. 2. I has been of great
value. But if we want to determine the sinusoidal current gi
V1ng minimal geometrical distortion, then the following questions
have to be answered first: How is geometrical distartion to be
defined 1n an objective manner, realising that the observer will
apply subjective criteria? Which current do we use as a reference
the theoretically determined current or the measured current?
-100-
The latter question is brought about by the small discrepancy
between the measured and the calculated current.
In the breadboard model described here we used k=l corresponding
to an amplitude of 2.63 ampère. This amplitude value only holds
for the horizontal deflection current in the xz-plane. The ampli
tude of the deflection current in the other planes parallel to
the xz-plane is determined by amplitude modulation, see par.
2.2.3. In this case (Ah 2.63 ampère) az, of equation (2.50) was
taken equal to -4650, while the other coëfficients of (2.50) were
taken zero.
In this breadboard model the ser1es resonance circuit, described
in par. 2.2.1, has been taken to generate the sinusoÏdal deflec
tion current. The only reason for this choice was that we then
simply could use an existing audio power ampl . In this re-
port no choice is made, however, a future choice should be made
by considering, inter al
- the power dissipation of the total deflection circuitry;
- the possibility of integration;
- the alignment involved.
The latter subject will be discussed 1n the following paragraph
(par. 4. 2).
4.2 Choice of al
With the hardware realisation discribed in chapter 3 it is also
possible to apply static linear alignment as described in par.
2.4. Therefore the clock signals CLO and CLOR have to be exchanged
by the clock signal CLI, the STIHM signal has to be exchanged by
the ST signal, and the VOE RAM RESET COUNTER (see figure 3.10)
has to be programmed with "1" rather than with "384".
With static linear alignment applied the distartion shown 1n fi
gure 2.18b 1s obtained. So the supposition (2.57) of par. 2.3
appears to be fulfilled and then the deflection current will not
satisfy the requirements a, b and c statet in par. 2.4.
-101-
The distorted grid eau be used to measure the phase changes in a1
(see par. 2.4). By choosing appropriate VOE functions and by ad
justing the delay of the RL line with respect to the LR line with
the RL DELAY unit (described in par. 3.3), the lines of the grid
shown in figure 2. 17b eau be made to coïncide everywhere on the
screen. Consider two different periods of the horizontal deflec
tion current in one field, one corresponding to i = i 1 and V V,
one to iv = iv, 2 . The change in a 1 (~a 1 ) between these two diffe-
rent periods is determined by the number of clock periods, TCLO'
which the RL line has to be shifted in order to bring the lines
of the grid together at the vertical positions corresponding to
1 = i 1 and i = i 2. If this is done for the deflection cur-v V, V V,
rent periods corresponding to the top and the middle of the screen,
then ~a 1 appears to be determined by 8 clock periods, giving:
8 x TCLO a -a = 4 x 360°.
l,middle l,top Th
The factor ! states the fact that a change in a 1 corresponding to
one clock period TCLO produces a fault on the screen of 2v.TCLO
(v being the spot velocity), just like a reset timing fault, see -6
par. 3.3. With fCLO 24 MHz and Th= 64.10 , we find for ~a 1 ,tm:
In the top of the screen, with the situation shown in figure 2.18b,
this corresponds to a difference of 6,7 mm between the LR and the
RL line (with au A66-540X tube). The change in a 1 per line duet,
approximated by:
0 .012 '
where n 1s the number of lines in one frame (n=625).
With line-duet static linear alignment, as described theoretically
in par. 2.4, and practically in par. 3.3, the distortien as shown
-102-
1n figure 2.18b will disappear. Thus the deflection current seems
to satisfy the requirements a, band c',described in par. 2.4.
It should, however, he noted that the phase difference a 1 1s deter
mined bye from (2.58b):
w(6L -61 ) res m
where the denominator Rt is given by:
Rt = Ri + Rh.
For the deflection coil used here (AT1270) Rh= 1.35Q. Ri is quite
large, however, since an extra resistor of 2.75Q is inserted in
the resonance circuit in order to match the deflection circuit to
the audio amplifier. By designing a power amplifier especially for
this resonance circuit, R. could he decreased, thereby increasing 1
the Q factor and decreasing the dissipated power. The decreasing
Ri howeve~would result in an increasing 6a 1, and hence it just
might he possible that the sinusoidal deflection current did not
satisfy conditions a, band c', stated in par. 2.4, anymore; so
that the application of line-duet static linear alignment would
give distortion. In this case line-duet dynamic linear alignment
could he considered, see par. 2.4.
4.3 Power diss and radiation 1n the sinusoÏdal
driven horizontal deflection circuit
In the breadboard model described here the losses in the horizontal
deflection yoke have been determined at a line frequency of 31.25 kHz.
The results are reproduced in table 4.2. The peak stored energy in
the yoke was 4.8 milijoules, about the same amount of energy as Bab
cock and Wedam kept up during their measurements [3] (see also par.
1.2). If the results obtained by Babcock and Wedam (table 1.1) are
compared with the results obtained with the breadboard model des
cribed here (table 4.2), it is noticed that in case of the sawtooth
-103-
Table 4.2: Losses ~n the horizontal deflection yoke
exited with a sinusoidal deflection current
ad 31,25 kHz
2 tot al eddy scanning p-p voltage total yoke yoke I R current & hys-frequency over Lh loss loss teresis loss
like deflection current the eddy current and hysteresis losses are
larger than in case of the sinusoidal deflection current. If, for
example, table !.la (at 31.5 kHz) is compared with table 4.2, the
total losses appear to be reduced from 15.2 1.Jatt, in the conven
tional case, to 8.75 Watt, in case of the sinusoidal deflection
current. The power dissipated in the amplifier end-stage has nat
been considered yet, but it appears that this dissipation will
be far less than the dissipation ~n the horizontal output tran
sistor and damper, especially at line frequencies of 31.25 kHz
and higher. There is also less power dissipated in the pin-cus
hion correction circuitry of the breadboard model than in the eer
responding circuitry of the conventional TV-set, because in the
farmer case the correction is realised on small signal level and
in the latter case on large signal level.
The high fly-back voltages in the conventional system limit the
self-inductance, Lh' of the horizontal deflection yoke. In the
system used by Babcock and Nedam [3J, see par. 1.2, the self
inductance had to be lowered to limit the high fly-back voltages.
These voltages had to be limited because the components would
nat have withstood higher voltages. At the scanning rate of
31.5 kHz, they used a self-inductance of 0.26 mH to limit the
fly-back voltage to 880 volt peak-to-peak (p-p). In the bread
board model described here, a self-inductance of I .35 mH was
used giving a voltage over Lh of only 700 volts p-p. Again the
deflection circuit using the sinusoidal current appears to be
superior.
~104-
The High Voltage (HV) for the picture tube in the conventional
system is being made by transforming the high fly-back voltages on
the isolation coil (see figure 1.6). In the breadboard model this
HV thus far has been made in the conventional way by using a dummy
coil in a waoden box. If syrnmetrical scanning actually is going to
be used a new HV souree has to be developped. This HV souree will
have to be developped independently from the horizontal deflection
circu • However, this will offer the possibility to optimize them
both separately (e.g. reduction of the internal resistance of the
HV source).
It is obvious that the radiation produced by the sinusoidal dri-
ven deflection circu is less than the radiation produced by the
conventional circuit, because the high frequency components due
to fly-back are eliminated.
-105-
CONCLUSIONS
It has been demonstrated, theoretically and experimentally that
a sinusoirlal current can be used for horizontal deflection ~n a
TV-tube without giving unacceptable geometrical distortion. The
alignrnent of the lines scanned in opposite direction was and will
be the major problem in a TV system, which uses a symrnetrical ho
rizontal deflection current. Special attention will have to be
payed to the alignment, if the ampl ier-end-stage changed
such, that the Q-factor of the horizontal deflection resonance
circuit is increased, because this will enlarge the phase shifts
of the horizontal deflection current.
Owing to the use of the sinusoirlal current the power losses and
the high voltage have been reduced considerably. At a line fre
quency of 31.25 kHz, as used in 100Hz TV-sets, the total losses
in the horizontal deflection yoke have been brought back from
15.2 watt, in the conventional case with the sawtooth like current
waveforrn, to 8.75 watt in the case of the sinusoirlal current at
the same line frequency. The peak-to-peak voltage over the hori
zontal deflection yoke has been reduced from ca. 2500 volt to
700 volt, while maintaining the same self-inductance (1.35 mH) as
in the conventional system at 15.625 kHz. The radiation caused by
the horizontal deflection circuit has also been reduced.
In the breadboard model the large signal part of the horizontal
deflection system can be reduced to an amplifier end-stage, a ca
pacitor a measurement resistor and the horizontal deflection coil.
If this compared with the large signal part of the horizontal
deflection system in the conventional TV (East-West correction in
cluded), it appears that the power dissipated in the large signal
part will be far less ~n the breadboard model than in the conven
tional TV; especially at frequencies of 31.25 kHz and higher.
The breadboard model, however, has extra small-signal circuitry
in the horizontal deflection system. Although the power dissipa-
ted this circuitry can be reduced by means of monolithic inte-
gration, it should be taken into account when the total power dis
sipation of the breadboard model is determined.
Befere a full compar~son, concerning picture quality and power
dissipation can be made the following needs to be clone:
- The vertical sawtooth current has to be modified into a stair
case current, see par. 2.5;
- a new way has to be found to generate the high voltage for the
picture tube independently from the horizontal deflection c
cuit;
- an amplifier end-stage, especially adapted to the horizontal
deflection circuit, used here, has to be designed.
The separation of the high voltage souree from the horizontal de
flection circuit makes it possible to optimize them both separa
tely {e.g. reduction of the internal resistance of the high vol
tage source).
-107-
REFERENCES
[1] P.S. Carnt and G.B. Townsend,
Colour Television: NTSC-systern,
life Books, Ltd., 1961.
[2] P.S. Carnt and G.B. Townsend,
Colour Television: PAL, SECAM and other systerns,
life Books, Ltd., 1969.
[3] Uwe E. Kraus,
"Verrneiding des Grossflächenflirrrrnerns 1n Fernseh- Heirn
empfängern",
Rundfunktechnische Mitteilungen, Jahrg. 25, 1981.
[4] W.E. Bab<".ock and W.F. Wedarn,
·~racticalconsiderations in the design of horizontal deflection
systerns for high definition television displays",
IEEE Transaction on Consurner Electronics, Vol. CE-29, No. 3,
August 1983.
[5] A. Bruce Carlson,
Corrrrnunication Systerns, second edition;
appendix C: "Television and Facsimile systerns",
Tokyo, McGraw-Hill, Kogakusha, 1975.
[~ N.K. Zworykin and G.A. Morton,
Television; chapter 5: "Fundarnentals of Television",
New York, John Wiley & Sans, Inc., 1954.
[7] Alan A. Liff
Color and Black & White Theory and Servicing;
chapter 15: "Horizontal Output Amplifier Systerns",
Englewoold Cliffs, N.J. Prentice-Hall, Inc., 1979.
-108-
[s] Donald G. Fink,
Television Engineering Handbook; paragraph 6.5:
"Geometrie aspect distortion",
New York, McGraw-Hill, Inc., 1957.
[9] J.Davidse,
Elektronische Beeldtechniek; paragraaf 3.1:
"Informatieinhoud van een beeld".
Utrecht/Antwerpen, Het Spectrum B.V., 1973.
[I 0] Donald G. Fink,
Television Engineering Handbook; paragraph 3.1:
"Fundamentals of magnetic deflection",
New York, McGraw-Hill, Inc., 1957.