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VC 2012 Wiley Periodicals, Inc.

EFFECTS OF THE BIASING VOLTAGE OFMODULATOR ON THE PHASE NOISE OFOPTO-ELECTRONIC OSCILLATOR

Jun Hong and Chun YangSchool of Electronic Science and Engineering,Southeast University, Nanjing, Jiangsu Province, China;Corresponding author: yangchun_seu@163.com

Received 30 May 2011

ABSTRACT: Low biasing technique has been used recently to reduce

the relative intensity noise of the microwave optical fiber links. In thisarticle, the possibility of reducing the phase noise of opto-electronic

oscillator (OEO) by using low-biased optical fiber link is verified.Theoretical and experimental results show that the single sideband(SSB) phase noise of OEO can be reduced at the maximum by 2.8 dB in

low-biased microwave optical fiber links when compared withconventional quadrature-biased links. The optimal biasing anglecorresponding to the minimum SSB phase noise increases with the

intensity of the optical power into modulator. VC 2012 Wiley Periodicals,

Inc. Microwave Opt Technol Lett 54:689–692, 2012; View this article

online at wileyonlinelibrary.com. DOI 10.1002/mop.26642

Key words: opto-electronic oscillator; biasing voltage; single sideband;phase noise

1. INTRODUCTION

Low-phase-noise microwave oscillators are at the core of high-

performance systems for communication, radar, and instruments

[1–3]. Microwave oscillator over GHz can be realized by syn-

thesizing a crystal oscillator, but frequency multiplication makes

phase noise worse. Dielectric resonator oscillator (DRO) is

another frequency source, but the phase noise increases rapidly

with the carrier frequency beyond 10 GHz. The opto-electronic

oscillator (OEO) is a novel hybrid oscillator with both electronic

and optical outputs, first introduced by Yao in 1995 [4, 5],

which has been proven to be a promising microwave source of

low phase noise, providing spectrally pure signals up to 80

GHz. Compared with traditional electronic oscillators, the most

significant characteristics of the OEO is that the quality factor

and thereby the phase noise are independent of the oscillation

frequency due to ultralow transmission loss and huge bandwidth

of optical fiber delay line.

Till now, much work on the OEO has been accomplished,

including studies on different structures (characterized by paral-

leled multiloop [6–8], injection-locked dual loop [9, 10], optical

filtering [11–13], and so on), noise analysis and suppression

[14–16], tenability [17], long-term stability [18], and even mea-

surement methods for low-phase noise signal [19]. However, in

above references, the effects of different biasing voltage of mod-

ulator on the SSB phase noise have never been studied

systematically.

In this article, we build the OEO model considering low

biasing condition of Mach-zehnder electro-optical modulator.

Based on this model, the SSB phase noise of the OEO is ana-

lyzed by changing the bias angle of the electro-optic modulator.

Theoretical and experimental results show that the high optical

power injection and the low-biasing voltage can lead to the

improvement of phase noise characteristics. In the following

sections, we first build the theoretical model for phase noise

under low biasing condition of single-loop OEO, and then verify

this model by experiments.

2. THEORY

The configuration of single-loop OEO is shown in Figure 1. The

setup consists of a laser diode (LD), a variable optical attenuator

(VOA), a Lithium niobate Mach-Zehnder intensity modulator

(MZM), a Single-mode optical fiber (SMF), a Pin-photodiode

(PD), a microwave amplifier, a tunable microwave attenuator, a

narrowband filter, and a microwave coupler. The light wave

(black line) from the LD is sent to the MZM, modulated by

oscillating signal originating from noise at the microwave input

port of the MZM, and then sent to the SMF fiber. After trans-

mission through the optical delay line, the optical signals turn

into electrical signals (dashed line) through the PD, after being

amplified and filtered, and then feed back to the electric port of

the MZM. Specific single-frequency signal, of which loop gain

is >1 and phase-shift is a multiple of 2p, is able to oscillate.

This loop can be considered to consist of two types of links:

one kind of link is the so-called microwave photonic link con-

taining LD, VOA, MZM, fiber, and PD and the another kind of

link is a typical microwave link composed by AMP, attenuator,

phase shifter, filter, and coupler.

The modulated optical signal after the MZM can be

expressed as

Pm ¼ 1

2aPo 1þ f cos

pVp

Vm

� �� �; (1)

where a is the insertion loss factor of the modulator, Vp is the

half-wave voltage, Vm is the MICROWAVE signal plus DC bias

voltage determined by Vm ¼ Vin þ Vb, Po is the input optical

power, and f determines the extinction ratio of the modulator by

(1 þ f)/(1 � f). In the following analysis, we assume f ¼ 1.

The MICROWAVE signal applied to the modulator is assumed

as a sinusoidal wave with an angular frequency x, and then

Vin ¼ VMW sin xt. The DC phase-shift amplitude is defined by

Øb ¼ pVb/Vp and the microwave phase-shift amplitude ØMW ¼pVMW/Vp. After transmission through the optical delay line, the

modulated optical signals convert to the electric signals though

PD and the photocurrent is then expressed as

Figure 1 Schematic diagram of the single-loop OEO. [Color figure

can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 54, No. 3, March 2012 689

Iph ¼ 1

2qapo 1þ cos

pVp

Vm

� �� �; (2)

where q is the responsivity of the PD. Expanding Eq. (2) with

the Bessel function of the first kind and the photocurrent is

Iph ¼ 1

2qapo½1þ J0ð/MWÞ cosð/bÞ� � qapo sinð/bÞ�

Xþ1

0

J2nþ1ð/MWÞ sinðð2nþ 1ÞxtÞ

þ qapo cosð/bÞXþ1

1

J2nð/MWÞ cosð2nxtÞ:

(3)

To simplify Eq. (3), Jnð/MWÞ � ð/MWÞn=ð2nn!Þ is used in

the case of small signal approximation for VMW < Vp. It can be

seen from Eq. (3) that the photocurrent comprises a DC term

and AC terms. The DC current is determined by

IDC ¼ Ioð1þ cos/bÞ; (4)

where Io is the DC photocurrent in the case of cosØb ¼ 0. The

total noise-density input to the oscillator is given by

qN ¼ kBT � Fþ 2qIDCRþ RINI2DCR; (5)

where qN is the total noise-power spectral density, kB is Boltz-

mann’s constant, T ¼ 290 K, F is the MICROWAVE ampli-

fier’s noise figure, q is the electron charge, RINI2DC is the laser’s

relative intensity noise, and R is the output impedance.

The output signal SSB phase noise is given by [4]

LðfmÞ ¼ d

ð2� d=sÞ � 2ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi1� d=s

pcosð2pfmsÞ

; (6)

where s is the circle delay time, fm is the offset frequency from

the carrier. d is the noise-to-signal ratio, determined by

d ¼ qNG2A=Posc, where GA is the amplifier’s voltage gain and it

is varied through a post-microwave attenuator, and Posc is the

oscillation signal’s power determined by

Posc ¼ 4V2p

p2R1� 1

Gs

� �: (7)

Gs is the open-loop gain determined by Gs ¼ GP � GA, con-

sisting of the gain of the microwave photonic link GP and the

gain of the microwave link GA. Note that we convert the loss of

other microwave devices to the GA for easy calculation. When

d=s << 1, Eq. (6) can be simplified by

LðfmÞ ¼ d2� 2 cosð2pfmsÞ : (8)

It could be concluded from Eq. (8) that the level of phase

noise is determined by d under a given fiber length that deter-

mines the value of delay time s. On the basis of the above anal-

ysis, we derive the ultimate expression of d as

d ¼ G3s

ðGs � 1Þ� fkBTFþ 2qI0Rð1þ cos/bÞ þ RIN � R½I0ð1þ cos/bÞ�2g

I20Rðsin/bÞ2(9)

and the SSB phase noise is

LðfmÞ ¼ G3s

ðGs � 1Þ� fkBTFþ 2qI0Rð1þ cos/bÞ þ RIN � R½I0ð1þ cos/bÞ�2g

I20Rðsin/bÞ2 � ½2� 2 cosð2pfmsÞ�:

(10)

G3s

ðGs�1Þ achieves the minimum value in the case of Gs ¼ 1:5. Fix-ing Gs and Io, the value of d is determined only by the biasing

angle /b normalized by b ¼ /b/p, where b is the so-called nor-

malized biasing angle, and it means that we can optimize the

characteristics of the phase noise of oscillation signal utilizing

the biasing voltage of the modulator. Given that the RIN equals

to �170 dBc/Hz and the half-wave of the MZM is 1.8 V at 10

GHz, the relationship between the normalized bias angle b and

d at different optical power injection is shown in Figure 2. It is

evident that high I0 and small /b responding low d, which

reduces the SSB phase noise of the OEO.

As we know from above analysis, the open-loop gain, deter-

mined by Gs ¼ GP � GA, is fixed at 1.5 for the lowest d or

LðfmÞ, therefore, increasing Gp means to reduce GA, while

increasing GA means to reduce GP. In equation d ¼ qNG2A=Posc,

Posc is determined by the nonlinearity of MZM and we can

deem it as a constant, so how to decrease the values of qN and

GA becomes the key point for reducing the phase noise of the

oscillating signal.

At the same level of biasing angle, higher injection optical

power corresponding to greater GP can reduce the value of GA

enormously and improve qN relatively small; therefore, as

shown in Figure 2, d decreases with the injection optical

power. Fixing the biasing angle and increasing the optical

power, when the optical power arrives at 50 mW, the decrease

of d is saturated due to the equivalent variations of both qNand G2

A.

At the same level of optical carrier, It is evident from Figure

2 that the minimum d usually appears at low-bias point (0.5 <b < 1) of the MZM, that is because low-biasing technology of

the opto-electronic modulator could suppress the optical carrier,

decrease the photocurrent, and further reduce qN. On the other

side, low-biasing technology of the opto-electronic modulator

could also degrade the electro-optic conversion efficiency of the

MZM, thereby reducing GP and enhancing GA. Whether qN can

Figure 2 Noise to signal ratio d as a function of the normalized bias

angle b

690 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 54, No. 3, March 2012 DOI 10.1002/mop

be optimized is decided by the variations of both qN and G2A.

From above analysis, we can conclude that the minimum value

of qN appears at the specific point of b when the variations of

both qN and G2A are the same, and we can also find from Figure

2 that the specific b corresponding to the minimum qN increases

with the injection optical power.

3. EXPERIMENT AND DISCUSSION

Experiments based on the schematic shown in Figure 1 are per-

formed. The LD used in experiments is a distributed feedback semi-

conductor laser (Ortel 1772), and we operated it at the full rated

power of 70 mW. A variable optical attenuator is used to adjust the

input optical power of the MZM, placed between the LD and the

MZM. As high output of the LD make its RIN relatively lower,

varying optical output power through the VOA can not only change

the output optical power but also make its RIN a constant. The

electro-optic modulator in this experiment is a low-Vp MZM (Con-

vega LN058), and its Vp is about 1.8 V at 10 GHz. The PD is a

high-saturation (35 mW) and high-speed (28 GHz) Pin-photodiode

(Optilab PD-30). The AMP’s gain and NF are 50 dB and 1.8 dB,

respectively, and we can adjust the gain through the rear adjustable

attenuator. The �3 dB bandwidth of the MICROWAVE filter

(K&L 3C60) is 20 MHz and the center frequency is 10 GHz.

Figure 3 shows the measured data of the SSB phase noise in

the case that Po ¼ 10mW; b ¼ 0:1 and Po ¼ 60mW; b ¼ 0:8.The condition of Po ¼ 10mW; b ¼ 0:1, corresponding to low-

injection optical power and high biasing point, causes worse

SSB phase noise characteristics, while the condition of

Po ¼ 60mW; b ¼ 0:8 corresponding to high-injection optical

power and low biasing point makes better SSB phase noise

characteristics inversely. The experimental phenomena could

also be well explained by above theoretical model Eq. (9).

Figure 4 gives the relationship between the normalized bias

angle b and the SSB phase noise L(fm) at fm ¼ 1kHz and

fm ¼ 10kHz at the optical power Po ¼ 60mW. Both solid and

dotted lines represent the theoretical data and the both square

and triangular dots are experimental data of SSB phase noise at

fm ¼ 1kHz and fm ¼ 10kHz, respectively.

Experimental data are well consistent with the theoretical

expression of Eq. (10). The minimum value of L(fm) is about

�115 dBc/Hz @ 10 kHz and �95 dBc/Hz @ 1 kHz at b ¼0.8, which are 2.8 dB lower than them at b ¼ 0.5. With the

increase of b from 0 to 0.5, the modulation efficiency of MZM

increases and GP improves, so the value of G2A decreases

accordingly. On the other side, enhancing b corresponds to

smaller photocurrent which reduces the value of qN; therefore,the SSB phase noise reduces with b from 0 to 0.5. Although

the modulation efficiency of MZM decreases with b from 0.5

to 0.8, the decreasing amount of qN is more than the increasing

amount of G2A, which makes the SSB phase noise continue to

reduce. When the value of b is bigger than 0.8, the decreasing

amount of qN is less than the increasing amount of G2A, so the

SSB phase noise starts to increase and, therefore, its minimum

value appears at b ¼ 0.8.

From the above analysis, it is evident that we can reduce the

SSB phase noise effectively utilizing the low-biasing technology

of opto-electronic modulator.

4. CONCLUSION

We have analyzed the influences of the biasing voltage of MZM

on the phase noise of the output signal of OEO. A new model

for the OEO considering the biasing angle of the MZM has also

been built and verified by experiments. Based on this model, the

low-biasing method was proposed for optimizing the SSB phase

noise of the OEO. Results suggest that high optical power injec-

tion and low biasing voltage lead to the improvement of the

phase noise characteristics of OEO.

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VC 2012 Wiley Periodicals, Inc.

NEW UWB ANTENNA DESIGN FORWIRELESS COMMUNICATIONS

M. Bithikh,1 R. Aksas,1 H. Kimouche,2 and A. Azrar31National Polytechnic School, El-Harrach, Algiers, Algeria2Military Polytechnic School (EMP), Bordj El-Bahri BP17,Algiers, Algeria3 Institute of Electrical and Electronic Engineering, UniversityM’Hamed Bougara (UMBB), Boumerdes, Algeria; Correspondingauthor: take_azrar_2007@yahoo.com

Received 2 June 2011

ABSTRACT: The presented work in this article concerns a new ultra-

wideband antenna design with an octagonal configuration and reduceddimensions. Different prototypes have been simulated and implemented

in the form of monopoles. These prototypes have shown an admissibleagreement between simulations and measurements and ultra-widebands running from 3.1 to 14 GHz for wireless applications. VC 2012

Wiley Periodicals, Inc. Microwave Opt Technol Lett 54:692–697,

2012; View this article online at wileyonlinelibrary.com.

DOI 10.1002/mop.26666

Key words: antennas; ultra-wideband; microstrip access; wirelesscommunications

1. INTRODUCTION

Nowadays, the potential of mobility, highly speed and quality

information access and sharing between portable devices has

grown with an extraordinary speed. Moreover, due to the rise of

computer networks, various electronic devices and mobile

phones, wireless applications have recently seen a tremendous

explosion.

It should be noted that the frequencies used by these applica-

tions are spread over several octaves and simultaneous access

from the same small and compact terminal to these frequencies

is not possible with conventional antennas.

Therefore, these various set up communication systems will

need increasingly compact and discrete antennas, functioning on

one or more frequencies, in particular allowing to ensure the

compatibility of different standards or to reach many services

starting from the same apparatus. To overcome these problems,

some authors have proposed compact antennas which generally

use certain geometrical structures. Thus technology is directed

towards new types of antennas which ensure the functionalities

for several applications at the same time, called antennas multi

and/or wide bands. The latter are the subject of many research

and development; especially, printed types whose form and

dimensions enable them to be integrated in the modules of trans-

mission or reception on the same substrate as well as their low

cost and also they are not bulky.

However, one of these antennas’ limiting factors is their nar-

row bandwidth. Hence, several configurations have been con-

ducted to increase the bandwidth of the planar antennas such as

the antennas ultra-wideband of rectangular shape [1–3], triangle

[4], circular [5, 6], elliptical [7], and several other antenna forms

[8–11].

The objective of this work is to design a microstrip antenna

wide in bandwidth capable to operate within the band [3.1–10.6]

GHz. Therefore, in this article, a new octagonal ultra-wideband

antenna [12–14] will be presented with some variations and

modifications and the obtained simulations and measurements

results are presented and discussed.

2. THE PROPOSED ANTENNA GEOMETRY

The proposed antenna as shown in Figure 1 is an octagonal pla-

nar antenna whose dimensions are given as: L ¼ 30 mm, L1 ¼10.75 mm, L2 ¼ 11.25 mm, L3 ¼ 15 mm, L4 ¼ 26.25 mm, W ¼30 mm, W1 ¼ 7.5 mm, W2 ¼ 15 mm, W3 ¼ 9 mm. The antenna

is fed by a microstrip line with g ¼ 3 mm and it is printed on a

dielectric substrate of an epoxy glass type with a relative permit-

tivity of er ¼ 4.4 and a thickness h ¼ 1.6 mm and a rectangular

ground plan printed on the other surface of the substrate.

Figure 1 Original form prototype (octagonal)

692 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 54, No. 3, March 2012 DOI 10.1002/mop