EE247 Lecture 7

EECS 247 Lecture 7: Filters © 2007 H.K.

EE247 Lecture 7

• Automatic on-chip filter tuning (continued from last lecture)– Continuous tuning

• Reference integrator locked to a reference frequency• DC tuning of resistive timing element

– Periodic digitally assisted filter tuning• Systems where filter is followed by ADC & DSP, existing hardware

can be used to periodically update filter freq. response• Continuous-time filters

– Highpass filters– Bandpass filters

• Lowpass to bandpass transformation• Example: 6th order bandpass filter• Gm-C BP filter using simple diff. pair

EECS 247 Lecture 7: Filters © 2007 H.K. Page 2

Summary last lecture• Continuous-time filters

– Opamp MOSFET-RC filters– Gm-C filters

• Frequency tuning for continuous-time filters– Trimming via fuses or laser – Automatic on-chip filter tuning

• Continuous tuning– Utilizing VCF built with replica integrators– Use of VCO built with replica integrators– Reference integrator locked to reference frequency


SummaryReference Integrator Locked to Reference Frequency

Feedback forces Gm to vary so that :

S2 S3Gm

C1 C2

Vref

A

int g

int g0

C1 N / fclkGmor

Gm fclk / NC1

τ

ω

= =

= =

Tuning error due to gm-cell offset voltage resolved

Advantage over previous schemes:

fclk can be chosen to be at much higher frequencies compared to filter bandwidth (N >1)

Feedthrough of clock attenuated by filter


DC Tuning of Resistive Timing Element

Vtune Tuning circuit Gm replica of Gm used in filter

Rext used to lock Gm to accurate off-chip R

Feedback forces Gm=1/Rext

Issues with DC offset

Account for capacitor variations in this gm-C implementation by trimming

Rext.

-

+ -

+

I

I

Gm

Ref: C. Laber and P.R. Gray, “A 20MHz 6th Order BiCMOS Parasitic Insensitive Continuous-time Filter and Second Order Equalizer Optimized for Disk Drive Read Channels,” IEEE Journal of Solid State Circuits, Vol. 28, pp. 462-470, April 1993


Digitally Assisted Frequency Tuning Example:Wireless Receiver Baseband Filters

• Systems where filter is followed by ADC & DSP– Take advantage of existing digital signal processor capabilities to

periodically update the filter critical frequency

– Filter tuned only at the outset of each data transmission session (off-line/periodic tuning) – can be fine tuned during times data is not transmitted

RF Amp

Osc.

A/D Digital Signal

Processor (DSP)

A/D

π 2IF Stage ( 0 to 2 )


Example: Seventh Order Tunable Low-Pass OpAmp-RC Filter


Digitally Assisted Filter Tuning Concept

Assumptions:– System allows a period of

time for the filter to undergo tuning (e.g. for a wireless transceiver during idle periods)

– An AC (e.g. a sinusoid) signal can be generated on-chip whose amplitude is a function of an on-chip DC source

• AC signal generator outputs a sinusoid with peak voltage equal to the DC signal source

• AC Signal Power =1/2 DC signal power @ the input of the filter

VP AC=VDC



VP AC=VDC

AC signal @ a frequency on the roll-off of the desired filter frequency response(e.g. -3dB frequency)

Provision can be made during the tuning cycle, the input of the filter is disconnected from the previous stage (e.g. mixer) and connected to:

1. DC source2. AC source

under the control of the DSP

( )desiredAC DC 3dBV V sin 2 f tπ −= ×



VP AC=VDC


2ΔΔ

Practical Implementation of Frequency TuningAC Signal Generation From DC Source

Vout

Clock

ClockB

Vout0+Δ

−Δ

Δ Vout=

Clock=high

+Δ Δ Vout= −Δ

ClockB=high


Δ 2ΔΔ

DC Measurement AC Measurement

A/D 4bit

10MHz

Digital Signal

ProcessorDSP1616

40MHzVref+Vref-

Filter

Register

CH

OP

TUN

E

FREQ

.C

ON

T.

625k

Hz

Practical Implementation of Frequency Tuning


2ΔΔAC

Measurement

Practical Implementation of Frequency TuningEffect of Using a Square Waveform

( ) ( )n 1,3,5,..

4Vin sin n tnt π ω=

∞ Δ= ∑

• Input signal chosen to be a square wave due to ease of generation• Filter input signal comprises a sinusoidal waveform @ the fundamental

frequency + its odd harmonics:

Key Point: The filter itself attenuates unwanted odd harmonics Inaccuracy incurred by the harmonics negligible

( ) ( )4 1Vout sin t2

t π ωΔ= ×


Simplified Frequency Tuning Flowchart


Digitally Assisted Offset Compensation


Filter Tuning Prototype Diagram



Chip Photo


Measured Tuning Characteristics


Off-line Digitally Assisted Tuning• Advantages:

– No reference signal feedthrough since tuning does not take place during data transmission (off-line)

– Minimal additional hardware– Small amount of programming

• Disadvantages:– If acute temperature change during data transmission,

filter may slip out of tune!• Can add fine tuning cycles during periods of data is not

transmitted or received

Ref: H. Khorramabadi, M. Tarsia and N.Woo, “Baseband Filters for IS-95 CDMA Receiver Applications Featuring Digital Automatic Frequency Tuning,” 1996 International Solid State Circuits Conference, pp. 172-173.


Summary: Continuous-Time Filter Frequency Tuning

• Trimming• Expensive & does not account for temperature and supply etc… variations

• Automatic frequency tuning– Continuous tuning

• Master VCF used in tuning loop– Tuning quite accurate– Issue reference signal feedthrough to the filter output

• Master VCO used in tuning loop– Design of reliable & stable VCO challenging– Issue reference signal feedthrough

• Single integrator in negative feedback loop forces time-constant to be a function of accurate clock frequency

– More flexibility in choice of reference frequency less feedthrough issues• DC locking of a replica of the integrator to an external resistor

– DC offset issues & does not account for integrating capacitor variations– Periodic digitally assisted tuning

– Requires digital capability + minimal additional hardware– Advantage of no reference signal feedthrough since tuning performed off-line


Integrator Based High-Pass Filters1st Order

• Conversion of simple high-pass RC filter to integrator-based type by using signal flowgraph technique

in

s CV Ros CV 1 R

=+

oV

R

C

inV


1st Order Integrator Based High-Pass FilterSGF

oV

R

C

inV+ VC - +

VR-

IC

IRV V VR in C1V IC C sC

V Vo R1I VR R R

I IC R

= −

= ×

=

= ×

=

1

1R

1sC

RICI

CV

inV

1−1

SFG

oV1VR


1st Order Integrator Based High-Pass FilterSGF

1sC R

−

oVinV 1 1oV

R

C

inV

oVinV

∫ -

SGF

Note: Addition of an integrator in the feedback path High pass frequency shaping

+

+

+ VC- + VR

-


Addition of Integrator in Feedback Path

oVinV

∫ -

a

1/sτ

Let us assume flat gain in forward path (a)Effect of addition of an integrator in the feedback path:

+

+

in

in

int gpole o

V aoV 1 af

sV aos sV 1 a / 1 / a

azero@ DC & pole @ a

ττ τ

ω ωτ

=+

= =+ +

→ = − = − ×

Note: For large forward path gain, a, can implement high pass function with low corner frequency without the need for large value of C or RAddition of an integrator in the feedback path zero @ DC + pole @ axω0

intg

This technique used for offset cancellation in systems where the low frequency content is not important and thus disposable


( )H jω

( )H jω

Lowpass Highpass

ω

( )H jω

ωω

Q<5

Q>5

• Bandpass filters two cases:1- Low Q (Q < 5)

Combination of lowpass & highpass

2- High Q or narrow-band (Q > 5)Direct implementation

ω

( )H jω

+

Bandpass Filters

Bandpass

Bandpass


Narrow-Band Bandpass FiltersDirect Implementation

• Narrow-band BP filters Design based on lowpass prototype• Same tables used for LPFs are also used for BPFs

Lowpass Freq. Mask Bandpass Freq. Mask

cc

s s2 s1c B2 B1

ss Qs

ωω

Ω Ω − ΩΩ Ω − Ω

⎡ ⎤× +⎢ ⎥⎣ ⎦

⇒ ⇒


Lowpass to Bandpass TransformationLowpass pole/zero (s-plane) Bandpass pole/zero (s-plane)

From: Zverev, Handbook of filter synthesis, Wiley, 1967- p.156.

PoleZero


Lowpass to Bandpass Transformation Table

From: Zverev, Handbook of filter synthesis, Wiley, 1967- p.157.

'

'

'

'

1

1

1 1

r r

r

r

r

r

r r

C QCRRL

QC

RL QL

CRQC

ω

ω

ω

ω

= ×

= ×

= ×

= ×

C

L

C’

LP BP BP Values

L CL’

Lowpass filter structures & tables used to derive bandpass filters

' 'C &L are normilzed LP values

filterQ Q=


Lowpass to Bandpass TransformationExample: 3rd Order LPF 6th Order BPF

• Each capacitor replaced by parallel L& C• Each inductor replaced by series L&C

oVL2 C2

RsC1

C3inV RLL1 L3

RsC1’ C3’

L2’

inV RL

oV

Lowpass Bandpass


Lowpass to Bandpass TransformationExample: 3rd Order LPF 6th Order BPF

'1 1

0

1 '01

2 '02

'2 2

0

'3 3

0

3 '03

1

1

1 1

1

1

C QCRRL

QC

CRQLRL QL

C QCRRL

QC

ω

ω

ω

ω

ω

ω

= ×

= ×

= ×

= ×

= ×

= ×

oVL2 C2

RsC1

C3inV RLL1 L3

Where:C1

’ , L2’ , C3

’ Normalized lowpass valuesQ Bandpass filter quality factor ω0 Filter center frequency


Lowpass to Bandpass TransformationSignal Flowgraph

oVL2 C2

RsC1

C3inV RLL1 L3

1- Voltages & currents named for all components2- Use KCL & KVL to derive state space description 3- To have BMFs in the integrator form

Cap voltage expressed as function of its current VC=f(IC)

Ind current as a function of its voltage IL=f(VL)4-Use state space description to draw SFG5- Convert all current nodes to voltage


Signal Flowgraph6th Order Bandpass Filter

1

*RRs

− *1

1sC R

1−

*

1

RsL

−

1−

1

*RRL

−*3

1sC R

*

3

RsL

−*

2

1sC R

−*

2

RsL

1−

Note: each C & L in the original lowpass prototype replaced by a resonatorSubstituting the bandpass L1, C1,….. by their normalized lowpass equivalent from page 29The resulting SFG is:

1

V1’

V2

V3’

V1

V2’

VoutVinV3


Signal Flowgraph6th Order BPF versus 3rd Order LPF

1−

*RRs

−*

1

1sC R

1

*RRs

− *1

1sC R

1−

*

1

RsL

−

1−

1

*RRL

−*

3

1sC R

*

3

RsL

−*

2

1sC R

−*

2

RsL

1

V1’

V2

V3’

V1

V2’

VoutVinV3

inV 1 1V oV1−1

1− 1V1’ V3’V2’

*

2

RsL

*RRL

−

V2

*3

1sC R

LPF

BPF



1

*RRs

− 0

1'QCs

ω

1−

'1 0QC

sω

−

1−

1

*RRL

−'3

0

Q Csω'

3 0Q C

sω

−2 0

'QL

sω

−

0

2'QLs

ω

1−

• Note the integrators different time constants• Ratio of time constants for two integrator in each resonator ~ Q2

Typically, requires high component ratiosPoor matching

• Desirable to modify SFG so that all integrators have equal time constants for optimum matching.

• To obtain equal integrator time constant use node scaling

1

V1’

V2

V3’

V1

V2’

VoutVin V3



'1

1QC

−

'2

1QL

*

'1

R 1Rs QC

− ×

0s

ω

1−

0s

ω−

'2

1QL

−

'3

1QC

*

3

R 1RL QC

− ×

0s

ω0s

ω−

0s

ω−0

sω

• All integrator time-constants equal• To simplify implementation choose RL=Rs=R*

1

V1’/(QC1’)

V2 /(QL2’)

V3’/(QC3’)

V1 V3

V2’

VinVout



'2

1QL

'1

1QC

− 0s

ω

1−

0s

ω−

'2

1QL

−

'3

1QC

'3

1

QC−0

sω

0s

ω−

0s

ω−0

sω

'1

1QC

−

Let us try to build this bandpass filter using the simple Gm-C structure

1VinVout


Second Order Gm-C FilterUsing Simple Source-Couple Pair Gm-Cell

• Center frequency:

• Q function of:

Use this structure for the 1st and the 3rd resonatorUse similar structure w/o M3, M4 for the 2nd resonatorHow to couple the resonators?

M1,2m

oint g

M1,2mM 3,4m

g2 C

gQg

ω = ×

=


Coupling of the Resonators1- Additional Set of Input Devices

Coupling of resonators:Use additional input source coupled pairs for the highlighted integrators For example, the middle integrator requires 3 sets of inputs

'2

1QL

'1

1QC

− 0s

ω

1−

0s

ω−

'2

1QL

−

'3

1QC

'3

1

QC−0

sω

0s

ω−

0s

ω−0

sω

'1

1QC

−

1VinVout


Example: Coupling of the Resonators1- Additional Set of Input Devices

int gC

Add one source couple pair for each additional input

Coupling level ratio of device widths

Disadvantage extra power dissipation

oV

maininV

+-

+

-

M1 M2M3 M4

-

+

couplinginV

+

--+

+

-

MainInput

CouplingInput


Coupling of the Resonators2- Modify SFG Bidirectional Coupling Paths

' '1 2

1Q C L

'1

1QC

− 0s

ω

inV 1−

0s

ω−

' '3 2

1Q C L

−

'1

' '3 2

CQC L

3

1QC'

−0s

ω0s

ω−

0s

ω−0

sω

1' 'Q C L1 2

−

Modified signal flowgraph to have equal coupling between resonators• In most filter cases C1

’ = C3’• Example: For a butterworth lowpass filter C1’ = C3’ =1 & L2’=2• Assume desired overall bandpass filter Q=10

outV1


Sixth Order Bandpass Filter Signal Flowgraph

γ

1Q

− 0s

ω

inV 1−

0s

ω−

1Q

−0s

ω0s

ω−

0s

ω−0

sω

outV1γ−

γγ−

1Q 2114

γ

γ

=

≈

• Where for a Butterworth shape

• Since Q=10 then:


Sixth Order Bandpass Filter Signal FlowgraphSFG Modification

1Q

−0s

ω

inV 1−

0s

ω−

1Q

−0s

ω0s

ω−0

sω

−0s

ω

outV1

γ−

20s

ωγ ⎛ ⎞⎜ ⎟⎝ ⎠

×

γ−

20s

ωγ ⎛ ⎞⎜ ⎟⎝ ⎠

×



20 1

ωω

⎛ ⎞ ≈⎜ ⎟⎝ ⎠

For narrow band filters (high Q) where frequencies within the passband are close to ω0 narrow-band approximation can be used:

Within filter passband:

The resulting SFG:

2200

js

ωωω

γ γ γ⎛ ⎞⎛ ⎞ = ≈⎜ ⎟⎜ ⎟⎝ ⎠ ⎝ ⎠

× × −



1Q

−0s

ω

inV 1−

0s

ω−

1Q

−0s

ω0s

ω−0

sω

−0s

ω

outV1

γ−

γ−

γ−

Bidirectional coupling paths, can easily be implemented with coupling capacitors no extra power dissipation

γ−


Sixth Order Gm-C Bandpass FilterUtilizing Simple Source-Coupled Pair Gm-Cell

Parasitic cap. at integrator output, if unaccounted for, will result in inaccuracy in γ

k

int g k

int gk

k int g

C2 C C

2 CC 1 1

2C C13

1 / 14

γ

γγ

+=

×

×=

−

→ =

=


Sixth Order Gm-C Bandpass FilterNarrow-Band versus Exact

Frequency Response Simulation

Q=10

Regular Filter

Response

Narrow-Band Approximation


Simplest Form of CMOS Gm-CellNonidealities

• DC gain (integrator Q)

• Where a denotes DC gain & θ is related to channel length modulation by:

• Seems no extra poles!

( )

M 1,2m

M 1,20 load

M 1,2

gag g

2LaV Vgs th

L

θ

θλ

=+

=−

=

Small Signal Differential Mode Half-Circuit


CMOS Gm-Cell High-Frequency Poles

• Distributed nature of gate capacitance & channel resistance results in infinite no. of high-frequency poles

Cross section view of a MOS transistor operating in saturation

Distributed channel resistance & gate capacitance


CMOS Gm-Cell High-Frequency Poles

• Distributed nature of gate capacitance & channel resistance results in an effective pole at 2.5 times input device cut-off frequency

High frequency behavior of an MOS transistor operating in saturation region

( )

M 1,2

M 1,2

effective2

i 2 i

effectivet2

M 1,2M 1,2m

t 2

1P1

P

P 2.5

V Vgs thg 3C2 / 3 WL 2 Lox

μ

ω

ω

∞

=

≈

≈

−= =

∑


Simple Gm-Cell Quality Factor

( )M 1,2effective2 2

V Vgs th15P4 L

μ −=( )M 1,2

2LaV Vgs thθ

=−

• Note that phase lead associated with DC gain is inversely prop. to L• Phase lag due to high-freq. poles directly prop. to L

For a given ωο there exists an optimum L which cancel the lead/lag phase error resulting in high integrator Q

( )( )

i1 1o pi 2

2M1,2 o

M1,2

int g. 1real

V Vgs th L1 4int g. 2L 15 V Vgs th

Q

a

Q

ω

θ ωμ

∞

=

≈−

−≈ −

−

∑


Simple Gm-Cell Channel Length for Optimum Integrator Quality Factor

( )1/ 32

M1,2

o

V Vgs th. 15opt. 4Lθμ

ω

⎡ ⎤−⎢ ⎥≈ ⎢ ⎥⎢ ⎥⎣ ⎦

• Optimum channel length computed based on process parameters (could vary from process to process)


Source-Coupled Pair CMOS Gm-Cell Transconductance

( ) ( )

( )

1/ 22i i

d ssM1,2 M1,2

i M 1,M 2dm

iM 1,2

di

i

v v1I I 1V V V V4gs th gs th

v INote : For small gV V vgs thINote : As v increases or the v

ef fect ive transconductance decreases

⎧ ⎫Δ⎡ ⎤ Δ⎡ ⎤⎪ ⎪Δ = ⎢ ⎥ − ⎢ ⎥⎨ ⎬− −⎢ ⎥ ⎢ ⎥⎪ ⎪⎣ ⎦ ⎣ ⎦⎩ ⎭

Δ⎡ ⎤ Δ→ =⎢ ⎥− Δ⎢ ⎥⎣ ⎦ΔΔ Δ

For a source-coupled pair the differential output current (ΔId)as a function of the input voltage(Δvi):

i i1 i2

d d1 d2

v V V

I I I

Δ = −

Δ = −


Source-Coupled Pair CMOS Gm-Cell Linearity

Ideal Gm=gm

• Large signal Gm drops as input voltage increasesGives rise to nonlinearity


Measure of Linearity

ω1 ω1 3ω1 ωω

2ω1−ω2 2ω2−ω1

Vin Voutω1 ωω2 ω1 ωω2

Vin Vout2 31 2 3

23

1

3

2 43 5

1 1

.............

3 . .3

1 ......4

3 .

3 25 ......4 8

Vout Vin Vin Vin

amplitude rd harmonicdist compHDamplitude fundamental

Vin

amplitude rd order IM compIMamplitude fundamental

Vin Vin

α α α

αα

α αα α

= + + +

=

= +

=

= + +


Source-Coupled Pair Gm-Cell Linearity

( ) ( )

( )

( )

( )

1/ 22i i

d ssM 1,2 M1,2

2 3d 1 i 2 i 3 i

ss1 2

M1,2

ss3 43

M 1,2

ss5

v v1I I (1)1V V V V4gs th gs th

I a v a v a v . . . . . . . . . . . . .

Series expansion used in (1)Ia & a 0

V Vgs thIa & a 0

8 V Vgs thIa

128 V Vgs th

⎧ ⎫Δ⎡ ⎤ Δ⎡ ⎤⎪ ⎪Δ = ⎢ ⎥ − ⎢ ⎥⎨ ⎬− −⎢ ⎥ ⎢ ⎥⎪ ⎪⎣ ⎦ ⎣ ⎦⎩ ⎭

Δ = ×Δ + ×Δ + ×Δ +

= =−

= − =−

= −−

65

M 1,2

& a 0=


Linearity of the Source-Coupled Pair CMOS Gm-Cell

• Key point: Max. signal handling capability function of gate-overdrive voltage

( ) ( )

( )

( )

2 43 5i i

1 1

1 32 4

i i

GS th GS th

i max GS th

rms3 GS th in

3a 25aˆ ˆIM 3 v v . . . . . . . . . . . .4a 8aSubst i tu t ing for a ,a ,. . . .

ˆ ˆv v3 25IM 3 . . . . . . . . . . . .32 1024V V V V

2v̂ 4 V V IM 33

ˆIM 1% & V V 1V V 230mV

≈ +

⎛ ⎞ ⎛ ⎞≈ +⎜ ⎟ ⎜ ⎟− −⎝ ⎠ ⎝ ⎠

≈ − × ×

= − = ⇒ ≈


Simplest Form of CMOS Gm CellDisadvantages

( )

( )( )

since

then

23 GS th

M 1,2m

oint g

o

IM V V

g2 C

W V VCg gs thm ox LV Vgs th

ω

μ

ω

−∝ −

=×

−=

−∝

•Max. signal handling capability function of gate-overdrive

•Critical freq. is also a function of gate-overdrive

Filter tuning affects max. signal handling capability!


Simplest Form of CMOS Gm CellRemoving Dependence of Maximum Signal Handling

Capability on Tuning

Dynamic range dependence on tuning removed (to 1st order)Ref: R.Castello ,I.Bietti, F. Svelto , “High-Frequency Analog Filters in Deep Submicron Technology ,

“International Solid State Circuits Conference, pp 74-75, 1999.

• Can overcome problem of max. signal handling capability being a function of tuning by providing tuning through :

– Coarse tuning via switching in/out binary-weighted cross-coupled pairs Try to keep gate overdrive voltage constant

– Fine tuning through varying current sources


Dynamic Range for Source-Coupled Pair Based Filter

( )3 1% & 1 230rmsGS th inIM V V V V mV= − = ⇒ ≈

• Minimum detectable signal determined by total noise voltage• It can be shown for the 6th order Butterworth bandpass filter

fundamental noise contribution is given by:

2o

int g

int grmsnoise

rmsmax

36

k Tv Q C

Assumin g Q 10 C 5pF

v 160 Vsince v 230mV

230x10Dynamic Range 20log 63dB160x10

3

μ

−−

≈

= =

≈=

= ≈


Improving the Max. Signal Handling Capability of the Source-Coupled Pair Gm-Cell

( )( )

( )( )

M 1,2ss1ss3

M 3,4

M 1,22

M 3,4

V Vgs thI b & a and thusI V Vgs th

WL b

W aL

−= =

−

=

• 2nd source-coupled pair added to subtract current proportional to nonlinear component associated with the main SCP


Improving the Max. Signal Handling Capability of the Source-Coupled Pair Gm

Ref: H. Khorramabadi, "High-Frequency CMOS Continuous-Time Filters," U. C. Berkeley, Department of Electrical Engineering, Ph.D. Thesis, February 1985 (ERL Memorandom No. UCB/ERL M85/19).


Improving the Max. Signal Handling Capability of the Source-Coupled Pair Gm

• Improves maximum signal handling capability by about 12dBDynamic range theoretically improved to 63+12=75dB


Simplest Form of CMOS Gm-Cell

• Pros– Capable of very high frequency

performance (highest?)– Simple design

• Cons– Tuning affects power dissipation– Tuning affects max. signal handling

capability (can overcome)– Limited linearity (possible to

improve)

Ref: H. Khorramabadi and P.R. Gray, “High Frequency CMOS continuous-time filters,” IEEE Journal of Solid-State Circuits, Vol.-SC-19, No. 6, pp.939-948, Dec. 1984.


Gm-CellSource-Coupled Pair with Degeneration

( )

( )dsV small

eff

M 3 M 1,2mds

M 1,2 M 3m ds

M 3eff ds

C Wox 2V VI 2 V Vgs thd ds ds2 L

I Wd V VCg gs thds oxV Lds1g 1 2

g g

for g g

g g

μ

μ

⎡ ⎤−= −⎣ ⎦

∂ −= ≈∂

=+

>>

≈

M3 operating in triode mode source degeneration determines overall gmProvides tuning through varing Vc


Gm-CellSource-Coupled Pair with Degeneration

•Pros– Moderate linearity– Continuous tuning provided

by Vc– Tuning does not affect power

dissipation

•Cons– Extra poles associated

with the source of M1,2 Low frequency

applications only

Ref: Y. Tsividis, Z. Czarnul and S.C. Fang, “MOS transconductors and integrators with high linearity,”Electronics Letters, vol. 22, pp. 245-246, Feb. 27, 1986


BiCMOS Gm-CellExample

• MOSFET in triode mode:

• Note that if Vds is kept constant:

• Linearity performance keep gm constantfunction of how constant Vds can be held

– Need to minimize Gain @ Node X

• Since for a given current, gm of BJT is larger compared to MOS- preferable to use BJT

• Extra pole at node X

( )

M 1m

C Wox 2V VI 2 V Vgs thd ds ds2 L

I Wd Cg Vox dsV Lgs

M 1 B1A g gx m m

μ

μ

⎡ ⎤−= −⎣ ⎦

∂= =

∂

=

B1

M1X

Iout

Is

Vcm+Vin

Vb

gm can be varied by changing Vb and thus Vds


Alternative Fully CMOS Gm-CellExample

• BJT replaced by a MOS transistor with boosted gm

• Lower frequency of operation compared to the BiCMOS version due to more parasitic capacitance at nodes A & B A B

+- +

-


• Differential- needs common-mode feedback ckt

• Freq.tuned by varying Vb

• Design tradeoffs:– Extra poles at the input device drain

junctions– Input devices have to be small to

minimize parasitic poles– Results in high input-referred offset

voltage could drive ckt into non-linear region

– Small devices high 1/f noise

BiCMOS Gm-C Integrator

-Vout+

Cintg/2

Cintg/2

EE247 Lecture 7

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Transcript of EE247 Lecture 7