Design and Characterization of a 6 x 6 PlanarReconfigurable Transmitarray

Jonathan Y. Lau, Sean V. HumThe Edward S. Rogers Sr. Dept. of Electrical and Computer Engineering, University of Toronto

10 King’s College Road, Toronto, Canada, M5S 3G4{jlau, svhum}@waves.utoronto.ca

Abstract—This paper presents the design and experimentalcharacterization of a planar 6 x 6 fully reconfigurable trans-mitarray at 5.7 GHz. First, the design of the array element,which consists of two varactor diode-loaded patches coupledby a varactor diode-loaded slot, is presented. The design isoptimized by full-wave simulation, followed by experimentalcharacterization in waveguide. The element exhibits the behaviorof a third-order tunable filter and achieves 245 degrees of phaseagility with less than 3 dB of variation in transmission magnitudein the tuning range. Next, the design and fabrication of aplanar array is described, and the varactor biasing scheme isdiscussed in detail. Finally, two-dimensional beamforming resultsare presented. Not only is this transmitarray able to demonstratefull two-dimensional reconfigurability and beamforming as anarray lens, it is also low-profile, low-cost, and easy to fabricate,making it very attractive for applications where high-gain beam-scanning is needed.

I. INTRODUCTION

A transmitarray, also known as an array lens, is a low-costparadigm for implementing a high-directivity reconfigurableaperture. The configuration of the transmitarray and feed hornis shown in Figure 1. Ideally, each transmitarray elementprovides 360◦ of phase tunability with no variation in thetransmission magnitude as the insertion phase is varied.

Fig. 1. A reconfigurable transmitarray configuration

The design of microwave transmitarrays is closely relatedto lens array techniques, often used for millimeter-wave appli-cations. Fixed lens arrays have been studied for many years,and have found applications in spatial power amplification [1],band-pass filter lens arrays [2], and transmitarray antennas [3].

It is only more recently, with advances in MEMS andmaterial technologies, that tunable lens arrays have been

proposed and demonstrated. A tunable liquid crystal frequencyselective surface (FSS) was presented in [4], and a varactor-based tunable FSS was presented in [5]. While these lensarrays are reconfigurable, in those investigations it is theoperating frequency of the entire lens that was tuned. Thatis, the elements of the arrays cannot be individually tuned, sothe surfaces are not suitable for beamforming applications astransmitarrays or lens arrays.

In [6], a reconfigurable lens array was proposed for thepurpose of antenna beamforming. In that design, MEMSswitches were used to manipulate the pole/zero response ofthe filter by path switching. With 2-bit control, each arrayelement was capable of reconfiguring to four discrete phases.With each row of elements individually controllable, one-dimensional beamforming was demonstrated experimentally atmillimeter wavelengths.

This work aims to experimentally demonstrate a fullyreconfigurable microwave transmitarray at 5.7 GHz that islow-cost, easy to fabricate, and can perform two-dimensionalbeamforming. The array element design is also low-profile,with a thickness of one-tenth of a wavelength, which issignificantly thinner than other multi-pole designs that areseveral wavelengths in thickness [7]. Furthermore, by usingvaractor diodes that provide a continuous variable capacitance,the phase of individual elements can be accurately tunedwithout the losses and side-lobe levels associated with phasequantization.

In this paper, we first present the design of a single arrayelement in Section II. Section III discusses the details of thedesign and fabrication of a 6×6 planar transmitarray. Finally,experimental results for beamforming tests are presented inSection IV.

II. ELEMENT DESIGN

The array element, based on the design first proposed in[8], consists of two patches on either side of a ground plane,coupled by a slot in the ground plane. Each patch is loadedwith two varactor diodes, and another varactor diode is alsoplaced across the slot in the ground plane. Together, the twopatches and the loaded slot form three tunable resonators,which can theoretically yield 360◦ of phase tunability withonly small variations in the transmission magnitude. Thisdesign, shown in Figure 2, has the advantage that it only

TABLE IELEMENT DIMENSIONS AND PARAMETERS

Parameter ValuePatch size 13.0 mm × 13.0 mmSubstrate Rogers Duroid 6002 (εr = 2.94, h = 3.0 mm)Slot size 5.0 mm × 2.0 mmSlot position Centered with respect to the patchBonding film Rogers 3001 bonding film (h = 0.04 mm)Varactor diodes Aeroflex GaAs MGV100-20

uses two dielectric layers, and does not require any inter-layer connections or vias, greatly reducing cost and fabricationcomplexity. The element was optimized using the SEMCAD-X FDTD simulation platform. The full-wave simulations in-cluded both dielectric losses and lumped component parasiticsassociated with the varactor diodes. Element dimensions andparameters are summarized in Table I.

Fig. 2. A reconfigurable transmitarray element

With the optimized design, a unit cell was then fabricatedon 3.0 mm Rogers 6002 Duroid with Aeroflex MetelicsMGV100-20 varactor diodes, and experimentally tested in aWR-187 rectangular waveguide. By image theory, waveguidetesting of a single element simulates an infinite array of suchelements. We can define two TE10 ports on either side ofthe element, and The S-parameters through the waveguideand test element reveal the behavior of plane wave incidence,reflection, and transmission on an infinite array, with an angleof incidence of 35◦. The experimental waveguide test setup isshown in Figure 3.

The element was populated with varactor diodes whichwere biased using a custom USB voltage controller. Thebias voltages of the patch varactors were controlled together,and the slot varactor was biased separately, resulting in twodegrees of freedom in the control of the element. Thus, eachconfiguration consisted of a particular patch and slot biasvoltage.

The S-parameters of the element were measured for a widerange of configurations, where both the patch and slot biases

Fig. 3. Waveguide measurement setup

were swept logarithmically from 1 V to 20 V in 130 steps.The magnitude and phase surfaces that were generated fromthe two-dimensional sweep are shown in Figures 4 and 5. Now,for each desired element transmission phase � S21 between 0◦

and 360◦, a number of configurations may have generatedthat particular element phase. So, for each desired phase,an optimal configuration can be found that maximizes thetransmission magnitude |S21|. The loci of the optimal voltageconfigurations are also shown in Figures 4 and 5. We note thatwhile the optimal locus is not linear with respect to the biasvoltages, the region over which � S21 varies the greatest, orthe majority of the phase tuning range, is almost linear.

0

10

20

0

10

20−60

−40

−20

0

Patch Varactor Bias [V]Slot Varactor Bias [V]

|S21

| [dB

]

Locus of Optimal Configurations

Fig. 4. Waveguide measured S21 (magnitude)

The transmission response of the element from the waveg-uide experiments using the optimal configurations are shown inFigure 6. For a maximum variation of |S21| less than 3 dB, theelement achieves a phase tuning range of 260◦ in simulationand 245◦ in experiment. The full 360◦ range is not achieveddue to the 2 Ω parasitic series resistance in the varactor diodes.However, while the phase range is less than the desired 360◦,we can consider allowing more amplitude variation since ithas less effect on the beam-steering capabilities of the arraythan phase variation. For a maximum variation of |S21| lessthan 10 dB over the tuning range, the element achieves 330◦

of phase tunability.

0

10

20

010

20−200

0

200

400

Patch Varactor Bias [V]Slot Varactor Bias [V]

∠ S

21 [D

eg]

Locus of Optimal Configurations

Fig. 5. Waveguide measured S21 (angle)

0 50 100 150 200 250 300 350 400 450−15

−10

−5

0

Configuration Number

Mag

nitu

de [d

B]

|S21

| (EXP)

|S21

| (SIM)

|S11

| (EXP)

|S11

| (SIM)

0 50 100 150 200 250 300 350 400 4500

100

200

300

Configuration Number

Pha

se [D

eg]

∠ S21

(EXP)

∠ S21

(SIM)

Fig. 6. Transmission characteristics of a single element

III. ARRAY DESIGN AND FABRICATION

Using this transmitarray element, a 6× 6 planar transmitar-ray was designed and experimentally tested. The size of 6×6was selected so that it could be easily and cost-effectivelyfabricated, but sufficiently large so that the directivity was highenough to be measured by a planar near-field scanner.

A. Patch Biasing

An inherent challenge to the bias network design of atransmitarray is that since waves are not guided by trans-mission lines, biasing mechanisms are fully exposed to thewaves travelling through the structure. Simulations showedthat resonances in non-resistive bias lines were easily excited,leading to undesired radiation, coupling, and scattering. Asone would expect, it was observed that horizontal (magnetic-field directed) bias lines were not excited and did not perturbthe fields in the structure if they were sufficiently far awayfrom the patch. On the other hand, resonances were easilyexcited in the vertical (electric-field directed) bias lines. Thus,the two patches of each element were biased with conductinglines in the manner shown in Figure 7, where short verticalsegments were connected with resistors. It was determinedthrough simulations that if the resistor values were too small,

the segments were not sufficiently isolated, leading to all ofthe bias lines being excited. On the other hand, if the resistorvalues were too large, power was reflected away from theresistors, leading to a build-up of power in the small segments.Ideally, all power coupled into the bias lines is dissipated bythe resistors.

An alternative solution was to use resistive material forthe bias lines. However, such an approach would introduceadditional complexity into the fabrication process, and a maingoal of this design was to minimize complexity.

B. Slot Biasing

The bias network design for the slot varactor was chal-lenging because for each element, a voltage needed to bedeveloped across a slot in the ground plane, where the groundplane needed to be electrically contiguous at high frequencies.Any perturbations in the slot created undesirable results in thetransmission response of the element. Furthermore, the useof lumped elements or inter-digitated capacitors to create RFcontiguity but DC isolation was not feasible due to the physicalsize of the slot and the unit cell.

However, since each layer was to be fabricated fromsubstrates with metallization on both sides, two layers ofmetallization were available to form the ground plane. Thus,as shown in Figure 7, the 5.0 mm × 2.0 mm slot was placedin the lower ground plane, with the two lengths of the slotelectrically isolated in DC, and a large 5.0 mm × 5.0 mmrectangular hole was placed in the upper ground plane. Thewidth of the rectangular hole was the same as the width ofthe slot, so that the biasing grooves do not electrically extendthe slot. The length of the hole was large enough such that avaractor diode could easily be soldered to the lower groundplane. A small cavity in the upper substrate (not shown)created room for the slot varactor diode to fit in betweenthe substrates. The two substrate layers were separated bya 0.04 mm Rogers 3001 dielectric bonding film. With thethin film in place, a large capacitance is created between theupper and lower ground planes, resulting in a biased slot thatappears electrically contiguous at high frequencies, but hassides that are isolated for DC voltages. Waveguide tests withelements biased in this way showed that this biasing schemehad minimal effect on the performance.

The fabricated 6 × 6 planar array is shown in Figure 8.On each substrate, one side had metallization containing thepatches, and the other side had metallization containing theground plane. Because large amount of the copper had tobe removed from the patch sides, they were patterned bychemical etching. The ground plane sides of the substrateswere patterned using a milling machine because only a slot andbiasing traces needed to be cut from the copper. Moreover, themilling machine facilitated the drilling of alignment and screwholes. Since a varactor diode needed to be inserted betweenthe two substrates, a 3.0 mm× 1.0 mm× 1.0 mm cavity wasmilled into underside of the upper substrate.

Fig. 7. Array biasing design (vertically exaggerated)

Fig. 8. Fabricated 6x6 array

IV. ARRAY MEASUREMENTS

A. Experimental Setup

The array was mounted onto a frame, which was coveredwith absorber foam on the feed side, as shown in Figure 9. Apyramidal feed horn with a directivity of 17.6 dB was placedsuch that the aperture of the horn was 300 mm from theground plane of the array, which corresponded to an f/D ratioof 1.67. The feed horn was positioned such that the array wasprime-focus fed. The feed horn was 180 mm long, with anaperture size of 140 mm × 130 mm. The entire setup wasplaced on a planar 5′ × 5′ Nearfield Systems Inc. (NSI) near-field scanner.

The patches and slot biases of each element were connectedto a series of USB voltage controllers. As there were 36 ele-

ments, 72 independent voltage channels were required. Three32-channel voltage controller boards were used to control theelements. Each channel had 8-bit digital-to-analog resolution,and produced a voltage between 0 V and 20 V.

Fig. 9. Experimental setup in near-field scanner

B. Beamforming

Figure 10 shows the far-field magnitude response of pencilbeams directed at θ0 = −20◦,−10◦, 0◦, 10◦, and 20◦ in theE-plane at 5.7 GHz, where θ0 is defined as the angle of thepencil beam from broadside. The broadside pencil beam hadside-lobe levels of −15.0 dB, and a half-power beamwidthof 17.6◦. Beamforming was also tested for pencil beams inthe H-plane, where the results are shown in Figure 11. Thebroadside pencil beam had side-lobe levels of −12.4 dB, and ahalf-power beamwidth of 15.5◦. The directivities of the pencilbeams are summarized in Table II.

From the plots, we can see that as θ0 deviated from broad-side, the magnitude response decreased dramatically. This isdue to the fact that as θ0 is increased, the range of requiredphase responses in the elements is also increased. Since theelements cannot produce large transmission magnitudes forevery phase, some elements effectively become disabled. Thus,there is a reduction in the overall amount of power transmittedthrough the array, and the gain is reduced.

We can also observe that as θ0 is increased, ripples appearin the shape of the main lobe. This is likely caused by surfacewaves and mutual coupling between the elements, since as θ0

is increased, the phase differences between adjacent elementsis also increased, leading to excitation of surface waves. Theripples in the far-field patterns are evidence that the elementsare interacting on the array aperture, since discrete elementsin a six element array would generate distinct lobes and nullsin the far-field.

Figure 12 shows a comparison between the broadside-beamformed 3D far-field pattern and the 3D far-field patternof a hypothetical uniform aperture with the same dimensionsas the array (180 mm × 180 mm). Given that the theoreticaldirectivity of the hypothetical aperture is 4π

λ2 ab = 21.7 dBi [9](p. 672), the directivity of this transmitarray at broadside is

−60 −40 −20 0 20 40 60−25

−20

−15

−10

−5

0

5

10

15

Elevation [deg]

Mag

nitu

de [d

B]

20°

10°

0°

−10°

−20°

Fig. 10. Measured far-field pattern (E-plane)

−60 −40 −20 0 20 40 60−25

−20

−15

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−5

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Azimuth [deg]

Mag

nitu

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20°

10°

0°

−10°

−20°

Fig. 11. Measured far-field pattern (H-plane)

only 0.9 dB less than that of an ideal aperture. Furthermore,the half-power beamwidths in both principal planes can becalculated as 50.6

a/λ = 14.8◦, which is very close to thebeamwidths achieved by this array. The theoretical first side-lobe level of a uniform aperture, −13.26 dB, is also veryclose to the side-lobe levels achieved by this array. In fact, themagnitude tapering due to the feed resulted in smaller E-planeside-lobes, but at the cost of slightly wider beamwidths. Atbroadside, the 6× 6 transmitarray demonstrates beamformingperformance that is very close to optimal.

TABLE IIPENCIL BEAM DIRECTIVITIES

θ0 E-plane H-plane−20◦ 19.4 dBi 17.9 dBi−10◦ 20.2 dBi 20.1 dBi0◦ 20.8 dBi 20.8 dBi10◦ 20.0 dBi 20.7 dBi20◦ 19.0 dBi 18.6 dBi

(a) Broadside Far-field Pattern (b) Uniform Aperture Far-fieldPattern

Fig. 12. Comparison between the broadside far-field pattern and the far-fieldpattern of a uniform aperture with the same dimensions as the array.

V. CONCLUSIONS

A fully reconfigurable 6 × 6 transmitarray has been ex-perimentally demonstrated at 5.7 GHz. To the best of ourknowledge, it is the first experimentally demonstrated lensarray where each element can be individual tuned and fulltwo-dimensional beamforming is achieved. Requiring onlytwo microstrip layers with no inter-layer connections, thisdesign is a low-cost and easy-to-fabricate candidate for high-gain reconfigurable beamforming. Furthermore, this designachieves a tunable multi-pole response with minimal arraythickness.

Future work will focus on improving the phase tuning rangeof the unit elements, which is limited by the loss in thevaractor diodes in the current design. While it is unlikely thatthe resistive loss of the diodes can be eliminated, differentpositioning of the diodes may improve the loss. We will alsoaim to reduce the two control voltages to one voltage perelement, since biasing complexity increases rapidly as arraysize increases. Finally, the possibility of increasing the orderof the element will be explored.

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