DECLARATION OF ORIGINALITY - University of...
Transcript of DECLARATION OF ORIGINALITY - University of...
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DECLARATION OF ORIGINALITY
NAME OF STUDENT MIINGI ANDREW MWENJA
REGISTRATION NUMBER F17/7859/2001
COLLEGE ARCHITECTURE AND ENGINEERING
DEPARTMENT ELECTRICAL AND INFORMATION
ENGINEERING
COURSE NAME BACHELOR OF SCIENCE IN ELECTRICAL
AND ELECTRONIC ENGINEERING
TITLE OF WORK DESIGN OF A TWO STAGE CMOS OP AMP
1. I understand what plagiarism is and I am aware of the University policy in this
regard
2. I declare that this final year project report is my original work and has not been
submitted elsewhere for examination, award of degree or publication where other
people’s work, or my own work has been used, this has properly been
acknowledged and referenced in accordance to the University of Nairobi
requirements
3. I have not sought or used the services of any professional agencies to produce this
work
4. I have not allowed, and shall not allow anyone to copy my work with the intention
of passing his/her work.
5. I understand that any false claim in respect of this work shall result in
disciplinary action; in accordance with university anti-plagiarism policy.
Signature:……………………….. Date:………………………..
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CERTIFICATION
This project report has been submitted for examination to the Department of Electrical
and Information Engineering, University of Nairobi with my approval as the supervisor
Professor Elijah Mwangi
Department of Electrical and Information Engineering
Signature:……………………….. Date:………………………..
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DEDICATION
I would like to dedicate this project to my family who have been very caring and
supportive. I also dedicate this to all those who have encouraged and supported me while
undertaking this course.
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ACKNOWLEDGEMENT
I express my gratitude to my supervisor, Prof. Elijah Mwangi for guiding me especially
on text book research.
I also take this opportunity to thank my classmates and friends for all their academic and
moral support.
I finally thank my parents, Dr. John Stephen Miingi and Mrs. Magdalene Muchoki
Miingi for constant financial support throughout my education.
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ABSTRACT
A two stage CMOS OP AMP has been designed with 0.5m technology using p-mos
and n-mos transistors. The op amp has a loop gain of 85.02 dB and gain bandwidth
product of IMHZ.
The project is made up of hand calculations for width to length ratios.
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LIST OF ABBREVIATIONS
1. OP AMP – Operational amplifier
2. CMOS- Complementary metal oxide semiconductor
3. dB- D
4. G- Gate
5. D- Drain
6. S-Source
7. K’ – process transconductance parameter
8. W- Transistor width
9. L- Transistor length
10. Cox- Capacitance per unit gate area
11. U- election or hole mobility
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CHAPTER ONE: INTRODUCTION
1.1. BACKGROUND
Op amps have been in use for a long time primarily on analog computation and
sophisticated instrumentation. It may contain large number of transistors on the same
silicon chip. They have high input impedance, do not amplify noise, are cheap. It is quite
easy to design circuits using the IC opamp.
It may contain one or more differential stages. The topology of the cmos opam is
discussed using 0.5μm technology.
Figure 1.1 a,b,c show various ways of using an op amp
Figure 1.1a : The non-inverting configuration
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Figure 1.b: The inverting configuration
Figure 1.1c: The unity gain buffer or voltage follower
1.2 OBJECTIVES
1.2.1 OVERALL OBJECTIVE
To design a two stage cmos opamp with gain bandwith product of 1MHZ, a view rate of
IV/μsec and CMRR of at least 60 . It should operate at low voltage of the ±3V range.
1.2.2 SPECIFIC OBJECTIVES
Design consists of mathematical calculations
1.3 PROJECT SCOPE
The project aims at coming up with a design of a two stage cmos opam such that
mathematical calculations followed by a simulation using pspice lead us to check if we
meet specifications.
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1.4 JUSTIFICATION OF STUDY
Op amp circuits are used in computation, instrumentation and other application, they are
popular due to cheap cost as a building block of modern electronic circuits.
1.5 ORGANIZATION OF REPORT
Chapters in the report are as follows: Chapter 1 is the introduction of the project and
general organization. Chapter 2 contains the literate review. Chapter 3 contains design
and analysis.
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CHAPTER TWO: LITERATURE REVIEW
2.0 ENHANCEMENT TYPE MOSFET
The enhancement type MOSFET (Metal-Oxide-Semiconductor-field Effect transistor is a
type of Field Effect Transistor (FET) where current is cut off until the input voltage
between its input terminals reaches a specific magnitude. The input voltage is actually
reverse biased with no direct contact between the input gate and the conducting channel,
with only the charge accumulated controlling the current flow. This is due to insulation at
its input. Current is conducted by only type of carrier – elections or holes. The fact that
an input voltage can control its output current without direct contact makes Field Effect
Transistors very popular. In fact the E-MOSFET (Enhancement type MOSFET) is the
most widely used FET.
Figure 2.1: nmos
The construction of the n-channel enhancement – type MOSFET is provided in figure
2.1/ a slab of p-type material is formed from a silicon base. This slab is at times called the
substance or body. The substrate at times can be connected to the source. The drain and
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source are two heavily doped regions created in the substrate. A thin layer of silicon
dioxide – an insulator is grown on the body. Finally metal is deposited on the SiO2
(silicon dioxide) layer to form the gate metallic platform from the region between the
drain and source. Between the metallic contacts and the body is the p-type substrate.
2.1.1 OPERATION WITH VGS = OV
With no bias voltage applied to the gate, two back to back diodes exist in series between
the drain and source. One diode is formed by the pn junction between the n+ (heavily
doped) drain region and the p-type substrate, and the other diode is formed by the pn
function between the p-type substrate and the n+ source region (also heavily doped).
They prevent current conduction from the drain to source when a voltage VDS is applied.
A very high resistance in the order of 1012Ω exists between both drain and source.
2.1.2 OPERATION WITH VGS>OV
Grounding the source and the drain and applying positive VGS causes free holes to be
repelled from the substance under the gate. These holes are pushed downwards into the
substrate leaving behind a camer-depletion region. The positive gate voltage attracts
electrons from the n+ source and drain regions into the channel region. This effectively is
connecting the drain and source region. This is shown in figure 2.2.
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Figure 2.2 : VGS> 0, VGS = 0
2.1.3 OPERATION WITH VDS>OV, VGS>OV (Small VDs)
With VDs at a small value (say 50mv), the voltage VDs causes current in to flow through
the induced n channel from drain to source.
For a VG = Vt the channel is just induced and the current conducted is still negligibly
small. As VGS is increased beyond Vt (thresholds) more electrons are attracted into the
channel. The conductance of the channel is proportional to the “excess gate voltage” VGS
– Vt, or “effective voltage” or “overdrive voltage.”
2.1.4 OPERATION WITH VDS INCREASED
As VDS increased we note that as we travel along the channel from source to drain the
voltage increases from 0 to VDs. Hence it is tapered and is narrowest at drain end
implying varying resistance.
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Figure 2.3 : VDS varied
When VDs is increased to the value that reduces the voltage between the gate and drain
end to Vt; that is VGD=Vt or VGS-VDS=Vt, or VDS=VGS – Vt the channel depth at the drain
end decreases to almost zero, and the channel is said to be “pinched off” and increasing
VDS beyond this value has little effect on channel width. Current remains constant at the
value reached for VDS = VGS – Vt. And the drain current ‘saturates’ at this value – the
saturation region has been reached. The voltage at which this occurs is
VDssat=VDs – Vt (2.1.4)
The region of the iD – VDs characteristics obtained for VDs<VDSsat is called the triode
region.
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Figure 2.4 : Saturation and triode regions
2.1.5 THE iD – VDSCHARACTERISTICS
The device is cut off when VGS<Vt. To operate the MOSFET in the trade region we
must first induce a channel.
VGS≥Vt (induced channel)
Then we keep VDS small enough so that the channel remain continuous. This is achieved
by ensuring that the gate-to drain voltage is
VGD>Vt (continuous channel)
Since
VGD = VGS + VSD = VGS – VDS, thus
VGS – VDS> Vt
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Or
VDS< VGS – Vt (continuous channel)
Thus the enhancement type MOSFET operates in the triode region when VGS is greater
than Vt and the drain voltage is lower than the gate voltage by atleast Vt volts.
In the triode region
= ( − ) − (2.1.5)
Where
= ′and K’ is the process transconductance parameter
=== ℎ== ℎ ℎ , = ℎ ℎ
To operate the transistor in the saturation region, a channel must be induced.
≥ ( ℎ )and pinched off at the drain end by raising VPS to a value that results in the gate-to-drain
voltage galling below Vt.
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≤ ( ℎ ℎOr
≥ − ( ℎ ℎThus the n channel enhancement type MOSFET operates in the saturation region when
VGS is greater than Vt and the drain voltage does not fall below the gate voltage by more
than Vt volts. Therefore
= 12With the depletion layer widening the channel length is in effect reduced, a phenomenon
known as channel length modulation
= ( − ) (1 +Where
== ℎ ℎ
ℎ ℎThe channel length modulation makes the output resistance in saturation finite.
ɤ =
11
Or
ɤ = 2 ( − )Thus
ɤ = 1Or
ɤ =Where VA is the early voltage
2.1.6 N-CHANNEL MOSFET CIRCUIT SYMBOL
Fig. 2.5(a) Fig. 2.5(b) Fig. 2.5(c)
N-CHANNEL CIRCUIT SYMBOLS
Figure 2.5
a) Circuit symbol for the n-channel enhancement type MOSFET
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b) Modified circuit symbol with an arrowhead on the source terminal to distinguish
it from the drain and to indicate device polarity (n-channel)
c) Simplified circuit symbol to be used when the source is connected to the body or
when the effect of the body on device operation is unimportant.
2.2. P-CHANNEL MOSFET (P-MOS)
The circuit symbol for the p-channel enhancement type MOSFET is shown in figure 2.6
(a), (b), (c). For the PMOS, Vt is negative, to induce a channel we apply a fate voltage
that is more negative than Vt, also VDS is negative. The current ID enters the source
terminal and leaves through the drain terminal.
Fig. 2.6(a) Fig. 2.6(b) Fig. 2.6(c)
P-channel circuit symbols
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Fig. 2.7: Current flow directions for PMOS
Figure 2.6 (a) shows the circuit symbol for the p channel enhancement type MOSFET
Figure 2.6 (b) modified symbol with an arrowhead on the source lead
(c) simplified circuit symbol for the case where the source is connected to the
body.
2.3 COMPLEMENTARY MOS OR CMOS
This technology employs MOS transistors of both polarities. Figure 2.8 shows a cross-
section of a CMOS chip illustrating how the PMOS and NMOS transistors are fabricated.
While the PMOS transistor is implemented directly in the p-type substrate, the P-MOS
transistor is implemented directly in the p-type substrate, the PMOS transistor is
fabricated in a specially created n region called the n well.
The two devices are isolated from each other by a thick region of oxide that functions as
a insulator.
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Fig. 2.8: Cross-section of CMOS integrated circuit
2.4 DC PARAMETERS OF THE OPAMP
2.4.1 OFFSET VOLTAGE AND CURRENTS
According to the ideal gain model of an opomp the dc output voltage should be zero
when the dc input voltage is zero. However in practice a dc offset is measured at the
output terminal even when the dc input voltage is zero.
The total output offset voltage is a function of two separate effects
(a) Input offset voltage
(b) Input bias currents
The input offset is as a result of a phenomenon arising from dc balance of the two points
The input bias currents are the actual current that must flow into or out of the two
terminals to ensure proper operation of the solid state circuitry.
=
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===
| | = ℎ=
| ( ) = | | + | |
= = | − |+2
2.4.2 INPUT COMMON MODE RANGE ICMR
It is the average voltage of inverting and non inverting terminal
2.4.5 OUTPUT IMPEDANCE
Let Rout be output impedance of OPAMP, that include connected resistances
= ɤ1 +Where ɤ is output impedance of opamp itself and is feedback factor.
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2.4.6 INPUT IMPEDANCE
Let the input impedance be Rin of the opamp including other connected resistance, rd be
input of impedance of opamp itself
= ɤ (! + )2.4.7 COMMON MODE REJECTION RATIO (CMRR)
CMRR gives the value by which the opamp suppresses common mode inputs.
= 20 logWhere Ad is differential gain (gain when differential input is applied).
Acm is the common mode gain (gain when common – mode inputs are applied)
2.4.8 SLEW RATE
The process which the output voltage of an opamp can change only at a finite range is
slew rate limitation. This effect is described quantitatively by specifying slew rate SR for
a give op amp.
The maximum rate of change of output voltage is SR
=
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2.4.9 GAIN – BANDWIDTH RELATIONSHIP
See figure 2.9 below. As the gain reaches IA|=1 . This frequency at which this condition
is met is called unity gain frequency and is denoted on the curve as B
Fig. 2.9: Form of the magnitude of open loop gain for many op amp as a function of
frequency. Both f and A are logarithmic
= 1The unit gain frequency is the product of the dc or low frequency gain and the 3db
frequency. For this reason, the “unity gain frequency” is also called the “gain bandwidth
product.” Both terms unity gain frequency and gain bandwidth product are used in
specifications.
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CHAPTER 3: DESIGN
3.1 CMOS OP AM design steps
1. Showing OP AMP SCHEMATICS
A schematic is created of all the eight transistors and connections.
Transistor ratios and ac currents selection
2. Poles and zeros obtained as well as dc gains
3.2 Design inputs
Requirements
1. Slew rate
2. Common mate rejection ratio
3. Gain bandwidth product
4. DC gain
5. Common mode input range
6. Output impedance
7. Input resistance
3.3 Design procedure for a 2 stage CMOS OP AMP
The first stage is a differential amplifier pair that is actively loaded with the current
mirror.
The second stage is a common source amplifier actively loaded with the current source
transistor.
The two stage CMOS OP AMP configuration is drawn in figure 3.1
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Figure 3.1: Two stage OPAMP configuration
3.4 A summary of design relationships for the two stage OP AMP
1) Slew rate =CC
I5
2) Gain bandwidth =C
t C
GMIW
3) Transmission zero =C
Z C
GMW 2
4) Dominant pole =221
1
1
RGMCRWP
C
5) Second non-dominant pole =2
22 C
GMWP
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6) Stage 1 gain )//( 040211 rrgA m
7) Stage 2 gain )//( 070662 rrgA m
3.5 OP AMP design procedure
1. Estimate the compensation capacitance CC by choosing or selecting a moderately
large load capacitance CL using the second non-dominant pole WP2 and transmission
zero Wz using similar triangles.
2. Determine I5 using slew rate relation; SR=I5|CC obtain tCWCGMguGM 11 sin
obtain (W/L) ratios of all transistors (note level 1 mo** parameters of 0.5 m
technology used.
3. Find overdrive voltage of Q1 Q2 Q6
Find r02, r04, r06, r07 using IDVr Ao ||
4. Find the voltage gains A1 and A2 and overall gain A obtain overall gain A in dB/
5. verify that ft is indeed lower than fz and fp2
3.4 Calculations
Step 1
2
2
2121
22 )( C
G
CCCCC
CGw m
C
Cmp
Where Lgddbdb CCCCC 7762
CC = Compensation capacitate
2211
1
RGCRwp
mC
C
mZ C
HW 2
21
Since Zpz WW
Say
wzWp 22
Since LCC 2 generally
L
mp C
GW 2
2
ThusC
m
L
m
C
G
C
G 22 2
CL CC
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CC = 2CL
Given a load capacitated CL = 12pf
CC = 2x12pf = 24pf
Choose CC = 20pf
Step 2
AAI
SVSNrateslewSR
SRCI C
201020101020
/101|
)(
66125
6
5
ft = 1MHZ (unity gain bandwidth/gain bandwidth product)
sradftwt /1028.62 6
Since tCt
worCw
Gm1
22
11
1
6
6121
/1256.0
/106.125
1028.61020
mm
m
tCm
Ggbut
VmAG
VA
wCG
Thus VAgm /1256.01
4
6
4
6
L
WL
W
I
Ior
46
4
6I
L
WL
W
I
But AII 665 1020
Choose (W/L)5 = (W/L)6 = 200
446
/
2001020
LW
I
And since 64 10102/16 I
1001020
1010200/
6
6
4
LW
Since Q3 and Q4 are matched:
34 // LWLW
To obtain (W/L)7
5
7
4
6
/
/2
/
/
LW
LW
LW
LW
23
200200
200200/
200
/2
100
200
7
7
LW
LW
Since Q1 and Q2 are matched choose (W/L)1 = (W/L)2 = 200
Step 3
Find overdrive voltage |VOV| at which Q1, Q2, Q6 are operating, find overdrive voltage.
L
WvovpKI nP
21,2
1
Or 1,/
2
pKLW
IDvov
n
To find |vov| of Q1 i.e. |vov, 1|
Given oxpCKp 1
But using level – 1 MOSFET model parameter cancel
]5./[100
115./115
105.9
]5./[1001155./115{
22
2
9
222
vmsvcmp
mtox
vmvcmp
ox the permittivity of 112 1045.3 SiO f/m
911 105.9/1045.3/ oxoxox tC
265
5
9
92
11'
/75.411075.4110175.4
105.9
10967.3
105.9100
1045.3115
VAK
CKp
p
oxp
24
V
I
KLW
IVov
p
D
0489.0
1039.2
103.8
1020
)1075.41)(200(
10102
)1075.41)(200(
2/2
)/(
2|1,|
3
3
6
6
6
6
11
1
Since Q1 and Q2 are matched
|Vov, 1| = |Vov, 2| = 0.0489V
To find |Vov| of Q6 i.e |Vov, 6)
Using level 1 mosfet model parameters since 460onu
That is pn u4 , in similar triangles
V
KLW
IVov
VAK
VAkk
n
D
n
np
034.0197.10334.0
1040
10167)200(
10202
')/(
26,
/167'
]/[1075.4144
6
6
6
6
6
2
26''
8m1 was already obtained as 8m1 = Gm1 = 8m2 = 0.1256MA/V
25
Find 8m for Q6
VMA
or
VA
VovL
Wk
m
m
nm
/36.18
/10136.1034.0200101678
6,,8
6
366
6
'6
Find r02, r04, r06, r07, using level 1 parameter 1.0,2.0 np
To find r02 = (r01)
Given VV pPA 202.0/1/1||
m
I
Vr
p
AP 11021021010
20|| 666
202
To find r04 = (r03)
MII
Vr
D
n
D
An 11011010
10|/1||| 66
4404
To find r06
kr
or
II
Ar
D
n
D
AN
500
105.01020
10|/1|||
06
66
6606
To find r07
kr
or
I
Vr
D
n
d
An
500
105.01020
10|/1|
7
||
07
66
7707
Step 4
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Find the voltage gains A1 and A2 and the overall gain A
The voltage gain of the first stage is determined from
The voltage gain of the second stage is determined from;
VV
k
kk
rrgA m
|284
25010136.1
)500||500(10136.1
)||(
3
3
070662
The overall dc open loop gain is
2.835,17
)284()8.62(21
A
AAA
In dB
Adb = 20log 17, 835.2 = 85.02dB
Step 5
Verify that ft is lower than fz and fp2
Since ft = 1MHZ
And gm6 = Gm2
VV
MIM
rrgA mi
|8.62
10500101256.0
)1||(101256.0
)||(
33
3
04021
27
MHZf
f
C
Gf
z
z
C
mz
044.9
10044.910256.1
1036.1
28.61020
10136.1
2
1
610
3
12
32
MHZf
Hz
C
Gf
CCnb
p
mp
L
074.15
10074.15
101228.6
1036.11
2
1
2
6
12
3
2
22
2
Thus ft is indeed lower than fz and fp2
Step 6
The lower limit of the input common mode range is the value of the input voltage at
which Q1 and Q2 leave saturation region. This occurs when the input voltage below the
voltage at the drain of Q1 by voltsVtp || . But vVtp 8.0|| .
xVV GSDD 3
To get VGS3
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VVx
VVx
VV
V
VVVov
VVVov
L
WVkI
DD
GSDD
GS
GS
tGS
tg
ovnD
834.0
834.08.0034.0
8.0034.0
||
||
2
1
3
3
3
33
333
2
33
3
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The lower limit of common mode range is VVICM 3min
VV
Vx
xV
DD
p
1.2
1.2|8.0|3
||3
The upper limit of the ICMR is the value of input voltage of which input voltage of Q5
leaves saturation region. Since for Q5 to operate in saturation, the voltage across it i.e.
55PV should be at least equal to the overdrive voltage at which it is operating i.e. 0.048v,
the highest permitted at the drain of Q5 should be vVVS 3.18.01.2055 . It follows
that;
Step 7
)1( OLdin ArR
Typically 1210dr
1612 10784.1)2.178351(10
/2.835,17
1
in
OL
R
VVA
30
REFERENCES
Adel S. Sedra, Kenneth C. Smith, “Microelectronic Circuits, Theory and Applications,
“Oxford University Press, 5th Edition, 2009
William D. Stanley, Operational Amplifiers with Linear Integrated Circuits. Pearson
Education, Fourth Edition, 2009.
Robert Boylestad Louis Nasheskly, Electronics Devices and Circuits Theory, 10th
Edition.