1

Abstract— In this paper, a simple approach has been proposed to determine parameters

of CPMLs with BGS. A break or slot in the ground plane raises the impedance of

microstrip lines. Firstly, the S-parameters of the transmission lines are obtained using an

electromagnetic simulator. Then the L and C values are extracted using circuit modeling

approaches. Measurements were performed using a vector network analyzer for the

fabricated CPMLs. Results show that the parallel or crossed break with decoupling

capacitors or ferrite beads enhances the noise isolation and lead to a reduction of signal

reflection and the radiated emission. Frequency characteristics such as crosstalk, return

loss and EMI radiated noise in the crossed break are degraded if capacitors or beads, are

not used.

Index Terms—break in ground structure (BGS), coupled pair of microstrip lines

(CPMLs), crosstalk effect, EMI noise, slot.

I. INTRODUCTION

Faster signal transients in high-density electronic systems are the current trend in new

high-speed circuit design. Using a cheaper substrate with a smaller number of routing

Crosstalk and EMI Noise Investigation for a Coupled

Pair of Microstrip Lines (CPMLs) with a Break in Ground

Structure (BGS)

Morteza Kazerooni1, Ahmad Cheldavi1 and Mahmoud Kamarei2

1College of Electrical Engineering, Iran University of Science and Technology, Tehran, Iran

2Faculty of Electrical and Computer Engineering University of Tehran, Tehran, Iran

2

layers is an attractive proposition at high frequencies. Routing on a severely size-

constrained circuit board can inadvertently create non-ideal conditions such as a

reference plane with breaks or slots produced by various power and ground islands.

Electromagnetic problems such as crosstalk and EMI noise, which are usually ignored at

relatively low frequencies in these circuits, are critical for overall system performance [1-

3]. Crosstalk or radiation coupling is also a fundamental topic for the estimation of

susceptibility and immunity since there are various types of transmission lines such as

microstrip lines and electronic equipments. However, crosstalk analysis of microstrip

lines including complex structures and a non-uniform ground plane (e.g. BGS) is usually

difficult. The models used for these structures are usually based on the transverse

electromagnetic (TEM) approximation [4], where the per-unit-length parameters are

determined first and then the crosstalk of the propagation signals is generally estimated

by using transmission line theory. Since the medium surrounding the microstrip is

inhomogeneous, numerical techniques such as the method of moments [5], finite-

difference solution technique [6], or finite element method [7] have been used to predict

the per-unit-length parameters. Other solutions have also been developed, e.g. [8] for

time-domain crosstalk between two parallel lossless lines.

In this paper, the problems of crosstalk, return loss and EMI noise in CPMLs with a

BGS are considered and investigated. First, the scattering parameters are obtained using

the full-wave EM simulator Ansoft HFSS version 11.0. Then the L and C values can be

extracted from a circuit modeling approach. These parameters are frequency dependent.

Generally, the solution method is based on the quasi-TEM assumption although in some

3

cases, this assumption is not valid. Finally, the simulation results are compared to

measurements of the structures. The simulated results show good agreement with

measurements. In this paper three examples of CPMLs are considered; one with uniform

and others with non-uniform ground planes.

II. BREAK IN GROUND STRUCTURE (BGS)

For high-speed signal transition systems, the power and ground planes are used as

important references. Therefore, when high-speed transmission lines cross a break in the

ground plane, crosstalk and radiated emission are greatly increased [9-14]. Fig. 1(a)

shows the existence of a crossed break in the ground plane. A crossed break in the

ground plane (Fig. 1(a)) differs from a crossed slot in the ground plane (Fig. 1(b)).

Current flows in the circuit that has the minimum impedance. This means that at high

frequencies, the current in the ground plane tends to flow directly under the signal trace.

Due to the break in the ground plane most of the return current takes a circuitous route

with minimum length around the break and increases the length electrically.

a crossed break in Ground Plane

Ground Plane

Direct Signal Trace 1

Crosstalked Current Flow

Direct Signal Trace 2

Direct Signal Trace 3

Return Signal Trace 1

Return Signal Trace 2

Return Signal Trace 3

(a)

4

a crossed slot in Ground Plane

Ground Plane

Direct Signal Trace 1

Crosstalked Current Flow

Direct Signal Trace 2

Direct Signal Trace 3

Return Signal Trace 1

Return Signal Trace 2

Return Signal Trace 3

(b)

Fig.1. Current distribution in a typical high density circuit. a) a crossed break in ground plane and b) a

crossed slot in ground plane

Increased crosstalk leads to unpredictable signal quality. Radiated emission, on the

other traces, may lead to electromagnetic interference (EMI) noise. Therefore, a ground

plane with a break can interrupt the return current and adversely affect EMI, crosstalk,

signal rise/fall times and characteristic impedance.

In most cases, using breaks in the ground plane is unavoidable. Therefore, some

discrete decoupling capacitors are used to reduce the signal integrity degradation, as

show in Fig. 2. A decoupling capacitor is sometimes placed near the signal trace across

the break to maintain the high frequency return current path, but this is a poor solution.

5

a complete crossed break Ground Plane

Ground Plane

Decoupling Capacitor

Fig. 2. Using decoupling capacitors in ground plane with a complete crossed break

Also, employing decoupling capacitors leads to additional inductance in the board

parameters and therefore creates an undesirable resonance in the frequency response as

well as increased manufacturing costs.

For high frequency design, it is desirable to have a uniform ground plane with no

breaks. However, breaks in the ground or power planes are a common feature in densely

packaged and multilayered PCBs. Breaks due to vias and thru holes divert the ground

plane return current are often used to isolate unintentional signals from intended signals

in a PCB.

The current must return to its source, so the return current follows the break parasitic

inductance and potential signal integrity (SI) problems can occur. Also, the current

through the equivalent inductance results in a noise voltage between the two pieces of the

ground plane that can lead to EMI events. Therefore:

1- A break is often cut in the ground plane to isolate the ground of the noisy drivers

from the sensitive loads by increasing the impedance of the noise route. Therefore, by

6

adjusting the dimensions of the break and changing the current distribution on the ground

plane, good susceptibility can be obtained. Fig. 3(a) shows an illustration of the current

flow scenario in a BGS design.

2- One can properly select the break or slot dimensions to create an electromagnetic

band gap (EBG) structure which suppresses the noise.

3- A complete crossed break in the ground plane can be placed on between the

connector and the microstrip as illustrated in Fig. 3(b). Using a ferrite bead to bridge the

break allows the intended low frequency signals to have a return current path, while

effectively maintaining the high impedance of the break to the high frequency

unintentional signals. High-speed I/O signals such as video should never be run over the

complete crossed break in the ground plane with a ferrite bead.

Noisy Driver Sensitive LoadDirect Current

Return current

A crossed break

Ground Plane

Capacitive current or displacement current is

proportional to the width of the break and

degree of charge density

Inductive current or conduction current is

proportional to the area of the break and degree

of current density

Effective area of the break

(a)

7

(b)

Fig. 3. a) Current flow scenario in a BGS design and b) one method for isolation of low and high

frequency PCB areas with the complete crossed break

III. EQUIVALENT CIRCIT OF A BREAK AND SLOT

Due to existence of a break or slot, an additional capacitance and inductance will be

created (Fig. 4(a)). A break and slot translate into a capactance due to the edge of the

defect and an increased inductance due to the longer current paths. Hence the increase in

area of the current loop causes an increase in EMI emissions or inductive noise since it is

directly proportional to the total defected area. For high speed signals, the signal trace

looks more like a transmission line and impedance mismatches cause reflections.

Therefore quantification of the increase in inductance is desirable for EMI estimation.

The break or slot section can be modeled exactly by lumped frequency dependent L and

C elements assuming that the wavelength is larger than the defect dimensions (Fig. 4(b)).

It is important to note that the transmission line is assumed lossless. The extraction of

simulated results shows that βα << allowing for the assumption of a lossless line. This

model is a real physical model. Therefore, all changes in the defect can be clearly

8

represented by values of the lumped elements. The total ABCD matrix for the

transmission line with the break or slot becomes the product of the three sections (Fig.

4(c)). Two sections are simple transmission lines and a lumped LC resonator has been

placed between them. Therefore, in spite of the existence of the break or slot, the TEM

approximation is correct.

The following equations present the approximate extracted equivalent circuit LC

parameters for the break or slot which are constant with frequency. The parallel

capacitance value in Fig. 4(b) for the given dimensions can be extracted from the

attenuation pole location. The capacitance of the equivalent circuit, can be obtained as:

)ff(2

1Z2fC 2

c200

c

−π⋅= (1)

Once the capacitance value of the equivalent circuit is extracted, the series equivalent

inductance for the given defected section can be calculated as:

Cf41L 2

02π

= (2)

To validate the circuit model, two transmission lines with a break and slot have been

simulated using Ansoft Designer 3.0. The proposed structures are designed on substrates

with relative permittivity 2.33. Also, the values of lA,, lB,, lC are 34.8 ,34.8 and

12.875mm, respectively. The width of the microstrip trace (W) on the board and its

length are 2.25 and 71.6mm, respectively. The dielectric substrate was 0.787 mm thick.

The width of the break or slot (wb) is 2mm and the length of the break and slot are 36.25

and 22mm, respectively. The extracted L, and C values are 4 nH, and 0.376 pF for the

break and 3.792 nH, and 0.247 pF for the slot, where fc and f0 are the resonant frequency

9

and 3-dB cutoff frequency, respectively. The simulation results for the amplitude show

relatively excellent agreement as illustrated in Fig. 4(d) up to 5-6 GHz. This means that

the model is valid up to this frequency. Therefore, above 5-6 GHz non-TEM modes are

dominant. The models can be improved by adding a resistor parallel to the LC resonator,

which represents the radiation from the break or slot.

Length

wb

lA lB

lCw

wb

lA lB

Length

lC

lC

w

(a)

(b)

The break or slot circuit

Port 2Port 1

Measurement plane

Circuit plane Circuit plane

Measurement plane

lA

p0 v,Zp0 v,Z

lB

(c)

10

0 1 2 3 4 5 6 7 8 9 10-60

-50

-40

-30

-20

-10

0

freq(GHz)

Am

plitu

de(d

B)

S parameters of the crossed break using full wave analysis and circuit model simulation

S11

S21

Red and solid lines are fromcircuit model simulation

0 1 2 3 4 5 6 7 8 9 10

-80

-70

-60

-50

-40

-30

-20

-10

0

freq(GHz)

Am

plitu

de(d

B)

S parameters of the crossed slot using full wave analysis and circuit model simulation

S11 S21

Red and solid lines are fromcircuit model simulation

(d)

Fig. 4. a) Dimensions of two microstrip circuits with a crossed break and slot in the ground planes, b)

the break or slot equivalent circuit model, c) schematic of embedded the break or slot by the transmission

lines (lA=lB=(Length-ws)/2) and d) comparison of the scattering parameters using full wave analysis and

circuit simulation

IV. EXTRACTION OF LINE PARAMETERS OF TRANSMISSION LINE WITH

BGS

Based on EM simulation results, the additional inductance and capacitance of a

transmission line with a BGS is established and these parameters are extracted using

circuit theory. The S-parameters as shown in (3) are obtained from Ansoft HFSS

simulator and can be used to derive the total L and C parameters of the transmission line.

This method assumes a TEM mode approximation. The procedure is detailed as:

[ ] ⎥⎦

⎤⎢⎣

⎡=

)f(S)f(S)f(S)f(S

)f(S2221

1211 (3)

The relations between the S-parameters, Z11 and Y11 and the propagation constant for

the lossless line are given as:

11

)l).f(cot(Z.j)f(S)f(S))f(S1)).(f(S1(

))f(S).f(S))f(S1)).(f(S1.((Z)f(Z 0

21122211

21122211p11 β−=

−−−

+−+= (4)

)l).f(cot(Y.j)f(S)f(S))f(S1)).(f(S1(

))f(S).f(S))f(S1)).(f(S1.((Y)f(Y 0

21122211

21122211p11 β−=

−++

++−= (5)

where Zp is the port impedance of 50Ω with Yp being the corresponding admittance.

β , l, Z0 and Y0 are the propagation constant, length, characteristic impedance and

admittance of the interconnect, respectively. Then:

11

110 Y

Z)f(Z = (6)

and

l/)Y.Z(cot)f( 11111−=β (7)

The relation between inductance and capacitance for a transmission line is given by:

εμ= .)f(C).f(L (8)

and

υ==

)f(C1

)f(C)f(L)f(Z0 (9)

therefore, the inductance and capacitance per unit length of a microstrip line are given

by:

)f()f().f(Z

f2

f2.)f(C

1

)f(C1)f(L 0

2 ωβ

=π

υπ

υ=

υ= (10)

and

12

)f(L1)f(C 2υ

= (11)

where

εμ=υ

.1 (12)

Simulations were performed for investigation of inductance and capacitance per unit

length for the proposed transmission line with and without the break and slot (Fig.5).

At low frequencies, the current is distributed uniformly throughout the conductor.

However, as the frequency rises, there will be a tendency for the current to concentrate at

the surface of the conductor. It is assumed that the current decreases exponentially inside

the conductor (skin effect). As a result, the outer portions of the conductor contribute less

than the inner parts to the overall inductance (current has more difficulty passing through

the inner parts due to the skin effect). Thus, if current is concentrating on the surface, the

inductance will be decreased. Therefore, with an increase of frequency, the inductance

decreases.

0.5 0.6 0.7 0.8 0.9 1 1.1 1.2 1.3 1.4 1.50

0.5

1

1.5

2

2.5

3

3.5x 10

-7

freq(GHz)

Indu

ctan

ce(H

/m)

Frequency dependent inductance

Line Without the crossed break and slot

Line with the crossed breakLine with the crossed slot

0.5 0.6 0.7 0.8 0.9 1 1.1 1.2 1.3 1.4 1.50

1

2

3

4

5

6

7

8

9x 10

-9

freq(GHz)

Cap

acita

nce(

F/m

)

Frequency dependent capacitance

Line without the crossed break and slot

Line With the crossed breakLine with the crossed slot

Fig. 5. Comparison of the frequency dependent inductance and capacitance per unit length for the

proposed transmission line with and without the break and slot

13

As can be seen from Fig. 5, with increasing frequency, the value of inductance for the

line is decreased. The current distribution is supposed to be a vector quantity and a

function of frequency. Hence, the inductance of the transmission line decreases to a

limited extent with an increase in frequency. According to this figure, the crossed break

has caused more values of inductance with respect to the standard transmission line. This

increase is more than 0.07 m/Hμ up to 0.7 GHz. The inductive noise in this frequency is

high. This means that the break in ground plane creates additional inductance in the

transmission lines. Also the results show almost no difference in capacitance line in these

traces at low frequencies. As can be seen, that the capacitive noise increases slowly at

very high frequencies.

V. EXAMPLES OF THREE TYPES OF BREAKS AND SLOTS IN THE GROUND

PLANES OF CPMLS

Fig. 6(a) illustrates types of the breaks and slots that were cut on the ground plane of

the test boards underneath the CPMLs. The cross sections of the CPMLs with the BGS

are shown in Fig. 6(b). For these test boards S=10mm, the break length in the CPMLs

with the parallel break is 71.6mm and the slot length in the crossed and parallel slot is 20

and 28mm, respectively.

14

Study board without break

Study board with parallel break

Study board with crossed break

Study board with parallel slot

Study board with crossed slot

(a) (b)

Fig. 6. a) Three basic types of the test boards for crosstalk assessment and b) the cross sections of the

CPMLs with the breaks

With a lumped representation, two coupled transmission lines are described only by L

and C matrices of size of 2 x 2. Two lossless, symmetrical reciprocal transmission lines

are described by:

LLLL

LLLL

LM

M

2221

1211 == (13)

and

CCCC

CCCC

CM

M

2221

1211

−−

=−

−=

(14)

where LM and CM are the mutual inductance and capacitance, respectively. As a second

method, this structure may be described by its eigenvalues, even-mode and odd-mode

15

characteristics. The wave can be decomposed to even and odd mode waves. Each wave

has its own characteristic impedance and propagation constant. The even-mode and odd-

mode characteristic impedances and propagation delays are:

M

Moe CC

LLZ−+

= M

Moo CC

LLZ+−

= (15)

and

)CC)(LL(t MMpde −+= )CC)(LL(t MMpdo ++= (16)

The relative capacitive mc and the magnetic ml coupling can be defined as:

CCc M

m = L

Ll mm = (17)

According to the above equations, in a homogenous medium, the relative coupling

capacitance and inductance are similar, therefore the two modes propagate with the same

speed: tpde = tpdo. If the two modes propagate with the same speed, no far-end crosstalk

will appear with matched terminations. This is the case in coupled planar striplines. In

surface microstrips, however, due to the inhomogeneity of the propagation medium, tpde

and tpdo are different. In surface microstrips, since the dielectric fills only the volume

bellow the strip, the capacitive coupling is weaker than the inductive coupling: lm > cm. In

the next section, the mutual couplings and effects of the breaks and slots on CPMLs are

investigated.

VI. INVESTIGATION OF MUTUAL COUPLINGS ON PROPOSED CPMLS

Mutual capacitance and inductance in two lossless, symmetrical reciprocal

16

transmission lines can be determined using simulation or experiment. At low frequencies

where the transmission lines are electrically short, either capacitive or inductive coupling

dominates for an open or short termination, respectively, of the driven line [15]. In either

case, the coupling is proportional to the mutual parameters, and the mutual capacitance

and inductance can be determined from S21 simulations or measurements. An equivalent

circuit model for this procedure is shown in Fig. 7.

ZS

ZLVS

Driven/Generator Line(TL1)-Z0

(1) (2)

ZFEZNE

Coupled/Receptor Line(TL2 )-Z0

(3) (4)

Near End Coupling

Far End Coupling

Length

Length

ZS

ZL

VS

ZNE ZFE

I1

jωLmI1

V1jωCmV1

Fig. 7. Schematic and equivalent circuit of symmetrical mutually CPMLs

The electrical parameters of CPMLs can be measured using a LCR meter or generally

calculated by well-known reference [16].

Fig. 7 shows the setup was used to determine S21 and derive expressions for mutual

inductance and mutual capacitance between Port 1 connected to one microstrip trace and

Port 2 attached alternately to the near-end and far-end of the other microstrip trace. To

calculate mutual capacitance, both traces were terminated with an open circuit. Also, to

calculate mutual inductance, both traces were terminated with a short circuit. The mutual

inductance and mutual capacitance were determined from S21 as follows [17]:

17

LengthZS

CPI

m ...221

ω= (18)

LengthZS

L PIm ..2

.21

ω= (19)

where ZPI is the port impedance (50 ohm). Fig. 8(a) shows the mutual coupling for

proposed structures. Whole of the cases, except for the crossed break, the CPMLs with

non uniform ground planes can be modeled using the symmetrical CPMLs with uniform

ground planes. In this case equations (13) to (19) can be used to determine the

characteristic impedances, phase constants and mutual couplings. From this figure, in the

CPMLs with a break or slot, the inductive coupling increases at low frequencies. Since

the return current is evenly distributed on the ground plane, existence of a slot causes a

disturbance. Cutting a break in the ground plane between and parallel the signal traces

can effectively isolate the driven circuit from the coupled circuit and reduce the inductive

coupling at higher frequencies e.g. above 1.05 GHz as illustrated in Fig.8(a).

At high frequencies with a parallel slot in the ground plane, the return current is

allowed to spread out across the ground plane just as it was able to do when there was a

solid ground plane. Since the width of slot is small, the return current is not adversely

disrupted and the mutual inductance is not too affected. When the return current is

restricted to the side under the source trace in the parallel break case, the inductive

coupling is decreased. However, in the parallel slot case, the isolation is generally

ineffective and can increase inductive coupling.

18

At lower frequencies, parallel breaking and slotting (bellow 0.9 GHz and 0.6 GHz,

respectively) in the ground plane can increase capacitive coupling but at higher

frequencies these defects can decrease it.

When a crossed slot is cut into the ground plane, the current distribution is abruptly

changed (Fig. 8(a)) and more of the electric and magnetic fluxes that would normally

couple to the ground plane from the source trace couple to the victim trace, thereby

increasing mutual inductance and capacitance. A similar effect occurs in the crossed

break as well. However, in this case, the mutual couplings are very much higher than the

crossed slot. Therefore, the line parameters of two mutual traces are not the same and the

CPMLs is not a symmetrical structure. Thus, one cannot use the above equations. The

proposed equivalent circuit of the crossed break is shown in Fig. 8(b). This equivalent

circuit contains two lumped inductances and two lumped capacitances and related

couplings such that LB1> LB2 and CB2 > CB1. To determine these values, the method

which is discussed in this paper can be used for each line individually. The extracted LB1,

LB2, CB1 and CB2 values are 5.16 nH, 1.89 nH, 0.292 pF and 0.881 pF, respectively.

Mutual couplings can be obtained from full wave simulation and circuit modeling by

comparison of standard CPMLs and CPMLs with the crossed break.

Fig. 8(c) illustrates how the breaks and slots change the distributions of return currents

at low or high frequencies. The magnitude of surface current distribution on the ground

of the test board with the crossed break is graphically presented in Fig. 8(d) at 1GHz. The

difference of current level due to the break is obvious.

19

20

(a)

(b)

21

(c)

(d)

Fig. 8. a) Mutual inductance and capacitance for the proposed CPMLs, b) the equivalent circuit for the

CPMLs with the crossed break, c) schematic of the current distribution in the proposed breaks and slots and

d) the magnitude of surface current at the 1 GHz for CPMLs with the crossed break in the ground plane.

22

VII. INVESTIGATION OF CROSSTALK EFFECT, RETURN LOSS AND EMI

NOISE IN PROPOSED CPMLS

In order to further investigate the conditions under which the crosstalk and return loss

is increased, a number of experiments and computer EM simulations were performed.

The far-end crosstalk cannot be neglected in the inhomogeneous medium case and can be

a dominant factor when the rise and fall times of the applied input signal are very short.

The effect of the parallel and crossed break on the near and far-end crosstalk of the

investigated test boards is shown in Figs. 9(a) and 9(b). Due to the crossed break in the

ground, the crosstalk in neighboring transmission lines is increased. Therefore, the

equivalent expression may be interpreted as referred to as inductive and capacitive

couplings. The influence of these parameters on the performance shows an impressive

destruction in signal integrity. This degradation is about 26dB for the near-end and 33dB

for the far-end. Also, the far-end crosstalk in the CPMLs with the parallel break is

improved. However, the near-end crosstalk in the board with the parallel break is slightly

increased at high frequencies.

Fig. 9(c) shows the measured and simulated results of S11 for the three proposed

CPMLs test boards. As shown in this figure, the S11 in the CPMLs is increased

significantly with the crossed break.

To survey the EMI noise in these test boards, the E-field pattern is a good gauge. As

can be seen from Fig.10, the realized pattern gain in the CPMLs with the crossed break

has been increased. This means that EMI noise in this case is higher than the other test

boards.

23

The proposed test boards in this paper were fabricated and measured from 0.5 to 1.5

GHz. Photographs of three fabricated circuits with the breaks are shown in Fig. 11.

0.5 0.6 0.7 0.8 0.9 1 1.1 1.2 1.3 1.4 1.5

x 109

-70

-60

-50

-40

-30

-20

-10Comparison of measurement and simulation

freq(GHz)

Nea

r-en

d cr

osst

alk(

dB)

measured

(a)

0.5 0.6 0.7 0.8 0.9 1 1.1 1.2 1.3 1.4 1.5

x 109

-55

-50

-45

-40

-35

-30

-25

-20

-15

-10Comparison of measurement and simulation

freq(GHz)

Far

-end

cro

ssta

lk(d

B)

measured

(b)

24

0.5 0.6 0.7 0.8 0.9 1 1.1 1.2 1.3 1.4 1.5

x 109

-55

-50

-45

-40

-35

-30

-25

-20

-15

-10

-5Comparison of measurement and simulation

freq(GHz)

Ret

urn

loss

(dB

)

measured

S11(

dB)

(c)

Fig. 9. Comparison of the measured and simulated of a) near-end crosstalk, b) far-end crosstalk and c)

return loss in the proposed test boards

-50.00

-40.00

-30.00

-20.00

90

60

30

0

-30

-60

-90

-120

-150

-180

150

120

Fig. 10. Comparison of the radiation pattern in E-plane between the three proposed CPMLs

25

The proposed breaks

Fig. 11. Photograph of the three fabricated CPMLs

VIII. CONCLUSIONS

A simple technique for extraction of L, C and mutual coupling parameters of CPMLs

with a break and slot in ground plane has been developed.

Breaking or slotting (defecting) the ground plane is an effective way to reduce or

suppress noise, but this approach may create some undesirable effects on the

transmission lines such as CMPLs. A narrow parallel slot cut into the ground plane may

be ineffective at reducing inductive coupling at high frequencies. However, it can reduce

the capacitive coupling at these frequencies. Adding a parallel break in the ground plane

actually decreases the mutual capacitance and inductance at high frequencies. Three

factors, namely the types, orientation and dimensions of the defect determine the effect

of the break or slot on capacitive and inductive couplings in CPMLs. Also, if the defect

crosses the traces, mutual couplings increase.

To reduce far-end crosstalk between two traces in the CPMLs, it may be advisable to

apply a break in ground plane such as using a parallel break.

26

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