Microelectronics Reliability 50 (2010) 1738–1743

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Microelectronics Reliability

journal homepage: www.elsevier .com/locate /microrel

Control technique for power device electro-thermal stress minimisationin non-linear load variable-frequency resonant power converters

F. Carastro *, A. Castellazzi, J. Clare, P. WheelerDepartment of Electrical and Electronic Engineering, University of Nottingham, Nottingham NG7 2RD, UK

a r t i c l e i n f o a b s t r a c t

Article history:Received 8 July 2010Accepted 13 July 2010Available online 7 August 2010

0026-2714/$ - see front matter � 2010 Elsevier Ltd. Adoi:10.1016/j.microrel.2010.07.044

* Corresponding author. Tel.: +44 (0) 115 84 6.E-mail address: fabio.carastro@nottingham.ac.uk (

This work proposes the development of a control technique to ensure soft-switching of the active devicesin a high-voltage high-frequency power supply, during all operational conditions. The study is carried outwith relevance to an industrial electrostatic precipitator system, based on a series-resonant convertertopology. The main operational features of the circuit are presented, detailing the requirements for highefficiency and high reliability. Based on theoretical analysis, the development of specific control solutionsis thoroughly discussed. Experimental results are presented to demonstrate the validity of the consideredapproach for maximizing circuit performance and reducing the risk of degradation of the power switches.

� 2010 Elsevier Ltd. All rights reserved.

1. Introduction

In a variety of industrial processes such as gas and oil firedpower stations, cement and steel production facilities, wasteincineration plants, high power electrostatic precipitators (EPS)are used to remove dust particles and reduce polluting emissions[1]. For most precipitators, a power supply with a high voltageoutput of 50–100 kV with high short-circuit capability is requiredto achieve the best efficiency of dust removal. Traditional powersupplies in use for high voltage EPS systems are based upon 50/60 Hz line frequency technology, which results in quite cumber-some designs [2]. Increasingly stringent environmental policiesacross the world are causing this market to grow rapidly andbroadly. This is pushing research towards novel power supplyarchitectures with improved performance (dynamic responseand efficiency), power density and reliability [2,3]. High-fre-quency resonant converter topologies are now being consideredto meet these targets. One of the advantages of these topologiesis the possibility to switch the active devices with virtuallynon-dissipative transitions (soft-switching), both at turn-on andturn-off, thus optimizing efficiency and minimising the electro-thermal stress of the devices [4]. However, due to the intrinsicallynon-linear nature of the load, sophisticated control techniques arerequired to ensure soft-switching resonant transitions, taking intoaccount that the regulation of the converter output voltage andcurrent require not only pulse-width modulation, but also vari-able frequency of operation.

ll rights reserved.

F. Carastro).

2. Circuit description and simulation results

This paper considers a converter rated at 100 kV, 2 A. Althoughresonant converter topologies have been established for some timein lower power applications, there are relatively few examples oftheir deployment in high power applications such as that describedhere. The power circuit design has been developed to be capable ofsupporting continuous operation of the EPS. Fig. 1 shows a sche-matic diagram of the converter and its control platform. The sys-tem consists of an input rectifier and a dc-link capacitor,followed by an H-bridge connected to a series resonant tank. Ahigh frequency transformer incorporated into the supply is usedto step-up the voltage to 25 kV. A two stage Cockcroft–Waltonvoltage multiplier is used to step-up the voltage to the nominal100 kV. The precipitator load, indicated as RL, changes accordingto the properties and quantity of the dust. When a spark occurs in-side the EPS, this effectively short-circuits the output of the powersupply. The power supply must be capable of surviving the effectsof this condition and must quickly reduce its current to zero in or-der to extinguish the spark. The short-circuit equivalent load oc-curs on average 50 times per minute. During recovery of thesteady state it is paramount to ensure soft-switching of the activedevices to ensure good dynamic performance and avoid switchelectro-thermal overload, degradation and failure.

The full electrical design of the proposed power supply has beencarried out assuming that the converter switches in a square wavemode to excite the resonant tank. A load quality factor (Q) of 3 hasbeen selected to achieve approximately sinusoidal tank waveformswithout excessively large tank components. In addition to simplifythe mathematical model, rectifier, filter and load have been re-placed with an equivalent AC resistance. The expression of theequivalent resistance can be calculated under the assumption of

C

Ltank

3-phase Supply

x6

Lf

Cf

RL

Voltage Multiplier

GATE DRIVEVOLTAGE

TRANSDUCER

A/D

Programmable Control Platform

VABVdc

V

I

S1

S2

GATE SIGNALS

PROTECTION AND CONTROL MEASUREMENTS

InputRectifier

Ctank

HVHF_TR

Cstack_1

Cstack_2

Cstack_3

Cstack_4

Itank

S4

A/D

Nonlinear load

A

B

PEAK DETECTOR

CURRENT TRANSDUCER

VOLTAGE

TRANSDUCER

S3

Fig. 1. Schematic of the single-phase power supply and control platform.

Table 1Power supply specifications and design results.

Description Symbol Value

Power supply specificationLoad voltage V 100 kVLoad voltage ripple Vr <3%Load current I 2AOutput power P 200 kWLoad voltage recovery V 1.5 kV/msQuality Factor Q 3Switching frequency fsw 20 kHzDC-link Vdc 600 V

Design resultsTank inductor Ltank 0.027 mHTank capacitor Ctank 2.04 lFNatural frequency f0 21.2 kHzEPS filter inductor Lf 3 mHEPS capacitance Cf 0.5 lFDC-link capacitor C 7 mFTransformer turn ratio N 1:47Voltage multiplier gain km 4

F. Carastro et al. / Microelectronics Reliability 50 (2010) 1738–1743 1739

ideal elements and absence of AC components in the load current.Table 1 summarizes the resulting power supply parameters basedon the specification.

To control the output voltage of the EPS a PI current controllerhas been designed. In order to calculate the PI controller parame-ters the toolbox SISO in Matlab has been used in conjunction withthe given power supply specifications. Direct control of the peaktank current allows fast regulation and recovery of the output volt-age after a sparking event. Furthermore, during the short-circuitcondition the controller has to follow zero current reference. Forthese reasons the current controller has to be designed in the dis-

crete domain to take into account all delays affecting the dynamicsof the control system.

Fig. 2a–g shows simulation results for the proposed power sup-ply topology. Initially the power supply operates in the normalcondition; the current follows the reference current and the final100 kV output voltage is reached after 0.09 s. Fig. 2g shows the de-tail of the small load voltage ripple at the final condition. Moreoverduring this voltage regulation, as detailed in Fig. 2f, the chosentopology and control allow for zero current switching (ZCS) turn-on and zero voltage switching (ZVS) turn-off of the IGBTs of thelagging leg; however, the IGBTs of the leading leg experiencehard-switched turn-off commutations and require additionalsnubber capacitors to achieve low dissipation operation. Att = 0.15 s a short-circuit event, representing a spark between theEPS electrodes, occurs and the spark controller reduces the tankcurrent reference to zero. Fig. 2c shows that the tank current israpidly reduced to zero by the current controller, with a very lim-ited current spike, avoiding possible damage to the precipitator aswell as the power supply. When the short-circuit dies away thecontrol returns to voltage regulation mode and the voltage recov-ers at the maximum rate (corresponding to rated tank currentlimit).

Control of the power supply is implemented by means of both adigital signal processor (DSP) and a field programmable gate array(FPGA). The control platform also includes voltage and currentmeasurement to control the output voltage and for circuit protec-tion from destructive overloads.

3. Control strategy development

To control the tank current, hence the DC output voltage of theESP, whilst minimising the semiconductor losses, a combined

Zoom

Time [s]

(a)

(b)

(c)

(d)

(e)

(f)

(g)

Fig. 2. Representative circuit waveforms during the regulation.

VA

VB

Vdc

Vdc

VABItankVAB_fund

Φψ

Vdc

Fig. 3. Theoretical waveforms for resonant tank using frequency and phase shiftcontrol.

0 0.2 0.4 0.6 0.8 1

1

0.9

1.1

1.2

1.3

1.4

1.5

m

F= fs

w /

fo

Fitting equation

F = 0.88*m(F)4 - 2.6*m(F)3 + 2.8*m(F)2 - 1.6*m(F) + 1.5

m(Q=3) calculated from Eq (2) 4th degree fitting

Fig. 4. Series-resonant converter gain and fitting curve.

1740 F. Carastro et al. / Microelectronics Reliability 50 (2010) 1738–1743

phase shift and frequency control has been adopted. Fig. 3 showstheoretical waveforms for a resonant tank operating with phaseand frequency control.

It must be observed that the control phase shift (/), betweenthe diagonal pair switches of the H-bridge, is twice the phase shift(w) between the tank voltage fundamental harmonic (VAB) and cur-rent (Itank); hence leg B will always switch at zero crossing point ofthe tank current, while leg A will always soft turn-on and hardturn-off. The phase shift between VAB and Itank in the series reso-nant configuration is given by

wðFÞ ¼ arctanðQðF � F�1ÞÞ ð1Þ

where Q is the resonator quality factor and F is the ratio betweenswitching frequency (fsw) and tank natural resonance frequency (f0).

For a given tank current target value, the controller output willdetermine target values for the converter voltage (VAB); so, themodulation index can be calculated by sampling the dc-link volt-age according to the well known relation

mðFÞ ¼ VAB � p4 � Vdc

ð2Þ

On the other hand, the modulation index (m) for a phase andfrequency controlled resonant converter with zero current switch-ing is related to the switching frequency as [5]

mðFÞ ¼ F2

ðF4Q 2 þ F2 þ Q2 � 2F2Q 2Þð3Þ

So, the value of m calculated after (2) will yield a well deter-mined value of F, to be calculated with (3). Here, as anticipated,the control is implemented in a DSP. So, to avoid computational de-lays the value of F is computed by means of a fourth order polyno-

(a)Converter output voltage VAB

Output Voltage

Tank Current

F. Carastro et al. / Microelectronics Reliability 50 (2010) 1738–1743 1741

mial approximation of Eq. (3), as indicated in Fig. 4. Note that theconverter operates with a modulation index approaching unity(corresponding to the resonance frequency) when the controller(of PI type in this case) reaches steady state.

m=0.18

fsw=25.6 kHz

(b)

4. Experimental setup and results

Based on the electrical design a prototype converter has beenconstructed, shown in Fig. 5. Based on the resonant tank specifica-tion (f0 = 20 kHz), an inductor rated at a peak current of 600 A is re-quired. The design adopted is a ferrite (3C90) inductor with adistributed gap. The devices used for the H-bridge are standard800 A dual-switch IGBT modules. The step-up high frequencytransformer was assembled using U ferrite cores and Litz-wire.For this application a dry film metallized polypropylene dc-linkcapacitor optimized for high power applications (low ESR) hasbeen used.

m=0.7

fsw=21.2 kHz

(c)

4.1. Open-loop experimental results

Fig. 6 shows experimental open-loop results demonstrating theachievement of soft-switching of the IGBTs at different values ofthe modulation index. For the test an equivalent resistive loaddirectly connected at the transformer secondary side. It is possibleto observe that the intended soft-switching performances (seeFig. 3) have been achieved for all values of the modulation indexvalue.

m=0.9

fsw=20.2 kHz

Fig. 6. Experimental results at different operating points.

4.2. Closed-loop experimental results without voltage multiplier

Fig. 7a–c shows closed-loop experimental results, without thevoltage multiplier. It is possible to observe that the designed con-troller is capable of tracking the tank current demand as Fig. 7ashows. Fig. 7b–c shows also the detailed PI regulation perfor-mances; it is possible to observe that soft-switching is kept duringboth transient and steady state conditions. Fig. 8 shows the de-tailed output waveforms at full power during closed-loop steadystate operation. It is possible to observe that a maximum outputvoltage of 27.8 kV has been obtained. During this condition thecontroller is following a peak tank current demand of 600 A, deliv-ering an output power of 220 kW. Furthermore stable soft-switch-ing operation is maintained during the steady state conditions.These results verify the correct operation, in line with the electricaldesign.

Gate driver

H-bridgeDC-link

Controlplatform

Inprecti

Fig. 5. Converter protot

4.3. Closed-loop experimental results with voltage multiplier

In order to verify that soft-switching operational condition weremaintain even in the presence of the high capacitance introducedby the final configuration, the voltage multiplier has been con-

Resonantinductor

Resonantcapacitor

ut fier

ype and test setup.

Demand tank current

Converter output voltage

Actual tank current

Peak detector output(a)

(C) Zoom steady state(b) Zoom transient

Fig. 7. Experimental results during tank current regulation.

Capacitor tank voltage 1.9 kV

(a)

(b) Zoom

Converter output voltage 600 V

Output voltage 27.8 kV

Tank current 600A

Fig. 8. Full power closed-loop experimental results.

1742 F. Carastro et al. / Microelectronics Reliability 50 (2010) 1738–1743

nected at the transformer secondary terminals. It is possible to ob-serve that the presence of the voltage multiplier does not affect theperformance of the controller furthermore soft-switching opera-

tion is maintained during both transient and steady state condi-tions also in the presence of the voltage multiplier as detailed inFig. 9.

Zoom

DC output voltage

Converter output voltage

Transformer secondary voltage

Tank current

Fig. 9. Closed-loop experimental results with voltage multiplier.

F. Carastro et al. / Microelectronics Reliability 50 (2010) 1738–1743 1743

5. Conclusions

This paper has presented a control solution to ensure soft-switching of the semiconductor devices during all operationalmodes of a variable-frequency resonant converter supplying ahighly non-linear load. An analytical formulation of the controlstrategy has been presented and experimental results have beenprovided to support the validity of the envisaged solution. Furtherto enabling the achievement of the desired dynamic regulation ofthe output variables, the proposed approach allows for the optimi-zation of efficiency and the reduction of the electro-thermal stresslevel affecting the power switches.

References

[1] Mizuno A. Electrostatic precipitation. In: IEEE transactions on dielectrics andelectrical insulation, vol. 7, No. 5; 2000. p. 615–24.

[2] Grass N, Hartmann W, Klockner M. Application of different types ofhigh-voltage supplies on industrial electrostatic precipitators. IEEEtransactions on industry applications, vol. 40, No. 6; November–December2004. p. 1513–20.

[3] Durga Prasad NVPR, Lakshminarayana T, Narasimham JRK, Verman TM,Krishnam Raju CSR. Automatic control and management of electrostaticprecipitator. In: IEEE transactions on industry applications, vol. 35, No. 3;May–June 1999. p. 561–7.

[4] Carastro F, Clare JC, Castellazzi A, Johnson CM, Bland MJ, Wheeler PW.Reliability considerations in pulsed power resonant conversion. In: 20thEuropean symposium on reliability of electron devices, failure physics andanalysis ESREF; 2009.

[5] Bland M, Clare J, Zanchetta P, Wheeler P, Pryzbyla J. A high frequency resonantpower converter for high power RF applications. In: Power electronics andapplications, 2005 European conference, 11–14 September 2005; p. 1–10.