Cellular-Phone and Hearing-Aid Interaction: An Antenna Solution

Cellular-Phone and Hearing-Aid Interaction:An Antenna Solution

Taeyoung Yang', William A. Davis', Warren L. Stutzman1, and Minh-Chau Huynh 2

'Virginia Tech Antenna Group, ECE Dept.-Ol 1IVirginia Tech, Blacksburg, VA 24061, USA

Tel: +1 (540) 231-6307; Fax: +1 (540) 231-3362; E-mail: {mindlink, wadavis, [email protected]

2Sony Ericsson Mobile Communications (USA) Inc.Research Triangle Park, NC 27709, USA

Tel: +1 (919) 472-1284; Fax: +1 (919) 472-7451; E-mail: [email protected]

Abstract

With the introduction of digital cellular phones, hearing-aid users have experienced a severe buzzing noise caused by the

interaction between digital cellular phones and hearing aids. The cellular-phone industry, the hearing-aid industry, and

consumers have been seeking a solution for the interference issue. Efforts reported in the literature have focused on

measurements, modeling, and evaluation of interference and RF emission, but not on methods to solve the problem. In this

paper, we focus on the causes of the interference and an understanding of the problem. We also present a method to reduce

near-field electromagnetic energy around a cellular phone, mitigating the interference between cellular phones and hearing

aids. The theoretical investigation of both the radiation mechanisms and fundamental limits on antennas suggested that a

low-Q antenna, such as an ultra-wideband antenna, could reduce the near-field intensity. Simulations and measurements

were performed at 900 and 1880 MHz, using both low- and high-Q test antennas mounted on a mock cellular phone. The

results showed that the peak electric and magnetic near-field strengths of the low-Q test antenna were lower than those of a

high-Q test antenna by at least 5 dBV/mn and 4 dBA/m, respectively. The improvement in the near-field performance for the

low-Q antenna was without any sacrifice in far-field performance. Furthermore, in the presence of a human head, the

simulation results showed that the radiation efficiency of the low-Q test antenna was better than that of the high-Q test

antenna.

Keywords: Mobile antennas; hearing aids; land mobile radio cellular systems; land mobile radio equipment; land mobile radio

interference; electromagnetic compatibility; electromagnetic radiative interference; ultra-wideband (UWB) antenna; finite

element methods; moment methods

1. Introduction

Dl emographers estimate that there are approximately 28 millionpeople in the US who suffer from hearing impairment, and

5.6 million of them wear hearing aids [1]. Unfortunately, these

people are troubled by a buzzing noise in the hearing aid caused by

interference from digital cellular phones. Although not all combi-

nations of digital cellular phones and hearing aids have this inter-

ference, the resulting buzzing noise is sometimes overwhelming,

and effectively prevents the user from hearing the cell-phone audio.

Interference can also be caused by a person using a digital cellular

phone in close proximity to a person wearing a hearing aid. The

hearing-aid industry, the cell-phone industry, and consumers have

been collaborating to solve this incompatibility problem. As a con-

sequence of the efforts, the American National Standards Institute

(ANSI) formed a task group (C63. 19) in 1996 [2] to address the

issue. The hearing-aid-compatibility (HAG) standard, developed in

2001 and updated in 2006 [3], provides acceptable performance

levels for the measurement and evaluation of wireless-phone near-

field strength. The standard also addresses immunity levels for

hearing aids.

IEEE Antennas and Propagation Magazine, Vol. 50, No. 3, June 2008

An FCC rule and order in 2003 required that phone manufac-turers and carriers offer at least two phones that meet the HAG

standard. Specifically, phones that can be used with hearing-aid

telecoils were required by 2006, and at least half of all mobile-tele-

phone handset models must not exceed the specified RF interfer-

ence level by February 18, 2008. Thus, the industry has been

seeking a solution to meet the FCC requirement [4]. Early solu-

tions, such as a neck loop or hands-free option, provide enough

distance between the cellular phone and the hearing aid to mitigate

the interference [2]. An emission blocker, which claims to create a

radiation null toward the hearing aid, is also available on the mar-

ket. These early solutions were supplied in the form of accessories,

and were often considered to be inconvenient. Furthermore, these

solutions do not satisfy the FCC requirements that address the RF

emission from the wireless phone. Many efforts shown in the lit-

erature have focused on measurement, modeling, and evaluation of

the interference and RF emission [5-7], and not on the fundamental

issue of the cause of the field-strength levels.

We begin with a summary of the structure of hearing aids, the

interference sources, and the HAG standard. Causation issues are

ISSN 1045-9243/2008/$25 (92008 IEEE 51

then addressed, identifying the radiation mechanism based on cur-rent-slope discontinuities and the fundamental limit theory olantennas. After this background material, we demonstrate, throughsimulations and measurements, a potential method using a low-Qantenna to reduce near-field electromagnetic energy generated froma cellular-phone antenna. The radiation efficiency of both low- andhigh-Q test antennas mounted on a mock cellular phone are com-pared in the presence of an actual human-head model. All thesimulations in this paper were conducted with the FEKO Moment-Method/Finite-Element Method hybrid code [8].

2. Structure of Hearing Aids andSource of Interference

There are three basic types of hearing aids: behind-the-ear(BTE), in-the-ear (ITE), and in-the-canal (ITC), named in accor-dance with placement. The structure of the conventional behind-the-ear hearing aid is shown in Figure 1. Either an acoustic modeor a telecoil mode can be selected. In the acoustic mode, a micro-phone converts acoustic energy to electrical energy. In the telecoilmode, the telecoil detects magnetic-field leakage from the phonespeaker and converts the detected magnetic energy to electricalenergy. The converted electric energy is amplified and convertedagain to acoustic energy by the receiver, finally being delivered tothe ear through the ear hook [9].

The advantage of the telecoil mode is that no audio back-ground noise is received. Therefore, hearing-aid users receive abetter audio signal in the telecoil mode than in the acoustic mode.However, the telecoil can act as an unintentional antenna and cou-ple energy from cellular phones in close proximity to the hearingaid. The energy coupled through this unintentional antenna isdemodulated -also unintentionally - in the amplifier circuits of thehearing aid, causing a buzzing noise to hearing-aid users. The typi-cal frequencies of the interfering acoustic signal are 50 Hz, 217 Hz,and their harmonics. In the acoustic mode, the microphonebecomes the unintentional receiving antenna [5, 10, 11].

The cellular-phone antenna is the major source of strong elec-tromagnetic energy, and causes interference in both the acousticand telecoil modes. In addition, any fuinctional operation associatedwith a varying current in a cellular phone - such as battery-surgecurrents, power-supply components, keyboard scanning, and dis-play refresh -can induce magnetic fields and create interference inthe telecoil mode [5, 11]. In this paper, we focus on RE-energyemission from the cellular-phone antenna.

3. Hearing Aid Compatibility (HAC)Standard

HAG standard ANSI C63.19-2006 [3] was updated recently,and covers wireless devices from 800 MHz to 3 GHz. The tele-phone emission below 960 MHz was lowered by 10 dB3 from the2001 HAC standard. Therefore, it is likely equipment will meet theHAC standard below 960 MHz, but a significant challenge remainsabove 960 MHz.

Figure 2 illustrates the 50-mm-square measurement areadefined in the standard. The meshed area located in the measure-ment plane (I cm above the reference plane) is evenly divided into

52

nine cells, and is centered on the wireless-device speaker. Field-strength measurements are made by scanning a probe over thismeshed area. Based on the measured peak near-field strength, thecellular phone is placed into one of the categories listed in Table 1.The categories are 5 dB3 apart, and GSM cellular phones have anadditional -5 dB weighting factor. In the far-field region, the elec-tric and magnetic fields are related by the intrinsic wave imped-ance. However, in the near-field region, the electric and magneticfields are not strongly coupled, and cannot he related in terms ofthe intrinsic wave impedance. In addition, there may be other mag-netic-interference sources, as explained in the previous section.Therefore, the HAG standards categorize the electric and magneticfields separately.

The standard for hearing-aid immunity to RE interference ismeasured using a balanced dipole antenna, and has four similarcategories (also denoted by Ml to M4). The FCC requires HAGcompliance for both hearing aids and cellular phones independ-ently, providing a variety of choices for users to select a combina-tion of hearing aids and cellular phones. The HAG standard treatsthe hearing aid and cellular phone together as a system, classifyingthe system's performnance by summing the category values of each,as in Table 2. For example, a phone with M2 emission level and ahearing aid with M3 immunity fall in the normal-use category(2 + 3 =5 ). A category sum below four suggests the system is notlikely to be useable.

4. Near-Field Theory

The electric field from a current distribution (J1) on an antenna isgiven by [ 12]

(1)- I 2 + ( V A ]

Table 1. Telephone emission limits and categories 13].

Category Telephone RF Parameters (above 960 MHz)

Ner ied E-Field Emission H-Field EmissionNearFiel (dB/m) dBA/m)

Category M1 46 to 51 + 0.5 x AWEF -4.4 to -0.6 + 0.5 x AWErCategory M2 41 to 46 + 0.5 x AWE -9.4 to -4.4 + 0.5 x AWETCategory M3 36 to 41 + 0.5 x AWF -14.4 to -9.4 + 0.5 x AWECategory M4 < 36 + 0.5 x AWE < -14.4 + 0.5 x AWE

AWFT(Articulated weighted factor): UMTS(WCDMA) = 0, CDMA = 0,TDMA =0, and GSM = 5.Note: The levels below 960 MHz are 10 dB below the values indicated.

Table 2. System performance classification 13].

Category SumSystem Classification (Hearing Aid Category

+ Telehone CategorUseable 4

Normal Use 5Excellent Performance 6


Volume controlReceiver

k Mficrophone

Telecoill.

Functions/w "

Electric howd IVtflI

1,047 . 015340e-0282100.02I 950e.02

4 820P-023 690er-02

Figure 7. A comparison of the calculated E-field amplitude dis-

tributions at 900 MHz (10 mm away from the antenna's sur-

face: (I) dipole and (r) fat dipole.

Figure 1. The structure of a behind-the-ear (BTE) hearing aid

191.

magirwic "W1 I~fm

2942 3

I 098

Figure 8. A comparison of the calculated H-field amplitude dis-

tributions at 900 MHz (10 mm away from the antenna's sur-

face): (1) dipole and (r) fat dipole.

Figure 2. The reference and measurement planes for field-

strength measurements complying with ANSI standards

C63.19-2006 131: (1) candy-bar type, (r) clamshell type.

4

-2

4

3

-8

at

Figure 5. The transient power calculated for an infinitesimal

dipole (normalized by average power).


where A is the magnetic vector potential and k =2;r/A . In freespace, the magnetic vector potential can be written as

Apfffj-dv'.V g

Radian Sphere --,

(2)

The first term of Equation (1), k 2 A , dominates the electric field inthe far field, while the second term, V ( v4) dominates in thenear field. Moreover, in the near field, Equation (1) increases dueto slope discontinuities of the magnetic vector potential, caused bythe second term having double derivatives (gradient of divergence).As discussed by King [ 13] and as can be seen from Equation (2),the magnetic vector potential (A) follows the spatial variation ofthe current distribution (j). To illustrate this effect, the normal-ized Afý and Jif distributions of z-axis-directed half- and quarter-

Figure 4. The field regions surrounding an antenna.

0

0N

I-

0.6i

01

0.2-

-950 -200 -160 -100 -42 0 42 100 150 200 250

L, Z [mm]

Figure 3. A comparison between the normalized magnitude ofthe current distribution and the magnetic vector potential (A)on the surface and at 10 rmm distance along the z axis(x=-10mm) for half- and quarter-wavelength cylindricaldipoles. The radius of the antenna cylinder was 0.01 mmn, andeach curve was normalized. (a, top) A 900 MHz resonant half-wavelength dipole (Lh /2 = 83.33 mmn); (b, bottom) A 900 MHzresonant quarter-wavelength dipole (Lq /2 = 41.67 mm).

54

wavelength dipoles are compared in Figure 3. The variations of HAand Ilare similar along the surface of the antenna cylinder, with

less variation of JAJ near the ends of the dipole. If the normalized

JIis evaluated at a larger radial distance from the antenna, thedistribution becomes smoother, but the normalized JAI still follows

the basic behavior of the normalized VIf- Therefore, the near fieldincreases near slope discontinuities in the current distribution.

As shown in Figure 3a, the conventional half-wavelengthdipole has a slope discontinuity in the current at each end of thewire. Thus, we can expect a strong electric near field, Af, aroundthe ends of the wire. For a quarter-wavelength dipole, there is anadditional slope discontinuity in current at the feed point (center),resulting from the approximately triangular current distribution(Figure 3b). As antenna length decreases, interference issuesincrease, due to the increasing current-slope discontinuity. Thiscurrent-slope discontinuity partially explains the following obser-vation by Kuk and Nielson [10]: "Cellular phones with built-in andshort antennas typically cause more interference than those withlong antennas."

To mitigate the slope discontinuities in antenna current distri-bution, we examined ways to create a smooth shape variation,including a fat dipole with a tapered feed. For the fat dipole, thecurrent spreads over the wide surface, reducing the slope-disconti-nuity effect. It is also well known that the impedance bandwidthincreases as the dipole becomes fatter.

The fundamental-limit theory on antennas [14] can also beused to address antenna near fields through the relationshipbetween the nonradiating stored energy around the antenna (nearfield) and the radiated power (far field), which defines the radiationQ. Suppose that we have an electrically small antenna without loss,as shown in Figure 4. Define the antenna sphere as the smallest


80r

75ý RAiu of aitnn *#ephforquWt X ipole

60

*5050

IL45

40

35

_-Ag- Peak IEI FtDipole

SPeak FAt V4 Dipole

_.r Peak tEl V4 Fat Dipote

Rakei of itactui Sph~eeJo spherical TMhi mode

s ~mo trnem sphwee

0.'I* M (lt

-... K011 PFA PkMMIe-DU~Wd PIFA8WueNeetWS

10

4 Figure 11. The measured and simulated VSWR as a function offrequency for a dual-band PIFA (frequency swept up to2.5 GHz) and for a spoon UWB.

20Ra~i o a~ru p Ai

-51,

pPeak 'H' ~pl

*Peak PIH1 DIoe~ ekPt24Fat Dipole

LN~~hafIM Dipol

Peak

tI -h Idpt

.Peak EIpoe- Peak tEI Felopo

--- Peak "I Dipole - Peak IIFtDpl

eh. Peak PIt IV4 Dipole - Peak tEt 214 FatDpolej

X [Imin

TT-

ii

II 11W

Figure 9. A comparison of the fields found by simulations for

half- and quarter-wavelength dipoles: (top) peak IEI, (middle)

peak Ifill, and (bottom) the difference of the peak field strength

as a function of the distance (x) from the antenna.


Figure 12. The near-field measurement setup: (1-r) mock cel-lular-phone dimensions, dual-band PIFA mock cellular phone,

spoon UWB mock cellular phone, and the definition of xdigt.

55

IL

'U

Ie0C

0

.3IL

1.

sphere that encloses the antenna structure, and assume that only thefundamental spherical mode (TM01) exists. Some of the energyprovided by the antenna structure then remains in the vicinity ofthe antenna as nonradiating energy, not contributing directly to theradiation process. This nonradiating stored energy dominates insidethe radian sphere of radius I/k, where k is the wavenumber [ 15].The radiation process dominates outside the radian sphere. Thenonradiating stored energy is trapped, and circulates between theantenna source and, approximately, the radian sphere. To visualizethis process, consider the total transient power of the infinitesimaldipole through the spherical surface at distance r, given by (seethe Appendix for details)

P(r~t)=zP. lcos2(o~t-kr-4\+ 1 sin 2 (wt-kr - 4)

(3)

where P%.,, is the average radiated power, and 4- = atan (i/kr) is theexcess phase delay of the radiated field. The normalized total tran-sient power is plotted in Figure 5. The positive sign for the totaltransient power represents outgoing propagation, whereas a nega-tive power represents power returning to the source. The energy-storage process is clearly observed inside the radian sphere(Icr <1I). There appears to be a periodic energy leak from the radiansphere to produce the radiation. This is the process of convertingreactive energy to radiated energy, which is discussed by Schantz[17]. Most cellular-phone services use frequencies below 2 GHz,where the corresponding radian-sphere radius is 37.5 mim, puttingthe hearing aid well into the reactive near-field region of the phone.If the stored energy and the radiated power are known, the radia-tion Q can be estimated from [ 14]:

Qd=2cmax[lWWm] 1,rad .d BW3dB

z(4)

where W,. and Wm, are the average nonradiating electric and mag-netic energies, respectively; Pad is the average radiated power;and BW3dB is the 3 dB fractional impedance bandwidth. This rela-tionship indicates that an antenna with a small near-field strengthhas a low radiation Q. In addition, the radiation Q is approximatelyinversely proportional to the impedance bandwidth. Therefore, anantenna with wide continuous impedance bandwidth inherently haslow nonradiating energy in the near field.

To summarize, investigations of both the radiation mecha-nism based on current-slope discontinuities and the fundamental-limit theory suggest that a low-Q antenna, such as an ultra-wide-band antenna, will provide low near-field energy around theantenna. In order to demonstrate this low-Q antenna approach, ahalf-wavelength dipole (high Q) and a planar fat circular dipole(low Q), shown in Figure 6, were investigated using simulations.The planar fat circular dipole is an ultra-wideband variation on theconventional dipole antenna [18]. The power delivered to bothantennas was 32 dBm. As shown in Figure 6b, the conventionalhalf-wavelength dipole was narrowband (high Q), with periodicresonances at odd harmonics of 900 MiHz. The planar fat circulardipole showed an ultra-wideband, high-pass characteristic. Bothantennas had standard dipole radiation patterns, with the radiationpattern of the fat dipole being slightly more directive in the planeof the antenna due to the slight focusing effect of the taper. Theazimuthal variation was less than 0.5 dB at 900 MHz, providing areasonably omnidirectional pattern. Tlhe E-field distribution at the

56

perpendicular distance x = 10mmn is plotted in Figure 7. As pre-dicted from the theory, the E field was concentrated near the wireends for the conventional dipole (Figure 7a), and was distributedover the circular surface of the circular fat dipole (Figure 7b).Similarly, the H-field strength of the fat dipole, shown in Figure 8,was significantly lower than the conventional dipole, due in largepart to the wider distribution of the current. As summarized inTable 3, the peak Etl and HI1 near field 10min away from the sur-face of the ultra-wideband circular fat dipole (low Q) were reducedfrom the values for the simple dipole (high Q) by approximately7 dBV/m and 4 dBA/m, respectively. Nevertheless, the far-fieldperformance was almost the same. Because there is 5 dB betweenHAG emission limits categories (see Table 1), the 7 dB reductionoffered a significant improvement.

-11-2a

L

rFigure 6a. A dipole antenna (narrowband, high Q) on the left,and a fat dipole antenna (ultra-wideband, low Q)on the right.

I I ---- ipole9-j Fat Dipole

> 6 I I

:~ II500 i000 1500 2000 2500 3000 3500 4000 4500 5000

Frequency [MHz]

Figure 6b. A comparison of the VSWR values for the antennasof Figure 6a. Note that L was the half wavelength at 900 MHz(166.67 mm) and a was 0.01 mm.


C9

Directrnty~ottd8lI2.40.9-0.7-2,2

-131

Z

O irectivityjotid8]3.52.31.20.0

-12

5.97.0

.21

Figure 13a. The radiation directivity pattern of the dual-bandPIFA at 900 MHz (Dm,. = 2.43 dB).

DirectMtyjotjdBlI24-44-3.1-5.9

Z-11.4-14.1

1 224-2511

Figure 13c. The radiation directivity pattern of the dual-bandPIFA at 1880 MHz (Dmx 3.52 dB).

Directiviy.jottdBI4.912,3-0.3-219-5.5

-10.1-13.4-16.0I-18.8-21.2

zxi

Figure 13b. The radiation directivity pattern of the spoonUWB antenna at 900 MHz (D,,,, = 2.39 dB).

Figure 13d. The radiation directivity pattern of the spoonUWB antenna at 1880 MHz (Dm,. = 4.94 dB).

IEEE Antennas and Propagation Magazine, Vol. 50, Na. 3, June 20085 57

Table 3. A comparison of Etl and HiI amplitudes for 900 MHz

conventional and fat dipoles at 10 mm distance from theantenna's surface (see Figures 6, 7, and 8).

xv 0-

10 20Y !mmI

Fat Dipole(Ultra-

Wideband)

IDifferencel

502.20 7.2d(54.02 dB) 7.2d

2.03 4.4d(6.15 dB) 41d

10X Etmm

Figure 10. The evolution of the spoon UWB antenna geometry(the height of the spoon UWB was 21.1 mm): (1) planar UWBantenna, (r) spoon UWB antenna.

In order to investigate the relationship between antenna sizeand the effectiveness of the low-Q antenna approach, the peak IEjl

and HIf values as a function of distance from the antenna were

compared for conventional and fat dipoles of both a half and aquarter wavelength. The quarter-wavelength antennas were tunedto deliver 32 dBm of power, as in the previous simulation. As seenin Figure 9, both half-wavelength low- and high-Q antennas hadnearly the same field strength beyond the antenna sphere (radiansphere < antenna sphere). Similarly, both quarter-wavelengthantennas had almost the same field strength beyond the radiansphere (radian sphere > antenna sphere). These observations sug-gested that a low-Q antenna lowers the near fields within the largerof the radian or antenna sphere, while retaining the same radiationperformance beyond the larger sphere.

5. Near-Field Measurements

To evaluate realistic systems, the two test antennas weremounted on identical mock cellular phones for testing. The firsttest antenna was a dual-band planar inverted-F antenna (PlEA),with overall dimensions of 4Ox10x10mm. (L xW xH). A mean-dered top plate, similar to one used in [19], was used to tune theantenna. The second test antenna, shown in Figure 10, was aspoon-shaped ultra-wideband (UWB) antenna with dimensions of

58

45.52 x20 x 21.1mm (L xW xH), designed by mapping a planarUWB antenna onto a curved surface to form a spoon shape for sizereduction. Both test antennas were mounted on the metallic mockcellula~r phones as shown in Figure 12, with dimensions of40 x 20 x 95 mm (L x W x H) and rounded corners.

In the simulation of the near-field scanning of the test anten-nas, the delivered power was set to 32 dBm at 900 MHz and to30 dBm at 1880 MHz, in order to remove the return-loss issues inthe comparisons. For the near-field measurements, the DASY3system [20] was used, with the available power at each antennaterminal set to 32 dBm at 900 MHz and 30 dBm at 1880 MHz.

The simulated and measured VSWrR values are compared inFigure 11. The dual-band PlEA had resonances at 900 and1880 MHz. The narrow impedance bandwidth about the resonantfrequencies suggested a rather high-Q structure. The intendedoperating frequencies for the spoon UWB antenna were from900 MHz to 10 GHz (VSWR < 4), providing a low-Q alternative.The dual-band PlEA had a lower VSWR than the spoon UWVB at900 MHz, while the VSWR was similar at 1880 MHz.

The simulated radiation patterns of the test antennas areshown in Figure 13. At 900 MHz, both test antennas had omnidi-rectional radiation patterns and similar directivity values: DMAX(dual-band PIFA) =2.43 dB and DMAX (spoon UWB) = 2.39 dB.However, at 1880 MHz, the radiation patterns were bidirectional,resulting from interaction with the cell phone. As seen in [21],even when the cellular phone with a dual-band antenna was mod-eled with great detail, the radiation pattern around 1800 MHz wasdistorted. The whole structure, including the phone body, acts as anunbalanced pseudo-dipole.

The mock test phones did not have a speaker, so the scanningarea with nine grids, as in the HAG standard, could not be defined.Therefore, the near-field strength was scanned over the entirestructure at a distance xdist, as shown in Figure 12. The xz cut ofthe radiation pattern for each antenna in Figure 14 indicated thatthe far-field patterns were close at 900 MHz, and deviated some-what at 1880 MHz (1.23 dB).

The simulated and measured near-field scanning results arecompared in Figures 15 to 18. In all cases, the spoon UWB antennashowed a lower peak near-field strength. The simulated and meas-ured peak field values differed because loss was not considered inthe simulation. However, the overall field distributions for thesimulations and the measurements were very similar, and demon-strated the same improvement using a UWB antenna structure.

The differences in the peak field strength are summarized inTable 4. At 900 MHz, the peak difference values of the measure-ments were overestimated because the mismatch loss in the meas-urement environment was included. The higher VSWR for themeasured values at 900 MHz corresponded to about 1.5 dB worsereturn loss; a 1.5 dB correction was included in Table 4 to accountfor this return-loss difference. In addition, as explained earlier, themaximum directivity of the spoon UWYB antenna in the measure-ment plane was lower than that of the dual-band PIFA by 1.23 dBat 1880 MHz. As a result, the difference in the peak field value at1880 MHz exceeded that at 900 MHz. In this particular experi-ment, both simulations and measurements suggested that the peak

1-41 and HIf of the spoon UWB antenna (low-Q antenna) wereapproximately 5 dB and 4 dB below the values for the dual-bandPIFA (high-Q antenna), respectively.


Vr----- Ous.b IandI PE-FIA (90 M h Itt)

20i--Dua5berndPEA (1880 MIz)--o Spoon UW 8 (O00 MHz)

0 30 60 90 120 150 180 210 240 270 300 330 3608 [POgW")

Figure 14. A comparison of the radiation directivity for the

dual-band PIFA and the spoon UWB antenna in the xz plane,

computed at 900 and 1880 MHz.

*8007208405M0

400320

240

Figure 15a. The simulated near-field distribution for the dual-

band PIFA at 900 MHz.

E, [Vh.) Oaat4wnd PUA

551 10.746

493 0.665

435 0.60

3190.4

Figre16.Temauenerfeddsrbtofothda-

badPFM t90Mz


Figure 15b. The simulated near-field distribution for the spoonUWB antenna at 900 MiHz.

H551 U493

435

377

319

261

103

145

87

29

Figure 16b. The measured near-field distribution for the spoon

UWB antenna at 900 MHz.

59

ETt IV/-]

.540

480400320I240

[00G10

Table 4. A comparison of the difference in peak field strengths for the dual-band PITA andspoon UWB antennas (APeak IrieldI = Peak IifeldIdual-bd PIFA -Peak Ifieldj~po UW)

900J MHz

(Pdel = 32 dBm)

Simulated 1MeasuredI

1880 MHz

(Pdel 30 dBm)900 MHz

(Paaii = 32 dBm)1880 MHz

(P.vaii = 30 dBm)

A Peak ElI (V/rn) 5.2 dB 6.8 dB 7.1 dB -1.5 dB - 5.6 dB1 6.5 dB

A Peak 1)l (M/m) 4.4 dB { 5.6 dB 5.6 dB -1.5 dB-=4.1ldB 4.0 dB

_o- PekIE . ea.I P WPekEDua~bad PIFA -Peak I EISpoo~n UWB* --A.- Peak HID H n pFA-Peak HI5 ,LFWVB

* Radius of Antenna Spherel* for Dual-band PIFA I

3 I,

~2

* * :~Radius of AntennaISphere for Spoon UWB

Cl) -A,

0- 2101 102

X dit [mm]

Figure 19a. A comparison of the difference in the computedpeak field strength between the dual-band PITA and the spoonUWB antenna at 900 M~z.

7-.Peak I E I Dua-b PIFA -Peak IjE ISpo LPNB

* --A- Pek I~Dualband PIA- Peak ''HSponLIRadius of Antenna SpeeII

S..~~ s rDual-band PIFA i

.~ 2 I!Radius of Antenna2- Sphere for Spoon UWB

02101 10

X dit [mm]

Figure 19b. A comparison of the difference in the computedpeak field strength between the dual-band PIFA and the spoonUWB antenna at 1880 Mlhz.

The peak field-strength difference as a function of distancefrom the antenna (xdi,,) for both test antennas is plotted in Fig-ure 19. As previously explained, the overall structures of the testantennas on the mock phones acted as asymmetric pseudo-dipoles.The effective radii of the antenna spheres for the spoon UWB andthe dual-band PIFA were 65.6 mmn and 61.3 mim, respectively. The

60

effect of using a low-Q antenna is to reduce the nonradiatingenergy within the antenna sphere. The field-strength difference at1880 MHz did not approach zero as expected, due to a 1.23 dB dif-ference in the directive gain off the side of the phone.

6. Radiation Efficiency in the Presenceof the Visible Human Head

When a cellular phone is used in close proximity to a humanhead, dielectric-loading effects can be expected. There may also bea detuning issue for narrowband antennas (high-Q antennas). Chen[22] reported that the impedance perfornance of a UWB antennawas only slightly affected by the presence of a human head. Duringnormal cellular-phone use, the human head is inside the antennasphere and is exposed to strong near fields. The strong near-fieldenergy increases the interaction with the human head, and can alterthe input impedance. It is reasonable to expect that UWVB antennaswould not experience as much detuning; as found with narrowbandantennas, due to the reduced near-field energy and associated inter-action with adjacent materials. Similarly, the efficiency of the low-Q antenna should be higher than that of a high-Q antenna whenincluding human-head effects.

In order to demonstrate improved performance of the low-Qantenna in the presence of a human head, simulations were per-formed using the test antennas on mock phones in the presence ofthe NIH (National Institutes of Health) head model, called the"Visible Human Head" [23]. A commercial Moment-Method/Finite-Element-Method hybrid code [8, 24] was used forthe simulations. The male human head was solved with the Finite-Element Method (tetrahedral meshes, 4 mm resolution), using thehead CAD model available from EM Software & Systems [8]. Asshown in Figure 20, the CAD model of the human male head wascomposed of 24 biological tissues. The electrical properties of thebiological tissues could be found on the Web site of the ItalianNational Research Council [25]. The test antennas on the mockphone were modeled with the Moment Method, neglecting ohmicloss. The test antennas were located 10 mmn from the male humanhead, as specified in the HAC standard. The simulated radiation-gain patterns at 900 and 1880 MHz, shown in Figure 21, weredirectional, due to head interaction. The antenna efficiency wasreduced by power absorption in the human head, and is summa-rized in Table 5. In this particular configuration of the cellularphone and human head, the spoon UWB antenna showedimprovements of 8% and a 20% in efficiency at 900 and1800 MHz, respectively.


rS IV*i ub ~lIm

320240

1 6 0

111,80Z2

Figure 17a. The simulated near-field distribution for the dual-band PIFA at 1880 MHz.

Figure 17b. The simulated near-field distribution for the spoon

UWB antenna at 1880 MHz.

Allo

.0cc. - Manuw

49mwa- Whit Mowtr

,Eye, lltmt

'FaM

ILYmphMucou MOWWMui

omwcwo04a"eSToudo

ToothVbccus Humor

Figure 18a. The measured near-field distribution for the dual-band PIFA at 1880 MHz.

]E. [Vt.

551

435

201

M0

145

V7

29

lA~I~

OJ24

#A"

&Ms

W.21

*AM0

Figure 18b. The measured near-field distribution for the spoonUWB antenna at 1880 MHz.

XY4US

x

ouA YZ-"u

Figure 20. The visible human male head model used for the

simulations.


551

435

S'"

3"19

241

145

07

29

61

xr. D

I

0.746

0.005

#A44

0.121

0"0

k:

I

Table 5. A comparison of the radiation performance for dual-band IPIFA and spoon UWB antennas at 900 and 1880 MHz,obtained by simulations with a human-head model.

I- 900 MHz 1880 MHzDual-band PIFA ISpoon UWB

Dual-band PIFA Spoon lIWf

7. Conclusions

A method for reducing near-field electromagnetic energyaround a cellular phone was presented, to mitigate the interactionbetween a cellular phone and a hearing aid. Both radiation mecha-nisms and the theory of the fundamental limit on antenna size wereused to understand the HAC problem. Examining the radiationprocess showed that current-slope discontinuities caused high near-field intensities. The fundamental-limit theory of antennas indi-cated that the problems could be reduced by using low-Q antennas,such as an ultra-wideband (UWB) antenna.

The low-Q antenna approach to reduce the near-field strengthwas demonstrated through both simulations and measurements.Simulations showed that the peak electric and magnetic near-fieldamplitudes were respectively 7 dBV/m and 4 dBA/m lower for theplanar UWB fat dipole (low Q) compared to a narrowband dipole(high Q). For more practical cellular-phone antennas, the simula-tion and measurement results were obtained for dual-band planarinverted-F (900 and 1880 MHz, high Q) and ultra-wideband(900 Mffz to 10 GHz, low Q) test antennas, mounted on mockphones. This showed that the peak electric and magnetic near-fieldstrengths were respectively reduced by at least 5 dBV/m and4 dBA/m for the UWB test antenna. Near-field strength reductionwas achieved without sacrificing far-field performance. In thepresence of a human head, simulation results also showed that theradiation efficiency of the ultra-wideband test antenna was betterthan that of the dual-band PIFA antenna.

The low-Q antenna approach for reducing near-field strengthand spreading the current distribution to reduce discontinuity peakscan be also applied to a variety of other applications. These includesafety issues of RE energy exposure, such as high-power radiatingsystems and medical electronic systems, and the efficiency of theantenna system. Future research on low-Q antennas is needed toreduce the antenna size while retaining the wide bandwidth andlow-Q nature of the antenna. This need is critical in an industrythat must consider the value of aesthetics and wide-scale produc-tion.

8. Acknowledgment

This research was partially supported by Sony Ericsson USA.

9. Appendix

In this appendix, we present a derivation of Equation (3)based on the work of Davis et al. [16].

62

The electric and magnetic fields of the spherical TM01 mode,under the assumption of Ikdl = 4zr (where dI is the length of theinfinitesimal dipole, k is the wavenumber, and I is the current),can be written in time-harmonic form, with e1i" time dependence[12,26], as

E(r, t) = 2q cosof sin(cot-kr) cos(wt-kr)j

-9~ro~w i) sin0in(wt -kr) cos (wt -kr)1r 'Cr k2

(5)

isrt)~in Cos(cot -kr) + sin (wt -krjjr I kCr]I'

(6)

where 77 is the intrinsic impedance of free space. The total tran-sient power through the spherical surface at distance r is given by

P(r,t)=# txfl.Fds

0000 2r 2 'Cr)

+ 2 1jin2~(cot-kr)} r2 sin Od~dtil

3~i [ (k,)2I r) k, (kr)3 3 i2atk)

(7)

Unfortunately, this expression does not clearly represent the sepa-ration between the radiated and reactive powers. Equation (7) maybe rearranged to give

P(r~t){4r[l+cos2(wt-.kr•ý)]+ 13 sin2(cot-kr-;)}

(8)

where =atan (1/kr) is the excess phase delay of the radiatedfield. The first and second terms of Equation (8) are related to theradiated power (cos 2 (-)), and the third term is related to reactivepower (sin (.)cos (.)). These radiated and reactive powers are 900


Gain Tot[dB]I1ý2-237-.41

-1215-14.6

16

185

Gain TOttd8

-2.8-519_-91

1-15.3

1-153~-18.51-21,6-.241-27.8

z

Figure 21a. The computed radiation gain pattern in the pres-ence of the visible human male head at 900 MHz for the dual-band PIFA (Gm = 1.25 dB).

Gain TatId~ll

-0 3-2.2-4.261

118.

[17.6

Z

Figure 21b. The computed radiation gain pattern in the pres-ence of the visible human male head at 900 MHz for the spoonIJWB antenna (Gm = 1.60 dB).

Figure 21c. The computed radiation gain pattern in the pres-ence of the visible human male head at 1880 MHz for the dual-band PIFA (Gm,,, 3.43 dB).

Gain Tot~dBJ

I 5-1 ý8

-11

_39

9z

Figure 21d, The computed radiation gain pattern in the pres-ence of the visible human male head at 1880 MHz for the spoonLJWB antenna (G,,,,, = 5.49 dB).

IEEE Antennas and Propagation Magazine, Vol. 50, No. 3, June 2008 663

out of phase, which is consistent with the circuit definitions of realand reactive power. The time average of the radiated power termsin Equation (8) is

Pa 747r3

10. References

1. National Institute on Deaffness and Other Communication Disor-ders (NIDCD), "Statistics About Hearing Disorders, Ear Infections,and Deafniess," available athttp://www.nidcd.nih.gov/health/statistics/heaiing.asp.

2. S. Berger, "Hearing Aid and Cellular Telephone Compatibility:Working Toward Solutions," Journal of the American Academy of'Audiology, June 2001, pp. 309-3 14.

3. American National Standards Institute, American National Stan-dard for Method of Measurement of Compatibility between Wire-less Communications Devices and Hearing Aids (ANSI C63.19-2006), New York, American National Standards Institute, 2006.

4. T. Victorian, "Hearing Aid Compatibility: Technical Update,"Audiology Online, 2004, available at http:I/www.audiologyonline.com/articles/arc-disp.asp?article-id=1 263.

5. M. Skopec, "Hearing Aid Electromagnetic Interference fromDigital Wireless Telephones," IEEE Transactions on Rehabilita-tion Engineering, 6, 2, June 1998, pp. 235-239.

6. M. Okoniewski and M. A. Stuchly, "Modeling of Interaction ofElectromagnetic Fields from a Cellular Telephone with HearingAids," IEEE Transactions on Microwave Theory and Techniques,MTT-46, 11, November 1998, pp. 1686-1693.

7. K. Caputa, M. A. Stuchly, M. Skopec, H. I. Bassen, P. Ruggera,and M. Kanda, "Evaluation of Electromagnetic Interference from aCellular Telephone with a Hearing Aid," IEEE Transactions onMicrowave Theory and Techniques, MTT-48, 11, November 2000,pp. 2148-2154.

8. EM Software & Systems, FEKO Suite v5.1, available athttp://www.feko.info.

9. H. Dillon, Hearing Aids, New York, Thieme, 200 1.

10. F. K. Kuk and K. H. Nielson, "Factors Affecting Interferencefrom Digital Cellular Telephones," The Hearing Journal, 50, Sep-tember 1997, pp. 32-34.

11. T. A. Victorian, "An Update on Digital Cellular TelephoneInterference and Hearing Aid Compatibility," The Hearing Jour-nal, 53, September 1998, pp. 53-60.

12. W. L. Stutzman and G. A. Thiele, Antenna Theory and Design,Second Edition, New York, John Wiley & Sons, 1998.

64

13. R. W. P. King, "Cylindrical Antennas and Arrays," in R. E.Collin and F. J. Zucker (eds.), Antenna Theory, Part 1, New York,McGraw-Hill, 1969, Chapter 9.

() 14. L. J. Chu, "Physical Limitations on Omni-directional Anten-() nas," J Appi. Phys., 19, 1948, pp. 1163-1175.

15. H. A. Wheeler, "The Radiansphere Around a Small Antenna,"Proceeding of the IRE, 47, August 1959, pp. 1325-133 1.

16. W. A. Davis, T. Yang, E. D. Caswell, and W. L. Stutzman,"Fundamental Limits on Antenna Size: A New Limit," submittedto IEEE Transactions on Antennas and Propagation.

17. H. G. Schantz, "Electromagnetic Energy around HertzianDipoles," IEEE Antennas and Propagation Magazine, 43, 2, April2001, pp. 50-62.

18. T. Yang, S.-Y. Suh, R. Nealy, W. A. Davis, and W. L.Stutzman, "Compact Antennas for UWvB Applications," IEEEAerospace and Electronic Systems Magazine, 19, May 2004, pp.16-20.

19. J. Ollikainen, 0. Kivekas, A. Toropainen, and P. Vainikainen,"Internal Dual-Band Patch Antenna for Mobile Phones," Proceed-ings of the AP2000 Millennium Conference on Antennas andPropagation, April 2000.

20. SPEAG, DASY3 Systems, available at http://www.speag.comlen/measurement/dasy5/index.php.

21. N. Chavannes, R. Tay, N. Nikoloski, and N. Kuster, "Suitabil-ity of FDTD-Based TCAD Tools RF Design of Mobile Phones,"IEEE Antennas and Propagation Magazine, 45, December 2003,pp. 52-66.

22. Z. N. Chen, A. Cai, T. S. P. See, Qing Xianming, and M. Y. W.Chia, "Small Planar UWB Antennas in Proximity of the HumanHead," IEEE Transactions on Microwave Theory and Techniques,MTT-54, 4, June 2006, pp. 1846-1857.

23. Visible Human Project, US National Library of Medicine,available at http://www.nlm.nih.gov/research/visible/visible human.html.

24. F. J. C. Meyer, D. B. Davidson, U. Jakobus, and M. A. Stuchly,"Human Exposure Assessment in the Near Field of GSM Base-Station Antennas Using a Hybrid Finite Element/Method ofMoments Technique," IEEE Transactions on Biomedical Engi-neering, 50, 2, February 2003, pp. 224-233.

25. Italian National Research Council, "Dielectric Properties ofBody Tissues," available at http://niremf.ifac.cnrit/tissprop/htmlclie/htmlclie.htm#atsftag.

26. R. F. Harrington, Time-Harmonic Electromagnetic Fields, NewYork, Wiley-IEEE Press, 2001.


Introducing the Feature Article Authors

Taeyoung Yang received his BS and MS degrees in Elec-tronic Engineering from Sung-kyun-kwan University in SouthKorea in 1997 and 1999, respectively. For his MS thesis, hedesigned a dual linear-polarized PCS base-station antenna using amulti-layer spectral-domain analysis approach, based on anequivalent circuit model. Following graduation, he joined Samsungfrom 1999 to 2000. While a research engineer at Samsung, he con-tributed to the development of miniaturized RE passive compo-nents, such as LTCG/HTCC filters and SAW duplexers for cellularphones and mobile devices. In 2001, he enrolled at Virginia Techand received an MSEE in 2003. Currently, he is a PhD candidate,working on the fundamental limits of antenna performance andmitigation of near-field interactions. Since 2002, he has been aresearch assistant in the Virginia Tech Antenna Group (VTAG),involved in advanced antenna research projects. During a 2005summer internship at Intel, he investigated potential interferenceissues between multiple radios on mobile platforms. He served as asession chair of the 2005 IEEE International Symposium onAntennas and Propagation, and currently serves as a reviewer forthe International Journal of Antennas and Propagation. In June2007, he was the recipient of the Best Poster Award at the WirelessPersonal Communication Symposium.

William A. Davis received the PhD degree from the Univer-sity of Illinois, Urbana, in 1974. He has held academic positions atboth the Air Force Institute of Technology and Virginia Polytech-nic Institute and State University, Blacksburg, where he is currentlya Professor and Director of the Virginia Tech Antenna Group. Hisprimary interests are antennas, measurements, scattering theory,and material characterization, with an emphasis on numericalmethods, electromagnetic theory, and microwave measurements.Dr. Davis specializes in antenna analysis and design. He has beenstrongly involved in numerical methods for electromagnetics, aswell as microwave measurements and material characterization.His current emphasis is on fundamental limits of antennas, UWBantennas, and applying UWB concepts to general antenna design.Related to the fundamental limits and UWB antenna developmentis basic work on modeling using related aspects of the SingularityExpansion Method (SEM) for representing antennas using polesand residues. This work has led to fundamental models for anten-nas based on measurements or simulations.


Warren L. Stutzman received his BS in Electrical Engineer-ing and AB in Mathematics degrees from the University of Illinoisin 1964, and received MS and PhD degrees in Electrical Engi-neening from Ohio State University in 1965 and 1969, respectively.Dr. Stutzman has been on the Electrical Engineering faculty of

Virginia Polytechnic Institute and State University since 1969. Heserved as the Director of the Antenna Group from its beginning in1983 until 2001. He was the Interim Department Head in 2000-01,Associate Head in 2001-02, and Interim Head from January 2003to July 2004. In 1983, he was a Visiting Professor at the PhysicalScience Laboratory of New Mexico State University. He is cur-rently a Research Professor in the Antenna Group.

Dr. Stutzman has worked in several areas of antennas andpropagation. His research activities include antennas for wirelesssystems, propagation, reflector antennas, phased-array design, andatmospheric effects on Earth-space communication links. He hasbeen involved in research totaling over $25,000,000 from severalsponsors in government and industry. He has authored over 80journal articles and made over 200 presentations at meetings. He iscoauthor, with Gary A. Thiele, of the textbook Antenna Theory andDesign (John Wiley, 1981 and 1998), and author of Polarization inElectromagnetic Systems (Artech House, 1993). He has twice wonthe annual Wheeler prize for the best applications paper in theIEEE Transactions on Antennas and Propagation.

He is a Life Fellow of the IEEE, and served as President ofthe IEEE Antennas and Propagation Society in 1992.

Minh-Chau T. Huynh received his BS, MS, and PhDdegrees in Electrical Engineering in 1997, 2000, and 2004, respec-tively. All degrees were from Virginia Polytechnic Institute andState University. He has been with Sony Ericsson Mobile Commu-nications (USA), Inc., since 2005. His work and research activitiesinclude small low-profile antennas, multi-band antennas forGSM/UMTS systems, and antenna-measurement techniques. Ir.

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Cellular-Phone and Hearing-Aid Interaction: An Antenna Solution

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