bandpass filter

98 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 61, NO. 1, JANUARY 2013

Microstrip Dual-Band Bandpass Filter Design WithClosely Specified Passbands

Cheng-Ying Hsu, Student Member, IEEE, Chu-Yu Chen, Member, IEEE, and Huey-Ru Chuang, SeniorMember, IEEE

Abstract—An efficient and practical design method for adual-band bandpass filter (BPF) is presented. The electricalspecifications of the filter, such as the center frequency, band-width, and transmission zero location, are controllable andadjustable. Once, the desired ratio of the resonant frequency ofthe two bands is given, the characteristic impedance of the linecorresponding to the specific ratio can be accurately determinedfrom the design curve. Compared with the conventional half-and quarter-wavelength stepped-impedance resonators (SIRs),the proposed unequal-length shunted-line stepped- impedanceresonator (shunted-line SIR) can provide an efficient way torealize the dual-band filter, especially when two passbands areclosely spaced. Furthermore, the multitransmission zeros and onecontrollable transmission zero are generated near the passbandsto improve the out-of-band rejection. The design examples of mi-crostrip dual-band BPFs operating at 2.2/3.45 and 1.95/2.65 GHzwith equal absolute bandwidths and high isolation are demon-strated to validate of the design method. In the illustrated cases,a small frequency ratio of 1.3 between two resonant frequenciescan be achieved. The proposed design method is very useful forthe dual-band BPF design, especially when the two passbands arevery close.

Index Terms—Bandpass filter (BPF), dual-band, microstrip,open-loop ring resonator (OLRR), unequal-length shunted-linestepped-impedance resonator (shunted-line SIR).

I. INTRODUCTION

R ECENTLY, multiband filters have been proposed and ex-ploited extensively as a key circuit block in multiband

wireless communication systems [1]–[3]. Among these designs,several papers have focused attention on multiband filters withcontrollable center frequencies, responses, or bandwidths. Dual-band filters can be designed easily using stepped-impedance res-onators (SIRs) because the spurious responses of the filters canbe controlled by properly adjusting the impedance ratios andelectrical lengths [4]–[6]. A dual-band bandpass filter (BPF)

Manuscript received June 05, 2012; accepted September 17, 2012. Date ofpublication December 03, 2012; date of current version January 17, 2013.C.-Y. Hsu and H.-R. Chuang are with the Institute of Computer and Commu-

nication Engineering, Department of Electrical Engineering, National ChengKung University, 413 Tainan, Taiwan, R.O.C (e-mail: [email protected]; [email protected]).C.-Y. Chen is with the Department of Electrical Engineering, National Uni-

versity of Tainan, 413 Tainan, Taiwan, R.O.C (e-mail: [email protected]).Color versions of one or more of the figures in this paper are available online

at http://ieeexplore.ieee.org.Digital Object Identifier 10.1109/TMTT.2012.2222912

with adjustable bandwidths using a uniform coupled line is pre-sented in [7]. The dual-band responses of the filter can be ac-curately synthesized using dual-band invertors with shunt res-onators. The embedded resonator method is also proposed torealize the dual-band bandpass filter [8]. The main resonatorwith two embedded resonators can excite two passbands forthe dual-band filter design. However, it was discovered that thesuppression between the two passbands is inadequate becausethe second spurious resonant frequency is adopted to achievethe second passband; the first spurious response is then excitedat mid-band. The dual-band BPFs using embedded spiral res-onators are presented in [9]. The extra passband can be ex-cited using the embedded spiral resonator, and the desired re-sponses of the passbands can be designed using the extractedcoupling coefficients and external quality factors. A dual-bandBPF using the parallel short-ended feed scheme is presentedin [10]. The proposed feed scheme can provide two couplingpaths for the RF signals, and the dual-band responses oper-ating at 1/1.44 GHz can be generated using two sets of thequarter-wavelength resonators operating at diverse frequencies.In the study of [11], a dual-band microstrip BPF using net-typeresonators operating at 1/2.05 GHz is proposed. Using the extracoupled resonator sections, the elliptic function response is ob-tained. Thus, the filter has a high mid-band rejection betweenthe two passbands. A design method of the dual-band BPF usinglow-temperature cofired ceramic (LTCC) technology is reportedin [12]. Three transmission zeros are generated to improve theskirt selectivities of the two passbands. The synthesis methodfor the dual-band BPF with fully controllable second passbandis proposed in [13]. The complete design procedures were in-troduced, and the limitations of the ratio of the second band tothe first band are currently being studied. A synthesized methodfor the dual-band BPF based on generalized branch-line hy-brids is proposed in [14]. The stub-loaded resonator can providea multiple-mode performance which has been used to designthe ultra-wideband filters [15], [16]. Furthermore, the open-loopresonator with loaded stubs can be applied to dual-band BPF de-sign [17], [18]. In [17], the filters with three different frequencyratios of the second band to the first band ( 1.6, 2.18,and 2.36) with different fractional bandwidths are designed todemonstrate the design procedure. In [18], a dual-band BPFwith a high suppression between the two passbands is demon-strated; the ratio of is approximately to 1.58. The dual-band BPFs using the multistage SIRs with a wide upper rejec-tion band are presented in [19]; the ratio of is 2.16 and2.5, respectively.

0018-9480/$31.00 © 2012 IEEE

HSU et al.: MICROSTRIP DUAL-BAND BBP DESIGN WITH CLOSELY SPECIFIED PASSBANDS 99

Fig. 1. Structures of three types of shunted-line SIRs: (a) half-wavelength, (b)quarter-wavelength, and (c) half-wavelength unequal length.

It is observed that most of the reported dual-band filterdesigns mainly focus on wide stopband, high isolation, or lowinsertion loss. A high performance dual-band BPF with closelyspecified passbands is seldom investigated. Waveguide filterscan easily design such a filter since the waveguide resonatorwith a very high unloaded can enable high selectivityand low insertion loss [20], [21] in filter design. A printedcircuit board (PCB) dual-band filter based on a defectedstepped-impedance resonator (DSIR) is presented in [22]. Byslotting two pairs of resonators, two passbands can be individu-ally designed to achieve a dual-band filter with close passbands

. However, the additional backside fabricationprocess is required.This paper presents a new design method for a planar mi-

crostrip dual-band BPF with two closely specified resonant fre-quencies using the unequal-length shunted-line SIR. The ratioof the second resonant frequency to the first resonant frequency

can be calculated from the even- and odd-mode resonanceconditions of the resonator. Section II investigates the behav-iors of the shunted-line SIRs, and Section III further explainsthe design procedures of the shunted-line SIR dual-band BPFand the control mechanism of transmission zeros. Finally, thetwo examples of microstrip dual-band BPFs with the closelyspecified passband ( 1.5 and 1.3) are demonstrated toverify this study. Section IV discusses the simulation and mea-surement results.

II. ANALYSIS OF SHUNTED-LINE SIR

Three types of shunted-line SIR called Type-A, Type-B,and Type-C are illustrated in Fig. 1. Type-A is an equal-lengthshunted-line SIR which is reported in [23]. The characteristic ofthe symmetric shunted-line SIR is the same as the conventionaltwo-section SIR when the two open-ended transmission lines atboth ends have the characteristic impedance of . Similarly,the conventional quarter-wavelength SIR called Type-B can be

Fig. 2. Relationships between impedance ratio and normalized spurious reso-nance frequencies.

equivalent to the quarter-wavelength shunted-line SIR illus-trated in Fig. 1(b). An unequal-length shunted-line SIR calledthe Type-C resonator is presented and shown in Fig. 1(c). Thestructural difference between Type-A and Type-C is that the un-equal-length shunted-line SIR is loaded with an unequal-lengthopen-ended line at both ends. The resonance conditions of theType-C resonator can be obtained from the input impedanceobserving the symmetric plane of the resonator as follows.

For odd resonant modes:

(1)

For even resonant modes:

(2)

The fundamental resonance frequency can be determinedfrom (1) and the second resonance frequency can be deter-mined from (2). From the above equations, the ratio ofcan be derived as follows:

(3)

The relationships between the impedance ratios andfor these three types of SIR are plotted in Fig. 2. Com-

pared with the half-wavelength resonator, the quarter-wave-length resonator (Type-B) shown in Fig. 1(b) has the advantageof small size when they are operated at the same fundamentalresonance frequency . However, the second resonancefrequency of the uniform quarter-wavelength resonator

is generated at three times of the fundamentalresonant frequency. Based on the analysis of the conventionalSIR theory, a dual-band filter with closely specified passbandscan be achieved using an extremely high impedance ratio. Asshown in Fig. 2, the Type-A resonator is more suitable for the


Fig. 3. Schematics of second-order (two-pole) dual-band BPF.

Fig. 4. Shunted-line SIR dual-band filters with the parallel-coupled structures.

Fig. 5. Simulation of the dual-band filters with different feed-line structure.

dual-band operation when the ratio of is low. However,Type-A resonator may cause some problematic issues if thepassbands are very close. For example, if is chosen to be1.25, the impedance ratio of must be approximately0.1. The non-negligible parasitic effects from the large stepchange may be generated. Compared with the resonators ofType-A and Type-B, the unequal-length shunted-line SIRmarked as Type-C can provide a lowest ratio of withthe same impedance ratio of . Moreover, the Type-Cresonator provides more degrees of freedom for bandwidth con-trol. The design methodology will be discussed in Section III.

III. DESIGN OF DUAL-BAND FILTER USING UNEQUAL-LENGTHSHUNTED-LINE SIR

A. Determination of Coupling Coefficient and ExternalQuality Factor

To demonstrate the applications of the studies in the previoussections, the design procedures of the dual-band filters usingthe unequal length shunted-line SIR are described. As shown in

Fig. 6. Transmission zero investigation of the sensitivity to tapping position.

Fig. 7. Frequency responses of dual-band BPF using parallel-coupled structure( in all cases).

Fig. 3, the parameters such as quality factors and coupling coef-ficients of the first and second band are required to be carefullydesigned, respectively. In this paper, the lowpass prototype withChebyshev response is adopted in all cases. Thus, the externalquality factors and coupling coefficients for the two passbandscan then be calculated from the low-pass prototype parametersas follows:

(4)

(5)

for to (6)

where and are the external quality factors of the res-onators at the input and output, respectively, and is thecoupling coefficient between the adjacent resonators. The ab-breviation is defined as the fractional bandwidth.


Fig. 8. Performance of different microstrip coupled-line structures.

Using EM simulator, it may be easier to find the couplingcoefficient, which is related to the odd-mode and even-moderesonance frequency for those coupled resonators. The couplingcoefficient can be calculated by using the following equation:

(7)

where and are the even- and odd-mode resonant frequen-cies, respectively. Thus two splitting modes can be observedfrom the simulation results. Additionally, the external qualityfactor is related to the tap position. The external quality factor

can be extracted from the simulated frequency response as

(8)

where is the absolute bandwidth between thepoints and the is the resonant frequency. Therefore, the ini-tial structural parameters of an unequal length shunted-line SIRcan be determined from Type-C resonator. The desired ratio ofthe can be calculated from Fig. 2 when the impedanceratio is determined.

B. Layout Arrangement and Transmission Zeros

Generally, several layout arrangements with different cou-pling and feeding structures are considered to realize a dual-band performance. Two different feeding structures includingtapped-line and coupled-line feed for different coupling mech-anisms are investigated. The filter using parallel-coupled struc-tures is shown in Fig. 4. The performance of the filter using atapped line and a coupled line are further shown in Fig. 5. Itis observed that, using the tapped feed line, an extra transmis-sion zero is excited between the two passbands. Furthermore,Fig. 6 shows the effect of the tap position on the location ofthe mid-band transmission zero. It is observed that, when thetapping position is close to the center of the resonator, thetransmission zero is shifted to a lower frequency without de-grading the performances of passbands. Fig. 7 shows the effect

Fig. 9. Schematic of proposed dual-band BPF using unequal-length shunted-line SIR.

of the impedance ratio of on the center frequencies oftwo passbands. It is observed that the larger impedance ratio of

results in the increase of the separation of the two pass-bands. The above observation closely agrees with the theoreticalresults. However, the selectivity and out-of-band rejections arenot sufficient for use. A straightforward approach to improve theband selectivity is to increase the filter order, but it will increasethe insertion loss and size. The selectivity and out-of-band re-jection can be effectively improved using the following coupledstructure.Four kinds of microstrip coupled structures shown in Fig. 8

are analyzed. From the simulations, the type-(i) with full-length coupling generates a transmission zero located at the highband rejection. The type-(ii) with partial-length coupling hasno transmission zero near the passband. The type-(iii) is a par-tial-length coupled line with a tapped feed line. It can generate atransmission zero at the low-band rejection. The type-(iv) called


Fig. 10. Even- and odd-modes distribution of the dual-band filter usingshunted-line SIR with 0 feeding structure.

both ends a coupled structure which can generate two trans-mission zeros at low and high band rejection and can furtherimprove the selectivities. In this paper, the type-(iv) configura-tion is used. The final schematic of the proposed dual-band BPFusing the unequal-length shunted-line SIR is shown in Fig. 9.Note that the and are the coupled and uncoupled sec-tions, respectively; and the total length is the sum of and. The length is designed as a quarter-wavelength open-

ended coupled line at the corresponding resonant frequency.To observe the characteristics of the spurious responses, two

simulations are plotted and compared. As shown in Fig. 10, theupper figure shows that even- and odd-mode distributions ofthe dual-band filter using a shunted-line SIR with 0 feedingstructure are determined through the simulation of the inputimpedance seen from the symmetric plane of the singleresonator. When two same type of resonators are coupled toeach other, the corresponding responses of the dual-bandBPF are shown in the lower part of Fig. 10. It is found that thesecond odd-mode and second even-mode resonant frequenciesare excited near 3.5 and 5.5 GHz, respectively. Also, adjustingthe length of the coupled-line section can control the loca-tion of the transmission zero and suppress the specified band.Fig. 11(a) and (b) shows the transmission-zero sensitivity versusthe coupled length of a dual-band BPF. It is clearly observed thata transmission zero moves to the higher frequency whenbecomes shorter. In particular, a sharper and deeper rejection

between the two adjacent passbands can also be achieved usingthe longer . Moreover, the passband performances are notdegraded by the shifting of such a transmission zero. The spec-ifications of the designed dual-band BPFs are listed in Table I.The of Filter-I and Filter-II are selected to be 1.5 and1.3, respectively. Note that Filter-II and Filter-III have the samespecifications but different locations of the transmission zero

. The multitransmission zeros of , , and arecreated by the 0 feeding structure, and the transmission zero of

is produced by the coupled-line section . Three exam-ples of dual-band filters are designed to have the same absolute

Fig. 11. Simulated frequency responses of dual-band BPF using differentlengths of parallel coupled-line section : (a) passband responses and (b)out-of-band responses. (All electrical lengths are defined at .)

bandwidths in both passbands, which are 80, 50, and 50 MHz,respectively. The coupling coefficients and external quality fac-tors are further listed in Table I.To summarize the design steps of the filters, a design flow

chart for the dual-band BPF based on coupled-shunted-line SIRsis shown in Fig. 12. The desired external quality factors of thetwo passbands can be controlled by adjusting , , and . Thecoupling coefficients are mainly related to the parametersand .

IV. SIMULATION AND MEASUREMENT RESULTS

The filters were fabricated on a Duroid RT6010 substrate withrelative permittivity of 10.2 and dielectric height of 25 mil.The EM solver IE3D is used for simulation and fine-tuning.Figs. 13 and 14 show the simulated current distributions andphotographs of the fabricated dual-band BPFs, respectively. It


TABLE ISPECIFICATIONS OF THE DESIGNED DUAL-BAND FILTERS

is observed that the current density in the high-impedance sec-tion is higher than those in the low-impedance center sectionand, hence, can generate a stronger magnetic field to enhancethe coupling strength between the resonators. The parametersof Filter-I are 18 , 49.5 , 38 ,

11.6 mm, 8 mm, 5.8 mm, 3.2 mm,3.7 mm, and 0.3 mm, where is the physical length

of and is the coupling spacing. The filter with 0 feedingstructure shown in Fig. 9 can provide upper and lower couplingpaths for the RF signal. Therefore, the multitransmission zeroscan be generated near the passbands and then further improvethe band selectivity [24]. Note that the 0 feeding structure areused in all fabricated filters. The narrowband and wideband fre-quency responses of Filter-I are shown in Fig. 15(a) and (b). Thefirst passband with a center frequency of 2.2 GHz has 1.8 dBof insertion loss and approximately 14 dB of return loss. Thesecond passbandwith a center frequency of 3.45GHz has 2.4 dBof insertion loss and approximately 15 dB return loss. The trans-mission zero of can also be created by the 0 feeding struc-ture. Both of the transmission zeros of and are appliedto suppress the third harmonic response. The filter has a widestopband from 4.5 to 7 GHz.

Fig. 12. Design flow for dual-band BPF using coupled-shunted-line SIRs.

Fig. 13. Simulated current distributions.

Fig. 14. Photographs of fabricated dual-band BPFs: (a) filter-I, (b) filter-II, and(c) filter-III.


Fig. 15. Simulation and measurement results of Filter-I: (a) passband responses and (b) out-of-band responses.

Fig. 16. Simulation and measurement results of Filter-II: (a) passband responses and (b) out-of-band responses.

The parameters of Filter-II are 14.5 , 54 ,40 , 12.5 mm, 10.7 mm, 9.5 mm,3.4 mm, 5.2 mm, and 0.3 mm. The narrow-

band and wideband frequency responses of Filter-II are shownin Fig. 16(a) and (b). The first passband with a center frequencyof 1.95 GHz has approximately 1 dB of insertion loss and ap-proximately 20 dB of return loss. The second passband with acenter frequency of 2.65 GHz has 1.7 dB of insertion loss andapproximately 12 dB of return loss. It is observed that the shaperrejection between the two adjacent passbands is generated be-cause the transmission zeros of produced by the longercoupled-line section is shifted to a lower frequency. The secondspurious response excited at 3.5 GHz can be improved throughthe adjustment of , which is demonstrated in Filter-III.

The parameters of Filter-III are 14.5 , 54 ,40 , 12.5 mm, 10.7 mm, 6.7 mm,3.4 mm, 5.2 mm, and 0.5 mm. The narrow

and wideband frequency responses of Filter-III are shown inFig. 17(a) and (b). The first passband with a center frequencyof 1.98 GHz has approximately 1.6 dB of insertion loss andapproximately 11 dB of return loss. The second passband witha center frequency of 2.67 GHz has 2.3 dB of insertion loss andapproximately 12 dB of return loss. The transmission zero of

is applied to suppress the third harmonic response. Thefilter has a wide stop band from 2.8 to 5.4 GHz.In the above discussions, all of the designed filters have low

insertion losses and high suppression between the two pass-bands. Three examples of the dual-band BPFs with closely spec-


Fig. 17. Simulation and measurement results of Filter-III: (a) passband responses and (b) out-of-band responses.

ified passbands and quasi-elliptic function responses have beencompleted. The simulation and measurement results are shownin favorable agreement.

V. CONCLUSION

The unequal-length shunted-line SIR that can be applied toeffectively realize the dual-band BPF is demonstrated in thispaper. Low-resonant frequency ratios of the second passbandto the first passband of the dual-band filter have been empha-sized. Three examples of microstrip dual-band BPFs are demon-strated to validate the design method. In the illustrated cases,the frequency ratios of 1.3 and 1.5 can be achieved. Withoutaffecting the performance of the passband, the sharp rejectionbetween the two adjacent passbands can be created by appro-priately placing the transmission zeros. In addition, by appro-priately selecting the coupled-line section, the upper band re-jection can be further improved. The proposed design methodusing the unequal-length shunted-line SIR is very useful for thedual-band filter design, especially when the two passbands areclosely spaced.

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Cheng-Ying Hsu, photograph and biography not available at the time ofpublication.

Chu-Yu Chen, photograph and biography not available at the time ofpublication.

Huey-Ru Chuang, photograph and biography not available at the time ofpublication.

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