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AN6077.3

An IC Operational Transconductance Amplifier(OTA) With Power Capability

In 1969, the first triple operational transconductance

amplifier or OTA was introduced. The wide acceptance of

this new circuit concept prompted the development of the

single, highly linear operational transconductance amplifier,the CA3080. Because of its extremely linear

transconductance characteristics with respect to amplifier

bias current, the CA3080 gained wide acceptance as a gain

control block. The CA3094 improved on the performance of

the CA3080 through the addition of a pair of transistors;

these transistors extended the current carrying capability to

300mA, peak. This new device, the CA3094, is useful in an

extremely broad range of circuits in consumer and industrial

applications; this paper describes only a few of the many

consumer applications.

What Is an OTA?

The OTA, operational transconductance amplifier, concept is

as basic as the transistor; once understood, it will broaden the

designer's horizons to new boundaries and make realizable

designs that were previously unobtainable. Figure 1 shows an

equivalent diagram of the OTA. The differential input circuit is

the same as that found on many modern operational

amplifiers. The remainder of the OTA is composed of current

mirrors as shown in Figure 2. The geometry of these mirrors is

such that the current gain is unity. Thus, by highly

degenerating the current mirrors, the output current is

precisely defined by the differential input amplifier. Figure 3

shows the output current transfer characteristic of the

amplifier. The shape of this characteristic remains constantand is independent of supply voltage. Only the maximum

current is modified by the bias current.

The major controlling factor in the OTA is the input amplifier

bias current IABC; as explained in Figure 1, the total output

current and gm are controlled by this current. In addition, the

input bias current, input resistance, total supply current, and

output resistance are all proportional to this amplifier bias

current. These factors provide the key to the performance of

this most flexible device, an idealized differential amplifier,i.e., a circuit in which differential input to single ended output

conversion can be realized. With this knowledge of the

basics of the OTA, it is possible to explore some of the

applications of the device.

DC Gain Control

The methods of providing DC gain control functions are

numerous. Each has its advantage: simplicity, low cost, high

level control, low distortion. Many manufacturers who have

nothing better to offer propose the use of a four quadrant

RIN

OTA

2RO

IABC

ein

V+

V-

2RO

+

-

6

4

5

7

gm ein

2

3

IOUT = gm (ein)

gm

=

(mS)

19.2 IABC(mA)

RO 7.5/IABC(M) (mA)

IOUT IABCMax(mA)

(mA)

FIGURE 1. EQUIVALENT DIAGRAM OF THE OTA

Y

W X

Z

4

6

OUTPUT

3NON-INVERTINGINPUT

2INVERTING

INPUTQ1 Q2

5

AMPLIFIERBIAS

CURRENTIABC

7

V-

V+

FIGURE 2. CURRENT MIRRORS W, X, Y AND Z USED IN

THE OTA

DIFFERENTIAL AMPLIFIERTRANSFER CHARACTERISTIC

Vbe (mV)

NORMALIZEDOUTPUTCURRENTAND

DEVIATIONFROMSTRAIGHTLINE

1.0

0.8

0.6

0.4

0.2

0

-0.2

-0.4

-0.6

-0.8

-1.0

-150 -100 -50 0 50 100 150

FIGURE 3. THE OUTPUT CURRENT TRANSFER

CHARACTERISTIC OF THE OTA IS THE SAME

AS THAT OF AN IDEALIZED DIFFERENTIAL

AMPLIFIER

Application Note October 2000

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1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Corporation 2000

OBSOLETEPRODUCT

NORECOMMENDEDREPLAC

EMENT

CallCentralApplications1-888

-INTERSIL

oremail: [email protected]

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2

multiplier. This is analogous to using an elephant to carry a

twig. It may be elegant but it takes a lot to keep it going!

When operated in the gain control mode, one input of the

standard transconductance multiplier is offset so that only

one half of the differential input is used; thus, one half of the

multiplier is being thrown away.

The OTA, while providing excellent linear amplifier

characteristics, does provide a simple means of gain control.For this application the OTA may be considered the

realization of the ideal differential amplifier in which the full

differential amplifier gm is converted to a single ended

output. Because the differential amplifier is ideal, its gm is

directly proportional to the operating current of the

differential amplifier; in the OTA the maximum output current

is equal to the amplifier bias current IABC. Thus, by varying

the amplifier bias current, the amplifier gain may be varied:

A = gm RL where RL is the output load resistance. Figure 4

shows the basic configuration of the OTA DC gain control

circuit.

As long as the differential input signal to the OTA remains

under 50mV peak-to-peak, the deviation from a linear

transfer will remain under 5 percent. Of course, the total

harmonic distortion will be considerably less than this value.

Signal excursions beyond this point only result in an

undesired compressed output. The reason for this

compression can be seen in the transfer characteristic of the

differential amplifier in Figure 3. Also shown in Figure 3 is a

curve depicting the departure from a linear line of this

transfer characteristic.

The actual performance of the circuit shown in Figure 4 isplotted in Figure 5. Both signal to noise ratio and total

harmonic distortion are shown as a function of signal input.

Figures 5B and 5C show how the signal handling capability of

the circuit is extended through the connection of diodes on

the input as shown in Figure 6 [2]. Figure 7 shows total

system gain as a function of amplifier bias current for several

values of diode current. Figure 8 shows an oscilloscope

reproduction of the CA3080 transfer characteristic as applied

to the circuit of Figure 4. The oscilloscope reproduction of

Figure 9 was obtained with the circuit shown in Figure 6. Note

the improvement in linearity of the transfer characteristic.

Reduced input impedance does result from this shunt

connection. Similar techniques could be used on the OTA

output, but then the output signal would be reduced and the

correction circuitry further removed from the source of non

linearity. It must be emphasized that the input circuitry is

differential.

3

54

7

OTA

CA3080A

251

51 +

-

6

SIGNALINPUT

VX

VM

GAINCONTROL

IABC

RM

-6V

+6V

10K

IO

IO = gm VXAMPLITUDEMODULATED

OUTPUT

FIGURE 4. BASIC CONFIGURATION OF THE OTA DC GAIN

CONTROL CIRCUIT

FIGURE 5A.

FIGURE 5B.

FIGURE 5C.

FIGURE 5. PERFORMANCE CURVES FOR THE CIRCUIT OF

FIGURES 4 AND 6

7

6

5

4

3

2

1

0

0.1 1.0 10 100 1.0V

INPUT VOLTAGE (mV)

THD(PERCENT) 100

80

60

40

20

0

S/NRATIO(dB)

THD

DIODE CURRENT = 0mA

500A

10A

IABCCA3080A

S/N RATIO

7

6

5

4

3

2

1

01 10 100 1V 10V

INPUT VOLTAGE (mV)

THD(PERCENT) 100

80

60

40

20

0

S/NRATIO(dB)

THD

DIODE CURRENT = 0.5mA

IABCCA3080A

S/N RATIO500A

10A

7

6

5

4

3

2

1

0

1 10 100 1V 10V

INPUT VOLTAGE (mV)

THD(PERCENT)

100

80

60

40

20

0

S/NR

ATIO(dB)

THD

DIODE CURRENT = 1mA

IABCCA3080A

S/N

500A

10A

RATIO

DISTORTION IS PRIMARILYA FUNCTION OF SIGNAL INPUT

60

Application Note 6077

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Simplified Differential Input to Single Ended

Output Conversion

One of the more exacting configurations for operational

amplifiers is the differential to single-ended conversion

circuit. Figure 10 shows some of the basic circuits that areusually employed. The ratios of the resistors must be

precisely matched to assure the desired common mode

rejection. Figure 11 shows another system using the

CA3080 to obtain this conversion without the use of

precision resistors. Differential input signals must be kept

under 126mV for better than 5 percent nonlinearity. The

common mode range is that of the CA3080 differential

amplifier. In addition, the gain characteristic follows the

standard differential amplifier gm temperature coefficient of-0.3%/oC. Although the system of Figure 11 does not

provide the precise gain control obtained with the standard

operational amplifier approach, it does provide a good

simple compromise suitable for many differential transducer

amplifier applications.

Transistors from CA3046 array.AGC System with extended input range.

FIGURE 6. A CIRCUIT SHOWING HOW THE SIGNAL

HANDLING CAPABILITY OF THE CIRCUIT OF

FIGURE 4 CAN BE EXTENDED THROUGH THE

CONNECTION OF DIODES ON THE INPUT

FIGURE 7. TOTAL SYSTEM GAIN vs AMPLIFIER BIAS

CURRENT FOR SEVERAL VALUES OF DIODE

CURRENT

FIGURE 8. CA3080 TRANSFER CHARACTERISTIC FOR THE

CIRCUIT OF FIGURE 4

FIGURE 9. CA3080 TRANSFER CHARACTERISTIC FOR THE

CIRCUIT OF FIGURE 6

3

CA3080A

251

2K

6

DIODECURRENT

10K

EO

EIN

2K +

V-

12

3

45

6

7

8

9

11

12 13

14

0mA

10A

100A

1mA

0.5mA

IABC (A)0 100 200 300 400 500

100

10

1.0

0.1

0.01

GAIN

DIODECURRENT

Horizontal: 25mV/Div. Vertical: 50A/Div., IABC = 100A Horizontal: 0.5V/Div. Vertical: 50A/Div., IABC = 100A,Diode Current = 1mA


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The CA3094

The CA3094 offers a unique combination of characteristics

that suit it ideally for use as a programmable gain block for

audio power amplifiers. It is a transconductance amplifier in

which gain and open-loop bandwidth can be controlled

between wide limits. The device has a large reserve of

output-current capability, and breakdown and power

dissipation ratings sufficiently high to allow it to drive acomplementary pair of transistors. For example, a 12W

power amplifier stage (8 load) can be driven with peakcurrents of 35mA (assuming a minimum output transistor

beta of 50) and supply voltages of 18V. In this application,the CA3094A is operated substantially below its supply

voltage rating of 44V max. and its dissipation rating of 1.6W

max. Also in this application, a high value of open-loop gain

suggests the possibility of precise adjustment of frequency

response characteristics by adjustment of impedances in the

feedback networks.

Implicit Tone Controls

In addition to low distortion, the large amount of loop gainand flexibility of feedback arrangements available when

using the CA3094 make it possible to incorporate the tone

controls into the feedback network that surrounds the entire

amplifier system. Consider the gain requirements of a

phonograph playback system that uses a typical high quality

magnetic cartridge[3]. A desirable system gain would result

in from 2W to 5W of output at a recorded velocity of 1cm/s.

Magnetic pickups have outputs typically ranging from 4mV to

10mV at 5cm/s. To get the desired output, the total system

needs about 72dB of voltage gain at the reference

frequency.

Figure 12 is a block diagram of a system that uses a passive orlosser type of tone control circuit that is inserted ahead of the

gain control. Figure 13 shows a system in which the tone

controls are implicit in the feedback circuits of the power

amplifier. Both systems assume the same noise input voltage

at the equalizer and main-amplifier inputs. The feedback

system shows a small improvement (3.8dB) in signal-to-noise

ratio at maximum gain but a dramatic improvement (20dB) at

the zero gain position. For purposes of comparison, the

assumption is made that the tone controls are set flat in both

cases.

Cost Advantages

In addition to the savings resulting from reduced parts countand circuit size, the use of the CA3094 leads to further

savings in the power supply system. Typical values of power

supply rejection and common-mode rejection are 90dB and

100dB, respectively. An amplifier with 40dB of gain and 90dB

of power supply rejection would require 316mV of power

supply ripple to produce 1mV of hum at the output. Thus, no

further filtering is required other than that given by the energy

storage capacitor at the output of the rectifier system.

R2

+-

R1

R3R4

OUTPUTDIFFERENTIAL

INPUT

R1R2

=R3R4

R5

+-

R4

R7

OUTPUTDIFFERENTIAL

INPUT

R4

R5=

R6

R7

R1

+-

R2

R3

+-

R6

R1 = R3

OUTPUT

DIFFERENTIALINPUT

R1

+-

R2

+-

R3

R4

R1

R2=

R4

R3

FIGURE 10. SOME TYPICAL DIFFERENTIAL TO SINGLE

ENDED CONVERSION CIRCUITS

24

CA3080

3

7

5

-

+

V-

2K

62K

10K

RL

OUTPUT

+VIABC

500A

A = gm RL at 500A, IABC:gm 10mS. A = 10mS x 10K = 100.

FIGURE 11. A DIFFERENTIAL TO SINGLE ENDED

CONVERSION CIRCUIT WITHOUT PRECISION

RESISTORS

DIFFERENTIALINPUT


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Power Amplifier Using the CA3094A complete power amplifier using the CA3094 and threeadditional transistors is shown schematically in Figure 14.The amplifier is shown in a single-channel configuration, butpower supply values are designed to support a minimum oftwo channels. The output section comprises Q1 and Q2,complementary epitaxial units connected in the familiarbootstrap arrangement. Capacitor C3 provides added basedrive for Q1 during positive excursions of the output. The

circuit can be operated from a single power supply as well asfrom a split supply as shown in Figure 15. The changesrequired for 14.4V operation with a 3.2 speaker are alsoindicated in the diagram.

The amplifier may also be modified to accept input fromceramic phonograph cartridges. For standard inputs(equalizer preamplifiers, tuners, etc.) C1 is 0.047F, R1 is250k, and R2 and C2 are omitted. For ceramic-cartridgeinputs, C1 is 0.0047F, R1 is 2.5M, and the jumper acrossC2 is removed.

Output Biasing

Instead of the usual two-diode arrangement for establishingidling currents in Q1 and Q2, a Vbe Multiplier, transistor Q3, is

used. This method of biasing establishes the voltage betweenthe base of Q1 and the base of Q2 at a constant multiple of thebase to emitter voltage of a single transistor while maintaining alow variational impedance between its collector and emitter(see Appendix A). If transistor Q3 is mounted in intimatethermal contact with the output units, the operating temperatureof the heat sink forces the Vbe of Q3 up and down inverselywith heat-sink temperature. The voltage bias between thebases of Q1 and Q2 varies inversely with heat sink temperatureand tends to keep the idling current in Q1 and Q2 constant.

A bias arrangement that can be accomplished at lower costthan those already described replaces the Vbe multiplier with a1N5391 diode in series with an 8.2 resistor. This arrangementdoes not provide the degree of bias stability of the Vbe

multiplier, but is adequate for many applications.

Tone-Controls

The tone controls, the essential elements of the feedbacksystem, are located in two sets of parallel paths. The bassnetwork includes R3, R4, R5, C4, and C5. C6 blocks the DC

from the feedback network so that the DC gain from input to thefeedback takeoff point is unity. The residual DC output voltageat the speaker terminals is then

where R1 is the source resistance. The input bias current is

then

The treble network consists of R7, R8, R9, R10, C7, C8, C9,and C10. Resistors R7 and R9 limit the maximum available cutand boost, respectively. The boost limit is useful in curtailing

heating due to finite turn-off time in the output units. The limit isalso desirable when there are tape recorders nearby. The cutlimit aids the stability of the amplifier by cutting the loop gain athigher frequencies where phase shifts become significant.

In cases in which absolute stability under all load conditions isrequired, it may be necessary to insert a small inductor in theoutput lead to isolate the circuit from capacitive loads. A 3Hinductor (1A) in parallel with a 22 resistor is adequate. Thederivation of circuit constants is shown in Appendix B. Curvesof control action versus electrical rotation are also given.

FIGURE 12. BLOCK DIAGRAM OF A SYSTEM USING A LOSSER TYPE TONE CONTROL CIRCUIT

FIGURE 13. A SYSTEM IN WHICH TONE CONTROLS ARE IMPLICIT IN THE FEEDBACK CIRCUIT OF THE POWER AMPLIFIER

EQUALIZER

AREF = 32dB

EN AT INPUT = 1 x 10-6

TONE

CONTROLS

-20dB

VOLUME

CONTROLS

POWER

AMP

PICKUPESIG =1mV

BUFFER STAGE


EN = 6.23 x 10-3EN = 4 x 10

-6EN = 4 x 10-6EN = 40 x 10

-6

ESIG = 40mV ESIG = 4mV ESIG = 4mV

ATOTAL = 60dB

8SPEAKER

TOTAL GAIN = 72dB

EO

EN

4

6.23 x 10-3= = 640 AT MAX VOL

EN = 4mV AT MIN VOL

EO = 4V

EQUALIZER

AREF = 32dB


VOLUME

CONTROLS

AMPLIFIER WITH FEED-

EN = 5 x 10-6

EN = 4.03 x 10-3

ESIG = 40mV ESIG = 40mV

ATOTAL = 60dB

EO

EN

4

4.03 x 10-3= = 990 AT MAX VOL

EN = 0.5mV AT MIN VOL

EO = 4V

EN = 40 x 10-6 EN = 40 x 10

-6

PICKUPESIG =1mV

BACK TONE CONTROLS

R1

R11 R12+

R12---------------------------

IAB C

IABC2

-------------VCC VBE( )

2R6----------------------------------=


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FIGURE 14. A COMPLETE POWER AMPLIFIER USING THE CA3094 AND THREE ADDITIONAL TRANSISTORS

FIGURE 15. A POWER AMPLIFIER OPERATED FROM A SINGLE SUPPLY

4

5

6

8

1

7

2

CA3094

+

-

ES

R1

3

68

1800

0.001F

0.12F

0.001F

15K

C1

CUT(CCW)

BOOST(CW)

5F

820

5600

3pF

2201W

220

1W

30

27

C3

+

Q3

Q2

Q1

1

0.47F

R6

0.2F

R4

1K25F 0.02F

R3

+

C20.47F

JUMPER

0.47

0.47

R11

R12

4700F

+4700F D1

D2

D3

D4

120V AC TO26.8VCT AT 1A

120V AC

R21.8M

0.01F

+ BOOST(CW)

CUT(CCW)

BASS

+

TREBLE

C9

R8

C8

R7C7R20

R9

-

C5

C4R5C6

-

100K 10K

680K

15F

330

47

T1

-

-

2N6292

2N6292

2N6107

C10

330

47

0.47F

1500F

+

-

Q2

Q1

15F

0.47

0.47

+

-

R12201W

R22201W

R330

R427

1

3pF

8

6

1

4

5

2

3

7

47K10K100KBASS

1K

1/2VCC

25F

+-

220K

0.02F

+

-25F1/2VCC

0.2F

0.22F

100K

220K

0.12F

1.8K

68

0.001F

0.01F

R51.2M

820

+36V

+

-5F

VCC

5.6K

220K5%

5%TREBLE

15K

CA3094

2N6292

2N6107

Q32N6292


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Performance

Figure 16 is a plot of the measured response of the complete

amplifier at the extremes of tone control rotation. A

comparison of Figure 16 with the computed curves of Figure

B4 (Appendix B) shows good agreement. The total harmonic

distortion of the amplifier with an unregulated power supply is

shown in Figure 17; IM distortion is plotted in Figure 18. Hum

and noise are typically 700V at the output, or 83dB down.

Companion RlAA Preamplifier

Many available preamplifiers are capable of providing the

drive for the power amplifier of Figure 14. Yet the uniquecharacteristics of the amplifier, its power supply, input

impedance, and gain make possible the design of an RIAA

preamplifier that can exploit these qualities. Since the input

impedance of the amplifier is essentially equal to the value of

the volume control resistance (250k), the preamplifier neednot have high output current capability. Because the gain of

the power amplifier is high (40dB) the preamplifier gain only

has to be approximately 30dB at the reference frequency

(1kHz) to provide optimum system gain.

Figure 19 shows the schematic diagram of a CA3080

preamplifier. The CA3080, a low cost OTA, provides

sufficient open-loop gain for all the bass boost necessary inRIAA compensation. For example, a gm of 10mS with a load

resistance of 250k provides an open-loop gain of 68dB,thus allowing at least 18dB of loop gain at the lowest

frequency. The CA3080 can be operated from the same

power supply as the main amplifier with only minimal

decoupling because of the high power supply rejection

inherent in the device circuitry. In addition, the high voltage

swing capability at the output enables the CA3080

preamplifier to handle badly over modulated (over-cut)

recordings without overloading. The accuracy of equalization

is within 1dB of the RIAA curve, and distortion is virtuallyimmeasurable by classical methods. Overload occurs at an

output of 7.5V, which allows for undistorted inputs of up to186mV (260mV peak).

60

55

50

45

40

35

30

25

20

15

10

102 3 6 8

1002 3 68

10002 3 6 8

10K 100K2 3 6 8

BASS CUT

TREBLE CUT

FLAT

TREBLE BOOSTBASS BOOST

FREQUENCY (Hz)

EO/ES(dB)

FIGURE 16. THE MEASURED RESPONSE OF THE AMPLIFIER

AT EXTREMES OF TONE CONTROL ROTATION

1.0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

2 4 6 8 10 12

OUTPUT POWER (W)

TOTALHARMON

ICDISTORTION(%)

2kHz

4kHz

16kHz

26kHz

1kHz

1kHz

26kHz

16kHz

4kHz

0

FIGURE 17. TOTAL HARMONIC DISTORTION OF THE

AMPLIFIER WITH AN UNREGULATED POWER

SUPPLY

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0 2 4 6 8 10 12 14

OUTPUT POWER (W)

INTERMODULA

TIONDISTORTION(%)

FIGURE 18. IM DISTORTION OF THE AMPLIFIER WITH AN

UNREGULATED SUPPLY

60Hz 12kHz60Hz 7kHz

60Hz 2kHz


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Appendix A - VbeMultiplier

The equivalent circuit for the Vbe multiplier is shown in

Figure A1. The voltage E1 is given by:

The value of Vbe is itself dependent on the emitter current of

the transistor, which is, in turn, dependent on the input

current I since:

The derivative of Equation A1 with respect to I yields the

incremental impedance of the Vbe multiplier:

where K3 is a constant of the transistor Q1 and can be foundfrom:

Equation A4 is but another form of the diode equation:

Using the values shown in Figure 14, plus data on the

2N6292 (a typical transistor that could be used in the circuit),

the dynamic impedance of the circuit at a total current of

40mA is found to be 4.6. In the actual design of the Vbemultiplier, the value of IR2 must be greater than Vbe or the

transistor will never become forward biased.

Appendix B - Tone Controls

Figure B1 shows four operational amplifier circuit

configurations and the gain expressions for each. The

asymptotic low frequency gain is obtained by letting S

approach zero in each case:

The asymptotic high-frequency gain is obtained by letting S

increase without limit in each expression:

3

CA3080

2

4

7

+

-3.9K

5.6K

V+

EOUT6

56K

5

0.002F 680pF

1.5M

120K

50F

5F

1F

ES

56K 5.6K

3

CA3080

2

4

7

+

-3.9K

EOUT

6

56K

5

0.002F 680pF

1.5M

120K

5F

1F

ES

56K

-

+50F

-

+

V-

FIGURE 19. A CA3080 PREAMPLIFIER

+

E1

R1 I

1+-------------= Vbe 1

R1R2 1+( )-------------------------++

(EQ. A1)

E1

IeR2

R1

VbeI

FIGURE A1. EQUIVALENT CIRCUIT FOR THE Vbe MULTIPLIER

Ie IVbeR2

----------= (EQ. A2)

dE1dI

---------- ZR1

1+------------- 1

BR1 1+( )R2

-------------------------+K3R2

R2Ie K3+--------------------------+= = (EQ. A3)

Vbe K3 ln Ie K2= (EQ. A4)

Ie ISe

qVbeKT

--------------- 1

=

(EQ. A5)

Bass Boost: ALOW

R1 R2 R3+ +

R2-----------------------------------=

Bass Cut: ALO WR1 R2 R3+ +

R2 R3+-----------------------------------=

Treble Boost: ALO W

C1 C4+

C4---------------------=

Treble Cut: ALO W

C1 C4+

C4---------------------=

Bass Boost: AHIGH

R1 R2+

R2---------------------=

Bass Cut: AHIGH

R1 R2+

R2---------------------=

Treble Boost: AHIGH 1 C1

C3 C4+

C3C4---------------------

+=

Treble Cut: AHIGH

C2

C1C4C1 C4+---------------------+

C1 C2+-----------------------------------=


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9

Note that the expressions for high frequency gain are

identical for both bass circuits, while the expressions for low

frequency gain are identical for the treble circuits.

Figure B2 shows cut and boost bass and treble controls that

have the characteristics of the circuits of Figure B1. The

value REFF in the treble controls of Figure B1 is derived from

the parallel combination of R1 and R2 of Figure B2 when the

control is rotated to its maximum counterclockwise position.When the control is rotated to its maximum clockwise

position, the value is equal to R1.

To compute the circuit constants, it is necessary to decide in

advance the amounts of boost and cut desired. The gain

expressions of Figure B1 indicate that the slope of the

amplitude versus frequency curve in each case will be 6dB

per octave (20dB per decade). If the ratios of boosted and

cut gain are set at 10, i.e.

:

then the following relationships result:

Bass Circuit: ALOW B oost( ) 10AMI D=

AL OW C ut( )

AMI D10

--------------=

Treble Circuit: AHIGH Boost( ) 10AMI D=

AH IGH C ut( )

AMI D10

--------------=

Bass Circuit: R1 10R2=

R3 99R2=

Treble Circuit: C1 10C4=

C2

10C499

--------------=

FIGURE B1 (A). BASS BOOST FIGURE B1 (B). BASS CUT

FIGURE B1 (C). TREBLE BOOST FIGURE B1 (D). TREBLE CUT

FIGURE B1. FOUR OPERATIONAL AMPLIFIER CIRCUIT CONFIGURATIONS AND THE GAIN EXPRESSIONS FOR EACH

+-

R2

R1R3

C1

ES

EO

AR1 R2 R3+ +

R2-----------------------------------

1 SC1

R2R3 R1R3+( )

R1 R2 R3+ +( )------------------------------------------+

1 SR3C1+----------------------------------------------------------------

=

ALOW FREQUENCY

R1 R2 R3+ +

R2-----------------------------------=

+-R2

R1

C2

ES

EO

R3

AR1 R2 R3+ +

R2 R3+-----------------------------------

1 SC2

R2R3 R1R3+( )

R1 R2 R3+ +( )------------------------------------------+

1 SC2R2R3

R2 R3+---------------------

+

----------------------------------------------------------------=

ALOW FREQUENCYR1 R2 R3+ +

R2 R3+-----------------------------------=

+-

REFF

C1

ES

EO

C4

C3

AC1 C4+

C4---------------------

1 SREFF

C1C4 C3C4 C1C3+ +( )C1 C4+( )

---------------------------------------------------------------+

1 SREFFC3+----------------------------------------------------------------------------------------------

=

ALOW FREQUENCY

C1 C4+

C4---------------------=

+-

REFF

ES

EO

C4

C2

C1

AC1 C4+

C4---------------------

1 SREF F

C1C4 C2C4 C1C2+ +( )C1 C4+( )

---------------------------------------------------------------+

1 SREF F C1 C2+( )+----------------------------------------------------------------------------------------------

=

ALOW FREQUENCY

C1 C4+

C4---------------------=


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The unaffected portion of the gain (AHIGH for the bass

control and ALOW for the treble control) is 11 in each case.

To make the controls work symmetrically, the low and high

frequency break points must be equal for both boost and cut.

Thus:

since R3 R2 + R3, C2 = 10C1

and

since

To make the controls work in the circuit of Figure 14, breaks

were set at 1000Hz:

for the base control

and for the treble control

Response and Control Rotation

In a practical design, it is desirable to make flat response

correspond to the 50% rotation position of the control, and to

have an aural sensation of smooth variation of response on

either side of the mechanical center. It is easy to show thatthe flat position of the bass control occurs when the wiper

arm is advanced to 91% of its total resistance. The

amplitude response of the treble control is, however, never

completely flat; a computer was used to generate response

curves as controls were varied.

Figure B3 is a plot of the response with bass and treble tone

controls combined at various settings of both controls. The

values shown are the practical ones used in the actual

design. Figure B4 shows the information of Figure B3

replotted as a function of electrical rotation. The ideal taper

for each control would be the complement of the 100Hz plot

for the bass control and the 10kHz response for the treble

control. The mechanical center should occur at the

crossover point in each case.

Bass Control:C1R3 R1 R2+( )

R1

R2

R3

+ +------------------------------------------

C2R2R3R

2

R3

+-----------------------=

and C1R3

C2R3 R1 R2+( )

R1 R2 R3+ +------------------------------------------=

Treble Control: R1

C1C4 C3C4 C1C3+ +( )

C1 C4+---------------------------------------------------------------

R1R2R1 R2+--------------------- C1 C2+( )=

R2C3

R1R2R1 R2+---------------------

C1C4 C2C4 C1C2+ +( )

C1 C4+( )---------------------------------------------------------------=

C1 100C2 C2, C3 and C1 10C4 R1, 9R2= = =

0.1C1R31

2 1000--------------------------=

R1C31

2 1000--------------------------=

R2 CW R3 CCW R1

C2C1

FIGURE B2 (A). BASS CONTROL

-+

C1 CW R1 CCW C4

C2

C3

R2

FIGURE B2 (B). TREBLE CONTROL

FIGURE B2. CUT AND BOOST BASS AND TREBLE CONTROLS

THAT HAVE THE CHARACTERISTICS OF THE

CIRCUITS IN FIGURE B1

-+

45

40

0

5

10

15

20

25

30

35

10 100 1000 10K 100K

FREQUENCY (Hz)

EOES

dB

68 15K 0.01F

0.0011.8K

0.12F

0.001F

820

EOES

0.020.2

1K

100K 10K

FF

Q = 1.0

Q = 0.99

Q = 0

Q = 0.98

Q = 0.96

Q =0.914

Q = 0.85

P = 1.0

P = 0.99

P = 0

P = 0.98

P = 0.96

P = 0.92

P = 0.8P = 0.5P = 0.3P = 0.1

P = 0.05

P

Q

F

FIGURE B3. A PLOT OF THE RESPONSE OF THE CIRCUIT OF

FIGURE 14 WITH BASS AND TREBLE TONE

CONTROLS COMBINED AT VARIOUS SETTINGS

OF BOTH CONTROLS

-+

Q = 0.75

Q = 0.5

Q = 0.2


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All Intersil semiconductor products are manufactured, assembled and tested under ISO9000 quality systems certification.

Intersil semiconductor products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time with-

out notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and

reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result

from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.

For information regarding Intersil Corporation and its products, see web site http://www.intersil.com

References

For Intersil documents available on the internet, see web site

http://www.intersil.com/

[1] AN6668 Application Note, Applications of the CA3080and CA3080A High Performance OperationalTransconductance Amplifiers, H. A. Wittlinger, IntersilCorporation.

[2] A New Wide-Band Amplifier Technique,B. Gilbert,IEEE Journal of Solid State Circuits, Vol. SC-3, No. 4,December, 1968.

[3] Trackability,James A. Kogar, Audio, December, 1966.

45

0

5

10

15

20

25

30

35

40

0.2 0.4 0.6 0.8 1.0

ELECTRICAL ROTATION OF BASS CONTROL

GA

IN(dB)

1000Hz

100Hz

31.6Hz

FIGURE B4 (A).

1kHz

0

5

10

15

20

25

30

35

40

0.2 0.4 0.6 0.8 1.0

ELECTRICAL ROTATION OF TREBLE CONTROL

GA

IN(dB)

31.6kHz10kHz

FIGURE B4 (B).

FIGURE B4. THE INFORMATION OF FIGURE B3 PLOTTED AS

A FUNCTION OF ELECTRICAL ROTATION


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