A Dual-Polarized Planar-Array Antenna forS-Band and X-Band Airborne Applications

S h i h - H s u n Hsu', Y u - J i u n Ren 2, a n d K a i C h a n g 1

"Department of Electrical and Computer Engineering, Texas A&M UniversityCollege Station, TX 77843-3128, USA

Tel: +1 (979) 845-5285; E-mail: shihhsun@ece.tamu.edu

21ntelligent Automation, Inc.Rockville, MD 20855, USA

Abstract

A new dual-frequency dual-polarized array antenna for airborne applications is presented in this paper. Two planar arrays

with thin substrates (RIT Ouroid 5880 substrate, with e; = 2.2 and a thickness of 0.13 mm) are integrated to provide

simultaneous operation at S band (3 GHz) and X band (10 GHz). Each 3 GHz antenna element is a large rectangular ring­resonator antenna, and has a 9.5 dBi gain that is about 3 dB higher than the gain of an ordinary ring antenna. The 10 GHzantenna elements are circular patches. They are combined to form the array with a gain of 18.3 dBi, using a series-fedstructure to save the space of the feeding line network. The ultra-thin array can be easily placed on an aircraft's fuselage, dueto its lightweight and conformal structure. It will be useful for wireless communication, radar, remote sensing, and surveillanceapplications.

Keywords: Antenna arrays; microstrip arrays; microwave antenna arrays; aircraft antennas; planar arrays; polarization

1. Introduction

M odern wireless communication systems demand low-profile,

lightweight, and relatively inexpensive antennas [1]. There

has been increasing interest in the development of dual-polarized

antennas/arrays, due to many advantages in performance improve­

ment for wireless communication and radar systems. Dual

polarization avoids the requirement of precise alignment needed in

single-polarized systems. Dual-polarized operation can also pro­

vide more information for radar systems, can increase the isolation

between the transmitting and receiving signals of transceivers and

transponders, and can double the capacity of communication sys­

tems by means of frequency reuse [2, 3]. In simultaneous transmit­

ting-receiving applications, dual-polarized antennas with two-port

connections offer an alternative to the commonly used bulky

circulator, or separate transmitting and receiving antennas. Further­

more, dual-polarized antennas can provide polarization diversity,

which prevents the system's performance degradation due to multi­

path fading in complex propagation environments [4]. Compared

with space diversity, the polarization-diversity technique has the

advantage of reducing the number and size of the antenna elements

in the system.

To create dual polarization, the antenna element has to be fed

at two orthogonal points or edges, such that two degenerate reso­

nant modes can be excited for the orthogonal polarizations, i.e.,

vertical and horizontal polarizations. Design techniques of dual­

polarized antennas can be classified into three main categories. The

first is to make an orthogonal arrangement of the radiation ele-

ments for two polarizations [5, 6]. The second is to properly

choose stacked structures of the radiating elements [7, 8]. The third

is to use special feeding techniques [9, 10]. In general, symmetric

structures tend to give better dual-polarization performance. At

higher frequencies, a dielectric-resonator antenna can be used to

reduce the metallic loss of the patch [11].

The dual-polarized antenna elements can be assembled to

construct a high-gain array. Although many dual-polarized anten­

nas have been proposed, not all of them are good candidates for

array design, due to their complex structures and feeding-line net­

works. Many of them are bulky and heavy, and not suitable for air­

borne applications. On the other hand, isolation is one of the

important parameters to be considered in dual-polarized array

design. Most reported dual-polarized arrays achieve at least 20 dB

isolation [12-15]. To minimize the coupling between the feed-line

networks, a proximity/aperture-coupling structure can be applied to

prevent radiation due to the microstrip line and radiator from

degrading the polarization performance. The multilayer structure

can also be used to enhance the isolation, in which isolations of

better than 20-30 dB can be obtained with more-complicated struc­

tures. However, most dual-polarized antenna designs will result in

a bulky array that is not suitable for use in aircraft, airships, or

unmanned aerial vehicles (DAVs).

In many airborne applications, an array antenna should have

good isolation, high efficiency, and ease of integration with the

aerial vehicle. A simple feeding-line network with lower loss and

high isolation is generally desired. Microstrip series-fed arrays

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have been shown to have a structure that enhances the antenna's

efficiency [1]. This is because the array feeding-line length is

significantly reduced, compared to the conventional corporate feed­

ing-line network. Such arrays can be either traveling-wave or reso­

nant arrays . A planar structure with a thin and flexible substrate is

a good choice , because it will not disturb the appearance of the air­

craft, and can be easily integrated with electronic devices for signal

processing.

In this paper, a dual-frequency dual-polarized array antenna is

presented for airborne antenna applications. A multilayer structure

is adopted for dual-band operation. The antenna arrays for the two

frequencies are separated on different layers. To reduce the array's

volume and weight, a series-fed network is used . An ultra-thin sub­

strate is chosen in order to make the array conformal, and the array

can be easily placed on an aircraft's fuselage, or inside the aircraft .

The parameters affecting the array 's characteristics are discussed,

and the measured return losses, radiation gains, and array patterns

are presented.

Two RTlDuroid 5880 substrates (61 =63 =2.2) and a foam

layer (62 = 1.06) form the multilayer structure. The thicknesses of

the substrates (hI and h 2 ) are both only 0.13 rom (5 mil) . These

ultra-thin and flexible substrates make it possible for the array to

be easily attached onto the aircraft's fuselage, or installed inside

the aircraft . The foam layer has a thickness of h 2 =1.6 mm . The

dimensions of the array were optimized by using the full-wave

electromagnetic simulator, IE3D [16] .

2.1 X-Band Antenna and Subarray

The X-band array uses the circular patch as its unit antenna

element. The patch radius, R, for the dominant TM)) mode at the

resonant frequency ( Ir' in GHz) can be calculated from [17]

2. Array Design

(1)

Unlike most other elements, the electrical parameters of the

substrate will be affected by the temperature and moisture varia­

tions occurring in airborne applications, and these affect the

performance of the antennas. The magnitude of the transmitting

power may also generate a large amount of heat, which results in a

significantly increased temperature. The choice of the substrate is

therefore an important factor in airborne-antenna design. The

configuration of the antenna element determines the complexity of

the array feeding-line network, which controls the size and mass of

the array. The configuration of the feeding-line networks may also

incur different levels of port isolation and pattern polarization.

Important design considerations of a dual-polarized airborne array

antenna are summarized in Table I. Our goal is to design a low­

mass conformal array antenna for airborne applications. An ultra­

thin substrate is used to achieve the conformal antenna.

DThe multilayer array structure for dual-band (S band and X

band) operation is shown in Figure I. The S-band antenna elements

sit on the top layer, and the X-band antennas are on the bottom

layer. A foam layer (h 2 ) serves as the spacer, and is sandwiched

between the two substrate layers. One of the important design

considerations for this multilayer dual-band array is that the S-band

antenna element should be nearly transparent to the X-band

antenna elements. Otherwise, the S-band element may degrade the

performance of the X-band antenna.

y.

x

Figure 1. The multilayer structure of the dual-band dual­

polarized array antenna.

Table 1. Design considerations for the airborne array antenna.

Factor Impact

TemperatureThis is determined by the aerial conditions and delivered power, which will affect the electrical

parameters ofthe substrate.

Moisture The effects due to moisture are similar to the temperature effects.

Electrical parameters This controls the antenna's nerformance and is mainly changed by large temperature variations.

Feeding-line networkThis determines the complexity of the array. Unsuitable antenna elements could result in a bulky

and heavy array.

Isolat ionPort-to-port isolation and cross-polarization level can be enhanced by using well-designed

feeing-line networks, such as proximity/aperture coupling and multilayer configurations.

CouplingFor dual-frequency operation, the interference between antennas operating at different

frequencies may affect radiation patterns and gain.

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where Or and h (in em) are the relative dielectric constant and

thickness of the substrate, respectively. At the operating frequency,

fr =10 GHz, an initial value of R of 5.82 mm, calculated from

Equation (I), was used. The optimized value of 5.95 mm was

obtained with the aid of IE3D.

The circular patches are fed with microstrip lines at the

circumferential edge, as shown in Figure 2a. For a single circular

patch, two microstrip feeding lines are used to feed the circular

patch to generate two orthogonally radiating TMt 1 modes for dual­

polarized operation. Two feed points are located at the edge of the

patch, 900 away from each other, so that the coupling between

these two ports can be minimized. The port isolation also depends

on the quality factor of the patch. Increasing the substrate's thick­

ness decreases the isolation [18]. Therefore, using thin substrates

could improve the quality of isolation.

with

F =8.79 I/(Jr F:) , (2)

, 2R , L" H-port <,'lL--- -- -- -;---f:O::::_'!:t------------------- -------------------

Figure 2a. The layout of the X-band antenna, where L = 106,

W =183, R =5.82, LxI =22, and Wxt =0.17 (all dimensions

are in mm),

..-.._..,---_._.._..L,5

------ ----------------------------- ------------------1

,,

.H-port :

-,

Figure 2b. The layout of the S-band antenna, where L = 106 ,

W=183 , Lst = 53.89 , Ls2 = 44.6 , Ls3 = 0.94 , Ls4 =19.31 ,

Lss =34.17, Wst =4.9, and Ws2 =7.8 (all dimensions are in

mm).

Figure 2a shows a 4 x 8 dual-polarized X-band array. The V

port and the H port are the input ports for the two orthogonal

polarizations (vertical and horizontal). The array is composed of

two 4 x 4 subarrays. The corporate-fed power-divider lines split the

input power at each port to the subarrays. Within each subarray, the

circular patches are configured into four 4 x 1 series-fed resonant

type arrays, which make the total array compact and have less

microstrip line losses than would a purely corporate-fed type of

array [19]. An open circuit is placed after the last patch of each

4 x 1 array. The spacing between adjacent circular-patch centers is

about one guided wavelength ( A g =21.5mm at 10 GHz). This is

equivalent to a 3600 phase shift between patches, such that the

main beam points to the broadside. The power coupled to each

patch can also be controlled by adjusting the size of the individual

patch to achieve a tapered amplitude distribution for a lower­

sidelobe design. Another advantage of using the series-fed array

configuration is that the array can be easily converted into a travel­

ing-wave array, with a matched termination at the end of the last

elements, if a steered-beam array is needed.

2.2 S-Band Antenna and Subarray

Figure 3. The geometry of the dual-band dual-polarized array

antenna.

As shown in Figure 2b, the S-band antenna elements are

printed on the top substrate, and are separated from the X-band ele­

ments by the foam layer. To reduce the blocking of the radiation

from the X-band elements at the bottom layer, the shape of the S­

band elements has to be carefully selected. A ring configuration

was a good candidate, since it uses less metallization than an

equivalent patch element. Here, a square-ring microstrip antenna is

used as the unit element of the S-band array. Because antenna ele­

ments at both frequency bands share the same aperture, it is also

preferred that the number of elements on the top layer be as small

as possible, to minimize the blocking effects.

The stacked X-band and S-band array antennas are shown in

Figure 3. As can be seen in the figure, the four sides of the square­

ring element are laid out in such a way that they only cover part of

the feeding lines on the bottom layer, but none of the radiating ele­

ments. Unlike an ordinary microstrip-ring antenna that has a mean

circumference equal to a guided wavelength, the antenna proposed

here has a mean circumference of about 2A g (A g =82.44mm at

V-port .(S-band) •

H-port "(X-band)

H-port(S-band)

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• Mean radius of the ring - 2 "'93GHz• Select two operating frequencies I------r----II

• Use very thin substrates

10 GHz

• Use Eq. (1) to calculate R

• Select IX1 - 1 "'9' wx1 = 0.17 mm• Based on (ls2 - Is1) < (lx1 - 2R)

and (ls2 + Is1) /2 - 2 "'9 ,select Is1 and Is2

• Select Is3 and Ws3 (50 ohm)

No

• Design the power divider• Interconnect the patch elements

No

Yes Yes

Overlay the 3 GHz elements on the 10 GHz elements with

minimum blockage of the circular patches

Figure 4. The design flowchart of the proposed antenna array.

3 GHz). Although the size of the proposed unit element is larger

than an ordinary ring antenna, its gain is about twice as high,

because of its larger radiation-aperture area. The ring is loaded by

two gaps at two of its parallel sides, and these make it possible to

achieve a 50 n input match at the edge of the third side without

using a small value of Ls2/LsI' as mentioned in [20]. For an edge-

fed microstrip ring, if a second feed line is added to the orthogonal

edge, the coupling between the two feeding ports will be high. The

V-port and H-port feeds are therefore placed at two individual ele­

ments, so that the coupling between the two ports can be signifi­

cantly reduced. Using separate elements seems to increase the

number of antenna elements within a given aperture. However, this

harmful effect could be minimized by reducing the number of ele­

ments with the use of larger-sized microstrip rings. A design flow­

chart of the proposed antenna array is provided in Figure 4.

3. Measurement and Discussion

A dual-band array prototype was fabricated and tested in the

anechoic chamber of the Texas A&M University. The array was

tested for one polarization at one frequency at a time, while the

other three ports were terminated with 50 n loads. Detailed simu­

lated and measured results are given and discussed in the follow­

ing.

3.1 X-band Array

top of it. The center frequency of both polarizations was at around

9.95 GHz, and the return losses (811 and 822 ) were better than

22 dB. 811 was the return loss for the V port, and 822 was for the

H port. The isolation (821 or 812 ) between the V and the H

polarizations was better than 30 dB at the resonant frequency, and

better than 25 dB over a wide frequency band. These results were

considered excellent for a dual-polarized array for which the feed­

ing lines for both polarizations were present at the same layer.

These results were similar to those reported in [4], and the physical

size of the array was reduced, due to the hybrid use of the corpo­

rate-fed and series-fed configurations.

Normalized measured radiation patterns are shown in Fig­

ure 6. Well-defined patterns were observed. Cross-polarization lev­

els in the E plane and H plane were 17 dB below the co-polarized­

beam peaks. It was noted that the dimensions of the ground plane

used for the array were about 18.3 em x 10.6 em, which were close

to those of the array's aperture. This could create strong edge

diffraction, and might account for the relatively higher cross­

polarization levels. The peak sidelobe levels (SLL) were -10 to

-13dB, which were normal for the arrays with a uniform ampli­

tude distribution. The asymmetric sidelobes of the H port were

caused by its feeding-line network asymmetry with respect to the

array's center. The maximum measured gain was 18.3 dBi, and the

averaged radiation efficiency was 31%. The half-power beamwidth

(HPBW) of the x-z plane was 9°, and that of the y-z plane was 17°.

This difference was due to the asymmetry of the 4 x 8 array

arrangement.

Figure 5 shows the return loss and the isolation of the X-band

array. The measurements were carried out with the S-band layer on

Theoretically, a microstrip antenna has a very good front-to­

back ratio (FBR), due to its infinite - or relatively large - ground

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plane . This is similar to the case when an airborne antenna is

mounted on an airframe. Here, with a finite ground plane , the

front-to-back radiation ratios of both the E and H planes were bet­

ter than 30 dB. The back radiation was very small , and hence not

shown. A ground plane with a larger dimension can provide a

higher front-to -back radiation ratio , since the coupling and diffrac­

tion of electromagnetic waves are reduced. Detailed specifications

of the X-band array are summarized in Table 2.

3.2 S-Band Array

- - Measured E-plane co-pol

- - - - - Measured E-plane x-pol_ -:iI' _ <-- _ Simulated E-plane co-pol

-- Measured H-plane co-pol

-- --- Measured H-plane x-pol

Simulated H-plane co-pol

Figure 6a. The radiation patterns of the X-band array: V-pert

feed, E plane.The return loss of the S-band array is shown in Figure 7. The

measurements were carried out with the X-band layer under the

S-band layer. About 19 dB return loss was obtained at the resonant

frequency of 2.96 GHz, for both polarizations. Since two separate

elements were used for two polarizations, the results show that the

port isolation was close to 30 dB. Normalized measured radiation

patterns for both polarizations are shown in Figure 8. A typical sin­

gle main beam with a wide 3 dB beam width was observed. The

cross-polarization levels were better than 20 dB. The maximum

gain of the antenna was measured to be 9.5 dBi, and the averaged

radiation efficiency was 81%. The HPBW of the S-band array was

about 56° on each plane. The front-to-back radiation ratios on both

planes were better than 34 dB for the V port and 25 dB for the H

port, which indicated that the dimensions of the ground plane were

acceptable for the S-band antenna.

(dB) -30 -20 -10

-- Measured E-plane co-pol

Measured E-plane x-pol

Simulated E-plane co-pol

Figure 6b. The radiation patterns of the X-band array: V-port

feed, H plane.

-20

- Measured S11

........Simulated S11

-><- Measured S12

-10 ...

o

·30 .

-20 -10

-- Measured H-plane co-pol

0° - - - - - Measured H-plane x-pol

Simulated H-plane co-pol

Figure 6c. The radiation patterns of the X-band array: H-port

feed, E plane.

-30 .

-20 .

-40

9.5 9.6 9.7 9.8 9.9 10 10.1 10.2 10.3 10.4 10.5

Frequency (GHz)

Figure Sa. The S parameters of the X-band array, where port 1

is the V-port feed for vertical polarization.

o

Figure Sb. The S parameters of the X-band array, where port 2

is the H-port feed for horizontal polarization.

Figure 6d. The radiation patterns of the X-band array: H-port

feed, H plane.

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Table 2. A summary of the measured and simulated results for the X-band and S-band array antennas.

X-Band S-Band

Polarization V Port H Port V Port HPort

Frequency (GHz)Measurement 9.8-9.98 9.81-10.0 2.94-2.96 2.94-2.96

Simulation 9.91-9.99 9.9-10.0 2.94-2.96 2.94-2.96

Bandwidth (%)Measurement 1.8 2.3 1.03 1.03

Simulation 1.0 1.0 1.03 1.03

Gain (dBi)Measurement 18.3 17.3 9.5 7.9

Simulation 17.6 17.6 8.72 8.73

Efficiency (%)Measurement 32.4 28.8 96.8 66.8

Simulation 27.5 27.5 80.9 80.9

HPBW (degrees)E plane 17° 8.9° 58° 59°Hplane 9.5° 17.4° 53° 52°

Peak SLL (dB)Measurement -12.3 -10.0 None None

Simulation -13.0 -13.0

FBR (dB) -30 -30 -34.8 -25.5

Isolation (dB)> 3I.l > 25.3 >36.4 >33 .8

X to S band X to S band S to X band S to X band

3.3 Mutual Coupling Effects of Two Layers

Figure 7. The S parameters of the S-band array, where port 1

is the V-port feed for vertical polarization, and port 2 is the H­

port feed for horizontal polarization.

- - Measured [-plane co-pol

-0-- Measured [-plane s-pot

-- Measured If-plane co-pol

-- Measur ed " ..plane x..pol

• Simulated E-plan e co-pol

Simulated H-plane co-pol

- - Measured E-plane co-pol

-0-- Measured [ -plane r-pe l

-- Measu red H-plane co-pol

- Measured H-plane s-pot

• Simulated [ -plane co-pol

Simulated If-plane co-pol

-90

Figure 8a. Radiation patterns ofthe S-band array: V-port feed.

Figure 8b. Radiation patterns of the S-band array: H-port feed.

ments for the S-band array, with or without the presence of the X­

band array, including the S parameters and the radiation patterns.

When the X band was presented, the isolation of S-to-X (from the

ports of the S-band antenna to the ports of the X-band) was

between 33 dB and 42 dB; the isolation of X-to-S was better than

25 dB.

It was observed that increasing the spacing between the S­

band and the X-band antennas did not significantly improve the

isolation. Instead, it raised the center frequency of the S-band

antenna, and vice versa. Consider the case where the foam-layer

thickness (h z) is changed within ±O.5 mm, If h z is increased by3.23.153.1

- - - - Measured 511

--Measured 522

- Measreud 521

-e-o-e-Simulated Sll

- Simulated S22

2.95 3 3.05

Frequency (GHz)

2.92.85

-20 - - - - - - - - - - - - - - - - - - - - - - -- - - - - - - -- -- -- - - - - - - - -

-30

· 10

-40

-50

2.8

CD

"

Simulated results of the return losses and radiation patterns

from IE3D are also shown for comparison. Good agreement was

observed for both frequency bands. Small discrepancies in the

resonant frequencies may be attributed to the accuracy of the

permittivity and the thickness dimension ofthe foam layer given by

the manufacturer. The former value played a more important role.

The resonant frequency shifts from 3 GHz to 2.95 GHz when the

dielectric constant of the foam layer (6Z) changes from 1.06 to

I.l2, a 5.7% change within the inaccuracy of the fabrication proc­

ess. The effects due to the thickness ofthe foam layer are described

in the following section. The specifications of the S-band array are

summarized in Table 2.

The results presented here for both the X and S bands were

measured with the concurrent presence of both antenna layers

(dual-band operation) . However, measurements of the single layer

without the presence of the other layer, i.e., single-band operation,

were also conducted, to investigate the mutual-coupling effects

between the elements of different frequency bands. It was found

that there were no distinct variations between the two measure-

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-- Dual layer

- - - X-band layer only

-90

Figure 9a. Comparisons of the X-band radiation patterns of

the If-port feed for the E plane.

5. Acknowledgment

The authors would like to thank the Rogers Corporation for

donating the high-frequency laminates, and Mr. Ming-Yi Li of

Texas A&M University for his technical assistance and helpful

suggestions.

6. References

1. A. Vallecchi and G. Gentili, "Design of Dual-Polarized Series­

Fed Microstrip Arrays with Low Losses and High Polarization

Purity," IEEE Transactions on Antennas and Propagation, AP-S3,

5, May 2005, pp. 1791-1798.

3. X. Qu, S. Zhong, Y. Zhang, and W. Wang, "Design of an SIXDual-Band Dual-Polarised Microstrip Antenna Array for SAR

Applications," lET Microwave, Antennas, and Propagation, 1, 2,

April 2007, pp. 513-517.

5. G. Chattopadhyay and J. Zmuidzinas, "A Dual-Polarized Slot

Antenna for Millimeter Waves," IEEE Transactions on Antennas

and Propagation, AP-46 , 5, May 1998, pp. 736-737.

2. S. Gao and A. Sambell, "Dual-Polarized Broad-Band Microstrip

Antennas Fed by Proximity Coupling," IEEE Transactions on

Antennas and Propagation, AP-S3, 1, January 2005, pp. 526-530.

4. S. Zhong, X. Yang, S. Gao, and J. Cui, "Comer-Fed Microstrip

Antenna Element and Arrays of Dual-Polarization Operation,"

IEEE Transactions on Antennas and Propagation, AP-SO, 10,

October 2002, pp. 1473-1480.-30(dB)

-- Dual layer

- - - X-band layer only

Figure 9b. Comparisons of the X-band radiation patterns of

the H-port feed for the H plane.

0.1 mm, the frequency is raised by 40 MHz; while if h 2 is

decreased by 0.1 mm, the frequency is reduced by about 60 MHz.

6. K. Mak, H. Wong, and M. Luk, "A Shorted Bowtie Patch

Antenna with a Cross Dipole for Dual Polarization," IEEE Anten­

nas and Wireless Propagation Letters, 6, 2007, pp. 126-129.

For the X-band array, only small variations in the sidelobe

levels were observed, and the measured peak gain dropped about

0.5 dB with the presence of the top S-band layer. Comparisons of

the radiation patterns of the H port are shown in Figure 9. The S­

band layer presented little effect on the performance of the X-band

layer. The statement that one antenna layer was transparent to the

other one was therefore confirmed.

7. S. Gao and A. Sambell, "Dual-Polarized Broad-Band Microstrip

Antennas Fed by Proximity Coupling," IEEE Transactions on

Antennas and Propagation, AP-S3, 1, January 2005, pp. 526-530.

8. K. Ghorbani and R. Waterhouse, "Dual Polarized Wide-Band

Aperture Stacked Patch Antennas ," IEEE Transactions on Anten­

nas and Propagation, AP-S2, 8, August 2004, pp. 2171-2174.

4. Conclusions

A dual-frequency (S-band and X-band) dual-polarization

array antenna has been developed. An ultra-thin structure was

adopted for the purpose of use with aircraft. The conformal array

can be installed on the airframe or inside the aircraft, due to its

small size and light weight. The X-band array used a series-fed

configuration to save the space of the feeding-line network. The S­

band array adopted a larger radiation aperture to decrease the num­

ber of elements, to reduce the blockage, and to enhance the radia­

tion gain. The V and H ports were put on two separate elements to

achieve high isolation. More subarrays could be assembled to

obtain higher gain with a narrower beamwidth. The newly devel­

oped dual-frequency dual-polarization array antenna should be use­

ful for future wireless communications, remote sensing, surveil­

lance, radar systems, and UAV applications .

9. D. Krishna, M. Gopikrishna, C. Aanandan, P. Mohanan, and K.Vasudevan, "Compact Dual-Polarized Square Microstrip Antenna

with Triangular Slots for Wireless Communication," lEE Electron­

ics Letters, 42,16, August 2006, pp. 894-895.

10. K. Wong and T. Chiou, "Design of Dual Polarized Patch

Antennas Fed by Hybrid Feeds," Proceedings of the IEEE Fifth

International Symposium on Antennas, Propagation, and EM The­

ory, Beijing, China, August 2000, pp. 22-25.

11. Y. Guo and K. Luk, "Dual-Polarized Dielectric Resonator,"

IEEE Transactions on Antennas and Propagation, AP-51, 5, May

2003,pp.1120-1123.

12. B. Lindmark, S. Lundgren, J. Sanford, and C. Beckman, "Dual­

Polarized Array for Signal-Processing Applications in Wireless

Communications," IEEE Transactions on Antennas and Propaga­

tion, AP-46, 6, June 1998, pp. 758-763.

76 IEEE Antennas and Propagation Magazine , Vol. 51, No.4, August 2009

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13. A Parfitt and N. Nikolic, "A Dual-Polarised Wideband Planar

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Introducing the Feature Article Authors

Texas -A&M University, he conducted research on wireless power

transmission, power combining, ultra-wideband antennas, and

phased arrays. In 2007, he joined Intelligent Automation Inc.,

Rockville, MD, where he is involved in research and development

of advanced antennas and RP/microwave systems, including

conformal antennas, metamaterial antennas, VAV collision avoid­

ance radar, airborne weather radar, through-the-wall noise radar,

and RPID sensors for biological-warfare agent detection and struc­

ture health monitoring.

Kai Chang received the BSEE degree from the National Tai­

wan University, Taipei, Taiwan; the MS degree from the State

University of New York at Stony Brook; and the PhD degree from

the University of Michigan, Ann Arbor; in 1970, 1972, and 1976,

respectively.

From 1972 to 1976, he worked for the Microwave Solid-State

Circuits Group, Cooley Electronics Laboratory of the University of

Michigan as a Research Assistant. From 1976 to 1978, he was

employed by Shared Applications, Inc., Ann Arbor, where he

worked in computer simulation of microwave circuits and micro­

wave tubes . From 1978 to 1981, he worked for the Electron

Dynamics Division, Hughes Aircraft Company, Torrance, CA,

Yu-Jiun Ren received the BSEE from National Chung­

Hsing University, Taiwan; the MS degree in Communication Engi­

neering from National Chiao-Tung University, Taiwan; and the

PhD degree from Texas A&M University at College Station; in

2000,2002, and 2007 , respectively. From 2002 to 2003, he was a

research assistant with the Radio Wave Propagation and Scattering

Laboratory, National Chiao-Tung University, and involved in

mobile-radio propagation, channel modeling, and cell planning. At

Shih-Hsu Hsu received the BSEE degree from National

Cheng-Kung University, Taiwan; the MS degree in Electrical Engi­

neering from the University of Wisconsin-Madison; and the PhD

degree from Texas A&M University at College Station; in 2000,

2004, and 2008 , respectively. At Texas A&M University, his

research activities involved microstrip reflectarrays, reconfigurable

antennas, and wideband antennas. In October 2008, he joined

Applied Optoelectronics, Inc., where he is involve in high-speed

laser design.

IEEE Antennas and Propagation Magazine, Vol. 51, No.4, August2009 77

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where he was involved in the research and development of

millimeter-wave solid-state devices and circuits, power combiners,

oscillators and transmitters. From 1981 to 1985, he worked for the

TRW Electronics and Defense, Redondo Beach, CA, as a Section

Head, developing state-of-the-art millimeter-wave integrated cir­

cuits and subsystems including mixers, VCOs, transmitters,

amplifiers, modulators, up-converters, switches, multipliers,

receivers, and transceivers. He joined the Electrical Engineering

Department of Texas A&M University in August 1985 as an

Associate Professor, and was promoted to a Professor in 1988. In

January 1990, he was appointed Raytheon E-Systems Endowed

Professor of Electrical Engineering. His current interests are in

microwave and millimeter-wave devices and circuits, microwave

integrated circuits, integrated antennas, wideband and active anten­

nas, phased arrays, microwave power transmission, and microwave

optical interactions.

Dr. Chang has authored and coauthored several books. He

served as the editor of the four-volume Handbook of Microwave

and Optical Components, published by John Wiley in 1989 and

1990 (second edition 2003), and the editor for the Wiley

Encyclopedia of RF and Microwave Engineering (six volumes,

2005). He is the editor of Microwave and Optical Technology Let­

ters, and the Wiley book series in Microwave and Optical

Engineering (over 70 books published). He has published over 450

papers, and many book chapters in the areas of microwave and

millimeter-wave devices, circuits, and antennas. He has graduated

over 25 PhD students and over 35 MS students.

Dr. Chang has served as technical committee member and

session chair for IEEE MTT-S, AP-S, and many international

conferences. He was the Vice General Chair for the 2002 IEEE

International Symposium on Antennas and Propagation. He

received the Special Achievement Award from TRW in 1984, the

Halliburton Professor Award in 1988, the Distinguished Teaching

Award in 1989, the Distinguished Research Award in 1992, and

the TEES Fellow Award in 1996 from the Texas A&M University.

Dr. Chang is a Fellow of the IEEE.

78

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