Active PFC Passive PFC

i

Jadavpur University M.E.E Thesis

Design and Implementation of UPFC based Boost Converter for Efficiency Optimization of Brushless

DC Motor Drive System

Thesis submitted in partial fulfillment of the requirements for the degree of

MASTER OF ELECTRICAL ENGINEERING

By

Subhendu Bikash Santra

Roll No.M4ELE12-15

Reg. No.113465

Under the guidance of

Dr. Debashis Chatterjee

DEPARTMENT OF ELECTRICAL ENGINEERING

FACULTY OF ENGINEERING AND TECHNOLOGY

JADAVPUR UNIVERSITY KOLKATA-700 032

2012

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Declaration of Originality and Compliance of Academic Ethics I hereby declare that this thesis contains literature survey and original research work by the undersigned candidate, as a part of his Master of Electrical Engineering studies. All information in this document have been obtained and presented in accordance with academic rules and ethical conduct. I also declare that, as required by these rules and conduct, I have fully cited and referred all materials and results that are not original to this work. Name: Subhendu Bikash Santra Roll Number: M4ELE12-15 Thesis Title: Design and Implementation of UPFC based Boost Converter for Efficiency Optimization of Brushless DC Motor Drive System. Signature with Date:

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JADAVPUR UNIVERSITY

Faculty of Engineering and Technology

CERTIFICATE

We hereby recommend that the thesis prepared under our supervision and guidance by

Subhendu Bikash Santra entitled “Design and Implementation of UPFC based Boost

Converter for Efficiency Optimization of Brushless DC Motor Drive System” be

accepted in partial fulfillment of the requirements for the award of the degree of “Master

of Electrical Engineering” at Jadavpur University. The project, in our opinion, is worthy

of its acceptance.

Supervisor

……………………………

Dr. Debashis Chatterjee Associate Professor

Department of Electrical Engg. Jadavpur University

Kolkata

Countersigned

…………………………………… Prof. Nirmal Kumar Deb Head of the Department

Department of Electrical Engg. Jadavpur University

Kolkata

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JADAVPUR UNIVERSITY

Faculty of Engineering and Technology

*

CERTIFICATE OF APPROVAL

The foregoing thesis is hereby approved as a creditable study of an

engineering subject carried out and presented in a satisfactory manner to

warrant its acceptance as pre-requisite to the degree for which it has been

submitted. It is notified to be understood that by this approval, the

undersigned do not necessarily endorse or approve any statement made,

opinion expressed and conclusion drawn therein but approve the thesis only

for the purpose for which it has been submitted.

Final Examination for the Board of Examiners

Evaluation of Thesis

.………………………………….

…………………………………..

…………………………………..

*Only in case thesis is approved (signature of the Examiners)

v


Dedicated

To

My parents, sister and my teachers.

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ACKNOWLEDGEMENTS

Truly speaking I believe that just a mere “thank you” doesn’t do proper justice to peoples’ contribution towards most of the endeavors. But usually, that remains the only thing we can offer, not just to acknowledge their efforts but also to pacify our consciences.

The most important thing we learn or try to learn over our life is how to think in simpler ways. First and foremost, I would like to thank Jadavpur University,for providing me such a graceful opportunity for studying M.E.E. and do a specialization in field of Electrical Machines. I believe the man who epitomizes this process is my guide Prof. Ashoke Kumar Ganguli. I was amazed to watch him think from the very basic and sort out complicated issues in very logical ways. My co-guide Dr. Debashis Chatterjee helped me in every possible ways whenever I got stuck. Both of them are truly workaholic. I feel opportune that they gave me prospect to carry out research under their guidance and supervision in spite of their busy schedule.

I would like to thank research scholar and friend Krishna Roy ,Dipten Maity, Rupak Bhowmick who gave me their precise time for improvement of this project. Here I would like to thank my friends Bikram Dutta, Subhendu Dutta, Abhinandan Basak, Aloke Raj Sarkar, Sutirtha Sen and Uddipta Bhaumik for being my comfort zone. I could resolve various so called trivial issues by discussing with them. Thank you very much. You all have been remarkable.

I would also like to express gratitude to all laboratory assistants in the Electrical Machines Lab in the Department of Electrical Engineering for always expending their helping hands.

I am from a family, which always encourages for higher studies. My parents despite their modest formal education always give highest preference to my studies. I feel deep respect about my family. I am fortunate and proud to be a member of such a family.

Most importantly, I would like to give God the glory for all the efforts I have put into the work and for giving me the physical strength and mental perseverance to carry out the work.

Kolkata: ………………………………..

May, 2012 (Subhendu Bikash Santra)

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CONTENTS

ACKNOWLEDGEMENTS………………………………………………………………vi

CONTENTS……………………………………………………………………………..vii

LIST OF SYMBOLS……………………………………………………………………...x

LIST OF ABBREVIATION……………………………………………………………...xi

1 Introduction .....................................................................................................................1

1.1Background………………………………………………………………………….....1

1.2 Objective………………………………………………………………………………2

1.3 Review of relevant literature…………………………………………………………..2

1.4 Thesis outline………………………………………………………………………….4

2 Boost Converter System…………………………………………………………….....6

2.1 Boost Converter……………………………………………………………………….6

2.1.1 Circuit analysis for continuous mode……………………………………………….8

2.1.2 Circuit analysis for discontinuous mode…………………………………………...10

2.2 Power Factor Correction……………………………………………………………..12

2.2.1 Causes of low power factor………………………………………………………...13

2.2.2 Standards for line current harmonics………………………………………………13

2.2.3 The need of PFC…………………………………………………………………...14

2.2.4 Types of PFC………………………………………………………………………15

2.2.4.1 Passive PFC……………………………………………………………………...15

2.2.4.2 Active PFC……………………………………………………………………….16

2.3 Current Mode Control………………………………………………………………..17

2.3.1 Purpose of current mode control…………………………………………………..18

2.3.2 Types of current mode control……………………………………………………..19

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2.4 Average current mode control……………………………………………………….19

2.4.1 Advantage of average current mode control……………………………………….20

2.4.2 Disadvantage of average current mode control……………………………………20

2.5 Active power factor controller using UC3854……………………………………….20

2.5.1 Feed-Forward Loop Compensation………………………………………………..23

2.5.2 Voltage Loop Bandwidth Selection………………………………………………..24

2.5.3 High PFC Controller Circuit……………………………………………………….25

2.6 Proposed Design……………………………………………………………………..29

3 Brushless DC Motor………………………………………………………………….37

3.1 Comparison with Brushed DC Motor………………………………………………..38

3.2 Brushless DC Motor vs Normal AC Induction Motor…………………………….....40

3.3 Components of a Brushless DC Motor………………………………………………41

3.4 Construction of Typical BLDC Motor……………………………………………….41

3.4.1 Stator……………………………………………………………………………….41

3.4.2 Rotor……………………………………………………………………………….43

3.4.3 Hall Sensors………………………………………………………………………..43

3.5 Working Principle of BLDC Motor………………………………………………….45

3.6 Torque/Speed Characteristics………………………………………………………..46

3.7 Typical BLDC Motor Application…………………………………………………...48

4 Harmonic Analysis of Back E.M.F and Phase Current Waveform of Trapezoidal BLDC Motor …………………………………………………………………………….51

4.1Back E.M.F of non sinusoidal BLDC machine………………………………………52

4.2 Phase Current of non sinusoidal BLDC machine……………………………………53

4.3 Different Conduction Mode of Trapezoidal BLDC Motor…………………………..53

4.4 Harmonic analysis of Ideal current waveform for different conduction mode………54

4.5 Rising Angle…………………………………………………………………………55

4.6 Harmonic analysis of phase current waveform considering delay angle………...…..55

5 Loss minimization and efficiency improvement.........................................................61

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5.1 Loss calculation……………………………………………………………………...61

5.1.1 Stator Resistive Losses…………………………………………………………….61

5.1.2 On state conduction loss across switch…………………………………………….64

5.1.3 Switching Loss……………………………………………………………………..64

5.1.4 Iron Loss…………………………………………………………………………...67

5.2 Efficiency Comparison and improvement through proposed technique……………..70

6 Simulation and Experimental Result..........................................................................73

6.1 Simulink M File……………………………………………………………………...74

6.2 PFC controller Block………………………………………………………………...75

6.3 Simulink results and Output Waveforms…………………………………………….76

6.4 Hardware Configuration……………………………………………………………..81

6.5 Experimental Result………………………………………………………………….82

7 Conclusions……………………………………………………………………………89

7.1 Conclusions…………………………………………………………………………..89

7.2 Scope for future work………………………………………………………………..89

References……………………………………………………………………………….90

x


LIST OF SYMBOLS

Ea

Back E.M.F of BLDC motor (V)

t Time (sec)

Ia

Phase current of BLDC motor (A)

I Maximum value of phase current (A)

P Average power (watt)

Vi Input voltage to boost converter (V)

IL

Inductor current (A)

Vo

Output voltage (V)

L Inductance (H)

E Stored energy (joules)

D Duty ratio

Io

Output current (A)

Id

Diode current (A)

Co

Output capacitance (F)

Iph

Peak inductor current (A)

∆I,Iripple

Inductor ripple current (A)

Rff

Feedforword resistance (ohms)

Greek letters

ω Speed of rotor (rad/s)

α Switching angle (degree)

xi


LIST OF ABBREVIATION

BLDC Brushless Direct Current Machine

E.M.F Electro Motive Force

PFC Power Factor Correction

EMI Electromagnetic Interference

ACMC Average Current Mode Control

PWM Pulse Width Modulation

CEA Current Error Amplifier

SMPS Switching-mode power supply

CCM Continuous Conduction Mode

THD Total Harmonic Distortion

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1

Introduction

This thesis covers the analysis, design and implementation of UPFC using UC3854

which is used to drive BLDC motor for efficiency optimization issue. In common

practice if we use simple rectifier to convert AC to DC voltage then input current

contains significant harmonics which is undesirable where efficiency of conversion is a

major topic.

1.1 Background:

Electronic equipments recently in use ( PCs, TVs, and Telecommunication Equipments

etc.) require power conditioning of some form, typically rectification, for their

proper working. But since they have non-linear input characteristics and they are

connected the electricity distribution network they produce a non-sinusoidal line

current. Current of frequency components which are multiples of the natural

frequency are produced that are otherwise called the line harmonics. With

constantly increasing demand of these kind of equipments at a high rate, line

current harmonics have become a significant problem. There has been an introduction

of a lot of international standards which pose limitations on the harmonic content in

the line currents of equipments connected to electricity distribution networks. This

calls for measures to reduce the line current harmonics which is also otherwise

known as Power Factor Correction - PFC.

There exist two kinds of power factor correction techniques – passive power factor

correction and active power factor correction. In this thesis we tried to devise an active

power factor correction method for improvement of the power factor. In this work

the advantages of a boost converter is combined with that of the average current

mode control to implement the technique. UC3854 was used to design the power

factor corrector. This integrated circuit had all the circuits necessary to control a power

factor corrector and was designed to implement the average current mode control.

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Permanent magnet brushless DC (PMBLDC) motors are the latest choice of researchers

due to their high efficiency, silent operation, compact size, high reliability and low

maintenance requirements. These motors are preferred for numerous applications;

however, most of them require sensorless control of these motors. The operation of

PMBLDC motors requires rotor-position sensing for controlling the winding currents.

The sensorless control would need estimation of rotor position from the voltage and

current signals, which are easily sensed. This thesis presents state of the art PMBLDC

motor drives with an improved efficiency. PMBLDC motors find applications in diverse

fields such as domestic appliances, automobiles, transportation, aerospace equipment,

power tools, toys, vision and sound equipment and healthcare equipment ranging from

microwatt to megawatts. Advanced control algorithms and ultra fast processors have

made PMBLDC motors suitable for position control in machine tools, robotics and high

precision servos, speed control and torque control in various industrial drives and process

control applications.

1.2 Objective:

BLDC based drive circuits are usually used in battery operated vehicles or system

requiring high overall efficiency. Since the input to the system is usually at lower voltage,

a boost topology is generally used. Thus the efficiency of the input system to the inverter

is also important for overall efficiency improvement of the system. In this thesis an

UPFC based boost converter topology using UC3854 is designed and implemented. The

results obtained tally with the simulation results. Also the switching of the inverter

introduces different order of harmonics to the machine which creates losses to the

machine. In this thesis two well known topologies e.g.120° and 180° switching schemes

are studied and compared losses due to harmonics to arrive at a composite switching

scheme which takes care of the torque ripple and overall efficiency of the system.

1.3 Review of relevant literature:

The use of permanent magnets (PMs) in electrical machines in place of electromagnetic

excitation results in many advantages such as no excitation losses, simplified

construction, improved efficiency, fast dynamic performance, and high torque or power

3


per unit volume. The PM excitation in electrical machines was used for the first time in

the early 19th century, but was not adopted due to the poor quality of PM materials. In

1932, the invention of Alnico revived the use of PM excitation systems, however it has

been limited to small and fractional horse power dc commutator machines. In the 20th

century, squirrel cage induction motors have been the most popular electric motors, due

to its rugged construction. Advancements in power electronics and coercive force as

compared to ceramic magnets are economical but their maximum energy density product

is low due to lower values of retentivity. however, Neodymium-Iron-Boron (Nd-Fe-B)

rare earth magnets are more in demand because they provide the highest energy density

and higher residual flux density than others [12-14].

PMBLDC motors are generally powered by a conventional three-phase voltage source

inverter (VSI) or current source inverter (CSI) which is controlled using rotor position.

The rotor position can be sensed using Hall sensors, resolvers, or optical encoders

[4][5][14].

Recently some additional applications of PMBLDC motors have been reported in electric

vehicles (EVs) and hybrid electric vehicles (HEVs) due to environmental concerns of

vehicular emissions. PMBLDC motors have been found more suitable for EVs/HEVs and

other low power applications, due to high power density, reduced volume, high torque,

high efficiency, easy to control, simple hardware and software and low maintenance [6-

8].

As the use of energy is increasing, the requirements for the quality of the supplied

electrical energy are more tighten. This means that power electronic converters must be

used to convert the input voltage to a precisely regulated DC voltage to the load.

Regulated DC power supplies are needed for PMBLDC motor drive system. Most power

supplies are designed to meet regulated output, isolation and multiple outputs. SMPS are

needed to convert electrical energy from AC to DC. SMPS are used as a re placement of

the linear power supplies when higher efficiency, smaller size or lighter weight is

required. Motors, electronic power supplies and fluorescent lighting consume the

majority of power in the world and each of these would benefit from power factor

4


correction. In the middle of 1990s, many of the countries of the world have adopted re

quirements for power factor correction for new products marketed [15].

Improved power quality converters are mostly required for many applications involving

power converters. PMBLDC motors are suitable for many low power applications due to

high efficiency and wide speed control [9-11]. Increasing concern of PQ problems in the

international power communities has prompted the use of power factor correction (PFC)

converters with a PMBLDCM. Since, these PMBLDCMs are fed from a single-phase AC

mains through a diode bridge rectifier (DBR) and a smoothening DC link capacitor,

which results in a pulsed current from AC mains having various power quality (PQ)

disturbances such as poor power factor (PF), increased total harmonic distortion (THD)

and high crest factor (CF) of current [9]. Moreover, various international PQ standards

for low power applications such as IEC 61000-3-2 [10] , emphasize on low THD of AC

mains current and near unity power factor. Therefore, use of a PFC converter topology

for a PMBLDC motor drive amongst various available topologies [11] is essential.

Some [16-21] application note are the invaluable source of understanding UPFC boost

converter design used to drive PMBLDC motor.

1.4 Thesis outline:

The thesis presents Efficiency Optimization Of BLDC Motor Drive Systems. Harmonics

contents of Back E.M.F waveform of BLDC machine with different conduction angle is

calculated. DC voltage is given to the BLDC machine. But normal rectifier with capacitor

has lower conversion efficiency as input AC current is peaky in nature rich with

harmonics. Thus with the increase of supply converter efficiency overall drive system

efficiency will increase. The different parts of the thesis work include:

Chapter 1, the introductory chapter it is discussed the importance of PMBLDC motor in

present day application and how efficiency can be improved.

Chapter 2, Boost Converter with UPFC is analyzed and designed using UC3854

Chapter 3, Elementary study of BLDC motor is done and its importance is analyzed.

5


Chapter 4, Harmonic contents of non-sinusoidal BLDC motor for both 120° and 180°

Phase current conduction mode is analyzed in this chapter.

Chapter 5, a discussion of loss minimization and efficiency improvement through

proposed technique is discussed in this chapter.

Chapter 6, a discussion on software simulated result and experimental result is presented.

Chapter 7, is the concluding chapter where conclusion of the thesis is drawn and scope of

future work is discussed.

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2

Boost Converter System

2.1 BOOST CONVERTER:

It is a type of power converter in which the DC voltage obtained at the output stage is

greater than that given at the input. It can be considered as a kind of switching-mode

power supply (SMPS). Although it can be formed in different configurations, the

basic structure must have at least two semiconductor switches (generally a diode and

a transistor) and one energy storing element must be used.

Among the three basic power converters buck, boost, buck-boost the boost converter is

the most suitable for use in implementing PFC. Because the boost inductor is in series

with the line input terminal, the inductor will achieve smaller current ripple and it is

easier to implement average current mode control. Buck converter has discontinuous

input current and would lose control when input voltage is lower than the output voltage.

The buck-boost converter can achieve average input line current, but it has higher voltage

and current stress, so it is usually used for low-power application . The power stage

adopted in this thesis is boost converter operating in continuous conduction mode. Figure

2.1 shows the circuit diagram of the boost PFC converter.

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Fig.2.1.Boost Converter

The Boost converter assumes two distinct states.

The On-state, in which the switch S in Fig 2.1 is closed, and then there is a constant

increase in the inductor current. The Off-state, in which the switch S is made open and

the inductor current now flows through the diode D, the load R and the capacitor C. In

this state, the energy that has been accumulated in the inductor gets transferred to the

capacitor. The input current and the inductor current are the same. Hence as one can

see clearly that current in a boost converter is continuous type and hence the design of

input filter is somewhat relaxed or it is of lower value.

Fig.2.2.The two states of a boost converter that change with change in state of the switch

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2.1.1 Circuit analysis for Continuous mode:

During continuous mode of operation of a boost converter, the inductor currentL

(I ) never

becomes zero during a commutation cycle.

Fig.2.3.Current and voltage waveforms while a boost converter operates in continuous mode.

The switch S is closed to start the On-state. This makes the input voltage

L(V ) appear

across the inductor, and that causes change in inductor currentL

(I ) during a finite time

period (t) which is given by the formula:

∆I VL i=∆t L

(2.1)

When the On-state reaches its end, the total increase inIL

is given by:

DT1 DT∆I = V dt= Vi iL L Lon 0

∫ (2.2)

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Where D is known as the duty cycle i.e. the ratio of time period for which the switch is

On and the total commutating time period T. Therefore D has a value between 0 ( that

indicates S is never on) and 1 ( that indicates S is always on).

When the switch S is made open the converter operates in Off-state. During that

time period the load serves as a path for the inductor current. If voltage drop in the diode

is neglected or assumed to be zero, and the capacitor is taken to be large enough

for maintaining a constant voltage, the equation ofIL

is given by:

dILV -V =Li o dt

(2.3)

During the time period for which the converter remains in Off state, the change in

IL

is given by:

(1-D)T V -V (V -V )(1-D)Ti o i o∆I = dt=Loff L L0

∫ (2.4)

As we consider that the converter operates in steady-state conditions, the amount

of energy stored in each of its components has to be the same at the beginning and at the

end of a commutation cycle. In particular, the energy stored in the inductor is given by:

1 2E = L I L2 (2.5)

Therefore, the inductor current has to be the same at the beginning and the end of the commutation cycle. This can be written as

I + I =0Lon Loff∆ ∆

(2.6)

Substituting and by their expressions yields:

(V -V )(1 -D )TV D T i oiI + I = + = 0L o n L o f f L L∆ ∆

(2.7)

which in turns reveals the duty cycle to be :

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ViD=(1- )Vo

(2.8)

From the above expression it is observable that the output voltage is always greater than the input voltage (as D is a number between 0 and 1), and that it increases as D increases. Theoretically it should approach infinity as D approaches 1. For this

reason boost converter is also known as step-up converter.

2.1.2 Circuit analysis for Discontinuous mode The only difference in the principle of discontinuous mode as compared to the

continuous mode is that the inductor is completely discharged at the end of the

commutation cycle. In this node of operation before the switch in the circuit is

opened the inductor current value reaches zero. This kind of case happens when the

energy to be transferred is very small and the process of transfer requires a time period

less than the commutating time period.

Fig.2.4.Waveforms of current and voltage in a boost converter operating in discontinuous mode

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As the inductor current at the beginning of the cycle is zero, its maximum value

V D TiI =L m a x L

(2.9)

During the off-period, IL

falls to zero after δT:

i o(V -V ) δT

I + =0Lm ax L

(2.10)

Using the two previous equations, δ is:

V Di δ =V - Vi o

(2.11)

The load current Io is equal to the average diode current (Id

). As can be seen on figure 4,

the diode current is equal to the inductor current during the off-state. Therefore

the output current can be written as:

o D

I Lm axI = I = δ2

(2.12)

Replacing andILmax

and δ by their respective expressions yields:

22D TV DT V D Vi i iI = × =o 2L V -V 2L(V -V )o i o i

(2.13)

Therefore, the output voltage gain can be written as flow

2V ViD To =1+V 2LIoi

2.14)

In comparison to the output voltage expression for the continuous mode, this expression

is much more complicated. Furthermore, in discontinuous operation, the output voltage

12


gain not only depends on the duty cycle, but also on the inductor value, the input

voltage, the switching frequency, and the output current.

The Active PFC method proposed in this thesis deals with the continuous mode of operation for its simplicity and easy design process.

2.2 Power factor Correction

The term, power factor can be defined as the ratio of real power to apparent power.

P.F= P

V ×Ir.m.s r.m.s =

AveragePowerApprentPower

(2.15)

Assuming an ideal sinusoidal input voltage source, PF can be expressed as the product of

two factors: the displacement factor Kθ

and the distortion factorKd

.The displacement

factorKθ

is the cosine of the displacement angle between the fundamental input current

and the input voltage. The distortion factorKd

is the ratio of the root-mean-square

(RMS) of the fundamental input current to the total RMS of input current. These

relationships are given as follows:

V I cosθrms rms1P.F K Kθ dV Irms rms

= = (2.16)

Where: Vrms

is the total RMS voltage value.

Irms

is the total RMS current value.

1

Irms

is the current fundamental harmonic RMS value.

θ is the displacement angle between the voltage and current fundamental harmonics.

K =cosθθ

is the displacement factor.

Kd

= 1

Irms

Irms

is the distortion factor.

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Power factor correction is a modern concept which deals with increasing the degraded

power factor of a power system by use of external equipments. The objective of

this described in plain words is to make the input to a power supply appear as a

simple resistor. As long as the ratio between the voltage and current is a constant the

input will be resistive and the power factor will be 1.0. When the ratio deviates from a

constant the input will contain phase displacement, harmonic distortion or both and

either one will degrade the power factor.

In simple words, Power factor correction (PFC) is a technique of counteracting the

undesirable effects of electric loads that create a power factor ( PF ) that is less than 1.

2.2.1Causes of low power factor

he power factor gets lowered as the real power decreases in comparison to the

apparent power. This becomes the case when more reactive power is drawn. This may

result from increase in the amount of inductive loads (which are sources of

Reactive Power) which include - Transformers, Induction motors, Induction generators

(wind mill generators), High intensity discharge (HID) lighting etc. However in such

a case the displacement power factor is affected and that in turn affects the power factor.

The other cause is the harmonic distortion which is due to presence of the non

linear loads in the power system. Due to the drawing of non-sinusoidal current there is

further reduction in the power factor.

2.2.2 Standards for line current harmonics

For limiting the line current harmonics in the current waveform standards are set

for regulating them. One such standard was IEC 555-2, which was published by

the International Electro-technical Committee in 1982. In 1987, European

Committee for Electro-Technical Standardization – CENELEC, adopted this as an

European Standard EN 60555-2. Then standard IEC 555-2 has been replaced by standard

IEC 1000-3-2 in 1995. The same has been adopted as an European standard EN

61000-3-2 by CENELEC.

14


Hence, these limitations are kept in mind while designing any instrument. So that there is

no violation and the negative effects of harmonics are not highly magnified.

2.2.3 The need of PFC

Constant increasing demand of consumer electronics has resulted in that the average

home has a huge variety of mains driven electronic devices. These electronic

devices have mains rectification circuits, which is the dominant reason of mains

harmonic distortion. A lot of modern electrical and electronic apparatus require to

convert ac to dc power supply within their architecture by some process. This causes

current pulses to be drawn from the ac network during each half cycle of the supply

waveform. Though a single apparatus (a domestic television for example) may not

draw a lot of reactive power or it cannot generate enough harmonics to affect the

supply system significantly, but within a typical phase connection there may exist 100s of

such devices connected to the same supply phase resulting in production of a significant

amount of reactive current flow and current harmonics.

With improvement in semiconductor devices field, the size and weight of control circuits

are on a constant decrease. This has also positively affected their performance and

functionality and thus power electronic converters have become increasingly popular in

industrial, commercial and residential applications. However this mismatch between

power supplied and power put to use cannot be detected by any kind of meter used for

charging the domestic consumers. It results in direct loss of revenues.

Furthermore 3-phase unbalance can also be created within a housing scheme since

different streets are supplied on different phases. The unbalance current flows in

the neutral line of a star configuration causing heating and in extreme cases cause burn

out of the conductor.

The harmonic content of this pulsating current causes additional losses and

dielectric stresses in capacitors and cables, increasing currents in windings of rotating

machinery and transformers and noise emissions in many products, and bringing about

early failure of fuses and other safety components. The major contributor to this problem

in electronic apparatus is the mains rectifier. In recent years, the number of

15


rectifiers connected to utilities has increased rapidly, mainly due to the growing use of

computers.

Hence it has become very necessary to somehow decrease the effect of this distortion.

Power factor correction is an extra loop added to the input of household applications to

increase the efficiency of power usage and decrease the degree of waste.

2.2.4 Types of Power Factor Correction (PFC) Power Factor Correction (PFC) can be classified as two types :

• Passive Power Factor Correction.

• Active Power Factor Correction.

2.2.4.1 Passive PFC In Passive PFC, only passive elements are used in addition to the diode bridge rectifier, to

improve the shape of the line current. By use of this category of power factor

correction, power factor can be increased to a value of 0.7 to 0.8 approximately.

With increase in the voltage of power supply, the sizes of PFC components increase in

size. The concept behind passive PFC is to filter out the harmonic currents by use of a

low pass filter and only leave the 50 Hz basic wave in order to increase the power factor.

Passive PFC power supply can only decrease the current wave within the standard and

the power factor cannot never be corrected to 1. And obviously the output

voltage cannot be controlled in this case.

Advantage Of Passive PFC Disadvantage Of Passive PFC

It has a simple structure. For achieving better power factor the

dimension of the filter increases.

It is reliable and rugged. Due to the time lag associated with the

passive elements it has a poor dynamic

response.

In this equipments used don’t generate The voltage cannot be regulated and the

16


high-frequency EMI. efficiency is somewhat lower.

Only the construction of a filter is

required which can be done easily. Hence

the cost is very low.

Due to presence of inductors and capacitors

interaction may take place between the

passive elements or they may interact with

the system and resonance may occur at

different frequencies.

The high frequency switching losses

are absent and it is insensitive to

noises and surges.

Although by filtering the harmonics can be

filtered out, the fundamental component

may get phase shifted excessively thus

reducing the power factor.

The shape of input current is dependent

upon the fact that what kind of load is

connected.

2.2.4.2 Active PFC

An active PFC is a power electronic system that is designed to have control over

the amount of power drawn by a load and in return it obtains a power factor as

close as possible to unity. Commonly any active PFC design functions by controlling

the input current of the load in order to make the current waveform follow the

mains voltage waveform closely (i.e. a sine wave). A combination of the reactive

elements and some active switches are in order to increase the effectiveness of the line

current shaping and to obtain controllable output voltage.

The switching frequency further differentiates the active PFC solutions into two classes.

A. Low frequency active PFC:

Switching takes place at low-order harmonics of the line-frequency and it is synchronized with the line voltage.

17


B. High frequency active PFC:

The switching frequency is much higher than the line frequency.

The power factor value obtained through Active PFC technique can be more than 0.9.

With a suitable design even a power factor of 0.99 can be reached easily. Active PFC

power supply can detect the input voltage automatically, supports 110V to 240V

alternative current, its dimension and weight is smaller than passive PFC power supply

which goes against the traditional view that heavier power supply is better.

Advantages of Active PFC Disadvantages of Active PFC

The weight of such a system is very less The layout design is bit more complex.

The dimension is also smaller and a

power factor value of over 0.95 can be

obtained through this method

Since it needs PFC control IC, high voltage

MOSFET, high voltage U-fast, choke and

other circuits; it is highly expensive.

Diminishes the harmonics to remarkably

low values

By this method automatic correction can

be obtained for the AC input voltage.

It is capable of operating in a full range

of voltage.

2.3 CURRENT MODE CONTROL

Current mode control uses the load current as feedback to regulate the output voltage. In

this approach there is direct control over the load current whereas output voltage is

controlled indirectly, hence it is called "current-mode programming"

In this control a functional block using local feedback is formed to create a voltage-to-

current converter. By using this voltage-to-current converter block inside an overall

Jadavpur University

voltage feedback loop, a voltage regulator can be produced where the control

voltage sets the load current rather than the switch duty cycle (as in the voltage

mode programming in which duty cycle is varied as it is directly proportional to

the control voltage). Figure 2.5

Fig.2.5. Block diagram of an ideal current mode converter

2.3.1 Purpose of Current Mode Control:

The current-mode approach offers the following advantages

• Since the output current is proportional to the control voltage

can be limited simply by clamping the control voltage.


oad current rather than the switch duty cycle (as in the voltage


he control voltage). Figure 2.5 is a block diagram of the concept.

. Block diagram of an ideal current mode converter

Purpose of Current Mode Control:

mode approach offers the following advantages

Since the output current is proportional to the control voltage

can be limited simply by clamping the control voltage.

18

M.E.E Thesis


oad current rather than the switch duty cycle (as in the voltage


. Block diagram of an ideal current mode converter

Since the output current is proportional to the control voltage, the output current

19


• The energy storage inductor is effectively absorbed into the current source. A

simpler compensation network can stabilize the control-to-output transfer

function.

When this is applied in higher power applications, parallel connection is used for the

power stages. The power stages can be made to share equal current by connecting them to

a common bus. This is possible because the output current is proportional to the control

voltage.

Last is the automatic feed forward from the line voltage. This particular feature is

actually more readily attained in voltage-mode converters by a technique known as

"ramp compensation". In fact, in current-mode converters perfect feed forward is

obtained only by a particular value of slope compensation.

2.3.2 Types of Current Mode Control:

There are various types of current control schemes. Generally a scheme would be

named based on the type of inductor current information being sensed and/or how the

information is used to control the power switches.

The various current mode control schemes are – average current control, peak current

control, hysteresis control, borderline control, valley current control, emulated

current control.

2.4 Average Current Mode Control (ACMC):

In this current mode control scheme the inductor current is sensed and filtered by

a current error amplifier and the output from it drives a PWM modulator. By doing

this extra step the inner current loop minimizes the error between the average input

current and its reference. This latter is obtained in the same way as in the peak current

control.

Average Current Mode Control is typically a two loop control method (inner loop,

current; outer loop, voltage) for power electronic converters. The main distinguishing

feature of ACMC, as compared with peak current mode control, is that ACMC

20


uses a high gain, wide bandwidth Current Error Amplifier (CEA) to force the average

of one current within the converter, typically the inductor current, to follow the

demanded current reference with very small error, as a controlled current source.

Below in Fig 2.7 the scheme for average current mode control is shown. This technique

of average current mode control overcomes the problems of peak current mode control by

introducing a high gain integrating current error amplifier (CEA) into the current loop.

The gain-bandwidth characteristic of the current loop can be tailored for optimum

performance by the compensation network around the CA. Compared with peak current

mode control, the current loop gain crossover frequency, can be made approximately the

same, but the gain will be much greater at lower frequencies.

2.4.1 Advantages of average current mode control :

• It also operates with a constant switching frequency.

• In this case any compensation ramp is not required.

• Since the current is filtered the control is less sensitive to commutation noises

unlike peak current mode control.

• Better input current waveforms than for the peak current control since, near the

zero crossing of the line voltage, the duty cycle is close to one.

2.4.2 Disadvantages of average current mode control :

• The inductor current needs to be sensed which is not easy.

• In this current mode control scheme a current error amplifier is needed. For this

error amplifier a compensation network needs to be designed in addition,

and that must account for different converter operating points.

2.5 Active power factor controller using UC3854:

As shown in Figure 2.6 (a), the controller has two tasks:Current tracking forces the

average inductor current to track the current reference so that it has the same shape as the

input voltage, as shown in Figure 2.6 (b). This task gives the input a unity PF.Voltage

21


regulation regulates the output voltage keeping the output voltage equal to 50V, which is

higher than the input voltage as shown in Fig.2.6.(c).

Fig.2.6.Boost PFC converter controller: (a) Boost PFC with controller, (b) Waveforms of input voltage and inductor current, (c) Waveforms of input voltage and output voltage.

22


The analog controller for PFC is often achieved by a current-mode PFC control chip.In

this thesis Unitrode UC3854 chip is used . The analog control structure for a single

switch CCM PFC boost converter is illustrated in Figure 2.7. The PFC converter has a

three-loop control structure. The fast current loop keeps the input current the shape of the

input voltage, which renders the unity PF. The input voltage feed-forward loop is to

compensate the input voltage variation . The voltage loop keeps the output voltage at

fixed value (in this thesis it is 50 volt) .The voltage loop is very slow to avoid introducing

2nd harmonic ripple into the current reference,illustrated in 2.5.2.

Fig.2.7.Analog average current control For boost-type PFC

23


2.5.1 Feed-forward Loop Compensation:

The current reference is given by:

2

ABI =ref C

(2.17)

Where A=K ×Vin in

( K :in

Input voltage gain), B=Vc

( Vc

: Voltage compensator

output) and C=K .Vff in_rms

( K :ff

Input voltage feed forward gain).

Assume the inductor current tracks the reference perfectly. The input current is

proportional to the input voltage, which means the voltage loop can be affected by input

voltage variation. The feed-forward loop is inserted to compensate the line voltage

variation . C is in proportion to the input-voltage RMS value. It is derived from a second

order low-pass filter (as shown in Fig.2.8).

Fig.2.8.voltage feedforword loop

Because both the input voltage and output voltage contain 2ndharmonic component, there

are ripples in B and C. The ripples in components B and C are passed into the current

reference. From Equation (2.17), it is derived that:

i B Cref = -2

Ci Bref

∆ ∆ ∆∆

. (2.18)

Although the phases of B ∆ and C∆ are unknown, the worst case occurs when they have

a 180 phase shift:

24


i B Cref = +2Ci B

ref

∆ ∆ ∆

(2.19)

If given a maximum acceptable THD of 1.5%, Equation (2.19) means that thedistortion

of C should be smaller than 0.5%, and the total distortion of B should be smaller than

0.5%. Selecting the cut-off frequency of the feed-forward low -pass filter and voltage

compensator gain should be based on this criterion.

2.5.2 Voltage Loop Bandwidth Selection:

Assuming that input voltage is

V = 2V sin(ωt) ,in rms that the input current isI = 2I sin(ω t) ,

in rms

ω=2πf angular frequency where f is the line frequency.

P =P =V I =V I (1-cos2ωt).o in in in m m where V = 2V

m rms

Assume that the output voltage varies small enough to be constant. Then the output

current, as shown in Figure 2.9 is

P V Io in inI = = (1-cos2ωt)o V Vout out (2.20)

Equation (2.20) indicates that the output current consists of a large 2nd harmonic

component, as shown in Figure 2.9 (b), which is given by:

V Iin inI =(- )cos2ωtripple Vout

(2.21)

This current ripple charges and discharges the output capacitor, leading to the2nd

harmonic ripple at the output voltage, such that:

V Iin inV = sin(2ωt+π)ripple 2V ωCout o (2.22)

25


V Iripple in inV =o 2V ωCout o (2.23)

Fig.2.9. (a) Input voltage and current waveform, (b) Average Diode forward current waveform.

The 2nd harmonic in the output voltage produces a fundamental component and3rd

harmonic distortions in the line current . The amplitude of the 3rdharmonic equals to half

of 2nd harmonic amplitude at the voltage compensator output . Another bandwidth

selection of the voltage loop is based on the total allowable 3rd harmonic distortions .

2.5.3 High power factor control circuit:

A block diagram of a boost power factor corrector is shown in Fig 2.10. The power

circuit of a boost power factor corrector is the same as that of a dc to dc boost

converter. A diode bridge is used to rectify the AC input voltage ahead of the inductor.

26


The capacitor generally used for this conversion is kept at the output of the converter and

even if any capacitor value is used in the input bridge, its value is very less and it

is only used to control any noise.

Fig.2.10.Basic configuration of a high power factor control circuit

A constant voltage is obtained at the boost converter output but the input voltage

by some programming forces the input current to be a half wave. The power obtained

by the output capacitor is in the form of a sine wave which has a frequency equal to twice

that of the line frequency and is never constant. This is illustrated in the Fig 2.11.

In the figure below, the voltage and current that goes into the power factor corrector are

indicated by the top waveform. The second waveform shows the power that flows

into and out of the capacitor in periods of its charging and discharging. When the

input voltage is higher than the voltage of the capacitor energy is stored in the

capacitor. When the input voltage drops below the capacitor voltage, to maintain the

output power flow the capacitor starts releasing energy. The third waveform in the figure

27


indicates the charging and discharging current. This current appears as if it is the

second harmonic component and is different in shape as compared to the input

current. Flow of energy reverses direction continuously and that in turn results in a

voltage ripple which is shown as the fourth waveform. Since the voltage ripple is

generated due to storage of reactive energy it is displaced by 90 degrees relative to

the current waveform above it. Ripple current of high frequencies are generated due

to the switching of the boost converter. The rating of the output capacitor should be

such that it can handle the second harmonic ripple current as well as the high frequency

ripple current.

Fig.2.11.Pre-regulator waveforms

28


The active power factor corrector is required to control both the input current and

the output voltage. The rectified line voltage programs the current loop in order to make

the converter input appear as resistive. Average amplitude of the current that is used as a

programming signal is changed to achieve control over the output voltage. The rectified

line voltage is multiplied with the output of voltage error amplifier by an analog

multiplier. This produces the current programming signal and provides it the shape

of the input voltage and average amplitude which helps control the output voltage.

Figure 2.10. is a block diagram which shows the basic control circuit arrangement

necessary for an active power factor corrector. The output of the multiplier is the current

programming signal and is called Imo for multiplier output current. A rectified line

voltage is shown as the multiplier input.

Figure 2.10 has a squarer as well as a divider along with the multiplier in the voltage

loop. The divider divides the output of the voltage error amplifier by the square of the

average input voltage. The resulting signal is then multiplied by the rectified input

voltage signal. The voltage loop gain is maintained at a constant value due to the

presence of the combination of these blocks. Otherwise the gain would have varied

with change in square of the average input voltage ( called feed forward voltage, Vff ).

This voltage only is squared by which the output of voltage error amplifier is later

divided.

For increasing the power factor to the maximum value possible, the rectified line voltage

and the current programming signal must match as closely as possible. The bandwidth of

the voltage loop should be maintained at a lower value than the input line frequency,

failing which huge distortion is produced in the input current. However on the other hand

for fast transient response of the output voltage the bandwidth needs to be made

as large as possible. In case of wide input voltage ranges, the bandwidth needs to be

as close as possible to line frequency. This is achieved by the action of the

squarer and divider circuits which help maintain the loop gain constant.

These circuits that maintain constant loop gain convert the output of voltage error

amplifier into a power control. Hence, now the power delivered to the load is controlled

29


by this output of the voltage error amplifier. Here we consider an example. Suppose that

the voltage error amplifier output is constant and then we double the input voltage. As

the programming signal depends on the input voltage it will also get doubled. Then it will

get divided by square of the feed forward voltage, which is equal to four times the input

now. This results in reducing the input current to half of its original value. Since the input

voltage is doubled, a factor of two is associated with it. Then it gets multiplied with half

of the input current. This results in no change in the input power and it remains same as

before. The output of the voltage error amplifier, then, controls the input power level of

the power factor corrector. This can be used to limit the maximum power which

the circuit can draw from the power line.

Provisions can be made for clamping the output of the voltage error amplifier at

some value which would correspond to some maximum power level. Then as long as the

input voltage is within its defined range, the active power factor corrector will not draw

more than that amount of power.

2.6 Proposed Design

For designing the proposed method of power factor correction, a power factor corrector

of output rated at 220 W is taken.

• Specifications:

Determination of the operating requirements for the active power factor corrector.

Pout (max) : 220W

Vin(range):22.5V-31.25 VAC(r.m.s)

Line Frequency:50Hz

Output Voltage:50Vdc

30


• Selection of switching frequency :

The switching frequency must be high enough to minimize the size of power circuit and

reduce distortion. On the other hand it should be less for greater

efficiency.Compromising between the two factors the value is selected as 2KHz.

• Inductor selection:

The inductor is selected from the value of maximum peak current which flows through it

when the input voltage has minimum value.

1. Maximum peak line current. P =Pin out(max) (2.24)

2×PinI =PK Vin(min)

(2.25)

220I =1.41× =10.34ampPK 30

(2.26)

2. Ripple current.

Ripple current is usually assumed to about 20% of the peak inductor current. In this case

it is arbitrarily selected to be 22% of it.

I =0.22×I =0.22×10.34=2.2748ampripple PK

(2.27)

3. Determination of the duty factor at Ipk where Vin(peak) is the peak of the

rectified line voltage at low line.

(V -V ) 50-42.420 in_PeakD= = =0.1514

V 500 (2.28)

4. Calculation of the inductance. fs is the switching frequency.

V ×D 42.426×0.1514inL= = =1.411mH

f ×I 2000×2.2748s ripple we take 1mH (2.29)

31


5. Selection of output capacitor. With hold-up time, the equation below was

used. Typical values for Co are 1uF to 2uF per watt. At is the hold-uptime in

seconds and V1 is the minimum output capacitor voltage.

2×P ×Vt 2×220×42.43msoutC = = =184.84mF0 2 2 2 2-V V 51 -500 1

(2.30)

6. Selection of current sensing resistor.This is required to sense the inductor

current. Sense resistor is the least expensive method and is suitable for low

power applications. Keeping the peak voltage across the resistor at a low value.

1.0V is a typical value for Vrs.

• Find ∆I 2.2748

I =I + =10.34+ =11.4747PK_max PK 2 2

(2.31)

• Calculating Current sensing resistor value.

V 1.00rsR = = =0.087;0.1ohms I 11.4747PK_max

(2.32)

7. Setting up independent peak current limit. Rpk1 and Rpk2 are the resistors in the

voltage divider. Choosing a peak current overload value, Ipk(ovld). A typical value for

Rpk1 is 10K.

V =I ×R =11.4747×0.1=1.14Vrs(ovld) PK(ovld) s

(2.33)

RPK 1.14×10K1R =V × = =1.52K

PK rs(ovld) V 7.52 ref (2.34)

8. Multiplier setup:

The operation of the multiplier is given by the following equation. Imo is

the multiplier output current, Km=1 , lac is the multiplier input current, Vff

is the feedforward voltage and Vvea is the output of the voltage error amplifier.

(V )vea-1I =K ×I ×mo m ac 2Vff

(2.35)

• Feedforward voltage divider. Changing Vin from RMS voltage to average

voltage of the rectified input voltage. At Vin (min) the voltage at Vff should

32


be 1.414 volts and the voltage at Vffc, the other divider node, should be about

7.5 volts. The average value of Vin is given by the following equation where

Vin(min) is the RMS value of the AC input voltage:

V =V ×0.9in(av) in(min)

(2.36)

The following two equations are used to find the values for the Vff divider

string. A value of 1Megohm is usually chosen for the divider input

impedance. The two equations must be solved together to get the resistor

values

Rff3V =1.414V=V ×ff in(av) R +R +Rff1 ff2 ff3

(2.37)

R +Rff2 ff3V =7.5V=V ×node in(av) R +R +Rff1 ff2 ff3

(2.38)

R =910Kff1

R =91Kff2

R =20Kff3

• Rvac selection. Maximum peak line voltage is found out.

V = 2×V = 2×31.25=44.19VPKmax inmax

(2.39)

Dividing by 600 microamps, the maximum multiplier input current.

VPKmaxR = =73.65Kvac 600E-6

, choosing 100K (2.40)

• Rb1 selection. This is the bias resistor. Treating this as a voltage divider

with Vref and Rvac and then solving for Rb1. The equation becomes:

R =0.25×R =0.25×100K=25Kb1 vac

(2.41)

Rset selection. Imo cannot be greater than twice the current through Rset.

Finding the multiplier input current, lac, with Vin(min). Then calculating the

value for Rset based on the value of lac just calculated.

V 44.19in(pk)I = = =0.4419mAmpacmin R 100Kvac

(2.42)

3.75 3.75×1000R = = =4.243K

set 2×I 2×0.4419acmin (2.43)

33


• Rmo selection. The voltage across Rmo must be equal to the voltage across

Rs at the peak current limit at low line input voltage

V ×1.12 1.14×1.12×1000rs(ovld)R = = =1.45K

mo 2×I 2×0.4419acmin (2.44)

9. Oscillator frequency:

Calculate Ct to give the desired switching frequency.

1.25 1.25

C = = =147.6uFt R ×f 4.234×2000set

(2.45)

10. Current error amplifier compensation:

• Amplifier gain at the switching frequency. Calculate the voltage across the

sense resistor due to the inductor current down slope and then divide by

the switching frequency.

50×0.1V = =2.5

rs 0.001×2000 (2.46)

This voltage must equal the peak to peak amplitude of Vs, the voltage

on the timing capacitor (5.2 volts). The gain of the error amplifier is

therefore given by:

V 5.2sG = = =2.08ca V 2.5rs

(2.47)

• Feedback resistors. Setting Rci equal to Rmo.

R =Rci mo

R =G ×R =0.832×1.45K=0.9568K;1Kcz ca ci

(2.48)

• Current loop crossover frequency.

V ×R ×R 50×0.1×1Kout s czf = = =105.59Hzci V ×2πL×R 5.2×2×3.14×0.001×1.45Ks ci

(2.49)

• Ccz selection. Choose a 45 degree phase margin. Setting the zero at the

loop crossover frequency.

34


1 1C = = =1.508uF

cz 2π×f ×R 2×3.14×105.59×1ci cz (2.50)

• Ccp selection. The pole must be above fs/2.

1 1C = = =79.57pF

cp 2π×f ×R 2π×2000×1s cz (2.51)

11. Voltage error amplifier compensation:

• Output ripple voltage. The output ripple is given by the following equation

where fr is the second harmonic ripple frequency.

P 220×1000outV = = =0.0947Vaco(pk) 2πf ×C ×V 2π×2000×184.84r o o

(2.52)

• Amplifier output ripple voltage and gain. Vo(pk) must be reduced to

the ripple voltage allowed at the output of the voltage error amplifier.

This sets the gain of the voltage error amplifier at the second harmonic

frequency.

The equation is:

∆V ×%Ripple 4×0.015vaoG = = =0.6335va V 0.0947o(pk)

(2.53)

• Feedback network values. Find the component values to set the gain of the

voltage error amplifier. The value of Rvi is reasonably arbitrary.

Choosing Rvi=100K

1 1

C = = =1.25pFvf 2π×f ×R ×G 2π×2000×100K×0.6335r vi va

(2.54)

• Setting DC output voltage.

R ×V 100K×7.5vi refR = = =17.64Kvd V -V 50-7.5o ref

(2.55)

• Finding pole frequency.

fvi = unity gain frequency of voltage loop.

35


P2 2outf = =(34.726) Hzvi 2∆V ×V ×R ×C ×C ×(2π)vao o vi o vf

f =34.726Hzvi

(2.56)

Finding Rvf

1 1R = = =366K

vf 2π×f ×C 2π×34.726×1.25pFvi vf

(2.57)

12. Feed forward voltage divider capacitors:

These capacitors determine the contribution of Vff to the third harmonic distortion

on the AC input current. Determine the amount of attenuation needed. The second

harmonic content of the rectified line voltage is 66.2%. %THD is the allowed percentage

of harmonic distortion budgeted to this input from step 10 above.

%THD 1.5G = = =0.0227

ff 66.2% 66.2 (2.58)

Using two equal cascaded poles. Find the pole frequencies. fr is the second harmonic

ripple frequency.

f = G f = 0.0227×100=15.06Hzffp r (2.59)

13. Select Cff1 and Cff2.

1 1C = = =0.259uF

ff1 2π×f ×R 2π×6.73×91Kp ff2 (2.60)

1 1C = = =1.18uF

ff2 2π×f ×R 2π×6.73×20Kp ff3 (2.61)

Inductor design for the boost converter:

∆II =I + =11.4747ampm pk 2

36


2 -3E=0.5×L×I =65.83×10 joulesm

2E 4A =A ×A = =3.65mmp w c K ×K ×J×Bw c m

(2.62)

2[B =0.2ferrite,J=3A/mm ,K =0.6,K =1]m w c

• Proper choice of core E 65/32/13

2 2 4A =2.66cm ,A =5.37cm ,A =14.284cmc w p

• No of turns

L×ImN= =21.56 22turnsA ×Bc m

≃

• Wire gauge

I 10.34pk 2A= = =3.446mmJ 3

For this SWG 9 is a proper choice.

• Cross Checking

The inequality A K >aNw w

has to be satisfied

A K =322w w

and AN=75.81

Hence the inequality is satisfied which means windings will fit into the available

window area.

37


3

Brushless DC Motor

A brushless dc motor is a rotating self-synchronous machine with a permanent

magnet rotor and with known rotor shaft positions for electronic commutation. A BLDC

motor meets this definition whether the drive electronics are integral with the motor or

separate from it.

Brushless DC motors compete with many other types of motors in the world of

motion control. However, for fractional-horsepower applications, brush and brushless DC

motors are often the main alternatives. Brushless motors are generally recognized as

being superior for a number of reasons including the elimination of maintenance,

increased efficiency, and reduced size and noise. In the past, for systems above a few 10's

of watts, brush motor systems have had a price advantage. This has changed. Advances in

magnet and electronic technologies, together with tooling investments by brushless motor

manufacturers, are reducing the costs of designing in brushless DC motors. In fact, many

high-volume applications such as automotive and appliances are moving to brushless.

Brushless DC motors (or BLDC motors) are generally recognized as being superior over

their brush counterparts. Their rich set of features includes high efficiency, greatly

reduced maintenance, high reliability, and elimination of brush debris, low acoustical and

electrical noise, small size, and a large speed range. Still, designers are reluctant to

specify brushless motors because of concerns about cost. However, the cost of using

brushless DC (BLDC) motors has been falling over the past few years so that in many

applications, they can compete on cost alone. This is due to three major factors: advances

in magnet technology, improvements in motor control electronics, and capital

investments by BLDC manufacturers. As a result, brushless DC motors are being used in

a wide range of cost sensitive applications including automotive, instrumentation,

38


printers, plotters, computer peripherals, document sorters, compressor and sewing

machines.

3.1 Comparison with brushed-DC motors:

BLDC motors offer several advantages over brushed DC-motors, including higher

efficiency and reliability, reduced noise, longer lifetime (no brush erosion), elimination of

ionizing sparks from the commutator, and overall reduction of electromagnetic

interference (EMI .) The maximum power that can be applied to a BLDC motor is

exceptionally high, limited almost exclusively by heat, which can damage the magnets.

BLDC's main disadvantage is higher cost, which arises from two issues. First, BLDC

motors require complex electronic speed controls to run. Brushed DC-motors can be

regulated by a comparatively trivial variable resistor (potentiometer or rheostat), which is

inefficient but also satisfactory for cost-sensitive applications.

BLDC motors are considered more efficient than brushed DC-motors. This means for the

same input power, a BLDC motor will convert more electrical power into mechanical

power than a brushed motor, mostly due to absence of friction of brushes. The enhanced

efficiency is greatest in the no-load and low-load region of the motor's performance

curve. Under high mechanical loads, BLDC motors and high-quality brushed motors are

comparable in efficiency.

BLDC motors are similar to brush PM motors. However, electronic commutation

eliminates the brushes and the mechanical commutator. This leads to many advantages.

Following table lists the advantages of BLDC motors compared to a Brushed DC motor:-

Features

BLDC Motors Brushed Dc Motor

Commutation

Electronic commutation based on Hall

position sensors.

Brushed commutation.

Maintenance

Less required due to absence of

brushes.

Periodic maintenance is

required.

Life Longer. Shorter.

Speed/Torque Flat – Enables operations at all speeds. Moderately flat – At higher.

39


Characteristics with rated load. speeds, brush friction

increases, thus reducing

useful torque.

Efficiency High – No voltage drop across

brushes.

Moderate.

Output

Power/Frame

size

High – Reduced size due to superior

thermal characteristics. Because

BLDC has windings on the stator,

which is connected to the case, the

heat dissipation is better.

Moderately low – The heat

produced by the armature is

dissipated in the air gap,

thus increasing the

temperature in the air gap

and limiting specs on the

output power/frame size.

Rotor Inertia Low – as it has permanent magnet on

the rotor. This improves dynamic

response.

Higher rotor inertia limits

dynamic characteristics.

Speed Range Higher – No mechanical limitations

imposed by brushes/ commutators.

Lower – Mechanical

limitations by brushes.

Electric Noise

generation

Low. Arcs in the brushes will

generate noise causing EMI

in the equipment nearby.

Cost of

building

Higher – Since it has permanent

magnets, building costs are higher.

Low.

Control

Complex and expensive. Simple and inexpensive.

Control

Requirement

A controller is always required to keep

the motor running. The same controller

can be used for variable speed control.

No controller is required for

fixed speed, a controller is

required only if variable

speed is desired.

40


3.2 Brushless motor vs Normal AC Induction Motor

Features BLDC Motors AC Induction motors

Speed/Torque

Characteristics

Flat – Enables operations at all

speeds with rated load.

Nonlinear – Lower torque at

lower speeds.

Output

Power/Frame size

High – Since it has permanent

magnet on the rotor, smaller

sizes can be achieved for a

given output power.

Moderate – Since both stator

and rotor have windings, the

output power to size is lower

than BLDC.

Rotor Inertia Low – Better dynamic

characteristics.

High – poor dynamic

characteristics.

Starting current Rated – No special starter

circuit required.

Approximately up to seven

times of rated – Starter circuit

rating should be carefully

selected. Normally uses a Star –

Delta starter.

Control

Requirement

A controller is always required

to keep the motor running. The

same controller can be used for

variable speed control.

No controller is required for

fixed speed, a controller is

required only if variable speed is

desired.

Slip No slip is experienced between

stator and rotor frequencies.

The rotor runs at a lower

frequency than stator by slip

frequency and slip increases

with load on the motor.

41


3.3 Components of a Brushless DC Motor:

(Refer Fig 3.1)

(1) Brushless motor - stator - stator windings - PM rotor or salient pole soft iron rotor (2) Controller - switches - driver (3) Shaft Position Sensor - sensor units - sensor armature

(4) Regulator Fig.3.1.Components of BLDC Motor

3.4 Construction of typical BLDC motor

3.4.1 STATOR:

Traditionally, the stator resembles that of an induction motor; however, the windings are

distributed in a different manner. There are two types of stator windings variants:

trapezoidal and sinusoidal motors. This differentiation is made on the basis of the

interconnection of coils in the stator windings to give the different types of back

Electromotive Force (EMF). As their names indicate, the trapezoidal motor gives a back

EMF in trapezoidal fashion and the sinusoidal motor’s back EMF is sinusoidal, as shown

in Figure 3.2 and Figure 3.3. In addition to the back EMF, the phase current also has

trapezoidal and sinusoidal variations in the respective types of motor. This makes the

torque output by a sinusoidal motor smoother than that of a trapezoidal motor. However,

this comes with an extra cost, as the sinusoidal motors take extra winding

interconnections because of the coils distribution on the stator periphery, thereby

increasing the copper intake by the stator windings

42


Fig 3.2: Trapezoidal Back E.M.F.

Fig 3.3: Sinusoidal back E.M.F.

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3.4.2 ROTOR:

The rotor is made of permanent magnet and can vary from two to eight pole pairs

with alternate North (N) and South (S) poles. Based on the required magnetic field

density in the rotor, the proper magnetic material is chosen to make the rotor. Ferrite

magnets are traditionally used to make permanent magnets. As the technology advances,

rare earth alloy magnets are gaining popularity. The ferrite magnets are less expe

but they have the disadvantage of low flux density for a given volume. In contrast, the

alloy material has high magnetic density per volume and enables the rotor to compress

further for the same torque. Also, these alloy magnets improve the size

and give higher torque for the same size motor using ferrite magnets. Neodymium (Nd),

Samarium Cobalt (SmCo) and the alloy of Neodymium, Iron and Boron (NdFeB) are

some examples of rare earth alloy magnets. Continuous research is going on to im

the flux density to compress the rotor further. Figure 1.4 shows cross sections of different

arrangements of magnets in a rotor.

Circular Core with magnets on

the periphery

3.4.3 Hall Sensors:

Unlike a brushed DC motor, the commutation of a BLDC motor is controlled

electronically. To rotate the BLDC motor, the stator windings should be energized in a

sequence. It is important to know the rotor position in order to understand which winding





rare earth alloy magnets are gaining popularity. The ferrite magnets are less expe



further for the same torque. Also, these alloy magnets improve the size



some examples of rare earth alloy magnets. Continuous research is going on to im


arrangements of magnets in a rotor.

Circular Core with magnets on Circular core with rectangular

magnets embedded in the

rotor

Circular core with rectangular

magnets

rotor core

Fig.3.4. Rotor magnet cross section

:




43

M.E.E Thesis





rare earth alloy magnets are gaining popularity. The ferrite magnets are less expensive



further for the same torque. Also, these alloy magnets improve the size-to-weight ratio



some examples of rare earth alloy magnets. Continuous research is going on to improve


Circular core with rectangular

magnets inserted into the

rotor core




Jadavpur University

will be energized following the energizing sequence. Rotor position is sensed using Hall

Effect sensors embedded into or outside the stator. Most BLDC motors have three Hall

sensors embedded into the stator on the non

magnetic poles pass near the Hall sensors, they give a high or low signal, indicating the N

or S pole is passing near the sensors. Based on the combination of these three Hall sensor

signals, the exact sequence of commutation can be determined. Figure 1.5

transverse section of a BLDC motor with a rotor that has alternate N and S permanent

magnets.

Fig.3

Hall sensors are embedded into the stationary part of the motor. Embedding the

Hall sensors into the stator is a complex process because any misalignment in these Hall

sensors, with respect to the rotor magnets, will generate an error in determination of t

rotor position. To simplify the process of mounting the Hall sensors onto the stator, some

motors may have the Hall sensor magnets on the rotor, in addition to the main rotor

magnets. These are a scaled down replica version of the rotor. Therefore, when

rotor rotates, the Hall sensor magnets give the same effect as the main magnets. The Hall

sensors are normally mounted on a PC board and fixed to the enclosure cap on the non

driving end. This enables users to adjust the complete assembly of Hall

with the rotor magnets, in order to achieve the best performance. Based on the physical

position of the Hall sensors, there are two versions of output. The Hall sensors may be at

electrical angle of 60° or 120° phase shift to each other.

e energized following the energizing sequence. Rotor position is sensed using Hall


sensors embedded into the stator on the non-driving end of the motor. Whenever the rotor



signals, the exact sequence of commutation can be determined. Figure 1.5


Fig.3.5.BLDC Motor transverse section



sensors, with respect to the rotor magnets, will generate an error in determination of t



magnets. These are a scaled down replica version of the rotor. Therefore, when


sensors are normally mounted on a PC board and fixed to the enclosure cap on the non

driving end. This enables users to adjust the complete assembly of Hall



electrical angle of 60° or 120° phase shift to each other. Based on this, the motor

44

M.E.E Thesis

e energized following the energizing sequence. Rotor position is sensed using Hall


driving end of the motor. Whenever the rotor



signals, the exact sequence of commutation can be determined. Figure 1.5 shows a




sensors, with respect to the rotor magnets, will generate an error in determination of the



magnets. These are a scaled down replica version of the rotor. Therefore, whenever the


sensors are normally mounted on a PC board and fixed to the enclosure cap on the non-

driving end. This enables users to adjust the complete assembly of Hall sensors, to align



Based on this, the motor

45


manufacturer defines the commutation sequence, which should be followed when

controlling the motor.

3.5 Working Principle Of BLDC Motor:

In conventional DC motors, the armature is rotor, and the field magnets are placed

in the stator. The construction of modern brushless DC (BLDC) motors is very similar to

the AC motor, known as the permanent magnet synchronous motor. The armature

windings are part of the stator, and the rotor is composed of one or more permanent

magnets. The windings in a BLDC motor are similar to those in a polyphase AC motor

and the most orthodox and efficient motor has a set of three-phase windings and is

operated in bipolar excitation. BLDC motors are different from AC synchronous motors

in that the former incorporates some means to detect the rotor position (or magnetic

poles) to produce signals to control the electronic switches.

Figure (3.6) shows a simple three-phase unipolar-operated motor that uses optical

sensors (phototransistors) as position detectors. Three phototransistors PT1, PT2, and

PT3 are placed on the end-plate at 120° intervals and exposed to light in sequence

through a revolving shutter coupled to the motor shaft.

Fig.3.6.Three Phase Unipolar-Driven BLDC motor

46


As shown in figure (3.6) the south pole of the rotor now faces the salient pole

P2 of the stator, and the phototransistor PT1 detects the light and turns transistor Tr1 on.

In this state, the south pole which is created at the salient pole P1 by the electrical current

flowing through the winding W1 is attracting towards the north pole of the rotor to move

it in the direction of the arrow (ccw).When the north pole comes to the position to face

the salient pole P1, the shutter, which is coupled to the rotor shaft, will shade PT1 and

PT2 will be exposed to the light and a current will flow through the transistor Tr2. When

a current flows through the winding W2, and creates a south pole on salient pole P2, then

the north pole in the rotor will revolve in the direction of the arrow and face the salient

pole P2.At this moment, the shutter shades PT2 and the phototransistor PT3 is exposed to

light. These actions steer the current from the winding 2 to W3. Thus salient pole P2 is

de-energized, while the salient pole P3 is energized and creates the south pole. Hence the

north pole on the rotor further travels from P2 to P3 without stopping. By repeating such

a switching action in a particular sequence the permanent magnet rotor revolves

continuously.

3.6 Torque/Speed Characteristics:

A motor which uses permanent magnets to supply the field flux can be represented by the

simple equivalent circuit of figure (3.7). This is a series circuit of the armature resistance,

Ra and the back e.m.f, E.

Fig.3.7 single phase equivalent circuit of BLDC motor

47


If the voltage drop across the brushes is ignored, the equation for the voltage is

V= Ra.Ia+KEΩ (3.1)

The armature current Ia is

Ia = (V- KEΩ)/ Ra (3.2)

Therefore, from equation (3.1) and (3.2) the torque T is

T= KTIa = KT/ Ra(V- KEΩ) (3.3)

Fig.3.8.Torque-Speed characteristics of BLDC motor

Figure (3.8) shows the relation between T (torque) and Ω (rotational speed) at two

voltages. The torque decreases linearly as the speed increases. The slope of this function

is a constant KTKE/Ra and is independent of the terminal voltage and the speed. Such

characteristics make the speed or position control of a dc motor easy. However, only dc

and brushless dc motors have this feature – ac and stepping motors do not.

Figure (3.9) shows an example of torque/speed characteristics. There are two

torque parameters used to define a BLDC motor, peak torque (TP) and rated torque (TR).

During continuous operations, the motor can be loaded up to the rated torque. In a BLDC

48


motor, the torque remains constant for a speed range up to the rated speed. The motor can

be run up to the maximum speed, which can be up to 150% of the rated speed, but the

torque starts dropping. Applications that have frequent starts and stops and frequent

reversals of rotation with load on the motor, demand more torque than the rated torque.

This requirement comes for a brief period, especially when the motor starts from a

standstill and during acceleration. During this period, extra torque is required to

overcome the inertia of the load and the rotor itself. The motor can deliver a higher

torque, maximum up to peak torque, as long as it follows the speed torque curve.

Peak Torque TP

Torque

Intermittent

torque zone

Rated Torque

TR Continuous torque zone

Rated Speed

Fig 3.9.Torque – Speed characteristics

3.7 TYPICAL BLDC MOTOR APPLICATIONS

BLDC motors find applications in every segment of the market. Automotive,

appliance, industrial controls, automation, aviation and so on, have applications for

BLDC motors. Out of these, we can categorize the type of BLDC motor control into three

major types:

• Constant load

• Varying loads

49


• Positioning applications

Applications with Constant Loads

These are the types of applications where a variable speed is more important than

keeping the accuracy of the speed at a set speed. In addition, the acceleration and

deceleration rates are not dynamically changing. In these types of applications, the load is

directly coupled to the motor shaft. For example, fans, pumps and blowers come under

these types of applications. These applications demand low-cost controllers, mostly

operating in open-loop.

Applications with Varying Loads

These are the types of applications where the load on the motor varies over a

speed range. These applications may demand a high-speed control accuracy and good

dynamic responses. In home appliances, washers, dryers and compressors are good

examples. In automotive, fuel pump control, electronic steering control, engine control

and electric vehicle control are good examples of these. In aerospace, there are a number

of applications, like centrifuges, pumps, robotic arm controls, gyroscope controls and so

on. These applications may use speed feedback devices and may run in semi-closed loop

or in total closed loop. These applications use advanced control algorithms, thus

complicating the controller. Also, this increases the price of the complete system.

Positioning Applications

Most of the industrial and automation types of application come under this

category. The applications in this category have some kind of power transmission, which

could be mechanical gears or timer belts, or a simple belt driven system. In these

applications, the dynamic response of speed and torque are important. Also, these

applications may have frequent reversal of rotation direction. A typical cycle will have an

accelerating phase, a constant speed phase and a deceleration and positioning phase. The

load on the motor may vary during all of these phases, causing the controller to be

complex. These systems mostly operate in closed loop. There could be three control

loops functioning simultaneously: Torque Control Loop, Speed Control Loop and

Position Control Loop. Optical encoder or synchronous resolvers are used for measuring

50


the actual speed of the motor. In some cases, the same sensors are used to get relative

position information. Otherwise, separate position sensors may be used to get absolute

positions. Computer Numeric Controlled (CNC) machines are a good example of this.

Process controls, machinery controls and conveyer controls have plenty of applications in

this category.

51


4

Harmonic analysis of Back E.M.F and Phase Current Waveform of Trapezoidal BLDC Motor

Cost minimization and performance improvement of the electrical machine drives is

more attractive for low cost applications where permanent magnet brushless dc (BLDC)

motors are widely used, now-a-days . Harmonics content present in the phase current and

back-EMF of BLDC motor are analyzed both for 120° and 180° mode of switching

schemes. A detailed analysis shown in this thesis reveals , efficiency is improved by

eliminating those current harmonics , both for 120° and 180° mode of switching

schemes and efficiency improvement is higher in case of 120° conduction mode of

inverter.

Brushless DC motors have been widely used in a variety of applications in industrial

automation and consumer appliances because of their high power density, compactness,

high efficiency, low maintenance and ease of control. Nowadays, many studies have been

focused on how to improve operating efficiency of BLDC motor without performance

degradation. The energy saving of variable speed drives such as BLDC motor drives is

accomplished by two approaches. One is the topological approach and the other is the

control approach. From a topology point of view, using high grade magnetic material and

design changes are required. In control approach eliminating harmonics content and

mechanical sensors are required for the inverter circuit to reduce conduction losses and

mechanical hazards. The back-EMF and phase current of BLDC motor both for 120° and

180° Conduction mode contains harmonics. It is quite obvious by eliminating these

current harmonics, torque ripple along with stator resistive losses minimized, thus

claiming maximum efficiency in the given speed range. Critical and central to achieving

such a performance is a good controller implementation based on back-EMF wave and

phase current characteristics, which is able to eliminate harmonics content present in the

52


phase current, optimizing machine efficiency. Accordingly, this chapter is centered on the

derivation of analytical expression for back-EMF and phase current with harmonic

contents for 120° and 180° mode of conduction.

Later in chapter 5 it is analyzed, which conduction mode is better for efficiency

improvement.

4.1 BACK EMF OF NON SINUSOIDAL BLDC MACHINES: Irrespective of the type of the winding (distributed, concentric, fractional or alternate

teeth wound) and the type of the rotor( surface mount or interior type),generalized phase

to neutral back EMF expression for three phase non sinusoidal BLDC machines is

described by equation (4.1)

E =a

4Eπα

( ) ( ) ( ) ( ) ( ) ( )1 1[sin α sin ωt + sin 3α sin 3ωt + sin 5α sin 5ωt +

2 23 5

( ) ( )1+ sin 7α sin 7ωt +.......nth term]

27

(4.1)

Fig 4.1 Back E.M.F waveform of trapezoidal BLDC Motor

53


4.2 Phase Current of non sinusoidal BLDC machine:

In the fig 4.1 at an angle α when switch is ‘on’ current flows through corresponding

phase winding. Ideally at switching time current should build instantaneously. It is shown

below in fig (4.2)

Fig.4.2. Ideal phase current waveform.

4.3 Different Conduction Mode of Trapezoidal BLDC Motor:

There are two types of current conduction mode in phase winding of BLDC motor.

• 120° conduction mode: In 120° conduction mode each phase winding conducts for 120° out of a half

period.

Fig 4.2 shows ideal 120° conduction mode of phase current when α=30°.

• 180° conduction mode: In 180° conduction mode each phase winding conducts for 180° out of a half

period.

Fig 4.3 shows ideal 180° conduction mode of phase current.

54


Fig.4.3. Ideal Phase Current waveform for 180°(α=60°) conduction mode

4.4 Harmonic analysis of Ideal current waveform for different conduction mode:

• 120° conduction mode: The expression for ideal current in 120°conduction mode is given by equation (4.2)

I =a

4I

π

( ) ( ) ( ) ( ) ( ) ( )1 1[cos α sin ωt + cos 3α sin 3ωt + cos 5α sin 5ωt +

3 5

( ) ( )1+ cos 7α sin 7ωt +……….nth term]

7

(4.2)

Putting α=30° in above equation (4.2) we get a harmonics spectrum given below in fig 4.4

55


Fig 4.4 Harmonic contents present in 120° ideal current conduction mode

• 180° conduction mode: The expression for ideal current in 180°conduction mode is given by equation (4.3)

( ) ( )3I 1 1 1( )[sin ωt + sin 5ωt + sin(7ωt)+ sin(11ωt)…+nth term]π 7 11

I =a 5

(4.3)

Fig.4.5.Harmonic contents present in 180° ideal current conduction mode

56


4.5 Rising Angle:

An angle made by the phase current with horizontal line after switch is ‘ON’

called current rising angle. Fig 4.6 illustrate this, given below.

Fig.4.6 Ideal and Actual Current Switching

Ideally when switch is on, current through the winding should reach to its maximum

value instantaneously but due to circuit inductance it requires specific time depends on

the circuit time constant.

4.6 Harmonic analysis of phase current waveform considering rising

delay angle:

In any conduction mode of trapezoidal BLDC motor, when inverter switch is on current

should build instantaneously to the rated value in corresponding phase winding. But due

to the winding inductance current can’t build or decay instantaneously after switching. It

takes time to build up.

• 120° conduction mode:

Fig 4.7 is the phase current waveform for 120° considering rising delay.

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Fig.4.7 Phase Current waveform for 120° conduction mode (

( 2 1α α− )rising delay.

For this current waveform in 120° conduction mode the current equation (4.4) is given below.

( ) ((4I 1[ sin

π α -α2 1I =a

The order of the harmonics (

in fig.4.8.

Current waveform for 120° conduction mode (

For this current waveform in 120° conduction mode the current equation (4.4) is given

) ( )) ( ) ( ) ((4I 1α -sin α sin ωt + sin 3α -sin 3α2 1 2 123

( ) ( )( ) ( )1+ sin 5α -sin 5α sin 5ωt +……+n term]2 125

The order of the harmonics (1 230 and 45α α= ° = ° ) for 120° conduction mode are given

57

M.E.E Thesis

Current waveform for 120° conduction mode ( 301

α °= ),considering

For this current waveform in 120° conduction mode the current equation (4.4) is given

)) ( )-sin 3α sin 3ωt +2 1 2 1

thωt +……+n term] (4.4)

) for 120° conduction mode are given

58


Fig.4.8. Harmonic contents present in 120° conduction mode.

The changes in the harmonic contents with the change in rising delay angle for this

conduction mode is given below.(fig.4.9)

Fig.4.9. Changes in the harmonic contents ,Y-axis with increase in angle (2 1α α− ) X-

axis, for 120° conduction mode.

0

0.2

0.4

0.6

0.8

1

1.2

1 2 3 4 5 6 7 8 9 10 11 order of harmonics

Harmonics content

0

0.05

0.1

0.15

0.2

0.25

3rd

5th

11th

3rd

5th

7 th

9th

5° 10° 15°

59


• 180° conduction mode:

Fig 4.10 is the phase current waveform for 180° considering rising delay. And for

180° mode of conduction current expression is given by equation (4.5)

( ) ( ) ( )( ) ( ) ( ) ( ) ( )( )2I 1( )[ sin α +sin α +α -sin α sin ωt + sin 3α +sin 3α +3α -sin 3α1 1 2 2 1I =

a 2 1 22πα 31

( ) ( ) ( ) ( )( ) ( )1 thsin 3ωt + sin 5α +sin 5α +5α -sin 5α sin 5ωt +…+n term]1 1 2 225

(4.5)

Fig.4.10.Phase current waveform for 180° conduction mode ( 602

α = ° ) taking 1

α

rising delay.

The order of the harmonics and its value (1 215 and 60 )α α= ° = ° corresponding to 180°

conduction mode is given below. (fig.4.11)

60


Fig.4.11. Harmonic contents present in 180° conduction mode.

The changes in the harmonic contents with the change in rising delay angle for this

conduction mode is given below.(fig.4.12)

Fig.4.12. Change in the harmonic contents ,Y-axis with increase in angle (1)X axisα − ,

for 180° conduction mode.

0

0.2

0.4

0.6

0.8

1

1.2

1 2 3 4 5 6 7 8 9 10 11Order of harmonics

Harmonics content

0

0.05

0.1

0.15

0.2

0.25

5th

7th

11th

5° 10° 15°

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5.1 Loss Calculation

From previous chapter

phase current in both the switching schemes the stator resistive losses and core losses

can be minimized and efficiency can be

5.1.1 Stator Resistive Losses

The r.m.s values of the phase current taking ideal waveform for 120° and 180°

conduction mode are respectively as following,

2I =Ir.m.s 3

For 120°

II =r.m.s 2

For 180°mode of conduction

For 120° mode of conduction two inverter switch will operate at a time. Thus tak

normal resistive circuit accordi

Efficiency and Loss Comparison

.1 Loss Calculation:

previous chapter it is clear that by eliminating the harmonics present in the


can be minimized and efficiency can be effectively improved.

Stator Resistive Losses:

lues of the phase current taking ideal waveform for 120° and 180°

are respectively as following,

For 120°mode of Conduction.

For 180°mode of conduction

mode of conduction two inverter switch will operate at a time. Thus tak

normal resistive circuit according to fig (5.1) when 1Q and 4

Q operate,

Fig.5.1.Inverter with resistive load.

61

M.E.E Thesis

5

Efficiency and Loss Comparison

it is clear that by eliminating the harmonics present in the


lues of the phase current taking ideal waveform for 120° and 180°

(5.1)

For 180°mode of conduction (5.2)

mode of conduction two inverter switch will operate at a time. Thus taking a

4operate,

62


Thus copper loss at any switching instant is given by equation (5.3)

42 2 2I ×2R= I R=1.33I Rr.m.s 3

(5.3)

For 180° mode of Conduction three inverter switch will operate at a time. Thus taking

a normal resistive circuit again according to fig.4.1 when switch 1Q ,3

Q and 4

Q will

operate,

Thus copper loss at any switching instant is given by equation (5.4)

32 2 2I ×3R= I R=1.5I Rr.m.s 2

(5.4)

Thus copper loss is higher in 180°conduction mode than 120° conduction when

current waveforms are ideal. But practically we can’t neglect coil inductance and

current takes time to build up.

Practically for any rise angle of current waveform of 120° Conduction mode have more harmonic contents. Let us take 15° rise angle and for 120°mode

Harmonic contents Percentage of fundamental(%)

3rd Harmonic 15.71

5th Harmonic 23.31

7th Harmonic 2.03

9th Harmonic 10.175

11thHarmonic 4.81

Similarly for 180° conduction mode

Harmonic contents Percentage of fundamental(%)

5th Harmonic 23.31

7th Harmonic 2.03

11th Harmonic 4.8

63


I =0.726997Ir.m.s °180

and °

I =0.73895Ir.m.s

120

Thus copper loss for 180° conduction mode is = 21.5855×I R and for 120° conduction

mode copper loss is = 21.0920×I R

From the above discussion it is clear that in 180° conduction mode copper loss is higher.

But the discussion above is at different torque, now we have to compare copper loss for same average torque.

Fig.5.2 Torque curve for ideal 120°(α=30°) conduction mode

2EIT =avg 3

is the average torque for 120°conduction mode (fig 5.2)

Fig.5.3.Torque curve for 180° conduction mode

64


EIT =avg 2

is the average torque for 180° conduction mode (fig 5.3)

Thus for same average torque, 2Copperloss 3I R180°= =1.52Copperloss 2I R120°

(5.5)

So, for same average torque copper loss is higher in 180° conduction mode than 120° conduction mode.

5.1.2 On State Conduction Loss Across Switch:

On state conduction loss across switches is same for 120° and 180° conduction mode.

Let, on state voltage drop=Vmin

On state current=Imax

For 120° conduction loss=2V Imin max

according to fig.4.1

For 180° conduction loss= ( ) ( )I Imax maxV I + V + V =2V Imin minmin max min max2 2

according to fig.5.1

Thus for both the conduction mode on state loss across switch is same.

5.1.3 Switching Loss:

• During Switch on time:

Fig.5.4 Voltage and Current waveform during switch ON

65


From fig.5.4. the phase current equation is

( )I -Imax minI t = t+I

minTon (5.6)

And the phase voltage equation is

( )V -Vmax minV t =- t+V

maxTon (5.7)

Now energy loss during ON time is

( ) ( )Ton

E = V t I t dton

0∫

Ton V -V I -Imax min max min= - t+V t+I dtmax minT Ton on0

∫

( )T V I -Ion V -V I -I V -Vmax max min2max min max min max min= - × t + t- I t+V I dtmin max minT T T Ton on on on0

∫

( ) V V +VVmax max minmin= I -I + T +I Tmax min on min on6 3 2

If we make, I =0min

and V =0min

just like an ideal condition, we get

I V Tmax max onE =on 6

(5.8)

66


• During switch off time:

Fig.5.5 Voltage and Current waveform during switch OFF

From fig.5.5. the phase current equation is

( )I -Imax minI t =- t+I

maxToff (5.9)

And the phase voltage equation is

( )V -Vmax minV t = t+V

minToff (5.10)

Now energy loss during ON time is

( ) ( )Toff

E = V t I t dtoff

0∫

I I +IImax max minmin=(V -V )( + )T +V T ( )

max min off min off6 3 2

If we make, I =0min

and V =0min

just like an ideal condition, we get

V I Tmax max offE =

off 6 (5.11)

Thus total energy loss within a period is

67


E=E +Eon off

V Imax max= (T +T )

on off6 (5.12)

Average Switching loss

V Imax maxW= (T +T )fon off switching6

(5.13)

As there are three switch operate at a time in 180° conduction mode switching loss is

more in 180° conduction mode than 120°. For one IRFZ44N, T =1µson

,T =1.5µsoff

The switching loss is (taking switching frequency=10KHz)=

50×10 -6 3(2.5×10 )10×10 =2.08watt6

.

5.1.4 Iron loss:

• Hysteresis Loss: This loss id due to the revarsal of magnetisation of the armature core.Every

portion of the stationary stator core passes under N and S pole

alternatively,thereby attaining S and N polarity respectively.The core

undergoes one complete cycle of magnetic revarsal after passing under one

pair of poles.

The loss depends upon the volume and grade of iron,maximum value of flux

density(Bmax

) and frequency of magnetic revarsal.Hysteresis loss is given

by Steinmentz formula.According to the formula,

1.6W =K B fh h max

watts

Where f=Fundamental frequency.

Kh

=Hysteresis constant.

Now taking harmonic components in account, we can write,

1.6 1.6 1.6 1.6W =K B fV+K VB f +K VB f +K VB f +.....h h max h 3max 3 h 5ma

nx 5 h 7max 7

th term

68


The percentage of hysteresis loss components for 120° phase current

conduction mode taking 15° rising angle delay is given in figure below (5.6).

Fig.5.6 Hysteresis loss components of total core loss for 120°

And similarly for 180° phase current current conduction mode percentage of

hysteresis loss components taking 15° riging angle delay is given below.Fig.5.7

Fig.5.7 Hysteresis loss components of total core loss for 180°

0

0.05

0.1

0.15

0.2

0.25

0.3

1 2 3 4 5 6 7 8 9 10 11

Y axis X 100%

X axis=Order of Harmonics

0

0.05

0.1

0.15

0.2

0.25

0.3

0.35

1 2 3 4 5 6 7 8 9 10 11

Y axis X 100%


69


• Eddy Current loss: When PM rotor rotates ,flux linkage changes in stator armature core.Thus

according to the laws of electromagnetic induction an e.m.f is induced in the

core body.This e.m.f though small, sets up large current in the body of the

core due to its small resistance.This current is known as eddy current.The

power loss due to the flow of this current is known as eddy current loss. Thus

the core is made of thin lamination to incrrease the current path resistance

thereby reducing eddy current loss.

It is found that eddy current loss We

is given by the following relation:

2 2W =K B fe e max

watt

V=Volume of the armature core, t=thickness of each lamination.

Now taking harmonic components in account, we can write,

2 2 2 2 2 2W =K B f +K B f +K B f +......e e max e 3max 3 e 5max 5

nth term

The percentage of eddy current loss components of total core loss for 120° phase current

conduction mode taking 15° riging angle delay is given in figure below 5.8

Fig.5.8 Eddy current loss components of total core loss for 120°

0

0.02

0.04

0.06

0.08

0.1

0.12

0.14

0.16

0.18

0.2

1 2 3 4 5 6 7 8 9 10 11

Y axis X 100%


70


And similarly for 180° phase current current conduction mode taking 15° rising angle

delay percentage of eddy current loss components is given below.Fig.5.9

Fig.5.9 Eddy current loss component of total core loss for 180°

5.2 Efficiency Comparison and Improvement through proposed technique:

Overall drive efficiency is dependent upon three categories

• UPFC converter efficiency.( )1

η

• Inverter efficiency.( )2

η

• Bldc motor efficiency.( )3

η

UPFC converter efficiency:

If we use narmal diode rectifier with capacitor at output then efficiency of the conversion

is not good along with line current harmonics which reduces power factor.In this

thesis (Chapter2) boost converter with PFC using UC3854 is used which increases the

supply converter efficiency with low line harmonics.

0

0.05

0.1

0.15

0.2

0.25

0.3

1 2 3 4 5 6 7 8 9 10 11


Y axis X 100%

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Inverter Efficiency:

Sinusoidal pulse width modulated inverter with selective harmonics elimination ensures

minimization of phase current harmonics which reduces copper loss and iron loss in

BLDC motor which in turn increases motor efficiency.

BLDC motor efficiency:

If we use high quality magnet for PM rotor and laminated silicon steel in stator core we

can minimise core losss.But this is design aspects.For a given motor only thing is left to

increase efficiency is to increase supply and inverter efficiency.

η =η ×η ×ηoverall 1 2 3

Fig.5.10 conventional system with large harmonic distortion

72


Fig 5.11 Active PFC circuit + PWM inverter

In BLDC motor the major part of losses is copper loss.In 180° phase current conduction

mode this copper loss, switching loss, core losses are more than 120° conduction mode.

Thus 120° phase current conduction mode with reduced harmonics is preferable for

BLDC motor efficiency improvement.

73


6

SIMULATION RESULTS

MATLAB software:

MATLAB is a software package for high performance numerical computation and visualization.It provides an interactive environment with hundreds of build-in-functions for technical computation,graphics and animation.Best of all, it also provides easy extensibility with its own high-level programming language.

In this thesis boost converter with power factor improvement is modelled in MATLAB simulink.

There are two stages of modelling:

• Power circuit modelling.

• Controller modelling for PFC.

Power circuit modelling is done similar as boost converter in practical (fig.5.1).And after controller is modelled according to chapter 3,described earlier.

Fig.6.1 Power circuit model

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6.1 Simulink M-File:

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6.2 PFC Controller Block:

After scaling of load voltage it is compared with a constant after desired load voltage the error is minimum,ideally zero.

Let this error voltage=A

Input Voltage gets fixed DC voltage after filter=C ,used for feedforward scaling.

Input voltage=B= 2 Sin(

Output of Divide block is=

current referance signal is compared with actual inductor current.This error signal helps to create pulse which in turn drive MOSFET so that inductor can track the actual referance current signal.

This will force the line current to follow the line voltage and PFC is achieved.

.2 PFC Controller Block:

Fig.6.2 controller model

After scaling of load voltage it is compared with a constant referance voltage such that after desired load voltage the error is minimum,ideally zero.

Let this error voltage=A


2 Sin(ωt)

Divide block is=A 2 Sin(ωt)

=K Sin(ωt)2C

and it current referance signal.This

current referance signal is compared with actual inductor current.This error signal helps to create pulse which in turn drive MOSFET so that inductor can track the actual

ance current signal.


75

M.E.E Thesis

referance voltage such that


and it current referance signal.This

current referance signal is compared with actual inductor current.This error signal helps to create pulse which in turn drive MOSFET so that inductor can track the actual


Jadavpur University

6.3 Simulink Result and Output Waveform:

• Output Waveforms:

Fig6

Fig.6.4 Input Voltage and Current Waveform at 2KHz frequency

.3 Simulink Result and Output Waveform:

Output Waveforms:

Fig6.3 Boost Converter Output Voltage.

.4 Input Voltage and Current Waveform at 2KHz frequency

76

M.E.E Thesis

.4 Input Voltage and Current Waveform at 2KHz frequency

77


Fig 6.5 Voltage PI controller output

Fig 6.6 Inductor Current at 2 KHz frequency

78


Fig 6.7. Pulse output from controller

When switching frequency is increased then current ripple will decrease as in this thesis it is observed in 100 Khz.

Fig 6.8 Input voltage and current waveform at 200KHz frequency

--Input Voltage ---Input Current

79


• Simulation Results for the Power Factor Corrector

Rating of the active power factor corrector

Maximum output power =220W

Operating Frequency = 50 Hz

Switching Frequency = 2 KHz

Range of input voltage =22.5-31 V AC (r.m.s)

Output voltage =50Volt

Load Resistance =11.35 ohms

Inductor value = 0.1mH

Output Capacitor = 2200 uF

Line Voltage (in V)

Converter input voltage [in V(r.m.s)]

Output Voltage (in V)

Power Factor

Total Harmonics Distortion (in %)

Efficiency (in %)

180

22.5

50.25

0.98425

17.96

75.96

190

23.75

50.4

0.9822

19.11

73.70

200

25

50.2

0.9812

19.66

74.62

210

26.25

50.23

0.9760

22.31

75.89

220

27.5

50.2

0.9756

22.46

74.81

230

28.75

50.1

0.9721

24.11

74.21

240

30

50.15

0.9689

25.51

82.34

250

31.25

50.1

0.9696

25.21

82.15

80


For switching frequency 100 KHz, result is given below.

Line Voltage (in V)

Converter input voltage (in V)

Output Voltage (in V)

Power Factor

Total Harmonics Distortion (in %)

Efficiency (in %)

180

22.5

50.25

0.9985

5.46

89.95

190

23.75

50.4

0.99825

5.92

93.757

200

25

50.2

0.99845

5.50

92.342

210

26.25

50.23

0.99866

5.17

91.295

220

27.5

50.2

0.99842

5.66

90.65

230

28.75

50.1

0.9990

4.21

90.27

240

30

50.15

0.9987

5.01

90.31

250

31.25

50.1

0.9986

5.22

86.70

81


6.4 Hardware Configuration:

82


6.5 Experimental Result:

For smooth operation of the inverter which is connected with UPFC boost converter, output voltage from boost converter should remain constant at different loading. This is verified by taking 100Ω variable resistance. It is seen that output voltage remains constant during variation of load. It is shown in output waveforms given below.

• Output Waveforms:

Fig.6.9. Boost Converter Output Voltage.

83


Fig.6.10. Input Voltage (Blue) and Current Waveform (yellow) at 12.92KHz frequency.

Fig.6.11.Input voltage (Blue) and current (yellow) in same frame at 12.92 KHz frequency.

Input Voltage

Input Current

84


Fig.6.12.Inductor Current waveform at 12.92 KHz frequency.

Fig.6.13.Input Current Waveform of UPFC Boost Converter.

85


Fig.6.14. Pulse Output from controller.

• Input voltage and current waveform is measured with a R-L load (R=100Ω variable and 25mH inductor) and the output waveform is given below.

Fig 6.15. Voltage (Blue) and Current Waveform (yellow) at 12.92 KHz with R-L load

86


Fig.6.16.Total UPFC Boost Converter Circuit using UC3854.

Fig 6.17. Controller Circuit of PFC

87


Fig.6.18.Power Circuit of Boost Converter

88


Fig.6.19 UPFC based BLDC drive set up

89


7 Conclusions

7.1 Conclusions:

An UC3854 based UPFC boost converter is designed and tested at the input side of

PMBLDC motor drive. A mathematical model of UPFC is designed and analyzed in

MATLAB and practical circuit from this is verified. By using active PFC at input side the

power quality and efficiency of the conversion is improved. It is also analyzed that

efficiency of the BLDC drive can be further improved if we use 120° phase current

conduction mode.

7.2 Scope for future work:

• More suitable design procedure can be implemented with soft switching scheme

at active PFC circuit to have very low THD in input current and higher conversion

efficiency. In this thesis it is concluded that 120° conduction mode is better for

efficiency improvement. But in 120° conduction mode torque ripple will increase

for a given speed range [1] [2] [3].

• So to have low torque ripple and improved efficiency both altogether hybride

switching is required. Thus the optimum conduction mode for both torque ripple

minimization and efficiency improvement is to be studied.

• The system can be implemented using DS1104 dspace module replacing the use

of UC3854.

90


References

[1] R. Carlson, M. Lajoie-Mazenc, and J.C.D.S. Fagundes,. “Analysis of torque ripple due to phase commutation in brushless DC machines,” IEEE Trans. Ind. Appl., vol. 28, no.3, pp. 632-638, May/June1992.

[2] J. Holtz and L. Springob, “Identification and compensation of torque ripple in high-precision permanent magnet motor drives,” IEEE Trans. Ind. Ecn., vol. 43, no.2, pp 309-320, April 1996.

[3] S.S.Bharatkar, R. Yanamshetti, D. Chatterjee, A.K. Ganguli, “Dual-mode switching technique for reduction of commutation torque ripple of brushless dc motor” Published in IET Electric Power Applications, 2011, Vol. 5, Iss. 1, pp. 193–202.

[4] R. Krishnan, “Electric Motor Drives: Modeling, Analysis, and Control,” Prentice Hall, New Jersey, 2001

[5] R Krishnan, “Permanent Magnet Synchronous and Brushless DC Motor Drives”,CRC Press, 2010.

[6] Parag Kshirsagar, R Krishnan,“Efficiency Improvement Evaluation of Non-Sinusoidal Back-EMF PMSM Machines Using Field Oriented Current Harmonic Injection Strategy”,IEEE Trans,On Power Electronics,Vol.12.No.3,November-2011.

[7] Tae-Hyung Kim, Mehrdad Ehsani,“ Sensorless Control of the BLDC Motors From Near-Zero to High Speeds” IEEE Trans. On Power Electronics, Vol. 19, No. 6, November 2004.

[8] T. H. Kim, B. K. Lee and M. Ehsani, “Sensorless control of the BLDC motors from near zero to high speed”, in Proc. IEEE Applied Power Electronics Conf. and Expo. , Vol. 1, pp. 306-312, 2003.

[9] C. L. Puttaswamy, Bhim Singh and B.P. Singh, “Investigations on dyanamic behavior of permanent magnet brushless DC motor drive,” Electric Power Comp. and Sys.,vol.23, no.6, pp. 689 - 701, Nov. 1995.

[10] Limits for Harmonic Current Emissions (Equipment input current less than 16 A per phase), International Standard IEC 61000-3-2, 2000.

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[11] B. Singh, B. N. Singh, A. Chandra, K. Al-Haddad, A. Pandey and D. P.Kothari, “A review of single-phase improved power quality AC-DCconverters,” IEEE Trans . Industrial Electron ., vol. 50, no. 5, pp. 962 –981, Oct. 2003.

[12] D. C. Hanselman, Brushless Permanent Magnet Motor Design , McGraw hill, New York, 1994.

[13] T. J. E. Miller, Brushless Permanent Magnet and Reluctance Motor Drive, Clarendon Press, oxford, 1989.

[14] T. Kenjo and S. Nagamori, Permanent Magnet Brushless DC Motors , Clarendon Press, oxford, 1985.

[15] ] F.C. Lee, “Power converter modeling and control,” Lecture notes for ECE 5254 at Virginia Tech, Spring 2003, http://www.cpes.vt.edu/public/courses/EE5254/contents.html.

[16] C. Silva, “Power factor correction with the UC 3854,” Application Notes, Unitrode Integrated Circuits.

[17] L. H. Dixon, “High Power Factor Preregualtor for offline power supplies”, Unitrode Design Seminars Manual, 1990.

[18] Hill 1987. 5. Phil Todd, "UC3854 Controlled Power Factor. Correction Circuit Design", Unitrode Application.

[19] Philip C. Todd, “UC3854 controlled power factor correction circuit design”.Texas Instruments, Inc., UC3854, High Power Factor Preregulator.

[20] UC 3854 datasheet, 1998, http://focus.ti.com/lit/ds/symlink/uc3854.pdf .

[21] Abraham I.Pressman , Switching Power Supply Design, Second Edition,McGraw-Hill,1998.

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