A New Configuration of Single Phase Symmetrical PWM AC Chopper Voltage Controller

942 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 46, NO. 5, OCTOBER 1999

A New Configuration of Single-Phase SymmetricalPWM AC Chopper Voltage Controller

Nabil A. Ahmed, Kenji Amei, and Masaaki Sakui,Member, IEEE

Abstract—With the increased availability of power MOSFET’sand insulated gate bipolar transistors, a new generation of simplechoppers for ac inductive loads is foreseen. These new powersemiconductors ease the use of forced commutations of thyristorswitches to improve the supply power factor, even with highlyinductive loads. The ac controllers with thyristor technologycan be replaced by pulsewidth modulation (PWM) ac choppercontrollers which have important advantages. In this paper, asymmetrical PWM ac chopper designed to operate with single-phase inductive loads with a reduced number of controlledswitches is described. The operation as a variable voltage sourceof this converter is evaluated. This includes the conversion char-acteristics, harmonic generation, harmonic distortion factor, andinput power factor. By digital simulation, these characteristicsare investigated theoretically, and to correlate the measurementswith theory, an experimental setup is presented to confirm theanalytical analysis.

Index Terms—AC chopper, duty cycle, harmonic distortion,switching function, symmetrical pulsewidth modulation, voltagecontroller.

NOMENCLATURE

Chopper duty cycle.Switching function of the chopper switch.Switching frequency.Instantaneous inductor current.Instantaneous load current.Instantaneous supply current.Load inductance.Load resistance.Total harmonic distortion factor.Peak value of the sinusoidal supply voltage.Instantaneous supply voltage.Instantaneous output voltage.Instantaneous intermediate modulated chopper volt-age.Supply angular frequency (rad/s).Switching angular frequency (rad/s).Load power factor angle at the supply frequency.Displacement angle.Filter inductor.

Manuscript received March 23, 1998; revised September 29, 1998. Abstractpublished on the Internet June 18, 1999.

N. A. Ahmed is with the Electrical and Electronic Department, Facultyof Engineering, Toyama University, Toyama 930-8555, Japan, and is alsowith the Electrical and Electronic Department, Faculty of Engineering, AssiutUniversity, Assiut 71516, Egypt.

K. Amei and M. Sakui are with the Electrical and Electronic Department,Faculty of Engineering, Toyama University, Toyama 930-8555, Japan (e-mail:[email protected]).

Publisher Item Identifier S 0278-0046(99)07252-4.

Filter capacitor.Input capacitor, for the purpose of power factorimprovement.

I. INTRODUCTION

T HE ac voltage regulator is used as one of the power elec-tronic systems to control an output ac voltage for power

ranges from a few watts (as in light dimmers) up to fractions ofmegawatts (as in starting systems for large induction motors).Phase-angle control line-commutated voltage controllers andintegral-cycle control of thyristors have been traditionallyused in these type of regulators. Some techniques offer suchadvantages as simplicity and the ability of controlling a largeamount of power economically. However, they suffer frominherent disadvantages, such as retardation of the firing angle,causing a lagging power factor at the input side, in particular,at large firing angles, and high low-order harmonic contentsin both load and supply voltages/currents [1]. Moreover, adiscontinuity of power flow appears at both the input andoutput sides.

The recent developments in the field of power electronicsmake it possible to improve the electrical power system utilityinterface. Line-commutated ac controllers can be replaced bypulsewidth modulation (PWM) ac chopper controllers, whichhave better overall performance, and the above problems canbe improved if these controllers are designed to operate in thechopping mode [2]–[6]. In this case, the input supply voltageis chopped into segments, and the output voltage level isdecided by controlling the duty cycle of the chopper switchingfunction. The advantages to be gained include nearly sinu-soidal input–output current/voltage waveforms, better inputpower factor, better transient response, elimination of the low-order harmonics and, consequently, smaller input–output filterparameters [7]–[10].

However, little attention has been given to the input powerfactor of ac chopper controllers. Most researchers who dealwith ac choppers have not considered the variation of the inputpower factor of such controllers [9], [11]–[13]. Others [14]insisted that the input power factor can be made to coincidewith the load power factor and that it is independent of theduty cycle. In fact, this claim is not true from the practical andtheoretical points of view due to the higher order harmoniccontents in the line current resulting from the nature of theswitching processes, in particular, at low values of duty cycle.

On the other hand, control by switching is often accompa-nied by extra losses due to the switching losses. The reductionin the number of switches is essential for control simplicity,

0278–0046/99$10.00 1999 IEEE

AHMED et al.: SINGLE-PHASE SYMMETRICAL PWM AC CHOPPER VOLTAGE CONTROLLER 943

(a)

(b)

(c)

Fig. 1. Proposed PWM ac chopper voltage controller. (a) Circuit configuration. (b) Switching patterns of the gating signals. (c) Block diagram ofthe control circuit.

cost, reliability, and high switching frequency with goodefficiency [11], [12].

This paper describes a new configuration of a symmetricalPWM ac chopper voltage controller for single-phase systems

[15], [16]. The proposed circuit employs only three switches.The modulated chopper switch is placed across a diode rectifierbridge connected in series with the load, and two parallelswitches are connected for the freewheeling purpose. The


Fig. 2. Equivalent circuits for the fundamental and harmonic voltages.

Fig. 3 Design factors of the output filter versus the duty cycle.

proposed controller is more economical, owing to a smallernumber of controlled switches and fewer switching losses.

II. CIRCUIT DESCRIPTION AND PRINCIPLE OF OPERATION

Fig. 1(a) shows the circuit configuration of the proposedsingle-phase symmetrical PWM ac chopper. This circuit hasthe following characteristics. The circuit can operate directlyfrom a single-phase line, the voltage across each switch islimited to the line voltage, and the number of switches hasbeen reduced to three. In the present scheme, the powercircuit is composed of a dc chopper switchacross a diodebridge rectifier connected in series with the load and twoswitches with two freewheeling diodes and connectedin parallel across the load. The series-connected switchisused periodically to connect and disconnect the load to thesupply, i.e., it regulates the power delivered to the load. Theparallel switches and provide a freewheeling path forthe load current to discharge the stored energy of the loadinductance when the series switch is turned off. The basicreason to use a diode in series with each parallel switch isto enable it to be used in a circuit where a reverse voltage isencountered and to complete the freewheeling current paths.The scheme of the present paper uses insulated gate bipolartransistor (IGBT) devices as controlled switches, and gating ofthese switches is based on equal PWM technique or constantpulsewidth method.

The switching patterns of the controlled switches are de-cided by the polarity of the source voltage and the loadcurrent in such a way as to provide a path for the load

TABLE ISWITCHING SEQUENCE OF THEDRIVING SIGNAL

Fig. 4. Normalized intermediate chopper voltage.

Fig. 5. Normalized output voltage.


Fig. 6. Harmonic spectra of intermediate chopper and output voltages.

Fig. 7. Percentage THD of the output voltage, output, and inductor currents.

current whatever its direction. Table I gives the sequence ofclosure and opening of switches, in which the chopper switch

is always modulated with a constant duty cycle. When thesupply voltage and the load current are of equal polarity,normal switching takes place, in which one of the parallelswitches or is completely turned on and the other iscompletely turned off according to the polarity of the supplyvoltage. In other words, when the supply voltage and theload current are positive, is turned on and is turnedoff, and vice-versa. This is not the case when voltage andcurrent are of different polarity, where the on switch from theparallel switches is gated by the complementary signal of themodulated switch instead of continuous conduction. Normalswitching is resumed at the instant when the load currentreverses its direction, as shown in Fig. 1(b). By such switchingpatterns, a continuous current path always exists, regardlessof the load current direction. Since only a single switch ismodulated and due to the fact that a single freewheeling switchis turned on during the majority of the half period of thevoltage source, the switching losses are significantly reducedand, consequently, high efficiency can be approached.

The operation modes are divided into two modes: activeand freewheeling modes. The active mode is defined when themodulated switch is turned on; during the active mode, theinductor current is forced to flow through the voltage sourcevia the modulated switch during its on-state periods. The

freewheeling mode is defined when the modulated switchisturned off and the inductor current paths can be formed by thedirection of the load current, i.e., in the freewheeling mode,the load current freewheels and naturally decays through theswitch with the help of the body diode of or throughthe switch with the help of the body diode of switchaccording to the direction of the load current.

The logic circuit for actuating the controlled switches isshown in Fig. 1(c). For the design of the control circuit, thefollowing requirements must be satisfied:

1) generating gating signals synchronized with the supplyand the load current, as shown in Fig. 1(b) and Table I;

2) giving the ability of changing the duty cycle of thegating pulses.

III. A NALYSIS

The input supply voltage is defined as

(1)

where and are the angular frequency and the peak valueof the input voltage, respectively.

When a switching function with a switching frequencyand constant duty cycle , defined by the Fourier series

of (2) [14], is applied to the chopper switch, the choppermodulated voltage appears in a PWM form at the load


terminals

(2)

Then, the chopper intermediate (modulated) voltage can begiven by

(3)

From (3), the peak value of the fundamental componentof the intermediate chopper voltage and its total harmoniccontents, , can be expressed as

(4)

According to (3) and (4), the fundamental component of theintermediate chopper voltage is proportional to the duty cycle

, which is defined by the ratio of the on time to the totalmodulated period . The dominant harmonicsare suppressed in proportion to the sum/difference betweenthe frequencies of the switching signal and the source voltage

. The value of the fundamental output voltage canbe adjusted according to the required duty cycle, which canbe obtained by comparing a triangular waveform with a dcreference signal.

By defining and setting , (3) showsthat the lowest order harmonics of the chopper modulatedvoltage occurs at the frequencies of and, at leastin theory, the size of the input/output filter components isinversely proportional to the value of . This implies that theswitching frequency in this type of controller should be kepthigh enough to raise the order of the dominant harmonics toa high level.

The output filter reduces the harmonic contents in the outputvoltage from that of the intermediate voltage, given by (3). Theequivalent circuits of the fundamental and harmonic voltagesare shown in Fig. 2. The fundamental component of the outputvoltage is given by

(5)

where

For the fundamental component and ,therefore, the fundamental output voltage can be simplified by

(6)

If the switching frequency is much higher than the supplyfrequency, , the harmonic frequency and the harmonic

Fig. 8. Experimental gating signals for the control switchesS; S1; andS2.

impedances are approximated as follows:

(7)

The harmonic components of the output voltage are given by

(8)where .


(a)

(b)

Fig. 9. Comparison of experimental and computed results of a resistive load. (a) and (b) Waveforms of the load voltage (upper trace), the load current(lower trace), and the input current forfs = 1:8 kHz without the output filter.

For a high switching frequency, and

, (8) can be simplified to

(9)

The total harmonic contents of the output voltage are given by

(10)

In the same way, the equation for the fundamental compo-

nent of the inductor current can be derived as follows:

(11)

If and , then the peak value

of the fundamental component is simplified as

(12)

where .

For the harmonic components of the inductor current

(13)

If and , then the peak values ofthe harmonic components are simplified by

(14)

The total harmonic contents of the inductor current are givenby

(15)

Using the above equations, the percentage THD’s of theoutput voltage and the inductor current can be


(c)

(d)

Fig. 9. (Continued.)Comparison of experimental and computed results of a resistive load. (c) and (d) Waveforms of the load voltage (upper trace), theload current (lower trace), and the input current forfs = 10 kHz with the output filter.

represented by

(16)

(17)

The corresponding input current has a PWM form dis-tributed over a whole cycle, and the difference between thesinusoidal load current and the supply current is the current ofthe freewheeling path. Then, the analytical expression for theinput current can be expressed as the product of the inductorcurrent times the switching function which can be put in theform

(18)

where is the input capacitor current, placed across thesupply terminals for the purpose of power factor improvement,

and is the load power factor at thesupply frequency .

From (18), the fundamental component of the input currentis

(19)

and the effective value is

(20)

Both and are expressed here in peak values.If is defined as the angle between the fundamental

component of input current and voltage, is called thedisplacement angle, then, the input power factor of the choppercan be expressed as

(21)


(a)

(b)

Fig. 10. Harmonic spectra of the load voltage/current and input current of a resistive load forfs = 10 kHz. (a) Unfiltered output. (b) Filtered output.

where

(22)

IV. FILTER DESIGN CRITERIA

The output filter parametersand can be designed withinthe THD’s and allowable in the system.

From (17), the filter inductor is given by

(23)

where

(24)

Substituting in (16), the filter capacitor is obtained as

(25)

where

(26)

Fig. 3 shows the design factors and versus the dutycycle for a switching frequency of 10 kHz.

In the same way, the input filter reduces the harmoniccontents in the input current given by (18). It can also bedesigned within the THD required in the input current byassuming that all the injected harmonics from the chopperflow through the input filter capacitor.

V. PERFORMANCE OFTESTED CONTROLLER

An experiment on a 1.1-kVA (110 V, 10 A, 60 Hz) labo-ratory model was performed in order to verify the feasibilityof the circuit and to investigate the validity of the simulatedresults for the proposed ac chopper. The load parameters usedwere mH and .

Using Fig. 3 and (23)–(26), the output filter parameters forkHz were selected as mH and F to

keep the THD’s % and % at a dutycycle of and to maintain % over all thecontrol range .


(a)

(b) (c)

(d)

Fig. 11. Results of an inductive load(� = 45�) for fs = 10 kHz. (a) Comparison of experimental and computed waveforms of output voltage (uppertrace) and load current (lower trace). (b), (c), and (d) Harmonic spectra of the load voltage, load, and input currents.

Although a proper firing sequence between chopper switchand freewheeling switches was provided, a small capacitorof 1 F, as a voltage suppressor, was placed across thefreewheeling path in order to avoid problems of high-voltagetransients that can occur if both switches are left open in thepresence of a reactive load. The test results for different loadconditions came close to the predicted values. The obtained

calculated and experimental results will be discussed in thefollowing.

Fig. 4 shows the variation of the normalized value of theintermediate voltage over a complete range of control,

to , including its fundamental component ,total harmonic contents , and the harmonic contents in thefiltered load voltage according to (4) and (10).


Fig. 5 demonstrates the normalized value of the outputvoltage versus the duty cycle for an inductive load with

. It is clear that the relation between the fundamentalcomponent of the output voltage and the duty cycleis almostlinear over most of the control range.

Fig. 6 shows the calculated harmonic contents of the inter-mediate chopper voltage, given by (4), and the filtered outputvoltage for a duty cycle of 0.7 and a switching frequencyof 10 kHz.

The relation between the THD’s of the output voltage, load current and filter inductor current

versus the duty cycle , calculated according to (16) and(17) for an inductive load , are shown in Fig. 7.A switching frequency of 10 kHz is considered for thecalculations. Fig. 7 shows a low total harmonic distortion inthe output voltage and current, less than 2% in both; this meansthat the harmonic contents of the output current are almostnegligible, which enhances the assumption that the load currentis approximately a sinusoidal current.

The experimental gating signals from the control circuit toactuate the switches and for pure resistive andinductive loads are shown in Fig. 8. When the load is purelyresistive, as in the control of a heater and light dimming,the freewheeling path and the parallel switches andcan be redundant. Therefore, the control circuit in this casebecomes very simple and the logic gates shown in Fig. 1(c)are dispensed with.

Fig. 9(a) shows a comparison of the dynamic simulationand experimental waveforms of the load voltage (upper trace)and the load current (lower trace) for a resistive load at aswitching frequency of 1.8 kHz and a duty cyclewithout connecting the output filter to the load terminals; alow switching frequency of 1.8 kHz is used for explanation.Fig. 9(b) shows the computed and experimental supply currentwaveforms for the same conditions of Fig. 9(a). Fig. 9(c)and (d) shows the computed and experimental waveforms offiltered output voltage, load, and supply currents at a switchingfrequency of 10 kHz and a duty cycle .

Fig. 10(a) and (b) demonstrates the respective rms harmonicspectra of the load voltage/current and supply current wave-forms of filtered and unfiltered output at a switching frequencyof 10 kHz. These figures indicate that the waveforms of theoutput voltage and current are close to pure sine waves underthe filtering conditions. Unlike the load current, the supplycurrent is significantly distorted by the higher order harmonics,as shown in Fig. 10(c).

The behavior of the chopper circuit with an inductive loadat a switching frequency of 10 kHz and a duty cycle

is shown in Fig. 11. The computed and experimentalwaveforms of the load voltage and load current with its rmsharmonic spectra are shown in Fig. 11(a)–(c), respectively. Itis clear that the harmonic contents of the load voltage andcurrents are almost negligible, and they practically coincidewith a sinusoidal waveform, as predicted. Fig. 11(d) showsthe rms harmonic spectrum of the input current for the sameconditions. In addition to the fundamental component, the linecurrent contains harmonics near the switching frequency andits multiplies.

Fig. 12. Variation of the input power factor, resistive, and inductive loads.

Figs. 10(b) and 11(d) corroborate the important predictedresult, given by (18), that the order of the lower harmonicspectrum in the input current is where .The same conclusion can also be drawn from Fig. 6, whichdepicts the same quantities, but for the intermediate choppervoltage. It is worth noting that the chopper shifts the linecurrent harmonic contents toward high-frequency values, whenhigh chopping frequency is chosen. These harmonics mayadversely affect the supply performance if not filtered out,in particular, at low values of the duty cycle.

The variations of the input power factor versus the dutycycle over the complete range of control and for different phaseload angles, pure resistive and inductiveat the supply frequency, are shown in Fig. 12. Input powerfactor measurement is done by recording the input power andrms values of the voltage and current at the supply side. Itshould be noted that the symmetrical PWM ac chopper showsa poor power factor due to the presence of supply currentdistortion, but it remains still better than that obtained bythyristor switches. More analysis and tests will be performedin the hope of identifying the best performance.

VI. CONCLUSION

This paper has presented an ac chopper circuit for single-phase systems with a reduced number of controlled switches(only three switches) and a simple control circuit. The circuitunder consideration provides a full range of ac power controlBesides the wide and continuous range of control, the relationbetween the fundamental component of the output voltageand the duty cycle is almost linear over most of the controlrange. Due to the nature of the switching process, highswitching frequency and lower order harmonics both in theload and supply sides are reduced, while harmonics nearthe switching frequency and its multipliers are canceled bya filter tuned with the switching frequency and practicallysinusoidal voltage and current waveforms can be obtained.The performance of the proposed circuit has been illustratedas applied to a single-phase voltage regulator by an exampleof pure resistive and inductive loads to verify the feasibility of


the proposed technique. A good agreement is obtained betweenthe experimental and predicted results.

It should be recalled that typical applications of the proposedac chopper consist of light dimming, heat control, and speedcontrol of single-phase induction motors. It is predicted thatthe chopper method allows also improvements in the motorefficiency, as it is not submitted to undesired harmonics atlow frequency, which is the subject of future work.

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[3] S. A. Bhat and J. Vithayathil, “A simple multiple pulse width modulatedAC chopper,”IEEE Trans. Ind. Electron., vol. IE-29, pp. 185–189, Aug.1982.

[4] S. Iida and S. Miyairi, “Effects of PWM applied in single-phase acchopper control,”Trans. Inst. Elect. Eng. Jpn., vol. 103-B, no. 1, pp.7–14, Jan. 1983.

[5] G. Roy, P. Poitevin, and G. Olivier, “A Comparative study of single-phase modulated AC choppers,”IEEE Trans. Ind. Applicat., vol. IA-20,pp. 1498–1506, Nov./Dec. 1984.

[6] G. H. Choe, A. K. Wallace, and M. H. Park, “An improved PWMtechnique for AC choppers,”IEEE Trans. Power Electron., vol. 4, pp.496–505, Oct. 1989.

[7] S. A. Hamed, “Steady-state modeling analysis, and performance oftransistor controlled AC power conditioning,”IEEE Trans. PowerElectron., vol. 5, pp. 305–313, July 1990.

[8] D. A. Deib and H. W. Hill, “Optimal harmonic reduction in ac/acchopper converters,” inProc. IEEE PESC’93, 1993, pp. 1055–1060.

[9] M. Mazzuccheli, L. Puglisi, G. Sciutto, and P. Tenti, “Improvingthe performance of AC/AC static converters with high frequency ACchopper control,” inProc. POWERCON 9, 1982, vol. I-3, pp. 1–9.

[10] D. H. Jang, J. S. Won, and G. H. Choe, “Asymmetrical PWM method ofac chopper with improved input power factor,” inProc. IEEE PESC’91,1991, pp. 838–845.

[11] L. Salazar, C. Vasquez, and E. Weichmann, “On the characteristics ofa PWM ac controller using four switches,” inProc. IEEE PESC’93,1993, pp. 307–313.

[12] P. D. Ziogas., D. Vincenti, and D. Joos, “A practical PWM ac choppertopology,” in Proc. IEEE IECON’91, 1991, pp. 880–887.

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[15] N. A. Ahmed, K. Amei, and M. Sakui, “Improved circuit of AC choppersfor single-phase systems,” inProc. Energy Conversion Conf. (PCC’97),Nagaoka, Japan, Aug. 3–6, 1997, pp. 907–9012.

[16] N. A. Ahmed, K. Amei, and M. Sakui, “A symmetrical PWM ACchopper controller fed single-phase induction motor,” inProc. AL-Azhar5th Int. Conf. (AEIC’97), Cairo, Egypt, Dec. 19–22, 1997, pp. 123–134.

[17] G. Joos and P. D. Ziogas, “A PWM AC controller-high current powersupply,” in Proc. IEEE IECON’92, 1992, pp. 554–559.

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Nabil A. Ahmed was born in Sohag, Egypt, in1966. He received the B.S. (with distinction) andM.S. degrees in electrical engineering from AssiutUniversity, Assiut, Egypt, in 1989 and 1994, respec-tively. He is currently working toward the Ph.D.degree in electrical engineering in the Electricaland Electronics Department, Toyama University,Toyama, Japan.

In 1989, he joined the Department of ElectricalEngineering, Assiut University, as a Teaching As-sistant, where he is currently an Assistant Lecturer.

His research interests include power electronics and electric drives.

Kenji Amei was born in Toyama, Japan, in 1966.He received the B.S. and M.S. degrees in electricalengineering from Nagaoka University of Technol-ogy, Nagaoka, Japan, in 1989 and 1991, respec-tively.

He was an Engineer with Nissin Electric Com-pany Ltd., Kyoto, Japan, from 1991 to 1994. Since1994, he has been with Toyama University, Toyama,Japan, where he is currently an Assistant Professor.He is engaged in research on power electronics.

Mr. Amei is a member of the Institute of Elec-trical Engineers of Japan.

Masaaki Sakui (M’88) was born in Toyama, Japan,in 1949. He received the B.S. and M.S. degreesin electrical engineering from Toyama University,Toyama, Japan, in 1972 and 1974, respectively,and the Ph.D. degree from Tokyo MetropolitanUniversity, Tokyo, Japan, in 1988.

Since 1974, he has been with Toyama University,where he is currently a Professor. He is engaged inresearch and education on power electronics.

Dr. Sakui is a member of the Institute of Elec-trical Engineers of Japan and Institute of Electrical

Installation Engineers of Japan.

A New Configuration of Single Phase Symmetrical PWM AC Chopper Voltage Controller

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Transcript of A New Configuration of Single Phase Symmetrical PWM AC Chopper Voltage Controller