5961309G

Research ArticlePattern Reconfigurable Wideband Stacked Microstrip PatchAntenna for 60GHz Band

Alexander Bondarik and Daniel Sjberg

Department of Electrical and Information Technology, Lund University, Box 118, 221 00 Lund, Sweden

Correspondence should be addressed to Alexander Bondarik; [email protected]

Received 18 September 2015; Revised 30 December 2015; Accepted 10 January 2016

Academic Editor: Giuseppe Castaldi

Copyright 2016 A. Bondarik and D. Sjoberg. This is an open access article distributed under the Creative Commons AttributionLicense, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properlycited.

A beam shift method is presented for an aperture coupled stacked microstrip antenna with a gridded parasitic patch. The griddedparasitic patch is formed by nine close coupled identical rectangular microstrip patches. Each of these patches is resonant at theantenna central frequency. Using four switches connecting adjacent parasitic patches in the grid, it is possible to realize a patternreconfigurable antenna with nine different beam directions in broadside, H-plane, E-plane, and diagonal planes. The switches aremodeled by metal strips and different locations for strips are studied. As a result an increase in the antenna coverage is achieved.Measurement results for fabricated prototypes correspond very well to simulation results. The antenna is designed for 60GHzcentral frequency and can be used in high speed wireless communication systems.

1. Introduction

The high frequency associated with millimeter-waves pro-vides high isolation between neighboring networks due tothe free space loss. This fact is further enhanced by rainattenuation in outdoor links, and the oxygen absorption peakat 60GHz, though this is of little impact for short range sce-narios (

2 International Journal of Antennas and Propagation

In [9] the antenna design described in [8] was imple-mented for 60GHz central frequency. The interconnectionswere realized by metal strips. However, only simulationresults were presented. In [10] a prototype of a reconfig-urable quasi-Yagi antenna design for the 60GHz band wasdescribed. In [11] a pattern reconfigurable patch antennaarray was presented for 60GHz. In the described antenna apatch was connected to a microstrip line stub shortened to aground plane. A p-i-n diodewas used to alter the length of thestub and to change the excitation phase of the patches. Thereported antenna bandwidth was narrow and correspondedto a single patch antenna bandwidth.

In this paper a beam shift realization is presented for anovel stacked microstrip patch antenna first reported in [12].In contrast to lower frequency designs presented in [57]the proposed antenna has a multilayer structure to achievea wide bandwidth specification for 60GHz wireless systems.The antenna consists of an aperture coupled driven patchand a parasitic gridded patch on the antenna top layer. Thegridded patch is composed of nine close coupled rectangularpatches resonant at a central frequency. The antenna has adouble resonance behavior and 22% measured impedancebandwidth.This differs the proposed antenna from the lowerfrequency realization described in [8] where nonresonantmetal pixels compose the parasitic layer; the antenna hasa single resonance behavior and 6% measured impedancebandwidth. In [9] a different antenna feeding as well asincreased substrate thickness with air gap is implementedfor the 60GHz antenna with pixels on the parasitic layer.Although reported simulated impedance bandwidth is widerthan in [8], it is difficult to judge the antenna without meas-urement confirmation.

The pattern reconfigurability property is realized by con-necting adjacent patches on the top layer. The connectionsare modeled by metal strips. Comparing to the realizationsin [57] where parasitic patches were connected to a ground,the antenna top layer connections aremuch easier to realize at60GHz, especially for a multilayer antenna structure. Com-paring to [8, 9] the number of used interconnections is less.

In the proposed design nine different beam directions inbroadside, H-plane, E-plane, and diagonal plane are realizedusing four switches. All eight shifted beams have wider beamwidth compared to the broadside direction beam for the ref-erence antenna without beam shift. This results in the con-siderable increase of beam coverage solid angle for the recon-figurable antenna compared to the reference antenna. Thereconfigurable antenna preserves wide return loss band-width, with about 1 dB gain difference for nine beam realiza-tions. The antenna prototype was fabricated and measured.Experimental results are in good agreement with simulations.It is the first paper for a wideband reconfigurable microstrippatch antenna design at 60GHz confirmed by measurementresults.

The reconfigurable antenna design is described inSection 2. The antenna realization and experimental resultsare in Section 3. The control elements implementation is dis-cussed in Section 4. Conclusions are made in Section 5.

Shorting pin

D C

BA

Microstrip lineGround planeFeeding patch Slot

z

y

x

z

x

p

d

S

Wp

Lp

hp

hm

hb

Figure 1: Top view and side view geometries of the reconfigurableaperture coupled stacked microstrip antenna with gridded parasiticpatch.

2. Antenna Design

2.1. Antenna Geometry. The proposed reconfigurable an-tenna geometry top view and side view are shown in Figure 1.A detailed description of the antenna without switches ispresented in [12]. The measured 10 dB return loss bandwidthfor the antenna is from 54GHz up to 67GHz. The measuredrealized antenna gain is close to 8 dB [12]. The antenna has athree-layer structure. To form it a PTFE (polytetrafluoroethy-lene) substrate and a bonding film are used. A microstripline is on the bottom side of the antenna. It feeds a singlerectangular patch (feeding patch) using aperture couplingthrough a slot in the ground plane. The bonding film actsas a substrate for the feeding patch. On the top, there is agridded parasitic patch formed by nine identical rectangularparts.The gridded patch is coupled electromagnetically to thefeeding patch. In Table 1 the antenna design parameters arelisted.

Shorting pins indicate switch positions for the reconfig-urable antenna in Figure 1. There are four different positions,designated as A, B, C, and D. In the current realizationswitches are replaced by metal strips having 0.1mm width.The pins are shifted on distance from the antenna edge.

2.2. Beam Shift Realization. In this section simulation resultsfrom CST Microwave Studio are presented to investigate thebehavior of the reconfigurable stackedmicrostrip antenna fordifferent shorting pins configurations. The influence of thedistance is studied as well. The antenna model size is 5mmby 5mm. For the antenna pattern reconfigurability evaluationa term called total scan pattern is used. The evaluationmethodwas introduced in [13] for antenna array performanceanalysis. The total scan pattern is obtained by evaluating, foreach fixed angular direction, the largest gain in that directionof all different implementations of switches. In Figure 2 theantenna radiation pattern level graphs are shown in azimuthand elevation angle coordinates for 60GHz. Azimuth is

International Journal of Antennas and Propagation 3

Table 1: Stacked microstrip antenna design parameters.

Description ValueSubstrate Taconic TLY-5

= 2.2, tan = 0.0009Parasitic substrate thickness,

0.127mm

Microstrip substrate thickness,

0.127mmBonding film Arlon CuClad 6700

= 2.35, tan = 0.0025Bonding film thickness,

0.0762mmMicrostrip line width 0.375mmMicrostrip line stub, 0.50mmSlot length 0.20mmSlot width 0.85mmFeeding patch length,

1.10mm

Feeding patch width,

1.29mmParasitic patch length,

1.16mm

Parasitic patch width,

1.20mmParasitic patches separation, 0.10mm

z

yx

No pinsp = 0.1mmp = 0.5mm

AD

AD

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B

BC

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CD

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n (d

eg.)

15 0 1530 30 4545

Azimuth (deg.)

El.

Az.yyxxx

ADD

AADDD

ABA

BB

BC

CCC

CDD

Az.A

Figure 2: Antenna radiation pattern level graph for different con-figurations of shorting pins at 60GHz. In the center of the graphnormalized radiation patterns are shown for = 0.5mm withminimum level 0.02 dB. Outer graphs show realized gain level ofradiation pattern for the reference antenna (no pins) and for thereconfigurable antenna total scan patternswhen all possible shortingpins configurations are applied. Realized gain levels correspond to3 dB from the reference antenna maximum realized gain.

the angle between an plane projection of a vector to aradiation pattern point and the -axis. Elevation is the anglebetween a vector to a radiation pattern point and the plane. In the central part of the graph there are normalizedradiation patterns for the antenna without shorting pins, or

0 0.2 0.3 0.4 0.5 0.60.1

p (mm)

1

1.1

1.2

1.3

1.4

1.5

1.6

/

ref

Figure 3: Relative beam coverage solid angle comparison for 3 dBlevel for the reference antenna ( = 0mm) and for the reconfig-urable antenna with different shorting pins distances from theantenna edge.

reference antenna (corresponding to a configurationwhen allswitches are in OFF state), and for different combinationsof shorting pins placed at = 0.5mm. Configuration A,for example, corresponds to a state when only shorting pinA is present or alternatively when switch A is in ON stateand switches B, C, and D are in OFF state. Normalizedpatterns scale is from 0 dB to 0.02 dB. The purpose of sucha representation is to show the position of the antenna beammaximum direction. Outer graphs in Figure 2 show realizedgain level for the reference antenna radiation pattern andfor the reconfigurable antenna total scan pattern, when allpossible shorting pins configurations are applied. Realizedgain level corresponds to 3 dB from the reference antennamaximum realized gain. For the comparison cases with =0.1mm and with = 0.5mm are presented. The purposeof these graphs is to show the increase in the reconfigurableantenna beam coverage.

Figure 3 shows the computed beam coverage solid anglefor realized gain level of radiation pattern for the antennawith different beam shift realizations; the common realizedgain level of 3 dB from the reference antenna maximumrealized gain is considered. The solid angle for the referenceantenna realized gain (this corresponds to = 0mm) istaken as unit and is compared to the solid angle for thereconfigurable antenna total scan pattern level for differentshorting pins distances from the antenna edge. It can be seenfrom the graph that it is possible for the beam coverage solidangle to increase from about 1.4 times for = 0.1mm toabout 1.6 times for = 0.5mm. Further increase in dis-tance increases the solid angle just slightly; however theantenna return loss becomes much worse. Figure 4 showsthe reconfigurable antenna reflection coefficient for differentbeam shift scenarios. The worst case is for the E-plane beamshift for = 0.5mm, where the return loss has about 8 dBlevel and there is a slight shift to higher frequencies. However,the antenna preserves wide bandwidth for all configurations.

The results for the reconfigurable antenna beam parame-ters for = 0.5mm are summarized in Table 2.The referenceantenna beam has a small shift in the elevation angle due to


Table 2: Reconfigurable antenna simulated pattern parameters.

Switch Position (degree) Gain {realizedgain} (dB)

Beam width (degree) Effic.(%)A B C D Az. El. Az. El.

0 2 9.4 {9.4} 60 56 95ON ON 11 2 8.8 {8.7} 71 58 94

ON ON 11 2 8.8 {8.7} 71 58 94ON ON 0 12 8.4 {7.9} 76 63 93

ON ON 0 8 8.2 {7.7} 76 65 92ON 7 2 8.9 {8.8} 68 58 94

ON 7 7 9.0 {8.9} 68 58 94ON 7 7 9.0 {8.9} 68 58 94

ON 7 2 8.9 {8.8} 68 58 94

SwitchNo pinsB, p = 0.5mmAB, p = 0.5mm

AD, p = 0.5mmAD, p = 0.1mm

8050 55 60 65 70 754540

Frequency (GHz)

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|S11|

(dB)

Figure 4: Reflection coefficient comparison for the reference an-tenna and for the reconfigurable antenna with different beam shiftconfigurations.

the antenna asymmetry caused by the microstrip line. Thisshift further occurs for all the shorting pins configurations.Configurations AB and CD shift the antenna beam on 11in azimuth plane; this corresponds to the antenna H-plane.Configurations AD and BC shift the antenna beam on 10(with respect to the reference antenna) in the elevation plane;this corresponds to the antenna E-plane. Configurations A,B, C, and D shift the antenna beam in the diagonal plane,however more in the azimuth plane than in the elevationplane. The highest gain drop is for the E-plane beam shiftand is 0.8 dB. The corresponding realized gain drop is 1.7 dB;this demonstrates worse impedancematching for the E-planebeam shift. The antenna efficiency remains nearly the sameand it is more than 90% for all configurations. The beamwidth for all the shifted beams is wider, and this results inthe antenna beam coverage increase.

Figure 5 shows a comparison for simulated absolute valueof the electric field distribution at 60GHz at the top surfaceof the reference antenna and antennas with different beam

shift configurations for = 0.5mm. In the reference antennathe electric field phase on the top layer central patch isdifferent from the phase of outer patches. The antenna toplayer excitation is caused by the feeding patch on the lowersubstrate level. As the feeding patch size is comparable withthe single part in the top layer grid, the central patch inthe grid is excited first. Then due to the capacitive loadingthe excitation appears slightly later at the outer patches ofthe grid. The symmetrical arrangement of the patch gridgenerates radiation in the broadside direction. However, byusing shorting pins in the grid, it is possible to introducean asymmetry in the radiation aperture. In the antennaAB configuration, shown in Figure 5, three outer patches inthe grid are connected by shorting pins and thereby haveweak excitation. These patches act as a reflector for theelectromagnetic field and the electric field amplitude is higheron remaining patches, when compared to the referenceantenna. Due to the different excitation phase for the centralpatch and for side patches, the antennamaximum radiation isshifted from the broadside direction in the antenna H-plane.As follows from Figure 4 and Table 2, the antenna AB con-figuration preserves the impedance matching and efficiency;that is, the antenna radiated power is roughly the same as forthe reference antenna and the gain drop in the antenna shiftedbeam is compensated by an increased beam width. Similarexplanations apply to the beam shift in the antenna E-planefor theADconnection configuration and the beam shift in theantenna diagonal plane for the A connection configuration.

3. Measurement Results

Photos of the fabricated antenna samples fragments areshown in Figure 6. The sample with AB interconnectionswas fabricated with = 0.5mm; the interconnection widthwas 0.1mm. Samples with AD and D interconnections werenot fabricated initially and were obtained from the referenceantenna using ribbon bonding. It was realized that manualprocedure for a wire placementmakes it difficult to put a wireexactly at = 0.5mm, and it was decided to make a ribbonbonding somewhere close to the patch edge.The sample withAD interconnection and the sample with D interconnectionwere roughly made with = 0.1mm. The wire width for


Table 3: Measured and simulated patterns parameters.

SwitchPosition (degree)

Meas. {Sim.} Realized gain (dB)Meas. {Sim.}

Beam width (degree)Meas. {Sim.}

Az. El. Az. El.No pins (reference antenna) 0 {0} 7 {9} 7.8 {8.4} 57 {59} 55 {57}AB (H-plane shift sample) 14 {10} 7 {9} 6.7 {7.4} 72 {73} 59 {61}AD (E-plane shift sample) 0 {0} 19 {20} 6.6 {7.3} 75 {77} 58 {62}D (D-plane shift sample) 2 {9} 8 {9} 7.1 {8.0} 60 {67} 64 {67}

No pins AB A AD

018183636545572739091109091272714545163641818220000

(V/m

)

Figure 5: Simulated absolute value of the electric field distribution at 60GHz at the antenna top surface for the reference antenna and antennaswith different beam shift configurations with = 0.5mm.The total antenna substrate size is 5mm by 5mm.

z

y

(a) (b)

Figure 6: Photos of the fabricated antenna samples fragments. (a) is the sample with fabricated interconnections for = 0.5mm. (b) is thesample with ribbon bonding connection for = 0.1mm.The antenna total substrate size is 20mm by 40mm.

the performed ribbon bonding was 0.075mm, which spec-ified the interconnection width. According to performedsimulations the interconnection width decrease in this casewill have only minor influence on the antenna performance.

The samples were measured using a 1.85mm end launchconnector. To be able to attach the connector and to decreaseits influence on measurements, the antenna sample size wasmade 20mm by 40mm. A connector model was made inCST and was used in combination with the antenna modelto verify measurements. More information about radiationpattern and realized gain measurements can be found in [12].

Figure 7 shows the normalized pattern measurements for60GHz. The H-plane shift sample corresponds to AB inter-connections. The E-plane shift sample corresponds to ADinterconnections, and the D-plane shift sample correspondstoD interconnection.Themeasured and simulated results aresummarized and compared in Table 3. It can be seen fromthe table that achieved beam shift angle is 14 for the H-plane shift sample with 1.1 dB realized gain drop compared to

the reference antenna and 12 for the E-plane shift samplewith 1.2 dB realized gain drop. Although the maximum radi-ation angle for the D-plane sample is not shifted significantlyas it was expected, there is a clear pattern shift in Figure 7.For all the measured beam shift samples there is an increasein beam width meaning increased antenna beam coverage.The simulated results from Table 2 are different since thefabricated sample model takes into account the influenceof a connector and different substrate size. The connectorinfluence can be clearly seen on the E-plane radiation pat-terns, which are not symmetrical, and there are considerableripples. The reason for this is an influence of a surface waveradiation and radiation from a connector, as discussed in [12].This is why the effect from interconnections in the antenna isnot well defined for E-plane patterns. However, good resultsagreement between simulation and measurements gives areason to state that forADandD interconnection the antennawithout connector influence will behave in the same wayas was shown in the previous section. The sample with D


Beam shift antenna. Meas.Reference antenna. Meas.Beam shift antenna. Sim.

Beam shift antenna. Meas.Reference antenna. Meas.Beam shift antenna. Sim.

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Theta (deg.) Theta (deg.)H-plane shift sample

E-plane shift sample

D-plane shift sample

60GHz H-plane 60GHz E-plane

60GHz H-plane60GHz E-plane

60GHz H-plane 60GHz E-plane

Figure 7: Normalized measured radiation patterns at 60GHz for the reference antenna compared with measured and simulated radiationpatterns for samples with beam shift. Patterns presented in the reference antenna H-plane and E-plane.


Simulation

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0

|S11|

(dB)

52 54 56 58 60 62 64 66 6850

Frequency (GHz)

Reference antennaH-plane shift sample

E-plane shift sampleD-plane shift sample

(a)

Reference antennaMeasurements

52 54 56 58 6662 64 6850 60

Frequency (GHz)

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0

|S11|

(dB)

H-plane shift sampleE-plane shift sampleD-plane shift sample

(b)

Figure 8: Simulated (a) and measured (b) reflection coefficient for the reference antenna (without shorting pins) and for samples with beamshift.

B

A

C

D

zy

20mm

0.1mm

0.8mm

1.5mm6.6mm

Figure 9: Fragment of the top layer geometry for the antenna with bias lines feeding network. The antenna total substrate size is 20mm by40mm.

interconnection has worse agreement with the simulationperhaps because of no good ribbon bonding connection.The antenna patterns at 57GHz and 64GHz (not includedfor brevity) show only slight deviations compared to thepattern at 60GHz. The measured cross polarization level forthe reconfigurable antenna is roughly the same as for thereference antenna and it is below 20 dB for a normalizedradiation pattern [14].

Figure 8 shows comparison for simulated and measuredreflection coefficient. It can be seen from the graphs that allbeam shift samples have approximately the same 10 dB returnloss bandwidth as the reference antenna sample. This widebandwidth covers entirely 60GHz communication band.

4. Control ElementsImplementation Consideration

In this paper, the connections aremodeled bymetal strips andshould be considered as a prototype for a further realizationusing p-i-n diodes, the applicability of which has been provenin [11]. An alternative control element can be RF MEMSswitches. The reason for using metal strips is to investigatethe pure effect of interconnections and to separate it from

different realizations of connecting switches. This is criticalat higher frequencies where it is difficult to predict all theparasitic effects. Moreover, to mount a control componentspecified for high frequencies is nontrivial.

4.1. Feeding Network Implementation. Simulations show thatthin bias lines, on the order of 0.1mm, can be connectedto the outer patches in the radiating grid with little impacton the field distribution by following two basic principles:the lines should be oriented along the -direction definedin Figure 1 (orthogonal to the dominating electric field), andthey should be connected to the center of the patches, wherethe amplitude of the electric field is low according to Figure 5.

In this section, a case of a bias lines feeding networkimplementation is considered for the fabricated antenna sam-ple having dimensions 20mm by 40mm. Figure 9 showsthe top layer geometry for the antenna with DC feedingnetwork. The bias lines connect the antenna to the metalpads on the substrate edge, where external wires can beconnected by soldering. The size of the metal pads is chosento allow for manual soldering. Figure 10 shows the simulatedabsolute value of the electric field distribution at 60GHz forthe sample with DC feeding network. The metal pads and


zy

018183636545572739091109091272714545163641818220000

(V/m

)

Figure 10: Simulated absolute value of the electric field distribution at 60GHz at the top surface for the antenna with bias lines feedingnetwork. The antenna total substrate size is 20mm by 40mm.

A and B: ONLR = 10

Control element inductance L (pH)1 10 100 1000 10000 100000

2

1

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8

Mai

n lo

be d

irect

ion

(deg

.)

60GHz H-plane

(a)

7

7.2

7.4

7.6

7.8

8

8.2

8.4

Real

ized

gai

n (d

B) A and B: ONLR = 10

10010 1000 10000 1000001

Control element inductance L (pH)

60GHz H-plane

(b)

Figure 11: H-plane radiation pattern main beam direction (a) and realized gain (b) at 60GHz for the reconfigurable antenna with A and Bcontrol elements in ON state.

the bias lines connected to the antenna center have negligibleelectric field excitation. The bias lines connected to bottomand top patches (according to Figure 10) in the antenna gridare slightly shifted from the corresponding patches center tohave a minimum influence on the antenna radiation.Thoughthere occurs some electric field leakage to the bias lines,the effect on the antenna performance is not considerable.The simulated realized gain difference between the antennawithout andwith bias lines andmetal pads is 0.1 dB at 60GHz.

4.2. Control Elements Investigation. The parasitics of theswitches, such as on-resistance and off-capacitance of p-i-ndiodes, can to some extent be included in the simulationsusing circuit models from the data sheets. However, thesemodels are not always validated at frequencies as high as60GHz, and further parasitic effects associated with themounting are expected. The parasitics are expected to influ-ence the performance of the reconfigurable antenna, whichmay require some further design iterations.

A p-i-n diode used as a switch has to have two definedstates: ON state and OFF state. The ON state is oftenrepresented by an inductor in series with a low ohmicresistor , typically having a resistance about 10, and theOFF state is represented by a capacitor in series with

an inductor [6, 8, 11]. In this section, simulation results arepresented aiming at investigating boundaries for resistance,inductance, and capacitance for the ON and OFF statesof the control element. The reconfigurable antenna modelhas substrate size 20mm by 40mm and includes the biaslines feeding network described in the previous section andlumped circuit elements.

We assume the inductance can be taken to be the same inboth states. To determine the inductance, we first considera structure having only control elements A and B in theON state, modeled by an inductor in series with a 10resistor, whereas control elements C andD are absent (perfectOFF state); see Figure 9. Figure 11(a) shows the main lobedirection in the antennaH-plane at 60GHz,when varying thecontrol element inductance. For inductance values less than100 pH the antenna radiation pattern is shifted in theH-planeby approximately 7. This indicates the state when controlelements A and Bwork as shorting pins.This behavior occurswhen the inductor absolute value impedance at 60GHz iscomparable to 10 resistance. For inductance values largerthan 10 nH, the control elements A and B work as insulators.Figure 11(b) indicates a drop in realized gain at 60GHzfor the case when elements A and B are in ON states.The conclusion that can be made from these results is that


L = 0.1pHL = 10pHL = 100pH

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C and D: OFF

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LC

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gai

n (d

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1003 5 10 20 30 5021


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(b)

Figure 12: H-plane radiation pattern main beam direction (a) and realized gain (b) at 60GHz for the reconfigurable antenna with A and Bcontrol elements in ON state, together with C and D control elements in OFF state.

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Figure 13: Radiation patterns in the antenna H-plane (a) and realized gain (b) at 60GHz for the reconfigurable antenna with A, B, C, and Dcontrol elements in OFF state.

the preferable inductance for the control element valueshould be less than 100 pH.

The next step is to study the capacitance of the OFFstate.We consider a structure where control elements A andBare in the ON state, and control elements C andD aremod-eled by an inductor in series with a capacitor, emulating theOFF state. Figure 12 shows the main lobe direction in theantenna H-plane and the realized gain at 60GHz, for varyingcontrol element capacitance and three values of inductance.From these results it follows that the preferable capacitancevalue for the OFF state should be less than 10 fF. It can benoticed that for capacitance around 10 fF beam shift angleincreases to about 40 with realized gain increase; howeverthis state is not stable and needs precise capacitance control,

and it is not clear that this can be achieved using p-i-n diodes.Instead varactors with proper range might be examined ascandidates for control elements.

To complete the control elements investigation we con-sider the case when all of them are in OFF state. This caseis assumed as a basic operational mode for the antenna withradiation in a broadside direction. Figure 13 shows radiationpatterns in the antenna H-plane and the realized gain at60GHz, for varying control element capacitance and threevalues of inductance. In contrast to the two operationalmodes discussed above, in case when all the control elementsare in OFF state, the antenna radiation aperture is sym-metric in the antenna H-plane, and the main lobe directiongraph is not relevant. From the results presented in Figure 13


the same recommendation for the capacitance value follows,as was made for the previous case.The preferable capacitancevalue for the OFF state should be less than 10 fF. For thiscondition the antenna has the highest realized gain in thebroadside direction, about 8.5 dB, and the antenna has goodimpedance matching. For the capacitance values between10 fF and 20 fF there is a sharp drop for the antenna gain, withtwo dominating side lobes in the antenna radiation pattern.For this condition the contribution to the antenna radiationis mainly from the outer patches in the grid structure. Forthe capacitance values more than 50 fF the control elementswork as shorting pins, and the contribution to the antennaradiation ismainly from the center patch in the grid structure.In this case the reconfigurable antenna realized gain is closeto 5 dB, which roughly corresponds to the single microstrippatch antenna gain. However, the reconfigurable antennaimpedance matching is poor in this mode.

5. Conclusion

A beam shift realization is presented for a novel widebandstacked microstrip antenna with gridded parasitic patchdesigned for 60GHz central frequency. The beam shift isobtained by connecting adjacent rectangular patches forminga gridded parasitic structure. Connections are performedby using metal strips and considered as a prototype for p-i-n diodes or RF MEMS switches implementation on theantenna top layer. Nine different beamdirections are realized.The reconfigurable antenna preserves wide bandwidth, about13GHz, for all beam shift scenarios. Measurement results arein good agreement with simulations. The measured beamshift angles are 14 and 12 in the H-plane and in the E-plane, respectively. Simulations show about 60% increase inthe beam coverage solid angle for 3 dB level of radiationpattern for the reconfigurable antenna compared to the ref-erence antenna. We believe that this would be a considerableimprovement for a wireless communication system. Themeasured reference antenna gain is about 8 dB, whereas evenshort range communication might require more. The gaincan be increased in several ways. A moderate increase isachieved by using the gridded parasitic patch antenna as asingle element in an array antenna with only a few elements.More significant increase is achieved bymaking a larger array,or by using the antenna as a feed combined with a reflectoror lens. The latter case may be necessary when consideringmillimeter-wave backhaul applications, where about 30 dBgain is desired.

Simulation confirms high efficiency for the reconfig-urable antenna. The 5mm by 5mm model has more than90% efficiency for all the antenna configurations, althoughthis is expected to be an overestimation compared to theman-ufactured antenna.

A realistic control element implementation is investi-gated. Simulations show that the bias lines implementationhas negligible effect on the reconfigurable antenna perfor-mance. For p-i-n diodes implementation as switches the pref-erable inductance for the ON and OFF states is less than100 pH. The preferable capacitance for the OFF state is lessthan 10 fF.These values are challenging to find in commercial

off-the-shelf diodes where the packaging introduces severalparasitics and may require a high level of integration of thecontrol elements in the manufacturing procedure in order tobe realized.

Conflict of Interests

The authors declare that there is no conflict of interestsregarding the publication of this paper.

References

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