30

3. POWER SEMICONDUCTOR DEVICES 3.1. General overview The era of semiconductor devices began in the 1940s, when three US scientists J. Bardeen, W.H. Brattain and W.B. Shockley, created the first types of transistors. Later by Nobel Prize were awarded all three. Before 1956, semiconductor engineering was regarded as part of low current and low voltage electrical engineering. The currents used in semiconductors were below 1 ampere and voltages only a few tens of volts. The era of power semiconductors began in 1956, when the silicon controlled rectifier or thyristor was founded. After that power semiconductors became an essential part of electric drives to control their speed and torque. In today’s semiconductor engineering, two branches are distinguished microelectronic components that belong to the field of signal processing and − power electronics components that belong to the field of energy conversion. Today’s electronic technologies include manufacture of microelectronic chips, printed circuits, power semiconductors and their modules as well as programming of memories and logic arrays.

Power semiconductors operate mostly in the switching mode. The most important properties of power semiconductors are:

1. Large breakdown voltage 2. Low on-state voltage and resistance 3. Fast turn-on and turn-off 4. Large power dissipation capability

None of power semiconductor device available has all of these properties. Ordinarily, there is trade-off between breakdown and on-state voltages. In bipolar devices, there is also trade-off between on-state losses and switching speeds. It means that in the case of different applications, different types of power semiconductors must be used. The requirements of different applications must be matched to the capabilities of the available semiconductor devices. In many cases, power semiconductors must be combined in series or parallel in order to control the high voltage or high current circuits.

This chapter reviews different parts of the Power Electronics textbook [Mohan]. Operation principles and different types of power semiconductors are described. Special attention is pointed to switching capabilities and external voltage-current characteristics of power semiconductors.

The most commonly used power semiconductors are: - Power diodes - Power bipolar transistors - Power field effect transistors or MOS-FETs - Thyristors or SCRs (semiconductor-controlled rectifiers) - Gate turn off thyristors - Insulated gate bipolar transistors or IGBTs Electric current will flow in a material as the result of existing charge carriers, usually electrons, and the applied electric field. The number of free carriers in different materials varies in an extremely wide range. In metals (e.g. in copper or silver), the density of free electrons is in order of 1023 / cm3. In insulators, e.g., in quartz or aluminium oxide, the free electron density is less than 103 / cm3. For this reason, the electrical conductivity of various materials (such as metal and isolators) is very different, more than 106 S/cm for metals and

31

less than 10-15 S/cm for insulators. Note that the conductivity unit S (siemens) can be replaced with mho, a unit frequently used, derived from the resistivity unit ohm, i.e. from symbol Ω. In the case of some materials, e.g. such as silicon, gallium arsenide or germanium, the density of free carriers and electrical conductivity is intermediate between that of insulators and metals. These materials are called semiconductors. In metals and in insulators, the density of free carriers is approximately constant and cannot be changed in a significant range. In the case of semiconductors, the free carrier density can be changed by many orders of magnitude. For this, either impurity must be introduced into the material or the electric field must be applied. This feature of semiconductors, the ability to manipulate the free carrier density, is very useful in many electrical applications. 3.2. Operation principles of semiconductors Electrons and holes The most popular semiconductor material, such as silicon, has valence electrons and is composed of a regular array (lattice) of silicon atoms (Fig. 3.1). Each silicon atom is bonded with four nearest neighbour atoms by covalent bonds. At the temperature above absolute zero, the temperature rise will result in more bonds broken by the random thermal motion of atoms. This process is called thermal ionization. The process creates a free electron that leaves behind a fixed positive charge on the nucleus of the silicon atom where the bond was broken. This charge is known as a hole in the semiconductor lattice. After some time, another negatively charged free electron might be attracted to the positive charge of the hole. The process of thermal ionization propagates over the whole lattice and creates approximately an equal number of free electrons and holes. The thermal equilibrium density of electrons and holes in a pure intrinsic semiconductor depends on the energy gap (the energy gap for silicon is about 1.1 eV), electron charge, Boltzmann’s constant and temperature. At room temperature (300 K), the free carrier approximate density is 310 /10 cmni = .

+

-Free electron

Ionizedsilicon atom

Covalent bond

Broken bond

Figure 3.1. A silicon lattice and creation of free electrons

Doping of semiconductors Adding of appropriate impurity atoms to the semiconductor can change the thermal equilibrium density of electrons and holes. The process is called doping. In the case of silicon, the appropriate impurities are elements from 3rd or 5th columns of the periodic table, e.g. such as boron and phosphorus. Boron has three valence electrons available for bonding

32

with other atoms in crystal. If boron is introduced into a silicon crystal, it needs an additional electron to bond to the four neighbouring silicon atoms. Boron will accept the required electron from the silicon lattice by capturing a free electron. This immobilizes the free electron and leaves a hole free to move through the crystal. As the result of doping by boron, silicon has more free holes, called majority carriers, than free electrons, termed as minority carriers. Silicon is said to be doped p-type with an acceptor impurity. Phosphorus (5th column) has five valence electrons, but only four are needed for bonding in the silicon lattice. The fifth electron becomes free. The resulting positive charge on the impurity represents the trapped or bound hole. Electrons are now the majority carriers and holes the minority carriers. The impurity (phosphorus) is called a donor, because it donates the fifth electron to the silicon lattice. Silicon is said to be doped n-type (Fig. 3.2).

A

- The acceptor (boron) atom Empty bond

D

- Extra valence electron

The donor (phosphorus) atom

a b

Figure 3.2. Doping of silicon by acceptors (a) and donors (b) In commonly used semiconductor materials, the density of impurity atoms is orders of magnitude smaller than the density of semiconductor atoms. Thus the impurities in semiconductor will not affect the rate of thermal ionization (the recombination of electrons and holes). The product of the thermal equilibrium hole density p0 and electron density n0 is equal to the square of the free carrier density in for the non-doped material, i.e. 2

00 innp = . A doped (extrinsic) semiconductor is electrically neutral even if p0 and n0 are not equal. Recombination In the conditions of thermal equilibrium, the processes of generation and disappearance of free electrons and holes have the same rate. For the minority-carrier devices, the process is characterized by the excess-carrier lifetime τ. In most situations, it is convenient to consider the lifetime τ as the constant of the material. But in two situations, the lifetime varies with the device’s operating conditions. First, in the case of power semiconductors, the excess-carrier lifetime will increase with the internal temperature of the device. This will lead to longer switching times of some devices, such as bipolar devices, thyristors, and Gatos. Secondly, at large excess-carrier densities, the lifetime becomes the function of density. Beginning from a certain value of density, the character of the recombination process is changing. Another process, called Auger recombination process, becomes important and causes the excess-carrier lifetime to decrease, when the density increases. The decrease in the excess-carrier lifetime will increase the on-state losses of some power devices at high current levels. Thus it will be a limiting factor in the operation of these devices. Larger values of the lifetime will minimize the on-state losses, but slow down the switching transition from the off to on-state and vice versa.

33

For the device manufacturers, the excess-carrier lifetime control during the fabrication process is very important. For this purpose, two methods are commonly used: doping with gold (or other heavy metal) and electron irradiation. Gold is an impurity in silicon devices that acts as a recombination centre. The higher the gold-doping density the shorter the excess-carrier lifetime will be. In the case of electron irradiation, the electrons penetrate deeply into a semiconductor and collisions with the crystalline lattice create imperfections that act as recombination centres. These technological measures help to control the excess-carrier lifetime during the manufacturing process and to create the device with the required features. Drift and diffusion Current flow in a semiconductor is the sum of the net flow of holes and the opposite net flow of electrons. The direction of the current is agreed on and is the same as the direction of the movement of the holes. When the external electric field is impressed to the semiconductor, the free holes are accelerated and move in the direction of the field, while the electrons acquire a velocity component opposite to the field (see Fig. 3.3). The free carriers can move via a drift and diffusion. The drift component of the current is the result of impressed electric field and is proportional to the strength of the field.

+

-

--

-

+

+

+ -

E

+

Figure 3.3. Drift of electrons and holes under the influence of an applied external electric field

In contrast, when there is a variation in the spatial density of free carriers, the carriers will move from the region of higher concentration to the region of lower concentration. This movement of free carriers due to the random thermal velocity is called the diffusion component of current. Such spatial variation in the carrier density could be obtained, for example, by variation of the doping density. In general, the both current components flow simultaneously. pn Junctions To form the pn junction of a semiconductor, an n-type region of the silicon crystal must be adjacent to or abuts a p-type region in the same crystal. For example, diffusing of acceptor impurities into the n-type silicon crystal or donor impurities into p-type silicon can form a junction. The junction is characterized by the changing of doping from n-type to p-type as the junction is crossed. The impurity density versus position can be an abrupt (step) function or a linearly graded function. If the acceptor density on the p-type side is much higher than the donor density on the n-type side, the junction is termed a p+n junction. The junctions of power semiconductor devices can have different variations of impurity densities (see Fig. 3.4, a and b).

34

Potential barrier at thermal equilibrium Some of majority carriers on either side of the pn junction will diffuse across the junction to the opposite side, where they are the minority carriers. In this way the space charge layer on either side of the junction will be formed (Fig 3.4, c). The diffusing carriers will leave behind ionised impurities that are immobile and not screened by a sufficient amount of free carriers for electrical neutrality. As a result of this process the space charge with its density ρ and specified variation of the electric field is formed. The increasing electric field across the depletion layer yields the magnitude of the barrier (contact potential). The electric field tends to retard the diffusion process because it acts against it and pushes the electrons back to the n-type side. The holes are pushed back to the p-type side. The equilibrium will be achieved when the hole and electron flux separately sum to zero. At room temperature the pn junction has specified contact potential, e.g. for silicon the approximate contact potential ϕc = 0,7...0,9 eV.

+

- -

-

-

+

+

+ -

Ionized acceptors

+

- -

+ +

Ionized donors

Space charge layer width w

Metallurgical junction

n p

x

-Nd

Na - Nd

Na

x

-Nd

Na - Nd

Na

a b c

Figure 3.4. pn junction with different impurity density versus position (a, b) and space charge or depletion layer (c)

The pn-junction with forward and reverse electric field This section discusses the semiconductor pn junction with applied voltage between p and n regions. When the external voltage is applied to the pn junction, the voltage appears entirely across the space charge region, because of the large resistance of the depletion layer as compared with the rest of the material. If the applied potential is positive on the p side of junction, it opposes the contact potential and reduces the potential barrier. In this case, the junction is forward biased. When the applied voltage makes the n side more positive, the junction is reverse side biased. In this case, the barrier height is increased (Fig. 3.5). The width of the space charge layer (width of depletion region) must also grow if the height of potential barrier grows. The change in the potential requires a change in the magnitude of the total amount of charge. Therefore, the variation in dimensions of the depletion region with the applied voltage is a very important factor for the design of power semiconductor devices. Charge control of a pn junction The reverse voltage increases the potential barrier and the probability that any carrier can diffuse across the junction becomes very small. The minority carrier density will be close to zero. When any of diffusing carriers reaches the depletion layer, the large electric field of

35

space charge will immediately sweep them into the electrical neutral region on the other side of the junction − electrons to the n-type side and holes to the p-type side. This small leakage current across the pn junction is called reverse saturation current Is. The current value is practically independent of the reverse voltage. However, the thermal ionisation process influences the value of the leakage current. The forward-applied (forward-bias) voltage lowers the potential barrier.

+-

--

-

+

+

+

-+

--

++

w

np

+

w0

+-

--

-

+

+

+

-+

--

++

w

np

+

w0

+

+

+

-

-

-

a b

Figure 3.5. pn junction with applied voltage in the forward (a) and the reverse (b) direction Avalanche breakdown The increase of the reverse-bias voltage UBD back over the specified value will cause a rapid increase of current. This process is called avalanche breakdown and the current is called reverse breakdown current. Operation of a diode in the breakdown area must be avoided. A simultaneous high current and voltage leads to the high power dissipation in a semiconductor and will quickly destroy the device. Physically, the breakdown is caused by the effect of impact ionization. Impact ionization. If some of the free electrons with sufficient kinetic energy strike a silicon atom, it can break the covalent bond and release a new electron from this bond. If the kinetic energy of an electron is gained by an applied electric field (applied reverse voltage across the space charge layer) the process is called impact ionization. This process can grow very quickly like a chain reaction (avalanche). Every newly liberated electron can gain enough energy from the electric field to break the next covalent bond when it strikes a silicon atom. As a result of this process the large current and power dissipation will destroy the device. The amount of kinetic energy required to break a bond is the energy gap. The electric field EBD needed for impact ionization is the function of the energy gap, the average time between collisions of electron with the lattice of crystal, the electron mass (m = 10-27 g) and charge. In the case of silicon, the value of EBD calculated from simple formula is approximately 300 kV/cm. The experimental value 200 kV/cm is determined from avalanche breakdown measurements in power devices (see Mohan, Undeland, Robbins). The breakdown voltage of a pn junction is the function of the width of the junction. In the case of power electronic devices, the maximal value of breakdown voltage is hundreds or a thousand of volts.

36

3.3. Power diodes Power semiconductor devices are more complicated in structure and operational characteristics than low-power semiconductor components. Their added complexity arises from the modifications made to the low-power devices to make them suitable for high-power applications. The nature of these modifications is common for all types of power semiconductors. The circuit symbol for the diode is shown in Fig. 3.6, a and the voltage-current i-u relationship for the power diode is shown in Fig. 3.6, b. In the case of power diodes, large currents cause a voltage drop. Instead of the conventional exponential i-u relationship, for low power diodes, the power diode forward bias characteristic is approximately linear, which means that the voltage drop is proportional to the ohmic resistance Ron and to the current. The maximum current in the forward bias is the function of area A of the pn junction. Today, the nominal currents of power diodes are in the range of several thousand amperes and the area of pn junction A may be tens of square centimetres. The cross-sectional view of a pn junction for a power diode is shown in Fig 3.6, c. The anode is connected to the p+ layer and the cathode to the substrate layer n+. In the case of power diodes, the n− layer exists between these two layers, which is termed the drift region. This is the structural feature of power diodes not found in low-power diodes. This layer can be quite wide for the diode with a large reverse breakdown voltage. This wide lightly doped region will add significant ohmic resistance to the forward biased diode and will cause large power dissipation in the diode when it is conducting the current.

≈ 1 ς u

i

UBD

1 Ron

Zoomed

Is

Anode Cathode

a b

Substrate

n−

n+

Drift layer

Anode

Cathode

p+ i c

0

Figure 3.6. The circuit symbol of a power diode (a), voltage-current characteristic of pn junction (b) and cross-sectional view of a pn-junction (c)

In the case of reverse bias voltage only the small leakage current Is flows through the diode. This current is independent of the reverse voltage until the reverse breakdown voltage UBD is reached. After that the diode voltage remains essentially constant while the current increases dramatically. Only the resistance of the external circuit limits the maximum value of current. The simultaneous large current and large voltage in the breakdown operation leads to the excessive power dissipation that quickly destroys the diode. Therefore the breakdown operation of the diode must be avoided.

37

The reverse breakdown voltage UBD determines the maximum allowed voltage for a power diode. To obtain a higher value of breakdown voltage one of two measures could be taken. First, for large breakdown voltages, lightly doped junctions are required because the breakdown voltage is lower if the doping density grows up. Second, the drift layer Wd of high voltage diodes must be sufficiently wide. For example, if a breakdown voltage is more than 1000 V, the doping level must be 1014 cm-3 or less and the minimum thickness of the drift region is about 100 µm. It is possible to have a shorter drift region (at the same breakdown voltage) if the depletion layer is elongated. In this case, the diode is called punch-through diode. One of the ways to obtain higher breakdown voltage is the boundary control of the depletion layer. But all these technological measures will result in more complex design of power diodes. On-state losses of a diode. Most of the power dissipated in a diode occurs in the forward-biased on-state operation. For low-power diodes, the forward voltage drop is approximately constant δUa = 0.7...1.0 V. The power dissipation in a diode can be calculated from a simple formula Ploss = (0.7...1.0) I, where I is the forward current of the diode. For power diodes, this formula is true only in the case of small currents. For large currents the effect of ohmic resistance must be added. Consequently, the voltage drop in a power diode: δU = δUa + Ron I, where Ron is ohmic resistance of the power diode in the forward bias. In the case of high frequency switching operation, the essential commutation losses (switching losses) will appear if the diode goes from the off-state to the on-state or vice versa. The real operation currents and voltages of power diodes are essentially restricted with power losses and thermal effect of power dissipation. Therefore the cooling of power semiconductor devices is very important.

i(t)

u(t)

diR/dt IF

UFmax

t1

UR

t3

t2

t5 t4

trr

URmax UR

t

t IRmax

Upäri

diF/dt

Turn on process

Turn off process Continuous operation

Figure 3.7. Voltage and current diagrams of a power diode during turn on and turn off processes in the case of given rate of current change (di/dt)

38

For power devices, the switching process is the most common operation mode. The power diode requires the finite time interval to switch over from the off state to the on state, or vice versa. During these transitions, the current and voltage of the circuit vary in a large range. The process is accompanied with energy conversion in the circuit components. A power circuit contains many components that can storage energy (inductance, capacitor, electric motor etc.). The energy level of every component cannot change instantaneously (the power used is restricted). Therefore the switching properties of power devices are studied at a given rate of current change di/dt, as shown in Fig. 3.7. Most essential characteristics of power switching are the forward voltage overshot UFmax of a diode (when the diode turns on) and the reverse current peak value (when the diode turns off). During the process, when space charge will be removed from the depletion region the ohmic and inductive resistances causes a relatively large (tens of volts) forward voltage overshot. The duration of the turn-on process of the power diode is the sum of two time intervals ton = t1 + t2, where t1 is the current rise time up to the steady state value of the diode forward current IF and t2 is the time up to stabilizing the forward on-state voltage. In the case of high voltage diodes (n 1000 V), the time interval t1 is approximately some hundreds of nanosecond and t2 about one microsecond. Commonly a shorter turn on transients and lower on-state losses cannot b achieved simultaneously.

The turn off current and voltage transient process duration ts, which is the sum of three time intervals ts = t3 + t4 + t5, where t3 is the forward current decreasing time, t4 the reverse current rise time and t5 is the stabilizing time of the reverse voltage UR. The maximum value of the reverse current is fixed at the end of the time interval t4. Then the current value drops quickly during interval t5. Note that shaded area in Fig. 3.7 is proportional to the charge of the negative current pulse. After the diode turns off the current drops close to zero (only a small leakage current flows). In the analysis of the power diode switching process the energy conversion processes should also be considered. The stored energy in the circuit inductance must be dissipated in resistances or restored in capacitors. When the capacitance is small, the energy will raise voltage up to a very high value.

The reverse recovery process continues during the time interval trr = t4 + t5, when the reverse current reaches its maximum value IRmax and then will decrease. A decrease in the diode reverse current raises the reverse voltage, the maximum value of which reaches up to URmax. Schottky diodes Schottky diodes are much faster than the pn junction diodes, but their breakdown voltage is relatively low. The operation of the Schottky diode is based on the fact that electrons in different materials have different absolute potential energies and the potential energy of electrons in materials is lower (electrons are bound) than the potential energy of free electrons. If a n-type semiconductor is in contact with a metal (whose electrons have a lower potential energy than electrons in the semiconductor), the flux of electrons from the semiconductor into the metal will be much larger (than the opposite flux) because of the higher potential energy of electrons in the semiconductor. As a result of the process, the metal will become negatively charged and the semiconductor will be charged positively. In that way the metal-semiconductor junction is formed, where the metal replaces the p type side of the pn junction. Compared with the pn junction bipolar devices with minority carrier current flow, in the Schottky diodes only the flow of majority carriers occurs. The on-state voltage drop of the Schottky diode is approximately 0.3 V, which is much less than the voltage drop of the pn junction diode. This will lead to smaller energy losses in Schottky diodes. The main advantage of Schottky diodes over pn junction diodes is the faster switching process.

39

3.4. Bipolar junction transistors (BJT) The bipolar transistor has three terminals: collector, emitter and base. In most power applications, the bipolar transistors are used in a circuit with a common emitter. The base is the input terminal and the collector is the output terminal. In power electronic circuits, the bipolar npn transistors are much more common than pnp transistors. A power-switching device must have high current and high voltage capabilities. Therefore the structure of the power bipolar junction transistors is substantially different from that of low power (logic level) bipolar transistors. The collector drift region of power transistors is relatively large (up to 200 µm) and their breakdown voltage is hundreds of volts. To reduce the effect of current crowding in a small area (unequal current density), the base and emitter of power transistors are composed of many parts interleaved between each other. This multiple-emitter layout reduces the ohmic resistance and power dissipation in the transistor. The base thickness of a transistor must be made as small as possible in order to have a high amplification effect, but too small base thickness will reduce the breakdown voltage capability of the transistor. Thus a compromise between these two considerations must be found. Therefore, as a rule, the current amplification gain of high voltage power transistors is essentially lower than in the case of low voltage transistors.

npn-transistorBase

Collector

Emitte r

pnp-transistorBase

Collector

Emitte r

C

B

E

C

E

B

Figure 3.8. The symbol of bipolar npn and pnp type transistors In many applications, the current gain of power bipolar transistors is too small (typically 5 - 10), unsuitable for the logic level control of power transistors. A Dralington-connected pair of bipolar transistors could increase the current gain of a power device. Commonly this connection is designed monolithic (Fig. 3.9). The summary current gain of the monolithic Darlington-connection of two transistors β = β1β2 + β1 +β2 is much higher than the current gain of a single transistor. To speed up the turn-off time of the main transistor, diode D1 is added. For bridge circuit applications, the anti-parallel diode D2 is added. The output voltage-current characteristics of a typical npn power transistor are shown in Fig. 3.10. The characteristics are given for different base currents. The figure shows only the characteristics for the forward bias. The reverse bias characteristics are not shown. Note that the voltage-current characteristics of the monolithic Darlington-connection are approximately the same. The allowed maximum voltage between the collector and the emitter UCEmax depends slightly on the base current (the breakdown voltage is higher on negative base currents). The operation of a power bipolar transistor in the primary breakdown region must be avoided because of simultaneous high voltage and current and large power dissipation that quickly destroys the transistor. For the same reason, the region of the secondary breakdown

40

must also be avoided. In this case the large power dissipation is localized within the semiconductor. The difference between power transistors and low current transistors, shown in Fig. 3.10, is the region labelled quasi saturation on the power transistor characteristics. For logic-level transistors, only the region of hard saturation is shown. If the ohmic resistance of the drift region is Rd, then the boundary between the active and quasi-saturation regions is the line described by the formula iC = UCE / Rd. The forward voltage drop and power dissipation of a transistor in the quasi-saturation region are larger than in the hard saturation region. The effect of the quasi-saturation operation appears in the switching processes when the transistor commutates from the off state to the on state or vice versa. Then the additional time interval is needed to move across the quasi saturation operation region and the resultant switching time of power transistors will be higher than for logic-level transistors.

npn type Darlington

Base

Collector

Emitte r

T1

T2

D1

D2

Figure 3.9. Power transistors in the Darlington configuration

IB = 0

IB1

IB4

IB3 > IB2

IB5

iC

UCE0

IB < 0

Primarybreakdown

Secondary breakdownQuasi saturation

Hard saturation

IB2 > IB1

iC = UCE / Rd

Activeregion

Figure 3.10. The voltage-current characteristic for the bipolar npn power transistor

41

Figure 3.11 shows the current and voltage transients of power BJT if the transistor turns on and turns off in the case of clamped inductive load circuit. The voltage between the base and the emitter rises with the base current, which has the maximum value IBon in on-state operation of a transistor. The collector current will increase after the moment when the voltage between the base and the emitter reaches the value UBon. In the on-state operation, the collector current has value I0. The on-state voltage of a transistor will decrease during two time periods. After the interval t1, the quasi saturation is entered and after the next time interval t2, the hard saturation will be stabilized. Thus, the steady value of the on-state voltage UCEsat will be stabilized. The turn-off process of a transistor begins with a decrease in the base current. Fig. 3.10 shows that a transistor gets out of the saturation also during two time intervals: t3 and t4. After restoring the supply voltage Ud between the collector and the emitter, the collector current decreases quickly and the transistor turns off. At the same time, the base current decreases to zero and the negative voltage between the base and the emitter will be restored.

t = 0

t

t

t

0

0

0

iBon

tdUBE

iB

iC

iBoff

tr

tUCE

t1

t2 t3

t4

0

UCEsat

-iB /dt

UBEoff

Ud Ud

Base current

Collector current

Turn on process Turn off process

UBEon

Figure 3.11. Power BJT current and voltage transients as the transistor turns on and turns off in the clamped inductive load circuit

In power circuits, the commutation losses must be decreased to a minimum and the switching time of transistors must be sufficiently short. The turn-off process of power transistors can be made much faster if the negative control current pulses with abrupt fronts are used. It is common that the safe operating area of currents and voltages for power devices is determined. To control the switching processes and protect the transistor, special snubber circuits are used.

42

3.4. Power MOSFETs Power MOSFETs or metal-oxide-semiconductor field effect power transistors have been used since the early 1980s. Today power MOSFETs are used in many fields instead of bipolar transistors where fast switching is obligatory. For the MOSFETs, like for bipolar transistors, also two types of doping are used. The circuit symbol of n-channel and p-channel MOSFETs is shown in Fig. 3.12. In the case of power MOSFETs, the semiconductor structure is composed of many thousands of cells connected in parallel to achieve a large gain and low on-state resistance. The transistors have three electrodes termed as drain, source and gate. The current between the source and the drain is controlled with the gate voltage.

Gate G

Drain D

Source S

a)

Gate G

Drain D

Source S

b)

UDS

UGS

iD iD

Figure 3.12. Circuit symbols of MOSFETs: a - n-channel MOSFET, b - p-channel MOSFET.

The output characteristic (the transfer curve) of the power MOSFETs (Fig. 3.13) and the voltage-current characteristic (Fig. 3.14) are very similar to the characteristics of bipolar transistors, but instead of the base current, the control signal is now the gate voltage. The function of the drain current versus the gate voltage (the input-output function) will characterize the gain of the transistor. If the gate-source voltage is lower than the threshold value UGS th (Fig. 3.13), the MOSFET is in the off state. Overall, the transfer curve of a power MOSFETs is quite linear compared with the parabolic transfer curve of the logic-level MOSFETs. When the MOSFET is driven by high gate-source voltage, it is working in the ohmic region and its gain value is approximately constant. The threshold voltage is in the range of a few volts. To keep the transistor in the off state when the gate voltage is zero, the drain-source voltage must not be higher than the maximum allowed value. The breakdown voltage UDSbd determines the latter. The MOSFET is in active region if the gate-source voltage is higher than the threshold value (UGS > UGS th). In this region, the drain current iD is independent of the drain-source voltage UDS and depends only on the gate-source voltage (the control signal). The operation point of the transistor depends on the external or load characteristic of the circuit (Fig. 3.14). The supply voltage of the circuit Us must be lower than the breakdown voltage. If the gate-source voltage UGS increases, the operation point of the MOSFET moves to the ohmic region, where the on-state voltage drop depends obviously on the drain current.

43

The switching processes of power MOSFETs are much faster than the switching processes of bipolar transistors, because they have no excess minority carriers that must be moved into or out of the device as it turns on or off. The switching intervals of the power MOSFETs are in the range of 100 to 300 ns. A detailed description of the switching process of the power MOSFET is quite complicated, because the inner capacitances of a transistor between the gate-source and the gate-drain, as well as the external circuit feature, must be considered. These processes could be characterized together with the IGBTs, because the switching operation of both types of transistors is quite similar. The IGBT transistors are described in section 3.7.

UGS

Linearized

iD

0 UGS th

Figure 3.13. Voltage-current transfer curve of an n-channel power MOSFET

UGS1

iD

UDS 0

UGS4 > UGS3

UGS4 > UGS3

UGS2 > UGS1

UGS3 > UGS2

MOSFET’s active region

MOSFET’s ohmic region

MOSFET’s breakdown-

voltage

UDSbd Ut

Load characteristic

Full saturation

Saturation boundary

MOSFET’s cut off

Figure 3.14. Voltage-current output characteristics of an n-channel power MOSFET Commonly, the power MOSFETs are packaged in different power modules (intelligent power module, IPM) with the integrated driver and snubber circuits.

44

3.5. Thyristors (SCR) Thyritor or, the silicon controlled rectifier (SCR) and the diode are among the oldest power semiconductor devices (the thyristor was invented in 1956). During a long period, thyristors were the main non-contact power-switching devices. Most of power converters were based on thyristors. A wide use of thyristors leads to the development of specific power converter circuits. Thyristors and power converters were also successfully developed in Estonia, in the Tallinn Factory of Electrical Equipment. The first thyristors and power converters were manufactured in Tallinn at the beginning of the 1960s. At the same time, the same types of thyristor rectifiers, AC voltage regulators and frequency converters with original circuits and control principles at the Department of electrical drives were developed. The thyristors manufactured in the Tallinn Factory of Electrical Equipment were high quality devices, which were able to compete with the thyristors from well-known international companies. Today’s thyristors are the largest semiconductor devices made, with diameters up to 10 cm. Thyristor (silicon controlled rectifier, SCR) is a power semiconductor device with three electrodes, which has two possible steady operation states. The circuit symbol of a thyristor is shown in Fig 3.15. The thyristor is in off state if there is no control signal on the gate. The short current pulse through the gate circuit will turn on the thyristor. Once turned on and the current higher than the hold current, flows from the anode to the cathode, the thyristor remains in this state after the control pulse is finished. Thyristors commonly used in the AC rectifier circuits where the line commutation is applied. In the case of DC circuits, the use of thyristors is complicated because of their main disadvantage: the turn off by gate circuit is not possible. Turn off is only possible with the forced commutation by the energy charge of external circuit components, when the anode current drops under the value of the hold current. Besides ordinary thyristors (SCR), the gate turn off (GTO) thyrsitors are produced. These devices have two controlled operations, they can be turned on or off by the gate pulse. Therefore the SCR can be termed as one operation thyristors or rectifier thyristors.

Cathode

Gate

Anode

iA + UAK -

Figure 3.15. The circuit symbol of a thyristor The voltage-current output characteristics of a thyristor for different gate currents are shown in Fig. 3.16. The voltage current characteristic of a thyrsitor in the reverse bias is very similar to the same curve of the diode, and a small leakage current flows through the thyristor. Maximum voltage of a thyristor reaches up to thousands of volts. In the forward blocking or off state a small leakage current also flows. After the thyristor turns on, the current (up to thousands of amperes) flows from the anode to the cathode and a small voltage drop (1-2 V) exists between them. The gate current will influence the form of the characteristic. The break-over voltage UB0 is the function of the gate current. Between the two stable states (on and off state) of the thyristor, the third unstable region of characteristics exists. After the control pulse on the gate circuit is given, the thyristor will be switched on. The thyristor remains opened

45

(switched on) if the anode current is higher than the holding current IH. The thyristor closes, when the current decreases below the holding current.

IH iG3 > iG2

iA

UAK0

iG = 0

The on-state operationpoint of a thyristor

IB0

iG1

UB0UH

-URM

iG2 > iG1

The off state operationpoint of a thyristor

Voltage drop

Load characteristic

Figure 3.16. Voltage-current output characteristics of a thyristor Help of an equivalent scheme shown in Fig 3.17 can model operation the thyristor. The equivalent scheme consists of two bipolar transistors.

A

Gate electrode

Anode

Cathode

K

IE2

IE2

IC1 G

Figure 3.17. An equivalent scheme of a thyristor

46

The dynamic properties of a thyristor can be described in the best way by the switching processes and diagrams. Figure 3.18 shows the current and voltage transients of a thyristor when it turns on after a gate pulse appears and turns off after the current direction is changed. The thyristor semiconductor crystal has the form of a large tablet and the gate electrode must reach all the areas of this tablet. During the thyristor opening (turn on) process, the anode current will be distributed to the whole crystal surface at the speed of 0.1 mm/µs. Current may be not distributed homogeneously and local overloading is possible. Therefore the growing rate of forward current iF or derivation diF/dt must be limited. The maximum allowed value for the current growing rate (diF/dt) is about hundreds of amperes per microsecond. For best control of the thyristor opening process, the gate electrode has a special distributed form. The whole turn on process includes three time intervals: the turn-on delay td, the current rise time t1 and the current spreading time t2.

t = 0

t

t

t

0

0

0

iGM

I0

td

iA

iG

UAK

tq

t2t1

trr

Irm

Urev

diFdt

The control pulse

diRdt

t3 t4 t5dUFdt

The turn-on process The turn-off process

Uon

Figure 3.18. Current and voltage transients of a thyristor when it turns on after the gate pulse appears

and turns off after the current direction is changed The turn off process of the one operation (SCR) thyristor is similar to that of a diode. For that the anode current must be decreased below the hold current Ih. The current decreasing rate depends on the circuit inductivity because di/dt = -u/L. The density of excess carriers will

47

decrease by the recombination. The thyristor remains open and the current direction changes the opposite. The thyristor is open until the current attains its peak negative value Irm. The voltage on the thyristor remains small and positive, i.e. the current flows (Fig. 3.18). During the next time interval − the reverse recovery time trr = t4 + t5 the thyristor will switch off and the voltage Urev will stabilize. At the end of the turn off process the excess carriers remain in the medium layers of the thyristor and must also be recombined until the forward voltage will be reapplied. The forward blocking properties of the thyristor will be restored during the turn off time tq. This time interval begins from the moment of current reversing to the moment when the positive forward voltage is allowed. The rate of rise of the reapplied forward voltage must not be higher than the specified value. The voltage change (derivative) du/dt will cause the capacitive current, which acts like the control current. This could be the reason of the unexpected opening of the thyristor. The allowed rise rate of the forward voltage (du/dt)all is the essential technical property of the thyristor. The value of the allowed rise rate of the forward voltage depends on the type of the thyristor, and can be up to a few hundred volts per microsecond. When the forward voltage is applied before the turn off time is finished, the thyristor will again turn on and the forward current will appear (without the control pulse being applied). 3.6. Gate turn off thyristors (GTO) The gate turn off thyristor (GTO) or the two-operation thyristor switches on as the ordinary SCR thyristor, i.e. after the control current pulse will be applied to the gate electrode (Fig. 3.19). The two-way arrow shows that the symbol belongs to the GTO. For the closing or turn-off, a high power negative voltage control pulse must be applied to the gate electrode. The use of GTO thyristors is more convenient than the SCR thyristors,. Therefore they are widely used in high power electrical drives. Another advantage is their much faster switching processes.

Cathode

Gate

Anode

iA

+ UAK -

Figure 3.19. The symbol of a gate turn off thyristor (GTO) The opening (turn-on) control pulse of the GTO must also be more powerful, because the GTO has no regenerative effect on the gate electrode. The control pulse must have a very short front and the duration must be longer than the minimal duration provided in the catalogue. This guarantees the full and fast switching and minimum switching losses of GTO. When working in the open state of the GTO, the positive small control current must be held to guarantee the open state on the small anode current. Note that with small anode currents and the distributed semiconductor structure of a GTO thyristor, some cathode areas can close and

48

other opened areas may be destroyed, because of the over-current in these areas. On danger, the anode current is rapidly decreasing and the thyristor can be closed, the control current of the turned on thyristor must be increased. If the temperature rises, the control current can be decreased. The current and voltage diagrams of the turn on and turn off switching of the GTO are shown in Fig. 3.20. Commonly the turn on process of the GTO is the same as for the ordinary SCR thyristor. The process includes the turn-on delay td and the current rise and stabilizing interval. For the turn off GTO, a powerful negative potential control pulse will be given to the gate electrode. The turn off process can be divided into three intervals. The first one is the storage time ts when the growing negative current is moving the charge stored near the cathode. The second interval includes the anode current decrease or fall time tfi, and the gate - cathode avalanche breakdown tw2, desirable for a short duration. During the third time interval ttail, the small anode current (termed anode tail current) will continue to flow between the anode and negatively biased gate. After the end of the tail current time, the anode current is close to zero. The gate electrode in the closed state of the GTO must be on the negative voltage to achieve the maximum blocking voltage of the GTO and minimize the influence of du/dt. The gate - cathode over-voltage can be reduced by means of an over-voltage snubber.

0

0

0

0

IGM IGT

td

iA

IG

UAK

UGK

tw1

t

t

t

t

du/dt

ts

iA

UAK

UGK

tgq Ud

tfi

ttail

tw2

Turn on process Turn off process

Uon

The decreasing control current

Figure 3.20. Current and voltage transients of a gate turn off (GTO) thyristor

49

3.7. Insulated gate bipolar transistors (IGBT) Bipolar transistors and MOSFETs have technical properties and characteristics that complement each other. Bipolar junction transistors have lower conduction losses in the on state, especially in the case of larger blocking voltages, but have longer switching times. The MOSFETs are much faster, but their on-state conduction losses are higher. Therefore, attempts were made to combine these two types of transistors on the same silicon wafer, to achieve better technical properties. These investigations resulted in the development of the insulated gate bipolar transistor. The semiconductor structure of an n-channel IGBT is similar to that of MOSFET. The difference is that p+ layer is added, which forms the drain (collector) of an IGBT. The possible circuit symbols of an n-channel IGBT are shown in Fig. 3.21. In most of cases, the second version on the right is used. The arrows for p-channel IGBT have opposite direction.

n-channel IGBT

Gate G

Dra in D

Source S

Gate G

Collector

Emitter

iD iC

Figure 3.21. The circuit symbols of the n-channel IGBT The voltage-current characteristics of the n-channel IGBT are very similar to those of the logic-level npn-type bipolar transistor. The difference is that the control signal of IGBTs is the gate voltage rather than the base current as for the bipolar transistors.

UGE1

iC

UCE 0

UGE4 > UGE3

UGE4 > UGE3

UCEbd

UGE2 > UGE1

UGE3 > UGE2

Breakdown voltage

Load characteristic

IGBT’s active region

IGBT’s cut off region

IGBT ohmic region

Figure 3.22. The voltage-current characteristics of an IGBT

50

The diagrams of the current and voltage transients, when IGBT turns on and off, are shown in Fig. 3.23. Commonly these processes are very similar to the transient processes of MOSFETs. The figure shows a switching process of an inductive load. In this case, the current in inductance cannot change very quickly and during the switching process, the load can be observed as the current source. The varying of the gate voltage will control the IGBT. During the turn on delay tdon, the gate - source voltage will rise up to the value UGEth. After that the IGBTs drain (collector) current ID will also be increasing. The steady state value of the drain (collector) current is I0. When the drain current stabilizes, the transistor’s on-state drain source (collector - emitter) voltage drops to the value UCEon. During the turn off process of the IGBT, the drain-source (collector-emitter) voltage will rise to the maximum value UCEff. After that the drain (collector) current will decrease during two intervals to zero. The first interval depends on the IGBTs input stage MOSFETs turn off and the second on the output stage BJTs turn off processes.

t = 0

t

0

0

UGE+

td

UGE

iC

tr

t UCE

t1 t2 t3 t4

0

UCEon

UGEoff

UCEoff UCEoff

t5

The turn on process The turn off process

UGE0 UGE th

MOSFET’scurrent Current of the bipolar transistor

t

IO

Figure 3.23. Diagrams of current and voltage transients of the IGBT, when its turns on and off The IGBTs are the most commonly used power devices in modern electric drives. In the range of smaller power units, the IGBTs are integrated in intelligent power modules (IPM) with the driver and snubber circuits. This module includes also different sensors for circuit protection and automatic control as well as components for galvanic separation of power and control circuits, commonly the optoelectronic converters and connections for fibre optic cables.