22893773 36 2 Field Orientated Control of a Multi Level PWM Inverter Fed Induction Motor

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School of Electrical and Computer Engineering

Department of Electrical Engineering

Final Year Thesis

Semester 1, 2002

Field Orientated Control

of a Multi-Level PWM Inverter Fed Induction Motor

Student Jason Monzu

I.D: 09711107

Supervisor: Dr. W. W. L. Keerthipala

Co-Supervisor: Dr. W. Lawrance

Due date: 31ST

May 2002


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Field Orientated Control of a Multi-Level PWM Inverter Fed Induction Motor

TITLE: FIELD ORIENTED CONTROL OF A MULTILEVEL PWM

INVERTER FED INDUCTION MOTOR.

AUTHOR:

FAMILY NAME: MONZU

GIVEN NAME: JASON CARMELO

DATE: SUPERVISOR:

31ST

MAY 2002 DR W.W.L KEERTHIPALA

DEGREE: OPTION:

BACHELOR OF ENGINEERING ELECTRICAL

ABSTRACT:

This thesis presents the simulation of a Field Orientated Control of a Multi level PWM

Inverter fed induction motor system and its implementation in terms of programming and

code in a real time operating system. Field Orientated Control allows precise

controllability and excellent transient behaviour when used to control an induction motor

by manipulating the angle and amplitude of the torque and speed producing current

vectors. A closed loop feedback control system is employed to give the system stability

and rapid response. The implementation of software code for the vector algorithms

through a rapid phototyping interface will also be investigated in this project.

INDEXING TERMS:

Field Orientated Control, Vector Control, Direct axis, Quadrature axis, Park

transformation, Clarke Transformation, Beta estimation, DSP, Rapid prototyping.

GOOD AVERAGE POOR

TECHINICAL WORK

REPORT PRESENTATION

EXAMINER:CO-EXAMINER:

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Mr Jason Carmelo Monzu

78 Lesouef Drive

Kardinya

Perth W.A. 6163

31st

May 2002

Prof. A Zoubir

Head of School

Electrical and Computer Engineering

Curtin University

P.O. Box U1987

Perth WA 6000

Dear Professor Zoubir,

Please find attached the project thesis titled Field Oriented Control of a multi-level PWM

Inverter Fed Induction Motor for the partial completion of the Bachelor Degree of

Electrical Engineering.

Yours Faithfully

Jason Monzu

09711107

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Table of Contents

LIST OF FIGURES ...........................................................................................................VI

LIST OF TABLES ..........................................................................................................VIII

NOMENCLATURE........................................................................................................... IX

CHAPTER 1 ..........................................................................................................................1

1 INTRODUCTION.........................................................................................................1

1.1 BACKGROUND..........................................................................................................11.2 PROJECT OBJECTIVE .................................................................................................2

1.3 PAST INVESTIGATIONS .............................................................................................3

CHAPTER 2 ..........................................................................................................................5

2 THEORY REVIEW......................................................................................................5

2.1 THREE PHASE INDUCTION MOTOR.............................................................................5

2.2 FIELD ORIENTATED CONTROL................................................................................12

2.2.1 Field Orientated Control overview..................................................................12

2.2.2 Motor Transformation Algorithms...................................................................162.2.2.1 Clarke Transformation .............................................................................16

2.2.2.2 Park Transformation ................................................................................17

2.2.2.3 Beta Estimation ........................................................................................20

2.2.3 Field Oriented Control Principal.....................................................................21

2.3 CURRENT FEEDBACK IN AC VARIABLE SPEED DRIVES ...........................................24

2.3.1 Methods of measuring current .........................................................................24

2.3.2 Current feedback in high performance Vector Drives.....................................25

2.4 SPEED FEEDBACK IN AC VARIABLE SPEED DRIVES ................................................26

2.4.1 Analogue Speed Transducer ............................................................................26

2.4.2 Digital Speed Transducer ................................................................................26

2.4.3 Digital Position Transducers ...........................................................................27

2.5 DIGITAL SIGNAL PROCESSORS IN AC VARIABLE SPEED DRIVES .............................28

2.6 PWM 5 LEVEL INVERTER .....................................................................................30

2.6.1 Full Bridge PWM Inverter Topology ...............................................................31

2.6.2 PWM Control ...................................................................................................33

2.6.3 Five Level PWM Inverter .................................................................................34

CHAPTER 3 ........................................................................................................................37

3 TECHNICAL REVIEW.............................................................................................37

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3.1 COMPARATIVE ANALYSIS OF TORQUE-CONTROLLED IM DRIVES WITH

APPLICATIONS IN ELECTRIC AND HYBRID VEHICLES..........................................................37

3.1.1 Integration of the stator voltage equation........................................................39

3.1.2 Calculation of the rotation angle .....................................................................42

3.1.3 Flux control loop..............................................................................................433.1.4 Torque control Strategies.................................................................................43

3.1.5 PI versus fuzzy control in the torque control loop ...........................................47

3.2 REVIEW CONCLUSION.............................................................................................48

CHAPTER 4 ........................................................................................................................50

4 VECTOR DRIVE CONTROL FEEDBACK LOOPS.............................................50

4.1 SPEED CONTROL LOOP ...........................................................................................54

4.2 TORQUE CONTROL LOOP STAGE 1 ..........................................................................55

4.3 TORQUE CONTROL LOOP STAGE 2 ..........................................................................564.4 TORQUE AND SPEED ERROR SIGNALS......................................................................57

4.5 INVERTER INPUT LOOP............................................................................................60

CHAPTER 5 ........................................................................................................................62

5 PHYSICAL IMPLEMENTATION...........................................................................62

5.1.1 Current Sensing................................................................................................62

5.1.1.1 HCPL-788J Optocoupler..........................................................................63

5.1.2 Speed sensor.....................................................................................................65

CHAPTER 6 ........................................................................................................................66

6 PROGRAMMING AND LOGIC IMPLEMENTATION .......................................66

6.1 RTAI REAL TIME SYSTEM OPERATION...................................................................67

6.2 HARDWARE............................................................................................................67

6.2.1 The parallel bus ...............................................................................................68

6.2.2 The interface board..........................................................................................68

6.3 SOFTWARE .............................................................................................................69

6.3.1 Prototype software code...................................................................................70

CHAPTER 7 ........................................................................................................................72

7 SIMULATION RESULTS .........................................................................................72

7.1 INDUCTION MOTOR MODEL.....................................................................................72

7.1.1 Induction motor simulation results ..................................................................74

7.2 FREQUENCY MAP....................................................................................................75

7.2.1 Frequency map simulation results ...................................................................76

7.3 3S 2R TRANSFORMATION ...................................................................................77

7.3.1 3S to 2R simulation results...............................................................................80

7.4 ROTOR FLUX GENERATION.....................................................................................83

7.4.1 Rotor flux estimation simulation results ..........................................................85

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7.5 TORQUE MAP..........................................................................................................87

7.5.1 Torque map simulation results.........................................................................88

7.5.2 Speed and Torque Error signals ......................................................................89

7.5.2.1 Speed error signal.....................................................................................89

7.5.2.2 Torque error signal...................................................................................907.5.2.3 Torque reference map ..............................................................................91

7.5.2.4 Speed and torque error results..................................................................93

CHAPTER 8 ........................................................................................................................94

8 CONCLUSION............................................................................................................94

8.1.1 Summary...........................................................................................................94

8.1.2 Recommendations for future work ...................................................................95

9 REFERENCES............................................................................................................97

10 APPENDICES .......................................................................................................102

10.1 APPENDIX A - PSCAD SYSTEM BLOCK DIAGRAMS .............................................103

10.2 APPENDIX B - PSCAD OUTPUT WAVEFORMS ......................................................104

10.3 APPENDIX C -AUTOCAD SYSTEM PLOTS ............................................................105

10.4 APPENDIX D - TEXAS INSTRUMENTS TMS320C55 DATA SHEETS........................106

10.5 APPENDIX E - C PROGRAM CODE ........................................................................107

10.5.1 PI controller...............................................................................................108

10.5.2 System routine ............................................................................................108

10.5.3 Convert phase current into i_alpha and i_Beta.........................................109

10.5.4 Transformation into rotating reference frame...........................................10910.5.5 rotor model in rotor flux coordinates ........................................................109

10.5.6 calculation of flux.......................................................................................109

10.5.7 Slip .............................................................................................................109

10.5.8 Angle of flux ...............................................................................................110

10.5.9 Speed control..............................................................................................110

10.5.10 Current control ..........................................................................................111

10.5.11 Back transformation into stator coordinates .............................................111

10.5.12 Calculation of u_alpha and u_Beta ...........................................................112

10.5.13 Calculation of times for space vector modulation .....................................112

10.5.14 Transfer data to modulator ........................................................................112

10.6 APPENDIX F - TYPICAL TORQUE CHARACTERISTICS FOR INDUSTRIAL MACHINERY112

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List of Figures

Figure 2-1: Equivalent Circuit of an AC Induction Motor ....................................................9

Figure 2-2: Torque Vs. Speed characteristics .....................................................................11

Figure 2-3: Typical Vector Transformation ........................................................................12

Figure 2-4: Phasor representation of a typical Vector Transformation..............................13

Figure 2-5: 3-Phase current space vector ...........................................................................15

Figure 2-6: Clarke space vector Transformation................................................................16

Figure 2-7: 2 variable Stationary space vector at o degrees ..............................................18

Figure 2-8: 2 variable rotating space vector at 60 degrees in the excitation frame ...........18

Figure 2-9: Beta estimation .................................................................................................20

Figure 2-10: FOC scheme for an AC-motor showing Park and Clarke Transformations ..22

Figure 2-11: Current load vectors in an AC induction motor.............................................24

Figure 2-12: Circuit Diagram of a Full Bridge PWM Inverter...........................................31

Figure 2-13: Diagram of PWM Output [10] .......................................................................32

Figure 2-14: Triangle comparison PWM implementation [10] ..........................................33

Figure 2-15: Five level staircase output voltage. [1]..........................................................34

Figure 2-16: Five Level PWM Control ................................................................................35

Figure 2-17: Five level PWM generation ............................................................................36

Figure 3-1: Step response (10-25Nm) of torque control loop with idsas reference at 2100

RPM [18] .............................................................................................................................45

Figure 3-2: Step response (0.34-0.2 Wb) of flux control loop with idsas reference at 2100

RPM [18] .............................................................................................................................45

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Figure 3-3: Step response (0.34-0.2 Wb) of the flux loop for torque and flux loops at 2100

RPM [18] .............................................................................................................................46

Figure 3-4: Step response (10-25Nm) of the flux loop for torque and flux loops at 2100

RPM [18] .............................................................................................................................46

Figure 3-5: Step torque modification for both PI and FL controllers. [18]........................48

Figure 5-1: Speed sensor from a shaft mounted transducer................................................54

Figure 5-2: Torque control loop -Rotor Flux and Beta estimations....................................55

Figure 5-3: Torque estimation algorithm ............................................................................56

Figure 5-4: Torque and speed error algorithm ...................................................................58

Figure 5-5: Inverter input algorithm ...................................................................................61

Figure 6-1: HCPL-788J Typical System Diagram .............................................................64

Figure 7-1: Interface structure of parallel bus. ...................................................................69

Figure 8-1: Induction motor simulation model(Appendix A) .............................................72

Figure 8-2: Voltage and Current inputs into the IM ..........................................................74

Figure 8-3: Frequency map simulation block diagram(Appendix A).................................75

Figure 8-4: Frequency Ramp over one second (Appendix B)..............................................76

Figure 8-5: abc to d-q voltage transformation block diagram (Appendix A).....................77

Figure 8-6: abc to d-q current transformation block diagram (Appendix A)......................78

Figure 8-7: d-q to D-Q current transformation block diagram (Appendix A) ....................79

Figure 8-8: IDSe

and IQSe

in the excitation frame (Appendix B) .......................................80

Figure 8-9: Dot product of IDSe

and IQSe

equals zero. (Appendix B) ..................................81

Figure 8-10: Rotor flux estimation block diagram (Appendix A)........................................83

Figure 8-11: Rotor flux calculated waveform (Appendix B) ..............................................85

Figure 8-12: Original Beta estimation(Appendix B) ...........................................................85

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Figure 8-13: New Beta estimation including Theta measurement (Appendix B) ................86

Figure 8-14: Torque map block diagram (Appendix A) ......................................................87

Figure 8-15: Comparison between calculated torque Vs. measure torque over 1 sec

(Appendix B). .......................................................................................................................88

Figure 8-16: Speed error schematic ....................................................................................89

Figure 8-17: Torque Error schematic .................................................................................90

Figure 8-18: Torque reference map.....................................................................................91

Figure 8-19: New loop current IDSenew .................................................................................93

List of Tables

Table 1: Induction motor parameters ..................................................................................73

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NOMENCLATURE

ia, ib, ic - Stator phase currents

i, i - Stator current components

id, iq - Stator current flux and torque components

id ref, iq ref - Stator current flux and torque reference vectors

e - Excitation frame

- Rotor Flux Angle

imR - Rotor Magnetizing Current

J - Moment of Inertia

K1, k2 - Proportional constant and integration constant of

PI controller respectively

Lm - Mutual Inductance between stator and rotor

N - Speed of Induction Motor in rpm

P - Output power of induction motor

DSP - Digital Signal Processor

r - Rotor Angular Speed

- Motor Speed

LR - Rotor Inductance

LS - Stator Inductance

RR - Rotor Resistance

RS - Stator Resistance

TR - Rotor Time Constant

TS - Stator Time Constant

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CHAPTER 1

1 Introduction

1.1 Background

The control of a DC or AC induction motor involves a manipulation of the vector

relationship in space of the air gap magnetic flux to the rotor current. From fundamental

mathematics, a vector represents both the magnitude and direction of a variable, such as

voltage or current. This type of AC variable speed drive gets it name from the fact that the

system attempts to separately measure and control the two vector components that make up

the overall stator current of the motor. Specifically, it attempts to measure and control the

torque-producing current in an AC motor. In a DC motor the switching action of the

commutator determines the position of the armature current in relation to the flux giving

control over the torque of the motor. It is this aspect of the DC motor that makes precise

control a relatively simple and effective procedure.

In an induction motor the rotating flux is responsible for setting up the rotor current and the

relationship between them is a function of the slip and certain other variables. This makes

controllability a more delicate process that requires some form of closed loop control

system to modify the field vectors of the motor. This closed loop system that modifies

vector components of the motor is what is known as Vector Control. Due to the vast

popularity of AC induction motors in industry this type of delicate control is used

frequently to control the speed and the torque of the induction motor.

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In applications with high dynamic requirements, where speed or load change rapidly, a

better form of control is necessary. The dynamic response has to be at least 10 times better

than that provided by standard variable voltage variable frequency drives. In the past, DC

drives have been effectively used for these difficult applications because of their inherent

ability to separately and directly control speed and torque. However, the high maintenance

requirements of DC drives has encouraged the development of alternate solutions. Vector

Control has evolved to provide a level of dynamic performance for AC drives which is

equivalent to or better than DC drives. Thanks to ever advancing technological

developments particularly in the area of semiconductor science and microprocessor and

DSP refinement the capabilities and knowledge are now accessible to allow greater control

over AC motor drives.

1.2 Project objective

The primary objective of this thesis was to simulate and test a Vector Control system with

the intent to produce some physical implementation. This thesis is based on the

continuation of past Field Orientated Control projects and provides corrections and

advancements made in the simulation and implementation.

Implementation initially consists of the hardware component specifications and

information of all sensors and equipment along with a complete blueprint of the Control

System. This blueprint was constructed in such a way that it allows for the progression of

this project in years to come. A large component of the implementation is the code

associated with the algorithms and the control system. A detailed prototype design of a



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Field Orientated Control system was produced in the C programming language but has not

been tested. As a secondary objective the project will also cover areas of Vector Control

code for real time operating using a PC in a Linux environment.

1.3 Past Investigations

This section discusses the problems encountered with precious project and the limitations

of the current project.

Initially the project required a great deal of refinement in terms of motor parameter values

and general system simulation setup. The values of inductance and resistances were

determined based on comparison to other similar models as the values obtained from

previous work were inconclusive. This led to the induction motor model and all

transformations being reconstructed to allow for the desired change to generate the

expected phase voltage and current waveforms.

Problems were also encountered in the Beta estimation schematic from the previous

project. A solution to this was the construction of a new rotor flux estimation to conform

with the new design and the respective changes. The value of rotor position angle theta was

discovered to be incorrect also due to the induction motor setup. This problem generated

incorrect values of current and voltage as the value Beta is used in many of the torque and

current estimation algorithms.

Considerable consideration into choice of Digital Signal Processors concluded that the use

of the TMS320C40 DSP was not financially viable. The actual DSP chip was but the



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CHAPTER 2

2 Theory Review

2.1 Three phase induction motor

For industrial and mining applications the 3-phase AC induction motor is the prime mover

for the vast majority of machines and processes. The beauty of such a device is that it can

be operated directly from the mains or controlled by adjustable frequency drives such as

PWM inverters. The importance of the AC induction motor to the economy is paramount

as they are used in more than 90% of all motor applications for example driving pumps,

fans, compressors, mixers, mills, conveyors and crushers. The popularity of such a motor

stems to its simplicity, reliability and low cost making it a very economically viable

choice. To clearly understand how a Vector Control system works it is essential to firstly

understand the principal operation of the squirrel cage induction motor.

The AC induction Motor is comprised of two electromagnetic parts:

1) Stator which is stationary

2) Rotor which rotates about the ends supported by bearings

The stator and rotor are each comprised of an electrical circuit made of insulated copper or

aluminium to carry current and a magnetic circuit usually made of laminated steel to carry

magnetic flux.



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The stator the outer stationary part of the motor consists of an outer cylindrical frame, a

magnetic path and insulated electrical windings. The outer cylindrical frame is made of

some metal alloy which incorporates and mountings or support brackets. The magnetic

path is comprised of a set of slotted steel laminations pressed into the cylindrical space

inside the outer frame which is laminated to reduce eddy currents and hence reduce losses.

The insulated electrical windings are placed inside the slots of the laminated magnetic path

and in the case of a 3-phase motor 3 sets of windings are required.

The rotor or rotating part of the motor consists of a set of slotted steel laminations pressed

together in the form of a cylindrical magnetic path and the electrical circuit. In the case of

this project specifically the squirrel cage induction motor is used as opposed to the wound

rotor type. This type of AC induction motor is comprised of a set of copper or aluminium

bars installed into the slots which are connected to an end ring at each end of the rotor.

Thus the construction of this type of motor resembles a cage hence the name squirrel

cage motor. The aluminium rotor bars are in direct contact with the steel laminations but

the rotor current tends to flow through the aluminium bars not the laminations.

The connection of the stator terminals of an AC induction motor to a 3-Phase AC power

supply induces a 3-Phase alternating current to flow in the stator windings. The presence

of these currents establishes a fluctuating magnetic flux that rotates around inside the

stator. This speed of rotation in synchronization with the frequency is named the

synchronous speed. In its simplest form the induction motor consists of 3 fixed stator

windings spaced 120 degrees apart. The flux completes one rotation for every cycle of the



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supply voltage and so on a 50Hz power supply the stator flux rotates at a speed of 50

revolutions per second or equivalently 3000 RPM. Therefore the number of poles of the

motor is inversely proportional to its operating speed. The synchronous speed is a function

of the number of poles of the motor and the supply frequency as shown in the relationship

below,

pf

n120

0

=

Where n0 Synchronous rotating speed in rev/min

f Power supply frequency in hertz

p Number of poles

Initially the voltage supplied from the magnetic field created by the stator current induces a

current flow in the rotor bars. The rotating stator magnetic flux passes from the stator iron

path, across the air-gap between the stator and rotor and penetrates the rotor iron path.

Therefore as the magnetic field rotates the lines of flux cut the rotor conductors and

consistent with Faradays law induces a voltage in the rotor windings which is relative to

the rate of change of flux. A magnetic field is set up by the current flow through the rotor

bars which is attributed to the short circuiting of the rotor bars by the end rings. It is this

magnetic field that interacts with the rotating stator flux to produce the rotational force and

in accordance with Lenzs law the rotor will accelerate to flow in the direction of rotating

flux.



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In the case of starting, the rotor is stationary and the magnetic flux cuts the rotor at

synchronous speed therefore inducing the highest rotor voltage and rotor current. As the

motor builds up speed the rate at which the magnetic flux cuts the rotor winding reduces

and therefore the induced rotor voltage decreases proportionally along with the frequency

of the voltage and current. Generally as the rotor speed becomes closer to the synchronous

speed the magnitude and frequency of the rotor voltage decreases showing a directly

proportional relationship. Therefore the closer the relationship between the synchronous

speed and the rotor speed the lesser the induced voltage and current in the rotor would be

and consequently the lower the rotor current the less torque produced by the motor.

Therefore for the motor to produce torque it must rotate at a speed slower or faster than the

synchronous speed. The speed of the rotor is called the slip speed and the difference in the

speed between the synchronous speed and the rotor speed is called the slip. Based on this

information the torque of an AC induction motor is a function of the slip and the amount of

slip is determined by the load torque which is the torque required to turn the rotor shaft.

As the shaft torque increases, the slip increases and more flux lines cut the rotor windings,

which in turn increases the rotor current and consequently the rotor magnetic field and

ultimately the rotor torque. A typical percentage variation for slip is approximately 1% of

the synchronous speed at no-load and 6% at full load.

The relationship for slip in per unit is as follows:

pf

n

nnn

sslip

120

)(

0

0

0

=

==



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Where n0 = Synchronous rotating speed in rev/min

n = Actual rotational speed in rev/min

s = Slip in per-unit

f = Frequency

p = Number of poles in the motor

To understand the performance of an ac induction motor it is useful to represent it in such a

manner that simplifies the system for calculations and observations, hence the equivalent

circuit model was constructed. The model varies in complexity and construction

depending on the type of induction motor and the range of parameters involved.

Figure 2-1 [2] below is a diagrammatic representation of the AC equivalent circuit,

Figure 2-1: Equivalent Circuit of an AC Induction Motor

Where V1 = Stator terminal voltage per phase

I1 = Stator current

I2 = Stator current

R1 = Stator winding resistance

X1 = Stator leakage reactance



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X2 = Rotor leakage reactance

R2 = Rotor resistance

Xm = Magnetizing inductance

Rc = Core losses

Pag = Air gap power

When choosing a motor to suit a certain application or load the primary considerations are

with respect to the torque and speed of the motor. The torque speed curve of an AC

induction motor can be determined using the equivalent circuit model and various

equations.

The following relationship for torque is:

602 n

PT

=

=

Where T = Torque

P = Mechanical Power

n = speed of shaft

Figure 2-2 [2] is a typical example of a torque Vs. speed curve for an induction motor:



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Figure 2-2: Torque Vs. Speed characteristics

On starting the torque must exceed the load breakaway torque for the induction motor to

pull away and thereafter the motor accelerates providing the motor torque always exceeds

the load torque. As the speed increases the torque will approach maximum level as seen in

the above characteristics then passing this point the torque is reduced as the speed

increases until the motor stalls. In reference to the torque-speed curve the final drive

speed stabilizes at the point where the load torque exactly equals the motor output torque.

In the instance where the load torque was to increase, the motor speed would in turn

decrease creating an increase in slip and stator current, hence the motor torque would

increase to match the load requirements.



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2.2 Field Orientated Control

2.2.1 Field Orientated Control overview

Field orientated control is the process of obtaining precise controllability over an induction

motor fed by a multilevel PWM inverter by manipulating the angle and amplitude

components of the stator field. The actual process involves a number of detailed

transformations to obtain a simplified model of an induction motor from a 3-phase time

and speed dependent system into a two co-ordinate time invariant system. Basically it

allows an induction motor to be controlled in a similar fashion to a dc motor by isolating

and simplifying the necessary variables for torque and speed control. The basis of the

control system is that stator current is referenced with respect to a synchronously rotating

frame and the torque (q) and flux components (d) aligned respectively to give

instantaneous controllability. Below depicts a ideal vector system for Field Orientated

Control [19],

Figure 2-3: Typical Vector Transformation



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Figure 2-4: Phasor representation of a typical Vector Transformation

The previous figures display the vector diagrams associated with the division of the stator

current in accordance with the d and q axis. Therefore this approach allows simple and



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accurate acquisition of constant reference values of torque and flux components to

maintain direct torque control. This is achieved in accordance to the following relationship

of torque in the (d,q) excitation reference frame [15],

SqRim

Where m = Torque

R = Rotor flux

iSq = Stator current vector

In essence the above relationship states that m and R iSq are directly proportional to each

other. Consequently maintaining a constant value of rotor flux will give a direct linear

relationship between the torque and torque component (iSq) allowing precise control by

governing the torque component of the stator current vector.

Vector analysis plays a major role in analysing the three-phase components of an AC

motor, namely the voltage currents and fluxes. Each phase of the stator current is

separated into it relative vector of the form ia, ib and ic. The complex stator current vector

is then determined using these three instantaneous stator currents and is represented as

follows [15],

Where

cbas iiii2 ++=

32

j

e=

34

2j

e=



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The following diagram shows the stator current complex space vector [15],

Figure 2-5: 3-Phase current space vector

The system now has to be transformed from the 3-phase rotating current state into a 2-

variable time invariant co-ordinate system, which are equivalent to the armature and field

currents of a DC motor. The transformation from this state is performed using the Clarke

and Park transformations with each step being performed independently with the intention

of reversibility.



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2.2.2 Motor Transformation Algorithms

2.2.2.1 Clarke Transformation

The intention of the Clarke Transform is to perform the first phase of the 2-variable

transformation by converting the 3-phase currents ia, ib and ic into an orthogonal reference

frame iqss

and idss

[15], These components are the in reference to the stator frame. The

diagram is as follows,

Figure 2-6: Clarke space vector Transformation



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This vector can now be represented in terms of a matrix equation to obtain a value for is

and is in terms of the 3-phase phasor currents, this relationship is as follows,

=

cs

bs

as

s

qs

s

ds

i

i

i

I

I

1112

3

2

30

2

1

2

11

This matrix is used as an algorithm to produce a 2-variable co-ordinate system that is

dependent on time and speed.

2.2.2.2 Park Transformation

The Park transformation is the most integral part of the control algorithm. It is the process

of converting the 2-variable iqss

and idss

system into another 2-variable D and Q rotating

reference frame. The vector diagram displays the d axis aligned with the motor flux and

representing the motor rotor flux angle. The Park transformation system is illustrated on

the following page.



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Figure 2-7: 2 variable Stationary space vector at o degrees

From the 2 variable stationary frame it is then the responsibility of the Park transformation

to convert the signal into a rotating excitation co ordinate system. A vector diagram of the

Park transformation is shown below,

Figure 2-8: 2 variable rotating space vector at 60 degrees in the excitation frame



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This allows the construction of the torque and flux components of the stator current and the

ability to manipulate these vectors to achieve direct torque control. These torque and flux

components of the stator current can be determined according to the following vector

matrix,

=

s

qs

s

ds

e

QS

e

DS

I

I

I

I

cossin

sincos

Using the above transformation the stator currents are now in the reference D-Q excitation

frame. These currents are used to control motors torque and flux independently. Similarly

the voltage matrix is as follows,

=

s

qs

s

ds

e

QS

e

DS

V

V

V

V

cossin

sincos



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2.2.2.3 Beta Estimation

Beta is the angle between the stationary reference frame to the rotating excitation frame. It

is essential in the Park transformation to transfer the signal into a rotational co ordinate

system. The angle Beta is derived as follows,

=

=

=

d

e

q

e

QS

e

QS

e

DS

e

DS

F

F

F

F

111 tantantan

Figure 2-9: Beta estimation



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2.2.3 Field Oriented Control Principal

The power circuit for a vector converter consists of firstly a diode rectifier to convert 3-

phase AC to a DC voltage with a DC link capacitor filter to provide a smooth and steady

DC voltage. Secondly a gate controlled semi-conductor inverter bridge to convert DC to a

PWM variable voltage variable frequency output suitable for a AC induction motor.

Finally and most importantly a microprocessor based digital control circuit to control the

switching and provide protection and a user interface. This type of Field Orientated

Control uses essentially a cascaded closed loop system with two separate control loops one

for speed and the second for current. The control strategy is similar to that used for the

control of a DC drive where the speed loop controls the output frequency proportional to

the speed and the torque loop controls the motor in-phase current proportional to the

torque. Following is a basic overview of the Field Orientated Control process including

the Park and Clarke transformations discussed previously [9],



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Figure 2-10: FOC scheme for an AC-motor showing Park and Clarke Transformations

Initially the loop begins with the measured values of current from the motor which are fed

into the primary Clarke transformation system converting the 3-phase currents into a 2-

variable stationary reference frame. The currents idsS

and iqsS

are then transferred into the

secondary Park transform [9] where they are transformed from a stationary system into the

d-q rotating reference frame [9] with rotation angle Beta. The currents iDSe

and iQSein the

excitation frame are then compared to referenced values of torque and flux needed to

produce the required values. The errors are processed and the resulting components

undergo the Inverse Park and Clarke Transformation to be reconverted to a 3-phase voltage

signal that is sent to the Inverter Bridge. The Inverter controls the switching in such a way



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that the desired voltage and frequency are generated at the output according to the PWM

algorithm.

From the previously discussed AC motor characteristics the stator current is comprised of

two components, the Magnetizing current and the load current [25]. The magnetizing

current (IM) is approximately constant over speed and lags the voltage by 90 degrees. The

load current is in phase with the voltage and is directly proportional to the torque over all

changes in load. Therefore the major objective of the Vector Controller is to continuously

calculate the value of the torque producing current.

Under no-load conditions, almost all the no load stator current (IS) comprises the

magnetizing current. Any torque producing current is only required to overcome the

windage and friction losses in the motor [21]. Slip is almost zero, stator current lags the

voltage by 90 degrees resulting in a power factor that is close to zero.

At low motor loads, the stator current (IS) is the vector sum of the magnetizing current (IM)

with a slightly increased active torque producing current. Stator current lags the voltage

hence power factor and slip is poor [23].

At high motor loads, the stator current is the vector sum of the magnetizing current with a

greatly increased active torque producing current, which increases in proportion to the

increase in load torque. Stator current lags the voltage by the angle , so power factor has

improved to be close to full load power factor. Below shows the relationship between the

vector diagrams of the system at low and high loads [27],



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Figure 2-11: Current load vectors in an AC induction motor

The central part of the Vector Control system is the Active motor model that continuously

models the conditions inside the motor in order to calculate the active torque producing

current, slip and magnetic flux.

2.3 Current Feedback in AC variable speed drives

2.3.1 Methods of measuring current

Current feedback is required in AC variable speed drives for a number of purposes [28],

Protection - Short circuit, earth fault and thermal overload in motor circuits

Metering - Metering and indication of the process control system

Control - Current limit and current loop control.



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Over the years several different methods have been developed to measure current and

convert it into an electronic form suitable for the drive controller. The method chosen

depends on the required accuracy of measurement and cost of implementation. The two

main methods of control are Current Shunt and the Hall effect sensor.

The Current Shunt principal is based on the current flowing through a link of pre-calibrated

resistance. The voltage measured across the link is directly proportional to the current

passing through it therefore according to the relationship V=IR the current can be

determined.

The Hall Effect sensor relies on the output voltage being DC which is directly proportional

to the current flowing through the sensor. High accuracy and stability over a wide current

and frequency range are amongst the main advantages of this device. This device is

commonly used with modern digital control circuits.

2.3.2 Current feedback in high performance Vector Drives

High performance drives that employ Field Oriented Control required some kind of current

feedback for the control loop to function correctly. In such cases the motor current varies

according to the load applied to the motor and the torque produced. The stator current for

each phase is used to construct a vector diagram from which requires the current

magnitude of all three phases. This can be achieved preferably with one Hall Effect CT in

each output phase or alternatively two in the output phases and one on the DC bus [29]. In

reality only two phases need to be measured as the final phase can be deduced from the



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relationship between the other current readings, however the DC bus current sensor is still

required for device protection.

2.4 Speed Feedback in AC variable speed drives

In closed loop speed control of electric motors and positioning systems, the speed and

position feedback from the rotating system is provided by transducers, which convert

mechanical speed into an electrical quantity compatible with the control system [7]. Some

of the more common techniques used today are Analogue speed Transducers, Digital

Speed Transducers and Digital Position Transducers.

2.4.1 Analogue Speed Transducer

Analogue Speed Transducer such as a Tachometer Generator which converts rotation

speed into an electrical voltage, which is proportional to the speed, and transferred to the

control system over a pair of screened wires.

2.4.2 Digital Speed Transducer

Digital Speed Transducers such as Rotary Incremental Encoders [17] that convert speed

into a series of pulses. The pulses are transferred to the control system over one or more

pairs of screened wires.



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2.4.3 Digital Position Transducers

Digital Position Transducers such as Rotary Absolute Encoder [17] that converts position

into a bit code whose value represents an angular position. The code is transferred

digitally to the control system over a screened parallel or serial communications link.



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2.5 Digital Signal Processors in AC variable speed drives

With the advances in microprocessor technology and DSP controllers there has been a host

of commercially available microprocessors that provide PWM module for control of

inverters. Typically the signals and algorithms associated with such a system can be very

complex and lengthy to compute. However through the use of DSP and the digitization of

the signal, calculation of the output, and the output to the D/A converter all must be

completed within the sample clock period. The speed at which this can be done determines

the maximum bandwidth that can be achieved with the system. For adequate dynamic

response or a Vector Control system the calculations associated with Field Orientated

Control completed around 2000 times per second which is less than 1ms [22]. The ability

to continuously model the induction motor at this speed only became viable recently the

development of the 16 bit microprocessor. Initially sufficient processing power was quite

expensive, but over a period of time, the cost of the processors have reduced and

processing speed has increased significantly.

Many different types of microprocessors have been utilized in the implementation of Field

Orientated Control. The MC68040 has been utilized in both field-orientated control and in

the implementation of ANN observer estimation of the rotor flux angle. The system also

includes 4 Mbytes of RAM, two 32-pin EPROM sockets, dual port MC68681 I.C. for

serial port communication, Local Resource Controller (LRC), VSB and VME bus

interfaces. One port of the MC68681 is connected to the 486 PC. The MC68040 operates

at 27.6 MIPS with a clock frequency of 33 MHz and 32-bit address/data bus [16].



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The TMS320 is a DSP from Texas instruments that has been specifically designed for

Field Orientated Control. The TMS320's high level of throughput results from the chip's

comprehensive instruction set and highly pipelined architecture. Based on a modified

Harvard Architecture, the TMS320 allows transfer between program and data spaces for

increased device flexibility. Constants can be stored in program memory, and program

branches based on data computations can be performed. Thus, parallel operations can

execute a complex instruction in one 200-nanosecond (ns) cycle. Competing chips

typically execute instructions in 250-, 300- or 400-ns cycles [16].

The TMS320's speed in enhanced by the arithmetic logic unit's (ALU's) 16 x 16-bit

multiplier that uses 16-bit, signed 2's complement numbers to form a 32-bit product in 200

ns. Although the TMS320 accepts 16-bit inputs and has a 16-bit output, it features a 32-bit

ALU/accumulator that carries out all arithmetic operations to 32 places for greater numeric

precision [16].



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2.6 PWM 5 level Inverter

Early forms of DC to AC conversion are derived from the basic buck converter, where a

power semiconductor is used to switch a DC signal into a square wave (Square Wave

Inverter). With the introduction of power storage components, such as inductors and

capacitors, this square wave will resemble a rough sinusoidal wave. The desired sinusoidal

output can be further refined with the use of logic control on the semiconductors, enabling

the positive and negative peaks of the square wave to be delayed (Phase Shifted Square

Wave inverters), creating a zero level. All these adjustment were made in the aid of

producing a perfect sinusoidal output or in other words decreasing the Total Harmonic

Distortion (THD) [28].

PWM inverters further refine the conversion of the DC input to an AC output. This

advancement in inverters was not possible until recent semiconductor technology

advancements, in this particular project the semiconductors must have a high power rating

combined with a high switching frequency. PWM inverters use high-speed semiconductor

switches to switch the DC signal at varied time intervals, this will create varied pulse

widths, hence the name Pulse Width Modulator.



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2.6.1 Full Bridge PWM Inverter Topology

The basic construction of a PWM inverter can be understood by using the single-phase full

bridge model depicted in the figure below.

Figure 2-12: Circuit Diagram of a Full Bridge PWM Inverter [Constructed using PSIM demo]

As you can see the DC input signal is feed into the two legs of the full bridge inverter and

with the aid of the high-speed semiconductor switches converted in to an ac signal. The

semiconductor switches are controlled but he PWM control logic, which switch the DC

input at varied time intervals, creating varied pulse widths. The adjustment function is

called a Modulation function, M(t). The Modulation function is defined as follows [28],



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carriertheofmagnatudewhereA

signalmodulatingtheofmagnatudeAwhere

2/)1()(

c

m

=

=

= cm

ANAtM

The Modulation function is used to determine the output, as shown in the equation below.

VintMtVd )()( =

The desired output is no longer the dc average value, but is a unique wanted component.

This should be a moving average to denote the variation of the average output with time.

The moving average can be defined by the integral.[10]

=

t

Tt dssfTtV )(

1

)(

The figure below depicts the PWM output and the moving average created by the output.

Figure 2-13: Diagram of PWM Output [10]



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2.6.2 PWM Control

In order to control the semiconductors of the PWM inverter, a triangular reference signal

along with a sinusoidal reference signal are used to determine the switching times for the

semiconductors. These inputs are compared and will control the ON/OFF states of the

semiconductors. The following diagram depicts the two inputs of the comparator.

Figure 2-14: Triangle comparison PWM implementation [10]

The PWM logic is created by comparison of the sinusoidal signal to the triangular signal.

When the reference sinusoidal signal is above the triangular sawtooth signal the PWM

logic is switched on and when the sinusoidal signal falls below the triangular sawtooth

signal the PWM logic switches off. The duration of this switching is dependent on the

length of time from when the sinusoidal signal remains above or below the triangular

sawtooth signal. This process is displayed in the above figure 2-14 [28].



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2.6.3 Five Level PWM Inverter

The intent of using the Five Level PWM Inverter is to reduce the THD of the output ac

signal. The Five Level PWM signal involves constructing the ac output with five discrete

voltage levels. However when doing this the design and operation of the inverter will

change. The figure below depicts the desired output of such an inverter, which the five

discrete voltage levels clearly show. It is important to point out that the pulse modulations

have been exaggerated to give a better understanding.

Figure 2-15: Five level staircase output voltage. [1]

Like the progression of the multilevel PWM inverter suggest a five level PWM is

controlled by four triangular reference signals and one sinusoidal reference signal, shown

in the figure below. It can be seen below that the four different triangular references, or

bands, have differing offsets with the same amplitude, this essential for the inverters

operation



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Figure 2-16: Five Level PWM Control

The basic circuit design is depicted in the PSCAD diagram below, which clearly shows the

importance of the offset carriers, which control the PWM inverter. This system is not

useful for driving an induction motor as the output will have voltage spikes due to the

inductive currents not being permitted to flow when the IGBT interrupts the current to the

mid levels.



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Figure 2-17: Five level PWM generation

Once the circuitry for a single-phase five level PWM inverter is fully understood, little

effort is needed to expand the circuit into the required three phase five level PWM inverter.

This was easily implemented by duplicating the single-phase inverter three times and

modifying the modulating phase of each respective modulating sine wave signal for each

phase [10]. Although the above circuit is a five level PWM inverter, this circuit is not

possible to be used in the implementation of this project as the predominately inductive

load of the motor, causes KCL violations as there is no current path for the reverse current

from the inductive load.



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CHAPTER 3

3 Technical Review

3.1 Comparative Analysis of Torque-Controlled IM Drives with

Applications in electric and Hybrid Vehicles.

This review presents torque-controlled drives control based on machine flux and torque

estimation. The theory surrounding this review has been credited to the IEEE paper on

Analysis of Torque-Controlled IM Drives with Applications in electric and Hybrid

Vehicles [18]. The theoretical aspects of the methods are discussed and a comparative

analysis is provided with emphasis on DSP Implementation and experimental results.

Problems in the application of these techniques to propulsion systems are also discussed

and possible solutions are presented.

An electric propulsion system is generally based on a torque-controlled electrical drive.

Many of the methods of Vector Control published require a torque feedback signal.

In this review a comparative analysis of the torque-controlled methods with current

controllers is presented. For the purpose of comparison, torque controlled methods based

on stator voltage orientation are being studied. The dependence of the system based on

implementation methods, calculation the phase angle and the effects of obtaining the d-q



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current references using open loop and closed loop flux and torque observers are

compared.

The topics that this paper focuses on are as follows:

1) Digital integration methods for obtaining the flux from

the stator voltages:

a) saturation of each flux component feedback.

b) low pass filter without any feedback.

c) saturated magnitude on the feedback path.

2) Open loop or closed loop torque controller:

a) calculation ofiqs current from the torque reference.

b) calculation ofiqs current from the closed loop torque PI controller.

c) calculation ofiqs current from the closed loop torque fuzzy logic based controller.

3) Open loop or closed loop flux controller:

a) ids reference without control of flux.

b) ids reference current from flux PI controller.

4) Calculation of the angle for Vector Control from:

a) flux coordinates.

b) estimated electrical frequency.



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The different combinations of the above control techniques influence the performance of

the system. The above techniques are discussed in detail in the following sections.

3.1.1 Integration of the stator voltage equation

Different techniques for the estimation of the torque produced by the induction motor are

presented in the literature. Among these, the torque estimation using the integration of

stator voltages depends less on the machine parameter variations. First, the flux

components are computed by integration of the voltages in the stationary reference frame

(,) using the following equations[18]:

dtiRv

dtiRv

ssss

sss

s

)(

)(

=

=

The accuracy of this calculation depends on the accurate value of the stator resistance as

well as on the integration method. Another aspect of the control is the electromagnetic

torque (Te) which is estimated using the following eqn. [18]:

[ ] ssss iiPp

Te =22

3

This estimated value of torque can be used for the torque control loop even if the induction

machine control method is based on rotor flux orientation.



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In the above flux equations the currents are obtained using current sensors, and the

voltages are the outputs of the PI controllers after d-q to a-transformation. This method

though quite reliable introduces errors in flux estimation, these are as follows,

1) A delay between the reference voltages used in estimation and the actual fundamental

components of the phase voltages.

2) Measured currents are passed through the A-to-D converter at a given sampling

frequency.

3) Flux integration by a digital low-pass filter introduces errors in phase and gain.

The common solution consists in the use of a low-pass filter that has the input-output

relation given by [18]

xs

yc

+

=

1

where c is chosen so that "s +c" ~ "s" for all the operating frequencies. If the lowest

stator frequency that should pass properly through filter is 8 Hz, then c


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The output can be further improved by re-writing the transfer function to introduce a

variable feedback within the low pass filter . This implementation leads to behaviour closer

to l/.s. Also, by saturating the feedback component (y), the dc bias component can be

limited.

xs

ys

yxs

ys

sx

ss

ys

xs

y

ccc

cc

c

+

++

=+

=

++

+

=

+

=

11

11

11

][1

]1[1

1][

]1[1

]1[1

1][

][][][

22

11

21

kxT

Tky

Tky

kyT

Tky

Tky

kykyky

sc

s

sc

sat

sc

sc

sc

+

++

=

+

+

+=

+=

Where ysat represents the saturated feedback.

Several digital integration methods have been tested in and a comparison of the current and

flux waveforms are presented further in the report (limitation of both flux components on

the feedback path, low-pass filter without any feedback and limitation of the flux

magnitude on the flux feedback). Experimental results have shown that the integration

method with limitation of the flux magnitude leads to reduced ripple.



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3.1.2 Calculation of the rotation angle

In any Vector Control method, an important item is the calculation or estimation of the

rotational angle or the SIN/COS function corresponding to this angle. The current

measurement resolution influences the angle estimation.

To obtain an estimate of stator flux orientation the following calculations can be used:

Electrical frequency [18],

=

=

dt

RivRiv

e

ss

ssssssss

e

22

)()(

Estimated flux components [18],

22

22

cos

sin

ss

s

ss

s

+=

+=

Precision ofdepends on the accuracy ofe that is a small dc value in low speeds (low

frequencies), obtained through a relationship in between small values (i or ). Accordingly,

at low speeds, accuracy of calculation is jeopardized by the large percentage of ripple in

e. For this reason, using the definition of SIN/COS based on the estimated flux

components in low speeds leads to better results than the calculation of electrical frequency



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(e). Using electrical frequency estimation at high speeds is better since it is essentially a

dc component that can be easily filtered to remove the inherent noise. Both methods were

implemented and tested and the experimental results confirmed the above statement.

3.1.3 Flux control loop

Since the dynamics of the flux regulator is not very important, the flux control loop is a

typical PI control loop based on the estimated flux magnitude signal. A Fuzzy Logic

controller would not introduce any major improvement for the performance of the overall

system. However, the output of the PI controller can be limited to different values. In order

to improve the flux loop dynamics, a negative ids current has been allowed during the

transients. The level of the positive limit at the controller output is a function of the

inverter output phase current ratings.

When the flux orientation is perfect, qs= 0 and , ds = | |. There are two possibilities of

developing the control based on ds or | |. The flux magnitude has been calculated and

used as the feedback signal in this control approach

3.1.4 Torque control Strategies

This review presents three different methods of torque control based on stator or rotor flux

orientation, these are as follows,

1) Open-loop torque control loop with closed loop stator flux control.

2) PI/Fuzzy torque control loop with closed loop stator flux

control based on rotor flux orientation.



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3) PI/Fuzzy torque control loop with closed loop stator flux control based on stator

flux orientation.

The influence of each control loop on the overall system performance is next discussed. In

this study, the current control loop sampling time was at 100 s, and the outer control

loops (torque and flux) are sampled at 300 s and, 600 s, respectively. In this particular

report to achieve better results special caution has been taken in software such that to do

not modify both flux and torque at the same time.

In the open-loop torque control, iqs reference is calculated by [18],

eqs

TKi =

The dynamic results will be affected whether iqsis calculated based upon flux reference or

estimated flux magnitude seen in the figures below. Figures 3-1 to 3-4 are presenting step

modification in flux or torque for the same base system, for different control structures.

Fig. 3-1 presents a control system containing a torque control loop while idsis introduced

as reference without any flux control loop. Fig. 3-2 presents a flux control loop with iqs

command without a torque control loop. Figs. 3-3 and 3-4 are introducing the full system

with both torque and flux control loops for step modification in either flux or torque. It can

also be seen the improved performance of the system containing both torque and flux

control loops.



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Figure 3-3: Step response (0.34-0.2 Wb) of the flux loop for torque and flux loops at 2100 RPM [18]

Figure 3-4: Step response (10-25Nm) of the flux loop for torque and flux loops at 2100 RPM [18]



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3.1.5 PI versus fuzzy control in the torque control loop

In propulsion systems, the torque control loop response is important. In order to decrease

the response time of the torque control loop and to minimize the influence of the induction

machine speed on the torque loop transients, a fuzzy logic control loop instead of the

conventional PI controller has been investigated. Fig. 3-5 [18] presents the torque control

loop responses at different speeds (500 RPM, 5000 RPM, and 7500 RPM) for both control

methods. The fuzzy logic controller has been developed in software using a simple

structure with two inputs, with seven triangular membership functions for each input,

linear de-fuzzification (Sugeno controller) [34] and triangular membership functions for

the iqs current (output of the FLC controller). The results are demonstrating that the FLC

has the same transient response at each speed. Measurement of the system efficiency has

been performed for different methods under study. The differences between these methods

at nominal speed are less than 2% for different torque levels.



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Figure 3-5: Step torque modification for both PI and FL controllers. [18]

3.2 Review conclusion

This paper presents a review of the torque-controlled IM drives based on the current-

controlled PWM voltage source converters and flux control loop. This review was found

to be quite relevant to the underlining themes of Field Orientated Control particularly as

they are directed at a propulsion system such as a hybrid vehicle. The review presents



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different integration methods of the stator voltage equation and compares them based on

the type of control method and the response of the system to these changes.

The advantages of stator flux control over rotor flux control are reviewed and the results

demonstrate the advantage of a system with two control loops (torque and flux) in terms of

stability and variable ripple despite the higher complexity. The report covers both PI

controllers and Fuzzy logic and focuses on the benefits of each. Essentially the Fuzzy

logic controller is more complex, but is less influenced by the parameter variations and less

dependent on speed.

This review concentrates on the most appropriate methods of control for application in a

hybrid and electric vehicle. The results obtained allow further insight into Field Orientated

Control and allow a greater depth of understanding into the control methods with respect to

a direct propulsion system.



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CHAPTER 4

4 Vector Drive Control Feedback loops

The term closed loop feedback control emphasizes the nature of the control system, where

feedback is provided from the output back to the input of the controller. A perfect example

of a closed loop feedback control system is a driver in a motor car. The speed of the car is

assessed by the drivers eye looking at the speedometer (speed transducer). This measured

speed is mentally compared to the desired speed (speed reference), which may be the set

limit for that stretch of road. Depending on the error, the driver (controller) may decide to

increase speed by further depressing the accelerator to adjust the speed reference. The

driver continually measures and re-evaluates the error between measured and desired speed

and adjusts the accelerator accordingly. At the same time the driver might be

simultaneously engaged in several other feedback control closed loop tasks such as

steering the car.

In the Vector Control system employs the use of an AC frequency converter to control the

voltage and frequency fed to the motor to suit the load characteristics . For example when

an operator selects a speed setting on a potentiometer, the system implements this selection

this selection by adjusting the output frequency and voltage to ensure that the motor runs at

its set speed. The accuracy of the control system and its response to the operators

command is determined by the type of control system employed.



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The levels of control are:

Simple open-loop control No feedback from the process

Closed-loop control Feedback of a process variable

Cascade closed-loop control Feedback from more than one variable

When load torque and load position have to be continuously and accurately controlled the

closed loop control method is the most effective means of control feedback and will be the

method used in this project. The best compromise for accuracy and efficiency is the use

two feedback loops, these being the Speed and Torque control loops.

The first feedback loop controller is know as the speed control loop and uses the speed

error from a speed sensor to calculate the desired current. The primary objective being to

either increase speed or decrease speed. The speed loop therefore controls the output

frequency proportional to speed. The system was based on the knowledge that the speed

was recorded from a speed transducer positioned at the shaft of the motor. Therefore this

reduces the need for complicated calculations used to estimate speed. This is not always a

desirable alternative as in many cases the installation of a speed transducer is difficult or

economically unjustified. The speed error signal becomes the set point for the torque

regulator and is processed by a PID algorithm. This signal is compared to the in-phase

current feedback from the motor circuit and the error signal determines whether the motor

needs to accelerate or decelerate.



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The speed reference signal is derived from the inputs of the controller for the specific

Vector Control application. For example a conveyor is needed to be driven at 0.5m/s, that

speed can then be translated into a shaft speed in RPM depending on the appropriate gear

ratios.

The second feedback loop controller is known as the torque control loop. The loop

compares the set torque with the output of the torque estimation algorithm, with the new

error current calculates the desired output voltages. The measured process variable in this

case is the measured motor current which is proportional to the motor torque. Therefore

this control loop is often called the current loop.

In the design of the torque control loop it is assumed that the rate of change of current is

higher than the rate of change of speed or essentially saying that the motor is operating at a

constant speed. The current loop must allow for a time delay between output frequency

and the current. It then determines the desired inverter output frequency and voltage which

is used by the PWM to determine switching logic. The torque estimation algorithm can be

seen in the simulation chapter.

The main functions of the Field Orientated Control sequence are:

To continuously calculate the value of the torque producing current. This is

achieved by implementing the following actions:

Continuously mod

22893773 36 2 Field Orientated Control of a Multi Level PWM Inverter Fed Induction Motor

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Transcript of 22893773 36 2 Field Orientated Control of a Multi Level PWM Inverter Fed Induction Motor